AD AD561T Low cost 10-bit monolithic d/a converter Datasheet

a
FEATURES
Complete Current Output Converter
High Stability Buried Zener Reference
Laser Trimmed to High Accuracy (1/4 LSB Max Error,
AD561K, T)
Trimmed Output Application Resistors for 0 V to +10 V,
65 V Ranges
Fast Settling – 250 ns to 1/2 LSB
Guaranteed Monotonicity Over Full Operating
Temperature Range
TTL/DTL and CMOS Compatible (Positive True Logic)
Single Chip Monolithic Construction
Available in Chip Form
MlL-STD-883-Compliant Versions Available
Low Cost 10-Bit
Monolithic D/A Converter
AD561
FUNCTIONAL BLOCK DIAGRAM
TO-116
PRODUCT DESCRIPTION
The AD561 is an integrated circuit 10-bit digital-to-analog
converter combined with a high stability voltage reference
fabricated on a single monolithic chip. Using ten precision highspeed current-steering switches, a control amplifier, voltage
reference, and laser-trimmed thin-film SiCr resistor network,
the device produces a fast, accurate analog output current.
Laser trimmed output application resistors are also included to
facilitate accurate, stable current-to-voltage conversion; they are
trimmed to 0.1% accuracy, thus eliminating external trimmers
in many situations.
Several important technologies combine to make the AD561 the
most accurate and most stable 10-bit DAC available. The low
temperature coefficient, high stability thin-film network is
trimmed at the wafer level by a fine resolution laser system to
0.01% typical linearity. This results in an accuracy specification
of ± 1/4 LSB max for the K and T versions, and 1/2 LSB max
for the J and S versions.
The AD561 also incorporates a low noise, high stability
subsurface zener diode to produce a reference voltage with
excellent long term stability and temperature cycle characteristics, which challenge the best discrete Zener references. A
temperature compensation circuit is laser-trimmed to allow
custom correction of the temperature coefficient of each device.
This results in a typical full-scale temperature coefficient of
15 ppm/°C; the TC is tested and guaranteed to 30 ppm/°C max
for the K and T versions, 60 ppm/°C max for the S, and
80 ppm/°C for the J.
The AD561 is available in four performance grades. The
AD561J and K are specified for use over the 0°C to +70°C
temperature range and are available in either a 16-pin
hermetically-sealed ceramic DIP or a 16-pin molded plastic
DIP. The AD561S and T grades are specified for the –55°C to
+125°C range and are available in the ceramic package.
PRODUCT HIGHLIGHTS
1. Advanced monolithic processing and laser trimming at the
wafer level have made the AD561 the most accurate 10-bit
converter available, while keeping costs consistent with large
volume integrated circuit production. The AD561K and T
have 1/4 LSB max relative accuracy and 1/2 LSB max
differential nonlinearity. The low TC R-2R ladder guarantees that all AD561 units will be monotonic over the entire
operating temperature range.
2. Digital system interfacing is simplified by the use of a
positive true straight binary code. The digital input voltage
threshold is a function of the positive supply level; connecting VCC to the digital logic supply automatically sets the
threshold to the proper level for the logic family being used.
Logic sink current requirement is only 25 µA.
3. The high speed current steering switches are designed to settle
in less than 250 ns for the worst case digital code transition.
This allows construction of successive-approximation A/D
converters in the 3 µs to 5 µs range.
4. The AD561 has an output voltage compliance range from
–2 V to +10 V, allowing direct current-to-voltage conversion
with just an output resistor, omitting the op amp. The 40 MΩ
open collector output impedance results in negligible errors
due to output leakage currents.
5. The AD561 is available in versions compliant with MILSTD-883. Refer to the Analog Devices Military Products
Databook or current AD561/883B data sheet for detailed
specifications.
REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700
World Wide Web Site: http://www.analog.com
Fax: 617/326-8703
© Analog Devices, Inc., 1997
AD561–SPECIFICATIONS (T = +258C, V
A
Model
Min
CC
AD561J
Typ
RESOLUTION
10 Bits
ACCURACY (Error Relative
to Full Scale)
± 1/4
(0.025)
DIFFERENTIAL NONLINEARITY
± 1/2
DATA INPUTS
TTL, VCC = +5 V
Bit ON Logic “1”
Bit OFF Logic “0”
CMOS, 10 V ≤ VCC ≤ 16.5 V
Bit ON Logic “ 1 “
Bit OFF Logic “0”
Logic Current (Each Bit) (T MIN to TMAX)
Bit ON Logic “1”
Bit OFF Logic “0”
OUTPUT
Current
Unipolar
Bipolar
Resistance (Exclusive of
Application Resistors)
Unipolar Zero (All Bits OFF)
Capacitance
Compliance Voltage
= –15 V, unless otherwise noted.)
Max
Min
AD561K
Typ
Max
Units
± 1/8
(0.012)
± 1/4
(0.025)
LSB
% of FS
± 1/4
± 1/2
LSB
*
V
V
*
V
V
*
*
*
*
nA
µA
*
*
*
*
*
*
mA
mA
*
*
*
*
*
*
Ω
% of FS
pF
V
10 Bits
± 1/2
(0.05)
+2.0
*
+0.8
70% VCC
*
30% VCC
+5
–5
+100
–25
1.5
± 0.75
2.0
± 1.0
2.4
± 1.2
–2
40 M
0.01
25
–3
0.05
+10
*
SETTLING TIME TO 1/2 LSB
All Bits ON-to-OFF or OFF-to-ON
250
POWER REQUIREMENTS
VCC, +4.5 V dc to +16.5 V dc
VEE, –10.8 V dc to –16.5 V dc
8
12
10
16
*
*
*
*
mA
mA
POWER SUPPLY GAIN SENSITIVITY
VCC, +4.5 V dc to +16.5 V dc
VEE, –10.8 V dc to –16.5 V dc
2
4
10
25
*
*
*
*
ppm of FS/%
ppm of FS/%
TEMPERATURE RANGE
Operating
Storage (“D” Package)
(“N” Package)
0 to +70
–65 to +150
–25 to +85
*
*
*
*
*
*
°C
°C
°C
TEMPERATURE COEFFICIENTS
With Internal Reference
Unipolar Zero
Bipolar Zero
Full Scale
Differential Nonlinearity
1
2
15
2.5
1
2
15
2.5
5
10
30
ppm of FS/°C
ppm of FS/°C
ppm of FS/°C
ppm of FS/°C
MONOTONICITY
*
10
20
80
Guaranteed Over Full Operating
Temperature Range
ns
Guaranteed Over Full Operating
Temperature Range
PROGRAMMABLE OUTPUT
RANGES
0 to +10
–5 to +5
*
*
V
V
CALIBRATION ACCURACY
Full-Scale Error with Fixed 25 Ω `
Resistor
Bipolar Zero Error with Fixed 10 Ω
Resistor
± 0.1
*
% of FS
± 0.1
*
% of FS
± 0.5
± 0.5
*
*
% of FS
% of FS
CALIBRATION ADJUSTMENT
RANGE
Full Scale (With 50 Ω Trimmer)
Bipolar Zero (With 50 Ω Trimmer)
NOTES
*Specifications same as AD561J specifications.
Specifications subject to change without notice.
–2–
REV. A
AD561
Model
Min
AD561S
Typ
RESOLUTION
10 Bits
ACCURACY (Error Relative
to Full Scale)
± 1/4
(0.025)
DIFFERENTIAL NONLINEARITY
± 1/2
DATA INPUTS
TTL, VCC = +5 V
Bit ON Logic “1”
Bit OFF Logic “0”
CMOS, 10 V ≤ VCC ≤ 16.5 V
Bit ON Logic “ 1 “
Bit OFF Logic “0”
Logic Current (Each Bit) (T MIN to TMAX)
Bit ON Logic “1”
Bit OFF Logic “0”
OUTPUT
Current
Unipolar
Bipolar
Resistance (Exclusive of
Application Resistors)
Unipolar Zero (All Bits OFF)
Capacitance
Compliance Voltage
Max
Min
AD561T
Typ
Max
Units
± 1/8
(0.012)
± 1/4
(0.025)
LSB
% of FS
± 1/4
± 1/2
LSB
**
V
V
**
V
V
**
**
**
**
nA
µA
**
**
**
**
**
**
mA
mA
**
**
**
**
**
**
Ω
% of FS
pF
V
10 Bits
± 1/2
(0.05)
+2.0
**
+0.8
70% VCC
**
30% VCC
+20
–25
+100
–100
1.5
± 0.75
2.0
± 1.0
2.4
± 1.2
–2
40 M
0.01
25
–3
0.05
+10
**
SETTLING TIME TO 1/2 LSB
All Bits ON-to-OFF or OFF-to-ON
250
POWER REQUIREMENTS
VCC, +4.5 V dc to +16.5 V dc
VEE, –10.8 V dc to –16.5 V dc
6
11
10
16
**
**
**
**
mA
mA
POWER SUPPLY GAIN SENSITIVITY
VCC, +4.5 V dc to +16.5 V dc
VEE, –10.8 V dc to –16.5 V dc
2
4
10
25
**
**
**
**
ppm of FS/%
ppm of FS/%
TEMPERATURE RANGE
Operating
Storage
–55 to +125
–65 to +150
**
**
**
**
°C
°C
TEMPERATURE COEFFICIENTS
With Internal Reference
Unipolar Zero
Bipolar Zero
Full Scale
Differential Nonlinearity
1
2
15
2.5
1
2
15
2.5
5
10
30
ppm of FS/°C
ppm of FS/°C
ppm of FS/°C
ppm of FS/°C
MONOTONICITY
**
10
20
60
Guaranteed Over Full Operating
Temperature Range
ns
Guaranteed Over Full Operating
Temperature Range
PROGRAMMABLE OUTPUT
RANGES
0 to +10
–5 to +5
**
**
V
V
CALIBRATION ACCURACY
Full-Scale Error with Fixed 25 Ω
Resistor
Bipolar Zero Error with Fixed 10 Ω
Resistor
± 0.1
**
% of FS
± 0.1
**
% of FS
CALIBRATION ADJUSTMENT
RANGE
Full Scale (With 50 Ω Trimmer)
Bipolar Zero (With 50 Ω Trimmer)
± 0.5
± 0.5
**
**
% of FS
% of FS
NOTES
**Specifications same as AD561S specifications.
Specifications subject to change without notice.
REV. A
–3–
AD561
THE AD561 OFFERS TRUE 10-BIT RESOLUTION OVER
FULL TEMPERATURE RANGE
Accuracy: Analog Devices defines accuracy as the maximum
deviation of the actual, adjusted DAC output (see page 5) from
the ideal analog output (a straight line drawn from 0 to FS – l
LSB) for any bit combination. The AD561 is laser trimmed to
1/4 LSB (0.025% of FS) maximum error at +25°C for the K
and T versions – 1/2 LSB for the J and S.
Monotonicity: A DAC is said to be monotonic if the output
either increases or remains constant for increasing digital inputs
such that the output will always be a single-valued function of the
input. All versions of the AD561 are monotonic over their full
operating temperature range.
Differential Nonlinearity: Monotonic behavior requires that
the differential nonlinearity error be less than
1 LSB both at +25°C and over the temperature range of
interest. Differential nonlinearity is the measure of the variation
in analog value, normalized to full scale, associated with a
1 LSB change in digital input code. For example, for a 10 volt
full scale output, a change of 1 LSB in digital input code should
result in a 9.8 mV change in the analog output (1 LSB = 10 V
× 1/1024 = 9.8 mV). If in actual use, however, a 1 LSB change
in the input code results in a change of only 2.45 mV (1/4 LSB)
in analog output, the differential nonlinearity error would be
7.35 mV, or 3/4 LSB The AD561K and T have a max differential linearity error of 1/2 LSB.
Figure 1. Chip Bonding Diagram
CONNECTING THE AD561 FOR BUFFERED VOLTAGE
OUTPUT
The standard current-to-voltage conversion connections using
an operational amplifier are shown here with the preferred
trimming techniques. If a low offset operational amplifier
(AD510, AD741L, AD301AL) is used, excellent performance
can be obtained in many situations without trimming. (A 5 mV
op amp offset is equivalent to 1/2 LSB on a 10 volt scale.) If a
25 Ω fixed resistor is substituted for the 50 Ω trimmer, unipolar
zero will typically be within ± 1/10 LSB (plus op amp offset),
and full scale accuracy will be within ± 1 LSB. Substituting a
25 Ω resistor for the 50 Ω bipolar offset trimmer will give a
bipolar zero error typically within ± 1 LSB.
The differential nonlinearity temperature coefficient must also
be considered if the device is to remain monotonic over its full
operating temperature range. A differential nonlinearity temperature coefficient of 2.5 ppm/°C could, under worst case conditions for a temperature change of +25°C to +125°C, add 0.025%
(100 3 2.5 ppm/°C of error). The resulting error could then be
as much as 0.025% + 0.025% = 0.05% of FS (1/2 LSB represents
0.05% of FS). To be sure of accurate performance all versions of
the AD561 are therefore 100% tested to be monotonic over the
full operating temperature range.
The AD509 is recommended for buffered voltage-output
applications that require a settling time to ± 1/2 LSB of one
microsecond. The feedback capacitor is shown with the
optimum value for each application; this capacitor is required to
compensate for the 25 picofarad DAC output capacitance.
ORDERING GUIDE
PIN CONFIGURATION
TOP VIEW
MODEL1
TEMP RANGE
ACCURACY
@ +258C
GAIN T C
(of FS/8C)
PACKAGE
OPTION2
AD561JD
AD561JN
AD561KD
AD561KN
AD561SD
AD561TD
AD561/883B
0°C to +70°C
0°C to +70°C
0°C to +70°C
0°C to +70°C
–55°C to +125°C
–55°C to +125°C
–55°C to +125°C
± 1/2 LSB max
± 1/2 LSB max
± 1/4 LSB max
± 1/4 LSB max
± 1/2 LSB max
± 1/4 LSB max
*
80 ppm max
80 ppm max
30 ppm max
30 ppm max
60 ppm max
30 ppm max
*
D-16
N-16
D-16
N-16
D-16
D-16
*
NOTES
1
For details on grade and package offerings screened in accordance with MIL-STD-883, refer to the
Analog Devices Military Products Databook or current AD561/883B data sheet.
2
D = Ceramic DIP; N = Plastic DIP.
*Refer to AD561/883B military data sheet.
–4–
REV. A
AD561
UNIPOLAR CONFIGURATION
This configuration, shown in Figure 2, will provide a unipolar
0 V to +10 V output range.
STEP I . . . ZERO ADJUST
Turn all bits OFF and adjust op amp trimmer, R1, until the
output reads 0.000 volts (1 LSB = 9.76 mV).
STEP 11. . . GAIN ADJUST
Turn all bits ON and adjust 50 Ω gain trimmer, R2, until the
output is 9.990 volts. (Full scale is adjusted to 1 LSB less than
nominal full scale of 10.000 volts.) If a 10.23 V full scale is desired
(exactly 10 mV/bit), insert a 120 Ω resistor in series with R2.
Figure 2. 0 V to +10 V Unipolar Voltage Output
BIPOLAR CONFIGURATION
This configuration, shown in Figure 3, will provide a bipolar
output voltage from –5.000 to +4.990 volts, with positive full
scale occurring with all bits ON (all 1s).
STEP 1. . . ZERO ADJUST
Turn ON MSB only, turn OFF all other bits. Adjust 50 Ω
trimmer R3, to give 0.000 output volts. For maximum resolution
a 120 Ω resistor may be placed in parallel with R3.
STEP 11. . . GAIN ADJUST
Turn OFF all bits, adjust 50 Ω gain trimmer to give a reading of
–5.000 volts.
Figure 3. ± 5 V Buffered Bipolar Voltage Output
Please note that it is not necessary to trim the op amp to obtain
full accuracy at room temperature. In most bipolar situations,
the op amp trimmer is unnecessary unless the untrimmed offset
drift of the op amp is excessive.
610 VOLT BUFFERED BIPOLAR OUTPUT
The AD561 can also be connected for a ± 10 volt bipolar range
with an additional external resistor as shown in Figure 4. A
larger value trimmer is required to compensate for tolerance in
the thin film resistors, which are trimmed to match the full-scale
current. For best full scale temperature coefficient performance,
the external resistors should have a TC of –50 ppm/°C.
CIRCUIT DESCRIPTION
A simplified schematic with the essential circuit features of the
AD561 is shown in Figure 5. The voltage reference, CR1, is a
buried Zener (or subsurface breakdown diode). This device
exhibits far better all-around performance than the NPN baseemitter reverse-breakdown diode (surface Zener), which is in
nearly universal use in integrated circuits as a voltage reference.
Greatly improved long-term stability and lower noise are the
major benefits the buried Zener derives from isolating the
breakdown point from surface stress and mobile oxide charge
effects. The nominal 7.5 volt device (including temperature
compensation circuitry) is driven by a current source to the
negative supply so the positive supply can be allowed to drop as
low as 4.5 volts. The temperature coefficient of each diode is
individually determined; this data is then used to laser trim a
compensating circuit to balance the overall TC to zero. The
typical resulting TC is 0 to ± 15 ppm/°C. The negative reference
level is inverted and scaled by A1 to give a +2.5 volt reference,
which can be driven by the low positive supply. The AD561,
packaged in the 16-pin DIP, has the +2.5 volt reference (REF
OUT) connected directly to the input of the control amplifier
(REF IN). The buffered reference is not directly available
externally except through the 2.5 kΩ bipolar offset resistor.
REV. A
Figure 4. ± 10 V Buffered Voltage Output
The 2.5 kΩ scaling resistor and control amplifier A2 then force a
1 mA reference current to flow through reference transistor Q1,
which has a relative emitter area of 8A. This is accomplished by
forcing the bottom of the ladder to the proper voltage. Since Q1
and Q2 have equal emitter areas and equal 5 kΩ emitter resistors,
Q2 also carries 1 mA. The ladder voltage drop constrains Q7
(with area 4A) to carry only 0.5 mA; Q8 carries 0.25 mA, etc.
The first four significant bit cells are exactly scaled in emitter
area to match Q1 for optimum VBE and VBE drift match, as well
as for beta match. These effects are insignificant for the lower
order bits, which account for a total of only 1/16 of full scale.
However, the 18 mV VBE difference between two matched
transistors carrying emitter currents in a ratio of 2:1 must be
corrected. This is achieved by forcing 120 µA through the
150 Ω interbase resistors. These resistors, and the R-2R ladder
resistors, are actively laser-trimmed at the wafer level to bring
total device accuracy to better than 1/4 LSB. Sufficient ratio
accuracy in the last two bits is obtained by simple emitter area
–5–
AD561
Figure 5. Circuit Diagram Showing Reference, Control Amplifier, Switching Cell, R-2R Ladder, and Bit Arrangement
of AD561
ratio such that it is unnecessary to use additional area for ladder
resistors. The current in Q16 is added to the ladder to balance it
properly, but is not switched to the output; thus, full scale is
1023/1024 3 2 mA.
will assume a “1” state (similar to TTL), but they are high
impedance and subject to noise pickup. Unused digital inputs
should be directly connected to ground or VCC, as desired.
The switching cell of Q3, Q4, Q5 and Q6 serves to steer the cell
current either to ground (BIT 1 low) or to the DAC output
(BIT 1 high). The entire switching cell carries the same current
whether the bit is on or off, minimizing thermal transients and
ground current errors. The logic threshold, which is generated
from the positive supply (see Digital Logic Interface), is applied
to one side of each cell.
SETTLING TIME
The high speed NPN current steering switching cell and
internally compensated reference amplifier of the AD561 are
specifically designed for fast settling operation. The typical
settling time to ± 0.05% (1/2 LSB) for the worst case transition
(major carry, 0111111111 to 1000000000) is less than 250 ns;
the lower order bits all settle in less than 200 ns. (Worst case
settling occurs when all bits are switched, especially the MSB.)
Full realization of this high speed performance requires strict
attention to detail by the user in all areas of application and
testing.
The settling time for the AD561 is specified in terms of the
current output, an inherently high speed DAC operating mode.
However, most DAC applications require a current-to-voltage
conversion at some point in the signal path, although an
unbuffered voltage level (not using an op amp) is suitable for
use in a successive-approximation A/D converter (see page 8),
or in many display applications. This form of conversion can
give very fast operation if proper design and layout is done. The
fastest voltage conversion is achieved by connecting a low value
resistor directly to the output, as shown in Figure 9. In this case,
the settling time is primarily determined by the cell switching
time and by the RC time constant of the AD561 output capacitance of 25 picofarads (plus stray capacitance) combined with the
output resistor value. Settling to 0.05% of full scale (for a fullscale transition) requires 7.6 time constants. This effect is
important for R > 1 kΩ.
Figure 6. Digital Threshold vs. Positive Supply
DIGITAL LOGIC INTERFACE
If an op amp must be used to provide a low impedance output
signal, some loss in settling time will be seen due to op amp
dynamics. The normal current-to-voltage converter op amp
circuits are shown in the applications circuits on page 5, using
the fast settling AD509. The circuits shown settle to ± 1/2 LSB
in 600 ns unipolar and 1.1 µs bipolar. The DAC output
capacitance, which acts as a stray capacitance at the op amp
inverting input, must be compensated by a feedback capacitor,
as shown. The value should be carefully chosen for each
application and each op amp type.
All standard positive supply logic families interface easily with
the AD561. The digital code is positive true binary (all bits
high, Logic “1,” gives positive full scale output). The logic input
load factor (100 nA max at Logic “1,” –25 µA max at Logic “0,”
3 pF capacitance), is less than one equivalent digital load for all
logic families, including unbuffered CMOS. The digital
threshold is set internally as a function of the positive supply, as
shown in Figure 6. For most applications, connecting VCC to the
positive logic supply will set the threshold at the proper level for
maximum noise immunity. For nonstandard applications, refer
to Figure 6 for threshold levels. Uncommitted bit input lines
–6–
REV. A
AD561
Fastest operation will be obtained by minimizing lead lengths,
stray capacitance and impedance levels. Both supplies should be
bypassed near the devices; 0.1 µF will be sufficient since the
AD561 runs at constant supply current regardless of input code.
for bipolar offset. For example, setting RX = 2.5 kΩ gives a ± 1
volt range with a 1 kΩ equivalent output impedance. A 0 to +10
volt output can be obtained by connecting the 5 kΩ gain resistor
to 9.99 volts; again the digital code is complementary binary.
POWER SUPPLY SELECTION
The AD561 will operate over a wide range of power supply
voltages, with a total supply from 15.3 to 33 volts. Symmetrical
supplies are not required, and in many applications not recommended. Maximum allowable supplies are ± 16.5 V.
The positive supply level determines the digital threshold level,
as explained on page 6 and shown in Figure 6. It is therefore
recommended that VCC be connected directly to the digital
supply for best threshold match.
Positive output voltage compliance range is unaffected by the
positive supply level because of the open collector output stage
design; thus the full +10 volt compliance is available even with a
+5 volt VCC level. Power supply rejection is excellent, so that
digital supply noise will not be reflected to the output. but use
of a 0.1 µF bypass capacitor near the AD561 is recommended
for decoupling.
The nominal negative supply level is –15 volts, with an allowable range of –10.8 to –16.5 volts. The negative supply level
affects the negative compliance range, as shown in Figure 7.
OUTPUT VOLTAGE COMPLIANCE
The AD561 has a typical output compliance range from –3 to
+10 volts. The output current is unaffected by changes in the
output terminal voltage over that range. This results from the
use of open collector output switching stages in a cascade
configuration, and gives an output impedance of 40 MΩ.
Positive compliance range is limited only by collector breakdown (and is independent of positive supply level), but the
negative range is limited by the required bias levels and resistor
ladder voltage. Negative compliance varies with negative supply,
as shown in Figure 7. The compliance range is guaranteed to be
–2 to +10 volts with VEE = –15 volts.
Figure 8. Unbuffered Bipolar Voltage Output
HIGH SPEED 10-BIT A/D CONVERTERS
The fast settling characteristics of the AD561 make it ideal for
high speed successive approximation A/D converters. The
internal reference and trimmed application resistors allow a
10-bit converter system to be constructed with a minimum parts
count. Shown here is a configuration using standard components; this system completes a full 10-bit conversion in 5.5 µs
unipolar or 12 µs bipolar. This converter will be accurate to
± 1/2 LSB of 10 bits and have a typical gain TC of 10 ppm/°C.
In the unipolar mode, the system range is 0 to 9.99 volts,
with each bit having a value of 9.76 mV. For true conversion
accuracy, an A/D converter should be trimmed so that a given
bit code output results from input levels from 1/2 LSB below to
1/2 LSB above the exact voltage which that code represents.
Therefore, the converter zero point should be trimmed with an
input voltage of +4.9 mV; trim R1 until the LSB just begins to
appear in the output code (all other bits “0”). For full scale, use
an input voltage of +9.985 volts (10 volts – 1 LSB – 1/2 LSB);
then trim R2 again until the LSB just begins to appear (all other
bits “1”).
The bipolar signal range is –5.0 to +4.99 volts. Bipolar offset
trimming is done by applying a +4.9 mV input signal and
trimming R1 for the LSB transition (MSB “1,” all other bits
“0.”) Full scale is set by applying –4.995 volts and trimming R2
for the LSB transition (all other bits “0”). In many applications,
the pretrimmed application resistors are sufficiently accurate
that external trimmers will be unnecessary, especially in
situations requiring less than full 10-bit ± 1/2 LSB accuracy.
Figure 7. Typical Negative Compliance Range vs.
Negative Supply
DIRECT UNBUFFERED VOLTAGE OUTPUT
The wide compliance range allows direct current-to-voltage
conversion with just an output resistor. Figure 8 shows a
connection using the gain and bipolar output resistors to give a
± 1.66 volt bipolar swing. In this situation, the digital code is
complementary binary. Other combinations of internal and
external output resistors (RX) can be used to scale to alternate
voltage ranges, simply by appropriately scaling the 0 to –2 mA
unipolar output current and using the 2.5 volt reference voltage
REV. A
For fastest operation, the impedance at the comparator summing node must be minimized, as mentioned in the section on
settling time. However, lowering the impedance will reduce the
voltage signal to the comparator (at an equivalent impedance of
1 kΩ, 1 LSB = 2 mV) to the point that comparator performance
will be sacrificed. A 1 kΩ resistor is the optimum value for this
application for 10-bit accuracy. The chart shown in the figure
gives the speed of the ADC for ± 1/2 LSB accuracy (and no
missing codes) for 6-, 8- and 10-bit resolution.
–7–
AD561
C394e–1–5/97
circuit will not produce both 1 to 5 volt and
4-to-20 mA outputs simultaneously.)
Figure 10. Digital 4-to-20 mA or 1-to-5 Volt Line Driver
DIGITALLY PROGRAMMABLE SETPOINT
COMPARATOR
Figure 11 demonstrates a high accuracy systems-oriented
setpoint comparator. The 2.5 volt reference is buffered and
amplified by the AD741K to produce an exact 10.000 volt
reference which could be used as a primary system reference for
several such circuits. The +10 volt compliance of the AD561
then allows it to generate a zero to +10 volt output swing
through the 5 kΩ application resistor without an additional op
amp. The digital code for this system will be complementary
binary (all 1s give 0.00 volts out).
Figure 9. Fast Precision Analog to Digital Converter
A much faster converter can be constructed by using higher
performance external components. Each individual high-order
bit settles in less than 250 ns; the low-order bits in less than
200 ns. Because of this, a staged clock, which speeds up for
lower bits will improve the speed. Also, a faster comparator and
Schottky TTL or ECL logic would be necessary. 10-bit converters in the 3 µs to 5 µs range could be built around the AD561
with these techniques.
DIGITAL 4-TO-20 mA OR 1-TO-5 VOLT CONVERTER
A direct digital 4-to-20 mA or 1-to-5 volt line driver can be built
with the AD561 as shown in Figure 10. The 2.5 volt reference is
divided to provide 1 volt at the op amp noninverting input – thus a
zero input code results in a 1 volt output at the Darlington emitter
(VOUT). The 2 k feedback resistance converts the nominal 2 mA
(± 20%) full-scale output from the AD561 to 4 volts, for a
total output of 5 volts FS. The voltage at the emitter forces a
proportional current through the 250 Ω (which appears at the
collector as IOUT) The AD561 current is added to the 4–20 mA
line; thus 5 volts full scale gives 22 mA in the current loop. For
exactly 20 mA, trim the 1 k pot for 4.5 V FS. (A single op amp
Figure 11. Digitally Programmable Set-Point Comparator
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
16-Pin Plastic Package
N-16
–8–
PRINTED IN U.S.A.
16-Pin Ceramic Package
D-16
REV. A