AD AD7864AS-1 4-channel, simultaneous sampling, high speed, 12-bit adc Datasheet

a
FEATURES
High Speed (1.65␣ ␮s) 12-Bit ADC
Four Simultaneously Sampled Inputs
Four Track/Hold Amplifiers
0.35␣ ␮s Track/Hold Acquisition Time
1.65 ␮s Conversion Time per Channel
HW/SW Select of Channel Sequence for Conversion
Single Supply Operation
Selection of Input Ranges:
ⴞ10 V, ⴞ5 V for AD7864-1
ⴞ2.5 V for AD7864-3
0 V to 2.5 V, 0 V to 5 V for AD7864-2
High Speed Parallel Interface Which Also Allows
Interfacing to 3 V Processors
Low Power, 90 mW Typ
Power Saving Mode, 20␣ ␮W Typ
Overvoltage Protection on Analog Inputs
APPLICATIONS
AC Motor Control
Uninterrupted Power Supplies
Data Acquisition Systems
Communications
GENERAL DESCRIPTION
The AD7864 is a high speed, low power, 4-channel simultaneous sampling 12-bit A/D converter that operates from a single
+5␣ V supply. The part contains a 1.65 µs successive approximation ADC, four track/hold amplifiers, 2.5 V reference, on-chip
clock oscillator, signal conditioning circuitry and a high speed
parallel interface. The input signals on four channels are
sampled simultaneously, thus preserving the relative phase information of the signals on the four analog inputs. The part accepts
analog input ranges of ± 10␣ V, ± 5 V (AD7864-1), 0 V to 2.5 V,
0 V to 5 V for AD7864-2 and ± 2.5␣ V (AD7864-3).
The part allows any subset of the four channels to be converted
in order to maximize the throughput rate on the selected sequence. The channels to be converted can be selected via either
hardware (channel select input pins) or software (programming
the channel select register).
A single conversion start signal (CONVST) simultaneously
places all the track/holds into hold and initiates conversion sequence for the selected channels. The EOC signal indicates the end
of each individual conversion in the selected conversion sequence.
The BUSY signal indicates the end of the conversion sequence.
Data is read from the part by means of a 12-bit parallel data
bus using the standard CS and RD signals. Maximum throughput for a single channel is 500 kSPS. For all four channels the
4-Channel, Simultaneous
Sampling, High Speed, 12-Bit ADC
AD7864
FUNCTIONAL BLOCK DIAGRAM
VREF
AVDD
TRACK/HOLD
34
STBY
VIN1A
VIN1B
SIGNAL
SCALING
VIN2A
VIN2B
SIGNAL
SCALING
SIGNAL
SCALING
VIN4A
VIN4B
SIGNAL
SCALING
12.5V
REFERENCE
DGND
AGND
AD7864
RD
12-BIT
ADC
OUTPUT
DATA
REGISTERS
SOFTWARE
LATCH
FRSTDATA
EOC
6kV
MUX
VIN3A
VIN3B
BUSY
DVDD VDRIVE
VREF GND
CONVERSION
CONTROL LOGIC
CONVST SL1 SL2 SL3 SL4 H/S
SEL
DB0–DB3
DB11
DB0
CS
WR
INT/EXT CLOCK
SELECT
INT
CLOCK
CLKIN INT/EXT AGND AGND
CLK
maximum throughput is 130 kSPS for the read during conversion sequence operation. The throughput rate for the read after
conversion sequence operation will depend on the read cycle
time of the processor. See Timing and Control section.
The AD7864 is available in a small (0.3 sq. inch area) 44-lead
MQFP.
PRODUCT HIGHLIGHTS
1. The AD7864 features four Track/Hold amplifiers and a fast
(1.65 µs) ADC allowing simultaneous sampling and then
conversion of any subset of the four channels.
2. The AD7864 operates from a single +5␣ V supply and consumes
only 90 mW typ making it ideal for low power and portable
applications. Also see Standby Mode Operation.
3. The part offers a high speed parallel interface for easy connection to microprocessors, microcontrollers and digital
signal processors.
4. The part is offered in three versions with different analog
input ranges. The AD7864-1 offers the standard industrial
input ranges of ± 10 V and ± 5 V; the AD7864-3 offers the
common signal processing input range of ± 2.5 V; the
AD7864-2 can be used in unipolar 0 V to 2.5 V, 0 V to 5 V
applications.
5. The part features very tight aperture delay matching between
the four input sample-and-hold amplifiers.
REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 1999
(VDD = +5 V ⴞ 5%, AGND = DGND = 0 V, VREF = Internal. Clock = Internal; all specifiMIN to T MAX unless otherwise noted.)
AD7864–SPECIFICATIONS cations T
Parameter
A Version1
B Version
Units
SAMPLE AND HOLD
–3 dB Full Power Bandwidth
Aperture Delay
Aperture Jitter
Aperture Delay Matching
3
20
50
4
3
20
50
4
MHz typ
ns max
ps typ
ns max
DYNAMIC PERFORMANCE2
Signal to (Noise + Distortion) Ratio3
@ +25°C
TMIN to TMAX
Total Harmonic Distortion3
Peak Harmonic or Spurious Noise3
Intermodulation Distortion3
2nd Order Terms
3rd Order Terms
Channel-to-Channel Isolation3
DC ACCURACY
Resolution
Relative Accuracy3
Differential Nonlinearity3
AD7864-1
Positive Gain Error3
Positive Gain Error Match3
Negative Gain Error3
Negative Gain Error Match3
Bipolar Zero Error
Bipolar Zero Error Match
AD7864-3
Positive Gain Error3
Positive Gain Error Match3
Negative Gain Error3
Negative Gain Error Match3
Bipolar Zero Error
Bipolar Zero Error Match
AD7864-2
Positive Gain Error3
Positive Gain Error Match3
Unipolar Offset Error
Unipolar Offset Error Match
ANALOG INPUTS
AD7864-1
Input Voltage Range
Input Resistance
AD7864-3
Input Voltage Range
Input Resistance
AD7864-2
Input Voltage Range
Input Current (0 V–2.5 V Option)
Input Resistance (0 V–5 V Option)
REFERENCE INPUT/OUTPUT
VREF IN Input Voltage Range
VREF IN Input Capacitance4
VREF OUT Output Voltage
VREF OUT Error @ +25°C
VREF OUT Error TMIN to TMAX
VREF OUT Temperature Coefficient
VREF OUT Output Impedance
Test Conditions/Comments
fIN = 100.0 kHz, fS = 500 kSPS
70
70
–80
–80
72
70
–80
–80
dB min
dB min
dB max
dB max
–80
–80
–80
–80
–80
–80
dB typ
dB typ
dB max
12
±1
± 0.9
12
± 1/2
± 0.9
Bits
LSB max
LSB max
±3
3
±3
3
±4
2
±3
±3
±3
±3
±3
±2
LSB max
LSB max
LSB max
LSB max
LSB max
LSB max
fa = 49 kHz, fb = 50 kHz
fIN = 50 kHz Sine Wave
Any Channel
±3
2
±3
2
±3
2
LSB max
LSB max
LSB max
LSB max
LSB max
LSB max
±3
3
±3
2
LSB max
LSB max
LSB max
LSB max
± 5, ± 10␣
9, 18
± 5,␣ ± 10␣
9, 18
Volts
kΩ min
± 2.5␣
4.5
± 2.5␣
4.5
Volts
kΩ min
+2.5, +5␣
± 100
9
+2.5, +5␣ ␣
± 100
9
Volts
nA max
kΩ min
2.375/2.625
10
2.5
± 10
± 20
25
6
2.375/2.625
10
2.5
± 10
± 20
25
6
VMIN/VMAX
pF max
V␣ nom
mV max
mV max
ppm/°C typ
kΩ typ
–2–
No Missing Codes
2.5 V ± 5%
See Reference Section
REV. A
AD7864
1
Parameter
A Version
B Version
Units
Test Conditions/Comments
LOGIC INPUTS
Input High Voltage, VINH
Input Low Voltage, VINL
Input Current, IIN
Input Capacitance, CIN4
2.4
0.8
± 10
10
2.4
0.8
± 10
10
V min
V max
µA max
pF max
VDD = 5 V ± 5%
VDD = 5 V ± 5%
4.0
0.4
4.0
0.4
V min
V max
ISOURCE = 400 µA
ISINK = 1.6 mA
± 10
10
± 10
10
µA max
pF max
LOGIC OUTPUTS
Output High Voltage, VOH
Output Low Voltage, VOL
DB11–DB0
High Impedance
Leakage Current
Capacitance4
Output Coding
AD7864-1, AD7864-3
AD7864-2
Twos Complement
Straight (Natural) Binary
CONVERSION RATE
Conversion Time
Track/Hold Acquisition Time2, 3
Throughput Time
1.65
0.35
130
1.65
0.35
130
µs max
µs max
kSPS max
+5
+5
V nom
± 5% for Specified Performance
(5 µA typ) Logic Inputs = 0 V or VDD
24
20
24
20
mA max
µA max
Typically 4␣ µA
120
100
120
100
mW max
µW max
Typically 90␣ mW
Typically 20␣ µW
POWER REQUIREMENTS
VDD
IDD
Normal Mode
Standby Mode
Power Dissipation
Normal Mode
Standby Mode
For One Channel
For All Four Channels
NOTES
1
Temperature ranges are as follows: A, B Versions: –40°C to +85°C. Note: The A Version is fully specified up to +105°C with degraded INL and DNL specifications
of ± 2 LSBs max.
2
Performance measured through full channel (SHA and ADC).
3
See Terminology.
4
Sample tested @ +25°C to ensure compliance.
Specifications subject to change without notice.
REV. A
–3–
AD7864
TIMING CHARACTERISTICS1, 2
(VD = +5 V ⴞ 5%, AGND = DGND = 0 V, V REF = Internal, Clock = Internal; all specifications
TMIN to TMAX unless otherwise noted.)
Parameter
A, B Versions
Units
Test Conditions/Comments
tCONV
1.65
13
2.6
0.34
No. of Channels
x (tCONV + t9) – t9
2
6
µs max
Clock Cycles
µs max
µs max
µs max
µs max
ms max
Conversion Time, Internal Clock
Conversion Time, External Clock
CLKIN = 5 MHz
Acquisition Time
Selected Number of Channels Multiplied by
(tCONV + EOC Pulsewidth)—EOC Pulsewidth
STBY Rising Edge to CONVST Rising Edge
STBY Rising Edge to CONVST Rising Edge
35
70
ns min
ns min
CONVST Pulsewidth
CONVST Rising Edge to BUSY Rising Edge
0
0
35
35
40
5
30
10
75
180
70
15
0
ns min
ns min
ns min
ns max
ns max
ns min
ns max
ns min
ns min
ns max
ns max
ns max
ns min
CS to RD Setup Time
CS to RD Hold Time
Read Pulsewidth
Data Access Time After Falling Edge of RD, VDRIVE = 5 V
Data Access Time After Falling Edge of RD, VDRIVE = 3 V
Bus Relinquish Time After Rising Edge of RD
20
0
0
5
5
ns min
ns min
ns min
ns min
ns min
WR Pulsewidth
CS to WR Setup Time
WR to CS Hold Time
Input Data Setup Time of Rising Edge of WR
Input Data Hold Time
tACQ
tBUSY
tWAKE-UP—External VREF
tWAKE-UP—Internal VREF3
t1
t2
Read Operation
t3
t4
t5
t6 4
t7 5
t8
t9
t10
t11
t12
Write Operation
t13
t14
t15
t16
t17
Time Between Consecutive Reads
EOC Pulsewidth
RD Rising Edge to FRSTDATA Edge (Rising or Falling)
EOC Falling Edge to FRSTDATA Falling Delay
EOC to RD Delay
NOTES
1
Sample tested at +25°C to ensure compliance. All input signals are measured with tr = tf = 1 ns (10% to 90% of +5 V) and timed from a voltage level of +1.6␣ V.
2
See Figures 7, 8 and 9.
3
Refer to the Standby Mode Operation section. The MAX specification of 6 ms is valid when using a 0.1 µF decoupling capacitor on the V REF pin.
4
Measured with the load circuit of Figure 2 and defined as the time required for an output to cross 0.8␣ V or 2.4 V.
5
These times are derived from the measured time taken by the data outputs to change 0.5␣ V when loaded with the circuit of Figure 2. The measured number is then
extrapolated back to remove the effects of charging or discharging the 50 pF capacitor. This means that the times quoted in the timing characteristics are the true bus
relinquish times of the part and as such are independent of external bus loading capacitances.
Specifications subject to change without notice.
1.6mA
TO
OUTPUT
11.6V
50pF
400mA
Figure 1. Load Circuit for Access Time and Bus Relinquish Time
–4–
REV. A
AD7864
ABSOLUTE MAXIMUM RATINGS*
Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C
Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . .+150°C
MQFP Package, Power Dissipation . . . . . . . . . . . . . . 450 mW
θJA Thermal Impedance . . . . . . . . . . . . . . . . . . . . . 95°C/W
Lead Temperature, Soldering
Vapor Phase (60 sec) . . . . . . . . . . . . . . . . . . . . . . +215°C
Infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . . . . .+220°C
(TA = +25°C unless otherwise noted)
VDD to AGND . . . . . . . . . . . . . . . . . . . . . . . . . –0.3␣ V to +7␣ V
VDD to DGND . . . . . . . . . . . . . . . . . . . . . . . . . –0.3␣ V to +7␣ V
Analog Input Voltage to AGND
AD7864-1 (± 10 V Input Range) . . . . . . . . . . . . . . . . ± 20 V
AD7864-1 (± 5 V Input Range) . . . . . . . . . . . –7 V to +20 V
AD7864-3 . . . . . . . . . . . . . . . . . . . . . . . . . . . –7 V to +20 V
AD7864-2 . . . . . . . . . . . . . . . . . . . . . . . . . . . –1 V to +20 V
Reference Input Voltage to AGND . . . –0.3 V to VDD + 0.3␣ V
Digital Input Voltage to DGND . . . . . –0.3 V to VDD + 0.3 V
Digital Output Voltage to DGND . . . . –0.3 V to VDD + 0.3 V
Operating Temperature Range
Commercial (A, B Version) . . . . . . . . . . . . –40°C to +85°C
*Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those listed in the operational
sections of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
ORDERING GUIDE
Model
Input
Ranges
Relative
Accuracy
Temperature
Range*
Package
Description
Package
Option
AD7864AS-1
AD7864BS-1
AD7864AS-3
AD7864AS-2
± 5 V, ± 10 V
± 5 V, ± 10 V
± 2.5 V
0 V to 2.5 V, 0 V to 5 V
± 1 LSB
± 0.5 LSB
± 1 LSB
± 1 LSB
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
Plastic Lead Quad Flatpack
Plastic Lead Quad Flatpack
Plastic Lead Quad Flatpack
Plastic Lead Quad Flatpack
S-44
S-44
S-44
S-44
*The A Version is fully specified up to +105°C with degraded INL and DNL specifications of ± 2 LSBs max.
DB6
VDRIVE
DVDD
DGND
DB4
DB5
DB2
DB3
DB0
DB1
EOC
PIN CONFIGURATION
44 43 42 41 40 39 38 37 36 35 34
33 DB7
BUSY 1
FRSTDATA 2
PIN 1
IDENTIFIER
32 DB8
CONVST 3
31 DB9
CS 4
30 DB10
RD 5
AD7864
WR 6
TOP VIEW
(Not to Scale)
SL1 7
29 DB11
28 CLKIN
27 INT/EXT CLK
SL2 8
26 AGND
SL3 9
25 AVDD
SL4 10
24 VREF
23 VREFGND
H/S SEL 11
STBY
VIN1A
VIN1B
VIN2A
VIN2B
AGND
VIN3B
VIN3A
VIN4A
VIN4B
AGND
12 13 14 15 16 17 18 19 20 21 22
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD7864 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
REV. A
–5–
WARNING!
ESD SENSITIVE DEVICE
AD7864
PIN FUNCTION DESCRIPTION
Pin
Mnemonic
Description
1
BUSY
2
FRSTDATA
3
CONVST
4
5
CS
RD
6
WR
7–10
SL1–SL4
11
H/S SEL
12
AGND
13–16
17
VIN4X, VIN3X
AGND
18–21
22
VIN2X, VIN1X
STBY
23
VREF GND
24
VREF
25
26
27
AVDD
AGND
INT/EXT CLK
28
CLKIN
29–34
DB11–DB6
35
DVDD
36
VDRIVE
37
DGND
38, 39
40–43
DB5, DB4
DB3–DB0
44
EOC
Busy Output. The busy output is triggered high by the rising edge of CONVST and remains high until
conversion is completed on all selected channels.
First Data Output. FRSTDATA is a logic output which, when high, indicates that the Output Data
Register Pointer is addressing Register 1—See Accessing the Output Data Registers.
Convert Start Input. Logic Input. A low-to-high transition on this input puts all track/holds into their
hold mode and starts conversion on the selected channels. In addition, the state of the Channel Sequence
Selection is also latched on the rising edge of CONVST.
Chip Select Input. Active Low Logic Input. The device is selected when this input is active.
Read Input. Active Low Logic Input that is used in conjunction with CS low to enable the data outputs. Ensure the WR pin is at logic high while performing a read operation.
Write Input. A rising edge on the WR input, with CS low and RD high, latches the logic state on DB0
to DB3 into the channel select register.
Hardware Channel Select. Conversion sequence selection can also be made via the SL1–SL4 pins if
H/S SEL is logic zero. The selection is latched on the rising edge of CONVST. See Selecting a Conversion Sequence.
Hardware/Software Select Input. When this pin is at a Logic 0, the AD7864 conversion sequence selection is controlled via the SL1–SL4 input pins. When this pin is at Logic 1, the sequence is controlled
via the channel select register. See Selecting a Conversion Sequence.
Analog Ground. General Analog Ground. This AGND␣ pin should be connected to the system’s
AGND plane.
Analog Inputs. See Analog Input section.
Analog Ground. Analog Ground reference for the attenuator circuitry. This AGND␣ pin should be
connected to the system’s AGND plane.
Analog Inputs. See Analog Input section.
Standby Mode Input. TTL-compatible input that is used to put the device into the power save or
standby mode. The STBY input is high for normal operation and low for standby operation.
Reference Ground. Ground reference for the part’s on-chip reference buffer. The V REF␣ GND pin
should be connected to the system’s AGND plane.
Reference Input/Output. This pin provides access to the internal reference (2.5 V ± 5%) and also
allows the internal reference to be overdriven by an external reference source (2.5 V). A 0.1 µF decoupling capacitor should be connected between this pin and AGND.
Analog Positive Supply Voltage, +5.0 V ± 5%.
Analog Ground. Analog Ground reference for the DAC circuitry.
Internal/External Clock Select Input. When this pin is at a Logic 0, the AD7864 uses its internally
generated master clock. When this pin is at Logic 1, the master clock is generated externally to the
device.
Conversion Clock Input. This is an externally applied clock that allows the user to control the conversion rate of the AD7864. Each conversion needs fourteen clock cycles in order for the conversion to be
completed and an EOC pulse to be generated. The clock should have a duty cycle that is no worse than
60/40. See Using an External Clock.
Data Bit 11 is the MSB, followed by Data Bit 10 to Data Bit 6. Three-state TTL outputs. Output
coding is 2s complement for the AD7864-1 and AD7864-3. Output coding is straight (natural) binary
for the AD7864-2.
Positive supply voltage for digital section, +5.0 V ± 5%. A 0.1 µF decoupling capacitor should be connected between this pin and AGND. Both DVDD and AVDD should be externally tied together.
This pin provides the positive supply voltage for the output drivers (DB0 to DB11), BUSY, EOC and
FRSTDATA. It is normally tied to DV DD. VDRIVE should be decoupled with a 0.1 µF capacitor. It allows
improved performance when reading during the conversion sequence. To facilitate interfacing to 3 V processors and DSPs the output data drivers may also be powered by a 3 V ± 10% supply .
Digital Ground. Ground reference for digital circuitry. This DGND pin should be connected to the
system’s AGND plane at the AGND pin.
Data Bit 5 to Data Bit 4. Three-state TTL Outputs.
Data Bit 3 to Data Bit 0. Bidirectional Data Pins. When a read operation takes place, these pins are threestate TTL outputs. The channel select register is programmed with the data on the DB0–DB3 pins with
standard CS and WR signals. DB0 represents Channel 1 and DB3 which represents Channel 4.
End-of-Conversion. Active low logic output indicating conversion status. The end of each conversion in
a conversion sequence is indicated by a low-going pulse on this line.
–6–
REV. A
AD7864
TERMINOLOGY
Channel-to-Channel Isolation
Signal to (Noise + Distortion) Ratio
Channel-to-Channel isolation is a measure of the level of
crosstalk between channels. It is measured by applying a fullscale 50␣ kHz sine wave signal to all nonselected input channels
and determining how much that signal is attenuated in the
selected channel. The figure given is the worst case across all four
channels.
This is the measured ratio of signal to (noise + distortion) at the
output of the A/D converter. The signal is the rms amplitude of
the fundamental. Noise is the rms sum of all nonfundamental
signals up to half the sampling frequency (fS/2), excluding dc.
The ratio is dependent upon the number of quantization levels
in the digitization process; the more levels, the smaller the quantization noise. The theoretical signal to (noise + distortion) ratio
for an ideal N-bit converter with a sine wave input is given by:
Relative Accuracy
Relative accuracy or endpoint nonlinearity is the maximum
deviation from a straight line passing through the endpoints of
the ADC transfer function.
Signal to (Noise + Distortion) = (6.02 N + 1.76) dB
Thus for a 12-bit converter, this is 74␣ dB.
Differential Nonlinearity
Total Harmonic Distortion
This is the difference between the measured and the ideal
1␣ LSB change between any two adjacent codes in the ADC.
Total harmonic distortion (THD) is the ratio of the rms sum of
harmonics to the fundamental. For the AD7864 it is defined as:
( )
THD dB = 20 log
Positive Full-Scale Error
This is the deviation of the last code transition (01 . . . 110 to
01 . . . 111) from the ideal 4 × VREF – 3/2 LSB (AD7864-1
± 10 V), 2 × VREF – 3/2 LSB (AD7864-1 ± 5 V range) or VREF
– 3/2 LSB (AD7864-3, ± 2.5 V range), after the Bipolar Offset
Error has been adjusted out.
V22 +V32 +V42 +V52 +V62
V1
where V1 is the rms amplitude of the fundamental and V2, V3,
V4 and V5 are the rms amplitudes of the second through the fifth
harmonics.
Positive Full-Scale Error (AD7864-2, 0 V to 2.5 V and 0 V to
5 V)
This is the deviation of the last code transition (11 . . . 110 to
11 . . . 111) from the ideal 2 × VREF – 3/2 LSB (AD7864-2 0 V
to 5 V range), or VREF – 3/2 LSB (AD7864-2 0 V to 2.5 V
range), after the Unipolar Offset Error has been adjusted out.
Peak Harmonic or Spurious Noise
Peak harmonic or spurious noise is defined as the ratio of the
rms value of the next largest component in the ADC output
spectrum (up to fS/2 and excluding dc) to the rms value of the
fundamental. Normally, the value of this specification is determined by the largest harmonic in the spectrum, but for parts
where the harmonics are buried in the noise floor, it will be a
noise peak.
Bipolar Zero Error (AD7864-1, ⴞ10/ⴞ5 V, AD7864-3 , ⴞ2.5 V)
This is the deviation of the midscale transition (all 0s to all 1s)
from the ideal AGND – 1/2 LSB.
Unipolar Offset Error (AD7864-2, 0 V to 2.5 V and 0 V to 5 V)
Intermodulation Distortion
This is the deviation of the first code transition (00 . . . 000 to
00 . . . 001) from the ideal AGND + 1/2 LSB.
With inputs consisting of sine waves at two frequencies, fa and
fb, any active device with nonlinearities will create distortion
products at sum and difference frequencies of mfa ± nfb where
m, n = 0, 1, 2, 3, etc. Intermodulation terms are those for which
neither m nor n are equal to zero. For example, the second
order terms include (fa + fb) and (fa – fb), while the third order
terms include (2 fa + fb), (2 fa – fb), (fa + 2 fb) and (fa – 2 fb).
Negative Full-Scale Error (AD7864-1, ⴞ10/ⴞ5 V, AD7864-3,
ⴞ2.5 V)
This is the deviation of the first code transition (10 . . . 000 to
10 . . . 001) from the ideal –4 × VREF + 1/2 LSB (AD7864-1
± 10 V), –2 × VREF + 1/2 LSB (AD7864-1 ± 5 V range) or
–VREF + 1/2 LSB (AD7864-3, ± 2.5 V range), after Bipolar
Zero Error has been adjusted out.
The AD7864 is tested using the CCIF standard, where two
input frequencies near the top end of the input bandwidth are
used. In this case, the second and third order terms are of different significance. The second order terms are usually distanced
in frequency from the original sine waves, while the third order
terms are usually at a frequency close to the input frequencies.
As a result, the second and third order terms are specified separately. The calculation of the intermodulation distortion is as
per the THD specification where it is the ratio of the rms sum of
the individual distortion products to the rms amplitude of the
fundamental expressed in dBs.
REV. A
Track/Hold Acquisition Time
Track/Hold acquisition time is the time required for the output of the track/hold amplifier to reach its final value, within
± 1/2 LSB, after the end of conversion (the point at which the
track/hold returns to track mode). It also applies to situations
where there is a step input change on the input voltage applied
to the selected VINXA/VINXB input of the AD7864. It means that
the user must wait for the duration of the track/hold acquisition
time after the end of conversion or after a step input change to
VINXA/VINXB before starting another conversion, to ensure that
the part operates to specification.
–7–
AD7864
CONVERTER DETAILS
Track/Hold Section
The AD7864 is a high speed, low power, 4-channel simultaneous sampling 12-bit A/D converter that operates from a single
+5␣ V supply. The part contains a 1.65␣ µs successive approximation ADC, four track/hold amplifiers, an internal +2.5␣ V reference and a high speed parallel interface. There are four analog
inputs that can be simultaneously sampled thus preserving the
relative phase information of the signals on all four analog inputs. Thereafter, conversions will be completed on the selected
subset of the four channels. The part accepts an analog input
range of ± 10␣ V or ± 5␣ V (AD7864-1), ± 2.5␣ V (AD7864-3) and
0 V–2.5 V or 0 V–5 V (AD7864-2). Overvoltage protection on
the analog inputs of the part allows the input voltage to go to
± 20 V, (AD7864-1 ± 10 V range), –7 V or +20 V (AD7864-1
± 5 V range) –1 V to +20 V (AD7864-2) and –7 V to +20 V
(AD7864-3) without causing damage or effecting a conversion in
progress. The AD7864 has two operating modes Reading Between
Conversions and Reading after the Conversion Sequence. These
modes are discussed in more detail in the Timing and Control
section.
The track/hold amplifiers on the AD7864 allows the ADCs to
accurately convert an input sine wave of full-scale amplitude to
12-bit accuracy. The input bandwidth of the track/hold is greater
than the Nyquist rate of the ADC even when the ADC is operated at its maximum throughput rate of 500 kSPS (i.e., the
track/hold can handle input frequencies in excess of 250␣ kHz).
The track/hold amplifiers acquire input signals to 12-bit accuracy in less than 350␣ ns. The operation of the track/holds are
essentially transparent to the user. The four track/hold amplifiers sample their respective input channels simultaneously, on
the rising edge of CONVST. The aperture time for the track/
holds (i.e., the delay time between the external CONVST signal
and the track/hold actually going into hold) is typically 15␣ ns
and, more importantly, is well matched across the four track/
holds on one device and also well matched from device to device. This allows the relative phase information between different input channels to be accurately preserved. It also allows
multiple AD7864s to sample more than four channels simultaneously. At the end of a conversion sequence, the part returns
to its tracking mode. The acquisition time of the track/hold
amplifiers begin at this point.
A conversion is initiated on the AD7864 by pulsing the CONVST
input. On the rising edge of CONVST, all four on-chip track/
holds are placed into hold simultaneously and the conversion
sequence is started on all the selected channels. Channel
selection is made via the SL1–SL4 pins if H/S SEL is logic zero
or via the channel select register if H/S SEL is logic one—see
Selecting a Conversion Sequence. The channel select register is
programmed via the bidirectional data lines DB0–DB3 and a
standard write operation. The selected conversion sequence is
latched on the rising edge of CONVST so changing a selection
will only take effect once a new conversion sequence is initiated.
The BUSY output signal is triggered high on the rising edge of
CONVST and will remain high for the duration of the conversion sequence. The conversion clock for the part is generated
internally using a laser-trimmed clock oscillator circuit. There is
also the option of using an external clock, by tying the INT/
EXT CLK pin logic high, and applying an external clock to the
CLKIN pin. However, the optimum throughput is obtained by
using the internally generated clock—see Using an External
Clock. The EOC signal indicates the end of each conversion in
the conversion sequence. The BUSY signal indicates the end of
the full conversion sequence and at this time all four Track and
Holds return to tracking mode. The conversion results can
either be read at the end of the full conversion sequence
(indicated by BUSY going low) or as each result becomes
available (indicated by EOC going low). Data is read from the
part via a 12-bit parallel data bus with standard CS and RD
signals—see Timing and Control.
Reference Section
The AD7864 contains a single reference pin, labelled VREF,
which either provides access to the part’s own +2.5␣ V reference
or to which an external +2.5␣ V reference can be connected to
provide the reference source for the part. The part is specified
with a +2.5␣ V reference voltage. Errors in the reference source
will result in gain errors in the AD7864’s transfer function and
will add to the specified full-scale errors on the part. On the
AD7864-1 and AD7864-3, it will also result in an offset error
injected in the attenuator stage, see Figures 2 and 4.
The AD7864 contains an on-chip +2.5␣ V reference. To use this
reference as the reference source for the AD7864, simply connect a 0.1␣ µF disc ceramic capacitor from the VREF pin to AGND.
The voltage that appears at this pin is internally buffered before
being applied to the ADC. If this reference is used externally to
the AD7864, it should be buffered, as the part has a FET switch
in series with the reference output resulting in a source impedance for this output of 6␣ kΩ nominal. The tolerance on the
internal reference is ± 10␣ mV at 25°C with a typical temperature
coefficient of 25␣ ppm/°C and a maximum error over temperature of ± 20␣ mV.
If the application requires a reference with a tighter tolerance or
the AD7864 needs to be used with a system reference, the user
has the option of connecting an external reference to this VREF
pin. The external reference will effectively overdrive the internal
reference and thus provide the reference source for the ADC.
The reference input is buffered before being applied to the ADC
with the maximum input current of ± 100␣ µA. Suitable reference
sources for the AD7864 include the AD680, AD780, REF192
and REF43 precision +2.5␣ V references.
Conversion time for each channel of the AD7864 is 1.65 µs and
the track/hold acquisition time is 0.35 µs. To obtain optimum
performance from the part, the read operation should not occur
during a channel conversion or during the 100 ns prior to the
next CONVST rising edge. This allows the part to operate at
throughput rates up to 130 kHz for all four channels and
achieve data sheet specifications.
CIRCUIT DESCRIPTION
Analog Input Section
The AD7864 is offered as three part types: the AD7864-1,
where each input can be configured for ± 10 V or a ± 5 V input
voltage range; the AD7864-3 which handles input voltage range
± 2.5 V; the AD7864-2, where each input can be configured to
have a 0␣ V to +2.5␣ V or 0␣ V to +5␣ V input voltage range.
–8–
REV. A
AD7864
AD7864-1
AD7864-2
Figure 2 shows the analog input section of the AD7864-1. Each
input can be configured for ± 5 V or ±10 V operation on the
AD7864-1. For ± 5 V (AD7864-1) operation, the VINXA and
VINXB inputs are tied together and the input voltage is applied to
both. For ± 10 V (AD7864-1) operation, the VINXB input is tied
to AGND and the input voltage is applied to the VINXA input.
The VINXA and VINXB inputs are symmetrical and fully interchangeable. Thus for ease of PCB layout on the ± 10 V range,
the input voltage may be applied to the VINXB input while the
VINXA input is tied to AGND.
Figure 3 shows the analog input section of the AD7864-2. Each
input can be configured for 0 V to +5 V operation or 0 V to
+2.5 V operation. For 0 V to +5 V operation, the VINXB input is
tied to AGND and the input voltage is applied to the VINXA
input. For 0 V to 2.5 V operation, the VINXA and VINXB inputs
are tied together and the input voltage is applied to both. The
VINXA and VINXB inputs are symmetrical and fully interchangeable. Thus for ease of PCB layout on the 0 V to +5 V range, the
input voltage may be applied to the VINXB input while the VINXA
input is tied to AGND.
For the AD7864-2, R1 = 6 kΩ and R2 = 6 kΩ. Once again, the
designed code transitions occur on successive integer LSB
values. Output coding is straight (natural) binary with 1 LSB =
FSR/4096 = 2.5 V/4096 = 0.61 mV, and 5 V/4096 = 1.22 mV,
for the 0 V to 2.5 V and the 0 V to 5 V options respectively.
Table II shows the ideal input and output transfer function for
the AD7864-2.
AD7864-1
6kV
12.5V
REFERENCE
VREF
R1
TO ADC
REFERENCE
CIRCUITRY
AD7864-2
R2
VIN1A
6kV
R3
T/H
VIN1B
TO INTERNAL
COMPARATOR
12.5V
REFERENCE
VREF
R4
TO ADC
REFERENCE
CIRCUITRY
AGND
Figure 2. AD7864-1 Analog Input Structure
R1
VIN1A
For the AD7864-1, R1 = 6 kΩ, R2 = 24 kΩ, R3 = 24 kΩ and
R4 = 12 kΩ. The resistor input stage is followed by the high
input impedance stage of the track/hold amplifier.
R2
T/H
VIN1B
The designed code transitions take place midway between successive integer LSB values (i.e., 1/2 LSB, 3/2 LSBs, 5/2 LSBs
etc.) LSB size is given by the formula, 1 LSB = FSR/4096. For
the ± 5 V range, 1 LSB = 10 V/4096 = 2.44 mV. For the ± 10 V
range, 1 LSB = 20 V/4096 = 4.88 mV. Output coding is twos
complement binary with 1 LSB = FSR/4096. The ideal input/
output transfer function for the AD7864-1 is shown in Table I.
TO INTERNAL
COMPARATOR
Figure 3. AD7864-2 Analog Input Structure
Table II. Ideal Input/Output Code Table for the AD7864-2
Analog Input1
Digital Output Code Transition
2
Digital Output Code Transition
+FSR – 3/2 LSB
+FSR – 5/2 LSB
+FSR – 7/2 LSB
111 . . . 110 to 111 . . . 111
111 . . . 101 to 111 . . . 110
111 . . . 100 to 111 . . . 101
+FSR/2 – 3/2 LSB
+FSR/2 – 5/2 LSB
+FSR/2 – 7/2 LSB
011 . . . 110 to 011 . . . 111
011 . . . 101 to 011 . . . 110
011 . . . 100 to 011 . . . 101
AGND + 5/2 LSB
AGND + 3/2 LSB
AGND + 1/2 LSB
000 . . . 010 to 000 . . . 011
000 . . . 001 to 000 . . . 010
000 . . . 000 to 000 . . . 001
AGND + 3/2 LSB
AGND + 1/2 LSB
AGND – 1/2 LSB
AGND – 3/2 LSB
000 . . . 001 to 000 . . . 010
000 . . . 000 to 000 . . . 001
111 . . . 111 to 000 . . . 000
111 . . . 110 to 111 . . . 111
NOTES
1
FSR is full-scale range and is 0 V to 2.5 V and 0 V to 5 V for AD7864-2 with
VREF = +2.5 V.
2
1 LSB = FSR/4096 and is 0.61 mV (0 V to 2.5 V) and 1.22 mV (0 V to 5 V) for
AD7864-2 with V REF = +2.5 V.
–FSR/2 + 5/2 LSB
–FSR/2 + 3/2 LSB
–FSR/2 + 1/2 LSB
100 . . . 010 to 100 . . . 011
100 . . . 001 to 100 . . . 010
100 . . . 000 to 100 . . . 001
Table I. Ideal Input/Output Code Table for the AD7864-1
Analog Inputl
2
NOTES
1
FSR is full-scale range and is 20 V for the ± 10 V range and 10 V for the ±5 V
range, with V REF = +2.5 V.
2
1 LSB = FSR/4096 = 4.883 mV (±10 V—AD7864-1) and 2.441 mV (± 5 V—
AD7864-1) with VREF = +2.5 V.
REV. A
–9–
AD7864
AD7864-3
SELECTING A CONVERSION SEQUENCE
Figure 4 shows the analog input section of the AD7864-3. The
analog input range is ± 2.5 V on the VIN1A input. The VIN1B
input can be left unconnected but if it is connected to a potential then that potential must be AGND.
Any subset of the four channels VIN1 to VIN4 can be selected
for conversion. The selected channels are converted in an
ascending order. For example if the channel selection includes
VIN4, VIN1 and VIN3 then the conversion sequence will be
VIN1, VIN3 and then VIN4. The conversion sequence selection
my be made by using either the hardware channel select input
pins (SL1 through SL4) or programming the channel select
register. A logic high on a hardware channel select pin (or logic
one in the channel select register) when CONVST goes logic
high, marks the associated analog input channel for inclusion in
the conversion sequence.
AD7864-3
6kV
12.5V
REFERENCE
VREF
Figure 5 shows the arrangement used. The H/S SEL controls a
multiplexer which selects the source of the conversion sequence
information, i.e., from the Hardware channel select pins (SL1 to
SL4) or from the channel selection register. When a conversion
is started the output from the multiplexer is latched until the
end of the conversion sequence. The data bus Bits DB0 to DB3
(DB0 representing Channel 1 through DB3 representing Channel 4) are bidirectional and become inputs to the channel select
register when RD is logic high and CS and WR are logic low.
The logic state on DB0 to DB3 is latched into the channel select
register when WR goes logic high.
TO ADC
REFERENCE
CIRCUITRY
R1
R2
VIN1A
T/H
TO INTERNAL
COMPARATOR
VIN1B
Figure 4. AD7864-3 Analog Input Structure
For the AD7864-3, R1 = 6 kΩ and R2 = 6 kΩ. As a result, the
VIN1A input should be driven from a low impedance source. The
resistor input stage is followed by the high input impedance
stage of the track/hold amplifier.
The designed code transitions take place midway between
successive integer LSB values (i.e., 1/2 LSB, 3/2 LSBs, 5/2 LSBs,
etc.) LSB size is given by the formula, 1 LSB = FSR/4096.
Output coding is 2s complement binary with 1 LSB = FSR/
4096 = 5 V/4096 = 1.22 mV. The ideal input/output transfer
function for the AD7864-3 is shown in Table III.
H/S
SL1
SL2
SL3
SL4
DATA BUS
D3 D2 D1 D0
CHANNEL SELECT
REGISTER
Digital Output Code Transition
+FSR/2 – 3/2 LSB2
+FSR/2 – 5/2 LSB
+FSR/2 – 7/2 LSB
011 . . . 110 to 011 . . . 111
011 . . . 101 to 011 . . . 110
011 . . . 100 to 011 . . . 101
AGND + 3/2 LSB
AGND + 1/2 LSB
AGND – 1/2 LSB
AGND – 3/2 LSB
000 . . . 001 to 000 . . . 010
000 . . . 000 to 000 . . . 001
111 . . . 111 to 000 . . . 000
111 . . . 110 to 111 . . . 111
–FSR/2 + 5/2 LSB
–FSR/2 + 3/2 LSB
–FSR/2 + 1/2 LSB
100 . . . 010 to 100 . . . 011
100 . . . 001 to 100 . . . 010
100 . . . 000 to 100 . . . 001
M
U
L
T
I
P
L
E
X
E
R
WR
Table III. Ideal Input/Output Code Table for the AD7864-3
Analog Inputl
SELECT INDIVIDUAL
TRACK-AND-HOLDS
FOR CONVERSION
HARDWARE CHANNEL
SELECT PINS
LATCH
SEQUENCER
TRANSPARENT WHILE WAITING FOR
CONVST. LATCHED ON THE RISING
EDGE OF CONVST AND DURING A
CONVERSION SEQUENCE.
CS
WR
Figure 5. Channel Select Inputs and Registers
RD
t 13
WR
t 15
t 14
CS
t 16
NOTES
1
FSR is full-scale range is 5 V, with V REF = +2.5 V.
2
1 LSB = FSR/4096 = 1.22 mV (±2.5 V - AD7864-3) with V REF = +2.5 V.
DATA
t 17
DATA IN
Figure 6. Channel Selection via Software Control
–10–
REV. A
AD7864
t ACQ
t1
CONVST
BUSY
t BUSY
QUIET
TIME
t2
t CONV
t CONV
t CONV
t CONV
t8
EOC
t 11
t 10
FRSTDATA
t 11
RD
t4
t3
t5
CS
t7
t6
VIN1
DATA
VIN2
VIN3
VIN4
100ns
H/S SEL
100ns
SL1–SL4
Figure 7. Timing Diagram for Reading During Conversion
TIMING AND CONTROL
Reading Between Each Conversion in the Conversion Sequence
Figure 7 shows the timing and control sequence required to
obtain the optimum throughput rate from the AD7864. To
obtain the optimum throughput from the AD7864 the user must
read the result of each conversion as it becomes available. The
timing diagram in Figure 7 shows a read operation each time the
EOC signal goes logic low. The timing in Figure 7 shows a
conversion on all four analog channels (SL1 to SL4 = 1, see
Channel Selection), hence there are four EOC pulses and four read
operations to access the result of each of the four conversions.
A conversion is initiated on the rising edge of CONVST. This
places all four track/holds into hold simultaneously. New data
from this conversion sequence is available for the first channel
selected (VIN1) 1.65␣ µs later. The conversion on each subsequent channel is completed at 1.65␣ µs intervals. The end of each
conversion is indicated by the falling edge of the EOC signal.
The BUSY output signal indicates the end of conversion for all
selected channels (four in this case).
Data is read from the part via a 12-bit parallel data bus with
standard CS and RD signals. The CS and RD inputs are internally gated to enable the conversion result onto the data bus.
The data lines DB0 to DB11 leave their high impedance state
when both CS and RD are logic low. Therefore, CS may be
permanently tied logic low and the RD signal used to access the
conversion result. Since each conversion result is latched into its
output data register prior to EOC going logic low a further
option would be to tie the EOC and RD pins together and use
the rising edge of EOC to latch the conversion result. Although
the AD7864 has some special features that permit reading during a conversion (e.g., a separate supply for the output data
REV. A
drivers, VDRIVE), for optimum performance it is recommended
that the read operation be completed when EOC is logic low,
i.e., before the start of the next conversion. Although Figure 8
shows the read operation taking place during the EOC pulse, a
read operation can take place at any time. Figure 8 shows a
timing specification called “Quiet Time.” This is the amount of
time that should be left after a read operation and before the
next conversion is initiated. The quiet time depends heavily on
data bus capacitance but a figure of 50 ns to 100 ns is typical.
The signal labeled FRSTDATA (First Data Word) indicates to
the user that the pointer associated with the output data registers is pointing to the first conversion result by going logic high.
The pointer is reset to point to the first data location (i.e., first
conversion result,) at the end of the first conversion (FRSTDATA
logic high). The pointer is incremented to point to the next
register (next conversion result) when that conversion result is
available. Hence, FRSTDATA in Figure 7 is seen to go low just
prior to the second EOC pulse. Repeated read operations during a conversion will continue to access the data at the current
pointer location until the pointer is incremented at the end of that
conversion. Note FRSTDATA has an indeterminate logic state
after initial power up. This means that for the first conversion
sequence after power up, the FRSTDATA logic output may
already be logic high before the end of the first conversion. This
condition is indicated by the dashed line in Figure 7. Also the
FRSTDATA logic output may already be high as a result of the
previous read sequence as is the case after the fourth read in
Figure 7. The fourth read (rising edge of RD) resets the
pointer to the first data location. Therefore, FRSTDATA is
already high when the next conversion sequence is initiated.
See Accessing the Output Data Registers.
–11–
AD7864
t1
CONVST
BUSY
t BUSY
QUIET
TIME
t2
EOC
t8
RD
t3
t4
CS
t7
t6
VIN1
DATA
VIN2
t 10
VIN3
VIN4
VIN1
t 10
FRSTDATA
Figure 8. Timing Diagram, Reading After the Conversion Sequence
Reading After the Conversion Sequence
Using an External Clock
Figure 8 shows the same conversion sequence as Figure 7. In
this case, however, the results of the four conversions (on VIN1
to VIN4) are read after all conversions have finished, i.e., when
BUSY goes logic low. The FRSTDATA signal goes logic high
at the end of the first conversion just prior to EOC going logic
low. As mentioned previously FRSTDATA has an indeterminate state after initial power up, therefore FRSTDATA may
already be logic high. Unlike the case when reading between
each conversion the output data register pointer is incremented
on the rising edge of RD because the next conversion result is
available. This means FRSTDATA will go logic low after the
first rising edge on RD.
The logic input INT/EXT CLK allows the user to operate the
AD7864 using the internal clock oscillator or an external clock.
The optimum performance is achieved by using the internal
clock on the AD7864. The highest external clock frequency
allowed is 5 MHz. This means a conversion time of 2.6 µs
compared to 1.65 µs using the internal clock. In some instances,
however, it may be useful to use an external clock when high
throughput rates are not required. For example, two or more
AD7864s may be synchronized by using the same external clock
for all devices. In this way there is no latency between output
logic signals like EOC due to differences in the frequency of the
internal clock oscillators. Figure 9 shows how the various logic
outputs are synchronized to the CLK signal. Each conversion
requires 14 clocks. The output data register pointer is reset to
point to the first register location on the falling edge of the 12
clock cycle of the first conversion in the conversion sequence—
See Accessing the Output Data Registers. At this point the logic
output FRSTDATA goes logic high. The result of the first
conversion is transferred to the output data registers on the
falling edge of the 13 clock cycle. The FRSTDATA signal is
reset on the falling edge of the 13 clock cycle of the next
conversion, i.e., when the result of the second conversion is
transferred to its output data register. As mentioned previously,
the pointer is incremented by the rising edge of the RD signal if
the result of the next conversion is available. The EOC signal
goes logic low on the falling edge of the 13 clock cycle and is
reset high again on the falling edge of the 14 clock cycle.
Successive read operations will access the remaining conversion
results in an ascending channel order. Each read operation
increments the output data register pointer. The read operation
that accesses the last conversion result causes the output data
register pointer to be reset so that the next read operation will
access the first conversion result again. This is shown in Figure
8 with the fifth read after BUSY goes low accessing the result of
the conversion on VIN1. Thus the output data registers act as a
circular buffer in which the conversion results may be continually accessed. The FRSTDATA signal will go high when the
first conversion result is available.
Data is enabled onto the data bus DB0 to DB11 using CS and
RD. Both CS and RD have the same functionality as described
in the previous section. There are no restrictions or performance implications associated to the position of the read operations after BUSY goes low. The only restriction is that there is
minimum time between read operations. Notice also that a
“Quiet Time” is needed before the start of the next conversion.
–12–
REV. A
AD7864
1 2 3 4 5 6 7 8 9 10 11 12 13 14 1 2 3 4 5 6 7 8 9 10 11 12 13 14 1 2
13 14
CLK
CONVST
FRSTDATA
EOC
RD
FIRST CONVERSION
COMPLETE
LAST CONVERSION
COMPLETE
BUSY
Figure 9. Using an External Clock
The AD7864 has a Standby Mode whereby the device can be
placed in a low current consumption mode (5 µA typ). The
AD7864 is placed in standby by bringing the logic input STBY
low. The AD7864 can be powered up again for normal operation by bringing STBY logic high. The output data buffers are
still operational while the AD7864 is in standby. This means
the user can still continue to access the conversion results while
the AD7864 is in standby. This feature can be used to reduce
the average power consumption in a system using low throughput rates. To reduce the average power consumption the AD7864
can be placed in standby at the end of each conversion sequence,
i.e., when BUSY goes low and taken out of standby again prior
the start of the next conversion sequence. The time it takes the
AD7864 to come out of standby is called the “wake up” time.
This wake-up time will limit the maximum throughput rate at
which the AD7864 can be operated when powering down between conversion sequences. The AD7864 will wake-up in
approximately 2 µs when using an external reference. The
“wake up” time is also 2 µs when the standby time is less than 1
millisecond while using the internal reference. Figure 11 shows
the wake-up time of the AD7864 for standby times greater than
1 millisecond. Note when the AD7864 is left in standby for
periods of time greater than 1 millisecond the part will require
more than 2 µs to wake up. For example after initial power up,
using the internal reference the AD7864 takes 6 ms to power
up. The maximum throughput rate that can be achieved when
powering down between conversions is 1/(tBUSY + 2 µs) = 100
kSPS, approximately. When operating the AD7864 in a
standby mode between conversions the power savings can be
significant. For example with a throughput rate of 10 kSPS the
AD7864 will be powered down (IDD = 5 µA) for 90 µs out of
every 100 µs. See Figure 10. Therefore the average power consumption drops to (125/10) mW or 12.5 mW approximately.
1
0.9
POWER-UP TIME – ms
Standby Mode Operation
0.8
0.7
0.6
0.5
+1058C
0.4
+258C
0.3
0.2
0.1
–408C
0
0.0001
0.001
0.01
0.1
STANDBY TIME – sec
Accessing the Output Data Registers
There are four Output Data Registers, one for each of the four
possible conversion results from a conversion sequence. The
result of the first conversion in a conversion sequence is placed
in Register 1 and the second result is placed in Register Number
2 and so on. For example if the conversion sequence VIN1,
VIN3 and VIN4 is selected (see Conversion Sequence Selection) then the results of the conversion on VIN1, VIN3 and
VIN4 are placed in Registers 1 to 3 respectively. The Output
Data register pointer is reset to point to Register 1 at the end of
the first conversion in the sequence, just prior to EOC going
t BUSY
t BUSY
7ms
BUSY
tWAKEUP
STBY
IDD = 20mA
2ms
Figure 10. Power-Down Between Conversion Sequences
REV. A
10
Figure 11. Power-Up Time vs. Standby Time Using the
On-Chip Reference (Decoupled with 0.1 µ F Capacitor)
100ms
CONVST
1
–13–
AD7864
low. At this point the logic output FRSTDATA will go logic
high to indicate that the output data register pointer is addressing Register Number 1. When CS and RD are both logic low
the contents of the addressed register are enabled onto the data
bus (DB0–DB11).
Positive Full-Scale Adjust
Apply a voltage of +9.9927 V (FS – 3/2 LSBs) at V1. Adjust R2
until the ADC output code flickers between 0111 1111 1110
and 0111 1111 1111.
Negative Full-Scale Adjust
When reading the output data registers after a conversion
sequence, i.e., when BUSY goes low, the register pointer is
incremented on the rising edge of the RD signal as shown in
Figure 12. However, when reading the conversion results during
the conversion sequence the pointer will not be incremented
until a valid conversion result is in the register to be addressed.
In this case the pointer is incremented when the conversion has
ended and the result has been transferred to the output data
register. This happens just prior to EOC going low, therefore
EOC may be used to enable the register contents onto the data
bus as described in Reading During the Conversion Sequence.
The pointer is reset to point to Register 1 on the rising edge of
the RD signal when the last conversion result in the sequence is
being read. In the example shown this means the pointer is set
to Register 1 when the contents of Register 3 are read.
Apply a voltage of –9.9976 V (–FS + 1/2 LSB) at V1 and adjust
R2 until the ADC output code flickers between 1000 0000
0000 and 1000 0000 0001.
An alternative scheme for adjusting full-scale error in systems
which use an external reference is to adjust the voltage at the
VREF pin until the full-scale error for any of the channels is
adjusted out. The good full-scale matching of the channels will
ensure small full-scale errors on the other channels.
INPUT
RANGE = 610V
V1
R1
10kV
R2
500V
FRSTDATA
VDRIVE
2-BIT
COUNTER
POINTER*
RESET
DECODE
OUTPUT DATA REGISTERS
OE #1
(VIN1)
OE #2
(VIN3)
OE #3
(VIN4)
OE #4
NOT VALID
O/P
DRIVERS
VINXA
R4
R3
10kV
10kV
R5
10kV
AD7864*
AGND
DB0 TO DB11
OE
*ADDITIONAL PINS OMITTED FOR CLARITY
AD7864
Figure 13. Full-Scale Adjust Circuit
RD
CS
*THE POINTER WILL NOT BE INCREMENTED BY A RISING EDGE ON RD UNTIL
THE CONVERSION RESULT IS IN THE OUTPUT DATA REGISTER. THE POINTER
IS RESET WHEN THE LAST CONVERSION RESULT IS READ
Figure 12. Output Data Registers
OFFSET AND FULL-SCALE ADJUSTMENT
In most Digital Signal Processing (DSP) applications, offset and
full-scale errors have little or no effect on system performance.
Offset error can always be eliminated in the analog domain by
ac coupling. Full-scale error effect is linear and does not cause
problems as long as the input signal is within the full dynamic
range of the ADC. Invariably, some applications will require
that the input signal span the full analog input dynamic range.
In such applications, offset and full-scale error will have to be
adjusted to zero.
Figure 13 shows a circuit which can be used to adjust the offset
and full-scale errors on the AD7864 (VA1 on the AD7864-1
version is shown for example purposes only). Where adjustment
is required, offset error must be adjusted before full-scale error.
This is achieved by trimming the offset of the op amp driving
the analog input of the AD7864 while the input voltage is a
1/2 LSB below analog ground. The trim procedure is as follows:
apply a voltage of –2.44 mV (–1/2 LSB) at V1 in Figure 13 and
adjust the op amp offset voltage until the ADC output code
flickers between 1111 1111 1111 and 0000 0000 0000.
Gain error can be adjusted at either the first code transition
(ADC negative full scale) or the last code transition (ADC positive full scale). The trim procedures for both cases are as follows.
DYNAMIC SPECIFICATIONS
The AD7864 is specified and 100% tested for dynamic performance specifications as well as traditional dc specifications such
as integral and differential nonlinearity. These ac specifications
are required for the signal processing applications such as
phased array sonar, adaptive filters and spectrum analysis.
These applications require information on the ADC’s effect on
the spectral content of the input signal. Hence, the parameters
for which the AD7864 is specified include SNR, harmonic distortion, intermodulation distortion and peak harmonics. These terms
are discussed in more detail in the following sections.
Signal-to-Noise Ratio (SNR)
SNR is the measured signal to noise ratio at the output of the
ADC. The signal is the rms magnitude of the fundamental.
Noise is the rms sum of all the nonfundamental signals up to
half the sampling frequency (fS/2) excluding dc. SNR is dependent upon the number of quantization levels used in the digitization process; the more levels, the smaller the quantization
noise. The theoretical signal to noise ratio for a sine wave input
is given by
␣ ␣ ␣ ␣ ␣ ␣ SNR = (6.02N + 1.76) dB
(1)
where N is the number of bits.
Thus for an ideal 12-bit converter, SNR = 74 dB.
Figure 14 shows a histogram plot for 8192 conversions of a dc
input using the AD7864 with 5 V supply. The analog input was
set at the center of a code. It can be seen that all the codes
appear in the one output bin indicating very good noise performance from the ADC.
–14–
REV. A
AD7864
9000
12
8000
11
–408C
EFFECTIVE NUMBERS OF BITS
7000
6000
5000
4000
3000
2000
10
+258C
9
8
+1058C
7
6
5
1000
0
4
1054 1055 1056 1057 1058 1059 1060 1061 1062 1063 1064
ADC CODE
Figure 14. Histogram of 8192 Conversions of a DC Input
0
500
1000
1500
2000
FREQUENCY – kHz
2500
3000
Figure 16.␣ Effective Numbers of Bits vs. Frequency
Intermodulation Distortion
With inputs consisting of sine waves at two frequencies, fa and
fb, any active device with nonlinearities will create distortion
products at sum and difference frequencies of mfa ± nfb where
m, n = 0, 1, 2, 3 . . ., etc. Intermodulation terms are those for
which neither m or n are equal to zero. For example, the second
order terms include (fa + fb) and (fa – fb) while the third order
terms include (2fa + fb), (2fa – fb), (fa + 2fb) and (fa – 2fb).
The output spectrum from the ADC is evaluated by applying a
sine wave signal of very low distortion to the analog input. A
Fast Fourier Transform (FFT) plot is generated from which the
SNR data can be obtained. Figure 15 shows a typical 4096 point
FFT plot of the AD7864 with an input signal of 99.9 kHz and a
sampling frequency of 500 kHz. The SNR obtained from this
graph is 72.6 dB. It should be noted that the harmonics are
taken into account when calculating the SNR.
Using the CCIF standard where two input frequencies near the
top end of the input bandwidth are used, the second and third
order terms are of different significance. The second order terms
are usually distanced in frequency from the original sine waves
while the third order terms are usually at a frequency close to
the input frequencies. As a result, the second and third order
terms are specified separately. The calculation of the intermodulation distortion is as per the THD specification where it is the
ratio of the rms sum of the individual distortion products to the
rms amplitude of the fundamental expressed in dBs. In this
case, the input consists of two, equal amplitude, low distortion
sine waves. Figure 17 shows a typical IMD plot for the AD7864.
0
AD7864-1 @ +258C
5V SUPPLY
SAMPLING AT 499712Hz
INPUT FREQUENCY OF 99857Hz
8192 SAMPLES TAKEN
–10
–20
–30
dB
–40
–50
–60
–70
–80
–90
0
–100
AD7864-1 @ +258C
5V SUPPLY
SAMPLING AT 131072Hz
INPUT FREQUENCIES OF
48928Hz AND 50016Hz
4096 SAMPLES TAKEN
–10
–110
0
50000
100000
150000
FREQUENCY – Hz
200000
250000
–20
–30
Figure 15. FFT Plot
The formula given in Equation 1 relates the SNR to the number
of bits. Rewriting the formula, as in Equation 2, it is possible to
get a measure of performance expressed in effective number of
bits (N).
SNR –1.76
N=
(2)
6.02
The effective number of bits for a device can be calculated directly from its measured SNR. Figure 16 shows a typical plot
of effective number of bits versus frequency for an AD7864-2.
REV. A
–15–
dB
–40
Effective Number of Bits
–50
–60
–70
–80
–90
–100
0
10000
20000
30000
40000
FREQUENCY – Hz
Figure 17. IMD Plot
50000
60000
AD7864
AC Linearity Plots
Measuring Aperture Jitter
The plots shown in Figure 18 below show typical DNL and INL
plots for the AD7864.
A convenient way to measure aperture jitter is to use the relationship it is known to have with SNR (Signal to Noise plus
distortion) given below:


1
SNRJITTER = 20 × log10 
 , where
2
(
×
π
×
f
×
σ
)


IN
SNR JITTER = Signal to Noise due to the rms time jitter
σ = rms time jitter
fIN = sinusoidal input frequency (1 MHz in this case)
3.00E–01
2.00E–01
DNL – LSB
1.00E–01
(3)
From Equation 3 above it can be seen that the signal-to-noise
ratio due to jitter degrades significantly with frequency. Therefore, at low input frequencies the measured SNR performance
of the AD7864 is indicative of noise performance due to quantization noise and system noise only (72 dBs used as a typical
figure here).
0.00E+00
–1.00E–01
–2.00E–01
–3.00E–01
0
500
1000
1500
2000
2500
ADC CODE
3000
3500
4000
2.50E–01
Therefore, by measuring the overall SNR performance (including noise due to jitter, system and quantization) of the AD7864 a
good estimation of the jitter performance of the AD7864 can be
calculated.
2.00E–01
1.50E–01
12
11
5.00E–02
0.00E+00
10
–5.00E–02
ENOBs
INL – LSB
1.00E–01
–1.00E–01
–1.50E–01
9
8
–2.00E–01
7
–2.50E–01
0
500
1000
1500
2000
2500
ADC CODE
3000
3500
4000
6
Figure 18. Typical DNL and INL Plots
5
900000
950000
1000000
FREQUENCY – Hz
1050000
1100000
Figure 19. ENOBs of the AD7864 at 1 MHz
From Figure 19 above the ENOBs of the AD7864 at 1 MHz is
approximately 11 bits. This is equivalent to 68 dBs SNR.
SNRTOTAL = SNR JITTER + SNRQUANT = 68 dBs
68 dBs = SNR JITTER + 72 dBs (at 100 kHz)
SNR JITTER = 70.2 dBs
From (3) above
70.2 dBs = 20 × log10 [1/(2 × π × 1 MHz × σ)]
σ = 49 ps where σ is the rms jitter of the AD7864
–16–
REV. A
AD7864
MICROPROCESSOR INTERFACING
TMS320C5x
The high speed parallel interface of the AD7864 allows easy
interfacing to most DSP processors and microprocessors.
The ADC7864 interface of the AD7864 consists of the data
lines (DB0–DB11), CS, RD, WR, EOC and BUSY.
ADDRESS
DECODE
Figure 20 shows an interface between the AD7864 and the
ADSP-2100. The CONVST signal can be generated by the
ADSP-2100 or from some other external source. Figure 20
shows the CS being generated by a combination of the DMS
signal and the address bus of the ADSP-2100. In this way
the AD7864 is mapped into the data memory space of the
ADSP-2100.
DB0–DB11
INTn
CONVST
PA0
The MC68000 AS and R/W outputs are used to generate a
separate RD input signal for the AD7864. RD is used to drive
the 68000 DTACK input to allow the processor to execute a
normal read operation to the AD7864. The conversion results
are read using the following 68000 instruction:
MOVE.W ADC,D0
CS
WR
BUSY
An interface between the AD7864 and the MC68000 is shown
in Figure 22. The conversion can be initiated from the MC68000
or from an external source. The AD7864 BUSY line can be
used to interrupt the processor or, alternatively, software delays
can ensure that conversion has been completed before a read to
the AD7864 is attempted. Because of the nature of its interrupts, the 68000 requires additional logic (not shown in Figure
22) to allow it to be interrupted correctly. For further information on 68000 interrupts, consult the 68000 Users Manual.
A0–A13
RD
D0–D15
AD7864–MC68000 Interface
DMS
WR
WE
Figure 21. AD7864–TMS320C5x Interface
ADSP-2100/1/2
RD
RD
WR
AD7864
where MR0 is the ADSP-2100 MR0 register and ADC is the
AD7864 address.
VIN1
VIN2
VIN3
VIN4
RD
DB0–DB11
The AD7864 BUSY line provides an interrupt to the ADSP2100 when the conversion sequence is complete on all the
selected channels. The conversion results can then be read from
the AD7864 using successive read operations. Alternately, one
can use the EOC pulse to interrupt the ADSP-2100 when the
conversion on each channel is complete when reading between
each conversion in the conversion sequence (Figure 7). The
AD7864 is read using the following instruction
MR0 = DM(ADC)
ADDRESS
DECODE
CS
VIN1
VIN2
VIN3
VIN4
AD7864–ADSP-2100/ADSP-2101/ADSP-2102 Interface
A0–A13
DS
where D0 is the 68000 D0 register and ADC is the AD7864
address.
D0–D24
AD7864
BUSY
CONVST
MC68000
IRQn
ADDRESS
DECODE
DT1/F0
Figure 20. AD7864–ADSP-2100 Interface
AD7864–TMS320C5x Interface
Figure 21 shows an interface between the AD7864 and the
TMS320C5x. As with the previous interfaces, conversion can be
initiated from the TMS320C5x or from an external source and
the processor is interrupted when the conversion sequence is
completed. The CS signal to the AD7864 is derived from the
DS signal and a decode of the address bus. This maps the AD7864
into external data memory. The RD signal from the TMS320 is
used to enable the ADC data onto the data bus. The AD7864
has a fast parallel bus so there are no wait state requirements.
The following instruction is used to read the conversion results
from the AD7864:
IN D,ADC
where D is Data Memory address and ADC is the AD7864
address.
REV. A
–17–
CS
VIN1
VIN2
VIN3
VIN4
A0–A15
DTACK
AS
RD
R/W
AD7864
DB0–DB11
CONVST
D0–D15
CLOCK
Figure 22. AD7864–MC68000 Interface
AD7864
Vector Motor Control
MULTIPLE AD7864s IN A SYSTEM
The current drawn by a motor can be split into two components: one produces torque and the other produces magnetic
flux. For optimal performance of the motor, these two components should be controlled independently. In conventional
methods of controlling a three-phase motor, the current (or
voltage) supplied to the motor and the frequency of the drive are
the basic control variables. However, both the torque and flux
are functions of current (or voltage) and frequency. This coupling effect can reduce the performance of the motor because,
for example, if the torque is increased by increasing the frequency, the flux tends to decrease.
Figure 24 shows a system where a number of AD7864s can be
configured to handle multiple input channels. This type of configuration is common in applications such as sonar, radar, etc.
The AD7864 is specified with maximum limits on aperture
delay match. This means that the user knows the difference in
the sampling instant between all channels. This allows the user
to maintain relative phase information between the different
channels. The AD7864 has a maximum aperture delay matching of 4 ns.
Vector control of an ac motor involves controlling phase in
addition to drive and current frequency. Controlling the phase
of the motor requires feedback information on the position of
the rotor relative to the rotating magnetic field in the motor.
Using this information, a vector controller mathematically transforms the three phase drive currents into separate torque and
flux components. The AD7864, with its four-channel simultaneous sampling capability, is ideally suited for use in vector
motor control applications.
All AD7864s use the same external SAR clock (5 MHz). Therefore, the conversion time for all devices will be the same and so
all devices may be read simultaneously. The example shown in
Figure 24 the data outputs of two AD7864s are enabled onto a
32-bit wide data bus when EOC goes low.
VIN1
EOC
VIN2
VIN3 AD7864
VIN4
REF193
A block diagram of a vector motor control application using the
AD7864 is shown in Figure 23. The position of the field is
derived by determining the current in each phase of the motor.
Only two phase currents need to be measured because the third
can be calculated if two phases are known. VIN1 and VIN2 of the
AD7864 are used to digitize this information.
VREF
CS
CLKIN
RD
VIN1
VIN2
VIN3 AD7864
VIN4
Simultaneous sampling is critical to maintain the relative phase
information between the two channels. A current sensing isolation amplifier, transformer or Hall effect sensor is used between
the motor and the AD7864. Rotor information is obtained by
measuring the voltage from two of the inputs to the motor. VIN3
and VIN4 of the AD7864 are used to obtain this information.
Once again the relative phase of the two channels is important.
A DSP microprocessor is used to perform the mathematical
transformations and control loop calculations on the information fed back by the AD7864.
VREF
CS
CLKIN
RD
ADSP-2106x
12
32
RD
12
ADDRESS
DECODE
5MHz
Figure 24. Multiple AD7864s in Multichannel System
DSP MICROPROCESSOR
IC
DAC
TORQUE & FLUX
CONTROL LOOP
CALCULATIONS &
TWO TO THREE
PHASE INFORMATION
TORQUE
SETPOINT
FLUX
SETPOINT
DRIVE
CIRCUITRY
DAC
IB
IA
3
PHASE
VA MOTOR
VB
DAC
+
+
ISOLATION
AMPLIFIERS
–
–
TRANSFORMATION
TO TORQUE &
FLUX CURRENT
COMPONENTS
VIN1
VIN2
AD7864*
VIN3
VIN4
VOLTAGE
ATTENUATORS
*ADDITIONAL PINS OMITTED FOR CLARITY
Figure 23. Vector Motor Control Using the AD7864
–18–
REV. A
AD7864
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
0.548 (13.925)
0.546 (13.875)
0.096 (2.44)
MAX
0.037 (0.94)
0.025 (0.64)
0.398 (10.11)
0.390 (9.91)
8°
0.8°
33
23
34
22
C3226a–5–6/99
44-Lead Plastic Quad Flatpack
(S-44)
SEATING
PLANE
TOP VIEW
(PINS DOWN)
44
0.040 (1.02)
0.032 (0.81)
0.040 (1.02)
0.032 (0.81)
12
1
11
0.033 (0.84)
0.029 (0.74)
0.016 (0.41)
0.012 (0.30)
PRINTED IN U.S.A.
0.083 (2.11)
0.077 (1.96)
REV. A
–19–
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