AD AD9874ABST If digitizing subsystem Datasheet

IF Digitizing Subsystem
AD9874*
FEATURES
10 MHz to 300 MHz Input Frequency
7.2 kHz to 270 kHz Output Signal Bandwidth
8.1 dB SSB NF
0 dBm IIP3
AGC Free Range up to –34 dBm
12 dB Continuous AGC Range
16 dB Front End Attenuator
Baseband I/Q 16-Bit (or 24-Bit) Serial Digital Output
LO and Sampling Clock Synthesizers
Programmable Decimation Factor, Output Format,
AGC, and Synthesizer Settings
370 ⍀ Input Impedance
2.7 V to 3.6 V Supply Voltage
Low Current Consumption: 20 mA
48-Lead LQFP Package (1.4 mm Thick)
GENERAL DESCRIPTION
APPLICATIONS
Multimode Narrow-Band Radio Products
Analog/Digital UHF/VHF FDMA Receivers
TETRA, APCO25, GSM/EDGE
Portable and Mobile Radio Products
Base Station Applications
SATCOM Terminals
The SPI port programs numerous parameters of the AD9874,
thus allowing the device to be optimized for any given application.
Programmable parameters include synthesizer divide ratios, AGC
attenuation and attack/decay time, received signal strength level,
decimation factor, output data format, 16 dB attenuator, and the
selected bias currents. The bias currents of the LNA and mixer
can be further reduced at the expense of degraded performance
for battery-powered applications.
The AD9874 is a general-purpose IF subsystem that digitizes a
low level 10 MHz to 300 MHz IF input with a signal bandwidth
ranging from 6.8 kHz to 270 kHz. The signal chain of the AD9874
consists of a low noise amplifier, a mixer, a band-pass sigma-delta
analog-to-digital converter, and a decimation filter with programmable decimation factor. An automatic gain control (AGC) circuit
gives the AD9874 12 dB of continuous gain adjustment. Auxiliary blocks include both clock and LO synthesizers.
The AD9874’s high dynamic range and inherent antialiasing
provided by the band-pass sigma-delta converter allow the
AD9874 to cope with blocking signals up to 95 dB stronger
than the desired signal. This attribute can often reduce the cost of
a radio by reducing its IF filtering requirements. Also, it enables
multimode radios of varying channel bandwidths, allowing the
IF filter to be specified for the largest channel bandwidth.
FUNCTIONAL BLOCK DIAGRAM
MXOP MXON IF2P IF2N
GCP GCN
DAC
AD9874
AGC
–16dB
IFIN
⌺-⌬ ADC
LNA
DECIMATION
FILTER
FORMATTING/SSI
DOUTA
DOUTB
FS
CLKOUT
FREF
CONTROL LOGIC
LO
SYN
IOUTL
VOLTAGE
REFERENCE
CLK SYN
LOP LON
LO VCO AND
LOOP FILTER
IOUTC
CLKP
CLKN
VREFP VCM VREFN
SPI
PC
PD
PE
SYNCB
LOOP FILTER
*Protected by U.S. Patent No. 5,969,657;
REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective companies.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
www.analog.com
Fax: 781/326-8703
© 2003 Analog Devices, Inc. All rights reserved.
AD9874* PRODUCT PAGE QUICK LINKS
Last Content Update: 02/23/2017
COMPARABLE PARTS
DESIGN RESOURCES
View a parametric search of comparable parts.
• AD9874 Material Declaration
• PCN-PDN Information
EVALUATION KITS
• Quality And Reliability
• Evaluation Board for AD9864 and AD9874
• Symbols and Footprints
DOCUMENTATION
DISCUSSIONS
Data Sheet
View all AD9874 EngineerZone Discussions.
• AD9874: IF Digitizing Subsystem Data Sheet
SAMPLE AND BUY
REFERENCE MATERIALS
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Technical Articles
• Low-Power IC Digitizes 300 MHz IF
TECHNICAL SUPPORT
• MS-2210: Designing Power Supplies for High Speed ADC
Submit a technical question or find your regional support
number.
• MS-2735: Maximizing the Dynamic Range of SoftwareDefined Radio
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AD9874
TABLE OF CONTENTS
AD9874—SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . 3
ABSOLUTE MAXIMUM RATINGS . . . . . . . . . . . . . . . . . 5
PIN CONFIGURATION/DESCRIPTION . . . . . . . . . . . . . 6
DEFINITION OF SPECIFICATIONS/
TEST METHODS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
TYPICAL PERFORMANCE CHARACTERISTICS . . . . . 8
SERIAL PERIPHERAL INTERFACE (SPI) . . . . . . . . . . . 13
SYNCHRONOUS SERIAL INTERFACE (SSI) . . . . . . . . 16
Synchronization Using SYNCB . . . . . . . . . . . . . . . . . . . . 18
Interfacing to DSPs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
POWER CONTROL . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
LO SYNTHESIZER . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Fast Acquire Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
CLOCK SYNTHESIZER . . . . . . . . . . . . . . . . . . . . . . . . . . 21
IF LNA/MIXER . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
BAND-PASS SIGMA DELTA (⌺-⌬) ADC . . . . . . . . . . . . 24
DECIMATION FILTER . . . . . . . . . . . . . . . . . . . . . . . . . . 26
VARIABLE GAIN AMPLIFIER WITH AGC . . . . . . . . . . 28
Variable Gain Control . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Automatic Gain Control . . . . . . . . . . . . . . . . . . . . . . . . . 29
System NF vs. VGA Control . . . . . . . . . . . . . . . . . . . . . . 31
APPLICATION CONSIDERATIONS . . . . . . . . . . . . . . . 32
Frequency Planning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Spurious Responses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
EXTERNAL PASSIVE COMPONENT
REQUIREMENTS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
APPLICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
Superheterodyne Receiver . . . . . . . . . . . . . . . . . . . . . . . . 34
Synchronization of Multiple AD9874s . . . . . . . . . . . . . . . 36
Split Path Rx Architecture . . . . . . . . . . . . . . . . . . . . . . . . 37
Hung Mixer Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
LAYOUT EXAMPLE
EVALUATION BOARD AND SOFTWARE . . . . . . . . . 38
OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . 39
REVISION HISTORY . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
–2–
REV. A
(VDDI = VDDF = VDDA = VDDC = VDDL = VDDD = VDDH = 2.7 V to 3.6 V,
AD9874–SPECIFICATIONS
VDDQ = VDDP = 2.7 V to 5.5 V, f = 18 MSPS, f = 109.65 MHz, f = 107.4 MHz, f = 16.8 MHz, unless otherwise noted.)
CLK
IF
LO
1
REF
Parameter
Temp
Test Level
SYSTEM DYNAMIC PERFORMANCE 2
SSB Noise Figure @ Min VGA Attenuation 3, 4
@ Max VGA Attenuation3, 4
Dynamic Range with AGC Enabled 3, 4
IF Input Clip Point @ Max VGA Attenuation 3
@ Min VGA Attenuation 3
Input Third Order Intercept (IIP3)
Gain Variation over Temperature
Full
Full
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
IV
IV
LNA + MIXER
Maximum RF and LO Frequency Range
LNA Input Impedance
Mixer LO Input Resistance
Full
25oC
25oC
IV
V
V
300
LO SYNTHESIZER
LO Input Frequency
LO Input Amplitude
FREF Frequency (for Sinusoidal Input ONLY)
FREF Input Amplitude
FREF Slew Rate
Minimum Charge Pump Current @ 5 V5
Maximum Charge Pump Current @ 5 V5
Charge Pump Output Compliance 6
Synthesizer Resolution
Full
Full
Full
Full
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
VI
VI
VI
IV
7.75
0.3
8
0.3
7.5
0.48
3.87
0.4
6.25
CLOCK SYNTHESIZER
CLK Input Frequency
CLK Input Amplitude
Minimum Charge Pump Output Current 5
Maximum Charge Pump Output Current 5
Charge Pump Output Compliance 6
Synthesizer Resolution
Full
Full
Full
Full
Full
Full
IV
IV
VI
VI
VI
IV
13
0.3
0.48
3.87
0.4
2.2
SIGMA-DELTA ADC
Resolution
Clock Frequency (fCLK)
Center Frequency
Pass-Band Gain Variation
Alias Attenuation
Full
Full
Full
Full
Full
IV
IV
V
IV
IV
16
13
GAIN CONTROL
Programmable Gain Step
AGC Gain Range (Continuous)
GCP Output Resistance
Full
Full
Full
V
V
IV
Full
OVERALL
Analog Supply Voltage
(VDDA, VDDF, VDDI)
Digital Supply Voltage
(VDDD, VDDC, VDDL)
Interface Supply Voltage 7
(VDDH)
Charge Pump Supply Voltage
(VDDP, VDDQ)
Total Current
High Performance Setting8
Low Power Mode8
Standby
Min
91
–20
–32
–5
Typ
Max
Unit
8.1
13
95
–19
–31
0
0.7
9.5
dB
dB
dB
dBm
dBm
dBm
dB
2
500
370//1.4
1
MHz
⍀//pF
k⍀
300
2.0
25
3
0.67
5.3
0.67
5.3
0.78
6.2
VDDP – 0.4
MHz
V p-p
MHz
V p-p
V/␮s
mA
mA
V
kHz
26
VDDC
0.78
6.2
VDDQ – 0.4
MHz
V p-p
mA
mA
V
kHz
24
26
Bits
MHz
MHz
dB
dB
fCLK/8
1.0
80
50
16
12
72.5
95
dB
dB
k⍀
VI
2.7
3.0
3.6
V
Full
VI
2.7
3.0
3.6
V
Full
VI
1.8
3.6
V
Full
VI
2.7
5.0
5.5
V
Full
Full
Full
VI
VI
VI
20
17
0.01
26.5
22
0.1
mA
mA
mA
+85
°C
OPERATING TEMPERATUR E RANGE
–40
NOTES
1
Standard operating mode: LNA/Mixer @ high bias setting, VGA @ Min ATTEN setting, synthesizers in normal (not fast acquire) mode, fCLK = 18 MHz, decimation
factor = 900, 16-bit digital output, and 10 pF load on SSI output pins.
2
This includes 0.9 dB loss of matching network.
3
AGC with DVGA enabled.
4
Measured in 10 kHz bandwidth.
5
Programmable in 0.67 mA steps.
6
Voltage span in which LO (or CLK) charge pump output current is maintained within 5% of nominal value of VDDP/2 (or VDDQ/2).
7
VDDH must be less than VDDD + 0.5 V.
8
Clock VCO off, add additional 0.7 mA with VGA @ Max ATTEN setting.
Specifications subject to change without notice.
REV. A
–3–
AD9874
DIGITAL SPECIFICATIONS
(VDDI = VDDF = VDDA = VDDC = VDDL = VDDD = VDDH = 2.7 V to 3.6 V, VDDQ = VDDP = 2.7 V to 5.5 V,
fCLK = 18 MSPS, fIF = 109.65 MHz, fLO = 107.4 MHz, fREF = 16.8 MHz, unless otherwise noted.)1
Parameter
Temp
Test Level
Min
DECIMATOR
Decimation Factor2
Pass-Band Width
Pass-Band Gain Variation
Alias Attenuation
Full
Full
Full
Full
IV
V
IV
IV
48
SPI-READ OPERATION (See Figure 1a)
PC Clock Frequency
PC Clock Period (tCLK)
PC Clock HI (tHI)
PC Clock LOW (tLOW)
PC to PD Setup Time (tDS)
PC to PD Hold Time (tDH)
PE to PC Setup Time (tS)
PC to PE Hold Time (tH)
Full
Full
Full
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
IV
IV
IV
SPI-WRITE OPERATION3 (See Figure 1b)
PC Clock Frequency
PC Clock Period (tCLK)
PC Clock HI (tHI)
PC Clock LOW (tLOW)
PC to PD Setup Time (tDS)
PC to PD Hold Time (tDH)
PC to PD (or DOUBT) Data Valid Time (tDV)
PE to PD Output Valid to Hi-Z (tEZ)
Full
Full
Full
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
IV
IV
IV
100
45
45
2
2
3
SSI (see Figure 2b)
CLKOUT Frequency
CLKOUT Period (tCLK)
CLKOUT Duty Cycle (tHI, tLOW)
CLKOUT to FS Valid Time (tV)
CLKOUT to DOUT Data Valid Time (tDV)
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
0.867
38.4
33
–1
–1
CMOS LOGIC INPUTS4
Logic “1” Voltage (VIH)
Logic “0” Voltage (VIL)
Logic “1” Current (VIH)
Logic “0” Current (VIL)
Input Capacitance
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
VDDH – 0.2
CMOS LOGIC OUTPUTS3, 4, 5
Logic “1” Voltage (VIH)
Logic “0” Voltage (VIL)
Full
Full
IV
IV
Typ
Max
Unit
960
50%
1.2
88
fCLKOUT
dB
dB
10
MHz
ns
ns
ns
ns
ns
ns
ns
10
MHz
ns
ns
ns
ns
ns
ns
ns
26
1153
67
+1
+1
MHz
ns
ns
ns
ns
100
45
45
2
2
5
5
8
3
50
0.5
10
10
3
VDDH – 0.2
0.2
V
V
µA
µA
pF
V
V
NOTES
1
Standard operating mode: high IIP3 setting, synthesizers in normal (not fast acquire) mode, f CLK = 18 MHz, decimation factor = 300, 10 pF load on SSI output pins:
VDDx = 3.0 V.
2
Programmable in steps of 48 or 60.
3
CMOS output mode with C LOAD = 10 pF and Drive Strength = 7.
4
Absolute Max and Min input/output levels are VDDH +0.3 V and –0.3 V.
5
IOL = 1 mA; specification is also dependent on Drive Strength setting.
Specifications subject to change without notice.
–4–
REV. A
AD9874
ABSOLUTE MAXIMUM RATINGS*
Parameter
With Respect to
Min
Max
Unit
VDDF, VDDA, VDDC, VDDD, VDDH,
VDDL, VDDI
GNDF, GNDA, GNDC, GNDD, GNDH,
GNDL, GNDI, GNDS
–0.3
+4.0
V
VDDF, VDDA, VDDC, VDDD, VDDH,
VDDL, VDDI
VDDR, VDDA, VDDC, VDDD, VDDH,
VDDL, VDDI
–4.0
+4.0
V
VDDP, VDDQ
GNDF, GNDA, GNDC, GNDD, GNDH,
GNDL, GNDI, GNDQ, GNDP, GNDS
GNDP, GNDQ
GNDF, GNDA, GNDC, GNDD, GNDH,
GNDL, GNDI, GNDQ, GNDP, GNDS
–0.3
–0.3
+6.0
+0.3
V
V
MXOP, MXON, LOP, LON, IFIN,
CXIF, CXVL, CXVM
GNDI
–0.3
VDDI + 0.3
V
PC, PD, PE, CLKOUT, DOUTA,
DOUTB, FS, SYNCB
GNDH
–0.3
VDDH + 0.3
V
IF2N, IF2P, GCP, GCN
VREFP, VREFN, RREF
IOUTC
IOUTL
CLKP, CLKN
FREF
Junction Temperature
Storage Temperature
Lead Temperature (10 sec)
GNDF
GNDA
GNDQ
GNDP
GNDC
GNDL
–0.3
–0.3
–0.3
–0.3
–0.3
–0.3
VDDF + 0.3
VDDA + 0.3
VDDQ + 0.3
VDDP + 0.3
VDDC + 0.3
VDDL + 0.3
150
+150
300
V
V
V
V
V
V
°C
°C
°C
–65
*Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied. Exposure to absolute maximum ratings for
extended periods may affect device reliability.
EXPLANATION OF TEST LEVELS
TEST LEVEL
THERMAL CHARACTERISTICS
Thermal Resistance
I.
48-Lead LQFP
JA = 76.2°C/W
100% production tested.
II. 100% production tested at 25°C and sample tested at
specified temperatures. AC testing done on sample basis.
JC = 17°C/W
III. Sample tested only.
IV. Parameter is guaranteed by design and/or
characterization testing.
V. Parameter is a typical value only.
VI. All devices are 100% production tested at 25°C; min and
max guaranteed by design and characterization for industrial
temperature range.
ORDERING GUIDE
Model
Temperature Range
Package Description
Package Option
AD9874ABST
AD9874EB
–40°C to +85°C
48-Lead Thin Plastic Quad Flatpack (LQFP)
Evaluation Board
ST-48
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although the
AD9874 features proprietary ESD protection circuitry, permanent damage may occur on devices
subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended
to avoid performance degradation or loss of functionality.
REV. A
–5–
AD9874
GNDP
IOUTL
VDDP
VDDL
CXVM
LON
LOP
CXVL
GNDI
CXIF
IFIN
VDDI
PIN CONFIGURATION
48 47 46 45 44 43 42 41 40 39 38 37
MXOP 1
MXON 2
36 GNDL
PIN 1
IDENTIFIER
35 FREF
GNDF 3
34 GNDS
IF2N 4
33 SYNCB
IF2P 5
VDDF 6
AD9874
GCP 7
TOP VIEW
(Not to Scale)
32 GNDH
31 FS
30 DOUTB
GCN 8
29 DOUTA
VDDA 9
28 CLKOUT
GNDA 10
27 VDDH
VREFP 11
26 VDDD
VREFN 12
25 PE
PC
PD
GNDC
CLKP
CLKN
GNDS
GNDD
RREF
VDDQ
IOUTC
GNDQ
VDDC
13 14 15 16 17 18 19 20 21 22 23 24
PIN FUNCTION DESCRIPTIONS
Pin
Mnemonic
Description
Pin
Mnemonic
Description
1
2
3
4
5
6
7
8
9
10
11
12
13
MXOP
MXON
GNDF
IF2N
IF2P
VDDF
GCP
GCN
VDDA
GNDA
VREFP
VREFN
RREF
27
28
29
30
VDDH
CLKOUT
DOUTA
DOUTB
31
32
33
FS
GNDH
SYNCB
34
35
GNDS
FREF
36
37
38
GNDL
GNDP
IOUTL
14
15
VDDQ
IOUTC
39
VDDP
16
GNDQ
40
41
VDDL
CXVM
17
18
19
VDDC
GNDC
CLKP
42
LON
43
LOP
20
CLKN
44
CXVL
21
22
23
24
25
26
GNDS
GNDD
PC
PD
PE
VDDD
Mixer Output, Positive.
Mixer Output, Negative.
Ground for Front End of ADC.
Second IF Input (to ADC), Negative.
Second IF Input (to ADC), Positive.
Positive Power Supply for Front End of ADC.
Filter Capacitor for ADC Full-Scale Control.
Full-Scale Control Ground.
Positive Power Supply for ADC Back End.
Ground for ADC Back End.
Voltage Reference, Positive.
Voltage Reference, Negative.
Reference Resistor: Requires 100 kΩ to
GNDA.
Positive Power Supply for Clock Synthesizer.
Clock Synthesizer Charge Pump Output
Current.
Ground for Clock Synthesizer Charge
Pump.
Positive Power Supply for Clock Synthesizer.
Ground for Clock Synthesizer.
Sampling Clock Input/Clock VCO Tank,
Positive.
Sampling Clock Input/Clock VCO Tank,
Negative.
Substrate Ground.
Ground for Digital Functions.
Clock Input for SPI Port.
Data I/O for SPI Port.
Enable Input for SPI Port.
Positive Power Supply for Internal Digital
Function.
45
46
GNDI
CXIF
47
48
IFIN
VDDI
Positive Power Supply for Digital Interface.
Clock Output for SSI Port.
Data Output for SSI Port.
Data Output for SSI Port (Inverted) or
SPI Port.
Frame Sync for SSI Port.
Ground for Digital Interface.
Resets SSI and Decimator Counters;
Active Low.
Substrate Ground.
Reference Frequency Input for Both
Synthesizers.
Ground for LO Synthesizer.
Ground for LO Synthesizer Charge Pump.
LO Synthesizer Charge Pump Output
Current Charge Pump.
Positive Power Supply for LO Synthesizer
Charge Pump.
Positive Power Supply for LO Synthesizer.
External Filter Capacitor; DC Output of
LNA.
LO Input to Mixer and LO Synthesizer,
Negative.
LO Input to Mixer and LO Synthesizer,
Positive.
External Bypass Capacitor for LNA Power
Supply.
Ground for Mixer and LNA.
External Capacitor for Mixer V-I Converter Bias.
First IF Input (to LNA).
Positive Power Supply for LNA and Mixer.
–6–
REV. A
AD9874
DEFINITION OF SPECIFICATIONS/TEST METHODS
Dynamic Range (DR)
Single-Sideband Noise Figure (SSB NF)
Dynamic range is the measure of a small target input signal
(PTARGET) in the presence of a large unwanted interferer signal
(PINTER). Typically, the large signal will cause some unwanted
characteristic of the component or system to degrade, thus
making it unable to detect the smaller target signal correctly. In
the case of the AD9874, it is often a degradation in noise figure
at increased VGA attenuation settings that limits its dynamic
range (refer to TPCs 15a, 15b, and 15c).
Noise figure (NF) is defined as the degradation in SNR performance (in dB) of an IF input signal after it passes through a
component or system. It can be expressed with the equation
Noise Figure = 10 × log(SNRIN SNROUT )
The term SSB is applicable for heterodyne systems containing a
mixer. It indicates that the desired signal spectrum resides on
only one side of the LO frequency (i.e., single sideband); thus a
“noiseless” mixer has a noise figure of 3 dB.
The test method for the AD9874 is as follows. The small target
signal (an unmodulated carrier) is input at the center of the IF
frequency, and its power level (PTARGET) is adjusted to achieve an
SNRTARGET of 6 dB. The power of the signal is then increased by
3 dB prior to injecting the interferer signal. The offset frequency
of the interferer signal is selected so that aliases produced by
the decimation filter’s response as well as phase noise from the LO
(due to reciprocal mixing) do not fall back within the measurement
bandwidth. For this reason, an offset of 110 kHz was selected.
The interferer signal (also an unmodulated carrier) is then
injected into the input and its power level is increased to the
point (PINTER) where the target signal SNR is reduced to 6 dB.
The dynamic range is determined with the equation:
The AD9874’s SSB noise figure is determined by the equation
{
}
SSB NF = PIN − 10 × log( BW ) − 174 dBm Hz − SNR
where PIN is the input power of an unmodulated carrier, BW is
the noise measurement bandwidth, –174 dBm/Hz is the thermal
noise floor at 293 K, and SNR is the measured signal-to-noise
ratio in dB of the AD9874.
Note that PIN is set to –85 dBm to minimize any degradation in
measured SNR due to phase noise from the RF and LO signal
generators. The IF frequency, CLK frequency, and decimation
factors are selected to minimize any spurious components
falling within the measurement bandwidth. Note also that a
bandwidth of 10 kHz is used for the data sheet specification.
Refer to Figures 22a and 22b for an indication of how NF varies
with BW. Also, refer to the TPCs to see how NF is affected by
different operating conditions. All references to noise figures
within this data sheet imply single-sideband noise figure.
DR = PINTER – PTARGET + SNRTARGET
Note that the AD9874’s AGC is enabled for this test.
IF Input Clip Point
The IF input clip point is defined as 2 dB below the input power
level (PIN), resulting in the clipping of the AD9874’s ADC.
Unlike other linear components that typically exhibit a soft
compression (characterized by its 1 dB compression point), an
ADC exhibits a hard compression once its input signal exceeds
its rated maximum input signal range. In the case of the AD9874,
which contains a - ADC, hard compression should be avoided
because it causes severe SNR degradation.
Input Third Order Intercept (IIP3)
IIP3 is a figure of merit used to determine a component’s or
system’s susceptibility to intermodulation distortion (IMD)
from its third order nonlinearities. Two unmodulated carriers at
a specified frequency relationship (f1 and f2) are injected into a
nonlinear system exhibiting third order nonlinearities producing
IMD components at 2f1 – f2 and 2f2 – f1. IIP3 graphically represents the extrapolated intersection of the carrier’s input power
with the third order IMD component when plotted in dB. The
difference in power (D in dBc) between the two carriers and the
resulting third order IMD components can be determined from
the equation
D = 2 × ( IIP 3 – PIN )
REV. A
–7–
AD9874–Typical Performance Characteristics
(VDDI = VDDF = VDDA = VDDC = VDDL = VDDD = VDDH = VDDx, VDDQ = VDDP = 5.0 V, fCLK = 18 MSPS, fIF = 109.56 MHz, fLO = 107.4 MHz,
TA = 25C, LO = –5 dBm, LO and CLK Synthesizer Disabled, 16-Bit Data with AGC and DVGA enabled, unless otherwise noted.)1
9.5
100
9.5
9.0
9.0
+85C
–40C
+25C
8.5
+85C
+85C
8.5
40
NF – dB
+25C
60
NF – dB
PERCENTAGE – %
80
8.0
7.5
8.0
+25C
7.5
–40C
7.0
7.0
6.5
6.5
20
–40C
0
7.2
7.5
7.8
8.1
8.4
8.7
NOISE FIGURE – dB
6.0
2.7
9.0
3.0
3.3
6.0
2.7
3.6
3.0
3.3
VDDx – V
TPC 1a. CDF of SSB Noise Figure
(VDDx = 3.0 V, High Bias2)
TPC 1b. SSB Noise Figure vs. Supply
(High Bias2)
TPC 1c. SSB Noise Figure vs. Supply
(Low Bias3)
1.5
100
3.6
VDDx – V
0
1.0
+85C
+25C
0
+85C
60
40
–0.5
–1.0
–1.5
–40C
–2.0
20
–4
+25C
IIP3 – dBm
–40C
IIP3 – dBm
PERCENTAGE – %
–2
0.5
80
+85C
–6
+25C
–8
–40C
–2.5
–10
–3.0
0
–2
–3
–1
0
IIP3 – dBm
1
–3.5
2.7
2
3.0
3.3
–12
2.7
3.6
TPC 2a. CDF of IIP3 (VDDx = 3.0 V,
High Bias2)
98
97
97
–40C
+25C
96
DR – dB
96
DR – dB
PERCENTAGE – %
80
40
95
+85C
+25C
94
–40C
95
94
+85C
20
93
93
+85C
+25C
0
92
3.6
TPC 2c. IIP3 vs. Supply (Low Bias3)
98
–40C
3.3
VDDx – V
TPC 2b. IIP3 vs. Supply (High Bias2)
100
60
3.0
VDDx – V
93
94
95
96
DYNAMIC RANGE – dB
97
TPC 3a. CDF of Dynamic Range
(VDDx = 3.0 V, High Bias2)
98
92
2.7
3.0
3.3
3.6
VDDx – V
TPC 3b. Dynamic Range vs. Supply
(High Bias2)
92
2.7
3.0
3.3
3.6
VDDx – V
TPC 3c. Dynamic Range vs. Supply
(Low Bias3)
Data taken with Toko FSLM series 10 µH inductors.
High Bias corresponds to LNA_Mixer Setting of 33 in SPI Register 0x01.
3
Low Bias corresponds to LNA_Mixer Setting of 12 in SPI Register 0x01.
1
2
–8–
REV. A
AD9874
(VDDI = VDDF = VDDA = VDDC = VDDL = VDDD = VDDH = VDDx, VDDQ = VDDP = 5.0 V, fCLK = 18 MSPS, fIF = 109.56 MHz, fLO = 107.4 MHz,
TA = 25ⴗC, LO = –5 dBm, LO and CLK Synthesizer Disabled, 16-Bit Data with AGC and DVGA enabled, unless otherwise noted.)1
PERCENTAGE – %
80
–40ⴗC
+25ⴗC
+85ⴗC
60
40
20
–17.5
–17.5
–18.0
–18.0
–18.5
INPUT CLIP POINT – dBm
INPUT CLIP POINT – dBm
100
+85ⴗC
–19.0
+25ⴗC
–19.5
–40ⴗC
–20.5
2.7
–19.4 –19.2 –19.0 –18.8 –18.6 –18.4
IFIN CLIP POINT – dBm
+25ⴗC
+85ⴗC
40
20
–31.6 –31.4 –31.2 –31.0 –30.8 –30.6 –30.4
–19.5
–40ⴗC
–20.5
2.7
3.6
3.0
3.3
3.6
VDDx – V
TPC 4c. Maximum VGA Attenuation
Clip Point vs. Supply (Low Bias3)
–29.5
–29.5
–30.0
–30.0
–30.5
+85ⴗC
–31.0
+25ⴗC
–31.5
–40ⴗC
–32.0
2.7
3.0
IFIN CLIP POINT – dBm
TPC 5a. CDF of Minimum VGA
Attenuation Clip Point (VDDx = 3.0 V,
High Bias2)
–30.5
+85ⴗC
–31.0
+25ⴗC
–31.5
3.3
3.6
–32.0
2.7
TPC 5b. Minimium VGA Attenuation
Clip Point vs. Supply (High Bias2)
+25ⴗC
+85ⴗC
60
40
18
20
ANALOG
(IDDA, IDDF, AND IDDI)
10
8
DIGITAL
(IDDD, IDDC, AND IDDL)
4
DIGITAL INTERFACE
(IDDH)
2
0
18.5 19.0 19.5 20.0 20.5 21.0 21.5 22.0
SUPPLY CURRENT – mA
TPC 6a. CDF of Supply Current
(VDDx = 3.0 V, High Bias2)
16
12
6
0
13
3.6
TPC 5c. Minimium VGA Attenuation
Clip Point vs. Supply (Low Bias3)
SUPPLY CURRENT – mA
–40ⴗC
SUPPLY CURRENT – mA
14
3.3
VDDx – V
16
80
3.0
VDDx – V
100
15
17
19
21
23
25
fCLK – MHz
TPC 6b. Supply Current vs. fCLK
(VDDx = 3.0 V, High Bias2)
Data taken with Toko FSLM series 10 µH inductors.
High Bias corresponds to LNA_Mixer Setting of 33 in SPI Register 0x01.
3
Low Bias corresponds to LNA_Mixer Setting of 12 in SPI Register 0x01.
1
2
REV. A
+25ⴗC
–40ⴗC
0
PERCENTAGE – %
3.3
TPC 4b. Maximum VGA Attenuation
Clip Point vs. Supply (High Bias2)
INPUT CLIP POINT – dBm
PERCENTAGE – %
–40ⴗC
60
–19.0
VDDx – V
100
80
3.0
INPUT CLIP POINT – dBm
TPC 4a. CDF of Maximum VGA
Attenuation Clip Point (VDDx = 3.0 V,
High Bias2)
+85ⴗC
–20.0
–20.0
0
–18.5
–9–
ANALOG
(IDDA, IDDF, AND IDDI)
14
12
10
8
6
DIGITAL
(IDDD, IDDC, AND IDDL)
4
DIGITAL INTERFACE
(IDDH)
2
0
2.7
3.0
3.3
3.6
VDDx – V
TPC 6c. Supply Current vs. Supply
(High Bias2)
AD9874
(VDDI = VDDF = VDDA = VDDC = VDDL = VDDD = VDDH = VDDx, VDDQ = VDDP = 5.0 V, fCLK = 18 MSPS, fIF = 109.56 MHz, fLO = 107.4 MHz,
TA = 25ⴗC, LO = –5 dBm, LO and CLK Synthesizer Disabled, 16-Bit Data with AGC and DVGA enabled, unless otherwise noted.)1
0.1
9.0
0
8.8
0
–12
–10
–15
–0.2
HIGH BIAS
–0.3
–0.4
–0.5
–0.6
–30
8.2
NF-LOW BIAS
–40
8.0
7.8
–50
IMD-LOW BIAS
7.6
–60
7.4
–0.7
IMD-HIGH BIAS
7.2
–0.8
–20
–17
–14
–11
LO DRIVE – dBm
–8
7.0
–20
–5
TPC 7a. Normalized Gain Variation
vs. LO Drive (VDDx = 3.0 V)
–15
–10
–5
LO DRIVE – dBm
–18
–30
–33
–36
–36
–80
5
–2
–4
–60
–6
dBFS
–40
–80
3.6V
3.3V
3.0V
2.7V
–8
–12
–12
–14
–30
80
TPC 8a. Complex FFT of Baseband
I/Q for Single-Tone (High Bias)
NBW = 3.66kHz
fCLK = 18MHz
MAX VGA ATTEN
DEC–BY–120
–18
TPC 8c. Gain Compression vs. IFIN
(Low Bias3)
–70
–15
–55
–76
–18
–61
–21
–67
–24
–73
–27
–79
IMD = 74dBc
–94
3.0V
–100
3.3V
–106
–112
–100
3.6V
–118
–30
–33
–60
–40 –20
0
20
40
FREQUENCY – kHz
60
–15
–18
80
TPC 9a. Complex FFT of Baseband
I/Q for Dual Tone IMD (High Bias
with Each IFIN Tone @ –35 dBm)
–130
–51
–48
–45
–42 –39
IFIN – dBm
–36
–33
TPC 9b. IMD vs. IFIN (High Bias2)
–21
2.7V
–24
–27
3.0V
3.3V
–91
–97
–39
–103
–42
–109
–45
–30
PIN
–85
–36
–120
–124
IMD – dBc
2.7V
–88
–60
–140
–80
–14
–30 –28 –26 –24 –22 –20 –18 –16 –14
IFIN – dBm
–16
PIN
IMD – dBc
dBFS
–20
–82
–40
–80
–24 –22
IFIN – dBm
PIN – dBFS
–20
–26
TPC 8b. Gain Compression vs. IFIN
(High Bias2)
0
–18.2dBFS OUTPUT
–28
3.0V
2.7V
–8
–120
60
3.6V
–6
–10
–40 –20
0
20
40
FREQUENCY – kHz
0
3.3V
–4
–10
–60
–6
ADC DOES NOT GO INTO
HARD COMPRESSION
–100
–140
–80
–18
–12
IFIN – dBm
–2
dBFS
–20
–24
0
ADC GOES INTO
HARD COMPRESSION
NBW = 3.66kHz
fCLK = 18MHz
MAX VGA ATTEN
DEC–BY–120
–30
TPC 7c. Gain Compression vs. IFIN
with 16 dB LNA Attenuator Enabled
0
–2.8dBFS OUTPUT
–24
–27
–70
0
LOW BIAS
–21
TPC 7b. Noise Figure and IMD
vs. LO Drive (VDDx = 3.0 V)
0
dBFS
–20
NF-HIGH BIAS
8.4
–115
–51
–30
–33
–36
3.6V
–39
–42
–48
–45
–42 –39
IFIN – dBm
–36
–33
–45
–30
TPC 9c. IMD vs. IFIN (Low Bias3)
Data taken with Toko FSLM series 10 µH inductors.
High Bias corresponds to LNA_Mixer Setting of 33 in SPI Register 0x01.
3
Low Bias corresponds to LNA_Mixer Setting of 12 in SPI Register 0x01.
1
2
–10–
REV. A
PIN – dBFS
NOISE FIGURE – dBc
GAIN VARIATION – dB
LOW BIAS
dBm
8.6
–0.1
IMD w/ IFIN = –36 dBm – dBc
HIGH BIAS
AD9874
(VDDI = VDDF = VDDA = VDDC = VDDL = VDDD = VDDH = VDDx, VDDQ = VDDP = 5.0 V, fCLK = 18 MSPS, fIF = 109.56 MHz, fLO = 107.4 MHz,
TA = 25ⴗC, LO = –5 dBm, LO and CLK Synthesizer Disabled, 16-Bit Data with AGC and DVGA enabled, unless otherwise noted.)1
10.0
16-BIT DATA
w/ DVGA
ENABLED
16-BIT
I/Q DATA
w/ DVGA
ENABLED
9.5
NOISE FIGURE – dB
9.5
9.0
8.5
8.0
16-BIT
DATA
16-BIT DATA
w/ DVGA
ENABLED
9.0
8.5
7.5
10
8.5
24-BIT
DATA
100
CHANNEL BANDWIDTH – kHz
7.5
10
1000
TPC 10a. Noise Figure vs. BW (Minimum Attenuation, fCLK = 13 MSPS)
100
CHANNEL BANDWIDTH – kHz
7.5
10
1000
TPC 10b. Noise Figure vs. BW (Minimum Attenuation, fCLK = 18 MSPS)
11.0
BW = 12.04kHz
(K = 0, M = 8)
9.0
BW = 6.78kHz
(K = 0, M = 15)
8.5
BW = 135.42kHz
(K = 1, M = 1)
13
BW = 75kHz
(K = 0, M = 1)
12
NOISE FIGURE – dB
NOISE FIGURE – dB
BW = 27.08kHz
(K = 0, M = 3)
10.0
1000
14
13
10.5
100
CHANNEL BANDWIDTH – kHz
TPC 10c. Noise Figure vs. BW (Minimum Attenuation, fCLK = 26 MSPS)
14
11.5
9.5
24-BIT
DATA
16-BIT
DATA
9.0
8.0
8.0
24-BIT
I/Q DATA
NOISE FIGURE – dB
NOISE FIGURE – dB
16-BIT
I/Q DATA
9.5
NOISE FIGURE – dB
10.0
10.0
BW = 50kHz
(K = 0, M = 2)
11
10
BW = 15kHz
(K = 0, M = 9)
9
12 BW = 90.28kHz
(K = 1, M = 2)
11
10
BW = 27.08kHz
(K = 1, M = 9)
9
8.0
8
7
0
7.0
6
3
9
VGA ATTENUATION – dB
0
12
TPC 11a. Noise Figure vs. VGA
Attenuation (fCLK = 13 MSPS)
6
3
9
VGA ATTENUATION – dB
7
0
12
TPC 11b. Noise Figure vs. VGA
Attenuation (fCLK = 18 MSPS)
6
3
9
VGA ATTENUATION – dB
12
TPC 11c. Noise Figure vs. VGA
Attenuation (fCLK = 26 MSPS)
–30
–5
–30
–5
–30
–5
–40
–10
–40
–10
–40
–10
–50
–50
–15
PIN
–70
LOW BIAS
–80
–25
HIGH BIAS
–90
–30
–100
–35
–110
–40
–120
–42
–39
–36
–33
–30
–27
–24
–45
–80
–25
HIGH BIAS
–90
–30
–35
–110
–40
–120
–42
–39
–36
–33
–30
–27
–24
–45
IFIN – dBm
TPC 12b. IMD vs. IFIN (fCLK = 18 MSPS)
Data taken with Toko FSLM series 10 µH inductors.
High Bias corresponds to LNA_Mixer Setting of 33 in SPI Register 0x01.
3
Low Bias corresponds to LNA_Mixer Setting of 12 in SPI Register 0x01.
1
2
REV. A
LOW BIAS
–35
IFIN – dBm
TPC 12a. IMD vs. IFIN (fCLK = 13 MSPS)
–20
–70
–100
–110
–130
–45
IMD – dBc
–30
HIGH BIAS
–100
–60
–20
PIN – dBFS
–25
–90
IMD – dBc
–80
POUT – dBFS
LOW BIAS
–15
PIN
–60
–20
–70
–130
–45
–50
–15
PIN
–60
IMD – dBc
8
–11–
–40
–120
–130
–45
–42
–39
–36
–33
–30
–27
–24
–45
IFIN – dBm
TPC 12c. IMD vs. IFIN (fCLK = 26 MSPS)
PIN – dBFS
7.5
AD9874
(VDDI = VDDF = VDDA = VDDC = VDDL = VDDD = VDDH = VDDx, VDDQ = VDDP = 5.0 V, fCLK = 18 MSPS, fIF = 109.56 MHz, fLO = 107.4 MHz,
TA = 25C, LO = –5 dBm, LO and CLK Synthesizer Disabled, 16-Bit Data with AGC and DVGA enabled, unless otherwise noted.)1
13
4
13
16-BIT w/DVGA
16-BIT w/DVGA
12
12
2
11
0
10
24-BIT
9
8
7
6
0
10
9
24-BIT
TPC 13a. Noise Figure vs. Frequency
(Minimum Attenuation, fCLK = 18 MSPS,
BW = 10 kHz, High Bias)
–4
–6
7
–8
0
–10
50 100 150 200 250 300 350 400 450 500
FREQUENCY – MHz
LOW BIAS
0
50 100 150 200 250 300 350 400 450 500
FREQUENCY – MHz
TPC 13b. Noise Figure vs. Frequency
(Minimum Attenuation, fCLK = 18 MSPS,
BW = 10 kHz, Low Bias)
13
–2
8
6
50 100 150 200 250 300 350 400 450 500
FREQUENCY – MHz
IIP3 – dBm
NOISE FIGURE – dB
NOISE FIGURE – dB
HIGH BIAS
11
TPC 13c. Input IP3 vs. Frequency
(fCLK = 18 MSPS)
2
13
16-BIT w/DVGA
16-BIT w/DVGA
10
9
8
0
11
–2
IIP3 – dBm
11
10
9
–4
–6
8
LOW BIAS
24-BIT
24-BIT
7
6
HIGH BIAS
12
NOISE FIGURE – dB
NOISE FIGURE – dB
12
–8
7
0
6
50 100 150 200 250 300 350 400 450 500
FREQUENCY – MHz
TPC 14a. Noise Figure vs. Frequency
(Minimum Attenuation, fCLK = 26 MSPS,
BW = 24 kHz, High Bias)
20.0
0
–10
0
50 100 150 200 250 300 350 400 450 500
FREQUENCY – MHz
FREQUENCY – MHz
TPC 14b. Noise Figure vs. Frequency
(Minimum Attenuation, fCLK = 26 MSPS,
BW = 24 kHz, Low Bias)
128
TPC 14c. Input IP3 vs. Frequency
(fCLK = 26 MSPS)
256
16
15
224
15
14
192
13
160
12
128
16
AGC
50 100 150 200 250 300 350 400 450 500
128
AGC ATTN
18.5
112
17.0
96
15.5
80
14.0
64
16
8.0
0
–55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
INTERFERER LEVEL – dBm
TPC 15a. Noise Figure vs. Interferer
Level (16-Bit Data, BW = 12.5 kHz,
AGCR = 1, fINTERFERER = fIF + 110 kHz)
10
64
9
32
8
0
–50 –45 –40 –35 –30 –25 –20 –15 –10
INTERFERER LEVEL – dBm
TPC 15b. Noise Figure vs. Interferer
Level (16-Bit Data with DVGA, BW =
12.5 kHz, AGCR = 1, fINTERFERER =
fIF + 110 kHz)
96
13
12
64
NOISE FIGURE
11
10
32
9
8
–65
–55
–45
–35
–25
–15
0
–5
INTERFERER LEVEL – dBm
TPC 15c. Noise Figure vs. Interferer
Level (24-Bit Data, BW = 12.5 kHz,
AGCR = 1, fINTERFERER = fIF + 110 kHz)
Data taken with Toko FSLM series 10 µH inductors.
High Bias corresponds to LNA_Mixer Setting of 33 in SPI Register 0x01.
3
Low Bias corresponds to LNA_Mixer Setting of 12 in SPI Register 0x01.
1
2
–12–
REV. A
MEAN AGC ATTN VALUE
32
9.5
96
NOISE FIGURE
NOISE FIGURE – dBc
11.0
11
MEAN AGC ATTN VALUE
48
NOISE FIGURE
NOISE FIGURE – dBc
12.5
MEAN AGC ATTN VALUE
NOISE FIGURE – dBc
AGC ATTN
14
AD9874
SERIAL PERIPHERAL INTERFACE (SPI)
The serial peripheral interface (SPI) is a bidirectional serial port. It is used to load configuration information into the registers listed
below as well as to read back their contents. Table I provides a list of the registers that may be programmed through the SPI port.
Addresses and default values are given in hexadecimal form.
Table I. SPI Address Map
Address Bit
(Hex)
Breakdown Width
Default Value Name
Description
POWER CONTROL REGISTERS
0x00
(7:0)
8
0xFF
STBY
Standby Control Bits (REF, LO, CKO, CK, GC, LNAMX, Unused,
and ADC).
0x01
(7:6)
(5:4)
(3:2)
(1:0)
2
2
2
2
0
0
0
0
LNAB
MIXB
CKOB
ADCB
LNA Bias Current (0 = 0.5 mA, 1 = 1 mA, 2 = 2 mA, 3 = 3 mA).
Mixer Bias Current (0 = 0.5 mA, 1 = 1.5 mA, 2 = 2.7 mA, 3 = 4 mA).
CK Oscillator Bias (0 = 0.25 mA, 1 = 0.35 mA, 2 = 0.40 mA, 3 = 0.65 mA).
Do not use.
0x02
(7:0)
8
0x00
TEST
Factory Test Mode. Do not use.
0x03
(7)
(6:0)
1
7
0
0x00
ATTEN
Apply 16 dB attenuation in the front end.
AGCG(14:8) AGC Attenuation Setting (7 MSB of a 15-Bit Unsigned Word).
0x04
(7:0)
8
0x00
AGCG(7:0) AGC Attenuation Setting (8 LSB of a 15-Bit Unsigned Word).
Default corresponds to maximum gain.
0x05
(7:4)
(3:0)
4
4
0
0
AGCA
AGCD
AGC Attack Bandwidth Setting. Default yields 50 Hz raw loop bandwidth.
AGC Decay Time Setting. Default is decay time = attack time.
0x06
(7)
(6:4)
(3)
(2:0)
1
3
1
3
0
0
0
0
AGCV
AGCO
AGCF
AGCR
Enable digital VGA to increase AGC range by 12 dB.
AGC Overload Update Setting. Default is slowest update.
Fast AGC (Minimizes resistance seen between GCP and GCN).
AGC Enable/Reference Level (Disabled, 3 dB, 6 dB, 9 dB, 12 dB, 15 dB
below Clip).
3
1
4
0
4
Unused
K
M
Decimation Factor = 60 (M + 1), if K = 0; 48 (M + 1), if K = 1.
Default is Decimate-by-300.
AGC
DECIMATION FACTOR
0x07
(7:5)
(4)
(3:0)
LO SYNTHESIZER
0x08
(5:0)
6
0x00
LOR(13:8)
Reference Frequency Divisor (6 MSB of a 14-Bit Word).
0x09
(7:0)
8
0x38
LOR(7:0)
Reference Frequency Divisor (8 LSB of a 14-Bit Word).
Default (56) yields 300 kHz from f REF = 16.8 MHz.
0x0A
(7:5)
(4:0)
3
5
0x5
0x00
LOA
LOB(12:8)
“A” Counter (Prescaler Control Counter).
“B” Counter MSB (5 MSB of a 13-Bit Word).
Default LOA and LOB values yield 300 kHz from 73.35 MHz to 2.25 MHz.
0x0B
(7:0)
8
0x1D
LOB(7:0)
“B” Counter LSB (8 LSB of a 13-Bit Word).
0x0C
(6)
(5)
(4:2)
(1:0)
1
1
3
2
0
0
0
3
LOF
LOINV
LOI
LOTM
Enable fast acquire.
Invert charge pump (0 = source current to increase VCO frequency).
Charge Pump Current in Normal Operation. IPUMP = (LOI + 1) 0.625 mA.
Manual Control of LO Charge Pump (0 = Off, 1 = Up, 2 = Down,
3 = Normal).
0x0D
(5:0)
4
0x0
LOFA(13:8) LO Fast Acquire Time Unit (6 MSB of a 14-Bit Word).
0x0E
(7:0)
8
0x04
LOFA(7:0)
REV. A
LO Fast Acquire Time Unit (8 LSB of a 14-Bit Word).
–13–
AD9874
Table I. SPI Address Map (continued)
Address Bit
(Hex)
Breakdown
Width
Default Value
Name
Description
CLOCK SYNTHESIZER
0x10
(5:0)
6
00
CKR(13:8) Reference Frequency Divisor (6 MSB of a 14-Bit Word).
0x11
(7:0)
8
0x38
CKR(7:0)
0x12
(4:0)
5
0x00
CKN(12:8) Synthesized Frequency Divisor (5 MSB of a 13-Bit Word).
0x13
(7:0)
8
0x3C
CKN(7:0)
Synthesized Frequency Divisor (8 LSB of a 13-Bit Word).
Default yields 300 kHz from fCLK = 18 MHz; Min = 3, Max = 8191.
0x14
(6)
(5)
(4:2)
(1:0)
1
1
3
2
0
0
0
3
CKF
CKINV
CKI
CKTM
Enable fast acquire.
Invert charge pump (0 = source current to increase VCO frequency).
Charge Pump Current in Normal Operation. IPUMP = (CKI + 1) 0.625 mA.
Manual Control of CLK Charge Pump (0 = Off, 1 = Up, 2 = Down,
3 = Normal).
0x15
(5:0)
6
0x0
CKFA(13:8) CK Fast Acquire Time Unit (6 MSB of a 14-Bit Word).
0x16
(7:0)
8
0x04
CKFA(7:0) CK Fast Acquire Time Unit (8 LSB of a 14-Bit Word).
Reference Frequency Divisor (8 LSB of a 14-Bit Word).
Default yields 300 kHz from fREF =16.8 MHz; Min = 3, Max = 16383.
SSI CONTROL
0x18
(7:0)
8
0x12
SSICRA
SSI Control Register A. See Table III. (Default is FS and CLKOUT
three-stated.)
0x19
(7:0)
8
0x07
SSICRB
SSI Control Register B. See Table III. (16-bit data, maximum drive strength.)
0x1A
(3:0)
4
1
SSIORD
Output Rate Divisor. f CLKOUT = fCLK/SSIORD.
ADC TUNING
0x1C
(1)
(0)
1
1
0
0
TUNE_LC Perform tuning on the LC portion of the ADC (cleared when done).
TUNE_RC Perform tuning on the RC portion of the ADC (cleared when done).
0x1D
(2:0)
3
0
CAPL1(2:0) Coarse Capacitance Setting for LC Tank (LSB is 25 pF, Differential).
0x1E
(5:0)
6
0x00
CAPL0(5:0) Fine Capacitance Setting for LC Tank (LSB is 0.4 pF, Differential).
0x1F
(7:0)
8
0x00
CAPR
Capacitance Setting for RC Resonator (64 LSB of Fixed Capacitance).
TEST REGISTERS AND SPI PORT READ ENABLE
0x37–
0x39
(7:0)
8
0x00
TEST
Factory Test Mode. Do not use.
0x3A
(7:4, 2:0)
(3)
7
1
0x0
0
TEST
SPIREN
Factory Test Mode. Do not use.
Enable read from SPI port.
0x3B
(7:4, 2:0)
(3)
7
1
0x0
0
TEST
TRI
Factory Test Mode. Do not use.
Three-state DOUTB.
0x3C–
0x3E
(7:0)
1
0x00
TEST
Factory Test Mode. Do not use.
0x3F
(7:0)
8
Subject to
Change
ID
Revision ID (Read-Only); A write of 0x99 to this register is equivalent to
a power-on reset.
–14–
REV. A
AD9874
shifted into the data pin (PD) on the rising edge of the next
eight clock cycles. PE stays low during the operation and goes
high at the end of the transfer. If PE rises before the eight clock
cycles have passed, the operation is aborted.
SERIAL PORT INTERFACE (SPI)
The serial port of the AD9874 has 3-wire or 4-wire SPI capability,
allowing read/write access to all registers that configure the
device’s internal parameters. The default 3-wire serial communication port consists of a clock (PC), peripheral enable (PE), and
bidirectional data (PD) signal. The inputs to PC, PE, and PD
contain a Schmitt trigger with a nominal hysteresis of 0.4 V
centered about the digital interface supply (i.e., VDDH/2).
If PE stays low for an additional eight clock cycles, the destination address is incremented and another eight bits of data are
shifted in. Again, should PE rise early, the current byte is
ignored. By using this implicit addressing mode, the entire
chip can be configured with a single write operation. Registers identified as being subject to frequent updates, namely
those associated with power control and AGC operation, have
been assigned adjacent addresses to minimize the time required
to update them. Note that multibyte registers are big-endian
(the most significant byte has the lower address) and are updated
when a write to the least significant byte occurs.
A 4-wire SPI interface can be enabled by setting the MSB of the
SSICRB register (Reg. 0x19, Bit 7), resulting in the output data
also appearing on the DOUTB pin. Note that since the default
power-up state sets DOUTB low, bus contention is possible for
systems sharing the SPI output line. To avoid any bus contention,
the DOUTB pin can be three-stated by setting the fourth control
bit in the three-state bit (Reg 0x3B, Bit 3). This bit can then be
toggled to gain access to the shared SPI output line.
Figure 1b illustrates the timing for a read operation to the SPI
port. Although the AD9874 does not require read access for
proper operation, it is often useful in the product development
phase or for system authentication. Note that the readback
enable bit (Register 0x3A, Bit 3) must be set for a read operation with a 3-wire SPI interface. After the peripheral enable
(PE) signal goes low, data (PD) pertaining to the instruction
header is read on the rising edges of the clock (PC). A read
operation occurs if the read/not-write indicator is set high. After
the address bits of the instruction header are read, the eight data
bits pertaining to the specified register are shifted out of the
data pin (PD) on the falling edges of the next eight clock cycles.
If the 4-wire SPI interface is enabled, the eight data bits will
also appear on the DOUTB pin with the same timing relationship as those appearing at PD. After the last data bit is shifted
out, the user should return PE high, causing PD to become
three-stated and return to its normal status as an input pin.
Since the auto increment mode is not supported for read operations, an instruction header is required for each register read
operation and PE must return high before initiating the next
read operation.
An 8-bit instruction header must accompany each read and
write SPI operation. Only the write operation supports an autoincrement mode, allowing the entire chip to be configured in a
single write operation. The instruction header is shown in
Table II. It includes a read/not-write indicator bit, six address
bits, and a don’t care bit. The data bits immediately follow the
instruction header for both read and write operations. Note that
the address and data are always given MSB first.
Table II. Instruction Header Information
MSB
LSB
I7
I6
I5
I4
I3
I2
I1
I0
R/W
A5
A4
A3
A2
A1
A0
X
Figure 1a illustrates the timing requirements for a write operation to the SPI port. After the peripheral enable (PE) signal goes
low, data (PD) pertaining to the instruction header is read on
the rising edges of the clock (PC). To initiate a write operation,
the read/not-write bit is set low. After the instruction header is
read, the eight data bits pertaining to the specified register are
tCLK
tS
tH
PE
tHI
tLOW
PC
tDS
tDH
PD
A5
R/W
A4
A0
DON’T
CARE
D7
D6
D1
D0
Figure 1a. SPI Write Operation Timing
tCLK
tS
PE
tHI
tLOW
PC
tDV
tDS
tEZ
tDH
R/W
PD
DOUTB
REV. A
DON’T
CARE
DON’T
CARE
A5
DON’T
CARE
A1
A0
DON’T
CARE
D7
D6
D1
D0
DON’T
CARE
DON’T
CARE
D7
D6
D1
D0
Figure 1b. SPI Read Operation Timing
–15–
DON’T
CARE
AD9874
The two optional bytes follow the I and Q data as a 16-bit
word provided that the AAGC bit of SSICRA is not set. If
the AAGC bit is set, the two bytes follow the I and Q data in
an alternating fashion. In this alternate AGC data mode, the
LSB of the byte containing the AGC attenuation is a 0, while
the LSB of the byte containing reset and RSSI information is
always a 1.
Table III. SSI Control Registers
Name Width Default
SSICRA (ADDR = 0x18)
Q (24:0)
24-Bit I AND Q, EAGC = 1, AAGC = 0:64 DATA BITS
I (24:0)
ATTN (7:0)
Q (24:0)
SSI(5:0)
RESET COUNT
1
1
1
1
1
1
1
1
0
0
0
1
0
0
1
0
Alternate AGC Data Bytes.
Embed AGC data.
Embed frame sync.
Three-state frame sync.
Invert frame sync.
Late Frame Sync (1 = Late, 0 = Early).
Three-state CLKOUT.
Invert CLKOUT.
SSICRB (ADDR = 0x19)
16-Bit I AND Q, EAGC = 0, AAGC = X:32 DATA BITS
I (15:0)
Q (15:0)
16-Bit I AND Q, EAGC = 1, AAGC = 0:48 DATA BITS
I (15:0)
Q (15:0)
ATTN (7:0)
SSI(5:0)
16-Bit I AND Q, EAGC = 1, AAGC = 1:40 DATA BITS
I (15:0)
Q (15:0)
ATTN (7:1) 0
I (15:0)
Q (15:0)
SSI(5:1) 1
4_SPI
1
0
DW
1
0
DS
3
7
Enable 4-Wire SPI Interface for SPI Read
operation via DOUTB.
I/Q Data-Word Width (0 = 16 bit, 1 bit–24 bit).
Automatically 16-bit when the AGCV = 1.
FS, CLKOUT, and DOUT Drive
Strength.
SSIORD (ADDR = 0x1A)
RESET COUNT
Figure 2. SSI Frame Structure
DIV
The two optional bytes are output if the EAGC bit of SSICRA
is set. The first byte contains the 8-bit attenuation setting (0 =
no attenuation, 255 = 24 dB of attenuation), while the second
byte contains a 2-bit reset field and 6-bit received signal
strength field. The reset field contains the number of modulator reset events since the last report, saturating at 3. The received
signal strength (RSSI) field is a linear estimate of the signal
strength at the output of the first decimation stage; 60 corresponds to a full-scale signal.
4
1
DW
DS_2
DS_1
DS_0
AAGC
EAGC
EFS
SFST
SFSI
SLFS
SCKT
SCKI
24-Bit I AND Q, EAGC = 0, AAGC = X: 48 DATA BITS
I (24:0)
Description
DIV_3
DIV_2
DIV_1
DIV_0
The primary output of the AD9874 is the converted I and Q
demodulated signal available from the SSI port as a serial bit
stream contained within a frame. The output frame rate is equal
to the modulator clock frequency (fCLK) divided by the digital
filter’s decimation factor that is programmed in the Decimator
Register (0x07). The bit stream consists of an I word followed
by a Q word, where each word is either 24 bits or 16 bits long
and is given MSB first in twos complement form. Two optional
bytes may also be included within the SSI frame following the
Q word. One byte contains the AGC attenuation and the other
byte contains both a count of modulator reset events and an
estimate of the received signal amplitude (relative to full scale
of the AD9874’s ADC). Figure 2 illustrates the structure of the
SSI data frames in a number of SSI modes.
In a 2-wire interface, the embedded frame sync bit (EFS) within
the SSICRA register is set to 1. In this mode, the framing information is embedded in the data stream, with each eight bits of
data surrounded by a start bit (low) and a stop bit (high), and
each frame ends with at least 10 high bits. FS remains either
low or three-stated (default), depending on the state of the
SFST bit. Other control bits can be used to invert the frame
sync (SFSI), to delay the frame sync pulse by one clock
period (SLFS), to invert the clock (SCKI), or to three-state the
clock (SCKT). Note that if EFS is set, SLFS is a don’t care.
AAGC
EAGC
EFS
SFST
SFSI
SLFS
SCKT
SCKI
The AD9874 provides a high degree of programmability of its
SSI output data format, control signals, and timing parameters
to accommodate various digital interfaces. In a 3-wire digital
interface, the AD9874 provides a frame sync signal (FS), a
clock output (CLKOUT), and a serial data stream (DOUTA)
signal to the host device. In a 2-wire interface, the frame sync
information is embedded into the data stream, thus only
CLKOUT and DOUTA output signals are provided to the
host device. The SSI control registers are SSICRA, SSICRB,
and SSIORD. Table III shows the different bit fields associated
with these registers.
4_SPI
SYNCHRONOUS SERIAL INTERFACE (SSI)
Output Bit Rate Divisor
fCLKOUT = fCLK/SSIORD.
The SSIORD register controls the output bit rate (fCLKOUT) of
the serial bit stream. fCLKOUT can be set to equal the modulator
clock frequency (fCLK) or an integer fraction of it. It is equal to
fCLK divided by the contents of the SSIORD register. Note that
fCLKOUT should be chosen such that it does not introduce harmful spurs within the pass band of the target signal. Users must
verify that the output bit rate is sufficient to accommodate the
required number of bits per frame for a selected word size
and decimation factor. Idle (high) bits are used to fill out
each frame.
–16–
REV. A
AD9874
CLKOUT
FS
DOUT
I0
I15
Q15
Q14
Q0
SCKI = 0, SCKT = 0, SLFS = 0, SFSI = 0, EFS = 0, SFST = 0, EAGC = 0
CLKOUT
FS
I15
DOUT
Q15
I0
Q14
Q0
SCKI = 0, SCKT = 0, SLFS = 1, SFSI = 0, EFS = 0, SFST = 0, EAGC = 0
CLKOUT
FS
DOUT
I0
I15
Q15
Q14
Q0
ATTN7
ATTEN6
RSSI0
SCKI = 0, SCKT = 0, SLFS = 0, SFSI = 0, EFS = 0, SFST = 0, EAGC = 1, AAGC = 0
CLKOUT
FS
HI-Z
START
BIT
DOUT
I15
I8
START
BIT
STOP
BIT
I7
I0
START
BIT
STOP
BIT
Q15
SCKI = 0, SCKT = 0, SLFS = X, SFSI = X, EFS = 1, SFST = 1, EAGC = 0
SCKI = 0, SCKT = 0, SLFS = X, SFSI = X, EFS = 1, SFST = 0, EAGC = 0: AS ABOVE, BUT FS IS LOW
IDLE (HIGH) BITS
Figure 3a. SSI Timing for Several SSICRA Settings with 16-Bit I/Q Data
Table IV.
Number of Bits per Frame for Different SSICR Settings
DW
EAGC
EFS
AAGC
Number of Bits
per Frame
0 (16-bit)
0
0
1
1
1
1
0
0
1
1
1
0
1
0
0
1
1
0
1
0
0
1
NA
NA
0
1
0
1
NA
NA
0
1
0
32
49*
48
40
69*
59*
48
69*
64
56
89*
1
1
1
79*
1 (24-bit)
An example helps illustrate how the maximum SSIORD setting
is determined. Suppose a user selects a decimation factor of 600
(Register 0x07, K = 0, M = 9) and prefers a 3-wire interface
with a dedicated frame sync (EFS = 0) containing 24-bit data
(DW = 1) with nonalternating embedded AGC data included
(EAGC = 1, AAGC = 0). Referring to Table IV, each frame
will consist of 64 data bits. Using Equation 1, the maximum
SSIORD setting is 9 (= TRUNC(600/64)). Thus, the user
can select any SSIORD setting between 1 and 9.
Figure 3a illustrates the output timing of the SSI port for several
SSI control register settings with 16-bit I/Q data, while Figure 3b
shows the associated timing parameters. Note that the same timing
relationship holds for 24-bit I/Q data, with the exception that I
and Q word lengths now become 24 bits. In the default mode of
the operation, data is shifted out on rising edges of CLKOUT
after a pulse equal to a clock period is output from the Frame
Sync (FS) pin. As described above, the output data consists of a
16- or 24-bit I sample followed by a 16- or 24-bit Q sample,
plus two optional bytes containing AGC and status information.
tCLK
*The number of bits per frame with embedded frame sync (EFS = 1) assume at
least 10 idle bits are desired.
tHI
The maximum SSIORD setting can be determined by the equation
SSIORD ≤ TRUNC{(Dec. Factor ) /
(# of Bits per Frame )}
CLKOUT
(1)
REV. A
tV
FS
tDV
where TRUNC is the truncated integer value.
Table IV lists the number of bits within a frame for 16-bit and
24-bit output data formats for all of the different SSICR settings. The decimation factor is determined by the contents of
Register 0x07.
tLOW
DOUT
I15
I14
Figure 3b. Timing Parameters for SSI Timing*
*Timing parameters also apply to inverted CLKOUT or FS modes, with t DV
relative to the falling edge of the CLK and/or FS.
–17–
AD9874
The AD9874 also provides the means for controlling the
switching characteristics of the digital output signals via the
DS (drive strength) field of the SSICRB. This feature is useful
in limiting switching transients and noise from the digital output that may ultimately couple back into the analog signal path,
potentially degrading the AD9874’s sensitivity performance.
Figures 3c and 3d show how the NF can vary as a function of
the SSI setting for an IF frequency of 109.65 MHz. The following two observations can be made from these figures:
Table V. Typical Rise/Fall Times (25%) with
a 10 pF Capacitive Load for Each DS Setting
• The NF becomes more sensitive to the SSI output drive
strength level at higher signal bandwidth settings.
• The NF is dependent on the number of bits within an SSI
frame, becoming more sensitive to the SSI output drive
strength level as the number of bits is increased. As a result,
one should select the lowest possible SSI drive strength setting that still meets the SSI timing requirements.
10.0
9.8
NOISE FIGURE – dB
9.6
16-BIT I/O DATA
9.4
9.2
9.0
24-BIT I/O DATA
8.8
8.6
8.4
16-BIT I/O DATA
w/DVGA ENABLED
8.2
8.0
1
2
3
4
5
6
SSI OUTPUT DRIVE STRENGTH SETTING
7
Figure 3c. NF vs. SSI Output Drive Strength
(VDDx = 3.0 V, fCLK = 18 MSPS, BW = 10 kHz)
13
NOISE FIGURE – dB
24-BIT I/O DATA
12
16-BIT I/O DATA
w/DVGA ENABLED
10
Typ (ns)
0
1
2
3
4
5
6
7
13.5
7.2
5.0
3.7
3.2
2.8
2.3
2.0
Synchronization Using SYNCB
Many applications require the ability to synchronize one or more
AD9874 in a way that causes the output data to be precisely
aligned to an external asynchronous signal. For example, receiver
applications employing diversity often require synchronization of
multiple AD9874 digital outputs. Satellite communication applications using TDMA methods may require synchronization
between payload bursts to compensate for reference frequency
drift and Doppler effects.
SYNCB can be used for this purpose. It is an active-low signal
that clears the clock counters in both the decimation filter and
the SSI port. The counters in the clock synthesizers are not
reset because it is presumed that the CLK signals of multiple
chips would be connected. SYNCB also resets the modulator,
resulting in a large-scale impulse that must propagate through
the AD9874’s digital filter and SSI data formatting circuitry
before recovering valid output data. At a result, data samples
unaffected by this SYNCB induced impulse can be recovered
12 output data samples after SYNCB goes high (independent of
the decimation factor).
Figure 4a shows the timing relationship between SYNCB and
the SSI port’s CLKOUT and FS signals. SYNCB is an asynchronous active-low signal that must remain low for at least half
an input clock period (i.e., 1/(2 fCLK)). CLKOUT remains
high while FS remains low upon SYNCB going low. CLKOUT
will become active within one to two output clock periods upon
SYNCB returning high. FS will reappear several output cycles
later, depending on the digital filter’s decimation factor and the
SSIORD setting. Note that for any decimation factor and
SSIORD setting, this delay is fixed and repeatable. To verify
proper synchronization, the FS signals of the multiple AD9874
devices should be monitored.
14
11
DS
16-BIT I/O DATA
9
SYNCB
8
CLKOUT
7
1
2
3
4
5
6
SSI OUTPUT DRIVE STRENGTH SETTING
FS
7
Figure 3d. NF vs. SSI Output Drive Strength
(VDDx = 3.0 V, fCLK = 18 MSPS, BW = 75 kHz)
Table V lists the typical output rise/fall times as a function of
DS for a 10 pF load. Rise/fall times for other capacitor loads
can be determined by multiplying the typical values presented
in Table V by a scaling factor equal to the desired capacitive
load divided by 10 pF.
Figure 4a. SYNCB Timing
Interfacing to DSPs
The AD9874 connects directly to an Analog Devices programmable
digital signal processor (DSP). Figure 4b illustrates an example
with the Blackfin® series of ADSP-2153x processors. The Blackfin
DSP series is a family of 16-bit products optimized for telecommunications applications with its dynamic power management feature,
making it well suited for portable radio products. The code
compatible family members share the fundamental core attributes
of high performance, low power consumption, and the ease-of-use
advantages of a microcontroller instruction set.
–18–
REV. A
AD9874
SPI
SSI
PC
PE
PD
DOUTB
CLKOUT
FS
DOUTA
The AD9874 also allows control over the bias current in the LNA,
mixer, and clock oscillator. The effects on current consumption
and system performance are described in the section dealing
with the affected block.
ADSP-2153x
AD9874
SCK
SEL
MOSI
MISO
SPI-PORT
RSCLK
RFS
DR
LO SYNTHESIZER
SERIAL
PORT
The LO Synthesizer shown in Figure 5 is a fully programmable
PLL capable of 6.25 kHz resolution at input frequencies up to
300 MHz and reference clocks of up to 25 MHz. It consists of a
low noise digital phase-frequency detector (PFD), a variable
output current charge pump (CP), a 14-bit reference divider,
programmable A and B counters, and a dual-modulus 8/9 prescaler. The A (3-bit) and B (13-bit) counters, in conjunction
with the dual 8/9 modulus prescaler, implement an N divider
with N = 8 B + A. In addition, the 14-bit reference counter
(R Counter) allows selectable input reference frequencies, fREF,
at the PFD input. A complete PLL (phase-locked loop) can be
implemented if the synthesizer is used with an external loop
filter and VCO (voltage controlled oscillator).
Figure 4b. Example of AD9874 and ADSP-2153x Interface
As shown in Figure 4b, AD9874’s synchronous serial interface
(SSI) links the receive data stream to the DSP’s Serial Port
(SPORT). For AD9874 setup and register programming, the
device connects directly to ADSP-2153x’s SPI port. Dedicated
select lines (SEL) allow the ADSP-2153x to program and read
back registers of multiple devices using only one SPI port. The
DSP driver code pertaining to this interface is available on the
AD9874 web page (http://www.analog.com/Analog_Root/
static/techSupport/designTools/evaluationBoards/
ad9874blackfinInterfacing.html).
POWER CONTROL
To allow power consumption to be minimized, the AD9874
possesses numerous SPI programmable power-down and bias
control bits. The AD9874 powers up with all of its functional
blocks placed into a standby state (i.e., STBY register default is
0xFF). Each major block may then be powered up by writing
a 0 to the appropriate bit of the STBY register. This scheme
provides the greatest flexibility for configuring the IC to a specific application as well as for tailoring the IC’s power-down and
wake-up characteristics. Table VI summarizes the function of
each of the STBY bits. Note that when all the blocks are in
standby, the master reference circuit is also put into standby,
and thus the current is reduced by a further 0.4 mA.
The A, B, and R counters can be programmed via the following
registers: LOA, LOB, and LOR. The charge pump output current is programmable via the LOI register from 0.625 mA to
5.0 mA using the equation
IPUMP = ( LOI + 1) × 0.625 mA
An on-chip fast acquire function (enabled by the LOF bit)
automatically increases the output current for faster settling
during channel changes. The synthesizer may also be disabled
using the LO standby bit located in the STBY register.
fREF
Table VI. Standby Control Bits
STBY
Bit
Effect
Current
Reduction Wake-Up
(mA)1
Time (ms)
Voltage reference OFF;
all biasing shut down.
0.6
<0.1 (CREF
= 4.7 nF)
6:LO
LO synthesizer OFF,
IOUTL three-state.
1.2
Note 2
5:CKO
Clock Oscillator OFF.
1.1
Note 2
4:CK
Clock synthesizer OFF,
IOUTC three-state. Clock
buffer OFF if ADC is OFF.
1.3
Note 2
3:GC
Gain control DAC OFF.
GCP and GCN three-state.
0.2
Depends
on CGC
2:LNAMX LNA and Mixer OFF. CXVM, 8.2
CXVL, and CXIF three-state.
<2.2
1:Unused
0:ADC
ADC OFF; Clock Buffer OFF 9.2
if CLK synthesizer OFF; VCM
three-state; Clock to the digital
filter halted; Digital outputs
static.
<0.1
NOTES
1
When all blocks are in standby, the master reference circuit is also put into
standby, and thus the current is further reduced by 0.4 mA.
2
Wake-up time is dependent on programming and/or external components.
REV. A
REF
BUFFER
fREF
ⴜR
LOR
7:REF
(2)
PHASE/
FREQUENCY
DETECTOR
fLO
TO EXTERNAL
LOOP
CHARGE FILTER
PUMP
FAST
ACQUIRE
LOA, LOB
A, B
COUNTERS
ⴜ8/9
LO
BUFFER
fLO
FROM
VCO
Figure 5. LO Synthesizer
The LO (and CLK) synthesizer works in the following manner.
The externally supplied reference frequency, fREF, is buffered
and divided by the value held in the R counter. The internal
fREF is then compared to a divided version of the VCO frequency, fLO. The phase/frequency detector provides UP and
DOWN pulses whose widths vary, depending upon the difference in phase and frequency of the detector’s input signals. The
UP/DOWN pulses control the charge pump, making current
available to charge the external low-pass loop filter when there is
a discrepancy between the inputs of the PFD. The output of the
low-pass filter feeds an external VCO whose output frequency,
fLO, is driven such that its divided down version, fLO, matches
that of fREF, thus closing the feedback loop.
The synthesized frequency is related to the reference frequency
and the LO register contents as follows:
fLO = (8 × LOB + LOA ) / LOR × fREF
(3)
Note that the minimum allowable value in the LOB register is 3
and its value must always be greater than that loaded into LOA.
–19–
AD9874
An example may help illustrate how the values of LOA, LOB,
and LOR can be selected. Consider an application employing
a 13 MHz crystal oscillator (i.e., fREF = 13 MHz) with the
requirement that fREF = 100 kHz and fLO = 143 MHz (i.e.,
high side injection with fIF = 140.75 MHz and fCLK = 18 MSPS).
LOR is selected to be 130 such that fREF = 100 kHz. The
N-divider factor is 1430, which can be realized by selecting
LOB = 178 and LOA = 6.
The stability, phase noise, spur performance, and transient
response of the AD9874’s LO (and CLK) synthesizers are
determined by the external loop filter, the VCO, the N-divide
factor, and the reference frequency, FREF. A good overview
of the theory and practical implementation of PLL synthesizers (featured as a three-part series in Analog Dialogue) can
be found at:
• www.analog.com/library/analogDialogue/archives/33-03/
phase/index.html
• www.analog.com/library/analogDialogue/archives/33-05/
phase_locked/index.html
• www.analog.com/library/analogDialogue/archives/33-07/
phase3/index.html
Also, a free software copy of the Analog Devices ADIsimPLL,
a PLL synthesizer simulation tool, is available at www.analog.com.
Note that the ADF4112 model can be used as a close approximation to the AD9874’s LO synthesizer when using this software tool.
LOP
84k⍀
LO
BUFFER
~VDDL/2
LON
FREF
TO MIXER
LO PORT
500⍀
500⍀
1.75V
BIAS
Fast Acquire Mode
The fast acquire circuit attempts to boost the output current
when the phase difference between the divided-down LO
(i.e., fLO) and the divided-down reference frequency (i.e., fREF)
exceeds the threshold determined by the LOFA register. The
LOFA register specifies a divisor for the fREF signal that determines the period (T) of this divided-down clock. This period
defines the time interval used in the fast acquire algorithm to
control the charge pump current.
Assume for the moment that the nominal charge pump current
is at its lowest setting (i.e., LOI = 0) and denote this minimum
current by I0. When the output pulse from the phase comparator exceeds T, the output current for the next pulse is 2I0.
When the pulse is wider than 2T, the output current for the
next pulse is 3I0, and so forth, up to eight times the minimum
output current. If the nominal charge pump current is more
than the minimum value (i.e., LOI > 0), the preceding rule is
only applied if it results in an increase in the instantaneous
charge pump current. If the charge pump current is set to its
lowest value (LOI = 0) and the fast acquire circuit is enabled,
the instantaneous charge pump current will never fall below 2I0
when the pulsewidth is less than T. Thus, the charge pump
current when fast acquire is enabled is given by:
IPUMP −FA = I0 × {1 + max(1, LOI , Pulsewidth T )}
(4)
The recommended setting for LOFA is LOR/16. Choosing a
larger value for LOFA will increase T. Thus, for a given phase
difference between the LO input and the fREF input, the instantaneous charge pump current will be less than that available for
a LOFA value of LOR/16. Similarly, a smaller value for LOFA
will decrease T, making more current available for the same
phase difference. In other words, a smaller value of LOFA will
enable the synthesizer to settle faster in response to a frequency
hop than will a large LOFA value. Care must be taken to choose
a value for LOFA that is large enough (values greater than 4
recommended) to prevent the loop from oscillating back and
forth in response to a frequency hop.
NOTES
1. ESD DIODE STRUCTURES OMITTED FOR CLARITY.
2. FREF STBY SWITCHES SHOWN WITH LO SYNTHESIZER ON.
Table VII. SPI Registers Associated with LO Synthesizer
Figure 6. Equivalent Input of LO and REF Buffers
Figure 6 shows the equivalent input structures of the synthesizers’ LO and REF buffers (excluding the ESD structures).
The LO input is fed to the LO synthesizer’s buffer as well as
the AD9874’s mixer’s LO port. Both inputs are self-biasing
and thus tolerate ac-coupled inputs. The LO input can be
driven with a single-ended or differential signal. Single-ended
dc-coupled inputs should ensure sufficient signal swing above
and below the common-mode bias of the LO and REF buffers
(i.e., 1.75 V and VDDL/2). Note that the fREF input is slew rate
dependent and must be driven with input signals exceeding
7.5 V/␮s to ensure proper synthesizer operation. If this condition can not be met, an external logic gate can be inserted
prior to the fREF input to “square-up” the signal thus allowing a
fREF input frequency approching dc.
Address
(Hex)
Bit
Breakdown
Default
Width Value
Name
0x00
(7:0)
1
0xFF
STBY
0x08
(5:0)
6
0x00
LOR(13:8)
0x09
(7:0)
8
0x38
LOR(7:0)
0x0A
(7:5)
(4:0)
3
5
0x5
0x00
LOA
LOB(12:8)
0x0B
(7:0)
8
0x1D
LOB(7:0)
0x0C
(6)
(5)
(4:2)
(1:0)
1
1
3
2
0
0
0
0
LOF
LOINV
LOI
LOTM
0x0D
(3:0)
4
0x0
LOFA(13:8)
0x0E
(7:0)
8
0x04
LOFA(7:0)
–20–
REV. A
AD9874
is approximately determined by LOSC and the series equivalent
capacitance of COSC and CVAR. As a result, LOSC, COSC, and
CVAR should be selected to provide a sufficient tuning range to
ensure proper locking of the clock synthesizer.
CLOCK SYNTHESIZER
The clock synthesizer is a fully programmable integer-N PLL
capable of 2.2 kHz resolution at clock input frequencies up to
18 MHz and reference frequencies up to 25 MHz. It is similar
to the LO synthesizer described in Figure 5 with the following
exceptions:
The bias, IBIAS, of the negative-resistance core has four programmable settings. Lower equivalent Q of the LC tank circuit
may require a higher bias setting of the negative-resistance core
to ensure proper oscillation. RBIAS should be selected so the
common-mode voltage at CLKP and CLKN is approximately
1.6 V. The synthesizer may be disabled via the CK standby bit
to allow the user to employ an external synthesizer and/or VCO
in place of those resident on the IC. Note that if an external
CLK source or VCO is used, the clock oscillator must be disabled via the CKO standby bit.
• It does not include an 8/9 prescaler nor an A counter.
• It includes a negative-resistance core that, when used in conjunction with an external LC tank and varactor, serves as the VCO.
The 14-bit reference counter and 13-bit N-divider counter can
be programmed via registers CKR and CKN. The clock
frequency, fCLK, is related to the reference frequency by the
equation
fCLK = (CKN CKR ) × fREF
(5)
The charge pump current is programmable via the CKI register
from 0.625 mA to 5.0 mA using the equation:
I PUMP = (CKI + 1) × 0.625 mA
(6)
The fast acquire subcircuit of the charge pump is controlled by
the CKFA register in the same manner as the LO synthesizer is
controlled by the LOFA register. An on-chip lock detect function (enabled by the CKF bit) automatically increases the
output current for faster settling during channel changes. The
synthesizer may also be disabled using the CK standby bit
located in the STBY register.
VDDC = 3.0 V
LOOP
FILTER
RBIAS
RD
RF
COSC
LOSC
The phase noise performance of the clock synthesizer is dependent on several factors, including the CLK oscillator IBIAS
setting, charge pump setting, loop filter component values, and
internal fREF setting. Figures 7b and 7c show how the measured
phase noise attributed to the clock synthesizer varies (relative to
an external fCLK) as a function of the IBIAS setting and charge
pump setting for a –31 dBm IFIN signal at 73.35 MHz with an
external LO signal at 71.1 MHz. Figure 7b shows that the optimum phase noise is achieved with the highest IBIAS (CKO)
setting, while Figure 7c shows that the higher charge pump
values provide the optimum performance for the given loop
filter configuration. The AD9874 clock synthesizer and oscillator were set up to provide an fCLK of 18 MHz from an external
fREF of 16.8 MHz. The following external component values
were selected for the synthesizer: RF = 390 Ω, RD = 2 kΩ,
CZ = 0.68 µF, CP = 0.1 µF, COSC = 91 pF, LOSC = 1.2 µH, and
CVAR = Toshiba 1SV228 Varactor.
0.1F
CP
CVAR
0
CZ
–10
–20
–30
CLKP
CLKN
–40
AD9874
–50
dBc/Hz
IOUTC
VCM = VDDC – RBIAS I BIAS > 1.6V
fOSC > 1/{2 (LOSC (C VARACTOR//COSC))1/2}
–60
–70
–80
CKO = 2
–90
CLK OSC. BIAS
2
–100
IBIAS = 0.15 mA, 0.25 mA,
0.40 mA, OR 0.65 mA
–110
CKO = 0
CKO = 3
CKO = 1
EXT CLK
–120
–130
Figure 7a. External Loop Filter, Varactor, and LC
Tank Are Required to Realize a Complete Clock
Synthesizer
–140
–25
–15
–10
–5
0
5
10
15
20
FREQUENCY OFFSET – kHz
Figure 7b. CLK Phase Noise vs. IBIAS Setting (CKO)
(IF = 73.35 MHz, IF = 71.1 MHz, IFIN = –31 dBm,
fCLK = 18 MHz, fREF = 16.8 MHz) (CLK SYN Settings:
CKI = 7, CLR = 56, and CLN = 60 with fREF = 300 kHz)
The AD9874 clock synthesizer circuitry includes a negativeresistance core so that only an external LC tank circuit with a
varactor is needed to realize a voltage controlled clock oscillator
(VCO). Figure 7a shows the external components required to
complete the clock synthesizer along with the equivalent input
circuitry of the CLK input. The resonant frequency of the VCO
REV. A
–20
–21–
25
AD9874
0
2.7V TO 3.6V
–10
50
–20
–30
–40
dBc/Hz
L
L
–50
C
–60
–70
VDDI
–80
MXOP
M X ON
CP = 0
–90
CP = 2
CP = 4
CP = 6
–100
–110
EXT CLK
–120
–130
–140
–25
RBIAS
–20
–15
–10
–5
0
5
10
FREQUENCY OFFSET – kHz
15
20
CXVL
25
LO INPUT =
0.3V p-p TO
1.0V p-p
RGAIN
Figure 7c. CLK Phase Noise vs. Charge Pump Setting Bias
(IF = 73.35 MHz, IF = 71.1 MHz, –31 dBm, fCLK = 18 MHz,
fREF = 16.8 MHz) (CLK SYN Settings: CKO Bias = 3, CKR = 56,
and CKN = 60 with fREF = 300 kHz)
MULTI-TANH
V–I STAGE
RF
CXIF
CXVM
IFIN
Table VIII. SPI Registers Associated with CLK Synthesizer
Address
(Hex)
Bit
Breakdown Width
Default
Value
Name
0x00
(7:0)
8
0xFF
STBY
0x01
(3:2)
2
0
CKOB
0x10
(5:0)
6
00
CKR(13:8)
0x11
(7:0)
8
0x38
CKR(7:0)
0x12
(4:0)
5
0x00
CKN(12:8)
0x13
(7:0)
8
0x3C
CKN(7:0)
0x14
(6)
(5)
(4:2)
(1:0)
1
1
3
1
0
0
0
0
CKF
CKINV
CKI
CKTM
0x15
(3:0)
4
0x0
CKFA(13:8)
0x16
(7:0)
8
0x04
CKFA(7:0)
DC SERVO
LOOP
Figure 8. Simplified Schematic of AD9874’s LNA/Mixer
600
LNA BIAS = 0
550
LNA BIAS = 1
RESISTANCE – LNA BIAS = 2
500
450
LNA BIAS = 3
400
350
300
0
50
100
150
250
200
FREQUENCY – MHz
300
350
Figure 9a. The Shunt Input Resistance vs. the
Frequency of the AD9874’s IF1 Input
The AD9874 contains a single-ended LNA followed by a Gilbert-type active mixer, shown in Figure 8 with the required
external components. The LNA uses negative shunt feedback to
set its input impedance at the IFIN pin, thus making it dependent on the LNA bias setting and input frequency. It can be
modeled as approximately 370 Ω//1.4 pF (620%) for the higher
bias settings below 100 MHz. Figures 9a and 9b show the
equivalent input impedance versus frequency characteristics of
the AD9874 with all the LNA bias settings. The increase in shunt
resistance versus frequency can be attributed to the reduction in
bandwidth, thus the amount of negative feedback of the LNA.
Note that the input signal into IFIN should be ac-coupled via a
10 nF capacitor since the LNA input is self-biasing.
2.5
LNA BIAS = 3
2.0
LNA BIAS = 2
CAPACITANCE – pF
IF LNA/MIXER
LNA BIAS = 1
1.5
1.0
LNA BIAS = 0
0.5
0
0
50
100
200
150
250
FREQUENCY – MHz
300
350
Figure 9b. The Shunt Capacitance vs.
the Frequency of the AD9874’s IF1 Input
–22–
REV. A
AD9874
The mixer’s differential LO port is driven by the LO buffer
stage shown in Figure 6, which can be driven single-ended or
differential. Since it is self-biasing, the LO signal level can be
ac-coupled and range from 0.3 V p-p to 1.0 V p-p with negligible
effect on performance. The mixer’s open-collector outputs,
MXOP and MXON, drive an external resonant tank consisting
of a differential LC network tuned to the IF of the band-pass
- ADC (i.e., fIF2_ADC = fCLK/8). The two inductors provide a
dc bias path for the mixer core via a series resistor of 50 Ω, which
is included to dampen the common-mode response. The mixer’s
output must be ac-coupled to the input of the band-pass - ADC,
IF2P, and IF2N via two 100 pF capacitors to ensure proper tuning
of the LC center frequency.
0
fIN = 109.65MHz
INPUT REFERRED POWER – dBm
–20
PIN
–40
–60
TOKO INDUCTOR
PIMD = 2.64 ⴛ PIN + 4.6
–80
–100
COILCRAFT
PIMD = 2.92 ⴛ PIN + 6.9
–120
–140
–54
The external differential LC tank forms the resonant element
for the first resonator of the band-pass - modulator, and so
must be tuned to the fCLK/8 center frequency of the modulator.
The inductors should be chosen such that their impedance at
fCLK/8 is about 140 Ω (i.e., L = 180/fCLK). An accuracy of 20%
is considered to be adequate. For example, at fCLK = 18 MHz,
L = 10 µH is a good choice. Once the inductors have been
selected, the required tank capacitance may be calculated using
the relation fCLK/8 = 1/{2 (2L C)1/2}.
–36
–42
–48
–30
–18
–24
Figure 10. IMD Performance between Different Inductors
with LNA and Mixer at Full Bias and fCLK of 18 MHz
The selection of the inductors is an important consideration in
realizing the full linearity performance of the AD9874. This is
true when operating the LNA and mixer at maximum bias and
low clock frequency. Figure 10 shows how the two-tone inputreferred IMD versus the input level performance at an IF of
109 MHz and fCLK of 18 MHz varies between Toko’s FSLM
series and Coilcraft’s 1812CS series inductors. The graph also
shows the extrapolated point of intersection used to determine
the IIP3 performance. Note that the Coilcraft inductor provides
a 7 dB to 8 dB improvement in performance and closely
approximates the 3:1 slope associated with a third order
linearity compared to the 2.65:1 slope associated with the
Toko inductor. The Coilcraft 1008CS series showed performance similar to that of the 1812CS series. It is worth noting
that the difference in IMD performance between these two
inductor families with an fCLK of 26 MHz is insignificant.
–20
13
12
–18
11
–16
10
–14
NOISE FIGURE
–12
9
8
1_0
1_1
1_2
1_3
2_0
2_1
2_2
2_3
3_0
3_1
3_2
–10
3_3
LNA_MIXER BIAS SETTING
Figure 11a. LNA/Mixer Noise Figure and
Conversion Gain vs. Bias Setting
9.50
5
8.25
LNA_MIXER CURRENT
–5
7.00
–10
5.75
IIP3
–15
4.50
–20
3.25
1_1
1_2 1_3
2_0 2_1
2_2
2_3 3_0
3_1 3_2
LNA_MIXER BIAS SETTING
Figure 11b. LNA/Mixer IIP3 and Current
Consumption vs. Bias Setting
–23–
2.00
3_3
IDDI – mA
INPUT IIP3 – dBm
0
–25
1_0
REV. A
CLIP POINT – dBm
CLIP POINT
NOISE FIGURE – dB
For example, at fCLK = 18 MHz and L = 10 µH, a capacitance of
250 pF is needed. However, in order to accommodate an inductor tolerance of 10%, the tank capacitance must be adjustable
from 227 pF to 278 pF. Selecting an external capacitor of
180 pF ensures that even with a 10% tolerance and stray capacitances as high as 30 pF, the total capacitance will be less than
the minimum value needed by the tank. Extra capacitance is
supplied by the AD9874’s on-chip programmable capacitor
array. Since the programming range of the capacitor array is at
least 160 pF, the AD9874 has plenty of range to make up for
the tolerances of low cost external components. Note that if fCLK
is increased by a factor of 1.44 MHz to 26 MHz so that fCLK/8
becomes 3.25 MHz, reducing L and C by approximately the
same factor (i.e., L = 6.9 µH and C = 120 pF) still satisfies the
requirements stated above.
Both the LNA and mixer have four programmable bias settings so
that current consumption can be minimized for a given application.
Figures 11a, 11b, and 11c show how the LNA and mixer’s noise
figure (NF), linearity (IIP3), IF clip point, current consumption,
and frequency response are affected for a given LNA/mixer bias
setting. The measurements were taken at an IF = 73.35 MHz and
LO = 71.1 MHz, with supplies set to 3 V.
AD9874
Based on these characterization curves, a LNA/mixer bias
setting of 3_3 is suitable for most applications since it will
provide the greatest dynamic range in the presence of multiple
unfiltered interferers. However, portable radio applications
demanding the lowest possible power may benefit by changing
the LNA/mixer bias setting based on the received signal
strength power (i.e., RSSI) available from the SSI output data.
For instance, selecting an LNA_Mixer bias setting of 1_2 for
nominal input strength conditions (i.e., <–45 dBm) would
result in 4 mA current savings (i.e., 18% reduction). If the
signal exceeds this level, a bias setting of 3_3 could be
selected. Refer to the Typical Performance Characteristics for
more performance graphs characterizing the LNA and mixer’s
effect upon the AD9874’s noise and linearity performance
under different operating conditions.
BAND-PASS SIGMA-DELTA (⌺-⌬) ADC
The ADC of the AD9874 is shown in Figure 12. The ADC
contains a sixth order multibit band-pass - modulator that
achieves very high instantaneous dynamic range over a narrow
frequency band. The loop filter of the band-pass - modulator
consists of two continuous-time resonators followed by a discretetime resonator, with each resonator stage contributing a pair of
complex poles. The first resonator is an external LC tank, while
the second is an on-chip active RC filter. The output of the LC
resonator is ac-coupled to the second resonator input via 100 pF
capacitors. The center frequencies of these two continuous-time
resonators must be tuned to fCLK/8 for the ADC to function
properly. The center frequency of the discrete-time resonator
automatically scales with fCLK, thus no tuning is required.
EXTERNAL
LC
0
–1
fCLK = 13 MSPS TO 26 MSPS
LNA_MIXER
3_3 SETTING
IF2P
–2
–3
dB
RC
RESONATOR
IF2N
MXOP
SC
RESONATOR
NINELEVEL
FLASH
MXON
–4
LNA_MIXER
1_2 SETTING
–5
–6
MIXER
OUTPUT
GAIN
CONTROL
–7
Figure 12. Equivalent Circuit of Sixth Order
Band-Pass - Modulator
–8
100
200
300
400
500
FREQUENCY – MHz
Figure 11c. LNA/Mixer Frequency Response vs. Bias Setting
A 16 dB step attenuator is also included within the LNA/
mixer circuitry to prevent large signals (i.e., > –18 dBm)
from overdriving the - modulator. In such instances, the
- modulator will become unstable, thus severely desensitizing
the receiver. The 16 dB step attenuator can be invoked by setting the ATTEN bit (Register 0x03, Bit 7), causing the mixer
gain to be reduced by 16 dB. The 16 dB step attenuator could
be used in applications in which a potential target or blocker
signal could exceed the IF input clip point. Although the LNA
will be driven into compression, it may still be possible to
recover the desired signal if it is FM. Refer to TPC 7c to see
the gain compression characteristics of the LNA and mixer
with the 16 dB attenuator enabled.
Figure 13a shows the measured power spectral density measured
at the output of the undecimated band-pass - modulator.
Note that the wide dynamic range achieved at the center frequency, fCLK/8, is achieved once the LC and RC resonators of
the - modulator have been successfully tuned. The out-ofband noise is removed by the decimation filters following
quadrature demodulation.
Table IX. SPI Registers Associated with LNA/Mixer
Address
(Hex)
Bit
Breakdown
0x00
0x01
0
–2dBFS OUTPUT
fCLK = 18MHz
NBW = 3.3kHz
–10
–20
–30
dBFS/NBW
0
TO DIGITAL
FILTER
ESL
DAC1
–40
–50
–60
–70
Width
Default
Value
Name
(7:0)
8
0xFF
STBY
(7:6)
2
0
LNAB
–80
–90
–100
0
1
2
3
4
5
6
7
8
9
FREQUENCY – MHz
0x01
(5:4)
2
0
MIXB
0x03
(7)
1
0
ATTEN
Figure 13a. Measured Undecimated Spectral Output of - Modulator ADC with fCLK = 18 MSPS
and Noise Bandwidth of 3.3 kHz
–24–
REV. A
AD9874
The signal transfer function of the AD9874 possesses inherent
antialias filtering by virtue of the continuous-time portions of
the loop filter in the band-pass - modulator. Figure 13b
illustrates this property by plotting the nominal signal transfer
function of the ADC for frequencies up to 2fCLK. The notches
that naturally occur for all frequencies that alias to the fCLK/8
pass band are clearly visible. Even at the widest bandwidth setting,
the notches are deep enough to provide greater than 80 dB of
alias protection. Thus, the wideband IF filtering requirements
preceding the AD9874 will be determined mostly by the mixer’s
image band, which is offset from the desired IF input frequency
by fCLK/4 (i.e., 2 3 fCLK/8) rather than any aliasing associated
with the ADC.
0
–10
–20
dB
–30
–50
–60
–70
0
0.5
1.0
1.5
When tuning the LC tank, the sampling clock frequency must
be stable and the LNA/mixer, LO synthesizer, and ADC must
all be placed in standby. Tuning is triggered when the ADC is
taken out of standby if the TUNE_LC bit of Register 0x1C has
been set. This bit will clear when the tuning operation is complete (less than 6 ms). The tuning codes can be read from the
3-bit CAPL1 (0x1D) and the 6-bit CAPL0 (0x1E) registers.
In a similar manner, tuning of the RC resonator is activated if
the TUNE_RC bit of Register 0x1C is set when the ADC is
taken out of standby. This bit will clear when tuning is complete. The tuning code can be read from the CAPR (0x1F)
register. Setting both the TUNE_LC and TUNE_RC bits tunes
the LC tank and the active RC resonator in succession. During
tuning, the ADC is not operational and neither data nor a clock
is available from the SSI port. Table X lists the recommended
sequence of the SPI commands for tuning the ADC, and Table XI
lists all of the SPI registers associated with band-pass - ADC.
NOTCH AT ALL ALIAS FREQUENCIES
–40
–80
Tuning of the - modulator’s two continuous-time resonators
is essential in realizing the ADC’s full dynamic range and must
be performed upon system startup. To facilitate tuning of the
LC tank, a capacitor array is internally connected to the MXOP
and MXON pins. The capacitance of this array is programmable from 0 pF to 200 pF 20% and can be programmed
either automatically or manually via the SPI port. The capacitors of the active RC resonator are similarly programmable.
Note that the AD9874 can be placed in and out of its standby
mode without retuning since the tuning codes are stored in the
SPI Registers.
2.0
NORMALIZED FREQUENCY – RELATIVE TO fOUT
Table X. Tuning Sequence
Figure 13b. Signal Transfer Function of the
Band-Pass - Modulator from 0 fCLK to 2 fCLK
Address Value Comments
Figure 13c shows the nominal signal transfer function magnitude for frequencies near the fCLK/8 pass band. The width of the
pass band determines the transfer function droop, but even at
the lowest oversampling ratio (48) where the pass band edges
are at fCLK/192 ( 0.005 fCLK), the gain variation is less than
0.5 dB. Note that the amount of attenuation offered by the
signal transfer function near fCLK/8 should also be considered
when determining the narrow-band IF filtering requirements
preceding the AD9874.
0x00
0x45
LO synthesizer, LNA/mixer, and ADC are
placed in standby.*
0x1C
0x03
Set TUNE_LC and TUNE_RC. Wait for
CLK to stabilize if CLK synthesizer used.
0x00
0x44
Take the ADC out of standby. Wait for
0x1C to clear (<6 ms). LNA/mixer can now
be taken out of standby.
*If external CLK VCO or source used, the CLK oscillator must also be disabled.
0
Table XI. SPI Registers Associated with Band-Pass - ADC
dB
–5
–10
Address
(Hex)
Bit
Breakdown
Width
Default
Value
0x00
(7:0)
8
0xFF
STBY
0x1C
(1)
(0)
1
1
0
0
TUNE_LC
TUNE_RC
0x1D
(2:0)
3
0
CAPL1(2:0)
0x1E
(5:0)
6
0x00
CAPL1(5:0)
0x1F
(7:0)
8
0x00
CAPR
Name
–15
–20
–0.10
–0.05
0
0.05
0.10
NORMALIZED FREQUENCY – RELATIVE TO fCLK
Figure 13c. Magnitude of the ADC’s Signal
Transfer Function near fCLK/8
REV. A
–25–
AD9874
Once the AD9874 has been tuned, the noise figure degradation
attributed solely to the temperature drift of the LC and RC
resonators is minimal. Since the drift of the RC resonator is
actually negligible compared to that of the LC resonator, the
external L and C components’ temperature drift characteristics
tend to dominate. Figure 13d shows the degradation in noise
figure as the product of the LC value is allowed to vary from
–12.5% to +12.5%. Note that the noise figure remains relatively
constant over a 3.5% range (i.e., 35,000 ppm), suggesting
that most applications will not be required to retune over the
operating temperature range.
Figure 15a shows the response of the decimation filter at a
decimation factor of 900 (K = 0, M = 14) and a sampling
clock frequency of 18 MHz. In this example, the output data
rate (fOUT) is 20 kSPS, with a usable complex signal bandwidth of 10 kHz centered around dc. As this figure shows,
the first and second alias bands (occurring at even integer
multiples of fOUT/2) have the least attenuation but provide at
least 88 dB of attenuation. Note that signals falling around
frequency offsets that are odd integer multiples of f OUT/2
(i.e., 10 kHz, 30 kHz, and 50 kHz) will fall back into the
transition band of the digital filter.
0
12
–20
BW = 75kHz
5.0kHz PASS BAND
FOLD–40 ING
POINT
dB
NF – dB
11
10
–88dB
–60
–88dB
–101dB
BW = 30kHz
–103dB
–80
9
BW = 10kHz
–100
–120
8
–15
–10
–5
0
5
10
0
15
10
20
30
LC ERROR – %
DECIMATION FILTER
The decimation filter shown in Figure 14 consists of an fCLK/8
complex mixer and a cascade of three linear phase FIR filters:
DEC1, DEC2, and DEC3. DEC1 downsamples by a factor of
12 using a fourth order comb filter. DEC2 also uses a fourth
order comb filter, but its decimation factor is set by the M field
of Register 0x07. DEC3 is either a decimate-by-5 FIR filter or a
decimate-by-4 FIR filter, depending on the value of the K bit
within Register 0x07. Thus, the composite decimation factor
can be set to either 60 M or 48 M for K equal to 0 or 1,
respectively.
DEC2
DEC3
SIN
SINC4
FILTER
12
SINC4 M + 1
FILTER
FIR
FILTER
100
–20
135.466kHz PASS BAND
dB
–40
–60
–98dB
–80
–115dB
–94dB
–100
I
DEC1
DATA
FROM -
MODULATOR
90
0
–120
K
80
Figure 15b shows the response of the decimation filter with a
decimation factor of 48 and a sampling clock rate of 26 MHz. The
alias attenuation is at least 94 dB and occurs for frequencies at the
edges of the fourth alias band. The difference between the alias
attenuation characteristics of Figure 15b and those of Figure 15a is
due to the fact that the third decimation stage decimates by a factor
of 5 for Figure 15a compared with a factor of 4 for Figure 15b.
The output data rate (fOUT) is equal to the modulator clock
frequency (fCLK) divided by the digital filter’s decimation factor.
Due to the transition region associated with the decimation
filter’s frequency response, the decimation factor must be
selected such that fOUT is equal to or greater than twice the
signal bandwidth. This ensures low amplitude ripple in the pass
band along with the ability to provide further application-specific digital filtering prior to demodulation.
M
70
Figure 15a. Decimation Filter Frequency Response
for fOUT = 20 kSPS (fCLK = 18 MHz, OSR = 900)
Figure 13d. Typical Noise Figure Degradation
from L and C Component Drift (fCLK = 18 MSPS,
fIF = 73.3501 MHz)
COS
40
50
60
FREQUENCY – kHz
0
0.5
1.0
1.5
FREQUENCY – MHz
2.0
2.5
Figure 15b. Decimation Filter Frequency Response
for fOUT = 541.666 kSPS (fCLK = 26 MHz, OSR = 48)
COMPLEX
4 DATA TO
OR SSI PORT
5 Q
Figure 14. Decimation Filter Architecture
–26–
REV. A
AD9874
Figures 16a and 16b show expanded views of the pass band for the
two possible configurations of the third decimation filter. When
decimating by 60n (K = 0), the pass-band gain variation is 1.2 dB;
when decimating by 48n (K = 1), the pass-band gain variation is
0.9 dB. Normalization of full scale at band center is accurate to
within 0.14 dB across all decimation modes. Figures 17a and 17b
show the folded frequency response of the decimator for K = 0
and K = 1, respectively.
0
–20
dB
–40
–60
3
–80
2
–100
MIN ALIAS ATTN = 87.7dB
PASS-BAND GAIN FREQUENCY = 1.2dB
–120
dB
1
0
0
0.25
NORMALIZED FREQUENCY – RELATIVE TO f OUT
0.50
Figure 17a. Folded Decimator Frequency Response for K = 0
–1
0
–2
–20
–3
0.125
NORMALIZED FREQUENCY – RELATIVE TO f OUT
–40
0.250
dB
0
Figure 16a. Pass-Band Frequency Response of
the Decimator for K = 0
–60
–80
3
MIN ALIAS ATTN = 97.2dB
–100
2
–120
PASS-BAND GAIN VARIATION = 0.9dB
0
dB
1
0.50
Figure 17b. Folded Decimator Frequency Response for K = 1
0
–1
–2
–3
0
0.125
NORMALIZED FREQUENCY – RELATIVE TO f OUT
0.250
Figure 16b. Pass-Band Frequency Response of
the Decimator for K = 1
REV. A
0.25
NORMALIZED FREQUENCY – RELATIVE TO f OUT
–27–
AD9874
- ADC
FS
DEC1
12
I/Q DATA
TO SSI
DEC2
AND
DEC3
DVGA
AGCR
REF LEVEL
I + Q
SELECT
LARGER
+
1
(1 – Z–1)
K
I + Q
AGCA/AGCD
SCALING
VGA
DAC
AGCV
SETTING
RSSI DATA
TO SSI
GCP
CDAC
Figure 18. Functional Block Diagram of VGA and AGC
VARIABLE GAIN AMPLIFIER OPERATION WITH
AUTOMATIC GAIN CONTROL
Variable Gain Control
The AD9874 contains both a variable gain amplifier (VGA) and
a digital VGA (DVGA) along with all of the necessary signal
estimation and control circuitry required to implement automatic gain control (AGC), as shown in Figure 18. The AGC
control circuitry provides a high degree of programmability,
allowing users to optimize the AGC response as well as the
AD9874’s dynamic range for a given application. The VGA is
programmable over a 12 dB range and implemented within the
ADC by adjusting its full-scale reference level. Increasing the
ADC’s full scale is equivalent to attenuating the signal. An
additional 12 dB of digital gain range is achieved by scaling the
output of the decimation filter in the DVGA. Note that a slight
increase in the supply current (i.e., 0.67 mA) is drawn from
VDDI and VDDF as the VGA changes from 0 dB to 12 dB
attenuation.
The lower 15 bits specify the attenuation in the remainder of
the signal path. If the DVGA is enabled, the attenuation range
is from –12 dB to +12 dB since the DVGA provides 12 dB of
digital gain. In this case, all 15 bits are significant. However,
with the DVGA disabled, the attenuation range extends from
0 dB to 12 dB and only the lower 14 bits are useful. Figure 19
shows the relationship between the amount of attenuation and
the AGC register setting for both cases.
12
ONLY
VGA ENABLED
AGC ATTENUATION – dB
The purpose of the VGA is to extend the usable dynamic range
of the AD9874 by allowing the ADC to digitize a desired signal
over a large input power range as well as recover a low level
signal in the presence of larger unfiltered interferers without
saturating or clipping the ADC. The DVGA is most useful in
extending the dynamic range in narrow-band applications
requiring a 16-bit I and Q data format. In these applications,
quantization noise resulting from internal truncation to 16 bits
as well as external 16-bit fixed point post-processing can
degrade the AD9874’s effective noise figure by 1 dB or more.
The DVGA is enabled by writing a 1 to the AGCV field. The
VGA (and the DVGA) can operate in either a user controlled
Variable Gain Mode or Automatic Gain Control (AGC) Mode.
The variable gain control is enabled by setting the AGCR field
of Register 0x06 to 0. In this mode, the gain of the VGA (and
the DVGA) can be adjusted by writing to the 16-bit AGCG
register. The maximum update rate of the AGCG register via
the SPI port is fCLK/240. The MSB of this register is the bit that
enables 16 dB of attenuation in the mixer. This feature allows
the AD9874 to cope with large level signals beyond the VGA’s
range (i.e., > –18 dBm at LNA input) to prevent overloading
of the ADC.
VGA
RANGE
6
DVGA AND
VGA ENABLED
0
DVGA
RANGE
–6
It is worth noting that the VGA imparts negligible phase error
upon the desired signal as its gain is varied over a 12 dB range.
This is due to the bandwidth of the VGA being far greater than
the downconverted desired signal (centered about fCLK/8) and
remaining relatively independent of gain setting. As a result,
phase modulated signals should experience minimal phase error
as the AGC varies the VGA gain while tracking an interferer or
the desired signal under fading conditions. Note that the envelope of the signal will still be affected by the AGC settings.
–12
0000
1FFF
3FFF
5FFF
7FFF
AGCG SETTING – HEX
Figure 19. AGC Gain Range Characteristics vs.
AGCG Register Setting with and without DVGA
Enabled
–28–
REV. A
AD9874
Referring to Figure 18, the gain of the VGA is set by an 8-bit control DAC that provides a control signal to the VGA appearing at
the gain control pin (GCP). For applications implementing automatic gain control, the DAC’s output resistance can be reduced
by a factor of 9 to decrease the attack time of the AGC response
for faster signal acquisition. An external capacitor, CDAC, from
GCP to analog ground is required to smooth the DAC’s output
each time it updates as well as to filter wideband noise. Note
that CDAC, in combination with the DAC’s programmable output resistance, sets the –3 dB bandwidth and time constant
associated with this RC network.
gain to ensure maximum digital gain while not exceeding the
programmable reference level.
A linear estimate of the received signal strength is performed at
the output of the first decimation stage (DEC1) and output of
the DVGA (if enabled) as discussed in the AGC section. This
data is available as a 6-bit RSSI field within an SSI frame with
60 corresponding to a full-scale signal for a given AGC attenuation setting. The RSSI field is updated at fCLK/60 and can be
used with the 8-bit attenuation field (or AGCG attenuation
setting) to determine the absolute signal strength.
Referring again to Figure 18, the majority of the AGC loop
operates in the discrete time domain. The sample rate of the
loop is fCLK/60; therefore, registers associated with the AGC
algorithm are updated at this rate. The number of overload and
ADC reset occurrences within the final I/Q update rate of the
AD9874, as well as the AGC value (8 MSB), can be read from
the SSI data upon proper configuration.
The accuracy of the mean RSSI reading (relative to the IF input
power) depends on the input signal’s frequency offset relative to
the IF frequency since both DEC1 filter’s response as well as
the ADC’s signal transfer function attenuate the mixer’s
downconverted signal level centered at fCLK/8. As a result, the
estimated signal strength of input signals falling within proximity to the IF is reported accurately, while those signals at
increasingly higher frequency offsets incur larger measurement errors. Figure 20 shows the normalized error of the
RSSI reading as a function of the frequency offset from the
IF frequency. Note that the significance of this error becomes
apparent when determining the maximum input interferer (or
blocker) levels with the AGC enabled.
0
MEASURED RSSI ERROR – dB
–3
–6
–9
–12
–15
–18
0
0.01
0.02
0.03
0.04
0.05
NORMALIZED FREQUENCY OFFSET – (fIN – fIF) fCLK
Figure 20. Normalized RSSI Error vs. Normalized
IF Frequency Offset
Automatic Gain Control (AGC)
The gain of the VGA (and DVGA) is automatically adjusted
when the AGC is enabled via the AGCR field of Register 0x06.
In this mode, the gain of the VGA is continuously updated at
fCLK/60 in an attempt to ensure that the maximum analog signal
level into the ADC does not exceed the ADC clip level and that
the rms output level of the ADC is equal to a programmable
reference level. With the DVGA enabled, the AGC control loop
also attempts to minimize the effects of 16-bit truncation noise
prior to the SSI output by continuously adjusting the DVGA’s
REV. A
This programmable level can be set at 3 dB, 6 dB, 9 dB, 12 dB,
and 15 dB below the ADC saturation (clip) level by writing
values from 1 to 5 to the 3-bit AGCR field. Note that the ADC
clip level is defined to be 2 dB below its full scale (i.e., –18 dBm
at the LNA input for a matched input and maximum attenuation). If AGCR is 0, automatic gain control is disabled. Since
clipping of the ADC input will degrade the SNR performance,
the reference level should also take into consideration the peakto-rms characteristics of the target (or interferer) signals.
The AGC performs digital signal estimation at the output of the
first decimation stage (DEC1) as well as the DVGA output that
follows the last decimation stage (DEC3). The rms power of the
I and Q signal is estimated by the equation
[]
( [ ])
( [ ])
Xest n = Abs I n + Abs Q n
(7)
Signal estimation after the first decimation stage allows the
AGC to cope with out-of-band interferers and in-band signals
that could otherwise overload the ADC. Signal estimation after
the DVGA allows the AGC to minimize the effects of the 16-bit
truncation noise.
When the estimated signal level falls within the range of the
AGC, the AGC loop adjusts the VGA (or DVGA) attenuation
setting so that the estimated signal level is equal to the programmed level specified in the AGCR field. The absolute signal
strength can be determined from the contents of the ATTN and
RSSI field that is available in the SSI data frame when properly
configured. Within this AGC tracking range, the 6-bit value in
the RSSI field remains constant while the 8-bit ATTN field
varies according to the VGA/DVGA setting. Note that the
ATTN value is based on the 8 MSB contained in the AGCG
field of Registers 0x03 and 0x04.
A description of the AGC control algorithm and the user adjustable parameters follows. First, consider the case in which the
in-band target signal is bigger than all out-of-band interferers
and the DVGA is disabled. With the DVGA disabled, a control
loop based only on the target signal power measured after
DEC1 is used to control the VGA gain, and the target signal
will be tracked to the programmed reference level. If the signal
is too large, the attenuation is increased with a proportionality
constant determined by the AGCA setting. Large AGCA values
result in large gain changes, thus rapid tracking of changes in
signal strength. If the target signal is too small relative to the
reference level, the attenuation is reduced; but now the proportionality constant is determined by both the AGCA and AGCD
settings. The AGCD value is effectively subtracted from AGCA,
so a large AGCD results in smaller gain changes and thus
slower tracking of fading signals.
The 4-bit code in the AGCA field sets the raw bandwidth of the
AGC loop. With AGCA = 0, the AGC loop bandwidth is at its
minimum of 50 Hz, assuming fCLK = 18 MHz. Each increment
of AGCA increases the loop bandwidth by a factor of 21/2, thus
–29–
AD9874
the maximum bandwidth is 9 kHz. A general expression for the
attack bandwidth is:
AGCA 2 )
(8)
BW = 50 × f
18 MHz × 2(
Hz
112
)
CLK
VGA ATTENUATION SETTING
(
A
128
and the corresponding attack time is:
AGCA 2 ) 
t
= 2.2 100 × π × 2(
(9)
 = 0.35 BWA
attack
assuming that the loop dynamics are essentially those of a
single-pole system.
The 4-bit code in the AGCD field sets the ratio of the attack
time to the decay time in the amplitude estimation circuitry.
When AGCD is zero, this ratio is one. Incrementing AGCD
multiplies the decay time constant by 21/2, allowing a 180:1
range in the decay time relative to the attack time. The decay
time may be computed from:
AGCD 2 )
t
=t
× 2(
(10)
decay attack
Figure 21a shows the AGC response to a 30 Hz pulse-modulated IF burst for different AGCA and AGCD settings.
96
AGCD = 8
64
48
AGCD = 0
32
16
VGA ATTENUATION SETTING
48
AGCD = 0
0
AGCA = 8
AGCD = 8
64
48
32
AGCD = 0
0
0
10
20
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1.0
TIME – ms
Figure 21b. AGC Response for Different AGCO
Settings with fCLK = 18 MSPS, fCLKOUT = 300 kSPS,
Decimate by 60, and AGCA = AGCD = 0
30
(11)
Now consider the case described above but with the DVGA
enabled to minimize the effects of 16-bit truncation. With the
DVGA enabled, a control loop based on the larger of the two
estimated signal levels (i.e., output of DEC1 and DVGA) is
used to control the DVGA gain. The DVGA multiplies the
output of the decimation filter by a factor of 1 to 4 (i.e., 0 dB to
12 dB). When signals are small, the DVGA gain is 4 and the
16-bit output is extracted from the 24-bit data produced by the
decimation filter by dropping 2 MSB and taking the next 16
bits. As signals get larger, the DVGA gain decreases to the point
where the DVGA gain is 1 and the 16-bit output data is simply
the 16 MSB of the internal 24-bit data. As signals get even
larger, attenuation is accomplished by the normal method of
increasing the ADC’s full scale.
16
16
AGCD = 0
where R is the resistance between the GCP pin and ground
(72.5 k 30% if AGCF = 0, < 8 k if AGCF = 1) and BW is
the raw loop bandwidth. Note that with C chosen at this upper
limit, the loop bandwidth increases by approximately 30%.
AGCD = 8
64
80
32
RC < 1 (8 πBW )
80
96
48
0
AGCA = 4
32
AGCO = 4
64
16
0
96
AGCO = 7
80
Lastly, the AGCF bit reduces the DAC source resistance by at
least a factor of 10. This facilitates fast acquisition by lowering
the RC time constant that is formed with the external capacitors
connected from the GCP pin-to-ground (GCN pin). For an
overshoot-free step response in the AGC loop, the capacitor
connected from the GCP pin to the GCN ground pin should be
chosen so that the RC time constant is less than one quarter of
the raw loop. Specifically:
AGCA = 0
80
96
40
50
TIME – ms
Figure 21a. AGC Response for Different AGCA
and AGCD Settings with fCLK = 18 MSPS,
fCLKOUT = 20 kSPS, Decimate by 900, and AGCO = 0
The 3-bit value in the AGCO field determines the amount of
attenuation added in response to a reset event in the ADC.
Each increment in AGCO doubles the weighting factor. At the
highest AGCO setting, the attenuation will change from 0 dB to
12 dB in approximately 10 µs, while at the lowest setting the
attenuation will change from 0 dB to 12 dB in approximately
1.2 ms. Both times assume fCLK = 18 MHz. Figure 21b shows
the AGC attack time response for different AGCO settings.
The extra 12 dB of gain range provided by the DVGA reduces
the input-referred truncation noise by 12 dB and makes the data
more tolerant of LSB corruption within the DSP. The price
paid for this extension to the gain range is that the start of AGC
action is 12 dB lower and that the AGC loop will be unstable if its
bandwidth is set too wide. The latter difficulty results from the
large delay of the decimation filters, DEC2 and DEC3, when
one implements a large decimation factor. As a result, given an
option, the use of 24-bit data is preferable to using the DVGA.
–30–
REV. A
AD9874
Table XII indicates which AGCA values are reasonable for
various decimation factors. The white cells indicate that the
(decimation factor/AGCA) combination works well; the light
gray cells indicate ringing and an increase in the AGC settling
time; and the dark gray cells indicate that the combination
results in instability or near instability in the AGC loop. Setting
AGCF = 1 improves the time-domain behavior at the expense
of increased spectral spreading.
Table XIII. SPI Registers Associated with AGC
Address
(Hex)
Bit
Breakdown
Width
Default
Value
Name
0x03
(7)
(6:0)
1
7
0
0x00
ATTEN
AGCG(14:8)
0x04
(7:0)
8
0x00
AGCG(7:0)
0x05
(7:4)
(3:0)
4
4
0
0x00
AGCA
AGCD
0x06
(7)
(6:4)
(3)
(2:0)
1
3
1
3
0
0
0
0
AGCV
AGCO
AGCF
AGCR
Table XII. AGCA Limits if the DVGA is Enabled
AGCA
M
60
6
7 8 9 10 11 12 13 14 15
0
System Noise Figure (NF) vs. VGA (or AGC) Control
120 1
300 4
540 8
900 E
Lastly, consider the case of a strong out-of-band interferer (i.e.,
–18 dBm to –32 dBm for matched IF input) that is larger than
the target signal and large enough to be tracked by the control
loop based on the output of the DEC1. The ability of the control loop to track this interferer and set the VGA attenuation to
prevent clipping of the ADC is limited by the accuracy of the
digital signal estimation occurring at the output of DEC1. The
accuracy of the digital signal estimation is a function of the
frequency offset of the out-of-band interferer relative to the IF
frequency as shown in Figure 20. Interferers at increasingly
higher frequency offsets incur larger measurement errors, potentially causing the control loop to inadvertently reduce the
amount of VGA attenuation that may result in clipping of the
ADC. Figure 21c shows the maximum measured interferer
signal level versus the normalized IF offset frequency (relative to
fCLK) tolerated by the AD9874 relative to its maximum target
input signal level (0 dBFS = –18 dBm). Note that the increase
in allowable interferer level occurring beyond 0.04 fCLK
results from the inherent signal attenuation provided by the
ADC’s signal transfer function.
The AD9874’s system noise figure is a function of the ACG
attenuation and output signal bandwidth. Figure 22a plots the
nominal system NF as a function of the AGC attenuation for
both narrow-band (20 kHz) and wideband (150 kHz) modes
with fCLK = 18 MHz. Also shown on the plot is the SNR that
would be observed at the output for a –2 dBFS input. The
high dynamic range of the ADC within the AD9874 ensures
that the system NF increases gradually as the AGC attenuation
is increased. In narrow-band (BW = 20 kHz) mode, the system
noise figure increases by less than 3 dB over a 12 dB AGC
range, while in wideband (BW = 150 kHz) mode, the degradation is about 5 dB. As a result, the highest instantaneous
dynamic range for the AD9874 occurs with 12 dB of AGC
attenuation, since the AD9874 can accommodate an additional 12 dB peak signal level with only a moderate increase
in its noise floor.
As Figure 22a shows, the AD9874 can achieve an SNR in
excess of 100 dB in narrow-band applications. To realize the
full performance of the AD9874 in such applications, it is recommended that the I/Q data be represented with 24 bits. If 16-bit
data is used, the effective system NF will increase because of the
quantization noise present in the 16-bit data after truncation.
RELATIVE TO CLIP POINT – dBFS
0
–3
15
SNR = 90.1dBFS
14
NOISE FIGURE – dB
DECIMATION FACTOR
4 5
13
BW = 50kHz
12
BW = 150kHz
11
SNR = 103.2dB
SNR = 82.9dBFS
10
–6
BW = 10kHz
9
–9
SNR = 95.1dBFS
8
0
6
9
12
Figure 22a. Nominal System Noise Figure and
Peak SNR vs. AGCG Setting (fIF = 73.35 MHz, fCLK =
18 MSPS, and 24-bit I/Q data)
–15
0
0.01
0.02
0.03
0.04
0.05
NORMALIZED FREQUENCY OFFSET = (fIN – fIF)/fCLK
Figure 21c. Maximum Interferer (or Blocker) Input
Level vs. Normalized IF Frequency Offset
REV. A
3
VGA ATTENUATION – dB
–12
–31–
AD9874
Figure 22b plots the nominal system NF with 16-bit output
data as a function of AGC in both narrow-band and wideband
mode. In wideband mode, the NF curve is virtually unchanged
relative to the 24-bit output data because the output SNR
before truncation is always less than the 96 dB SNR that 16-bit
data can support.
APPLICATION CONSIDERATIONS
Frequency Planning
However, in narrow-band mode, where the output SNR
approaches or exceeds the SNR that can be supported with 16-bit
data, the degradation in system NF is more severe. Furthermore, if the signal processing within the DSP adds noise at the
level of an LSB, the system noise figure can be degraded even
more than Figure 22b shows. For example, this could occur in a
fixed 16-bit DSP whose code is not optimized to process the
AD9874’s 16-bit data with minimal quantization effects. To
limit the quantization effects within the AD9874, the 24-bit
data undergoes noise shaping just prior to 16-bit truncation,
thus reducing the in-band quantization noise by 5 dB (with 23
oversampling). This explains why 98.8 dBFS SNR performance
is still achievable with 16-bit data in a 10 kHz BW.
17
SNR = 98.8dBFS
16
15
NOISE FIGURE – dB
BW = 10kHz
14
13
BW = 150kHz
12
11
SNR = 89.9dBFS
SNR = 94.1dBFS
10
BW = 50kHz
9
SNR = 83dBFS
8
0
3
6
VGA ATTENUATION – dB
9
The LO frequency (and/or ADC clock frequency) must be
chosen carefully to prevent known internally generated spurs
from mixing down along with the desired signal, thus degrading the SNR performance. The major sources of spurs in the
AD9874 are the ADC clock and digital circuitry operating at
1/3 of fCLK. Thus, the clock frequency (fCLK) is the most
important variable in determining which LO (and therefore
IF) frequencies are viable.
Many applications have frequency plans that take advantage of
industry-standard IF frequencies due to the large selection of
low cost crystal or SAW filters. If the selected IF frequency and
ADC clock rate result in a problematic spurious component, an
alternative ADC clock rate should be selected by slightly modifying the decimation factor and CLK synthesizer settings (if
used) such that the output sample rate remains the same. Also,
applications requiring a certain degree of tuning range should
take into consideration the location and magnitude of these
spurs when determining the tuning range as well as optimum IF
and ADC clock frequency.
Figure 23a plots the measured in-band noise power as a function of the LO frequency for fCLK = 18 MHz and an output
signal bandwidth of 150 kHz when no signal is present. Any LO
frequency resulting in large spurs should be avoided. As this
figure shows, large spurs result when the LO is fCLK/8 = 2.25 MHz
away from a harmonic of 18 MHz (i.e., n fCLK fCLK/8). Also
problematic are LO frequencies whose odd order harmonics
(i.e., m fLO) mix with harmonics of fCLK to fCLK/8. This spur
mechanism is a result of the mixer being internally driven by a
squared-up version of the LO input consisting of the LO frequency and its odd order harmonics. These spur frequencies
can be calculated from the relation
m fLO = (n ± 1 8) fCLK
12
Figure 22b. Nominal System Noise Figure and Peak SNR
vs. AGCG Setting (fIF = 73.35 MHz, fCLK = 18 MSPS, and
16-bit I/Q data)
(12)
where m = 1, 3, 5... and n = 1, 2, 3...
A second source of spurs is a large block of digital circuitry that
is clocked at fCLK/3. Problematic LO frequencies associated with
this spur source are given by:
f LO = fCLK /3 + n fCLK ± fCLK 8
(13)
where n = 1, 2, 3 ...
IN-BAND POWER – dBFS
–50
–60
–70
–80
–90
0
50
100
150
200
250
300
LO FREQUENCY – MHz
Figure 23a. Total In-Band Noise + Spur Power with No Signal Applied as a Function of the LO Frequency
(fCLK = 18 MHz and Output Signal Bandwidth of 150 kHz)
–32–
REV. A
AD9874
IN-BAND POWER – dBFS
–50
–60
–70
–80
–90
0
50
100
150
200
300
250
LO FREQUENCY – MHz
Figure 23b. Same as Figure 23a Excluding LO Frequencies Known to Produce Large In-Band Spurs
Figure 23b shows that omitting the LO frequencies given by
Equation 12 for m = 1, 3, and 5 and by Equation 13 accounts
for most of the spurs. Some of the remaining low level spurs can
be attributed to coupling from the SSI digital output. As a
result, users are also advised to optimize the output bit rate
(fCLKOUT via the SSIORD register) and the digital output driver
strength to achieve the lowest spurious and noise figure performance for a particular LO frequency and fCLK setting. This is
especially the case for particularly narrow-band channels in
which low level spurs can degrade the AD9874’s sensitivity
performance.
Spurious Responses
The spectral purity of the LO (including its phase noise) is an
important consideration since LO spurs can mix with undesired
signals present at the AD9874’s IFIN input to produce an in-band
response. To demonstrate the low LO spur level introduced within
the AD9874, Figure 25 plots the demodulated output power as a
function of the input IF frequency for an LO frequency of
71.1 MHz and a clock frequency of 18 MHz.
0
D = fCLK/4 = 4.5MHz
–20
Despite the many spurs, sweet spots in the LO frequency are
generally wide enough to accommodate the maximum signal
bandwidth of the AD9874. As evidence of this property, Figure 24 shows that the in-band noise is quite constant for LO
frequencies ranging from 70 MHz to 71 MHz.
dBFS
–40
DESIRED
RESPONSES
–60
–80
–50
IN-BAND POWER – dBFS
–100
–60
–120
50
60
70
80
90
100
IF FREQUENCY – MHz
–70
Figure 25. Response of AD9874 to a –20 dBm IF
Input when fLO = 71.1 MHz
–80
–90
70.0
70.5
71.0
LO FREQUENCY – MHz
Figure 24. Expanded View from 70 MHz to 71 MHz
The two large –10 dBFS spikes near the center of the plot are
the desired responses at fLO, fIF2_ADC, where fIF2_ADC = fCLK/8,
i.e., at 68.85 MHz and 73.35 MHz. LO spurs at fLO fSPUR
would result in spurious responses at offsets of fSPUR around the
desired responses. Close-in spurs of this kind are not visible on
the plot, but small spurious responses at fLO fIF2_ADC fCLK, i.e.,
at 50.85 MHz, 55.35 MHz, 86.85 MHz, and 91.35 MHz, are
visible at the –90 dBFS level. This data indicates that the AD9874
does an excellent job of preserving the purity of the LO signal.
Figure 25 can also be used to gauge how well the AD9874
rejects undesired signals. For example, the half-IF response (at
69.975 MHz and 72.225 MHz) is approximately –100 dBFS,
giving a selectivity of 90 dB for this spurious response. The
largest spurious response at approximately –70 dBFS occurs
with input frequencies of 70.35 MHz and 71.85 MHz. These
spurs result from third order nonlinearity in the signal path
(i.e., abs [3 fLO – 3 fIF_Input] = fCLK/8).
REV. A
–33–
AD9874
EXTERNAL PASSIVE COMPONENT REQUIREMENTS
10nF
1nF
3 GNDF
4 IF2N
5 IF2P
2.2nF
SYNCB 33
6 VDDF
7 GCP
GNDH 32
FS 31
AD9874
DOUTB 30
8 GCN
DOUTA 29
9 VDDA
CLKOUT 28
VDDH 27
10 GNDA
GNDD
GNDS
CLKN
CLKP
GNDC
VDDC
GNDQ
IOUTC
VDDQ
100pF
VDDD 26
RREF
11 VREFP
12 VREFN
10nF
PE 25
PC
100pF
GNDP
VDDP
VDDL
GNDL 36
FREF 35
GNDS 34
PD
100pF
IOUTL
1 MXOP
2 MXON
100
pF
CXVM
LOP
LON
GNDI
CXVL
IFIN
CXIF
48 47 46 45 44 43 42 41 40 39 38 37
VDDI
180pF
10nF
100nF
10H
10H
LC TANK
50
100nF
10nF
Figure 26 shows an example circuit using the AD9874 and
Table XIV shows the nominal dc bias voltages seen at the different pins. The purpose is to show the various external passive
components required by the AD9874, along with nominal dc
voltages for troubleshooting purposes.
13 14 15 16 17 18 19 20 21 22 23 24
100k
10nF
10nF
Figure 26. Example Circuit Showing Recommended
Component Values
Table XIV. Nominal DC Bias Voltages
Pin Number
Mnemonic
Nominal DC Bias (V)
1
2
4
5
11
12
13
19
20
35
41
42
43
44
46
47
MXOP
MXON
IF2N
IF2P
VREFP
VREFN
RREF
CLKP
CLKN
FREF
CXVM
LON
LOP
CXVL
CXIF
IFIN
VDDI – 0.2
VDDI – 0.2
1.3 – 1.7
1.3 – 1.7
VDDA/2 + 0.250
VDDA/2 – 0.250
1.2
VDDC – 1.3
VDDC – 1.3
VDDC/2
1.6 – 2.0
1.65 – 1.9
1.65 – 1.9
VDDI – 0.05
1.6 – 2.0
0.9 – 1.1
The LO, CLK, and IFIN signals are coupled to their respective
inputs using 10 nF capacitors. The output of the mixer is coupled
to the input of the ADC using 100 pF. An external 100 kΩ resistor
from the RREF pin to GND sets up the AD9874’s internal bias
currents. VREFP and VREFN provide a differential reference
voltage to the AD9874’s - ADC and must be decoupled by
a 0.01 µF differential capacitor along with two 100 pF capacitors to
GND. The remaining capacitors are used to decouple other sensitive internal nodes to GND.
Although power supply decoupling capacitors are not shown,
it is recommended that a 0.1 µF surface-mount capacitor be
placed as close as possible to each power supply pin for maximum effectiveness. Also not shown is the input impedance
matching network used to match the AD9874’s IF input to the
external IF filter. Lastly, the loop filter components associated
with the LO and CLK synthesizers are not shown.
LC component values for fCLK = 18 MHz are given on the diagram. For other clock frequencies, the two inductors and the
capacitor of the LC tank should be scaled in inverse proportion to
the clock. For example, if fCLK = 26 MHz, then the two inductors
should be = 6.9 µH and the capacitor should be about 120 pF. A
tolerance of 10% is sufficient for these components since tuning
of the LC tank is performed upon system startup.
APPLICATIONS
Superheterodyne Receiver Example
The AD9874 is well suited for analog and/or digital narrowband radio systems based on a superheterodyne receiver
architecture. The superheterodyne architecture is noted for
achieving exceptional dynamic range and selectivity by using
two or more downconversion stages to provide amplification
of the target signal while filtering the undesired signals. The
AD9874 greatly simplifies the design of these radio systems
by integrating the complete IF strip (excluding the LO VCO)
while providing an I/Q digital output (along with other system
parameters) for the demodulation of both analog and digital
modulated signals. The AD9874’s exceptional dynamic range
often simplifies the IF filtering requirements and eliminates the
need for an external AGC.
Figure 27 shows a typical dual conversion superheterodyne
receiver using the AD9874. An RF tuner is used to select and
downconvert the target signal to a suitable first IF for the
AD9874. A preselect filter may precede the tuner to limit the
RF input to the band of interest. The output of the tuner
drives an IF filter that provides partial suppression of adjacent channels and interferers that could otherwise limit the
receiver’s dynamic range. The conversion gain of the tuner
should be set such that the peak IF input signal level into the
AD9874 is no greater than –18 dBm to prevent clipping. The
AD9874 downconverts the first IF signal to a second IF that
is exactly 1/8 of the - ADC’s clock rate (i.e., fCLK/8) to simplify the digital quadrature demodulation process.
–34–
REV. A
AD9874
VDDA
AD9874
DAC AGC
–16dB
DOUTA
IFIN
LNA
GCN
GCP
II-2P
II-2N
IF CRYSTAL OR
SAW FILTER
TUNER
VXON
PRESELECT
RF
FILTER
INPUT
VXOP
IF2 = fCLK/8
- ADC
LNA
DECIMATION
FILTER
FORMATTING/SSI
DOUTB
FS
TO
DSP
CLKOUT
CONTROL LOGIC
VCO
SAMPLE CLOCK
SYNTHESIZER
LO
SYNTH.
LOOP
FILTER
VCO
SYNCB
PE
PD
PC
RREF
VREFN
VREFP
CLKN
CLKP
IOUTC
LON
LOP
REFIN
SPI
IOUTC
ADF42xx
PLL SYN
VOLTAGE
REFERENCE
LOOP
FILTER
VDDC
CRYSTAL
OSCILLATOR
FROM DSP
Figure 27. Typical Dual Conversion Superheterodyne Application Using the AD9874
This second IF signal is then digitized by the - ADC, demodulated into its quadrature I and Q components, filtered via matching
decimation filters, and reformatted to enable a synchronous serial
interface to a DSP. In this example, the AD9874’s LO and CLK
synthesizers are both enabled, requiring some additional passive
components (for the synthesizer’s loop filters and CLK oscillator)
and a VCO for the LO synthesizer. Note that not all of the
required decoupling capacitors are shown. Refer to the previous
section and Figure 26 for more information on required external
passive components.
The selection of the first IF frequency is often based on the
availability of low cost standard crystal or SAW filters as well as
system frequency planning considerations. In general, crystal
filters are often used for narrow-band radios having channel
bandwidths below 50 kHz with IFs below 120 MHz, while SAW
filters are more suited for channel bandwidths greater than
50 kHz with IFs greater than 70 MHz. The ultimate stop-band
rejection required by the IF filter will depend on how much
suppression is required at the AD9874’s image band resulting
from downconversion to the second IF. This image band is
offset from the first IF by twice the second IF frequency (i.e.,
fCLK/4, depending on high or low side injection).
The selectivity and bandwidth of the IF filter will depend on
both the magnitude and frequency offset(s) of the adjacent
REV. A
channel blocker(s) that could overdrive the AD9874’s input
or generate in-band intermodulation components. Further
suppression is performed within the AD9874 by its inherent
band-pass response and digital decimation filters. Note that
some applications will require additional application-specific
filtering performed in the DSP that follows the AD9874 to
remove the adjacent channel and/or implement a matched
filter for optimum signal detection.
The output data rate of the AD9874, fOUT, should be chosen
to be at least twice the bandwidth or symbol rate of the desired
signal to ensure that the decimation filters provide a flat passband response as well as to allow for postprocessing by a DSP.
Once fOUT is determined, the decimation factor of the digital
filters should be set such that the input clock rate, fCLK, falls
between the AD9874’s rated operating range of 13 MHz to
26 MHz and no significant spurious products related to fCLK fall
within the desired pass band, resulting in a reduction in sensitivity performance. If a spurious component is found to limit the
sensitivity performance, the decimation factor can often be
modified slightly to find a spurious free pass band. Selecting a
higher fCLK is typically more desirable given a choice, since
the first IF’s filtering requirements often depend on the transition region between the IF frequency and the image band
(i.e., fCLK/4 ). Lastly, the output SSI clock rate, fCLKOUT,
–35–
AD9874
and digital driver strength should be set to their lowest possible settings to minimize the potential harmful effects of
digital induced noise while preserving a reliable data link to
the DSP. Note that the SSICRA, SSICRB, and SSIORD
registers (i.e., 0x18, 0x19, and 0x1A) provide a large degree
of flexibility for optimization of the SSI interface.
VDDC
RBIAS
0.1F
LOOP
FILTER
COSC
RD
LOSC
CVAR
CP
RF
CZ
Synchronization of Multiple AD9874s
Some applications such as receiver diversity and beam steering
may require two or more AD9874s operating in parallel while
maintaining synchronization. Figure 28 shows an example of
how multiple AD9874s can be cascaded, with one device serving as the master and the other devices serving as the slaves. In
this example, all of the devices have the same SPI register configuration since they share the same SPI interface to the DSP.
Since the state of each of the AD9874’s internal counters is
unknown upon initialization, synchronization of the devices is
required via a SYNCB pulse (see Figure 4) to synchronize their
digital filters and ensure precise time alignment of the data
streams.
15
IOUTC
fREF
35
FROM
CRYSTAL
OSCILLATION
19 CLKP
20 CLKN
47 IFIN
AD9874
MASTER
FS 31
DOUTA 29
TO DSP
CLKOUT 28
43 LOP
PE 25
42 LON
PD 24
Although all of the devices’ synthesizers are enabled, the LO
and CLK signals for the slaves(s) are derived from the masters’
synthesizers and are referenced to an external crystal oscillator.
All of the necessary external components (i.e., loop filters,
varactor, LC, and VCO) required to ensure proper closed-loop
operation of these synthesizers are included.
FROM
DSP
PC 23
SYNCB 33
IOUTL
38
VCO
LOOP
FILTER
Note that although the VCO output of the LO synthesizer is
ac-coupled to the slave’s LO input(s), all of the CLK inputs of
the devices must be dc-coupled if the AD9874’s CLK oscillators
are enabled. This is due to the dc current required by the CLK
oscillators in each device. In essence, these negative impedance
cores are operating in parallel, increasing the effective Q of the
LC resonator circuit. Note that RBIAS should be sized such
that the sum of the oscillators’ dc bias currents maintains a
common-mode voltage of around 1.6 V.
15
IOUTC
PE 25
47 IFIN
PD 24
PC 23
43 LOP
42 LON
SYNCB 33
AD9874
SLAVE
TO OTHER
AD9874s
19 CLKP
20 CLKN
FS 31
DOUTA 29
TO OTHER
AD9874s
CLKOUT 28
fREF
TO
DSP
35
Figure 28. Example of Synchronizing Multiple AD9874s
–36–
REV. A
AD9874
VDDC
LOOP
FILTER
RBIAS
COSC
0.1F
RD
LOSC
RF
CP
CVAR
CZ
ATTENUATED PATH WITH
CLIP POINT = 7.0dBm
15
19 CLKP
13MHz
IOUTC
fREF 35
20 CLKN
FS 31
47 IFIN
DOUTA 29
CLKOUT 28
36dB
PAD
43 LOP
42 LON
AD9874
PE 25
MASTER
PD 24
PC 23
SYNCB 33
IOUTL
38
VCO
LOOP
FILTER
DUPLEXER PRESELECT
IF SAW 1
X
LNA
IF SAW 2
15
IOUTC
IF
AMP
PE 25
47 IFIN
PD 24
MIXER
PC 23
43 LOP
GAIN = –2dB
NF = 2dB
GAIN = 22dB
NF = 1dB
GAIN = –3dB
NF = 3dB
DSP
OR
ASIC
GAIN = 5dB GAIN = 15dB GAIN = –9dB
NF = 12dB NF = 2dB
NF = –9dB
SYNCB 33
42 LON
AD9874
DIRECT PATH WITH
CLIP POINT = –17dBm
SLAVE
19 CLKP
20 CLKN
FS 31
DOUTA 29
CLKOUT 28
fREF 35
Figure 29. Example of Split Path Rx Architecture to Increase Receiver Dynamic Range Capabilities
stage consists of two SAW filters isolated by a 15 dB gain stage.
The cascaded SAW filter response must provide sufficient
blocker rejection in order for the receiver to meet its sensitivity
requirements under worst-case blocker conditions. A composite
response having 27 dB, 60 dB, and 100 dB rejection at frequency
offsets of 0.8 MHz, 1.6 MHz, and 6.5 MHz, respectively,
provides enough blocker suppression to ensure that the AD9874
with the lower clip point will not be overdriven by any blocker.
This configuration results in the best possible receiver sensitivity
under all blocking conditions.
Split Path Rx Architecture
A split path Rx architecture may be attractive for those applications whose instantaneous dynamic range requirements exceed
the capability of a single AD9874 device. To cope with these
higher dynamic range requirements, two AD9874s can be operated in parallel with their respective clip points offset by a fixed
amount. Adding a fixed amount of attenuation in front of the
AD9874 and/or programming the attenuation setting of its
internal VGA can adjust the input-referred clip point. To save
power and simplify hardware, the LO and CLK circuits of the
device can also be shared. Connecting the SYNCB pins of the two
devices and pulsing this line low synchronizes the two devices.
An example of this concept for possible use in a GSM base station
is shown in Figure 29. The signal chain consists of a high linearity
RF front end and IF stage followed by two AD9874s operating in
parallel. The RF front end consists of a duplexer and preselect
filter to pass the GSM RF band of interest. A high performance
LNA isolates the duplexer from the preselect filter while providing
sufficient gain to minimize system NF. An RF mixer is used to
downconvert the entire GSM band to a suitable IF, where much of
the channel selectivity is accomplished. The 170.6 MHz IF is
chosen to avoid any self-induced spurs from the AD9874. The IF
REV. A
The output of the last SAW filters drives the two AD9874s via a
direct signal path and an attenuated signal path. The direct path
corresponds to the AD9874 having the lowest clip point and
provides the highest receiver sensitivity with a system noise
figure of 4.7 dB. The VGA of this device is set for maximum
attenuation, so its clip point is approximately –17 dBm. Since
conversion gain from the antenna to the AD9874 is 19 dB, the
digital output of this path will nominally be selected unless the
target signal’s power exceeds –36 dBm at the antenna. The
attenuated path corresponds to the AD9874 having the highest
input-referred clip point, and its digital output point of this path
is set to 7 dBm by inserting a 30 dB attenuator and setting the
AD9874’s VGA to the middle of its 12 dB range. This setting
–37–
AD9874
Since GSM is based on a TDMA scheme, digital data (or path)
selection can occur on a slot-by-slot basis. The AD9874 would
be configured to provide Serial I and Q data at a frame rate of
541.67 kSPS, as well as additional information including a 2-bit
reset field and a 6-bit RSSI field. These two fields contain the
information needed to decide whether the direct or attenuated
path should be used for the current time slot.
Hung Mixer Mode
The AD9874 can be operated in the hung mixer mode by tying
one of the LO’s self-biasing inputs to ground (i.e., GNDI) or
the positive supply (VDDI). In this mode, the AD9874 acts as a
narrow-band, band-pass - ADC, since its mixer passes the
IFIN signal without any frequency translation. The IFIN signal
must be centered about the resonant frequency of the - ADC
(i.e., fCLK/8) and the clock rate, fCLK, and decimation factors
must be selected to accommodate the bandwidth of the desired
input signal. Note that the LO synthesizer can be disabled
because it is no longer required.
Since the mixer does not have any losses associated with the
mixing operation, the conversion gain through the LNA and
mixer is higher resulting in a nominal input clip point of
–24 dBm. The linearity or IIP3 performance of the LNA and
mixer remains roughly unchanged and similar to that shown
in Figure 11b. The SNR performance is dependent of the
VGA attenuation setting, I/Q data resolution, and output
bandwidth as shown in Figure 30. Applications requiring the
highest instantaneous dynamic range should set the VGA for
maximum attenuation. Also, several extra decibels in SNR
performance can be gained at lower signal bandwidths by
using 24-bit I/Q data.
105
fCLK = 18MSPS
100
SNR – dB
MAX ATTEN w/
24-BIT I/Q DATA
LAYOUT EXAMPLE, EVALUATION BOARD, AND
SOFTWARE
The evaluation board and its accompanying software provide
a simple way to evaluate the AD9874. The block diagram in
Figure 31 shows the major blocks of the evaluation board,
which is designed to be flexible, allowing configuration for
different applications.
The power supply distribution block provides filtered, adjustable
voltages to the various supply pins of the AD9874. In the IF
input signal path, component pads are available to implement
different IF impedance matching networks. The LO and CLK
signals can be externally applied or internally derived from a
user-supplied VCO module interface daughter board. The reference for the on-chip LO and CLK synthesizers can be applied
via the external fREF input or an on-board crystal oscillator.
The evaluation board is designed to interface to a PC via a
National Instruments NI 6533 digital IO card. An XILINX
FPGA formats the data between the AD9874 and digital
I/O card.
IF
LO
INPUT INPUT
AD9874
MIXER
OUTPUT
VCO
MODULE
INTERFACE
FREF
INPUT
DUT
CRYSTAL
OSCILLATOR
(OPTIONAL)
XILINX
SPARTON
FPGA
POWER SUPPLY
DISTRIBUTION
CLK
INPUT
IDT
FIFO
(OPTIONAL)
NIDAQ 68-PIN
CONNECTOR
results in a 6 dB adjustment of the clip point, allowing the clip
point difference to be calibrated to exactly 24 dB, so that a
simple 5-bit shift would make up the gain difference. The
attenuated path can handle signal levels up to –12 dB at the
antenna before being overdriven. Since the SAW filters provide
sufficient blocker suppression, the digital data from this path
need only be selected when the target signal exceeds –36 dBm.
Although the sensitivity of the receiver with the attenuated path
is 20 dB lower than the direct path, the strong target signal
ensures a sufficiently high carrier-to-noise ratio.
EPROM
Figure 31. Evaluation Board Platform
Software developed using National Instruments’ LabVIEW™
(and provided as Microsoft® Windows® executable programs)
is supplied for the configuration of the SPI port registers and
evaluation of the AD9874 output data. These programs have
a convenient graphical user interface that allows for easy access
to the various SPI port configuration registers and real-time
frequency analysis of the output data.
For more information on the AD9874 evaluation board, including an example layout, please refer to the EVAL-AD9874EB
Data Sheet.
95
MAX ATTEN w/
16-BIT I/Q DATA
90
MIN ATTEN w/
16-BIT I/Q DATA
85
MIN ATTEN w/
24-BIT I/Q DATA
80
0
20
40
60
80
100
120
140
160
BW – kHz
Figure 30. Hung Mixer SNR vs. BW and VGA
–38–
REV. A
AD9874
OUTLINE DIMENSIONS
48-Lead Low Profile Quad Flat Package [LQFP]
(ST-48)
Dimensions shown in millimeters
1.60 MAX
0.75
0.60
0.45
PIN 1
INDICATOR
9.00 BSC
37
48
36
1
1.45
1.40
1.35
0.15
0.05
0.20
0.09
SEATING
PLANE
SEATING
PLANE
7ⴗ
3.5ⴗ
0ⴗ
0.08 MAX
COPLANARITY
VIEW A
25
12
13
0.50
BSC
VIEW A
ROTATED 90ⴗ CCW
COMPLIANT TO JEDEC STANDARDS MS-026BBC
REV. A
7.00
BSC
TOP VIEW
(PINS DOWN)
–39–
24
0.27
0.22
0.17
AD9874
Revision History
Location
Page
3/03—Data sheet changed from REV. 0 to REV. A
Changes to FUNCTIONAL BLOCK DIAGRAM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Changes to SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Changes to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Replaced Figure 1b . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Deleted Synchronization section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Added Synchronization Using SYNCB section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Changes to LO SYNTHESIZER section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Changes to Figure 7b . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Changes to Figure 7c . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Changes to Table X . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Changes to Automatic Gain Control section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Changes to Figure 29 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
Changes to LAYOUT EXAMPLE, EVALUATION BOARD, and SOFTWARE section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
Updated OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
–40–
REV. A
C02639–0–3/03(A)
Change to FEATURES . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1