AD AD8067ART-REEL High gain bandwidth product, precision fast fetâ ¢ op amp Datasheet

High Gain Bandwidth Product,
Precision Fast FET™ Op Amp
AD8067
FEATURES
CONNECTION DIAGRAM (TOP VIEW)
FET input amplifier: 0.6 pA input bias current
Stable for gains ≥8
High speed
54 MHz, −3 dB bandwidth (G = +10)
640 V/μs slew rate
Low noise
6.6 nV/√Hz
0.6 fA/√Hz
Low offset voltage (1.0 mV max)
Wide supply voltage range: 5 V to 24 V
No phase reversal
Low input capacitance
Single-supply and rail-to-rail output
Excellent distortion specs: SFDR 95 dBc @ 1 MHz
High common-mode rejection ratio: −106 dB
Low power: 6.5 mA typical supply current
Low cost
Small packaging: SOT-23-5
SOT-23-5 (RT-5)
5 +VS
VOUT 1
–VS 2
4 –IN
+IN 3
Figure 1.
APPLICATIONS
Photodiode preamplifiers
Precision high gain amplifiers
High gain, high bandwidth composite amplifiers
GENERAL DESCRIPTION
The AD8067 is designed to work in applications that require
high speed and low input bias current, such as fast photodiode
preamplifiers. As required by photodiode applications, the laser
trimmed AD8067 has excellent dc voltage offset (1.0 mV max)
and drift (15 μV/°C max).
The FET input bias current (5 pA max) and low voltage noise
(6.6 nV/√Hz) also contribute to making it appropriate for precision
applications. With a wide supply voltage range (5 V to 24 V)
and rail-to-rail output, the AD8067 is well suited for a variety of
applications that require wide dynamic range and low distortion.
The AD8067 amplifier is available in a SOT-23-5 package and is
rated to operate over the industrial temperature range of –40°C
to +85°C.
28
G = +20
26
24
22
G = +10
GAIN – dB
The AD8067 FastFET amp is a voltage feedback amplifier with
FET inputs offering wide bandwidth (54 MHz @ G = +10) and
high slew rate (640 V/μs). The AD8067 is fabricated in a
proprietary, dielectrically isolated eXtra Fast Complementary
Bipolar process (XFCB) that enables high speed, low power, and
high performance FET input amplifiers.
20
18
G = +8
16
14
12
10
8
0.1
1
10
FREQUENCY – MHz
100
Figure 2. Small Signal Frequency Response
Rev. A
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
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One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
©2006 Analog Devices, Inc. All rights reserved.
AD8067
TABLE OF CONTENTS
Features .............................................................................................. 1
Resistor Selection for Wideband Operation............................... 14
Applications....................................................................................... 1
DC Error Calculations............................................................... 15
General Description ......................................................................... 1
Input and Output Overload Behavior ..................................... 15
Connection Diagram (Top View)................................................... 1
Input Protection ......................................................................... 16
Revision History ............................................................................... 2
Capacitive Load Drive ............................................................... 16
Specifications for ±5 V..................................................................... 3
Layout, Grounding, and Bypassing Considerations .............. 16
Specifications for +5 V..................................................................... 4
Applications..................................................................................... 18
Specifications for ±12 V................................................................... 5
Wideband Photodiode Preamp ................................................ 18
Absolute Maximum Ratings............................................................ 6
Using the AD8067 at Gains of Less Than 8 ............................ 19
Maximum Power Dissipation ..................................................... 6
Single-Supply Operation ........................................................... 20
ESD Caution.................................................................................. 6
High Gain, High Bandwidth Composite Amplifier .............. 20
Typical Performance Characteristics ............................................. 7
Outline Dimensions ....................................................................... 22
Test Circuits..................................................................................... 12
Ordering Guide .......................................................................... 22
Theory of Operation ...................................................................... 13
Basic Frequency Response ........................................................ 13
REVISION HISTORY
5/06—Rev. 0 to Rev. A
Changes to Figure 51...................................................................... 18
Changes to Figure 54...................................................................... 19
Changes to Figure 57...................................................................... 21
Updated Outline Dimensions ....................................................... 22
Changes to Ordering Guide .......................................................... 22
11/02—Revision 0: Initial Version
Rev. A | Page 2 of 24
AD8067
SPECIFICATIONS FOR ±5 V
VS = ±5 V (@ TA = +25°C, G = +10, RF = RL =1 kΩ, unless otherwise noted.)
Table 1.
Parameter
DYNAMIC PERFORMANCE
−3 dB Bandwidth
Bandwidth for 0.1 dB Flatness
Output Overdrive Recovery Time (Pos/Neg)
Slew Rate
Settling Time to 0.1%
NOISE/DISTORTION PERFORMANCE
Spurious-Free Dynamic Range (SFDR)
Input Voltage Noise
Input Current Noise
DC PERFORMANCE
Input Offset Voltage
Input Offset Voltage Drift
Input Bias Current
Conditions
Min
Typ
VO = 0.2 V p-p
VO = 2 V p-p
VO = 0.2 V p-p
VI = ±0.6 V
VO = 5 V step
VO = 5 V step
39
54
54
8
115/190
640
27
MHz
MHz
MHz
ns
V/μs
ns
fC = 1 MHz, 2 V p-p
fC = 1 MHz, 8 V p-p
fC = 5 MHz, 2 V p-p
fC = 1 MHz, 2 V p-p, RL = 150 Ω
f = 10 kHz
f = 10 kHz
95
84
82
72
6.6
0.6
dBc
dBc
dBc
dBc
nV/√Hz
fA/√Hz
TMIN to TMAX
0.2
1
0.6
25
0.2
1
119
500
Input Offset Current
Open-Loop Gain
INPUT CHARACTERISTICS
Common-Mode Input Impedance
Differential Input Impedance
Input Common-Mode Voltage Range
Common-Mode Rejection Ratio (CMRR)
OUTPUT CHARACTERISTICS
Output Voltage Swing
Output Current
Short Circuit Current
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current
Power Supply Rejection Ratio (PSRR)
TMIN to TMAX
VO = ±3 V
103
Max
1.0
15
5
1
1000||1.5
1000||2.5
VCM = –1 V to +1 V
RL = 1 kΩ
RL = 150 Ω
SFDR > 60 dBc, f = 1 MHz
−5.0
−85
−4.86 to +4.83
30% overshoot
Rev. A | Page 3 of 24
mV
μV/°C
pA
pA
pA
pA
dB
−106
GΩ||pF
GΩ||pF
V
dB
−4.92 to +4.92
−4.67 to +4.72
30
105
120
V
V
mA
mA
pF
2.0
5
−90
Unit
6.5
−109
24
6.8
V
mA
dB
AD8067
SPECIFICATIONS FOR +5 V
VS = +5 V (@ TA = +25°C, G = +10, RL = RF = 1 kΩ, unless otherwise noted.)
Table 2.
Parameter
DYNAMIC PERFORMANCE
−3 dB Bandwidth
Bandwidth for 0.1 dB Flatness
Output Overdrive Recovery Time (Pos/Neg)
Slew Rate
Settling Time to 0.1%
NOISE/DISTORTION PERFORMANCE
Spurious-Free Dynamic Range (SFDR)
Input Voltage Noise
Input Current Noise
DC PERFORMANCE
Input Offset Voltage
Input Offset Voltage Drift
Input Bias Current
Conditions
Min
Typ
VO = 0.2 V p-p
VO = 2 V p-p
VO = 0.2 V p-p
VI = +0.6 V
VO = 3 V step
VO = 2 V step
36
54
54
8
150/200
490
25
MHz
MHz
MHz
ns
V/μs
ns
86
74
60
72
6.6
0.6
dBc
dBc
dBc
dBc
nV/√Hz
fA/√Hz
390
fC = 1 MHz, 2 V p-p
fC = 1 MHz, 4 V p-p
fC = 5 MHz, 2 V p-p
fC = 1 MHz, 2 V p-p, RL = 150 Ω
f = 10 kHz
f = 10 kHz
TMIN to TMAX
Input Offset Current
Open-Loop Gain
INPUT CHARACTERISTICS
Common-Mode Input Impedance
Differential Input Impedance
Input Common-Mode Voltage Range
Common-Mode Rejection Ratio (CMRR)
OUTPUT CHARACTERISTICS
Output Voltage Swing
Output Current
Short Circuit Current
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current
Power Supply Rejection Ratio (PSRR)
VO = 0.5 V to 4.5 V
100
0.2
1
0.5
25
0.1
117
Max
1.0
15
5
1
1000||2.3
1000||2.5
VCM = 0.5 V to 1.5 V
RL = 1 kΩ
RL =150 Ω
SFDR > 60 dBc, f = 1 MHz
0
−81
0.07 to 4.89
30% overshoot
Rev. A | Page 4 of 24
mV
μV/°C
pA
pA
pA
dB
−98
GΩ||pF
GΩ||pF
V
dB
0.03 to 4.94
0.08 to 4.83
22
95
120
V
V
mA
mA
pF
2.0
5
−87
Unit
6.4
−103
24
6.7
V
mA
dB
AD8067
SPECIFICATIONS FOR ±12 V
VS = ±12 V (@ TA = +25°C, G = +10, RL = RF = 1 kΩ, unless otherwise noted.)
Table 3.
Parameter
DYNAMIC PERFORMANCE
–3 dB Bandwidth
Bandwidth for 0.1 dB Flatness
Output Overdrive Recovery Time (Pos/Neg)
Slew Rate
Settling Time to 0.1%
NOISE/DISTORTION PERFORMANCE
Spurious-Free Dynamic Range (SFDR)
Input Voltage Noise
Input Current Noise
DC PERFORMANCE
Input Offset Voltage
Input Offset Voltage Drift
Input Bias Current
Conditions
Min
Typ
VO = 0.2 V p-p
VO = 2 V p-p
VO = 0.2 V p-p
VI = ±1.5 V
VO = 5 V step
VO = 5 V step
39
54
53
8
75/180
640
27
MHz
MHz
MHz
ns
V/μs
ns
92
84
74
72
6.6
0.6
dBc
dBc
dBc
dBc
nV/√Hz
fA/√Hz
500
fC = 1 MHz, 2 V p-p
fC = 1 MHz, 20 V p-p
fC = 5 MHz, 2 V p-p
fC = 1 MHz, 2 V p-p, RL = 150 Ω
f = 10 kHz
f = 10 kHz
TMIN to TMAX
Input Offset Current
Open-Loop Gain
INPUT CHARACTERISTICS
Common-Mode Input Impedance
Differential Input Impedance
Input Common-Mode Voltage Range
Common-Mode Rejection Ratio (CMRR)
OUTPUT CHARACTERISTICS
Output Voltage Swing
Output Current
Short Circuit Current
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current
Power Supply Rejection Ratio (PSRR)
VO = ±10 V
107
0.2
1
1.0
25
0.2
119
Max
1.0
15
5
1
1000||1.5
1000||2.5
VCM = –1 V to +1 V
RL = 1 kΩ
RL = 500 Ω
SFDR > 60 dBc, f = 1 MHz
−12.0
−89
−11.70 to +11.70
30% overshoot
Rev. A | Page 5 of 24
mV
μV/°C
pA
pA
pA
dB
−108
GΩ||pF
GΩ||pF
V
dB
−11.85 to +11.84
−11.31 to +11.73
26
125
120
V
V
mA
mA
pF
+9.0
5
−86
Unit
6.6
−97
24
7.0
V
mA
dB
AD8067
ABSOLUTE MAXIMUM RATINGS
PD = Quiescent Power + (Total Drive Power − Load Power)
Table 4.
Rating
26.4 V
See Figure 3
VEE – 0.5 V to VCC + 0.5 V
1.8 V
–65°C to +125°C
–40°C to +85°C
300°C
150°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
MAXIMUM POWER DISSIPATION
The associated raise in junction temperature (TJ) on the die
limits the maximum safe power dissipation in the AD8067
package. At approximately 150°C, which is the glass transition
temperature, the plastic changes its properties. Even temporarily
exceeding this temperature limit can change the stresses that the
package exerts on the die, permanently shifting the parametric
performance of the AD8067. Exceeding a junction temperature
of 175°C for an extended period can result in changes in the
silicon devices, potentially causing failure.
⎛V V
PD = (VS × I S )+ ⎜⎜ S × OUT
RL
⎝ 2
⎞ VOUT 2
⎟–
⎟ R
L
⎠
If RL is referenced to VS− as in single-supply operation, then the
total drive power is VS × IOUT.
If the rms signal levels are indeterminate, then consider the
worst case, when VOUT = VS/4 for RL to midsupply:
PD = (VS × I S ) +
(VS /4 )2
RL
In single-supply operation with RL referenced to VS−, worst case
is VOUT = VS/2.
Airflow increases heat dissipation effectively, reducing θJA. In
addition, more metal directly in contact with the package leads
from metal traces, through holes, ground, and power planes
reduces the θJA.
Figure 3 shows the maximum safe power dissipation in the
package vs. the ambient temperature for the SOT-23-5
(180°C/W) package on a JEDEC standard 4-layer board. θJA
values are approximations.
It should be noted that for every 10°C rise in temperature, IB
approximately doubles (see Figure 22).
B
2.0
MAXIMUM POWER DISSIPATION – W
Parameter
Supply Voltage
Power Dissipation
Common-Mode Input Voltage
Differential Input Voltage
Storage Temperature Range
Operating Temperature Range
Lead Temperature (Soldering 10 sec)
Junction Temperature
The power dissipated in the package (PD) is the sum of the
quiescent power dissipation and the power dissipated in the
package due to the load drive. The quiescent power is the
voltage between the supply pins (VS) times the quiescent
current (IS). Assuming the load (RL) is referenced to midsupply,
the total drive power is VS/2 × IOUT, some of which is dissipated
in the package and some in the load (VOUT × IOUT). The
difference between the total drive power and the load power is
the drive power dissipated in the package. RMS output voltages
should be considered.
1.5
1.0
SOT-23-5
0.5
0
–40 –30 –20 –10
0
10
20
30
40
50
60
70
80
AMBIENT TEMPERATURE – °C
Figure 3. Maximum Power Dissipation vs. Temperature for a 4-Layer Board
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. A | Page 6 of 24
AD8067
TYPICAL PERFORMANCE CHARACTERISTICS
Default Conditions: VS = ±5 V (@ TA = +25°C, G = +10, RL = RF = 1 kΩ, unless otherwise noted.)
20.7
28
VOUT = 0.2V p-p
VOUT = 200mV p-p
G = +20
26
20.6
24
20.5
22
20.4
VOUT = 1.4V p-p
GAIN – dB
G = +10
20
GAIN – dB
VOUT = 0.7V p-p
G = +8
18
G = +6
16
20.3
20.2
20.1
14
20.0
12
19.9
10
19.8
8
1
10
FREQUENCY – MHz
1
100
Figure 4. Small Signal Frequency Response for Various Gains
100
Figure 7. 0.1 dB Flatness Frequency Response
24
22
VOUT = 200mV p-p
CL = 100pF
VOUT = 200mV p-p
VS = +5V
21
23
VS = ±5V
22
20
CL = 25pF
21
GAIN – dB
VS = ±12V
19
GAIN – dB
10
FREQUENCY – MHz
18
17
20
CL = 100pF
RSNUB = 24.9Ω
19
18
17
16
16
CL = 5pF
15
15
14
1
14
1
10
FREQUENCY – MHz
100
Figure 5. Small Signal Frequency Response for Various Supplies
100
Figure 8. Small Signal Frequency Response for Various CLOAD
22
22
VOUT = 2V p-p
VS = +5V
21
21
VS = ±5V
20
VOUT = 0.2V p-p, 2V p-p
20
VOUT = 4V p-p
VS = ±12V
19
GAIN – dB
GAIN – dB
10
FREQUENCY – MHz
18
19
18
17
17
16
16
15
15
14
1
10
FREQUENCY – MHz
14
100
1
Figure 6. Large Signal Frequency Response for Various Supplies
10
FREQUENCY – MHz
100
Figure 9. Frequency Response for Various Output Amplitudes
Rev. A | Page 7 of 24
AD8067
22
VOUT = 200mV p-p
RF = 2kΩ
21
90
120
80
90
RF = 1kΩ
70
60
20
GAIN – dB
GAIN – dB
RF = 499Ω
19
18
17
60
30
50
0
40
–30
GAIN
30
–60
20
–90
16
10
–120
15
0
–150
14
–10
0.01
1
10
FREQUENCY – MHz
100
0.1
Figure 10. Small Signal Frequency Response for Various RF
–180
1000
100
Figure 13. Open-Loop Gain and Phase
–40
–40
HD2 RLOAD = 150Ω
–50
G = +10
VOUT = 2V p-p
–50
–60
–60
–70
DISTORTION – dBc
DISTORTION – dBc
1
10
FREQUENCY – MHz
HD3 RLOAD = 150Ω
–80
HD2
RLOAD = 1kΩ
–90
–100
–110
–70
HD2 VS = ±12V
–80
–90
HD2 VS = ±5V
–100
–110
VOUT = 2V p-p
HD3 RLOAD = 1kΩ
–120
HD3 VS = ±12V
–120
G = +10
VS = ±5V
–130
–140
0.1
1
10
FREQUENCY – MHz
–130
–140
0.1
100
Figure 11. Distortion vs. Frequency for Various Loads
–20
HD3 VS = ±5V
1
10
FREQUENCY – MHz
100
Figure 14. Distortion vs. Frequency for Various Supplies
–30
VS = ±12V
G = +10
VS = ±12V
f = 1MHz
G = +10
–40
–40
–50
DISTORTION – dBc
DISTORTION – dBc
HD2 RLOAD = 150Ω
–60
–80
HD2 VOUT = 20V p-p
HD3 VOUT = 2V p-p
–100
–70
HD3 RLOAD = 150Ω
–80
–90
HD2 RLOAD = 1kΩ
–100
HD2 VOUT = 2V p-p
HD3 VOUT = 20V p-p
–120
–60
–110
HD3 RLOAD = 1kΩ
–120
–140
0.1
1
10
FREQUENCY – MHz
–130
100
0
Figure 12. Distortion vs. Frequency for Various Amplitudes
2
4
6
8
10 12 14 16 18
OUTPUT AMPLITUDE – V p-p
20
22
24
Figure 15. Distortion vs. Output Amplitude for Various Loads
Rev. A | Page 8 of 24
PHASE – Degrees
PHASE
AD8067
G = +10
VIN = 20mV p-p
G = +10
VIN = 20mV p-p
CL = 100pF
CL = 0pF
1.5V
50mV/DIV
50mV/DIV
25ns/DIV
Figure 16. Small Signal Transient Response 5 V Supply
10VIN
2V/DIV
Figure 19. Small Signal Transient Response ± 5 V Supply
VS = ±12V
VIN = 2V p-p
G = +10
G = +10
VOUT
25ns/DIV
200ns/DIV
5V/DIV
Figure 17. Output Overdrive Recovery
50ns/DIV
Figure 20. Large Signal Transient Response
VOUT (1V/DIV)
G = +10
VIN (100mV/DIV)
VOUT – 10VIN (5mV/DIV)
+0.1%
+0.1%
VIN (100mV/DIV)
VOUT – 10VIN (5mV/DIV)
–0.1%
–0.1%
t=0
5μs/DIV
Figure 18. Long-Term Settling Time
5ns/DIV
Figure 21. 0.1% Short-Term Settling Time
Rev. A | Page 9 of 24
AD8067
14
10
8
VS = ±12V
INPUT BIAS CURRENT – pA
INPUT BIAS CURRENT – pA
12
10
8
6
VS = ±12V
4
VS = ±5V
VS = +5V
6
4
2
0
–2
–4
–6
2
–8
VS = ±5V
0
25
35
45
55
65
TEMPERATURE – °C
75
–10
–14 –12 –10 –8 –6 –4 –2 0 2 4 6 8
COMMON-MODE VOLTAGE – V
85
Figure 25. Input Bias Current vs. Common-Mode Voltage
Figure 22. Input Bias Current vs. Temperature
1800
5
N = 12255
SD = 0.203
MEAN = –0.033
4
INPUT OFFSET VOLTAGE – mV
1600
1400
1200
COUNT
10 12 14
1000
800
600
400
200
VS = ±12V
3
VS = ±5V
2
1
VS = +5V
0
–1
–2
–6
–4
–5
–14 –12 –10 –8 –6 –4 –2 0 2 4 6 8
COMMON-MODE VOLTAGE – V
0
–1
0
INPUT OFFSET VOLTAGE – mV
1
Figure 23. Input Offset Voltage Histogram
10 12 14
Figure 26. Input Offset Voltage vs. Common-Mode Voltage
1000
–40
–50
CMRR – dB
NOISE – nV/ Hz
–60
100
10
–70
–80
–90
–100
–110
1
1
10
100
1k
10k
100k
FREQUENCY – Hz
1M
10M
100M
Figure 24. Voltage Noise
–120
0.1
1
10
FREQUENCY – MHz
Figure 27. CMRR vs. Frequency
Rev. A | Page 10 of 24
100
AD8067
6.7
100
VS = ±12V
G = +10
QUIESCENT CURRENT – mA
6.6
OUTPUT IMPEDANCE – Ω
10
1
0.1
VS = ±5V
6.5
VS = +5V
6.4
6.3
6.2
6.1
0.01
6.0
–40
0.001
0.01
0.1
1
10
FREQUENCY – MHz
100
–20
0
1000
20
40
TEMPERATURE – °C
60
80
Figure 31. Quiescent Current vs. Temperature for Various Supply Voltages
Figure 28. Output Impedance vs. Frequency
0.30
200
OUTPUT SATURATION VOLTAGE – mV
OUTPUT SATURATION VOLTAGE – V
RL = 1kΩ
0.25
VCC – VOH
0.20
VOL – VEE
0.15
0.10
0.05
180
(VCC – VOH), (VOL – VEE), VS = ±12V
160
140
120
100
(VCC – VOH), (VOL – VEE), VS = ±5V
80
VCC – VOH, VS = +5V
60
40
VOL – VEE, VS = +5V
20
0
0
5
10
15
20
25
ILOAD – mA
30
35
40
0
–40
Figure 29. Output Saturation Voltage vs. Output Load Current
–20
0
20
40
TEMPERATURE – °C
60
80
Figure 32. Output Saturation Voltage vs. Temperature
0
140
–10
130
–20
120
–PSRR
OPEN-LOOP GAIN – dB
PSRR – dB
–30
–40
–50
–60
+PSRR
–70
–80
100
90
VS = ±5V
80
VS = +5V
70
–90
–100
0.01
VS = ±12V
110
60
0.1
1
FREQUENCY – MHz
10
100
50
0
5
10
15
20
25
ILOAD – mA
30
35
40
Figure 33. Open-Loop Gain vs. Load Current for Various Supplies
Figure 30. PSRR vs. Frequency
Rev. A | Page 11 of 24
AD8067
TEST CIRCUITS
+VCC
10μF
10μF
+
0.1μF
+
0.1μF
110Ω
1kΩ
110Ω
AD8067
49.9Ω
1kΩ
VIN
5
4
VIN
+VCC
VOUT
1
2
VOUT
AD8067
110Ω
1
3
RL = 1kΩ
3
5
4
1kΩ
2
0.1μF
0.1μF
1kΩ
10μF
10μF
+
+
AV = 10
–VEE
–VEE
Figure 34. Standard Test Circuit
Figure 37. CMRR Test Circuit
+VCC
VIN
10μF
110Ω
1kΩ
+VCC
+
0.1μF
110Ω
V–
1kΩ
5
4
AD8067
5
4
3
AD8067
100Ω
3
VOUT
1
1kΩ
2
VOUT
1
0.1μF
100Ω
1kΩ
2
10μF
0.1μF
+
10μF
VOUT
AOL =
V–
+
–VEE
–VEE
Figure 35. Open-Loop Gain Test Circuit
Figure 38. Positive PSRR Test Circuit
+VCC
+VCC
10μF
10μF
+
0.1μF
+
0.1μF
110Ω
110Ω
4
VIN
1kΩ
1kΩ
AD8067
49.9Ω
4
5
RSNUB
VOUT
1
3
CLOAD
2
5
AD8067
100Ω
VOUT
1
NETWORK ANALYZER
3
2
1kΩ
0.1μF
0.1μF
10μF
10μF
+
+
AV = 10
–VEE
–VEE
Figure 36. Test Circuit for Capacitive Load
Figure 39. Output Impedance Test Circuit
Rev. A | Page 12 of 24
AD8067
THEORY OF OPERATION
The combination of low noise, dc precision, and high
bandwidth makes the AD8067 uniquely suited for wideband,
very high input impedance, high gain buffer applications. It is
also useful in wideband transimpedance applications, such as a
photodiode interface, that require very low input currents and
dc precision.
BASIC FREQUENCY RESPONSE
The AD8067’s typical open-loop response (see Figure 41) shows
a phase margin of 60° at a gain of +10. Typical configurations
for noninverting and inverting voltage gain applications are
shown in Figure 40 and Figure 42.
The closed-loop frequency response of a basic noninverting
gain configuration can be approximated by:
Closed Loop–3 dB Frequency = (GBP ) ×
RG
(RF + RG )
DC Gain = RF/RG + 1
GBP is the gain bandwidth product of the amplifier. Typical
GBP for the AD8067 is 300 MHz. See Table 5 for the
recommended values for RG and RF.
Noninverting Configuration Noise Gain =
90
120
80
90
70
60
60
30
50
0
40
–30
GAIN
30
–60
20
–90
10
–120
0
–150
–10
0.01
0.1
1
10
FREQUENCY – MHz
100
–180
1000
Figure 41. Open-Loop Frequency Response
The bandwidth formula only holds true when the phase margin
of the application approaches 90°, which it will in high gain
configurations. The bandwidth of the AD8067 used in a
G = +10 buffer is 54 MHz, considerably faster than the 30 MHz
predicted by the closed loop –3 dB frequency equation. This
extended bandwidth is due to the phase margin being at 60°
instead of 90°. Gains lower than +10 show an increased amount
of peaking, as shown in Figure 4. For gains lower than +7, use
the AD8065, a unity gain stable JFET input op amp with a unity
gain bandwidth of 145 MHz, or refer to the Applications section
for using the AD8067 in a gain of 2 configuration.
Table 5. Recommended Values of RG and RF
Gain
10
20
50
100
RG (Ω)
110
49.9
20
10
RF
+1
RG
RF (kΩ)
1
1
1
1
+VS
0.1μF
BW (MHz)
54
15
6
3
+
10μF
RX
+
+VS
0.1μF
RS
AD8067
+
10μF
RLOAD
–
RX
+
0.1μF
AD8067
VI
RLOAD
–
0.1μF
SIGNAL
SOURCE
10μF
+
RS
+
VOUT
–
RG
10μF
+
+
VOUT
–
–VS
RF
VI
–VS
RF
RG
PHASE – Degrees
PHASE
GAIN – dB
The AD8067 is a low noise, wideband, voltage feedback
operational amplifier that combines a precision JFET input
stage with Analog Devices’ dielectrically isolated eXtra Fast
Complementary Bipolar (XFCB) process BJTs. Operating
supply voltages range from 5 V to 24 V. The amplifier features a
patented rail-to-rail output stage capable of driving within
0.25 V of either power supply while sourcing or sinking 30 mA.
The JFET input, composed of N-channel devices, has a
common-mode input range that includes the negative supply
rail and extends to 3 V below the positive supply. In addition,
the potential for phase reversal behavior was eliminated for all
input voltages within the power supplies.
SIGNAL
SOURCE
FOR BEST PERFORMANCE,
SET RS + RX = RG || RF
FOR BEST PERFORMANCE, SET RX = (RS + RG) || RF
Figure 40. Noninverting Gain Configuration
Figure 42. Inverting Gain Configuration
Rev. A | Page 13 of 24
AD8067
For inverting voltage gain applications, the source impedance of
the input signal must be considered because it sets the application’s
noise gain as well as the apparent closed-loop gain. The basic
frequency equation for inverting applications is
RG + RS
Closed-Loop –3 dB Frequency = (GBP ) ×
R F + R G + RS
+
CM
CD
CM
–
–
+
VOUT
–
RF
SIGNAL SOURCE
CPAR
RG
where GBP is the gain bandwidth product of the amplifier, and
RS is the signal source resistance.
RF + RG + RS
RG + RS
It is important that the noise gain for inverting applications be
kept above 6 for stability reasons. If the signal source driving
the inverter is another amplifier, take care that the driving
amplifier shows low output impedance through the frequency
span of the expected closed-loop bandwidth of the AD8067.
RESISTOR SELECTION FOR WIDEBAND OPERATION
Voltage feedback amplifiers can use a wide range of resistor
values to set their gain. Proper design of the application’s
feedback network requires consideration of the following issues:
•
Poles formed by the amplifier’s input capacitances with the
resistances seen at the amplifier’s input terminals
•
Effects of mismatched source impedances
•
Resistor value impact on the application’s output
voltage noise
•
CPAR
VI
RF
DC Gain = –
RG + RS
Inverting Configuration Noise Gain =
+
RS
Amplifier loading effects
The AD8067 has common-mode input capacitances (CM) of
1.5 pF and a differential input capacitance (CD) of 2.5 pF. This is
illustrated in Figure 43. The source impedance driving the
positive input of a noninverting buffer forms a pole primarily
with the amplifier’s common-mode input capacitance as well as
any parasitic capacitance due to the board layout (CPAR). This
limits the obtainable bandwidth. For G = +10 buffers, this
bandwidth limit becomes apparent for source impedances >1 kΩ.
Figure 43. Input and Board Capacitances
There is a pole in the feedback loop response formed by
the source impedance seen by the amplifier’s negative input
(RG ⎢⎢RF) and the sum of the amplifier’s differential input
capacitance, common-mode input capacitance, and any board
parasitic capacitance. This decreases the loop phase margin and
can cause stability problems, that is, unacceptable peaking and
ringing in the response. To avoid this problem, it is recommended
that the resistance at the AD8067’s negative input be kept below
200 Ω for all wideband voltage gain applications.
Matching the impedances at the inputs of the AD8067 is also
recommended for wideband voltage gain applications. This
minimizes nonlinear common-mode capacitive effects that can
significantly degrade settling time and distortion performance.
The AD8067 has a low input voltage noise of 6.6 nV/√Hz.
Source resistances greater than 500 Ω at either input terminal
notably increases the apparent referred-to-input (RTI) voltage
noise of the application.
The amplifier must supply output current to its feedback
network, as well as to the identified load. For instance, the
load resistance presented to the amplifier in Figure 40 is
RLOAD ⎪⎪ (RF + RG). For an RLOAD of 100 Ω, RF of 1 kΩ, and RG of
100 Ω, the amplifier is driving a total load resistance of about
92 Ω. This becomes more of an issue as RF decreases. The
AD8067 is rated to provide 30 mA of low distortion output
current. Heavy output drive requirements also increase the
part’s power dissipation and should be taken into account.
Rev. A | Page 14 of 24
AD8067
DC ERROR CALCULATIONS
INPUT AND OUTPUT OVERLOAD BEHAVIOR
Figure 44 illustrates the primary dc errors associated with a
voltage feedback amplifier. For both inverting and noninverting
configurations:
A simplified schematic of the AD8067 input stage is shown in
Figure 45. This shows the cascoded N-channel JFET input pair,
the ESD and other protection diodes, and the auxiliary NPN
input stage that eliminates phase inversion behavior.
⎛ R + RF
Output Voltage Error due to VOS = VOS ⎜⎜ G
⎝ RG
⎞
⎟
⎟
⎠
⎛ R + RG
Output Voltage Error due to I B = I B + × RS ⎜⎜ F
⎝ RG
⎞
⎟ – I B– × RF
⎟
⎠
Total error is the sum of the two.
DC common-mode and power supply effects can be added by
modeling the total VOS with the expression:
VOS (tot ) = VOS (nom) +
ΔVS ΔVCM
+
PSR CMR
where:
VOS (nom) is the offset voltage specified at nominal conditions
(1 mV max).
When the common-mode input voltage to the amplifier is
driven to within approximately 3 V of the positive power
supply, the input JFET’s bias current turns off, and the bias of
the NPN pair turns on, taking over control of the amplifier. The
NPN differential pair now sets the amplifier’s offset, and the
input bias current is now in the range of several tens of
microamps. This behavior is illustrated in Figure 25 and Figure 26.
Normal operation resumes when the common-mode voltage
goes below the 3 V from the positive supply threshold.
The output transistors have circuitry included to limit the
extent of their saturation when the output is overdriven. This
improves output recovery time. A plot of the output recovery
time for the AD8067 used as a G = +10 buffer is shown in
Figure 17.
ΔVS is the change in power supply voltage from nominal
conditions.
VCC
TO REST OF AMP
VTHRESHOLD
PSR is power supply rejection (90 dB minimum).
ΔVCM is the change in common-mode voltage from nominal test
conditions.
SWITCH
CONTROL
VCC
VCC
VN
VP
CMR is the common-mode rejection (85 dB minimum for the
AD8067).
RF
VEE
VEE
+VOS–
RG
–
+ VOUT –
IB–
– VI +
RS
VEE
Figure 45. Simplified Input Schematic
+
IB+
Figure 44. Op Amp DC Error Sources
Rev. A | Page 15 of 24
VBIAS
AD8067
INPUT PROTECTION
The inputs of the AD8067 are protected with back-to-back
diodes between the input terminals as well as ESD diodes to
either power supply. The result is an input stage with picoamp
level input currents that can withstand 2 kV ESD events
(human body model) with no degradation.
Excessive power dissipation through the protection devices
destroys or degrades the performance of the amplifier.
Differential voltages greater than 0.7 V result in an input
current of approximately (| V+ – V− | − 0.7 V)/(RI + RG)),
where RI and RG are the resistors (see Figure 46). For input
voltages beyond the positive supply, the input current is about
(VI – VCC – 0.7 V)/RI. For input voltages beyond the negative
supply, the input current is about (VI – VEE + 0.7 V)/RI. For any
of these conditions, RI should be sized to limit the resulting
input current to 50 mA or less.
–
VI
+ RI
AD8067
RI > ( |V+ – V– | –0.7V)/50mA
FOR LARGE |V+ – V– |
RG
RF
RI > (VI – VEE + 0.7V)/50mA
RI > (VI – VCC – 0.7V)/50mA
FOR VI BEYOND
+ SUPPLY VOLTAGES
VOUT
–
Figure 46. Current Limiting Resistor
CAPACITIVE LOAD DRIVE
Capacitive load introduces a pole in the amplifier loop response
due to the finite output impedance of the amplifier. This can
cause excessive peaking and ringing in the response. The
AD8067 with a gain of +10 handles up to a 30 pF capacitive
load without an excessive amount of peaking (see Figure 8). If
greater capacitive load drive is required, consider inserting a
small resistor in series with the load (24.9 Ω is a good value to
start with). Capacitive load drive capability also increases as the
gain of the amplifier increases.
LAYOUT, GROUNDING, AND BYPASSING
CONSIDERATIONS
Layout
In extremely low input bias current amplifier applications, stray
leakage current paths must be kept to a minimum. Any voltage
differential between the amplifier inputs and nearby traces sets
up a leakage path through the PCB. Consider a 1 V signal and
100 GΩ to ground present at the input of the amplifier. The
resultant leakage current is 10 pA; this is 10× the input bias
current of the amplifier. Poor PCB layout, contamination, and
the board material can create large leakage currents. Common
contaminants on boards are skin oils, moisture, solder flux, and
cleaning agents. Therefore, it is imperative that the board be
thoroughly cleaned and the board surface be free of contaminants
to fully take advantage of the AD8067’s low input bias currents.
To significantly reduce leakage paths, a guard-ring/shield
around the inputs should be used. The guard-ring circles the
input pins and is driven to the same potential as the input
signal, thereby reducing the potential difference between pins.
For the guard ring to be completely effective, it must be driven
by a relatively low impedance source and should completely
surround the input leads on all sides, above, and below, using a
multilayer board (see Figure 47). The SOT-23-5 package
presents a challenge in keeping the leakage paths to a minimum.
The pin spacing is very tight, so extra care must be used when
constructing the guard ring (see Figure 48 for recommended
guard-ring construction).
GUARD RING
GUARD RING
NONINVERTING
INVERTING
Figure 47. Guard-Ring Configurations
VOUT
+V
VOUT
AD8067
AD8067
–V
–V
+IN
–IN
INVERTING
+IN
–IN
NONINVERTING
Figure 48. Guard-Ring Layout SOT-23-5
Rev. A | Page 16 of 24
+V
AD8067
Grounding
Power Supply Bypassing
To minimize parasitic inductances and ground loops in high
speed, densely populated boards, a ground plane layer is critical.
Understanding where the current flows in a circuit is critical in
the implementation of high speed circuit design. The length of
the current path is directly proportional to the magnitude of the
parasitic inductances and thus the high frequency impedance of
the path. Fast current changes in an inductive ground return
creates unwanted noise and ringing.
Power supply pins are actually inputs and care must be taken to
provide a clean, low noise dc voltage source to these inputs. The
bypass capacitors have two functions:
The length of the high frequency bypass capacitor leads is
critical. A parasitic inductance in the bypass grounding works
against the low impedance created by the bypass capacitor.
Because load currents flow from supplies as well as ground, the
load should be placed at the same physical location as the
bypass capacitor ground. For large values of capacitors, which
are intended to be effective at lower frequencies, the current
return path length is less critical.
• Provide a low impedance path for unwanted frequencies
from the supply inputs to ground, thereby reducing the effect
of noise on the supply lines
• Provide localized charge storage—this is usually
accomplished with larger electrolytic capacitors
Decoupling methods are designed to minimize the bypassing
impedance at all frequencies. This can be accomplished with a
combination of capacitors in parallel to ground. Good quality
ceramic chip capacitors (X7R or NPO) should be used and
always kept as close to the amplifier package as possible. A
parallel combination of a 0.1 μF ceramic and a 10 μF electrolytic,
covers a wide range of rejection for unwanted noise. The 10 μF
capacitor is less critical for high frequency bypassing, and in
most cases, one per supply line is sufficient.
Rev. A | Page 17 of 24
AD8067
APPLICATIONS
WIDEBAND PHOTODIODE PREAMP
The preamp’s output noise over frequency is shown in Figure 50.
CF
Table 6. RMS Noise Contributions of Photodiode Preamp
RMS
Noise
(μV)1
152
RF
–
CM
CS
IPHOTO
VOUT
CD
RSH = 1011Ω
+
VB
CM
Contributor
RF × 2
Expression
Amp to f1
VNOISE × f 1
Amp (f2 − f1)
AD8067
CF + CS
Amp (Past f2)
RF
1
Figure 49 shows an I/V converter with an electrical model of a
photodiode.
CF
(C S + C M + C F + 2C D ) ×
CF
708
1
I PHOTO × RF
The stable bandwidth attainable with this preamp is a function
of RF, the gain bandwidth product of the amplifier, and the total
capacitance at the amplifier’s summing junction, including CS
and the amplifier input capacitance. RF and the total capacitance
produce a pole in the amplifier’s loop transmission that can
result in peaking and instability. Adding CF creates a zero in the
loop transmission that compensates for the pole’s effect and
reduces the signal bandwidth. It can be shown that the signal
bandwidth resulting in a 45° phase margin (f(45)) is defined by
GBP
2π × RF × C S
GBP is the unit gain bandwidth product, RF is the feedback
resistance, and CS is the total capacitance at the amplifier
summing junction (amplifier + photodiode + board parasitics).
VOLTAGE NOISE – nV/ Hz
f2 =
1 + sC F RF
1
2 πR F C F
GBP
f3 = (C + C + 2C + C )/C
S
M
D
F
F
RF NOISE
f2
f3
VEN (C F + C S + C M + 2C D )/C F
f1
VEN
NOISE DUE TO AMPLIFIER
FREQUENCY – Hz
Figure 50. Photodiode Voltage Noise Contributions
Figure 51 shows the AD8067 configured as a transimpedance
photodiode amplifier. The amplifier is used in conjunction with
a JDS uniphase photodiode detector. This amplifier has a
bandwidth of 9.6 MHz, as shown in Figure 52, and is verified by
the design equations shown in Figure 50.
0.33pF
The value of CF that produces f(45) can be shown to be
49.9kΩ
+5V
CS
2π × R F × GBP
10μF
0.1μF
–5V
The frequency response in this case shows about 2 dB of
peaking and 15% overshoot. Doubling CF and cutting the
bandwidth in half results in a flat frequency response, with
about 5% transient overshoot.
684
f 3 × 1.57
f1 = 2 πR (C + C + C + 2C )
F F
S
M
D
where IPHOTO is the output current of the photodiode, and the
parallel combination of RF and CF sets the signal bandwidth.
CF =
96
f 2 – f1
RMS noise with RF = 50 kΩ, CS = 0.67 pF, CF = 0.33 pF, CM = 1.5 pF, and CD = 2.5 pF.
The basic transfer function is
f ( 45 ) =
(C S + C M + C F + 2C D ) ×
VNOISE ×
VNOISE ×
4.3
RSS Total
Figure 49. Wideband Photodiode Preamp
VOUT =
2 × 4kT × R F × f 2 × 1.57
EPM 605 LL
50Ω
AD8067
0.1μF
0.33pF
NOTES
ID @ –5V = 0.074nA
CD @ –5V = 0.690pF
RB @ 1550nm = –49dB
49.9kΩ
10μF
–5V
Figure 51. Photodiode Preamplifier
Rev. A | Page 18 of 24
VOUT
AD8067
Test data for the preamp is shown in Figure 52 and Figure 53.
A common technique used to stabilize de-compensated
amplifiers is to increase the noise gain, independent of the
signal gain. The AD8067 can be used in applications where the
signal gain is less than 8, if proper care is taken to ensure that
the noise gain of the amplifier is set to at least the recommended
minimum signal gain of 8 (see Figure 54).
100
95
TRANSIMPEDANCE GAIN – dB
USING THE AD8067 AT GAINS OF LESS THAN 8
90
85
80
The signal and noise gain equations for a noninverting
amplifier are:
75
Signal Gain = 1 +
70
65
Noise Gain = 1 +
60
0.01
0.1
1
FREQUENCY – MHz
10
R3
R1
R3
R1
100
Figure 52. Photodiode Preamplifier Frequency Response
The addition of resistor R2 modifies the noise gain equation.
Note the signal gain equation has not changed.
Noise Gain = 1 +
R3
R1 || R2
C1 RISE
31.2ns
R3
600Ω
+5V
T
R1
301Ω
C1 FALL
31.6ns
CH1 500mV
M 50ns CH1
VIN
830mV
4
R2
50Ω
5
AD8067
3
2
–5V
C1
10μF
C2
0.1μF
1
R4
51Ω
C4
0.1μF
VOUT
RL
C3
10μF
Figure 54. Gain of Less than 2 Schematic
Figure 53. Photodiode Preamplifier Pulse Response
This technique allows the designer to use the AD8067 in gain
configurations of less than 8. The drawback to this type of
compensation is that the input noise and offset voltages are
also amplified by the value of the noise gain. In addition, the
distortion performance is degraded. To avoid excessive
overshoot and ringing when driving a capacitive load, the
AD8067 should be buffered by a small series resistor; in this
case, a 51 Ω resistor was used.
Rev. A | Page 19 of 24
AD8067
Reference network:
VOUT
V+REF − 3 dB Bandwidth =
VIN
T
1
2π(R2 || R3)C2
Resistors R4 and R1 set the gain, in this case, an inverting gain
of 10 was selected. In this application, the input and output
bandwidths were set for approximately 10 Hz. The reference
network was set for a tenth of the input and output bandwidth,
at approximately 1 Hz.
R4
2.7kΩ
CH1 200mV
CH2 200mV
M 50ns CH1
C1
47μF
Figure 55. Gain of 2 Pulse Response
R1
300Ω
4
VIN
3
The AD8067 is well suited for low voltage single-supply
applications, given its N-channel JFET input stage and rail-torail output stage. It is fully specified for 5 V supplies. Successful
single-supply applications require attention to keep signal
voltages within the input and output headroom limits of the
amplifier. The input stage headroom extends to 1.7 V
(minimum) on a 5 V supply. The center of the input range is
0.85 V. The output saturation limit defines the hard limit of the
output headroom. This limit depends on the amount of current
the amplifier is sourcing or sinking, as shown in Figure 29.
Traditionally, an offset voltage is introduced in the input
network replacing ground as a reference. This allows the output
to swing about a dc reference point, typically midsupply.
Attention to the required headroom of the amplifier is
important, in this case, the required headroom from the
positive supply is 3 V; therefore, 1.5 V was selected as a
reference, which allows for a 100 mV signal at the input. Figure 56
shows the AD8067 configured for 5 V supply operation with a
reference voltage of 1.5 V. Capacitors C1 and C5 ac couple the
signal into and out of the amplifier and partially determine the
bandwidth of the input and output structures.
R2
70kΩ
5
C4
0.1μF
C5
15μF
VOUT
1
2
R3
30kΩ
RL
1kΩ
+5V
C2
6.8μF
Figure 56. Single-Supply Operation Schematic
HIGH GAIN, HIGH BANDWIDTH COMPOSITE
AMPLIFIER
The composite amplifier takes advantage of combining key
parameters that can otherwise be mutually exclusive of a
conventional single amplifier. For example, most precision
amplifiers have good dc characteristics but lack high speed ac
characteristics. Composite amplifiers combine the best of both
amplifiers to achieve superior performance over their single op
amp counterparts. The AD8067 and the AD8009 are well suited
for a composite amplifier circuit, combining dc precision with
high gain and bandwidth. The circuit runs off a ±5 V power
supply at approximately 20 mA of bias current. With a gain of
approximately 40 dB, the composite amplifier offers <1 pA
input current, a gain bandwidth product of 6.1 GHz, and a slew
rate of 630 V/μs.
1
2πR1C1
VOUTPUT – 3 dB Bandwidth =
C3
10μF
AD8067
SINGLE-SUPPLY OPERATION
VINPUT – 3 dB Bandwidth =
+5V
288mV
1
2πRL C5
Resistors R2 and R3 set a 1.5 V output bias point for the output
signal to swing about. It is critical to have adequate bypassing to
provide a good ac ground for the reference voltage. Generally,
the bandwidth of the reference network (R2, R3, and C2) is
selected to be one tenth that of the input bandwidth. This
ensures that any frequencies below the input bandwidth do not
pass through the reference network into the amplifier.
Rev. A | Page 20 of 24
AD8067
44
R2
4.99kΩ
+5V
C8
0.1μF
C7 +5V
10μF
40
38
4
5
AD8067
INPUT
3
2
–5V
C2
1 0.1μF
C4
0.1μF
C3
10μF
3
C5
5pF
7
AD8009
6
C6
0.001μF
4
C10
0.001μF
C9
10μF –5V
C11
0.01μF
2
36
OUTPUT
dB
R1
51.1Ω
42
C1
10μF
R5
50Ω
34
32
30
R4
200Ω
28
26
R3
21.5Ω
24
0.1
Figure 57. AD8067/AD8009 Composite Amplifier AV = 100, GBWP = 6.1 GHz
1
10
FREQUENCY – MHz
100
Figure 58. Gain Bandwidth Response
The composite amplifier is set for a gain of 100. The overall gain
is set by
VO R2
=
+1
VI R1
C1 AMPL
4V
The output stage is set for a gain of 10; therefore, the AD8067
has an effective gain of 10, thereby allowing it to maintain a
bandwidth in excess of 55 MHz.
The circuit can be tailored for different gain values; keeping the
ratios roughly the same ensures that the bandwidth integrity is
maintained. Depending on the board layout, Capacitor C5 can
be required to reduce ringing on the output. The gain bandwidth
and pulse responses are shown in Figure 58, Figure 59, and
Figure 60.
T
CH1 1V
M 25ns CH1
0V
Figure 59. Large Signal Response
Layout of this circuit requires attention to the routing and
length of the feedback path. It should be kept as short as
possible to minimize stray capacitance.
C1 AMPL
480mV
T
CH1 200mV
M 25ns CH1
Figure 60. Small Signal Response
Rev. A | Page 21 of 24
0V
AD8067
OUTLINE DIMENSIONS
2.90 BSC
5
4
2.80 BSC
1.60 BSC
1
2
3
PIN 1
0.95 BSC
1.90
BSC
1.30
1.15
0.90
1.45 MAX
0.15 MAX
0.50
0.30
0.22
0.08
SEATING
PLANE
10°
5°
0°
0.60
0.45
0.30
COMPLIANT TO JEDEC STANDARDS MO-178-AA
Figure 61. 5-Lead Small Outline Transistor Package [SOT-23}
(RT-5)
Dimensions shown in millimeters
ORDERING GUIDE
Model
AD8067ART-REEL
AD8067ART-REEL7
AD8067ART-R2
AD8067ARTZ-REEL 1
AD8067ARTZ-REEL71
AD8067ARTZ-R21
1
Temperature Range
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
Package Description
5-Lead SOT-23
5-Lead SOT-23
5-Lead SOT-23
5-Lead SOT-23
5-Lead SOT-23
5-Lead SOT-23
Z = Pb-free part, # denotes lead-free product may be top or bottom marked.
Rev. A | Page 22 of 24
Package Option
RT-5
RT-5
RT-5
RT-5
RT-5
RT-5
Branding
HAB
HAB
HAB
HAB#
HAB#
HAB#
AD8067
NOTES
Rev. A | Page 23 of 24
AD8067
NOTES
©2006 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
C03205–0–5/06(A)
Rev. A | Page 24 of 24