Intersil ISL97516IUZ-TK 600khz/1.2mhz pwm step-up regulator Datasheet

ISL97516
®
Data Sheet
March 16, 2007
600kHz/1.2MHz PWM Step-Up Regulator
Features
The ISL97516 is a high frequency, high efficiency step-up
voltage regulator operated at constant frequency PWM
mode. With an internal 2.0A, 200mΩ MOSFET, it can deliver
up to 1A output current at over 90% efficiency. The
selectable 600kHz and 1.2MHz allows smaller inductors and
faster transient response. An external compensation pin
gives the user greater flexibility in setting frequency
compensation allowing the use of low ESR Ceramic output
capacitors.
• >90% Efficiency
When shut down, it draws <1µA of current and can operate
down to 2.3V input supply. These features along with
1.2MHz switching frequency makes it an ideal device for
portable equipment and TFT-LCD displays.
FN9261.2
• 2.0A, 200mΩ Power MOSFET
• 2.3V to 5.5V Input
• Up to 25V Output
• 600kHz/1.2MHz Switching Frequency Selection
• Adjustable Soft-Start
• Internal Thermal Protection
• 1.1mm Max Height 8 Ld MSOP Package
• Pb-free Plus Anneal Available (RoHS compliant)
Applications
The ISL97516 is available in an 8 Ld MSOP package with a
maximum height of 1.1mm. The device is specified for
operation over the full -40°C to +85°C temperature range.
• TFT-LCD displays
Pinout
• PCMCIA cards
• DSL modems
• Digital cameras
ISL97516
(8 LD MSOP)
TOP VIEW
• GSM/CDMA phones
• Portable equipment
COMP 1
8 SS
FB 2
7 FSEL
EN 3
6 VDD
GND 4
5 LX
• Handheld devices
Ordering Information
PART
NUMBER
(Note)
PART
MARKING TAPE AND REEL
PACKAGE
(Pb-Free)
PKG.
DWG. #
ISL97516IUZ
7516Z
bulk pack (tubes)
8 Ld MSOP
MDP0043
ISL97516IUZ-T
7516Z
13” (2,500 pieces) 8 Ld MSOP
MDP0043
13” (1k pieces)
MDP0043
ISL97516IUZ-TK 7516Z
8 Ld MSOP
NOTE: Intersil Pb-free plus anneal products employ special Pb-free material
sets; molding compounds/die attach materials and 100% matte tin plate
termination finish, which are RoHS compliant and compatible with both SnPb
and Pb-free soldering operations. Intersil Pb-free products are MSL classified
at Pb-free peak reflow temperatures that meet or exceed the Pb-free
requirements of IPC/JEDEC J STD-020.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2006, 2007. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL97516
Absolute Maximum Ratings (TA = +25°C)
LX to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .27V
VDD to GND. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6V
COMP, FB, EN, SS, FSEL to GND . . . . . . . . . -0.3V to (VDD +0.3V)
Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C
Operating Ambient Temperature . . . . . . . . . . . . . . . .-40°C to +85°C
Operating Junction Temperature . . . . . . . . . . . . . . . . . . . . . . +135°C
Power Dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . See Curves
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
IMPORTANT NOTE: All parameters having Min/Max specifications are guaranteed. Typical values are for information purposes only. Unless otherwise noted, all tests
are at the specified temperature and are pulsed tests, therefore: TJ = TC = TA
Electrical Specifications
PARAMETER
VIN = 3.3V, VOUT = 12V, IOUT = 0mA, FSEL = GND, TA = -40°C to +85°C unless otherwise specified.
DESCRIPTION
CONDITIONS
MIN
TYP
MAX
UNIT
1
5
µA
IQ1
Quiescent Current - Shutdown
EN = 0V
IQ2
Quiescent Current - Not Switching
EN= VDD, FB = 1.3V
0.7
IQ3
Quiescent Current - Switching
EN = VDD, FB = 1.0V
3
4
mA
VFB
Feedback Voltage
1.294
1.309
V
IB-FB
Feedback Input Bias Current
0.01
0.5
µA
VDD
Input Voltage Range
5.5
V
DMAX-600kHz
Maximum Duty Cycle
FSEL = 0V
85
92
%
DMAX-1.2MHz
Maximum Duty Cycle
FSEL = VDD
85
90
%
1.5
2.0
A
1.272
2.3
mA
ILIM
Current Limit - Max Peak Input Current
IEN
Shutdown Input Bias Current
EN = 0V
0.01
rDS(ON)
Switch ON Resistance
VDD = 2.7V, ILX = 1A
0.2
ILX-LEAK
Switch Leakage Current
VSW = 27V
0.01
ΔVOUT/ΔVIN
Line Regulation
3V < VIN < 5.5V, VOUT = 12V
0.2
%
ΔVOUT/ΔIOUT
Load Regulation
VIN = 3.3V, VOUT = 12V, IO = 30mA to 200mA
0.3
%
FOSC1
Switching Frequency Accuracy
FSEL = 0V
500
620
740
kHz
FOSC2
Switching Frequency Accuracy
FSEL = VDD
1000
1250
1500
kHz
0.5
V
VIL
EN, FSEL Input Low Level
VIH
EN, FSEL Input High Level
GM
Error Amp Tranconductance
VDD-ON
VDD UVLO On Threshold
HYS
VDD UVLO hyeteresis
ISS
Soft-Start Charge Current
OTP
Over Temperature Protection
2
0.5
Ω
3
1.5
ΔI = 5µA
µA
µA
V
70
130
150
1µ/Ω
2.1
2.2
2.3
V
100
4
6
150
mV
8
µA
°C
FN9261.2
March 16, 2007
Block Diagram
EN
FSEL
REFERENCE
GENERATOR
VDD
OSCILLATOR
SS
SHUTDOWN &
START-UP
CONTROL
LX
PWM LOGIC
CONTROLLER
FET
DRIVER
COMPARATOR
CURRENT
SENSE
GND
FB
GM
AMPLIFIER
COMP
Pin Descriptions
PIN NUMBER
PIN NAME
DESCRIPTION
1
COMP
Compensation pin. Output of the internal error amplifier. Capacitor and resistor from COMP pin to ground.
2
FB
Voltage feedback pin. Internal reference is 1.294V nominal. Connect a resistor divider from VOUT.
VOUT = 1.294V (1 + R1/R2). See Typical Application Circuit.
3
EN
Shutdown control pin. Pull EN low to turn off the device.
4
GND
5
LX
6
VDD
Analog power supply input pin.
7
FSEL
Frequency select pin. When FSEL is set low, switching frequency is set to 620kHz. When connected to
high or VDD, switching frequency is set to 1.25MHz.
8
SS
Analog and power ground.
Power switch pin. Connected to the drain of the internal power MOSFET.
Soft-start control pin. Connect a capacitor to control the converter start-up.
Typical Application Circuit
R3
3.9kΩ
C5
4.7nF
1 COMP
R1
R2
10kΩ
2 FB
FSEL 7
3 EN
VDD 6
4 GND
S1
3
SS 8
85.2kΩ
LX 5
C3
27nF
C4
2.3V TO 5.5V
+ C1
0.1µF
22µF
10µH
D1
+ C2
12V
22µF
FN9261.2
March 16, 2007
Typical Performance Curves
95
92
90
90
VIN = 3.3V, VO = 9V,
fs = 620kHz
88
EFFICIENCY(%)
EFFICIENCY(%)
85
VIN = 5V, VO = 12V, fs = 1.25 MHz
80
VIN = 5V, VO = 12V, fs = 620 kHz
75
VIN = 5V, VO = 9V, fs = 620 kHz
70
86
84
82
VIN = 3.3V, VO = 12V,
fs = 620kHz
VIN = 3.3V, VO = 12V,
80
fs = 1.25MHz
78
65
VIN = 5V, VO = 9V, fs = 1.25MHz
VIN = 3.3V, VO = 9V,
fs = 1.25MHz
76
60
74
0
200
400
600
800
1000
0
100
IOUT (mA)
FIGURE 1. BOOST EFFICIENCY vs IOUT
200
300
IOUT (mA)
400
500
FIGURE 2. BOOST EFFICIENCY vs IOUT
0.7
0.9
0.8
VIN = 5V, VO = 12V,
VIN = 5V, VO = 9V,
fs = 1.25MHz
fs = 1.25MHz
0.6
VIN = 3.3V, VO = 12V,
VIN = 3.3V, VO = 9V,
fs = 1.25MHz
fs = 1.25MHz
0.6
LOAD REGULATION (%)
LOAD REGULATION(%)
0.7
VIN = 5V, VO = 9V,
fs = 620kHz
0.5
0.4
0.3
0.2
fs = 620kHz
200
400
fs = 1.25kHz
0.3
0.2
VIN = 3.3, VO = 12V,
fs = 620kHz
0
0
0
VIN = 3.3, VO = 9V,
0.4
0.1
VIN = 5V, VO = 12V,
0.1
0.5
600
800
1000
0
100
IOUT (mA)
200
300
400
500
IOUT (mA)
FIGURE 3. LOAD REGULATION vs IOUT
FIGURE 4. LOAD REGULATION vs IOUT
0.6
VO = 12V
VO = 9V, IO = 80mA
0.5
IO = 50mA to 300mA
LINE REGULATION (%)
fs = 1.25MHz
0.4
VO = 9V, IO = 100mA
fs = 620kHz
0.3
VIN = 3.3V
fs = 600kHz
VO = 12V, IO = 80mA
fs = 1.25MHz
0.2
0.1
0
VO = 12V, IO = 80mA
fs = 620kHz
-0.1
2
3
4
5
6
VIN (V)
FIGURE 5. LINE REGULATION vs VIN
4
FIGURE 6. TRANSIENT RESPONSE
FN9261.2
March 16, 2007
Typical Performance Curves
(Continued)
JEDEC JESD51-7 HIGH EFFECTIVE THERMAL
CONDUCTIVITY TEST BOARD
IO = 50mA to 300mA
1.0
VO = 12V
VIN = 3.3V
POWER DISSIPATION (W)
0.9
fs = 1.2MHz
870mW
0.8
0.7
θ
JA
0.6
0.5
0.4
=
M
SO
+1 P8
15
°C
/W
0.3
0.2
0.1
0
0
25
50
75 85
100
125
AMBIENT TEMPERATURE (°C)
FIGURE 7. TRANSIENT RESPONSE
FIGURE 8. PACKAGE POWER DISSIPATION vs AMBIENT
TEMPERATURE
JEDEC JESD51-3 LOW EFFECTIVE THERMAL
CONDUCTIVITY TEST BOARD
POWER DISSIPATION (W)
0.6
0.5
486mW
0.4
θ
JA
=
0.3
M
SO
+2 P8
06
°C
/W
0.2
0.1
0.0
0
25
50
75 85
100
125
AMBIENT TEMPERATURE (°C)
FIGURE 9. PACKAGE POWER DISSIPATION vs AMBIENT TEMPERATURE
Applications Information
The ISL97516 is a high frequency, high efficiency boost
regulator operated at constant frequency PWM mode. The
boost converter stores energy from an input voltage source
and deliver it to a higher output voltage. The input voltage
range is 2.3V to 5.5V and output voltage range is 5V to 25V.
The switching frequency is selectable between 600kHz and
1.2MHz allowing smaller inductors and faster transient
response. An external compensation pin gives the user
greater flexibility in setting output transient response and
tighter load regulation. The converter soft-start characteristic
can also be controlled by external CSS capacitor. The EN pin
allows the user to completely shutdown the device.
the boost converter operates in two cycles. During the first
cycle, as shown in Figure 11, the internal power FET turns
on and the Schottky diode is reverse biased and cuts off the
current flow to the output. The output current is supplied
from the output capacitor. The voltage across the inductor is
VIN and the inductor current ramps up in a rate of VIN/L, L is
the inductance. The inductance is magnetized and energy is
stored in the inductor. The change in inductor current is:
V IN
ΔI L1 = ΔT1 × --------L
D
ΔT1 = -----------F SW
D = Duty Cycle
Boost Converter Operations
Figure 10 shows a boost converter with all the key
components. In steady state operating and continuous
conduction mode where the inductor current is continuous,
5
I OUT
ΔV O = ---------------- × ΔT 1
C OUT
(EQ. 1)
FN9261.2
March 16, 2007
During the second cycle, the power FET turns off and the
Schottky diode is forward biased, (Figure 12). The energy
stored in the inductor is pumped to the output supplying
output current and charging the output capacitor. The
Schottky diode side of the inductor is clamp to a Schottky
diode above the output voltage. So the voltage drop across
the inductor is VIN - VOUT. The change in inductor current
during the second cycle is:
L
D
VOUT
VIN
COUT
CIN
ISL97516
IL
ΔIL2
ΔT2
V IN – V OUT
ΔI L = ΔT2 × -------------------------------L
ΔVO
1–D
ΔT2 = ------------F SW
FIGURE 12. BOOST CONVERTER - CYCLE 2, POWER
SWITCH OPEN
(EQ. 2)
For stable operation, the same amount of energy stored in
the inductor must be taken out. The change in inductor
current during the two cycles must be the same.
ΔI1 + ΔI2 = 0
V IN 1 – D V IN – V OUT
D
------------ × --------+ ------------- × -------------------------------- = 0
L
F SW
L
F SW
V OUT
1
---------------- = ------------1–D
V IN
Output Voltage
An external feedback resistor divider is required to divide the
output voltage down to the nominal 1.294V reference
voltage. The current drawn by the resistor network should be
limited to maintain the overall converter efficiency. The
maximum value of the resistor network is limited by the
feedback input bias current and the potential for noise being
coupled into the feedback pin. A resistor network less than
100k is recommended. The boost converter output voltage is
determined by the relationship:
(EQ. 3)
L
D
VOUT
VIN
CIN
(EQ. 4)
The nominal VFB voltage is 1.294V.
Inductor Selection
COUT
ISL97516
FIGURE 10. BOOST CONVERTER
L
VOUT
VIN
R 1⎞
⎛
V OUT = V FB × ⎜ 1 + -------⎟
R
⎝
2⎠
The inductor selection determines the output ripple voltage,
transient response, output current capability, and efficiency.
Its selection depends on the input voltage, output voltage,
switching frequency, and maximum output current. For most
applications, the inductance should be in the range of 2µH to
33µH. The inductor maximum DC current specification must
be greater than the peak inductor current required by the
regulator. The peak inductor current can be calculated:
:
COUT
CIN
ISL97516
I OUT × V OUT
V IN × ( V OUT – V IN )
I L ( PEAK ) = ------------------------------------ + 1 ⁄ 2 × ----------------------------------------------------V IN
L × V OUT × FREQ
(EQ. 5)
Output Capacitor
IL
ΔIL1
ΔT1
ΔVO
FIGURE 11. BOOST CONVERTER - CYCLE 1, POWER
SWITCH CLOSED
Low ESR capacitors should be used to minimized the output
voltage ripple. Multilayer ceramic capacitors (X5R and X7R)
are preferred for the output capacitors because of their lower
ESR and small packages. Tantalum capacitors with higher
ESR can also be used. The output ripple can be calculated
as:
I OUT × D
ΔV O = --------------------------- + I OUT × ESR
F SW × C O
(EQ. 6)
For noise sensitive application, a 0.1µF placed in parallel
with the larger output capacitor is recommended to reduce
the switching noise coupled from the LX switching node.
6
FN9261.2
March 16, 2007
Schottky Diode
Maximum Output Current
In selecting the Schottky diode, the reverse break down
voltage, forward current and forward voltage drop must be
considered for optimum converter performance. The diode
must be rated to handle 2.0A, the current limit of the
ISL97516. The breakdown voltage must exceed the
maximum output voltage. Low forward voltage drop, low
leakage current, and fast reverse recovery will help the
converter to achieve the maximum efficiency.
The MOSFET current limit is nominally 2.0A and guaranteed
1.7A. This restricts the maximum output current, IOMAX,
based on the following formula:
Input Capacitor
I L = I L-AVG + ( 1 ⁄ 2 × ΔI L )
(EQ. 7)
where:
IL = MOSFET current limit
IL-AVG = average inductor current
The value of the input capacitor depends the input and
output voltages, the maximum output current, the inductor
value and the noise allowed to put back on the input line. For
most applications, a minimum 10µF is required. For
applications that run close to the maximum output current
limit, input capacitor in the range of 22µF to 47µF is
recommended.
ΔIL = inductor ripple current
V IN × [ ( V O + V DIODE ) – V IN ]
ΔI L = -----------------------------------------------------------------------------L × ( V O + V DIODE ) × F S
VDIODE = Schottky diode forward voltage, typically, 0.6V
FS = switching frequency, 600kHz or 1.2MHz
The ISL97516 is powered from the VIN. A High frequency
0.1µF bypass cap is recommended to be close to the VIN
pin to reduce supply line noise and ensure stable operation.
I OUT
I L-AVG = ------------1–D
Loop Compensation
D = MOSFET turn-on ratio:
The ISL97516 incorporates a transconductance amplifier in
its feedback path to allow the user some adjustment on the
transient response and better regulation. The ISL97516
uses current mode control architecture which has a fast
current sense loop and a slow voltage feedback loop. The
fast current feedback loop does not require any
compensation. The slow voltage loop must be compensated
for stable operation. The compensation network is a series
RC network from COMP pin to ground. The resistor sets the
high frequency integrator gain for fast transient response
and the capacitor sets the integrator zero to ensure loop
stability. For most applications, the compensation resistor in
the range of 2k to 7.5k and the compensation capacitor in
the range of 3nF to 10nF.
V IN
D = 1 – -------------------------------------------V OUT + V DIODE
Soft-Start
The soft-start is provided by an internal 6µA current source
charges the external CSS, the peak MOSFET current is
limited by the voltage on the capacitor. This in turn controls
the rising rate of the output voltage. The regulator goes
through the start-up sequence as well after the EN pin is
pulled to HI.
Frequency Selection
The ISL97516 switching frequency can be user selected to
operate at either constant 620kHz or 1.25MHz. Connecting
FSEL pin to ground sets the PWM switching frequency to
620kHz. When connecting FSEL high or VDD, the switching
frequency is set to 1.25MHz.
(EQ. 8)
(EQ. 9)
(EQ. 10)
Table 1 gives typical maximum IOUT values for 1.2MHz
switching frequency and 10µH inductor.
TABLE 1.
VIN (V)
VOUT (V)
IOMAX (mA)
2.5
5
870
2.5
9
500
2.5
12
380
3.3
5
1150
3.3
9
655
3.3
12
500
5
9
990
5
12
750
Cascaded MOSFET Application
An 25V N-channel MOSFET is integrated in the boost
regulator. For the applications where the output voltage is
greater than 25V, an external cascaded MOSFET is needed
as shown in Figure 12. The voltage rating of the external
MOSFET should be greater than AVDD.
Shutdown Control
When the EN pin is pulled down, the ISL97516 is shutdown
reducing the supply current to <1µA.
7
FN9261.2
March 16, 2007
DC PATH BLOCK APPLICATION
AVDD
VIN
LX
FB
INTERSIL
ISL97516
Note that there is a DC path in the boost converter from the
input to the output through the inductor and diode, hence the
input voltage will be seen at output with a forward voltage
drop of diode before the part is enabled. If this voltage is not
desired, the following circuit can be inserted between input
and inductor to disconnect the DC path when the part is
disabled.
TO INDUCTOR
INPUT
EN
FIGURE 13. CASCADED MOSFET TOPOLOGY FOR HIGH
OUTPUT VOLTAGE APPLICATIONS
8
FIGURE 14. CIRCUIT TO DISCONNECT THE DC PATH OF
BOOST CONVERTER
FN9261.2
March 16, 2007
Mini SO Package Family (MSOP)
0.25 M C A B
D
MINI SO PACKAGE FAMILY
(N/2)+1
N
E
MDP0043
A
E1
MILLIMETERS
PIN #1
I.D.
1
B
(N/2)
e
H
C
SEATING
PLANE
0.10 C
N LEADS
SYMBOL
MSOP8
MSOP10
TOLERANCE
NOTES
A
1.10
1.10
Max.
-
A1
0.10
0.10
±0.05
-
A2
0.86
0.86
±0.09
-
b
0.33
0.23
+0.07/-0.08
-
c
0.18
0.18
±0.05
-
D
3.00
3.00
±0.10
1, 3
E
4.90
4.90
±0.15
-
E1
3.00
3.00
±0.10
2, 3
e
0.65
0.50
Basic
-
L
0.55
0.55
±0.15
-
L1
0.95
0.95
Basic
-
N
8
10
Reference
-
0.08 M C A B
b
Rev. D 2/07
NOTES:
1. Plastic or metal protrusions of 0.15mm maximum per side are not
included.
L1
2. Plastic interlead protrusions of 0.25mm maximum per side are
not included.
A
3. Dimensions “D” and “E1” are measured at Datum Plane “H”.
4. Dimensioning and tolerancing per ASME Y14.5M-1994.
c
SEE DETAIL "X"
A2
GAUGE
PLANE
L
A1
0.25
3° ±3°
DETAIL X
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
9
FN9261.2
March 16, 2007
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