LINER LT3757 Monolithic high voltage isolated flyback converter no external start-up resistor Datasheet

LT3511
Monolithic High Voltage
Isolated Flyback Converter
Features
n
n
n
n
n
n
n
n
n
Description
4.5V to 100V Input Voltage Range
Internal 240mA, 150V Power Switch
Boundary Mode Operation
No Transformer Third Winding or
Opto-Isolator Required for Regulation
Improved Primary-Side Winding Feedback
Load Regulation
VOUT Set with Two External Resistors
BIAS Pin for Internal Bias Supply and Power
Switch Driver
No External Start-Up Resistor
16-Lead MSOP Package
The LT®3511 is a high voltage monolithic switching regulator specifically designed for the isolated flyback topology.
No third winding or opto-isolator is required for regulation as the part senses output voltage directly from the
primary-side flyback waveform. The device integrates a
240mA, 150V power switch, high voltage circuitry, and
control into a high voltage 16-lead MSOP package with
four leads removed.
Applications
n
n
n
Isolated Telecom Power Supplies
Isolated Auxiliary/Housekeeping Power Supplies
Isolated Industrial, Automotive and Medical Power
Supplies
The LT3511 operates from an input voltage range of 4.5V
to 100V and delivers up to 2.5W of isolated output power.
Two external resistors and the transformer turns ratio
easily set the output voltage. Off-the-shelf transformers
are available for several applications. The high level of
integration and the use of boundary mode operation results
in a simple, clean, tightly regulated application solution to
the traditionally tough problem of isolated power delivery.
L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks
and No RSENSE is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners. Protected by U.S. Patents, including 5438499, 7471522.
Typical Application
48V to 5V Isolated Flyback Converter
1µF
4:1
1M
EN/UVLO
43.2k
VIN
LT3511
RFB
RREF
169k
10k
TC
SW
VC
69.8k
•
300µH
•
5.25
VOUT+
5V
0.3A
19µH
5.20
5.15
22µF
VOUT–
5.10
VOUT (V)
VIN
36V TO 72V
Output Load and Line Regulation
VIN = 48V
5.05
VIN = 36V
5.00
VIN = 72V
4.95
4.90
4.85
GND BIAS
4.80
16.9k
3.3nF
4.7µF
3511 TA01a
4.75
0
50
150
200
100
LOAD CURRENT (mA)
250
300
3511 TA01b
3511fa
1
LT3511
Absolute Maximum Ratings
(Note 1)
Pin Configuration
SW (Note 4).............................................................150V
VIN, EN/UVLO...........................................................100V
RFB.............................................................100V, VIN ±6V
BIAS....................................................................VIN, 20V
RREF, TC, VC.................................................................6V
Operating Junction Temperature Range (Note 2)
LT3511E, LT3511I................................ –40°C to 125°C
LT3511H.............................................. –40°C to 150°C
LT3511MP........................................... –55°C to 150°C
Storage Temperature Range................... –65°C to 150°C
TOP VIEW
EN/UVLO 1
16 SW
VIN 3
14 RFB
GND
BIAS
NC
GND
5
6
7
8
12
11
10
9
RREF
TC
VC
GND
MS PACKAGE
16(12)-LEAD PLASTIC MSOP
θJA = 90°C/W
Order Information
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3511EMS#PBF
LT3511EMS#TRPBF
3511
16-Lead Plastic MSOP
–40°C to 125°C
LT3511IMS#PBF
LT3511IMS#TRPBF
3511
16-Lead Plastic MSOP
–40°C to 125°C
LT3511HMS#PBF
LT3511HMS#TRPBF
3511
16-Lead Plastic MSOP
–40°C to 125°C
LT3511MPMS#PBF
LT3511MPMS#TRPBF
3511
16-Lead Plastic MSOP
–55°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
Electrical Characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V unless otherwise noted.
PARAMETER
MAX
UNITS
100
15
V
V
2.7
0
3.5
mA
µA
1.15
1.21
1.27
V
2.0
2.6
0
3.3
µA
µA
Maximum Current Limit
240
330
430
mA
Minimum Current Limit
35
60
90
mA
1.18
1.17
1.20
1.215
1.23
V
V
0.01
0.03
%/V
80
400
nA
Input Voltage Range
CONDITIONS
MIN
l
VIN = BIAS
Quiescent Current
Not Switching
VEN/UVLO = 0.2V
EN/UVLO Pin Threshold
EN/UVLO Pin Voltage Rising
EN/UVLO Pin Current
VEN/UVLO = 1.1V
VEN/UVLO = 1.4V
l
6
4.5
Maximum Switching Frequency
Switch VCESAT
650
ISW = 100mA
l
RREF Voltage Line Regulation
6V < VIN < 100V
RREF Pin Bias Current
(Note 3)
Error Amplifier Voltage Gain
∆I = 2µA
kHz
0.3
RREF Voltage
Error Amplifier Transconductance
TYP
l
V
150
V/V
140
µmhos
3511fa
2
LT3511
Electrical Characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
Minimum Switching Frequency
TC Current into RREF
RTC = 53.6k
BIAS Pin Voltage
Internally Regulated
3
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3511E is guaranteed to meet performance specifications
from 0°C to 125°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LT3511I is guaranteed to meet performance specifications from –40°C to
125°C operating junction temperature range. The LT3511H is guaranteed
Output Voltage
5.20
Quiescent Current
5.15
3
2
4.90
4.85
VIN = 24V
VIN = 48V
VIN = 100V
4.80
0
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
0
Switch VCESAT
Switch Current Limit
350
25 50 75 100 125 150
TEMPERATURE (°C)
Quiescent Current vs VIN
4
MAXIMUM CURRENT LIMIT
3
300
250
IQ (mA)
800
CURRENT LIMIT (mA)
SWITCH VCESAT VOLTAGE (mV)
0
3511 G03
400
200
VIN = 24V, 10mA
VIN = 24V, NO LOAD
3511 G02
1000
400
V
3.0
2.0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
3511 G01
600
3.2
2.5
1
4.75
–50 –25
3.1
3.5
BIAS VOLTAGE (V)
IQ (mA)
VOUT (V)
4.95
µA
BIAS Pin Voltage
4
5.00
9.5
4.0
5.10
5.05
kHz
TA = 25°C, unless otherwise noted.
5
VIN = 48V
UNITS
to meet performance specifications from –40°C to 150°C operating
junction temperature range. The LT3511MP is guaranteed over the full
–55°C to 150°C operating junction range. High junction temperatures
degrade operating lifetimes. Operating lifetime is derated at junction
temperatures greater than 125°C.
Note 3: Current flows out of the RREF pin.
Note 4: The SW pin is rated to 150V for transients. Operating waveforms
of the SW pin should keep the pedestal of the flyback waveform below
100V as shown in Figure 5.
Typical Performance Characteristics
5.25
MAX
40
200
2
150
100
1
MINIMUM CURRENT LIMIT
50
0
0
50
100 150 200 250
SWITCH CURRENT (mA)
300
350
3511 G04
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3511 G05
0
0
20
60
40
VOLTAGE (V)
80
100
3511 G06
3511fa
3
LT3511
Typical Performance Characteristics
EN/UVLO Pin (Hysteresis) Current
vs Temperature
EN/UVLO = 1.2V
EN/UVLO PIN CURRENT (µA)
EN/UVLO PIN CURRENT (µA)
4
3
2
1
0
–50 –25
0
EN/UVLO Threshold
vs Temperature
EN/UVLO Pin Current
vs VEN/UVLO
30
3.0
25
2.5
20
15
10
2.0
1.5
1.0
0.5
5
0
25 50 75 100 125 150
TEMPERATURE (°C)
EN/UVLO THRESHOLD (V)
5
TA = 25°C, unless otherwise noted.
1
20
40
60
80
VEN/UVLO VOLTAGE (V)
3511 G07
0
–50 –25
100
0
25 50 75 100 125 150
TEMPERATURE (°C)
3511 G09
3511 G08
Maximum Frequency
vs Temperature
Minimum Frequency
vs Temperature
1000
100
800
80
EN/UVLO Shutdown Threshold
vs Temperature
0.9
600
400
200
EN/UVLO THRESHOLD (V)
MINIMUM FREQUENCY (kHz)
MAXIMUM FREQUENCY (kHz)
0.8
60
40
20
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3511 G10
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3511 G11
3511 G14
Light Load Discontinuous
Mode Waveform
Boundary Mode Waveform
20V/DIV
20V/DIV
1µs/DIV
3511 G12
2µs/DIV
3511 G13
3511fa
4
LT3511
Pin Functions
EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The EN/
UVLO pin is used to start up the LT3511. Pull the pin to 0V
to shut down the LT3511. This pin has an accurate 1.21V
threshold and can be used to program an undervoltage
lockout (UVLO) threshold using a resistor divider from
supply to ground. A 2.6μA pin current hysteresis allows
the programming of undervoltage lockout (UVLO) hysteresis. EN/UVLO can be directly connected to VIN. If left
open circuit the part will not power up.
VIN (Pin 3): Input Supply Pin. This pin supplies current to
the internal start-up circuitry, and serves as a reference
voltage for the DCM comparator and feedback circuitry.
Must be locally bypassed with a capacitor.
GND (Pin 5, 8, 9): Ground Pins. All three pins should be
tied directly to the local ground plane.
BIAS (Pin 6): Bias Voltage. This pin supplies current to
the switch driver and internal circuitry of the LT3511.
This pin may also be connected to VIN if a third winding
is not used and if VIN < 20V. The part can operate down
to 4.5V when BIAS and VIN are connected together. If a
third winding is used, the BIAS voltage should be lower
than the input voltage and greater than 3.3V for proper
operation. BIAS must be bypassed with a 4.7μF capacitor
placed close to the pin.
VC (Pin 10): Compensation Pin for Internal Error Amplifier.
Connect a series RC from this pin to ground to compensate
the switching regulator. An additional 100pF capacitor from
this pin to ground helps eliminate noise.
TC (Pin 11): Output Voltage Temperature Compensation. Connect a resistor to ground to produce a current
proportional to absolute temperature to be sourced into
the RREF node.
ITC = 0.55V/RTC.
RREF (Pin 12): Input Pin for External Ground-Referred
Reference Resistor. The resistor at this pin should be 10k.
For nonisolated applications, a traditional resistor voltage
divider from VOUT may be connected to this pin.
RFB (Pin 14): Input Pin for External Feedback Resistor.
This pin is connected to the transformer primary (VSW).
The ratio of this resistor to the RREF resistor, times the
internal bandgap reference, determines the output voltage
(plus the effect of any non-unity transformer turns ratio).
For nonisolated applications, this pin should be connected
to GND with a 1M resistor.
SW (Pin 16): Switch Pin. Collector of the internal power
switch. Minimize trace area at this pin to minimize EMI
and voltage spikes.
3511fa
5
LT3511
Block Diagram
D1
VIN
VOUT +
T1
C1
L1A
L1B
C2
R3
VOUT –
N:1
TC
CURRENT
TC
VIN
Q3
SW
RFB
FLYBACK
ERROR
AMP
Q2
R5
I2
1.2V
–g
m
+
–
+
ONE
SHOT
CURRENT
COMPARATOR
A2
–
A1
+
S
S
BIAS
DRIVER
BIAS
R1
EN/UVLO
R
1.2V
R2
+
A5
–
3µA
Q4
INTERNAL
REFERENCE
AND
REGULATORS
Q1
Q
MASTER
LATCH
C4
–
VIN
RREF
R4
+
V1
120mV
+
–
A4
RSENSE
0.02Ω
GND
OSCILLATOR
VC
R6
C3
3511 BD
3511fa
6
LT3511
Operation
The LT3511 is a current mode switching regulator IC designed specifically for the isolated flyback topology. The
key problem in isolated topologies is how to communicate
information regarding the output voltage from the isolated
secondary side of the transformer to the primary side.
Historically, optoisolators or extra transformer windings
communicate this information across the transformer.
Optoisolator circuits waste output power, and the extra
components increase the cost and physical size of the
power supply. Optoisolators can also exhibit trouble due
to limited dynamic response, nonlinearity, unit-to-unit
variation and aging over life. Circuits employing an extra
transformer winding also exhibit deficiencies. Using an
extra winding adds to the transformer’s physical size and
cost, and dynamic response is often mediocre.
In the LT3511, the primary-side flyback pulse provides
information about the isolated output voltage. In this manner, neither optoisolator nor extra transformer winding is
required for regulation. Two resistors program the output
voltage. Since this IC operates in boundary mode, the part
calculates output voltage from the switch pin when the
secondary current is almost zero.
The Block Diagram shows an overall view of the system.
Many of the blocks are similar to those found in traditional
switching regulators including internal bias regulator, oscillator, logic, current amplifier, current comparator, driver,
and output switch. The novel sections include a special
flyback error amplifier and a temperature compensation
circuit. In addition, the logic system contains additional
logic for boundary mode operation.
The LT3511 features boundary mode control, where the
part operates at the boundary between continuous conduction mode and discontinuous conduction mode. The
VC pin controls the current level just as it does in normal
current mode operation, but instead of turning the switch
on at the start of the oscillator period, the part turns on the
switch when the secondary-side winding current is zero.
Boundary Mode Operation
Boundary mode is a variable frequency, current mode
switching scheme. The switch turns on and the inductor
current increases until a VC pin controlled current limit.
After the switch turns off, the voltage on the SW pin rises
to the output voltage divided by the secondary-to-primary
transformer turns ratio plus the input voltage. When the
secondary current through the diode falls to zero, the SW
pin voltage falls below VIN. A discontinuous conduction
mode (DCM) comparator detects this event and turns the
switch back on.
Boundary mode returns the secondary current to zero every
cycle, so parasitic resistive voltage drops do not cause load
regulation errors. Boundary mode also allows the use of a
smaller transformer compared to continuous conduction
mode and does not exhibit subharmonic oscillation.
At low output currents, the LT3511 delays turning on the
switch, and thus operates in discontinuous mode. Unlike
traditional flyback converters, the switch has to turn on
to update the output voltage information. Below 0.6V on
the VC pin, the current comparator level decreases to
its minimum value, and the internal oscillator frequency
decreases. With the decrease of the internal oscillator,
the part starts to operate in DCM. The output current is
able to decrease while still allowing a minimum switch off
time for the flyback error amplifier. The typical minimum
internal oscillator frequency with VC equal to 0V is 40kHz.
3511fa
7
LT3511
Applications Information
PSUEDO DC THEORY
In the Block Diagram, RREF (R4) and RFB (R3) are external
resistors used to program the output voltage. The LT3511
operates similar to traditional current mode switchers,
except in the use of a unique error amplifier, which derives
its feedback information from the flyback pulse.
Operation is as follows: when the output switch, Q1, turns
off, its collector voltage rises above the VIN rail. The amplitude of this flyback pulse, i.e., the difference between
it and VIN, is given as:
VFLBK = (VOUT + VF + ISEC • ESR) • NPS
VF = D1 forward voltage
ISEC = Transformer secondary current
ESR = Total impedance of secondary circuit
NPS = Transformer effective primary-to-secondary turns
ratio
RFB and Q2 convert the flyback voltage into a current. Nearly
all of this current flows through RREF to form a groundreferred voltage. The resulting voltage forms the input
to the flyback error amplifier. The flyback error amplifier
samples the voltage information when the secondary side
winding current is zero. The bandgap voltage, 1.20V, acts
as the reference for the flyback error amplifier.
The relatively high gain in the overall loop will then cause
the voltage at RREF to be nearly equal to the bandgap
reference voltage VBG. The resulting relationship between
VFLBK and VBG approximately equals:
⎛ VFLBK ⎞ VBG
⎛R ⎞
or VFLBK = VBG ⎜ FB ⎟
⎜
⎟=
⎝ RFB ⎠ RREF
⎝ RREF ⎠
VBG = Internal bandgap reference
Combination of the preceding expression with earlier
derivation of VFLBK results in the following equation:
⎛ R ⎞⎛ 1 ⎞
VOUT = VBG ⎜ FB ⎟ ⎜
⎟ – VF – ISEC (ESR)
⎝ RREF ⎠ ⎝ NPS ⎠
The expression defines VOUT in terms of the internal reference, programming resistors, transformer turns ratio
and diode forward voltage drop. Additionally, it includes
8
the effect of nonzero secondary output impedance (ESR).
Boundary control mode minimizes the effect of this impedance term.
Temperature Compensation
The first term in the VOUT equation does not have temperature dependence, but the diode forward drop has a
significant negative temperature coefficient. A positive
temperature coefficient current source connects to the
RREF pin to compensate. A resistor to ground from the
TC pin sets the compensation current.
The following equation explains the cancellation of the
temperature coefficient:
δVF
R
1 δVTC
= – FB •
•
or,
δT
RTC NPS δT
RTC =
δV
–RFB
1
R
•
• TC ≈ FB
NPS δVF / δT δT
NPS
(δVF/δT) = Diode’s forward voltage temperature coefficient
(δVTC/δT) = 2mV
VTC = 0.55V
Experimentally verify the resulting value of RTC and adjust as
necessary to achieve optimal regulation over temperature.
The addition of a temperature coefficient current modifies
the expression of output voltage as follows:
⎛ R ⎞⎛ 1 ⎞
VOUT = VBG ⎜ FB ⎟ ⎜
⎟ – VF
⎝ RREF ⎠ ⎝ NPS ⎠
⎛V ⎞ R
– ⎜ TC ⎟ • FB – ISEC (ESR)
⎝ RTC ⎠ NPS
Output Power
A flyback converter has a complicated relationship between the input and output current compared to a buck
or a boost. A boost has a relatively constant maximum
input current regardless of input voltage and a buck has a
relatively constant maximum output current regardless of
input voltage. This is due to the continuous nonswitching
behavior of the two currents. A flyback converter has both
discontinuous input and output currents which makes it
3511fa
LT3511
Applications Information
similar to a nonisolated buck-boost. The duty cycle will
affect the input and output currents, making it hard to
predict output power. In addition, the winding ratio can
be changed to multiply the output current at the expense
of a higher switch voltage.
One design example would be a 5V output converter with
a minimum input voltage of 36V and a maximum input
voltage of 72V. A four-to-one winding ratio fits this design
example perfectly and outputs close to 1.6W at 72V but
lowers to 1W at 36V.
The graphs in Figures 1-4 show the typical maximum output
power possible for the output voltages 3.3V, 5V, 12V and
24V. The maximum power output curve is the calculated
output power if the switch voltage is 100V during the offtime. 50V of margin is left for leakage voltage spike. To
achieve this power level at a given input, a winding ratio
value must be calculated to stress the switch to 100V,
resulting in some odd ratio values. The following curves
are examples of common winding ratio values and the
amount of output power at given input voltages.
The equations below calculate output power:
Power = η • VIN • D • IPEAK • 0.5
Efficiency = η = ~85%
Duty Cycle = D =
Peak switch current = IPEAK = 0.26AV
3.0
3.5
3.0
N = NPS(MAX)
2.0
N = 15 N = 12
N = 10
N=8
OUTPUT POWER (W)
OUTPUT POWER (W)
2.5
N=6
1.5
N=4
1.0
N=2
0.5
0
( VOUT + VF ) • NPS
( VOUT + VF ) • NPS + VIN
N=5
2.5
N=4
N = NPS(MAX)
N=3
2.0
N=2
1.5
N=1
1.0
0.5
0
20
40
60
INPUT VOLTAGE (V)
0
100
80
0
20
40
60
INPUT VOLTAGE (V)
80
3511 F01
3511 F03
Figure 1. Output Power for 3.3V Output
Figure 3. Output Power for 12V Output
3.0
3.0
OUTPUT POWER (W)
N = NPS(MAX)
2.0
1.5
N=3
N=2
1.0
N=1
0.5
0
20
40
60
INPUT VOLTAGE (V)
80
N = NPS(MAX)
2.5
N=7
N=6
N=5
N=4
OUTPUT POWER (W)
N=8
2.5
0
100
N=2
2.0
N=1
1.5
1.0
0.5
100
3511 F02
Figure 2. Output Power for 5V Output
0
0
20
40
60
INPUT VOLTAGE (V)
80
100
3511 F04
Figure 4. Output Power for 24V Output
3511fa
9
LT3511
Applications Information
TRANSFORMER DESIGN CONSIDERATIONS
Successful application of the LT3511 relies on proper
transformer specification and design. Carefully consider
the following information in addition to the traditional
guidelines associated with high frequency isolated power
supply transformer design.
Linear Technology has worked with several leading magnetic component manufacturers to produce pre-designed
flyback transformers for use with the LT3511. Table 1
shows the details of these transformers.
Table 1. Predesigned Transformers
TRANSFORMER
PART NUMBER
LPRI (μH)
LEAKAGE (μH)
NP:NS:NB
ISOLATION (V)
SATURATION
CURRENT (mA)
VENDOR
750311558
300
1.5
4:1:1
1500
500
Würth Elektronik
48V to 5V, 0.3A
24V to 5V, 0.2A
12V to 5V, 0.13A
48V to 3.3V, 0.33A
24V to 3.3V, 0.28A
12V to 3.3V, 0.18A
750311019
400
5
6:1:2
1500
750
Würth Elektronik
24V to 5V, 0.26A
12V to 5V, 0.17A
48V to 3.3V, 0.43A
24V to 3.3V, 0.35A
12V to 3.3V, 0.2A
750311659
300
2
1:1:0.2
1500
560
Würth Elektronik
48V to 24V, 0.07A
750311660
350
3
2:1:0.33
1500
520
Würth Elektronik
48V to 15V, 0.1A
48V to 12V, 0.12A
24V to 15V, 0.09A
12V to 15V, 0.045A
750311838
350
3
2:1:1
1500
520
Würth Elektronik
48V to ±15V, 0.05A
48V to ±12V, 0.06A
24V to ±15V, 0.045A
750311963
200
0.4
1:5:5
1500
650
Würth Elektronik
12V to ±70V, 0.004A
12V to ±100V, 0.003A
12V to ±150V, 0.002A
750311966
120
0.45
1:5:0.5
1500
900
Würth Elektronik
12V to +120V and
–12V, 0.002A
10396-T024
300
2.0
4:1:1
1500
500
Sumida
48V to 5V, 0.3A
24V to 5V, 0.2A
12V to 5V, 0.13A
48V to 3.3V, 0.33A
24V to 3.3V, 0.28A
12V to 3.3V, 0.18A
10396-T026
300
2.5
6:1:2
1500
500
Sumida
24V to 5V, 0.26A
12V to 5V, 0.17A
48V to 3.3V, 0.43A
24V to 3.3V, 0.35A
12V to 3.3V, 0.2A
01355-T057
250
2.0
1:1:0.2
1500
500
Sumida
48V to 24V, 0.07A
10396-T022
300
2.0
2:1:0.33
1500
500
Sumida
48V to 15V, 0.1A
48V to 12V, 0.12A
24V to 15V, 0.09A
12V to 15V, 0.045A
10396-T028
300
2.5
2:1:1
1500
500
Sumida
48V to ±15V, 0.05A
48V to ±12V, 0.06A
24V to ±15V, 0.045A
TARGET
APPLICATIONS
3511fa
10
LT3511
Applications Information
Turns Ratio
Saturation Current
Note that when using an RFB/RREF resistor ratio to set
output voltage, the user has relative freedom in selecting
a transformer turns ratio to suit a given application. In
contrast, the use of simple ratios of small integers, e.g.,
1:1, 2:1, 3:2, provides more freedom in setting total turns
and mutual inductance.
The current in the transformer windings should not exceed its rated saturation current. Energy injected once the
core is saturated will not be transferred to the secondary
and will instead be dissipated in the core. Information on
saturation current should be provided by the transformer
manufacturers. Table 1 lists the saturation current of the
transformers designed for use with the LT3511.
Typically, choose the transformer turns to maximize available output power. For low output voltages (3.3V or 5V), a
N:1 turns ratio can be used with multiple primary windings
relative to the secondary to maximize the transformer’s
current gain (and output power). However, remember that
the SW pin sees a voltage that is equal to the maximum
input supply voltage plus the output voltage multiplied by
the turns ratio. In addition, leakage inductance will cause
a voltage spike (VLEAKAGE) on top of this reflected voltage.
This total quantity needs to remain below the absolute
maximum rating of the SW pin to prevent breakdown of
the internal power switch. Together these conditions place
an upper limit on the turns ratio, N, for a given application.
Choose a turns ratio low enough to ensure:
N<
150V – VIN(MAX) – VLEAKAGE
VOUT + VF
For larger N:1 values, a transformer with a larger physical
size is needed to deliver additional current and provide a
large enough inductance value to ensure that the off-time
is long enough to accurately measure the output voltage.
For larger N:1 values, choose a transformer with a larger
physical size to deliver additional current. In addition,
choose a large enough inductance value to ensure that
the off-time is long enough to measure the output voltage.
For lower output power levels, choose a 1:1 or 1:N transformer for the absolute smallest transformer size. A 1:N
transformer will minimize the magnetizing inductance
(and minimize size), but will also limit the available output
power. A higher 1:N turns ratio makes it possible to have
very high output voltages without exceeding the breakdown
voltage of the internal power switch.
The turns ratio is an important element in the isolated
feedback scheme. Make sure the transformer manufacturer
guarantees turns ratio accuracy within ±1%.
Primary Inductance Requirements
The LT3511 obtains output voltage information from the
reflected output voltage on the switch pin. The conduction
of secondary winding current reflects the output voltage
on the primary. The sampling circuitry needs a minimum
of 400ns to settle and sample the reflected output voltage.
In order to ensure proper sampling, the secondary winding
needs to conduct current for a minimum of 400ns. The following equation gives the minimum value for primaryside
magnetizing inductance:
LPRI ≥
tOFF(MIN) • NPS • ( VOUT + VF )
IPEAK(MIN)
tOFF(MIN) = 400ns
IPEAK(MIN) = 55mA
Leakage Inductance and Clamp Circuits
Transformer leakage inductance (on either the primary or
secondary) causes a voltage spike to appear at the primary
after the output switch turns off. This spike is increasingly
prominent at higher load currents where more stored energy must be dissipated. When designing an application,
adequate margin should be kept for the effect of leakage
voltage spikes. In most cases the reflected output voltage
on the primary plus VIN should be kept below 100V. This
leaves at least 50V of margin for the leakage spike across
line and load conditions. A larger voltage margin will be
needed for poorly wound transformers or for excessive
leakage inductance. Figure 5 illustrates this point. Minimize
transformer leakage inductance.
A clamp circuit is recommended for most applications.
3511fa
11
LT3511
Applications Information
VSW
VSW
<150V
<150V
<140V
VLEAKAGE
<100V
<100V
t OFF > 400ns
t OFF > 400ns
TIME
tSP < 150ns
tSP < 150ns
without Clamp
3511 F05
TIME
with Clamp
Figure 5. Maximum Voltages for SW Pin Flyback Waveform
Two circuits that can protect the internal power switch
include the RCD (resistor-capacitor-diode) clamp and the
DZ (diode-Zener) clamp. The clamp circuits dissipate the
stored energy in the leakage inductance. The DZ clamp
is the recommended clamp for the LT3511. Simplicity of
design, high clamp voltages, and low power levels make the
DZ clamp the preferred solution. Additionally, a DZ clamp
ensures well defined and consistent clamping voltages.
Figure 5 shows the clamp effect on the switch waveform
and Figure 6 shows the connection of the DZ clamp.
Proper care must be taken when choosing both the diode
and the Zener diode. Schottky diodes are typically the best
choice, but some PN diodes can be used if they turn on
fast enough to limit the leakage inductance spike. Choose
a diode that has a reverse-voltage rating higher than the
maximum input voltage. The Zener diode breakdown voltage should be chosen to balance power loss and switch
voltage protection. The best compromise is to choose the
largest voltage breakdown. Use the following equation to
make the proper choice:
VZENER(MAX) ≤ 150V – VIN(MAX)
For an application with a maximum input voltage of 72V,
choose a 68V VZENER which has VZENER(MAX) at 72V, which
will be below the 78V maximum.
The power loss in the clamp will determine the power rating of the Zener diode. Power loss in the clamp is highest
at maximum load and minimum input voltage. The switch
LS
Z
D
3511 F06
Figure 6. DZ Clamp
current is highest at this point along with the energy stored
in the leakage inductance. A 0.5W Zener will satisfy most
applications when the highest VZENER is chosen. Choosing
a low value for VZENER will cause excessive power loss as
shown in the following equations:
1
• L • IPK(VIN(MIN))2 • fSW •
2 
⎛
⎞
NPS • ( VOUT + VF )
⎜⎜1+
⎟⎟
⎝ VZENER – NPS • ( VOUT + VF ) ⎠
L  = Leakage Inductance
VOUT • IOUT • 2
IPK(VIN(MIN)) =
η • VIN(MIN) • DVIN(MIN)
DZ Power Loss =
fSW =

1
1
=
tON + tOFF LPRI • IPK(VIN(MIN)) LPRI • IPK(VIN(MIN))
+
VIN(MIN)
NPS • ( VOUT + VF )
3511fa
12
LT3511
Applications Information
Table 2 and 3 show some recommended diodes and Zener
diodes.
Table 2. Recommended Zener Diodes
VZENER
(V)
POWER
(W)
CASE
VENDOR
MMSZ5266BT1G
68
0.5
SOD-123
On Semi
MMSZ5270BT1G
91
0.5
SOD-123
CMHZ5266B
68
0.5
SOD-123
CMHZ5267B
75
0.5
SOD-123
BZX84J-68
68
0.5
SOD323F NXP
BZX100A
100
0.5
SOD323F
PART
Central
Semiconductor
Table 3. Recommended Diodes
pulse. The smaller flyback pulse results in a higher regulated
output voltage. The inductive divider effect of secondary
leakage inductance is load independent. RFB/RREF ratio
adjustments can accommodate this effect to the extent
secondary leakage inductance is a constant percentage
of mutual inductance (over manufacturing variations).
Winding Resistance Effects
Resistance in either the primary or secondary will reduce
overall efficiency (POUT/PIN). Good output voltage regulation will be maintained independent of winding resistance
due to the boundary mode operation of the LT3511.
Bifilar Winding
PART
I (A)
VREVERSE
(V)
BAV19W
0.625
100
SOD-123 Diodes Inc.
BAV20W
0.625
150
SOD-123
CASE
VENDOR
Leakage Inductance Blanking
When the power switch turns off, the flyback pulse appears. However, a finite time passes before the transformer primary-side voltage waveform approximately
represents the output voltage. Rise time on the SW node
and transformer leakage inductance cause the delay. The
leakage inductance also causes a very fast voltage spike
on the primary side of the transformer. The amplitude of
the leakage spike is largest when power switch current is
highest. Introduction of an internal fixed delay between
switch turn-off and the start of sampling provides immunity to the phenomena discussed above. The LT3511
sets internal blanking to 150ns. In certain cases leakage
inductance spikes last longer than the internal blanking,
but will not significantly affect output regulation.
Secondary Leakage Inductance
In addition to primary leakage inductance, secondary leakage inductance exhibits an important effect on application
design. Secondary leakage inductance forms an inductive
divider on the transformer secondary. The inductive divider
effectively reduces the size of the primary-referred flyback
A bifilar, or similar winding technique, is a good way to
minimize troublesome leakage inductances. However, remember that this will also increase primary-to-secondary
capacitance and limit the primary-to-secondary breakdown
voltage, so bifilar winding is not always practical. The
Linear Technology applications group is available and
extremely qualified to assist in the selection and/or design
of the transformer.
APPLICATION DESIGN CONSIDERATIONS
Iterative Design Process
The LT3511 uses a unique sampling scheme to regulate
the isolated output voltage. The use of this isolated scheme
requires a simple iterative process to choose feedback
resistors and temperature compensation. Feedback resistor values and temperature compensation resistance
is heavily dependent on the application, transformer and
output diode chosen.
Once resistor values are fixed after iteration, the values
will produce consistent output voltages with the chosen
transformer and output diode. Remember, the turns ratio
of the transformer must be guaranteed within ±1%. The
transformer vendors mentioned in this data sheet can
build transformers to this specification.
3511fa
13
LT3511
Applications Information
Selecting RFB and RREF Resistor Values
The following section provides an equation for setting
RFB and RREF values. The equation should only serve
as a guide. Follow the procedure outlined in the Design
Procedure to set accurate values for RFB, RREF and RTC
using the iterative design procedure.
Rearrangement of the expression for VOUT in the Temperature Compensation section, developed in the Operations
section, yields the following expression for RFB:
RFB =
RREF • NPS ⎡⎣( VOUT + VF ) + VTC ⎤⎦
VBG
where:
VOUT = Output voltage
VF = Switching diode forward voltage
NPS = Effective primary-to-secondary turns ratio
VTC = 0.55V
This equation assumes:
RTC
R
= FB
NPS
The equation assumes the temperature coefficients of
the diode and VTC are equal, which is a good first order
approximation.
Strictly speaking, the above equation defines RFB not as an
absolute value, but as a ratio of RREF. So the next question
is, what is the proper value for RREF? The answer is that
RREF should be approximately 10k. The LT3511 is trimmed
and specified using this value of RREF. If the impedance of
RREF varies considerably from 10k, additional errors will
result. However, a variation in RREF of several percent is
acceptable. This yields a bit of freedom in selecting standard 1% resistor values to yield nominal RFB/RREF ratios.
Undervoltage Lockout (UVLO)
A resistive divider from VIN to the EN/UVLO pin implements undervoltage lockout (UVLO). Figure 7 shows this
configuration. The EN/UVLO pin threshold is set at 1.21V.
VIN
R1
EN/UVLO
RUN/STOP
CONTROL
(OPTIONAL)
R2
LT3511
GND
3511 F07
Figure 7. Undervoltage Lockout (UVLO)
In addition, the EN/UVLO pin draws 2.6μA when the voltage at the pin is below 1.21V. This current provides user
programmable hysteresis based on the value of R1. The
effective UVLO thresholds are:
VIN(UVLO,RISING) =
1.2V • (R1+ R2)
VIN(UVLO,FALLING) =
+ 2.6µA • R1
R2
1.2V • (R1+ R2)
R2
Figure 7 also shows the implementation of external shutdown control while still using the UVLO function. The
NMOS grounds the EN/UVLO pin when turned on, and
puts the LT3511 in shutdown with quiescent current draw
of less than 1μA.
Minimum Load Requirement
The LT3511 recovers output voltage information using the
flyback pulse. The flyback pulse occurs once the switch
turns off and the secondary winding conducts current. In
order to regulate the output voltage, the LT3511 needs to
sample the flyback pulse. The LT3511 delivers a minimum
amount of energy even during light load conditions to
ensure accurate output voltage information. The minimum
delivery of energy creates a minimum load requirement
of 10mA to 15mA depending on the specific application.
Verify minimum load requirements for each application.
A Zener diode with a Zener breakdown of 20% higher
than the output voltage can serve as a minimum load if
pre-loading is not acceptable. For a 5V output, use a 6V
Zener with cathode connected to the output.
3511fa
14
LT3511
Applications Information
BIAS Pin Considerations
The BIAS pin powers the internal circuitry of the LT3511.
Three unique configurations exist for regulation of the BIAS
pin. In the first configuration, the internal LDO drives the
BIAS pin internally from the VIN supply. In the second setup,
the VIN supply directly drives the BIAS pin through a direct
connection bypassing the internal LDO. This configuration
will allow the part to operate down to 4.5V and up to 15V.
In the third configuration, an external supply or third winding drives the BIAS pin. Use this option when a voltage
supply exists lower than the input supply. Drive the BIAS
pin with a voltage supply higher than 3.3V to disable the
internal LDO. The lower voltage supply provides a more
efficient source of power for internal circuitry.
LT3511
VIN
6V TO 100V
LDO
3V
BIAS
LT3511
VIN
4.5V TO 15V
BIAS
OPTIONAL
VIN
Use the following design procedure as a guide to designing applications for the LT3511. Remember, the unique
sampling architecture requires an iterative process for
choosing correct resistor values.
VIN(MIN) = 36V, VIN(NOM) = 48V, VIN(MAX) = 72V, VOUT =
15V and IOUT = 100mA
LDO
BIAS
An external resistor-capacitor network compensates the
LT3511 on the VC pin. Typical compensation values are in
the range of RC = 20k and CC = 2.2nF (see the numerous
schematics in the Typical Applications section for other possible values). Proper choice of both RC and CC is important
to achieve stability and acceptable transient response. For
example, vulnerability to high frequency noise and jitter
result when RC is too large. On the other hand, if RC is
too small, transient performance suffers. The inverse is
true with respect to the value of CC. Transient response
suffers with too large of a CC, and instability results from
too small a CC. The specific value for RC and CC will vary
based on the application and transformer choice. Verify
specific choices with board level evaluation and transient
response performance.
The design example involves designing a 15V output with
a 100mA load current and an input range from 36V to 72V.
6V TO 100V
3.3V < BIAS < 20V
Loop Compensation
DESIGN PROCEDURE/DESIGN EXAMPLE
LDO
LT3511
an additional winding (often called a third winding) improves overall system efficiency. Design the third winding
to output a voltage between 3.3V and 12V. For a typical
48VIN application, overdriving the BIAS pin improves
efficiency 4% to 5%.
EXTERNAL
SUPPLY
3511 F08
Figure 8. BIAS Pin Configurations
Overdriving the BIAS Pin with a Third Winding
The LT3511 provides excellent output voltage regulation
without the need for an opto-coupler, or third winding, but
for some applications with higher input voltages (>20V),
Step 1: Select the transformer turns ratio.
NPS <
VSW(MAX) – VIN(MAX) – VLEAKAGE
VOUT + VF
VSW(MAX) = Max rating of internal switch = 150V
VLEAKAGE = Margin for transformer leakage spike = 40V
VF = Forward voltage of output diode = assume approximately ~ 0.5V
3511fa
15
LT3511
Applications Information
Example:
150V – 72V – 40V
15V + 0.5V
< 2.45
NPS <
NPS
NPS = 2
The choice of turns ratio is critical in determining output
power as shown earlier in the Output Power section. At
this point, a third winding can be added to the transformer
to drive the BIAS pin of the LT3511 for higher efficiencies.
Choose a turns ratio that sets the third winding voltage
to regulate between 3.3V and 6V for maximum efficiency.
Choose a third winding ratio to drive BIAS winding with
5V. (Optional)
Example:
NTHIRD VTHIRD 5V
=
=
= 0.33
NS
VOUT
15V
The turns ratio of the transformer chosen is as follows
NPRIMARY: NSECONDARY: NTHIRD = 2:1:0.33.
Step 2: Calculate maximum power output at minimum
VIN.
POUT(VIN(MIN)) = η • VIN(MIN) • IIN = η • VIN(MIN) • D •
IPEAK • 0.5
D=
( VOUT + VF ) • NPS
( VOUT + VF ) • NPS + VIN(MIN)
η = Efficiency = ~75%
IPEAK = Peak switch current = 0.26A
Example:
D = 0.46
POUT(VIN(MIN)) = 1.62
IOUT(VIN(MIN)) = POUT(VIN(MIN))/VOUT = 0.11A
The chosen turns ratio satisfies the output current requirement of 100mA. If the output current was too low,
the minimum input voltage could be adjusted higher. The
turns ratio in this example is set to its highest ratio given
switch voltage requirements and margin for leakage inductance voltage spike.
16
Step 3: Determine primary inductance, switching frequency and saturation current.
Primary inductance for the transformer must be set
above a minimum value to satisfy the minimum off time
requirement.
tOFF(MIN) • NPS • ( VOUT + VF )
LPRI ≥
IPEAK(MIN)
tOFF(MIN) = 400ns
IPEAK(MIN) = 55mA
Example:
LPRI ≥
400ns • 2 • (15 + 0.5)
0.055
LPRI ≥ 225µH
In addition, primary inductance will determine switching
frequency.
fSW =
1
1
=
tON + tOFF LPRI • IPEAK + LPRI • IPEAK
VIN
NPS • ( VOUT + VF )
IPEAK =
VOUT • IOUT • 2
η • VIN • D
Example:
Let’s calculate switching frequency at our nominal VIN
of 48V.
D=
(15 + 0.5) • 2 = 0.39
(15 + 0.5) • 2 + 48
IPEAK =
15V • 0.1A • 2
= 0.21A
0.75 • 48V • 0.39
Let’s choose LPRI = 350μH. Remember, most transformers specify primary inductance with a tolerance of ±20%.
fSW = 256kHz
Finally, the transformer needs to be rated for the correct
saturation current level across line and load conditions.
In the given example, the worst-case condition for switch
current is at minimum VIN and maximum load.
3511fa
LT3511
Applications Information
Step 5: Choose an output capacitor.
VOUT • IOUT • 2
η • VIN • D
15V • 0.1A • 2
=
= 0.24A
0.75 • 36V • 0.46
IPEAK =
IPEAK
The output capacitor choice should minimize output voltage
ripple and balance the trade-off between size and cost for
a larger capacitor. Use the equation below at nominal VIN:
Ensure that the saturation current covers steady-state
operation, start-up and transient conditions. To satisfy
these conditions, choose a saturation current 50% or more
higher than the steady-state calculation. In this example, a
saturation current between 400mA and 500mA is chosen.
Table 1 presents a list of pre-designed flyback transformers. For this application, the Würth 750311660 transformer
will be used.
Step 4: Choose the correct output diode.
The two main criteria for choosing the output diode include
forward current rating and reverse voltage rating. The
maximum load requirement is a good first-order guess
at the average current requirement for the output diode.
A better metric is RMS current.
IRMS = IPEAK(VIN(MIN)) • NPS •
1– DVIN(MIN)
3
Example:
IRMS = 0.24 • 2 •
1– 0.46
= 0.2A
3
Next calculate reverse voltage requirement using maximum VIN:
VREVERSE = VOUT +
VIN(MAX)
NPS
Example:
VREVERSE = 15V +
72V
= 51V
2
A 0.5A, 60V diode from Diodes Inc. (SBR0560S1) will
be used.
C=
IOUT • D
ΔVOUT • fSW
Example:
Design for ripple levels below 50mV.
C=
0.1A • 0.39
= 3.1µF
0.05V • 256kHz
A 10μF, 25V output capacitor is chosen. Remember ceramic capacitors lose capacitance with applied voltage.
The capacitance can drop to 40% of quoted capacitance
at the max voltage rating.
Step 6: Design clamp circuit.
The clamp circuit protects the switch from leakage inductance spike. A DZ clamp is the preferred clamp circuit. The
Zener and the diode need to be chosen.
The maximum Zener value is set according to the maximum VIN:
VZENER(MAX) ≤ 150V – VIN(MAX)
Example:
VZENER(MAX) ≤ 150V – 72V
VZENER(MAX) ≤ 78V
In addition, power loss in the clamp circuit is inversely
related to the clamp voltage as shown previously. Higher
clamp voltages lead to lower power loss.
A 68V Zener with a maximum of 72V will provide optimal
protection and minimize power loss. Half-watt Zeners will
satisfy most clamp applications involving the LT3511.
Power loss can be calculated using the equations presented
in the Leakage Inductance and Clamp Circuit section.
The Zener chosen is a 68V 0.5W Zener from On Semiconductor (MMSZ5266BT1G).
3511fa
17
LT3511
Applications Information
Choose a diode that is fast and has sufficient reverse
voltage breakdown:
VREVERSE > VIN(MAX)
Example:
VREVERSE > 72V
Step 7: Compensation.
Compensation will be optimized towards the end of the
design procedure. Connect a resistor and capacitor from
the VC node to ground. Use a 20k resistor and a 2.2nF
capacitor.
Step 8: Select RFB and RTC Resistors.
Use the following equations to choose starting values for
RFB and RTC. Set RREF to 10k.
( VOUT + VF + 0.55V ) • NPS • RREF
1.2V
RREF = 10k
R
= FB
NPS
RTC
Power up the application with application components
connected and measure the regulated output voltage.
Readjust RFB based on the measured output voltage.
RFB(NEW) =
The diode needs to handle the peak switch current of the
switch which was determined to be 0.24A. A 100V, 0.6A
diode from Diodes Inc. (BAV19W) is chosen.
RFB =
Step 9: Adjust RFB based on output voltage.
VOUT
VOUT(MEAS)
• RFB(OLD)
Example:
RFB(NEW) =
15V
• 267k = 237k
16.8V
Step 10: Remove RTC and measure output voltage over
temperature.
Measure output voltage in a controlled temperature environment like an oven to determine the output temperature
coefficient. Measure output voltage at a consistent load
current and input voltage, across the temperature range
of operation. This procedure will optimize line and load
regulation over temperature.
Calculate the temperature coefficient of VOUT:
ΔVOUT VOUT(HOT) – VOUT(COLD)
=
ΔTemp
THOT(°C) – TCOLD(°C)
Example:
VOUT measured at 100mA and 48VIN
Example:
RFB =
RTC
(15 + 0.5 + 0.55V ) • 2 • 10k = 267k
1.2V
267k
=
= 133k
2
ΔVOUT 15.70V – 15.37V
=
= 1.9mV/°C
ΔTemp 125°C – ( –50°C)
3511fa
18
LT3511
Applications Information
Step 11: Calculate new value for RTC.
RTC(NEW) =
RFB 1.85mV / °C
•
ΔVOUT
NPS
ΔTemp
Example:
RTC(NEW) =
237k 1.85
•
= 118k
2
1.9
VOUT
VOUT(MEAS)
• RFB(OLD)
Example:
RFB(NEW) =
Check minimum load requirement at maximum input
voltage. The minimum load occurs at the point where the
output voltage begins to climb up as the converter delivers
more energy than what is consumed at the output.
Example:
Step 12: Place new value for RTC, measure VOUT, and
readjust RFB due to RTC change.
RFB(NEW) =
Step 15: Ensure minimum load.
The minimum load at an input voltage of 72V is:
7mA
Step 16: EN/UVLO resistor values.
Determine amount of hysterysis required.
Voltage hysteresis = 2.6μA • R1
Example:
Choose 2V of hysteresis.
15V
• 237k = 237k
15V
Step 13: Verify new values of RFB and RTC over
temperature.
R1=
2V
= 768k
2.6µA
Determine UVLO Threshold.
Measure output voltage over temperature with RTC
connected.
VIN(UVLO,FALLING) =
Step 14: Optimize compensation.
R2 =
Now that values for RFB and RTC are fixed, optimize the
compensation. Compensation should be optimized for
transient response to load steps on the output. Check
transient response across the load range.
Example:
The optimal compensation for the application is:
RC = 22.1k, CC = 4.7nF
1.2V • (R1+ R2)
R2
1.2V • R1
VIN(UVLO,FALLING) – 1.2V
Set UVLO falling threshold to 30V.
1.2V • 768k
= 32.4k
30V – 1.2V
1.2V • (R1+ R2)
VIN(UVLO,FALLING) =
R2
1.2V • (768k + 32.4k )
=
= 30V
32.4k
R2 =
VIN(UVLO,RISING) = VIN(UVLO,FALLING) + 2.6μA • R1 = 30V
+ 2.6μA • 768k = 32V
3511fa
19
LT3511
Typical Applications
48V to 5V Isolated Flyback Converter
VIN
36V TO 72V
4:1:1
C1
1µF
R1
1M
Z1
VIN
R2
43.2k
EN/UVLO
RFB
RREF
LT3511
TC
D3
R3
169k
T1
300µH
R5
69.8k
GND
VOUT–
C1: TAIYO YUDEN HMK316B7105KL-T
C3: TAIYO YUDEN EMK212B7475KG
C4: MURATA GRM32ER71C226KE18B
D1, D2: DIODES INC. SBR140S3
D3: DIODES INC. BAV19W
T1: WÜRTH 750311558
Z1: ON SEMI MMSZ5266BT1G
R4
10k
BIAS
D2
R6
16.9k
C2
3.3nF
VOUT+
5V
0.3A
C4
22µF
19µH
SW
VC
D1
L1C
19µH
C3
4.7µF
3511 TA02
OPTIONAL THIRD
WINDING FOR
HV OPERATION
48V to 15V Isolated Flyback Converter
VIN
36V TO 72V
C1
1µF
D1
2:1
R1
1M
Z1
VIN
R2
43.2k
T1
350µH
EN/UVLO
RFB
RREF
LT3511
TC
R4
10k
SW
VC
R5
97.6k
R3
237k
GND
BIAS
R6
13k
C2
6.8nF
D2
88µH
VOUT+
15V
0.1A
C4
10µF
VOUT–
C1: TAIYO YUDEN HMK316B7105KL-T
C3: TAIYO YUDEN EMK212B7475KG
C4: MURATA GRM31CR71E106KA12
D1: DIODES INC. SBR0560S1
D2: DIODES INC. BAV19W
T1: WÜRTH 750311660
Z1: ON SEMI MMSZ5266BT1G
C3
4.7µF
3511 TA03
3511fa
20
LT3511
Typical Applications
48V to 24V Isolated Flyback Converter
VIN
36V TO 72V
C1
1µF
Z1
VIN
R2
43.2k
T1
300µH
EN/UVLO
R3
187k
RFB
RREF
LT3511
TC
R5
200k
GND
C4
4.7µF
VOUT–
C1: TAIYO YUDEN HMK316B7105KL-T
C3: TAIYO YUDEN EMK212B7475KG
C4: MURATA GRM32ER71H475KA88B
D1: DIODES INC. SBR1U150SA
D2: DIODES INC. BAV19W
T1: WÜRTH 750311659
Z1: ON SEMI MMSZ5266BT1G
R4
10k
BIAS
R6
33.2k
C2
3.3nF
300µH
D2
SW
VC
VOUT+
24V
65mA
D1
1:1
R1
1M
C3
4.7µF
3511 TA04
24V to 5V Isolated Flyback Converter
VIN
20V TO 30V
C1
4.7µF
D1
6:1
R1
1M
Z1
VIN
R2
80.6k
EN/UVLO
LT3511
RFB
RREF
SW
TC
VC
R5
73.2k
GND
BIAS
R6
9.31k
C2
15nF
R3
249k
R4
10k
D2
T1
300µH
8µH
VOUT+
5V
0.25A
C4
22µF
VOUT–
C1: MURATA GRM31CR71H475KA12B
C3: TAIYO YUDEN EMK212B7475KG
C4: MURATA GRM32ER71C226KE18B
D1: DIODES INC. SBR2A30P1
D2: DIODES INC. BAV19W
T1: SUMIDA 10396-T026
Z1: ON SEMI MMSZ5270BT1G
C3
4.7µF
3511 TA05
3511fa
21
LT3511
Typical Applications
24V to 15V Isolated Flyback Converter
VIN
20V TO 30V
C1
4.7µF
R1
1M
Z1
VIN
EN/UVLO
R2
80.6k
R3
237k
RFB
RREF
LT3511
D2
T1
350µH
VC
R5
133k
GND
VOUT–
C1: MURATA GRM31CR71H475KA12B
C3: TAIYO YUDEN EMK212B7475KG
C4: MURATA GRM31CR71E106KA12B
D1: DIODES INC. SBR140S3
D2: DIODES INC. BAV19W
T1: WÜRTH 750311660
Z1: ON SEMI MMSZ5270BT1G
R4
10k
BIAS
R6
20k
C2
4.7nF
C4
10µF
88µH
SW
TC
VOUT+
15V
0.09A
D1
2:1
C3
4.7µF
3511 TA06
12V to 15V Isolated Flyback Converter
VIN
8V TO 20V
C1
4.7µF
2:1
R1
1M
R2
562k
Z1
VIN
EN/UVLO
LT3511
RFB
RREF
SW
TC
VC
R5
133k
GND
BIAS
R6
26.1k
C2
4.7nF
C3
4.7µF
R3
237k
R4
10k
D2
T1
350µH
VOUT+
15V
40mA
D1
88µH
C4
4.7µF
Z2
VOUT–
OPTIONAL
MINIMUM LOAD
C1: MURATA GRM31CR71H475KA12B
C3: TAIYO YUDEN EMK212B7475KG
C4: MURATA GRM31CR71H475KA12
D1: DIODES INC. SBR130S3
D2: DIODES INC. BAV19W
T1: WÜRTH 750311660
Z1: ON SEMI MMSZ5270BT1G
3511 TA08
3511fa
22
LT3511
Typical Applications
12V to ±70V Isolated Flyback Converter
VIN
8V TO 20V
1:5:5
C1
2.2µF
R1
1M
VIN
Z1
EN/UVLO
R2
562k
LT3511
D3
R3
105k
RFB
RREF
D1
T1
80µH
C4
0.47µF
D2
R4
10k
VC
R5
191k
GND
BIAS
R6
90k
C2
6.8nF
VOUT1–
VOUT2+
4mA
C5
0.47µF
VOUT2–
–70V
SW
TC
VOUT1+
70V
4mA
C1: MURATA GRM21BR71E225KA73B
C3: TAIYO YUDEN EMK212B7475KG
C4, C5: NIPPON CHEMI-CON KTS251B474M43N0T00
D1, D2: CENTRAL SEMI CRM1U-06M
D3: DIODES INC. BAV19W
T1: WÜRTH 750311692
Z1: NXP BZX100A
C3
4.7µF
3511 TA07
48V to 3.3V Non-Isolated Flyback Converter
VIN
36V TO 72V
C1
1µF
D1
6:1
R1
1M
Z1
VIN
R2
43.2k
RFB
EN/UVLO
LT3511
D2
8.66k
RREF
11µH
VOUT
R4
5k
VC
GND
C1: TAIYO YUDEN HMK316B7105KL-T
C3: TAIYO YUDEN EMK212B7475KG
C4: TAIYO YUDEN LMK325B7476MM-TR
D1: DIODES INC. SBR2A30P1
D2: DIODES INC. BAV19W
T1: WÜRTH 750311019
Z1: ON SEMI MMSZ5266BT1G
BIAS
R6
8k
C2
4.7nF
C4
47µF
VOUT–
SW
TC
R5
1M
R3
1M
T1
400µH
VOUT
3.3V
0.4A
C3
4.7µF
3511 TA09
48V to 12V Isolated Flyback Converter
VIN
36V TO 72V
C1
1µF
D1
2:1
R1
1M
Z1
VIN
R2
43.2k
EN/UVLO
LT3511
RFB
RREF
GND
BIAS
TC
R4
10k
SW
VC
R5
143k
R3
191k
R6
15k
C2
6.8nF
D2
T1
300µH
75µH
VOUT+
12V
0.1A
C4
4.7µF
VOUT–
C1: TAIYO YUDEN HMK316B7105KL-T
C3: TAIYO YUDEN EMK212B7475KG
C4: MURATA GRM31CR71H475KA12
D1: DIODES INC. SBR0560S1
D2: DIODES INC. BAV19W
T1: SUMIDA 10396-T022
Z1: ON SEMI MMSZ5266BT1G
C3
4.7µF
3511 TA10
3511fa
23
LT3511
Package Description
MS Package
Varitation: MS16 (12)
16-Lead Plastic MSOP with 4 Pins Removed
(Reference LTC DWG # 05-08-1847 Rev A)
1.0
(.0394)
BSC
5.23
(.206)
MIN
0.889 ± 0.127
(.035 ± .005)
3.20 – 3.45
(.126 – .136)
4.039 ± 0.102
(.159 ± .004)
(NOTE 3)
16 14 121110 9
0.50
(.0197)
BSC
0.305 ± 0.038
(.0120 ± .0015)
TYP
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
0.280 ± 0.076
(.011 ± .003)
REF
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
4.90 ± 0.152
(.193 ± .006)
DETAIL “A”
0° – 6° TYP
1
GAUGE PLANE
0.53 ± 0.152
(.021 ± .006)
DETAIL “A”
0.18
(.007)
SEATING
PLANE
1.10
(.043)
MAX
0.17 – 0.27
(.007 – .011)
TYP
0.50
(.0197)
BSC
3 567 8
1.0
(.0394)
BSC
0.86
(.034)
REF
0.1016 ± 0.0508
(.004 ± .002)
MSOP (MS12) 0510 REV A
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
3511fa
24
LT3511
Revision History
REV
DATE
DESCRIPTION
A
4/11
Added MP-grade.
PAGE NUMBER
2, 3
Revised RFB pin description in the Pin Functions section.
5
Updated efficiency equation and Table 1 in the Applications Information section.
9, 10
Revised the Typical Applications drawings.
20, 21
3511fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
25
LT3511
Typical Application
48V to ±15V Isolated Flyback Converter
VIN
36V TO 72V
2:1:1
C1
1µF
R1
1M
Z1
VIN
R2
43.2k
EN/UVLO
RFB
LT3511
VC
D3
R3
237k
RREF
TC
R5
154k
D1
T1
350µH
88µH
C4
4.7µF
VOUT1–
D2
R4
10k
88µH
VOUT2+
50mA
C5
4.7µF
SW
GND
BIAS
R6
20k
C2
6.8nF
C3
4.7µF
3511 TA11
VOUT1+
15V
50mA
C1: TAIYO YUDEN HMK316B7105KL-T
C3: TAIYO YUDEN EMK212B7475KG
C4, C5: MURATA GRM31CR71H475KA12
D1, D2: DIODES INC. SBR0560S1
D3: DIODES INC. BAV19W
T1: WÜRTH 750311838
Z1: CENTRAL SEMI CMHZ5266B
VOUT2–
–15V
Related Parts
PART NUMBER
DESCRIPTION
COMMENTS
LT3958
High Input Voltage Boost, Flyback, SEPIC and
Inverting Converter
5V ≤ VIN ≤ 80V, 3.3A/84V Onboard Power Switch, 5mm × 6mm QFN-36
with High Voltage Pin Spacing
LT3748
100V Isolated Flyback Controller
5V ≤ VIN ≤ 100V, No Opto-Isolator or “Third Winding” Required, Onboard
Gate Driver, MSOP-16 with High Voltage Pin Spacing
LT3957
Boost, Flyback, SEPIC and Inverting Converter
3V ≤ VIN ≤ 40V, 5A/40V Onboard Power Switch, 5mm × 6mm QFN-36 with
High Voltage Pin Spacing
LT3956
Constant-Current, Constant-Voltage Boost, Buck,
Buck-Boost, SEPIC or Flyback Converter
4.5V ≤ VIN ≤ 80V, 3.3A/84V Onboard Power Switch, True PWM Dimming,
5mm × 6mm QFN-36 with High Voltage Pin Spacing
LT3575
Isolated Flyback Switching Regulator with 60V/2.5A
Integrated Switch
3V ≤ VIN ≤ 40V, No Opto-Isolator or “Third Winding” Required, Up to 14W,
TSSOP-16E
LT3573
Isolated Flyback Switching Regulator with 60V/1.25A
Integrated Switch
3V ≤ VIN ≤ 40V, No Opto-Isolator or “Third Winding” Required, Up to 7W,
MSOP-16E
LT3574
Isolated Flyback Switching Regulator with 60V/0.65A
Integrated Switch
3V ≤ VIN ≤ 40V, No Opto-Isolator or “Third Winding” Required, Up to 3W,
MSOP-16
LT3757
Boost, Flyback, SEPIC and Inverting Controller
2.9V ≤ VIN ≤ 40V, 100kHz to 1MHz Programmable Operating Frequency,
3mm × 3mm DFN-10 and MSOP-10E Package
LT3758
Boost, Flyback, SEPIC and Inverting Controller
5.5V ≤ VIN ≤ 100V, 100kHz to 1MHz Programmable Operating Frequency,
3mm × 3mm DFN-10 and MSOP-10E Package
LTC1871/LTC1871-1/ No RSENSE™ Low Quiescent Current Flyback, Boost
and SEPIC Controller
LTC1871-7
2.5V ≤ VIN ≤ 36V, Burst Mode® Operation
3511fa
26 Linear Technology Corporation
LT 0411 REV A • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
 LINEAR TECHNOLOGY CORPORATION 2010
Similar pages