LTC3801/LTC3801B Micropower Constant Frequency Step-Down DC/DC Controllers in ThinSOT U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO The LTC®3801/LTC3801B are constant frequency current mode step-down DC/DC controllers in a low profile (1mm max) 6-lead SOT-23 (ThinSOTTM) package. The parts provide excellent AC and DC load and line regulation with ±1.5% output voltage accuracy. The LTC3801 consumes only 195µA of quiescent current in normal operation, dropping to 16µA under no-load conditions. High Efficiency: Up to 94% Very Low No-Load Quiescent Current: Only 16µA (LTC3801) High Output Currents Easily Achieved Internal Soft-Start Wide VIN Range: 2.4V to 9.8V Low Dropout: 100% Duty Cycle Constant Frequency 550kHz Operation Burst Mode® Operation for High Efficiency at Light Loads (LTC3801) Burst Mode Operation Disabled for Lower Output Ripple at Light Loads (LTC3801B) Output Voltage as Low as 0.8V ±1.5% Voltage Reference Accuracy Current Mode Operation for Excellent Line and Load Transient Response Only 6µA Supply Current in Shutdown (LTC3801) Low Profile (1mm) SOT-23 Package The LTC3801/LTC3801B incorporate an undervoltage lockout feature that shuts down the device when the input voltage falls below 2.2V. The LTC3801 automatically switches into Burst Mode operation at light loads which enhances efficiency at low output current. In the LTC3801B, Burst Mode operation is disabled for lower output ripple at light loads. To further maximize the life of a battery source, the external P-channel MOSFET is turned on continuously in dropout (100% duty cycle). High switching frequency of 550kHz allows the use of a small inductor. U APPLICATIO S ■ ■ ■ 1- or 2-Cell Li-Ion Battery-Powered Applications Wireless Devices Portable Computers Distributed Power Systems U ■ , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a registered trademark of Linear Technology Corporation. ThinSOT is a trademark of Linear Technology Corporation. TYPICAL APPLICATIO LTC3801 Efficiency vs Load Current* 100 95 550kHz Micropower Step-Down DC/DC Controller VIN 2.7V TO 9.8V 10k VIN ITH/RUN LTC3801/ LTC3801B GND SENSE – VFB 10µF 0.025Ω VIN = 4.2V VIN = 3.3V 90 EFFICIENCY (%) 220pF VOUT = 2.5V 85 VIN = 6.6V 80 75 VIN = 9.8V 70 VIN = 8.4V 65 PGATE 402k 60 866k 4.7µH + 47µF VOUT 2.5V 2A 3801 TA01 55 50 0.1 1 10 100 1000 LOAD CURRENT (mA) 10000 3801 TA02 *SEE NO-LOAD IQ vs INPUT VOLTAGE ON THE LAST PAGE OF THIS DATA SHEET 3801f 1 LTC3801/LTC3801B U W U U W W W ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION (Note 1) Input Supply Voltage (VIN)........................ – 0.3V to 10V SENSE –, PGATE Voltages ............ – 0.3V to (VIN + 0.3V) VFB, ITH/RUN Voltages ............................. – 0.3V to 2.4V PGATE Peak Output Current (<10µs) ........................ 1A Operating Temperature Range (Note 2) .. – 40°C to 85°C Junction Temperature (Note 3) ............................ 150°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER TOP VIEW 6 PGATE ITH/RUN 1 LTC3801ES6 LTC3801BES6 5 VIN GND 2 4 SENSE – VFB 3 S6 PART MARKING S6 PACKAGE 6-LEAD PLASTIC TSOT-23 LTACR LTAHN TJMAX = 150°C, θJA = 230°C/W Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The ● indicates specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise noted. (Note 2) PARAMETER Input Voltage Range Input DC Supply Current Normal Operation SLEEP Mode Shutdown UVLO Undervoltage Lockout Threshold Start-Up Current Source Shutdown Threshold (at ITH/RUN) Regulated Feedback Voltage Feedback Voltage Line Regulation Feedback Voltage Load Regulation VFB Input Current Overvoltage Protect Threshold Overvoltage Protect Hysteresis Oscillator Frequency Normal Operation Output Short Circuit Gate Drive Rise Time Gate Drive Fall Time Peak Current Sense Voltage Peak Current Sense Voltage in Burst Mode Operation Default Soft-Start Time CONDITIONS ● Typicals at VIN = 4.2V (Note 4) 2.4V ≤ VIN ≤ 9.8V, VITH/RUN = 1.3V 2.4V ≤ VIN ≤ 9.8V (LTC3801 Only) 2.4V ≤ VIN ≤ 9.8V, VITH/RUN = 0V (LTC3801) 2.4V ≤ VIN ≤ 9.8V, VITH/RUN = 0V (LTC3801B) VIN < UVLO Threshold VIN Rising VIN Falling VITH/RUN = 0V (LTC3801) VITH/RUN = 0V (LTC3801B) VITH/RUN Rising 0°C ≤ TA ≤ 85°C (Note 5) –40°C ≤ TA ≤ 85°C (Note 5) 2.4V ≤ VIN ≤ 9.8V (Note 5) ITH/RUN Sinking 5µA (Note 5) ITH/RUN Sourcing 5µA (Note 5) (Note 5) Measured at VFB VFB = 0.8V VFB = 0V CLOAD = 3000pF CLOAD = 3000pF Duty Cycle < 40% (Note 6) LTC3801 LTC3801B LTC3801 Only Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC3801ES6/LTC3801BES6 are guaranteed to meet specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: MIN 2.4 TYP MAX 9.8 UNITS V 195 16 6 8 1 1.8 1.7 1 2 0.6 0.800 0.800 0.05 2 2 2 0.880 40 300 30 15 17 2 2.3 2.2 1.5 3.0 0.95 0.812 0.812 µA µA µA µA µA V V µA µA V V V mV/V mV/µA mV/µA nA V mV 500 550 210 40 40 650 kHz kHz ns ns 109 95 117 104 26 0.6 125 113 mV mV mV ms ● ● ● ● 0.5 1.0 0.3 0.788 0.780 0.850 ● ● 10 0.910 TJ = TA + (PD • θJA°C/W) Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: The LTC3801/LTC3801B are tested in a feedback loop that servos VFB to the output of the error amplifier while maintaining ITH/RUN at the midpoint of the current limit range. Note 6: Peak current sense voltage is reduced dependent on duty cycle as given in Figure 1. 3801f 2 LTC3801/LTC3801B U W TYPICAL PERFOR A CE CHARACTERISTICS Input DC Supply Current (Normal) vs Input Voltage 225 Input DC Supply Current (SLEEP) vs Input Voltage (LTC3801 Only) 20 TA = 25°C VITH/RUN = 1.3V 205 16 15 TA = 25°C LTC3801B IIN (µA) 14 6 185 12 3 10 2 3 4 5 6 7 8 9 10 2 3 4 5 6 LTC3801 0 7 8 9 10 2 3 4 5 VIN (V) VIN (V) 3801 G01 6 7 8 9 3801 G03 Shutdown Threshold vs Temperature 2.2 800 10 VIN (V) 3801 G02 Undervoltage Lockout Threshold vs Temperature Regulated Feedback Voltage vs Temperature 812 VIN = 4.2V VIN = 4.2V 808 2.0 700 1.8 VIN FALLING 1.6 804 VFB (mV) VITH/RUN (mV) VIN RISING VIN (V) 9 195 175 TA = 25°C VITH/RUN = 0V 12 IIN (µA) 18 IIN (µA) 215 Input DC Supply Current (Shutdown) vs Input Voltage 600 800 796 500 1.4 792 1.2 –50 –30 30 –10 10 50 TEMPERATURE (°C) 400 –50 90 70 –30 50 –10 10 30 TEMPERATURE (°C) 70 Regulated Feedback Voltage vs Input Voltage 600 TA = 25°C 590 80 90 3801 G06 Oscillator Frequency vs Temperature 808 Oscillator Frequency vs Input Voltage 560 VIN = 4.2V 580 TA = 25°C 555 570 800 796 560 fOSC (kHz) fOSC (kHz) 804 VFB (mV) 30 50 –10 10 TEMPERATURE (°C) 3801 G05 3801 G04 812 788 –50 –30 90 550 540 530 550 545 520 792 510 788 2 3 4 5 7 6 VIN (V) 8 9 10 3801 G07 500 –50 540 –30 30 –10 10 50 TEMPERATURE (°C) 70 90 3801 G08 2 3 4 5 7 6 VIN (V) 8 9 10 3801 G09 3801f 3 LTC3801/LTC3801B U U U PI FU CTIO S SENSE – (Pin 4): Current Sense Pin. An external sense resistor is connected between this pin and VIN (Pin 5). ITH/RUN (Pin 1): This pin performs two functions. It serves as the error amplifier compensation point as well as the run control input. Nominal voltage range for this pin is 0.7V to 1.9V. Forcing this pin below 0.6V causes the device to be shut down. In shutdown, all functions are disabled and the PGATE pin is held high. VIN (Pin 5): Supply Pin. This pin must be closely decoupled to GND (Pin 2). PGATE (Pin 6): Gate Drive for the External P-Channel MOSFET. This pin swings from 0V to VIN. GND (Pin 2): Ground Pin. VFB (Pin 3): Receives the feedback voltage from an external resistor divider across the output. W FU CTIO AL DIAGRA U U 4 SENSE – 15mV (LTC3801B) 5 VIN UNDERVOLTAGE LOCKOUT 1µA (LTC3801) 2µA (LTC3801B) 1 ITH/RUN – UV VOLTAGE REFERENCE 0.8V BURST DEFEAT (LTC3801B) BURST CLAMP + CURRENT COMPARATOR SHUTDOWN COMPARATOR 0.3V + + SLOPE COMPENSATION – ILIM ITH BUFFER SHDN – 550kHz OSCILLATOR RS R S LATCH Q – BURST DEFEAT (LTC3801B) VIN SLEEP COMPARATOR SWITCHING LOGIC AND BLANKING CIRCUIT SLEEP + FREQUENCY FOLDBACK PGATE 6 0V SOFT-START CLAMP OVERVOLTAGE COMPARATOR 0.15V – ERROR AMPLIFIER 0.225V + + 0.3V 0.88V – + 1.2V SHORT-CIRCUIT DETECT – VFB 3 0.8V GND 2 3801 FD 3801f 4 LTC3801/LTC3801B U OPERATIO (Refer to the Functional Diagram) Main Control Loop (Normal Operation) The LTC3801/LTC3801B are constant frequency current mode step-down switching regulator controllers. During normal operation, an external P-channel MOSFET is turned on each cycle when the oscillator sets the RS latch and turned off when the current comparator resets the latch. The peak inductor current at which the current comparator trips is controlled by the voltage on the ITH/RUN pin, which is the output of the error amplifier. The negative input to the error amplifier is the output feedback voltage VFB which is generated by an external resistor divider connected between VOUT and ground. When the load current increases, it causes a slight decrease in VFB relative to the 0.8V reference, which in turn causes the ITH/RUN voltage to increase until the average inductor current matches the new load current. The main control loop is shut down by pulling the ITH/RUN pin to ground. Releasing the ITH/RUN pin allows an internal 1µA current source (2µA on LTC3801B) to charge up the external compensation network. When the ITH/ RUN pin voltage reaches approximately 0.6V, the main control loop is enabled and the ITH/RUN voltage is pulled up by a clamp to its zero current level of approximately one diode voltage drop (0.7V). As the external compensation network continues to charge up, the corresponding peak inductor current level follows, allowing normal operation. The maximum peak inductor current attainable is set by a clamp on the ITH/RUN pin at 1.2V above the zero current level (approximately 1.9V). and the load will eventually cause the error amplifier output to start drifting higher. When the error amplifier output rises to 0.225V above its zero current level (approximately 0.925V), the sleep comparator will untrip and normal operation will resume. The next oscillator cycle will turn the external MOSFET on and the switching cycle will repeat. Low Load Current Operation (LTC3801B Only) Under very light load current conditions, the ITH/RUN pin voltage will be very close to the zero current level of 0.85V. As the load current decreases further, an internal offset at the current comparator input will ensure that the current comparator remains tripped (even at zero load current) and the regulator will start to skip cycles, as it must, in order to maintain regulation. This behavior allows the regulator to maintain constant frequency down to very light loads, resulting in less low frequency noise generation over a wide load current range. Figure 1 illustrates this result for the circuit on the front page of this data sheet using both an LTC3801 (in Burst Mode operation) and an LTC3801B (with Burst Mode operation disabled). At an output current of 100mA, the LTC3801 exhibits an output ripple of 81.6mVP-P, whereas the LTC3801B has an output ripple of only 17.6mVP-P. At lower output current levels, the improvement is even greater. This comes at a tradeoff of lower efficiency for the non Burst Mode part at light load currents (see Figure 2). Also notice the constant frequency operation of the LTC3801B, even at 5% of maximum output current. Burst Mode Operation (LTC3801 Only) Dropout Operation The LTC3801 incorporates Burst Mode operation at low load currents (<25% of IMAX). In this mode, an internal clamp sets the peak current of the inductor at a level corresponding to an ITH/RUN voltage 0.3V above its zero current level (approximately 1V), even though the actual ITH/RUN voltage is lower. When the inductor’s average current is greater than the load requirement, the voltage at the ITH/RUN pin will drop. When the ITH/RUN voltage falls to 0.15V above its zero current level (approximately 0.85V), the sleep comparator will trip, turning off the external MOSFET. In sleep, the input DC supply current to the IC is reduced to 16µA from 195µA in normal operation. With the switch held off, average inductor current will decay to zero When the input supply voltage decreases towards the output voltage, the rate of change of inductor current during the on cycle decreases. This reduction means that at some input-output differential, the external P-channel MOSFET will remain on for more than one oscillator cycle (start dropping off-cycles) since the inductor current has not ramped up to the threshold set by the error amplifier. Further reduction in input supply voltage will eventually cause the external P-channel MOSFET to be turned on 100%, i.e., DC. The output voltage will then be determined by the input voltage minus the voltage drop across the sense resistor, the MOSFET and the inductor. 3801f 5 LTC3801/LTC3801B U OPERATIO (Refer to the Functional Diagram) VOUT Ripple for Front Page Circuit Using the LTC3801 (with Burst Mode Operation) 20mVAC/DIV VOUT Ripple for Front Page Circuit Using the LTC3801B (Burst Mode Operation Disabled) 20mVAC/DIV VIN = 4.2V VOUT= 2.5V IOUT = 100mA 5µs/DIV 3801 F01a VIN = 4.2V VOUT= 2.5V IOUT = 100mA 5µs/DIV 3801 F01b Figure 1. Output Ripple Waveforms for the Front Page Circuit 100 95 90 EFFICIENCY (%) This lower frequency allows the inductor current to safely discharge, thereby preventing current runaway. After the short is removed, the oscillator frequency will gradually increase back to 550kHz as VFB rises through 0.3V on its way back to 0.8V. VOUT = 2.5V VIN = 3.3V 85 80 75 70 65 VIN = 4.2V VIN = 6.6V VIN = 9.8V 60 Overvoltage Protection VIN = 8.4V 55 50 0.1 1 100 1000 10 LOAD CURRENT (mA) 10000 3801 F02 If VFB exceeds its regulation point of 0.8V by more than 10% for any reason, such as an output short circuit to a higher voltage, the overvoltage comparator will hold the external P-channel MOSFET off. This comparator has a typical hysteresis of 40mV. Figure 2. LTC3801B Efficiency vs Load Current Slope Compensation and Inductor’s Peak Current Undervoltage Lockout Protection To prevent operation of the external P-channel MOSFET with insufficient gate drive, an undervoltage lockout circuit is incorporated into the LTC3801/LTC3801B. When the input supply voltage drops below approximately 1.7V, the P-channel MOSFET and all internal circuitry other than the undervoltage block itself are turned off. Input supply current in undervoltage is approximately 1µA. Short-Circuit Protection If the output is shorted to ground, the frequency of the oscillator is folded back from 550kHz to approximately 210kHz while maintaining the same minimum on time. The switch on duty cycle in normal operation is given by: Duty Cycle = VOUT + VD VIN + VD where VD is the forward voltage drop of the external diode at the average inductor current. For duty cycles less than 40%, the inductor’s peak current is determined by: IMAX = VITH/RUN – 0.7 V 10 RSENSE However, for duty cycles greater than 40%, slope compensation begins and effectively reduces the peak 3801f 6 LTC3801/LTC3801B U OPERATIO (Refer to the Functional Diagram) 100 115 90 105 Soft-Start An internal default soft-start circuit is employed at powerup and/or when coming out of shutdown. The soft-start circuit works by internally clamping the voltage at the ITH/RUN pin to the corresponding zero current level and gradually raising the clamp voltage such that the minimum time required for the programmed switch current to reach its maximum is approximately 0.6ms. After the soft-start circuit has timed out, it is disabled until the part is put in shutdown again or the input supply is cycled. TRIP VOLTAGE (mV) LTC3801 95 80 LTC3801B 85 70 75 60 65 50 55 35 40 VIN = 4.2V TA = 25°C 45 20 30 40 LTC3801 SLOPE FACTOR (%) inductor current. The amount of reduction is given by the curve in Figure 3. 50 60 70 80 DUTY CYCLE (%) 90 30 100 3801 F03 Figure 3. Maximum Current Limit Trip Voltage vs Duty Cycle U W U U APPLICATIO S I FOR ATIO The basic LTC3801/LTC3801B application circuit is shown on the front page of this data sheet. External component selection is driven by the load requirement and begins with the selection of the inductor and RSENSE. Next, the power MOSFET and the output diode are selected followed by the input bypass capacitor CIN and output bypass capacitor COUT. However, for operation that is above 40% duty cycle, slope compensation effect has to be taken into consideration to select the appropriate value to provide the required amount of current. Using Figure 3, the value of RSENSE is: RSENSE Selection for Output Current where SF is the “Slope Factor.” RSENSE is chosen based on the required output current. With the current comparator monitoring the voltage developed across RSENSE, the threshold of the comparator determines the inductor’s peak current. The output current the LTC3801 can provide is given by: Inductor Value Calculation IOUT = 0.117 IRIPPLE − RSENSE 2 where IRIPPLE is the inductor peak-to-peak ripple current (see Inductor Value Calculation section). For the LTC3801B use 104mV in the previous equation and follow through the analysis using that number. A reasonable starting point for setting ripple current is IRIPPLE = (0.4)(IOUT). Rearranging the above equation, it becomes: RSENSE = 1 for Duty Cycle < 40% (10)(IOUT ) RSENSE = SF (10)(IOUT )(100) The operating frequency and inductor selection are interrelated in that higher operating frequencies permit the use of a smaller inductor for the same amount of inductor ripple current. However, this is at the expense of efficiency due to an increase in MOSFET gate charge losses. The inductance value also has a direct effect on ripple current. The ripple current, IRIPPLE, decreases with higher inductance or frequency and increases with higher VIN or VOUT. The inductor’s peak-to-peak ripple current is given by: IRIPPLE = VIN − VOUT VOUT + VD f(L) VIN + VD where f is the operating frequency. Accepting larger values of IRIPPLE allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is 3801f 7 LTC3801/LTC3801B U W U U APPLICATIO S I FOR ATIO IRIPPLE = 0.4(IOUT(MAX)). Remember, the maximum IRIPPLE occurs at the maximum input voltage. In Burst Mode operation on the LTC3801, the ripple current is normally set such that the inductor current is continuous during the burst periods. Therefore, the peakto-peak ripple current must not exceed: IRIPPLE ≤ 0.03 RSENSE This implies a minimum inductance of: LMIN = VIN − VOUT VOUT + VD 0.03 VIN + VD f RSENSE (Use VIN(MAX) = VIN) A smaller value than LMIN could be used in the circuit; however, the inductor current will not be continuous during burst periods. Inductor Core Selection manufacturer is Kool Mµ. Toroids are very space efficient, especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more difficult. However, new designs for surface mount that do not increase the height significantly are available. Power MOSFET Selection An external P-channel power MOSFET must be selected for use with the LTC3801/LTC3801B. The main selection criteria for the power MOSFET are the threshold voltage VGS(TH) and the “on” resistance RDS(ON), reverse transfer capacitance CRSS and total gate charge. Since the LTC3801/LTC3801B are designed for operation down to low input voltages, a sublogic level threshold MOSFET (RDS(ON) guaranteed at VGS = 2.5V) is required for applications that work close to this voltage. When these MOSFETs are used, make sure that the input supply to the LTC3801/LTC3801B is less than the absolute maximum VGS rating, typically 8V. The required minimum RDS(ON) of the MOSFET is governed by its allowable power dissipation. For applications that may operate the LTC3801/LTC3801B in dropout, i.e., 100% duty cycle, at its worst case the required RDS(ON) is given by: Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool Mµ® cores. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! where PP is the allowable power dissipation and δp is the temperature dependency of RDS(ON). (1 + δp) is generally given for a MOSFET in the form of a normalized RDS(ON) vs temperature curve, but δp = 0.005/°C can be used as an approximation for low voltage MOSFETs. Molypermalloy (from Magnetics, Inc.) is a very good, low loss core material for toroids, but it is more expensive than ferrite. A reasonable compromise from the same where DC is the maximum operating duty cycle of the LTC3801/LTC3801B. RDS(ON)DC=100% = PP (IOUT(MAX) )2 (1+ δp) In applications where the maximum duty cycle is less than 100% and the LTC3801/LTC3801B are in continuous mode, the RDS(ON) is governed by: RDS(ON) ≅ PP (DC)IOUT2 (1+ δp) Kool Mµ is a registered trademark of Magnetics, Inc. 3801f 8 LTC3801/LTC3801B U W U U APPLICATIO S I FOR ATIO Output Diode Selection The catch diode carries load current during the off-time. The average diode current is therefore dependent on the P-channel switch duty cycle. At high input voltages the diode conducts most of the time. As VIN approaches VOUT the diode conducts only a small fraction of the time. The most stressful condition for the diode is when the output is short-circuited. Under this condition the diode must safely handle IPEAK at close to 100% duty cycle. Therefore, it is important to adequately specify the diode peak current and average power dissipation so as not to exceed the diode ratings. Under normal load conditions, the average current conducted by the diode is: V −V ID= IN OUT I OUT VIN + VD The allowable forward voltage drop in the diode is calculated from the maximum short-circuit current as: VF ≈ PD ISC(MAX) where PD is the allowable power dissipation and will be determined by efficiency and/or thermal requirements. A fast switching diode must also be used to optimize efficiency. Schottky diodes are a good choice for low forward drop and fast switching times. Remember to keep lead length short and observe proper grounding (see Board Layout Checklist) to avoid ringing and increased dissipation. An additional consideration in applications where low noload quiescent current is critical is the reverse leakage current of the diode at the regulated output voltage. A leakage greater than several microamperes can represent a significant percentage of the total input current. CIN and COUT Selection In continuous mode, the source current of the P-channel MOSFET is a square wave of duty cycle (VOUT + VD)/ (VIN + VD). To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: 1/ 2 VOUT ( VIN − VOUT )] [ CIN Required IRMS ≈ IMAX VIN This formula has a maximum value at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may be paralleled to meet the size or height requirements in the design. Due to the high operating frequency of the LTC3801/LTC3801B, ceramic capacitors can also be used for CIN. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (∆VOUT) is approximated by: 1 ∆VOUT ≈ IRIPPLE ESR + 8 fCOUT where f is the operating frequency, COUT is the output capacitance and IRIPPLE is the ripple current in the inductor. The output ripple is highest at maximum input voltage since ∆IL increases with input voltage. Manufacturers such as Nichicon, United Chemicon and Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR (size) product of any aluminum electrolytic at a somewhat higher price. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. In surface mount applications, multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in 3801f 9 LTC3801/LTC3801B U W U U APPLICATIO S I FOR ATIO surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS, AVX TPSV and KEMET T510 series of surface mount tantalum, available in case heights ranging from 2mm to 4mm. Other capacitor types include Sanyo OS-CON, Nichicon PL series and Panasonic SP. Setting Output Voltage The LTC3801/LTC3801B develop a 0.8V reference voltage between the feedback (Pin 3) terminal and ground (see Figure 4). By selecting resistor R1, a constant current is caused to flow through R1 and R2 to set the overall output voltage. The regulated output voltage is determined by: R2 VOUT = 0.8 1 + R1 For most applications, an 80k resistor is suggested for R1. In applications where low no-load quiescent current is critical, R1 should be made >400k to limit the feedback divider current to approximately 10% of the chip quiescent current. If R2 then results in a very high impedance, it may be beneficial to bypass R2 with a 5pF to 10pF capacitor. To prevent stray pickup, locate resistors R1 and R2 close to LTC3801/LTC3801B. LTC3801/ LTC3801B 3 VFB VOUT R2 R1 3801 F04 Figure 4. Setting Output Voltage Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC3801/LTC3801B circuits: 1) LTC3801/ LTC3801B DC bias current, 2) MOSFET gate charge current, 3) I2R losses and 4) voltage drop of the output diode. 1. The VIN current is the DC supply current, given in the electrical characteristics, that excludes MOSFET driver and control currents. VIN current results in a small loss which increases with VIN. 2. MOSFET gate charge current results from switching the gate capacitance of the power MOSFET. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of VIN which is typically much larger than the DC supply current. In continuous mode, IGATECHG = (f)(dQ). 3. I2R losses are predicted from the DC resistances of the MOSFET, inductor and current shunt. In continuous mode the average output current flows through L but is “chopped” between the P-channel MOSFET (in series with RSENSE) and the output diode. The MOSFET RDS(ON) plus RSENSE multiplied by duty cycle can be summed with the resistances of L and RSENSE to obtain I2R losses. 4. The output diode is a major source of power loss at high currents and gets worse at high input voltages. The diode loss is calculated by multiplying the forward voltage times the diode duty cycle multiplied by the load current. For example, assuming a duty cycle of 50% with a Schottky diode forward voltage drop of 0.4V, the loss increases from 0.5% to 8% as the load current increases from 0.5A to 2A. 5. Transition losses apply to the external MOSFET and increase at higher operating frequencies and input voltages. Transition losses can be estimated from: Transition Loss = 2(VIN)2IO(MAX)CRSS(f) Other losses including CIN and COUT ESR dissipative losses, and inductor core losses, generally account for less than 2% total additional loss. Efficiency = 100% – (η1 + η2 + η3 + ...) where η1, η2, etc. are the individual losses as a percentage of input power. 3801f 10 LTC3801/LTC3801B U W U U APPLICATIO S I FOR ATIO Foldback Current Limiting LTC3801/ LTC3801B As described in the Output Diode Selection, the worstcase dissipation occurs with a short-circuited output when the diode conducts the current limit value almost continuously. To prevent excessive heating in the diode, foldback current limiting can be added to reduce the current in proportion to the severity of the fault. VOUT R2 ITH /RUN VFB + DFB1 R1 DFB2 3801 F05 Figure 5. Foldback Current Limiting Foldback current limiting is implemented by adding diodes DFB1 and DFB2 between the output and the ITH/RUN pin as shown in Figure 5. In a hard short (VOUT = 0V), the current will be reduced to approximately 50% of the maximum output current. U PACKAGE DESCRIPTIO S6 Package 6-Lead Plastic TSOT-23 (Reference LTC DWG # 05-08-1636) 0.62 MAX 2.90 BSC (NOTE 4) 0.95 REF 1.22 REF 3.85 MAX 2.62 REF 1.4 MIN 2.80 BSC 1.50 – 1.75 (NOTE 4) PIN ONE ID RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.30 – 0.45 6 PLCS (NOTE 3) 0.95 BSC 0.80 – 0.90 0.20 BSC 0.01 – 0.10 1.00 MAX DATUM ‘A’ 0.30 – 0.50 REF NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 0.09 – 0.20 (NOTE 3) 1.90 BSC S6 TSOT-23 0302 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193 3801f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 11 LTC3801/LTC3801B U TYPICAL APPLICATIO 550kHz Micropower Step-Down DC/DC Controller VIN ITH/RUN LTC3801/ LTC3801B GND SENSE – VFB 25 VIN 2.7V TO 9.8V 10k 10µF 0.025Ω VIN SUPPLY CURRENT (µA) 220pF LTC3801 No-Load IQ vs Input Voltage* PGATE 402k 4.7µH 866k VOUT 2.5V 2A + 47µF VOUT = 2.5V FRONT PAGE APPLICATION 23 21 19 17 3801 TA01 15 3 4 7 6 5 8 VIN INPUT VOLTAGE (V) 9 10 3801 TA04 *SEE THE FRONT PAGE OF THIS DATA SHEET FOR THE EFFICIENCY vs LOAD CURRENT CURVE RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1147 Series High Efficiency Step-Down Switching Regulator Controllers 100% Duty Cycle, 3.5V ≤ VIN ≤ 16V LTC1622 Low Input Voltage Current Mode Step-Down DC/DC Controller VIN 2V to 10V, IOUT Up to 4.5A, Synchronizable to 750kHz Optional Burst Mode Operation, 8-Lead MSOP LTC1624 High Efficiency SO-8 N-Channel Switching Regulator Controller N-Channel Drive, 3.5V ≤ VIN ≤ 36V LTC1625 No RSENSETM Synchronous Step-Down Regulator 97% Efficiency, No Sense Resistor LTC1702A 550kHz, 2 Phase, Dual Synchronous Controller Two Channels; Minimum CIN and COUT, IOUT up to 15A LTC1733 Li-Ion Linear Battery Charger Standalone Charger with Charge Termination, Integrated MOSFET, Thermal Regulator Prevents Overheating LT®1765 25V, 2.75A (IOUT), 1.25MHz Step-Down Converter 3V ≤ VIN ≤ 25V, VOUT ≥ 1.2V, SO-8 and TSSOP16 Packages LTC1771 Ultra-Low Supply Current Step-Down DC/DC Controller 10µA Supply Current, 93% Efficiency, 1.23V ≤ VOUT ≤ 18V; 2.8V ≤ VIN ≤ 20V LTC1772/LTC1772B 550kHz ThinSOT Step-Down DC/DC Controllers 2.5V ≤ VIN ≤ 9.8V, VOUT ≥ 0.8V, IOUT ≤ 6A LTC1778/LTC1778-1 No RSENSE Current Mode Synchronous Step-Down Controllers 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ (0.9)(VIN), IOUT Up to 20A LTC1779 250mA Monolithic Step-Down Converter in ThinSOT 2.5V ≤ VIN ≤ 9.8V, 550kHz, VOUT ≥ 0.8V LTC1872/LTC1872B 550kHz ThinSOT Step-Up DC/DC Controllers 2.5V ≤ VIN ≤ 9.8V; 90% Efficiency LTC3411/LTC3412 1.25/2.5A Monolithic Synchronous Step-Down Converter 95% Efficiency, 2.5V ≤ VIN ≤ 5.5V, VOUT ≥ 0.8V, TSSOP16 Exposed Pad Package LTC3440 600mA (IOUT), 2MHz Synchronous Buck-Boost DC/DC Converter 2.5V ≤ VIN ≤ 5.5V, Single Inductor No RSENSE is a trademark of Linear Technology Corporation. 3801f 12 Linear Technology Corporation LT/TP 1103 1K • PRINTED IN THE USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LINEAR TECHNOLOGY CORPORATION 2003