LINER LTC3801 Micropower constant frequency step-down dc/dc controllers in thinsot Datasheet

LTC3801/LTC3801B
Micropower
Constant Frequency Step-Down
DC/DC Controllers in ThinSOT
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FEATURES
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DESCRIPTIO
The LTC®3801/LTC3801B are constant frequency current mode step-down DC/DC controllers in a low profile
(1mm max) 6-lead SOT-23 (ThinSOTTM) package. The
parts provide excellent AC and DC load and line regulation with ±1.5% output voltage accuracy. The LTC3801
consumes only 195µA of quiescent current in normal
operation, dropping to 16µA under no-load conditions.
High Efficiency: Up to 94%
Very Low No-Load Quiescent Current:
Only 16µA (LTC3801)
High Output Currents Easily Achieved
Internal Soft-Start
Wide VIN Range: 2.4V to 9.8V
Low Dropout: 100% Duty Cycle
Constant Frequency 550kHz Operation
Burst Mode® Operation for High Efficiency
at Light Loads (LTC3801)
Burst Mode Operation Disabled for Lower Output
Ripple at Light Loads (LTC3801B)
Output Voltage as Low as 0.8V
±1.5% Voltage Reference Accuracy
Current Mode Operation for Excellent Line and Load
Transient Response
Only 6µA Supply Current in Shutdown (LTC3801)
Low Profile (1mm) SOT-23 Package
The LTC3801/LTC3801B incorporate an undervoltage lockout feature that shuts down the device when the input
voltage falls below 2.2V. The LTC3801 automatically
switches into Burst Mode operation at light loads which
enhances efficiency at low output current. In the LTC3801B,
Burst Mode operation is disabled for lower output ripple at
light loads.
To further maximize the life of a battery source, the
external P-channel MOSFET is turned on continuously in
dropout (100% duty cycle). High switching frequency of
550kHz allows the use of a small inductor.
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APPLICATIO S
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1- or 2-Cell Li-Ion Battery-Powered Applications
Wireless Devices
Portable Computers
Distributed Power Systems
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, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a registered trademark of Linear Technology Corporation.
ThinSOT is a trademark of Linear Technology Corporation.
TYPICAL APPLICATIO
LTC3801 Efficiency vs Load Current*
100
95
550kHz Micropower Step-Down DC/DC Controller
VIN
2.7V TO 9.8V
10k
VIN
ITH/RUN
LTC3801/
LTC3801B
GND
SENSE –
VFB
10µF
0.025Ω
VIN = 4.2V
VIN = 3.3V
90
EFFICIENCY (%)
220pF
VOUT = 2.5V
85
VIN = 6.6V
80
75
VIN = 9.8V
70
VIN = 8.4V
65
PGATE
402k
60
866k
4.7µH
+
47µF
VOUT
2.5V
2A
3801 TA01
55
50
0.1
1
10
100
1000
LOAD CURRENT (mA)
10000
3801 TA02
*SEE NO-LOAD IQ vs INPUT VOLTAGE ON THE LAST PAGE OF THIS DATA SHEET
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LTC3801/LTC3801B
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
(Note 1)
Input Supply Voltage (VIN)........................ – 0.3V to 10V
SENSE –, PGATE Voltages ............ – 0.3V to (VIN + 0.3V)
VFB, ITH/RUN Voltages ............................. – 0.3V to 2.4V
PGATE Peak Output Current (<10µs) ........................ 1A
Operating Temperature Range (Note 2) .. – 40°C to 85°C
Junction Temperature (Note 3) ............................ 150°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
TOP VIEW
6 PGATE
ITH/RUN 1
LTC3801ES6
LTC3801BES6
5 VIN
GND 2
4 SENSE –
VFB 3
S6 PART MARKING
S6 PACKAGE
6-LEAD PLASTIC TSOT-23
LTACR
LTAHN
TJMAX = 150°C, θJA = 230°C/W
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● indicates specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise noted. (Note 2)
PARAMETER
Input Voltage Range
Input DC Supply Current
Normal Operation
SLEEP Mode
Shutdown
UVLO
Undervoltage Lockout Threshold
Start-Up Current Source
Shutdown Threshold (at ITH/RUN)
Regulated Feedback Voltage
Feedback Voltage Line Regulation
Feedback Voltage Load Regulation
VFB Input Current
Overvoltage Protect Threshold
Overvoltage Protect Hysteresis
Oscillator Frequency
Normal Operation
Output Short Circuit
Gate Drive Rise Time
Gate Drive Fall Time
Peak Current Sense Voltage
Peak Current Sense Voltage in Burst Mode Operation
Default Soft-Start Time
CONDITIONS
●
Typicals at VIN = 4.2V (Note 4)
2.4V ≤ VIN ≤ 9.8V, VITH/RUN = 1.3V
2.4V ≤ VIN ≤ 9.8V (LTC3801 Only)
2.4V ≤ VIN ≤ 9.8V, VITH/RUN = 0V (LTC3801)
2.4V ≤ VIN ≤ 9.8V, VITH/RUN = 0V (LTC3801B)
VIN < UVLO Threshold
VIN Rising
VIN Falling
VITH/RUN = 0V (LTC3801)
VITH/RUN = 0V (LTC3801B)
VITH/RUN Rising
0°C ≤ TA ≤ 85°C (Note 5)
–40°C ≤ TA ≤ 85°C (Note 5)
2.4V ≤ VIN ≤ 9.8V (Note 5)
ITH/RUN Sinking 5µA (Note 5)
ITH/RUN Sourcing 5µA (Note 5)
(Note 5)
Measured at VFB
VFB = 0.8V
VFB = 0V
CLOAD = 3000pF
CLOAD = 3000pF
Duty Cycle < 40% (Note 6)
LTC3801
LTC3801B
LTC3801 Only
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LTC3801ES6/LTC3801BES6 are guaranteed to meet specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
MIN
2.4
TYP
MAX
9.8
UNITS
V
195
16
6
8
1
1.8
1.7
1
2
0.6
0.800
0.800
0.05
2
2
2
0.880
40
300
30
15
17
2
2.3
2.2
1.5
3.0
0.95
0.812
0.812
µA
µA
µA
µA
µA
V
V
µA
µA
V
V
V
mV/V
mV/µA
mV/µA
nA
V
mV
500
550
210
40
40
650
kHz
kHz
ns
ns
109
95
117
104
26
0.6
125
113
mV
mV
mV
ms
●
●
●
●
0.5
1.0
0.3
0.788
0.780
0.850
●
●
10
0.910
TJ = TA + (PD • θJA°C/W)
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 5: The LTC3801/LTC3801B are tested in a feedback loop that servos
VFB to the output of the error amplifier while maintaining ITH/RUN at the
midpoint of the current limit range.
Note 6: Peak current sense voltage is reduced dependent on duty cycle as
given in Figure 1.
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LTC3801/LTC3801B
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TYPICAL PERFOR A CE CHARACTERISTICS
Input DC Supply Current (Normal)
vs Input Voltage
225
Input DC Supply Current (SLEEP)
vs Input Voltage (LTC3801 Only)
20
TA = 25°C
VITH/RUN = 1.3V
205
16
15
TA = 25°C
LTC3801B
IIN (µA)
14
6
185
12
3
10
2
3
4
5
6
7
8
9
10
2
3
4
5
6
LTC3801
0
7
8
9
10
2
3
4
5
VIN (V)
VIN (V)
3801 G01
6
7
8
9
3801 G03
Shutdown Threshold
vs Temperature
2.2
800
10
VIN (V)
3801 G02
Undervoltage Lockout Threshold
vs Temperature
Regulated Feedback Voltage
vs Temperature
812
VIN = 4.2V
VIN = 4.2V
808
2.0
700
1.8
VIN FALLING
1.6
804
VFB (mV)
VITH/RUN (mV)
VIN RISING
VIN (V)
9
195
175
TA = 25°C
VITH/RUN = 0V
12
IIN (µA)
18
IIN (µA)
215
Input DC Supply Current
(Shutdown) vs Input Voltage
600
800
796
500
1.4
792
1.2
–50
–30
30
–10 10
50
TEMPERATURE (°C)
400
–50
90
70
–30
50
–10 10
30
TEMPERATURE (°C)
70
Regulated Feedback Voltage
vs Input Voltage
600
TA = 25°C
590
80
90
3801 G06
Oscillator Frequency
vs Temperature
808
Oscillator Frequency
vs Input Voltage
560
VIN = 4.2V
580
TA = 25°C
555
570
800
796
560
fOSC (kHz)
fOSC (kHz)
804
VFB (mV)
30
50
–10 10
TEMPERATURE (°C)
3801 G05
3801 G04
812
788
–50 –30
90
550
540
530
550
545
520
792
510
788
2
3
4
5
7
6
VIN (V)
8
9
10
3801 G07
500
–50
540
–30
30
–10 10
50
TEMPERATURE (°C)
70
90
3801 G08
2
3
4
5
7
6
VIN (V)
8
9
10
3801 G09
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LTC3801/LTC3801B
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PI FU CTIO S
SENSE – (Pin 4): Current Sense Pin. An external sense
resistor is connected between this pin and VIN (Pin 5).
ITH/RUN (Pin 1): This pin performs two functions. It
serves as the error amplifier compensation point as well as
the run control input. Nominal voltage range for this pin is
0.7V to 1.9V. Forcing this pin below 0.6V causes the
device to be shut down. In shutdown, all functions are
disabled and the PGATE pin is held high.
VIN (Pin 5): Supply Pin. This pin must be closely decoupled to GND (Pin 2).
PGATE (Pin 6): Gate Drive for the External P-Channel
MOSFET. This pin swings from 0V to VIN.
GND (Pin 2): Ground Pin.
VFB (Pin 3): Receives the feedback voltage from an external resistor divider across the output.
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FU CTIO AL DIAGRA
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4
SENSE –
15mV (LTC3801B)
5
VIN
UNDERVOLTAGE
LOCKOUT
1µA (LTC3801)
2µA (LTC3801B)
1
ITH/RUN
–
UV
VOLTAGE
REFERENCE
0.8V
BURST
DEFEAT
(LTC3801B)
BURST
CLAMP
+
CURRENT
COMPARATOR
SHUTDOWN
COMPARATOR
0.3V
+
+
SLOPE
COMPENSATION
–
ILIM
ITH
BUFFER
SHDN
–
550kHz
OSCILLATOR
RS R S
LATCH
Q
–
BURST
DEFEAT
(LTC3801B)
VIN
SLEEP
COMPARATOR
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
SLEEP
+
FREQUENCY
FOLDBACK
PGATE
6
0V
SOFT-START
CLAMP
OVERVOLTAGE
COMPARATOR
0.15V
–
ERROR
AMPLIFIER
0.225V
+
+
0.3V
0.88V
–
+
1.2V
SHORT-CIRCUIT
DETECT
–
VFB
3
0.8V
GND
2
3801 FD
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LTC3801/LTC3801B
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OPERATIO
(Refer to the Functional Diagram)
Main Control Loop (Normal Operation)
The LTC3801/LTC3801B are constant frequency current
mode step-down switching regulator controllers. During
normal operation, an external P-channel MOSFET is turned
on each cycle when the oscillator sets the RS latch and
turned off when the current comparator resets the latch.
The peak inductor current at which the current comparator
trips is controlled by the voltage on the ITH/RUN pin, which
is the output of the error amplifier. The negative input to
the error amplifier is the output feedback voltage VFB
which is generated by an external resistor divider connected between VOUT and ground. When the load current
increases, it causes a slight decrease in VFB relative to the
0.8V reference, which in turn causes the ITH/RUN voltage
to increase until the average inductor current matches the
new load current.
The main control loop is shut down by pulling the ITH/RUN
pin to ground. Releasing the ITH/RUN pin allows an
internal 1µA current source (2µA on LTC3801B) to charge
up the external compensation network. When the ITH/
RUN pin voltage reaches approximately 0.6V, the main
control loop is enabled and the ITH/RUN voltage is pulled
up by a clamp to its zero current level of approximately
one diode voltage drop (0.7V). As the external compensation network continues to charge up, the corresponding
peak inductor current level follows, allowing normal operation. The maximum peak inductor current attainable is
set by a clamp on the ITH/RUN pin at 1.2V above the zero
current level (approximately 1.9V).
and the load will eventually cause the error amplifier output to start drifting higher. When the error amplifier output
rises to 0.225V above its zero current level (approximately
0.925V), the sleep comparator will untrip and normal operation will resume. The next oscillator cycle will turn the
external MOSFET on and the switching cycle will repeat.
Low Load Current Operation (LTC3801B Only)
Under very light load current conditions, the ITH/RUN pin
voltage will be very close to the zero current level of 0.85V.
As the load current decreases further, an internal offset at
the current comparator input will ensure that the current
comparator remains tripped (even at zero load current)
and the regulator will start to skip cycles, as it must, in
order to maintain regulation. This behavior allows the
regulator to maintain constant frequency down to very
light loads, resulting in less low frequency noise generation over a wide load current range.
Figure 1 illustrates this result for the circuit on the front
page of this data sheet using both an LTC3801 (in Burst
Mode operation) and an LTC3801B (with Burst Mode
operation disabled). At an output current of 100mA, the
LTC3801 exhibits an output ripple of 81.6mVP-P, whereas
the LTC3801B has an output ripple of only 17.6mVP-P. At
lower output current levels, the improvement is even
greater. This comes at a tradeoff of lower efficiency for the
non Burst Mode part at light load currents (see Figure 2).
Also notice the constant frequency operation of the
LTC3801B, even at 5% of maximum output current.
Burst Mode Operation (LTC3801 Only)
Dropout Operation
The LTC3801 incorporates Burst Mode operation at low
load currents (<25% of IMAX). In this mode, an internal
clamp sets the peak current of the inductor at a level corresponding to an ITH/RUN voltage 0.3V above its zero
current level (approximately 1V), even though the actual
ITH/RUN voltage is lower. When the inductor’s average
current is greater than the load requirement, the voltage at
the ITH/RUN pin will drop. When the ITH/RUN voltage falls
to 0.15V above its zero current level (approximately 0.85V),
the sleep comparator will trip, turning off the external
MOSFET. In sleep, the input DC supply current to the IC is
reduced to 16µA from 195µA in normal operation. With the
switch held off, average inductor current will decay to zero
When the input supply voltage decreases towards the
output voltage, the rate of change of inductor current
during the on cycle decreases. This reduction means that
at some input-output differential, the external P-channel
MOSFET will remain on for more than one oscillator cycle
(start dropping off-cycles) since the inductor current has
not ramped up to the threshold set by the error amplifier.
Further reduction in input supply voltage will eventually
cause the external P-channel MOSFET to be turned on
100%, i.e., DC. The output voltage will then be determined
by the input voltage minus the voltage drop across the
sense resistor, the MOSFET and the inductor.
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LTC3801/LTC3801B
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OPERATIO (Refer to the Functional Diagram)
VOUT Ripple for Front Page Circuit Using the LTC3801
(with Burst Mode Operation)
20mVAC/DIV
VOUT Ripple for Front Page Circuit Using the LTC3801B
(Burst Mode Operation Disabled)
20mVAC/DIV
VIN = 4.2V
VOUT= 2.5V
IOUT = 100mA
5µs/DIV
3801 F01a
VIN = 4.2V
VOUT= 2.5V
IOUT = 100mA
5µs/DIV
3801 F01b
Figure 1. Output Ripple Waveforms for the Front Page Circuit
100
95
90
EFFICIENCY (%)
This lower frequency allows the inductor current to safely
discharge, thereby preventing current runaway. After the
short is removed, the oscillator frequency will gradually
increase back to 550kHz as VFB rises through 0.3V on its
way back to 0.8V.
VOUT = 2.5V
VIN = 3.3V
85
80
75
70
65
VIN = 4.2V
VIN = 6.6V
VIN = 9.8V
60
Overvoltage Protection
VIN = 8.4V
55
50
0.1
1
100
1000
10
LOAD CURRENT (mA)
10000
3801 F02
If VFB exceeds its regulation point of 0.8V by more than
10% for any reason, such as an output short circuit to a
higher voltage, the overvoltage comparator will hold the
external P-channel MOSFET off. This comparator has a
typical hysteresis of 40mV.
Figure 2. LTC3801B Efficiency vs Load Current
Slope Compensation and Inductor’s Peak Current
Undervoltage Lockout Protection
To prevent operation of the external P-channel MOSFET
with insufficient gate drive, an undervoltage lockout circuit is incorporated into the LTC3801/LTC3801B. When
the input supply voltage drops below approximately 1.7V,
the P-channel MOSFET and all internal circuitry other than
the undervoltage block itself are turned off. Input supply
current in undervoltage is approximately 1µA.
Short-Circuit Protection
If the output is shorted to ground, the frequency of the
oscillator is folded back from 550kHz to approximately
210kHz while maintaining the same minimum on time.
The switch on duty cycle in normal operation is given by:
Duty Cycle =
VOUT + VD
VIN + VD
where VD is the forward voltage drop of the external diode
at the average inductor current. For duty cycles less than
40%, the inductor’s peak current is determined by:
IMAX =
VITH/RUN – 0.7 V
10 RSENSE
However, for duty cycles greater than 40%, slope compensation begins and effectively reduces the peak
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LTC3801/LTC3801B
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OPERATIO
(Refer to the Functional Diagram)
100
115
90
105
Soft-Start
An internal default soft-start circuit is employed at powerup and/or when coming out of shutdown. The soft-start
circuit works by internally clamping the voltage at the
ITH/RUN pin to the corresponding zero current level and
gradually raising the clamp voltage such that the minimum
time required for the programmed switch current to reach
its maximum is approximately 0.6ms. After the soft-start
circuit has timed out, it is disabled until the part is put in
shutdown again or the input supply is cycled.
TRIP VOLTAGE (mV)
LTC3801
95
80
LTC3801B
85
70
75
60
65
50
55
35
40
VIN = 4.2V
TA = 25°C
45
20
30
40
LTC3801 SLOPE FACTOR (%)
inductor current. The amount of reduction is given by the
curve in Figure 3.
50 60 70 80
DUTY CYCLE (%)
90
30
100
3801 F03
Figure 3. Maximum Current Limit Trip Voltage vs Duty Cycle
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APPLICATIO S I FOR ATIO
The basic LTC3801/LTC3801B application circuit is shown
on the front page of this data sheet. External component
selection is driven by the load requirement and begins with
the selection of the inductor and RSENSE. Next, the power
MOSFET and the output diode are selected followed by the
input bypass capacitor CIN and output bypass capacitor
COUT.
However, for operation that is above 40% duty cycle, slope
compensation effect has to be taken into consideration to
select the appropriate value to provide the required amount
of current. Using Figure 3, the value of RSENSE is:
RSENSE Selection for Output Current
where SF is the “Slope Factor.”
RSENSE is chosen based on the required output current.
With the current comparator monitoring the voltage
developed across RSENSE, the threshold of the comparator determines the inductor’s peak current. The output
current the LTC3801 can provide is given by:
Inductor Value Calculation
IOUT =
0.117 IRIPPLE
−
RSENSE
2
where IRIPPLE is the inductor peak-to-peak ripple current
(see Inductor Value Calculation section). For the LTC3801B
use 104mV in the previous equation and follow through
the analysis using that number.
A reasonable starting point for setting ripple current is
IRIPPLE = (0.4)(IOUT). Rearranging the above equation, it
becomes:
RSENSE =
1
for Duty Cycle < 40%
(10)(IOUT )
RSENSE =
SF
(10)(IOUT )(100)
The operating frequency and inductor selection are interrelated in that higher operating frequencies permit the use
of a smaller inductor for the same amount of inductor
ripple current. However, this is at the expense of efficiency
due to an increase in MOSFET gate charge losses.
The inductance value also has a direct effect on ripple current. The ripple current, IRIPPLE, decreases with higher inductance or frequency and increases with higher VIN or
VOUT. The inductor’s peak-to-peak ripple current is given by:
IRIPPLE =
VIN − VOUT  VOUT + VD 


f(L)  VIN + VD 
where f is the operating frequency. Accepting larger values
of IRIPPLE allows the use of low inductances, but results in
higher output voltage ripple and greater core losses. A
reasonable starting point for setting ripple current is
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LTC3801/LTC3801B
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APPLICATIO S I FOR ATIO
IRIPPLE = 0.4(IOUT(MAX)). Remember, the maximum IRIPPLE
occurs at the maximum input voltage.
In Burst Mode operation on the LTC3801, the ripple
current is normally set such that the inductor current is
continuous during the burst periods. Therefore, the peakto-peak ripple current must not exceed:
IRIPPLE ≤
0.03
RSENSE
This implies a minimum inductance of:
LMIN =
VIN − VOUT  VOUT + VD 


 0.03   VIN + VD 
f

 RSENSE 
(Use VIN(MAX) = VIN)
A smaller value than LMIN could be used in the circuit;
however, the inductor current will not be continuous
during burst periods.
Inductor Core Selection
manufacturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire.
Because they generally lack a bobbin, mounting is more
difficult. However, new designs for surface mount that do
not increase the height significantly are available.
Power MOSFET Selection
An external P-channel power MOSFET must be selected
for use with the LTC3801/LTC3801B. The main selection
criteria for the power MOSFET are the threshold voltage
VGS(TH) and the “on” resistance RDS(ON), reverse transfer
capacitance CRSS and total gate charge.
Since the LTC3801/LTC3801B are designed for operation
down to low input voltages, a sublogic level threshold
MOSFET (RDS(ON) guaranteed at VGS = 2.5V) is required
for applications that work close to this voltage. When
these MOSFETs are used, make sure that the input supply
to the LTC3801/LTC3801B is less than the absolute maximum VGS rating, typically 8V.
The required minimum RDS(ON) of the MOSFET is governed
by its allowable power dissipation. For applications that may
operate the LTC3801/LTC3801B in dropout, i.e., 100% duty
cycle, at its worst case the required RDS(ON) is given by:
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ® cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires
more turns of wire and therefore copper losses will increase. Ferrite designs have very low core losses and are
preferred at high switching frequencies, so design goals
can concentrate on copper loss and preventing saturation.
Ferrite core material saturates “hard,” which means that
inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in
inductor ripple current and consequent output voltage
ripple. Do not allow the core to saturate!
where PP is the allowable power dissipation and δp is the
temperature dependency of RDS(ON). (1 + δp) is generally
given for a MOSFET in the form of a normalized RDS(ON) vs
temperature curve, but δp = 0.005/°C can be used as an
approximation for low voltage MOSFETs.
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive
than ferrite. A reasonable compromise from the same
where DC is the maximum operating duty cycle of the
LTC3801/LTC3801B.
RDS(ON)DC=100% =
PP
(IOUT(MAX) )2 (1+ δp)
In applications where the maximum duty cycle is less than
100% and the LTC3801/LTC3801B are in continuous
mode, the RDS(ON) is governed by:
RDS(ON) ≅
PP
(DC)IOUT2 (1+ δp)
Kool Mµ is a registered trademark of Magnetics, Inc.
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LTC3801/LTC3801B
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APPLICATIO S I FOR ATIO
Output Diode Selection
The catch diode carries load current during the off-time.
The average diode current is therefore dependent on the
P-channel switch duty cycle. At high input voltages the
diode conducts most of the time. As VIN approaches VOUT
the diode conducts only a small fraction of the time. The
most stressful condition for the diode is when the output
is short-circuited. Under this condition the diode must
safely handle IPEAK at close to 100% duty cycle. Therefore,
it is important to adequately specify the diode peak current
and average power dissipation so as not to exceed the
diode ratings.
Under normal load conditions, the average current conducted by the diode is:
V −V 
ID=  IN OUT  I OUT
 VIN + VD 
The allowable forward voltage drop in the diode is calculated from the maximum short-circuit current as:
VF ≈
PD
ISC(MAX)
where PD is the allowable power dissipation and will be
determined by efficiency and/or thermal requirements.
A fast switching diode must also be used to optimize
efficiency. Schottky diodes are a good choice for low
forward drop and fast switching times. Remember to keep
lead length short and observe proper grounding (see
Board Layout Checklist) to avoid ringing and increased
dissipation.
An additional consideration in applications where low noload quiescent current is critical is the reverse leakage
current of the diode at the regulated output voltage. A
leakage greater than several microamperes can represent
a significant percentage of the total input current.
CIN and COUT Selection
In continuous mode, the source current of the P-channel
MOSFET is a square wave of duty cycle (VOUT + VD)/
(VIN + VD). To prevent large voltage transients, a low ESR
input capacitor sized for the maximum RMS current must
be used. The maximum RMS capacitor current is given by:
1/ 2
VOUT ( VIN − VOUT )]
[
CIN Required IRMS ≈ IMAX
VIN
This formula has a maximum value at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations
do not offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet the
size or height requirements in the design. Due to the high
operating frequency of the LTC3801/LTC3801B, ceramic
capacitors can also be used for CIN. Always consult the
manufacturer if there is any question.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering.
The output ripple (∆VOUT) is approximated by:

1 
∆VOUT ≈ IRIPPLE  ESR +


8 fCOUT 
where f is the operating frequency, COUT is the output
capacitance and IRIPPLE is the ripple current in the inductor. The output ripple is highest at maximum input voltage
since ∆IL increases with input voltage.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
product of any aluminum electrolytic at a somewhat
higher price. Once the ESR requirement for COUT has been
met, the RMS current rating generally far exceeds the
IRIPPLE(P-P) requirement.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in
3801f
9
LTC3801/LTC3801B
U
W
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APPLICATIO S I FOR ATIO
surface mount configurations. In the case of tantalum, it is
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS, AVX TPSV and KEMET T510 series of surface mount
tantalum, available in case heights ranging from 2mm to
4mm. Other capacitor types include Sanyo OS-CON,
Nichicon PL series and Panasonic SP.
Setting Output Voltage
The LTC3801/LTC3801B develop a 0.8V reference voltage
between the feedback (Pin 3) terminal and ground (see
Figure 4). By selecting resistor R1, a constant current is
caused to flow through R1 and R2 to set the overall output
voltage. The regulated output voltage is determined by:
 R2 
VOUT = 0.8  1 + 
 R1
For most applications, an 80k resistor is suggested for R1.
In applications where low no-load quiescent current is
critical, R1 should be made >400k to limit the feedback
divider current to approximately 10% of the chip quiescent
current. If R2 then results in a very high impedance, it may
be beneficial to bypass R2 with a 5pF to 10pF capacitor. To
prevent stray pickup, locate resistors R1 and R2 close to
LTC3801/LTC3801B.
LTC3801/
LTC3801B
3
VFB
VOUT
R2
R1
3801 F04
Figure 4. Setting Output Voltage
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC3801/LTC3801B circuits: 1) LTC3801/
LTC3801B DC bias current, 2) MOSFET gate charge current, 3) I2R losses and 4) voltage drop of the output diode.
1. The VIN current is the DC supply current, given in the
electrical characteristics, that excludes MOSFET driver
and control currents. VIN current results in a small loss
which increases with VIN.
2. MOSFET gate charge current results from switching the
gate capacitance of the power MOSFET. Each time a
MOSFET gate is switched from low to high to low again,
a packet of charge dQ moves from VIN to ground. The
resulting dQ/dt is a current out of VIN which is typically
much larger than the DC supply current. In continuous
mode, IGATECHG = (f)(dQ).
3. I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current shunt. In continuous
mode the average output current flows through L but is
“chopped” between the P-channel MOSFET (in series
with RSENSE) and the output diode. The MOSFET RDS(ON)
plus RSENSE multiplied by duty cycle can be summed with
the resistances of L and RSENSE to obtain I2R losses.
4. The output diode is a major source of power loss at high
currents and gets worse at high input voltages. The
diode loss is calculated by multiplying the forward
voltage times the diode duty cycle multiplied by the load
current. For example, assuming a duty cycle of 50%
with a Schottky diode forward voltage drop of 0.4V, the
loss increases from 0.5% to 8% as the load current
increases from 0.5A to 2A.
5. Transition losses apply to the external MOSFET and
increase at higher operating frequencies and input
voltages. Transition losses can be estimated from:
Transition Loss = 2(VIN)2IO(MAX)CRSS(f)
Other losses including CIN and COUT ESR dissipative
losses, and inductor core losses, generally account for
less than 2% total additional loss.
Efficiency = 100% – (η1 + η2 + η3 + ...)
where η1, η2, etc. are the individual losses as a percentage of input power.
3801f
10
LTC3801/LTC3801B
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APPLICATIO S I FOR ATIO
Foldback Current Limiting
LTC3801/
LTC3801B
As described in the Output Diode Selection, the worstcase dissipation occurs with a short-circuited output
when the diode conducts the current limit value almost
continuously. To prevent excessive heating in the diode,
foldback current limiting can be added to reduce the
current in proportion to the severity of the fault.
VOUT
R2
ITH /RUN VFB
+
DFB1
R1
DFB2
3801 F05
Figure 5. Foldback Current Limiting
Foldback current limiting is implemented by adding diodes DFB1 and DFB2 between the output and the ITH/RUN
pin as shown in Figure 5. In a hard short (VOUT = 0V), the
current will be reduced to approximately 50% of the
maximum output current.
U
PACKAGE DESCRIPTIO
S6 Package
6-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1636)
0.62
MAX
2.90 BSC
(NOTE 4)
0.95
REF
1.22 REF
3.85 MAX 2.62 REF
1.4 MIN
2.80 BSC
1.50 – 1.75
(NOTE 4)
PIN ONE ID
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45
6 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.20 BSC
0.01 – 0.10
1.00 MAX
DATUM ‘A’
0.30 – 0.50 REF
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
0.09 – 0.20
(NOTE 3)
1.90 BSC
S6 TSOT-23 0302
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
3801f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
11
LTC3801/LTC3801B
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TYPICAL APPLICATIO
550kHz Micropower Step-Down DC/DC Controller
VIN
ITH/RUN
LTC3801/
LTC3801B
GND
SENSE –
VFB
25
VIN
2.7V TO 9.8V
10k
10µF
0.025Ω
VIN SUPPLY CURRENT (µA)
220pF
LTC3801 No-Load IQ vs Input Voltage*
PGATE
402k
4.7µH
866k
VOUT
2.5V
2A
+
47µF
VOUT = 2.5V
FRONT PAGE APPLICATION
23
21
19
17
3801 TA01
15
3
4
7
6
5
8
VIN INPUT VOLTAGE (V)
9
10
3801 TA04
*SEE THE FRONT PAGE OF THIS DATA SHEET FOR THE EFFICIENCY vs LOAD CURRENT CURVE
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1147 Series
High Efficiency Step-Down Switching Regulator Controllers
100% Duty Cycle, 3.5V ≤ VIN ≤ 16V
LTC1622
Low Input Voltage Current Mode Step-Down DC/DC Controller
VIN 2V to 10V, IOUT Up to 4.5A, Synchronizable to
750kHz Optional Burst Mode Operation, 8-Lead MSOP
LTC1624
High Efficiency SO-8 N-Channel Switching Regulator Controller
N-Channel Drive, 3.5V ≤ VIN ≤ 36V
LTC1625
No RSENSETM Synchronous Step-Down Regulator
97% Efficiency, No Sense Resistor
LTC1702A
550kHz, 2 Phase, Dual Synchronous Controller
Two Channels; Minimum CIN and COUT, IOUT up to 15A
LTC1733
Li-Ion Linear Battery Charger
Standalone Charger with Charge Termination, Integrated
MOSFET, Thermal Regulator Prevents Overheating
LT®1765
25V, 2.75A (IOUT), 1.25MHz Step-Down Converter
3V ≤ VIN ≤ 25V, VOUT ≥ 1.2V, SO-8 and TSSOP16 Packages
LTC1771
Ultra-Low Supply Current Step-Down DC/DC Controller
10µA Supply Current, 93% Efficiency,
1.23V ≤ VOUT ≤ 18V; 2.8V ≤ VIN ≤ 20V
LTC1772/LTC1772B
550kHz ThinSOT Step-Down DC/DC Controllers
2.5V ≤ VIN ≤ 9.8V, VOUT ≥ 0.8V, IOUT ≤ 6A
LTC1778/LTC1778-1
No RSENSE Current Mode Synchronous Step-Down Controllers
4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ (0.9)(VIN), IOUT Up to 20A
LTC1779
250mA Monolithic Step-Down Converter in ThinSOT
2.5V ≤ VIN ≤ 9.8V, 550kHz, VOUT ≥ 0.8V
LTC1872/LTC1872B
550kHz ThinSOT Step-Up DC/DC Controllers
2.5V ≤ VIN ≤ 9.8V; 90% Efficiency
LTC3411/LTC3412
1.25/2.5A Monolithic Synchronous Step-Down Converter
95% Efficiency, 2.5V ≤ VIN ≤ 5.5V, VOUT ≥ 0.8V,
TSSOP16 Exposed Pad Package
LTC3440
600mA (IOUT), 2MHz Synchronous Buck-Boost DC/DC Converter
2.5V ≤ VIN ≤ 5.5V, Single Inductor
No RSENSE is a trademark of Linear Technology Corporation.
3801f
12
Linear Technology Corporation
LT/TP 1103 1K • PRINTED IN THE USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
 LINEAR TECHNOLOGY CORPORATION 2003
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