TI1 LM3402 0.5a constant current buck regulator for driving high power led Datasheet

LM3402, LM3402HV
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SNVS450D – SEPTEMBER 2006 – REVISED FEBRUARY 2010
LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High Power LEDs
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FEATURES
DESCRIPTION
•
•
•
•
•
•
•
The LM3402/02HV are monolithic switching
regulators designed to deliver constant currents to
high power LEDs. Ideal for automotive, industrial, and
general lighting applications, they contain a high-side
N-channel MOSFET switch with a current limit of 735
mA (typical) for step-down (Buck) regulators.
Hysteretic control with controlled on-time coupled with
an external resistor allow the converter output voltage
to adjust as needed to deliver a constant current to
series and series - parallel connected arrays of LEDs
of varying number and type, LED dimming by pulse
width modulation (PWM), broken/open LED
protection, low-power shutdown and thermal
shutdown complete the feature set.
1
2
•
•
•
Integrated 0.5A N-channel MOSFET
VIN Range from 6V to 42V (LM3402)
VIN Range from 6V to 75V (LM3402HV)
500 mA Output Current Over Temperature
Cycle-by-Cycle Current Limit
No Control Loop Compensation Required
Separate PWM Dimming and Low Power
Shutdown
Supports All-ceramic Output Capacitors and
Capacitor-less Outputs
Thermal Shutdown Protection
VSSOP, SO PowerPAD Packages
APPLICATIONS
•
•
•
•
•
LED Driver
Constant Current Source
Automotive Lighting
General Illumination
Industrial Lighting
Typical Application
CB
L1
VIN
VIN
CIN
BOOT
SW
RON
D1
IF
RON
LM3402/02HV
CS
RSNS
DIM
GND
VCC
CF
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2006–2010, Texas Instruments Incorporated
LM3402, LM3402HV
SNVS450D – SEPTEMBER 2006 – REVISED FEBRUARY 2010
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Connection Diagram
1
8
VIN
SW
2
BOOT
7
1
6
2
VCC
3
DIM
RON
4
3
5
GND
4
CS
Figure 1. 8-Lead Plastic VSSOP-8 Package
See Package Number DGK (S-PDSO-G8)
VIN
SW
BOOT
DIM
GND
DAP
VCC
RON
CS
8
7
6
5
Figure 2. 8-Lead Plastic SO PowerPAD-8 Package
See Package Number DDA0008B
PIN DESCRIPTIONS
Pin(s)
Name
1
SW
2
BOOT
3
DIM
4
GND
5
CS
6
Description
Application Information
Switch pin
Connect this pin to the output inductor and Schottky diode.
MOSFET drive bootstrap pin
Connect a 10 nF ceramic capacitor from this pin to SW.
Input for PWM dimming
Connect a logic-level PWM signal to this pin to enable/disable the power FET and
reduce the average light output of the LED array.
Ground pin
Connect this pin to system ground.
Current sense feedback pin
Set the current through the LED array by connecting a resistor from this pin to
ground.
RON
On-time control pin
A resistor connected from this pin to VIN sets the regulator controlled on-time.
7
VCC
Output of the internal 7V linear
regulator
Bypass this pin to ground with a minimum 0.1 µF ceramic capacitor with X5R or
X7R dielectric.
8
VIN
Input voltage pin
Nominal operating input range is 6V to 42V (LM3402) or 6V to 75V (LM3402HV).
DAP
GND
Thermal Pad
Connect to ground. Place 4 to 6 vias from DAP to bottom layer ground plane.
ABSOLUTE MAXIMUM RATINGS(LM3402)
(1)
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors
for availability and specifications.
VALUE / UNIT
VIN to GND
–0.3V to 45V
BOOT to GND
–0.3V to 59V
SW to GND
–1.5V
BOOT to VCC
–0.3V to 45V
BOOT to SW
–0.3V to 14V
VCC to GND
–0.3V to 14V
DIM to GND
–0.3V to 7V
CS to GND
–0.3V to 7V
RON to GND
–0.3V to 7V
Junction Temperature
150°C
Storage Temp. Range
–65°C to 125°C
ESD Rating
(2)
2kV
Soldering Information
Lead Temperature (Soldering, 10sec)
260°C
Infrared/Convection Reflow (15sec)
235°C
(1)
(2)
2
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but specific performance is not ensured. For ensured specifications and the test
conditions, see Electrical Characteristics.
The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.
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OPERATING RATINGS(LM3402)
(1)
VALUE / UNIT
VIN
6V to 42V
−40°C to +125°C
Junction Temperature Range
Thermal Resistance θJA (VSSOP-8 Package)
(2)
200°C/W
Thermal Resistance θJA (SO PowerPAD-8 Package)
(1)
(3)
50°C/W
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but specific performance is not ensured. For ensured specifications and the test
conditions, see Electrical Characteristics.
VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading.
θJA of 50°C/W with DAP soldered to a minimum of 2 square inches of 1 oz. copper on the top or bottom PCB layer.
(2)
(3)
ABSOLUTE MAXIMUM RATINGS(LM3402HV)
(1)
VALUE / UNIT
VIN to GND
−0.3V to 76V
BOOT to GND
−0.3V to 90V
−1.5V
SW to GND
BOOT to VCC
−0.3V to 76V
BOOT to SW
−0.3V to 14V
VCC to GND
−0.3V to 14V
DIM to GND
−0.3V to 7V
CS to GND
−0.3V to 7V
RON to GND
−0.3V to 7V
Junction Temperature
150°C
Storage Temp. Range
−65°C to 125°C
ESD Rating
(2)
2kV
Soldering Information
Lead Temperature (Soldering, 10sec)
260°C
Infrared/Convection Reflow (15sec)
235°C
(1)
(2)
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but specific performance is not ensured. For ensured specifications and the test
conditions, see Electrical Characteristics.
The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.
OPERATING RATINGS(LM3402HV)
(1)
VALUE UNIT
VIN
6V to 75V
Junction Temperature Range
Thermal Resistance θJA (VSSOP-8 Package)
–40°C to +125°C
(2)
200°C/W
Thermal Resistance θJA (SO PowerPAD-8 Package)
(1)
(2)
(3)
(3)
50°C/W
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but specific performance is not ensured. For ensured specifications and the test
conditions, see Electrical Characteristics.
VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading.
θJA of 50°C/W with DAP soldered to a minimum of 2 square inches of 1 oz. copper on the top or bottom PCB layer.
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ELECTRICAL CHARACTERISTICS LM3402
VIN = 24V unless otherwise indicated. Typicals and limits appearing in plain type apply for TA = TJ = +25°C. (1) Limits
appearing in boldface type apply over full Operating Temperature Range. Datasheet min/max specification limits are ensured
by design, test, or statistical analysis.
Symbol
Parameter
Conditions
Min
Typ
Max
Units
SYSTEM PARAMETERS
(1)
tON-1
On-time 1
VIN = 10V, RON = 200 kΩ
2.1
2.75
3.4
µs
tON-2
On-time 2
VIN = 40V, RON = 200 kΩ
490
650
810
ns
Typical specifications represent the most likely parametric norm at 25°C operation.
ELECTRICAL CHARACTERISTICS LM3402HV
Symbol
Parameter
Conditions
Min
Typ
Max
Units
SYSTEM PARAMETERS
tON-1
On-time 1
VIN = 10V, RON = 200 kΩ
2.1
2.75
3.4
µs
tON-2
On-time 2
VIN = 70V, RON = 200 kΩ
290
380
470
ns
ELECTRICAL CHARACTERISTICS LM3402/LM3402HV
Symbol
Parameter
Conditions
Min
Typ
Max
Units
194
200
206
mV
REGULATION AND OVER-VOLTAGE COMPARATORS
VREF-REG
CS Regulation Threshold
CS Decreasing, SW turns on
VREF-0V
CS Over-voltage Threshold
CS Increasing, SW turns off
300
mV
ICS
CS Bias Current
CS = 0V
0.1
µA
VSD-TH
Shutdown Threshold
RON / SD Increasing
VSD-HYS
Shutdown Hysteresis
RON / SD Decreasing
40
mV
Minimum Off-time
CS = 0V
300
ns
SHUTDOWN
0.3
0.7
1.05
V
OFF TIMER
tOFF-MIN
INTERNAL REGULATOR
VCC-REG
VCC Regulated Output
VIN-DO
VIN - VCC Dropout
ICC = 5 mA, 6.0V < VIN < 8.0V
VCC-BP-TH
VCC Bypass Threshold
VIN Increasing
8.8
V
VCC-BP-HYS
VCC Bypass Hysteresis
VIN Decreasing
225
mV
VCC-Z-6
VCC Output Impedance
(0 mA < ICC < 5 mA)
VIN = 6V
55
Ω
VIN = 8V
50
VIN = 24V
0.4
VCC-Z-8
6.6
VCC-Z-24
7
7.4
300
V
mV
VCC-LIM
VCC Current Limit (Note 3)
VIN = 24V, VCC = 0V
16
mA
VCC-UV-TH
VCC Under-voltage Lock-out Threshold
VCC Increasing
5.25
V
VCC-UV-HYS
VCC Under-voltage Lock-out Hysteresis
VCC Decreasing
150
mV
VCC-UV-DLY
VCC Under-voltage Lock-out Filter Delay
100 mV Overdrive
IIN-OP
IIN Operating Current
Non-switching, CS = 0V
600
900
µA
IIN-SD
IIN Shutdown Current
RON / SD = 0V
90
180
µA
735
940
mA
3
µs
CURRENT LIMIT
ILIM
Current Limit Threshold
530
DIM COMPARATOR
VIH
Logic High
DIM Increasing
VIL
Logic Low
DIM Decreasing
IDIM-PU
DIM Pull-up Current
DIM = 1.5V
2.2
V
0.8
75
V
µA
N-MOSFET AND DRIVER
RDS-ON
Buck Switch On Resistance
ISW = 200mA, BOOT-SW = 6.3V
VDR-UVLO
BOOT Under-voltage Lock-out Threshold
BOOT–SW Increasing
4
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1.7
0.7
1.5
Ω
3
4
V
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ELECTRICAL CHARACTERISTICS LM3402/LM3402HV (continued)
Symbol
VDR-HYS
Parameter
Conditions
BOOT Under-voltage Lock-out Hysteresis
BOOT–SW Decreasing
Min
Typ
Max
Units
400
mV
THERMAL SHUTDOWN
TSD
Thermal Shutdown Threshold
165
°C
TSD-HYS
Thermal Shutdown Hysteresis
25
°C
VSSOP-8 Package
200
°C/W
SO PowerPAD-8 Package
50
THERMAL RESISTANCE
θJA
Junction to Ambient
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TYPICAL PERFORMANCE CHARACTERISTICS
6
VREF vs Temperature (VIN = 24V)
VREF vs VIN, LM3402 (TA = 25°C)
Figure 3.
Figure 4.
VREF vs VIN, LM3402HV (TA = 25°C)
Current Limit vs Temperature (VIN = 24V)
Figure 5.
Figure 6.
Current Limit vs VIN, LM3402 (TA = 25°C)
Current Limit vs VIN, LM3402HV (TA = 25°C)
Figure 7.
Figure 8.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
TON vs VIN,
RON = 100 kΩ (TA = 25°C)
TON vs VIN, (TA = 25°C)
Figure 9.
Figure 10.
TON vs VIN, (TA = 25°C)
TON vs RON, LM3402 (TA = 25°C)
Figure 11.
Figure 12.
TON vs RON, LM3402HV (TA = 25°C)
VCC vs VIN (TA = 25°C)
Figure 13.
Figure 14.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
8
VO-MAX vs fSW, LM3402 (TA = 25°C)
VO-MIN vs fSW, LM3402 (TA = 25°C)
Figure 15.
Figure 16.
VO-MAX vs fSW, LM3402HV (TA = 25°C)
VO-MIN vs fSW, LM3402HV
(TA = 25°C)
Figure 17.
Figure 18.
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Block Diagram
7V BIAS
REGULATOR
VIN
VIN
SENSE
VCC
UVLO
BYPASS
SWITCH
0.7V
VCC
THERMAL
SHUTDOWN
+
300 ns MIN
OFF TIMER
Complete
ON TIMER
RON
RON
Complete
5V
BOOT
Start
Start
GATE DRIVE
UVLO
75 PA
DIM
1.5V
0.2V
+
-
SD
VIN
LEVEL
SHIFT
+
-
LOGIC
CS
SW
0.3V
+
-
CURRENT
LIMIT OFF
TIMER
GND
+
-
0.735A
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BUCK
SWITCH
CURRENT
SENSE
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APPLICATION INFORMATION
THEORY OF OPERATION
The LM3402 and LM3402HV are buck regulators with a wide input voltage range, low voltage reference, and a
fast output enable/disable function. These features combine to make them ideal for use as a constant current
source for LEDs with forward currents as high as 500 mA. The controlled on-time (COT) architecture is a
combination of hysteretic mode control and a one-shot on-timer that varies inversely with input voltage.
Hysteretic operation eliminates the need for small-signal control loop compensation. When the converter runs in
continuous conduction mode (CCM) the controlled on-time maintains a constant switching frequency over the
range of input voltage. Fast transient response, PWM dimming, a low power shutdown mode, and simple output
overvoltage protection round out the functions of the LM3402/02HV.
CONTROLLED ON-TIME OVERVIEW
Figure 1 shows the feedback system used to control the current through an array of LEDs. A voltage signal,
VSNS, is created as the LED current flows through the current setting resistor, RSNS, to ground. VSNS is fed back
to the CS pin, where it is compared against a 200 mV reference, VREF. The on-comparator turns on the power
MOSFET when VSNS falls below VREF. The power MOSFET conducts for a controlled on-time, tON, set by an
external resistor, RON, and by the input voltage, VIN. On-time is governed by the following equation:
tON = 1.34 x 10-10 x
RON
VIN
(1)
At the conclusion of tON the power MOSFET turns off for a minimum off-time, tOFF-MIN, of 300 ns. Once tOFF-MIN is
complete the CS comparator compares VSNS and VREF again, waiting to begin the next cycle.
VO
LED 1
IF
LM3402/02HV
LED n
CS
Comparator
One-shot
+
+
-
VF
VSNS
CS
VREF
VF
IF
RSNS
Figure 19. Comparator and One-Shot
The LM3402/02HV regulators should be operated in continuous conduction mode (CCM), where inductor current
stays positive throughout the switching cycle. During steady-state operationin the CCM, the converter maintains
a constant switching frequency, which can be selected using the following equation:
fSW =
VO
1.34 x 10-10 x RON
VO = n x VF + 200 mV
(2)
(3)
VF = forward voltage of each LED, n = number of LEDs in series
10
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AVERAGE LED CURRENT ACCURACY
The COT architecture regulates the valley of ΔVSNS, the AC portion of VSNS. To determine the average LED
current (which is also the average inductor current) the valley inductor current is calculated using the following
expression:
IL-MIN =
0.2 VO x tSNS
L
RSNS
(4)
In this equation tSNS represents the propagation delay of the CS comparator, and is approximately 220 ns. The
average inductor/LED current is equal to IL-MIN plus one-half of the inductor current ripple, ΔiL:
IF = IL = IL-MIN + ΔiL / 2
(5)
Detailed information for the calculation of ΔiL is given in the Design Considerations section.
MAXIMUM OUTPUT VOLTAGE
The 300 ns minimum off-time limits on the maximum duty cycle of the converter, DMAX, and in turn ,the maximum
output voltage VO(MAX) is determined by the following equations:
DMAX =
tON
tON + tOFF-MIN
VO(max) = DMAX x VIN
(6)
The maximum number of LEDs, nMAX, that can be placed in a single series string is governed by VO(MAX) and the
maximum forward voltage of the LEDs used, VF(MAX), using the expression:
VO(max) - 200 mV
nMAX =
VF(MAX)
(7)
At low switching frequency the maximum duty cycle and output voltage are higher, allowing the LM3402/02HV to
regulate output voltages that are nearly equal to input voltage. The following equation relates switching frequency
to maximum output voltage.
VO(MAX) = VIN x
TSW - 300 ns
TSW
TSW = 1/fSW
(8)
MINIMUM OUTPUT VOLTAGE
The minimum recommended on-time for the LM3402/02HV is 300 ns. This lower limit for tON determines the
minimum duty cycle and output voltage that can be regulated based on input voltage and switching frequency.
The relationship is determined by the following equation:
VO(MIN) = VIN x
300 ns
TSW
(9)
HIGH VOLTAGE BIAS REGULATOR
The LM3402/02HV contains an internal linear regulator with a 7V output, connected between the VIN and the
VCC pins. The VCC pin should be bypassed to the GND pin with a 0.1 µF ceramic capacitor connected as close
as possible to the pins of the IC. VCC tracks VIN until VIN reaches 8.8V (typical) and then regulates at 7V as
VIN increases. Operation begins when VCC crosses 5.25V.
INTERNAL MOSFET AND DRIVER
The LM3402/02HV features an internal power MOSFET as well as a floating driver connected from the SW pin to
the BOOT pin. Both rise time and fall time are 20 ns each (typical) and the approximate gate charge is 3 nC. The
high-side rail for the driver circuitry uses a bootstrap circuit consisting of an internal high-voltage diode and an
external 10 nF capacitor, CB. VCC charges CB through the internal diode while the power MOSFET is off. When
the MOSFET turns on, the internal diode reverse biases. This creates a floating supply equal to the VCC voltage
minus the diode drop to drive the MOSFET when its source voltage is equal to VIN.
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FAST SHUTDOWN FOR PWM DIMMING
The DIM pin of the LM3402/02HV is a TTL logic compatible input for low frequency PWM dimming of the LED. A
logic low (below 0.8V) at DIM will disable the internal MOSFET and shut off the current flow to the LED array.
While the DIM pin is in a logic low state the support circuitry (driver, bandgap, VCC) remains active in order to
minimize the time needed to turn the LED array back on when the DIM pin sees a logic high (above 2.2V). A 75
µA (typical) pull-up current ensures that the LM3402/02HV is on when DIM pin is open circuited, eliminating the
need for a pull-up resistor. Dimming frequency, fDIM, and duty cycle, DDIM, are limited by the LED current rise time
and fall time and the delay from activation of the DIM pin to the response of the internal power MOSFET. In
general, fDIM should be at least one order of magnitude lower than the steady state switching frequency in order
to prevent aliasing.
PEAK CURRENT LIMIT
The current limit comparator of the LM3402/02HV will engage whenever the power MOSFET current (equal to
the inductor current while the MOSFET is on) exceeds 735 mA (typical). The power MOSFET is disabled for a
cool-down time that is 10x the steady-state on-time. At the conclusion of this cool-down time the system re-starts.
If the current limit condition persists the cycle of cool-down time and restarting will continue, creating a low-power
hiccup mode, minimizing thermal stress on the LM3402/02HV and the external circuit components.
OVER-VOLTAGE/OVER-CURRENT COMPARATOR
The CS pin includes an output over-voltage/over-current comparator that will disable the power MOSFET
whenever VSNS exceeds 300 mV. This threshold provides a hard limit for the output current. Output current
overshoot is limited to 300 mV / RSNS by this comparator during transients.
The OVP/OCP comparator can also be used to prevent the output voltage from rising to VO(MAX) in the event of
an output open-circuit. This is the most common failure mode for LEDs, due to breaking of the bond wires. In a
current regulator an output open circuit causes VSNS to fall to zero, commanding maximum duty cycle. Figure 2
shows a method using a zener diode, Z1, and zener limiting resistor, RZ, to limit output voltage to the reverse
breakdown voltage of Z1 plus 200 mV. The zener diode reverse breakdown voltage, VZ, must be greater than the
maximum combined VF of all LEDs in the array. The maximum recommended value for RZ is 1 kΩ.
As discussed in the Maximum Output Voltage section, there is a limit to how high VO can rise during an output
open-circuit that is always less than VIN. If no output capacitor is used, the output stage of the LM3402/02HV is
capable of withstanding VO(MAX) indefinitely, however the voltage at the output end of the inductor will oscillate
and can go above VIN or below 0V. A small (typically 10 nF) capacitor across the LED array dampens this
oscillation. For circuits that use an output capacitor, the system can still withstand VO(MAX) indefinitely as long as
CO is rated to handle VIN. The high current paths are blocked in output open-circuit and the risk of thermal stress
is minimal, hence the user may opt to allow the output voltage to rise in the case of an open-circuit LED failure.
CB
VIN
VIN
CIN
BOOT
L1
SW
RON
D1
Z1
RON
LM3402/02HV
RZ
CS
RSNS
DIM
GND
VCC
CF
Figure 20. Output Open Circuit Protection
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LOW POWER SHUTDOWN
The LM3402/02HV can be switched to a low power state (IIN-SD = 90 µA) by grounding the RON pin with a signallevel MOSFET as shown in Figure 3. Low power MOSFETs like the 2N7000, 2N3904, or equivalent are
recommended devices for putting the LM3402/02HV into low power shutdown. Logic gates can also be used to
shut down the LM3402/02HV as long as the logic low voltage is below the over temperature minimum threshold
of 0.3V. Noise filter circuitry on the RON pin can cause a few pulses with a longer on-time than normal after RON
is grounded or released. In these cases the OVP/OCP comparator will ensure that the peak inductor or LED
current does not exceed 300 mV / RSNS.
CB
L1
VIN
VIN
BOOT
SW
RON
CIN
D1
RON
IF
LM3402/02HV
ON/OFF
CS
Q1
2N7000 or
equivalent
RSNS
DIM
GND
VCC
CF
Figure 21. Low Power Shutdown
THERMAL SHUTDOWN
Internal thermal shutdown circuitry is provided to protect the IC in the event that the maximum junction
temperature is exceeded. The threshold for thermal shutdown is 165°C with a 25°C hysteresis (both values
typical). During thermal shutdown the MOSFET and driver are disabled.
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DESIGN CONSIDERATIONS
SWITCHING FREQUENCY
Switching frequency is selected based on the tradeoffs between efficiency (better at low frequency), solution
size/cost (smaller at high frequency), and the range of output voltage that can be regulated (wider at lower
frequency.) Many applications place limits on switching frequency due to EMI sensitivity. The on-time of the
LM3402/02HV can be programmed for switching frequencies ranging from the 10’s of kHz to over 1 MHz. The
maximum switching frequency is limited only by the minimum on-time requirement.
LED RIPPLE CURRENT
Selection of the ripple current, ΔiF, through the LED array is analogous to the selection of output ripple voltage in
a standard voltage regulator. Where the output ripple in a voltage regulator is commonly ±1% to ±5% of the DC
output voltage, LED manufacturers generally recommend values for ΔiF ranging from ±5% to ±20% of IF. Higher
LED ripple current allows the use of smaller inductors, smaller output capacitors, or no output capacitors at all.
The advantages of higher ripple current are reduction in the solution size and cost. Lower ripple current requires
more output inductance, higher switching frequency, or additional output capacitance. The advantages of lower
ripple current are a reduction in heating in the LED itself and greater range of the average LED current before
the current limit of the LED or the driving circuitry is reached.
BUCK CONVERTERS WITHOUT OUTPUT CAPACITORS
The buck converter is unique among non-isolated topologies because of the direct connection of the inductor to
the load during the entire switching cycle. By definition an inductor will control the rate of change of current that
flows through it, and this control over current ripple forms the basis for component selection in both voltage
regulators and current regulators. A current regulator such as the LED driver for which the LM3402/02HV was
designed focuses on the control of the current through the load, not the voltage across it. A constant current
regulator is free of load current transients, and has no need of output capacitance to supply the load and
maintain output voltage. Referring to the Typical Application circuit on the front page of this datasheet, the
inductor and LED can form a single series chain, sharing the same current. When no output capacitor is used,
the same equations that govern inductor ripple current, ΔiL, also apply to the LED ripple current, ΔiF. For a
controlled on-time converter such as LM3402/02HV the ripple current is described by the following expression:
'iL = 'iF =
VIN - VO
L
tON
(10)
A minimum ripple voltage of 25 mV is recommended at the CS pin to provide good signal-to-noise ratio (SNR).
The CS pin ripple voltage, ΔVSNS, is described by the following:
ΔVSNS = ΔiF x RSNS
(11)
BUCK CONVERTERS WITH OUTPUT CAPACITORS
A capacitor placed in parallel with the LED or array of LEDs can be used to reduce the LED current ripple while
keeping the same average current through both the inductor and the LED array. This technique is demonstrated
in Design Example 1. With this topology the output inductance can be lowered, making the magnetics smaller
and less expensive. Alternatively, the circuit could be run at lower frequency but keep the same inductor value,
improving the efficiency and expanding the range of output voltage that can be regulated. Both the peak current
limit and the OVP/OCP comparator still monitor peak inductor current, placing a limit on how large ΔiL can be
even if ΔiF is made very small. A parallel output capacitor is also useful in applications where the inductor or
input voltage tolerance is poor. Adding a capacitor that reduces ΔiF to well below the target provides headroom
for changes in inductance or VIN that might otherwise push the peak LED ripple current too high.
Figure 4 shows the equivalent impedances presented to the inductor current ripple when an output capacitor, CO,
and its equivalent series resistance (ESR) are placed in parallel with the LED array. The entire inductor ripple
current flows through RSNS to provide the required 25 mV of ripple voltage for proper operation of the CS
comparator.
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'iL
CO
rD
'iC
'iF
ESR
'iL
RSNS
Figure 22. LED and CO Ripple Current
To calculate the respective ripple currents the LED array is represented as a dynamic resistance, rD. LED
dynamic resistance is not always specified on the manufacturer’s datasheet, but it can be calculated as the
inverse slope of the LED’s VF vs. IF curve. Note that dividing VF by IF will give an incorrect value that is 5x to 10x
too high. Total dynamic resistance for a string of n LEDs connected in series can be calculated as the rD of one
device multiplied by n. Inductor ripple current is still calculated with the expression from Buck Regulators without
Output Capacitors. The following equations can then be used to estimate ΔiF when using a parallel capacitor:
'iF =
'iL
1+
rD
ZC
ZC = ESR +
1
2S x fSW x CO
(12)
The calculation for ZC assumes that the shape of the inductor ripple current is approximately sinusoidal.
Small values of CO that do not significantly reduce ΔiF can also be used to control EMI generated by the
switching action of the LM3402/02HV. EMI reduction becomes more important as the length of the connections
between the LED and the rest of the circuit increase.
INPUT CAPACITORS
Input capacitors at the VIN pin of the LM3402/02HV are selected using requirements for minimum capacitance
and rms ripple current. The input capacitors supply pulses of current approximately equal to IF while the power
MOSFET is on, and are charged up by the input voltage while the power MOSFET is off. Switching converters
such as the LM3402/02HV have a negative input impedance due to the decrease in input current as input
voltage increases. This inverse proportionality of input current to input voltage can cause oscillations (sometimes
called ‘power supply interaction’) if the magnitude of the negative input impedance is greater the the input filter
impedance. Minimum capacitance can be selected by comparing the input impedance to the converter’s negative
resistance; however this requires accurate calculation of the input voltage source inductance and resistance,
quantities which can be difficult to determine. An alternative method to select the minimum input capacitance,
CIN(MIN), is to select the maximum voltage ripple which can be tolerated. This value,ΔvIN(MAX), is equal to the
change in voltage across CIN during the converter on-time, when CIN supplies the load current. CIN(MIN) can be
selected with the following:
CIN(MIN) =
IF x tON
'VIN(MAX)
(13)
A good starting point for selection of CIN is to use an input voltage ripple of 5% to 10% of VIN. A minimum input
capacitance of 2x the CIN(MIN) value is recommended for all LM3402/02HV circuits. To determine the rms current
rating, the following formula can be used:
IIN(rms) = IF x D(1 - D)
(14)
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Ceramic capacitors are the best choice for the input to the LM3402/02HV due to their high ripple current rating,
low ESR, low cost, and small size compared to other types. When selecting a ceramic capacitor, special
attention must be paid to the operating conditions of the application. Ceramic capacitors can lose one-half or
more of their capacitance at their rated DC voltage bias and also lose capacitance with extremes in temperature.
A DC voltage rating equal to twice the expected maximum input voltage is recommended. In addition, the
minimum quality dielectric which is suitable for switching power supply inputs is X5R, while X7R or better is
preferred.
RECIRCULATING DIODE
The LM3402/02HV is a non-synchronous buck regulator that requires a recirculating diode D1 (see the Typical
Application circuit) to carrying the inductor current during the MOSFET off-time. The most efficient choice for D1
is a Schottky diode due to low forward drop and near-zero reverse recovery time. D1 must be rated to handle the
maximum input voltage plus any switching node ringing when the MOSFET is on. In practice all switching
converters have some ringing at the switching node due to the diode parasitic capacitance and the lead
inductance. D1 must also be rated to handle the average current, ID, calculated as:
ID = (1 – D) x IF
(15)
This calculation should be done at the maximum expected input voltage. The overall converter efficiency
becomes more dependent on the selection of D1 at low duty cycles, where the recirculating diode carries the
load current for an increasing percentage of the time. This power dissipation can be calculated by checking the
typical diode forward voltage, VD, from the I-V curve on the product datasheet and then multiplying it by ID. Diode
datasheets will also provide a typical junction-to-ambient thermal resistance, θJA, which can be used to estimate
the operating die temperature of the Schottky. Multiplying the power dissipation (PD = ID x VD) by θJA gives the
temperature rise. The diode case size can then be selected to maintain the Schottky diode temperature below
the operational maximum.
LED CURRENT DURING DIM MODE
The LM3402 contains high speed MOSFET gate drive circuitry that switches the main internal power MOSFET
between “on” and “off” states. This circuitry uses current derived from the VCC regulator to charge the MOSFET
during turn-on, then dumps current from the MOSFET gate to the source (the SW pin) during turn-off. As shown
in the block diagram, the MOSFET drive circuitry contains a gate drive under-voltage lockout (UVLO) circuit that
ensures the MOSFET remains off when there is inadequate VCC voltage for proper operation of the driver. This
watchdog circuitry is always running including during DIM and shutdown modes, and supplies a small amount of
current from VCC to SW. Because the SW pin is connected directly to the LEDs through the buck inductor, this
current returns to ground through the LEDs. The amount of current sourced is a function of the SW voltage, as
shown in Figure 23.
25
SW CURRENT (PA)
20
15
10
5
0
0
1
2
3
4
5
6
SW VOLTAGE (V)
Figure 23. LED Current From SW Pin
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Though most power LEDs are designed to run at several hundred milliamps, some can be seen to glow with a
faint light at extremely low current levels, as low as a couple microamps in some instances. In lab testing, the
forward voltage was found to be approximately 2V for LEDs that exhibited visible light at these low current levels.
For LEDs that did not show light emission at very low current levels, the forward voltage was found to be around
900mV. It is important to remember that the forward voltage is also temperature dependent, decreasing at higher
temperatures. Consequently, with a maximum Vcc voltage of 7.4V, current will be observed in the LEDs if the
total stack voltage is less than about 6V at a forward current of several microamps. No current is observed if the
stack voltage is above 6V, as shown in Figure 23. The need for absolute darkness during DIM mode is also
application dependent. It will not affect regular PWM dimming operation.
The fix for this issue is extremely simple. Place a resistor from the SW pin to ground according to the chart
below.
Number of LEDs
Resistor Value (kΩ)
1
20
2
50
3
90
4
150
5
200
>5
300
The luminaire designer should ensure that the suggested resistor is effective in eliminating the off-state light
output. A combination of calculations based on LED manufacturer data and lab measurements over temperature
will ensure the best design.
Transient Protection Considerations
Considerations need to be made when external sources, loads or connections are made to the switching
converter circuit due to the possibility of Electrostatic Discharge (ESD) or Electric Over Stress (EOS) events
occurring and damaging the integrated circuit (IC) device. All IC device pins contain zener based clamping
structures that are meant to clamp ESD. ESD events are very low energy events, typically less than 5µJ
(microjoules). Any event that transfers more energy than this may damage the ESD structure. Damage is
typically represented as a short from the pin to ground as the extreme localized heat of the ESD / EOS event
causes the aluminum metal on the chip to melt, causing the short. This situation is common to all integrated
circuits and not just unique to the LM340X device.
CS PIN PROTECTION
When hot swapping in a load (e.g. test points, load boards, LED stack), any residual charge on the load will be
immediately transferred through the output capacitor to the CS pin, which is then damaged as shown in
Figure 24 below. The EOS event due to the residual charge from the load is represented as VTRANSIENT.
From measurements, we know that the 8V ESD structure on the CS pin can typically withstand 25mA of direct
current (DC). Adding a 1kΩ resistor in series with the CS pin, shown in Figure 25, results in the majority of the
transient energy to pass through the discrete sense resistor rather than the device. The series resistor limits the
peak current that can flow during a transient event, thus protecting the CS pin. With the 1kΩ resistor shown, a
33V, 49A transient on the LED return connector terminal could be absorbed as calculated by:
V = 25mA * 1kΩ + 8V = 33V
I = 33V / 0.67Ω = 49A
(16)
(17)
This is an extremely high energy event, so the protection measures previously described should be adequate to
solve this issue.
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LM3402
SW
Module
Connector
Module
Connector
VTRANSIENT
CS
8V
~ 0.675
GND
Figure 24. CS Pin, Transient Path
LM3402
SW
Module
Connector
Module
Connector
CS
VTRANSIENT
1 k5
8V
~ 0.675
GND
Figure 25. CS Pin, Transient Path with Protection
Adding a resistor in series with the CS pin causes the observed output LED current to shift very slightly. The
reason for this is twofold: (1) the CS pin has about 20pF of inherent capacitance inside it which causes a slight
delay (20ns for a 1kΩ series resistor), and (2) the comparator that is watching the voltage at the CS pin uses a
pnp bipolar transistor at its input. The base current of this pnp transistor is approximately 100nA which will cause
a 0.1mV change in the 200mV threshold. These are both very minor changes and are well understood. The shift
in current can either be neglected or taken into consideration by changing the current sense resistance slightly.
CS PIN PROTECTION WITH OVP
When designing output overvoltage protection into the switching converter circuit using a zener diode, transient
protection on the CS pin requires additional consideration. As shown in Figure 26, adding a zener diode from the
output to the CS pin (with the series resistor) for output overvoltage protection will now again allow the transient
energy to be passed through the CS pin’s ESD structure thereby damaging it.
Adding an additional series resistor to the CS pin as shown in Figure 27 will result in the majority of the transient
energy to pass through the sense resistor thereby protecting the LM340X device.
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LM3402
SW
Module
Connector
Module
Connector
VTRANSIENT
CS
1 k5
8V
~ 0.675
GND
Figure 26. CS Pin with OVP, Transient Path
LM3402
SW
Module
Connector
Module
Connector
VTRANSIENT
CS
1 k5
5005
8V
~ 0.675
GND
Figure 27. CS Pin with OVP, Transient Path with Protection
VIN PIN PROTECTION
The VIN pin also has an ESD structure from the pin to GND with a breakdown voltage of approximately 80V. Any
transient that exceeds this voltage may damage the device. Although transient absorption is usually present at
the front end of a switching converter circuit, damage to the VIN pin can still occur.
When VIN is hot swapped in, the current that rushes in to charge CIN up to the VIN value also charges (energizes)
the circuit board trace inductance as shown in Figure 28. The excited trace inductance then resonates with the
input capacitance (similar to an under-damped LC tank circuit) and causes voltages at the VIN pin to rise well in
excess of both VIN and the voltage at the module input connector as clamped by the input TVS. If the resonating
voltage at the VIN pin exceeds the 80V breakdown voltage of the ESD structure, the ESD structure will activate
and then “snap-back” to a lower voltage due to its inherent design. If this lower snap-back voltage is less than
the applied nominal VIN voltage, then significant current will flow through the ESD structure resulting in the IC
being damaged.
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An additional TVS or small zener diode should be placed as close as possible to the VIN pins of each IC on the
board, in parallel with the input capacitor as shown in Figure 29. A minor amount of series resistance in the input
line would also help, but would lower overall conversion efficiency. For this reason, NTC resistors are often used
as inrush limiters instead.
LM3402
Board Trace
Inductance
VIN
Module
Connector
80V
VIN
TVS
CIN
GND
Module
Connector
Figure 28. VIN Pin with Typical Input Protection
LM3402
Board Trace
Inductance
VIN
Module
Connector
80V
VIN
TVS
CIN
TVS or
smaller
zener diode
GND
Module
Connector
Figure 29. VIN Pin with Additional Input Protection
GENERAL COMMENTS REGARDING OTHER PINS
Any pin that goes “off-board” through a connector should have series resistance of at least 1kΩ to 10kΩ in series
with it to protect it from ESD or other transients. These series resistors limit the peak current that can flow (or
cause a voltage drop) during a transient event, thus protecting the pin and the device. Pins that are not used
should not be left floating. They should instead be tied to GND or to an appropriate voltage through resistance.
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Design Example 1: LM3402
The first example circuit will guide the user through component selection for an architectural accent lighting
application. A regulated DC voltage input of 24V ±10% will power a single 1W white LED at a forward current of
350 mA ±5%. The typical forward voltage of a 1W InGaN LED is 3.5V, hence the estimated average output
voltage will be 3.7V. The objective of this application is to place the complete current regulator and LED in the
compact space formerly occupied by an MR16 halogen light bulb. (The LED will be on a separate metal-core
PCB.) Switching frequency will be as fast as the 300 ns tON limit allows, with the emphasis on space savings over
efficiency. Efficiency cannot be ignored, however, as the confined space with little air-flow requires a maximum
temperature rise of 40°C in each circuit component. A complete bill of materials can be found in Table 1 at the
end of this datasheet.
CB
L1
VIN = 24V
VIN
BOOT
SW
RON
CIN
D1
RON
CO
LED1
IF = 350 mA
LM3402/02HV
CS
DIM
RSNS
VCC
GND
CF
Figure 30. Schematic for Design Example 1
RON and tON
To select RON the expression relating tON to input voltage from the Controlled On-time Overview section can be
re-written as:
RON =
tON x VIN
1.34 x 10-10
(18)
Minimum on-time occurs at the maximum VIN, which is 24V x 110% = 26.4V. RON is therefore calculated as:
RON = (300 x 10-9 x 26.4) / 1.34 x 10-10 = 59105 Ω
(19)
The closest 1% tolerance resistor is 59.0 kΩ. The switching frequency of the circuit can then be found using the
equation relating RON to fSW:
fSW = 3.7 / (59000 x 1.34 x 10-10) = 468 kHz
(20)
USING AN OUTPUT CAPACITOR
The inductor will be the largest component used in this design. Because the application does not require any
PWM dimming, an output capacitor can be used to greatly reduce the inductance needed without worry of
slowing the potential PWM dimming frequency. The total solution size will be reduced by using an output
capacitor and small inductor as opposed to one large inductor.
OUTPUT INDUCTOR
Knowing that an output capacitor will be used, the inductor can be selected for a larger current ripple. The
desired maximum value for ΔiL is ±30%, or 0.6 x 350 mA = 210 mAP-P. Minimum inductance is selected at the
maximum input voltage. Re-arranging the equation for current ripple selection yields the following:
LMIN =
VIN(MAX) - VO
'iL
x tON
(21)
(22)
LMIN = [(26.4 – 3.7) x 300 x 10-9] / (0.6 x 0.35) = 32.4 µH
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The closest standard inductor value is 33 µH. Off-the-shelf inductors rated at 33 µH are available from many
magnetics manufacturers.
Inductor datasheets should contain three specifications which are used to select the inductor. The first of these is
the average current rating, which for a buck regulator is equal to the average load current, or IF. The average
current rating is given by a specified temperature rise in the inductor, normally 40°C. For this example, the
average current rating should be greater than 350 mA to ensure that heat from the inductor does not reduce the
lifetime of the LED or cause the LM3402 to enter thermal shutdown.
The second specification is the tolerance of the inductance itself, typically ±10% to ±30% of the rated inductance.
In this example an inductor with a tolerance of ±20% will be used. With this tolerance the typical, minimum, and
maximum inductor current ripples can be calculated:
ΔiL(TYP) = [(26.4 – 3.7) x 300 x 10-9] / 33 x 10-6 = 206 mAP-P
ΔiL(MIN) = [(26.4 – 3.7) x 300 x 10-9] / 39.6 x 10-6 = 172 mAP-P
ΔiL(MAX) = [(26.4 – 3.7) x 300 x 10-9] / 26.4 x 10-6 = 258 mAP-P
(23)
(24)
(25)
The third specification for an inductor is the peak current rating, normally given as the point at which the
inductance drops off by a given percentage due to saturation of the core. The worst-case peak current occurs at
maximum input voltage and at minimum inductance, and can be determined with the equation from the Design
Considerations section:
IL(PEAK) = IF +
'iL(MAX)
2
(26)
(27)
IL(PEAK) = 0.35 + 0.258 / 2 = 479 mA
For this example the peak current rating of the inductor should be greater than 479 mA. In the case of a short
circuit across the LED array, the LM3402 will continue to deliver rated current through the short but will reduce
the output voltage to equal the CS pin voltage of 200 mV. Worst-case peak current in this condition is equal to:
ΔiL(LED-SHORT) = [(26.4 – 0.2) x 300 x 10-9] / 26.4 x 10-6 = 298 mAP-P IL(PEAK) = 0.35 + 0.149 = 499 mA
(28)
In the case of a short at the switch node, the output, or from the CS pin to ground the short circuit current limit
will engage at a typical peak current of 735 mA. In order to prevent inductor saturation during these short circuits
the inductor’s peak current rating must be above 735 mA. The device selected is an off-the-shelf inductor rated
33 µH ±20% with a DCR of 96 mΩ and a peak current rating of 0.82A. The physical dimensions of this inductor
are 7.0 x 7.0 x 4.5 mm.
RSNS
The current sensing resistor value can be determined by re-arranging the expression for average LED current
from the LED Current Accuracy section:
RSNS =
0.2 x L
IF x L + VO x tSNS - VIN - VO x tON
2
(29)
(30)
RSNS = 0.74Ω, tSNS = 220 ns
Sub-1Ω resistors are available in both 1% and 5% tolerance. A 1%, 0.75Ω resistor will give the best accuracy of
the average LED current. To determine the resistor size the power dissipation can be calculated as:
PSNS = (IF)2 x RSNS PSNS = 0.352 x 0.75 = 92 mW
(31)
Standard 0805 size resistors are rated to 125 mW and will be suitable for this application.
To select the proper output capacitor the equation from Buck Regulators with Output Capacitors is re-arranged to
yield the following:
ZC =
'iF
'iL - 'iF
x rD
(32)
The target tolerance for LED ripple current is ±5% or 10%P-P = 35 mAP-P, and the LED datasheet gives a typical
value for rD of 1.0Ω at 350 mA. The required capacitor impedance to reduce the worst-case inductor ripple
current of 258 mAP-P is therefore:
ZC = [0.035 / (0.258 - 0.035] x 1.0 = 0.157Ω
(33)
A ceramic capacitor will be used and the required capacitance is selected based on the impedance at 468 kHz:
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CO = 1/(2 x π x 0.157 x 4.68 x 105) = 2.18 µF
(34)
This calculation assumes that impedance due to the equivalent series resistance (ESR) and equivalent series
inductance (ESL) of CO is negligible. The closest 10% tolerance capacitor value is 2.2 µF. The capacitor used
should be rated to 10V or more and have an X7R dielectric. Several manufacturers produce ceramic capacitors
with these specifications in the 0805 case size. A typical value for ESR of 1 mΩ can be read from the curve of
impedance vs. frequency in the product datasheet.
INPUT CAPACITOR
Following the calculations from the Input Capacitor section, ΔvIN(MAX) will be 1%P-P = 240 mV. The minimum
required capacitance is:
CIN(MIN) = (0.35 x 300 x 10-9) / 0.24 = 438 nF
(35)
In expectation that more capacitance will be needed to prevent power supply interaction a 1.0 µF ceramic
capacitor rated to 50V with X7R dielectric in a 1206 case size will be used. From the Design Considerations
section, input rms current is:
IIN-RMS = 0.35 x Sqrt(0.154 x 0.846) = 126 mA
(36)
Ripple current ratings for 1206 size ceramic capacitors are typically higher than 1A, more than enough for this
design.
RECIRCULATING DIODE
The first parameter for D1 which must be determined is the reverse voltage rating. Schottky diodes are available
at reverse ratings of 30V and 40V, often in the same package, with the same forward current rating. To account
for ringing a 40V Schottky will be used.
The next parameters to be determined are the forward current rating and case size. In this example the low duty
cycle (D = 3.7 / 24 = 15%) requires the recirculating diode D1 to carry the load current much longer than the
internal power MOSFET of the LM3402. The estimated average diode current is:
ID = 0.35 x 0.85 = 298 mA
(37)
Schottky diodes are available at forward current ratings of 0.5A, however the current rating often assumes a
25°C ambient temperature and does not take into account the application restrictions on temperature rise. A
diode rated for higher current may be needed to keep the temperature rise below 40°C.To determine the proper
case size, the dissipation and temperature rise in D1 can be calculated as shown in the Design Considerations
section. VD for a small case size such as SOD-123 in a 40V, 0.5A Schottky diode at 350 mA is approximately
0.4V and the θJA is 206°C/W. Power dissipation and temperature rise can be calculated as:
PD = 0.298 x 0.4 = 119 mW TRISE = 0.119 x 206 = 24.5°C
(38)
According to these calculations the SOD-123 diode will meet the requirements. Heating and dissipation are
among the factors most difficult to predict in converter design. If possible, a footprint should be used that is
capable of accepting both SOD-123 and a larger case size, such as SMA. A larger diode with a higher forward
current rating will generally have a lower forward voltage, reducing dissipation, as well as having a lower θJA,
reducing temperature rise.
CB and CF
The bootstrap capacitor CB should always be a 10 nF ceramic capacitor with X7R dielectric. A 25V rating is
appropriate for all application circuits. The linear regulator filter capacitor CF should always be a 100 nF ceramic
capacitor, also with X7R dielectric and a 25V rating.
EFFICIENCY
To estimate the electrical efficiency of this example the power dissipation in each current carrying element can
be calculated and summed. This term should not be confused with the optical efficacy of the circuit, which
depends upon the LEDs themselves.
Total output power, PO, is calculated as:
PO = IF x VO = 0.35 x 3.7 = 1.295W
(39)
Conduction loss, PC, in the internal MOSFET:
PC = (IF2 x RDSON) x D = (0.352 x 1.5) x 0.154 = 28 mW
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(40)
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LM3402, LM3402HV
SNVS450D – SEPTEMBER 2006 – REVISED FEBRUARY 2010
www.ti.com
Gate charging and VCC loss, PG, in the gate drive and linear regulator:
PG = (IIN-OP + fSW x QG) x VIN PG = (600 x 10-6 + 468000 x 3 x 10-9) x 24 = 48 mW
(41)
Switching loss, PS, in the internal MOSFET:
PS = 0.5 x VIN x IF x (tR + tF) x fSW PS = 0.5 x 24 x 0.35 x (40 x 10-9) x 468000 = 78 mW
(42)
AC rms current loss, PCIN, in the input capacitor:
PCIN = IIN(rms)2 x ESR = (0.126)2 x 0.006 = 0.1 mW (negligible)
(43)
DCR loss, PL, in the inductor
PL = IF2 x DCR = 0.352 x 0.096 = 11.8 mW
(44)
Recirculating diode loss, PD = 119 mW
Current Sense Resistor Loss, PSNS = 92 mW
Electrical efficiency, η = PO / (PO + Sum of all loss terms) = 1.295 / (1.295 + 0.377) = 77%
DIE TEMPERATURE
TLM3402 = (PC + PG + PS) x θJA TLM3402 = (0.028 + 0.05 + 0.078) x 200 = 31°C
(45)
Design Example 2: LM3402HV
The second example application is an RGB backlight for a flat screen monitor. A separate boost regulator
provides a 60V ±5% DC input rail that feeds three LM3402HV current regulators to drive one series array each of
red, green, and blue 1W LEDs. The target for average LED current is 350 mA ±5% in each string. The monitor
will adjust the color temperature dynamically, requiring fast PWM dimming of each string with external, parallel
MOSFETs. 1W green and blue InGaN LEDs have a typical forward voltage of 3.5V, however red LEDs use
AlInGaP technology with a typical forward voltage of 2.9V. In order to match color properly the design requires
14 green LEDs, twice as many as needed for the red and blue LEDs. This example will follow the design for the
green LED array, providing the necessary information to repeat the exercise for the blue and red LED arrays.
The circuit schematic for Design Example 2 is the same as the Typical Application on the front page. The bill of
materials (green array only) can be found in Table 2 at the end of this datasheet.
OUTPUT VOLTAGE
Green Array: VO(G) = 14 x 3.5 + 0.2 = 49.2V
Blue Array: VO(B) = 7 x 3.5 + 0.2 = 24.7V
Red Array: VO(R) = 7 x 2.9 + 0.2 = 20.5V
(46)
(47)
(48)
RON and tON
A compromise in switching frequency is needed in this application to balance the requirements of magnetics size
and efficiency. The high duty cycle translates into large conduction losses and high temperature rise in the IC.
For best response to a PWM dimming signal this circuit will not use an output capacitor; hence a moderate
switching frequency of 300 kHz will keep the inductance from becoming so large that a custom-wound inductor is
needed. This design will use only surface mount components, and the selection of off-the-shelf SMT inductors for
switching regulators is poor at 1000 µH and above. RON is selected from the equation for switching frequency as
follows:
RON =
VO
1.34 x 10-10 x fSW
-10
RON = 49.2 / (1.34 x 10
(49)
(50)
5
x 3 x 10 ) = 1224 kΩ
The closest 1% tolerance resistor is 1.21 MΩ. The switching frequency and on-time of the circuit can then be
found using the equations relating RON and tON to fSW:
fSW = 49.2 / (1210000 x 1.34 x 10-10) = 303 kHz
tON = (1.34 x 10-10 x 1210000) / 60 = 2.7 µs
(51)
(52)
USING AN OUTPUT CAPACITOR
This application is dominated by the need for fast PWM dimming, requiring a circuit without any output
capacitance.
24
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LM3402, LM3402HV
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SNVS450D – SEPTEMBER 2006 – REVISED FEBRUARY 2010
OUTPUT INDUCTOR
In this example the ripple current through the LED array and the inductor are equal. Inductance is selected to
give the smallest ripple current possible while still providing enough ΔvSNS signal for the CS comparator to
operate correctly. Designing to a desired ΔvSNS of 25 mV and assuming that the average inductor current will
equal the desired average LED current of 350 mA yields the target current ripple in the inductor and LEDs:
ΔiF = ΔiL = ΔvSNS / RSNS, RSNS = VSNS / IF
ΔiF = 0.025 / 0.57 = 43.8 mA
(53)
(54)
With the target ripple current determined the inductance can be chosen:
LMIN =
VIN - VO
'iF
x tON
(55)
(56)
LMIN = [(60 – 49.2) x 2.7 x 10-6] / (0.044) = 663 µH
The closest standard inductor value is 680 µH. As with the previous example, the average current rating should
be greater than 350 mA. Separation between the LM3402HV drivers and the LED arrays mean that heat from the
inductor will not threaten the lifetime of the LEDs, but an overheated inductor could still cause the LM3402HV to
enter thermal shutdown.
The inductance itself of the standard part chosen is ±20%. With this tolerance the typical, minimum, and
maximum inductor current ripples can be calculated:
ΔiF(TYP) = [(60 - 49.2) x 2.7 x 10-6] / 680 x 10-6 = 43 mAP-P
ΔiF(MIN) = [(60 - 49.2) x 2.7 x 10-6] / 816 x 10-6 = 36 mAP-P
ΔiF(MAX) = [(60 - 49.2) x 2.7 x 10-6] / 544 x 10-6 = 54 mAP-P
(57)
(58)
(59)
The peak LED/inductor current is then estimated:
IL(PEAK) = IL + [ΔiL(MAX)] / 2
IL(PEAK) = 0.35 + 0.027 = 377 mA
(60)
(61)
In the case of a short circuit across the LED array, the LM3402HV will continue to deliver rated current through
the short but will reduce the output voltage to equal the CS pin voltage of 200 mV. Worst-case peak current in
this condition would be equal to:
ΔiF(LED-SHORT) = [(63 – 0.2) x 2.7 x 10-6] / 544 x 10-6 = 314 mAP-P IF(PEAK) = 0.35 + 0.156 = 506 mA
(62)
In the case of a short at the switch node, the output, or from the CS pin to ground the short circuit current limit
will engage at a typical peak current of 735 mA. In order to prevent inductor saturation during these fault
conditions the inductor’s peak current rating must be above 735 mA. A 680 µH off-the shelf inductor rated to
1.2A (peak) and 0.72A (average) with a DCR of 1.1Ω will be used for the green LED array.
RSNS
A preliminary value for RSNS was determined in selecting ΔiL. This value should be re-evaluated based on the
calculations for ΔiF:
RSNS =
0.2 x L
IF x L + VO x tSNS - VIN - VO x tON
2
(63)
Sub-1Ω resistors are available in both 1% and 5% tolerance. A 1%, 0.56Ω device is the closest value, and a
0.125W, 0805 size device will handle the power dissipation of 69 mW. With the resistance selected, the average
value of LED current is re-calculated to ensure that current is within the ±5% tolerance requirement. From the
expression for LED current accuracy:
IF = 0.19 / 0.56 + 0.043 / 2 = 361 mA, 3% above 350 mA
(64)
INPUT CAPACITOR
Following the calculations from the Input Capacitor section, ΔvIN(MAX) will be 1%P-P = 600 mV. The minimum
required capacitance is:
CIN(MIN) = (0.35 x 2.7 x 10-6) / 0.6 = 1.6 µF
Copyright © 2006–2010, Texas Instruments Incorporated
Product Folder Links: LM3402 LM3402HV
(65)
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25
LM3402, LM3402HV
SNVS450D – SEPTEMBER 2006 – REVISED FEBRUARY 2010
www.ti.com
In expectation that more capacitance will be needed to prevent power supply interaction a 2.2 µF ceramic
capacitor rated to 100V with X7R dielectric in an 1812 case size will be used. From the Design Considerations
section, input rms current is:
IIN-RMS = 0.35 x Sqrt(0.82 x 0.18) = 134 mA
(66)
Ripple current ratings for 1812 size ceramic capacitors are typically higher than 2A, more than enough for this
design.
RECIRCULATING DIODE
The input voltage of 60V ±5% requires Schottky diodes with a reverse voltage rating greater than 60V. Some
manufacturers provide Schottky diodes with ratings of 70, 80 or 90V; however the next highest standard voltage
rating is 100V. Selecting a 100V rated diode provides a large safety margin for the ringing of the switch node and
also makes cross-referencing of diodes from different vendors easier.
The next parameters to be determined are the forward current rating and case size. In this example the high duty
cycle (D = 49.2 / 60 = 82%) places less thermals stress on D1 and more on the internal power MOSFET of the
LM3402. The estimated average diode current is:
ID = 0.361 x 0.18 = 65 mA
(67)
A Schottky with a forward current rating of 0.5A would be adequate, however at 100V the majority of diodes have
a minimum forward current rating of 1A. To determine the proper case size, the dissipation and temperature rise
in D1 can be calculated as shown in the Design Considerations section. VD for a small case size such as SOD123F in a 100V, 1A Schottky diode at 350 mA is approximately 0.65V and the θJA is 88°C/W. Power dissipation
and temperature rise can be calculated as:
PD = 0.065 x 0.65 = 42 mW TRISE = 0.042 x 88 = 4°C
(68)
CB AND CF
The bootstrap capacitor CB should always be a 10 nF ceramic capacitor with X7R dielectric. A 25V rating is
appropriate for all application circuits. The linear regulator filter capacitor CF should always be a 100 nF ceramic
capacitor, also with X7R dielectric and a 25V rating.
EFFICIENCY
To estimate the electrical efficiency of this example the power dissipation in each current carrying element can
be calculated and summed. Electrical efficiency, η, should not be confused with the optical efficacy of the circuit,
which depends upon the LEDs themselves.
Total output power, PO, is calculated as:
PO = IF x VO = 0.361 x 49.2 = 17.76W
(69)
Conduction loss, PC, in the internal MOSFET:
PC = (IF2 x RDSON) x D = (0.3612 x 1.5) x 0.82 = 160 mW
(70)
Gate charging and VCC loss, PG, in the gate drive and linear regulator:
PG = (IIN-OP + fSW x QG) x VIN PG = (600 x 10-6 + 3 x 105 x 3 x 10-9) x 60 = 90 mW
(71)
Switching loss, PS, in the internal MOSFET:
PS = 0.5 x VIN x IF x (tR + tF) x fSW PS = 0.5 x 60 x 0.361 x 40 x 10-9 x 3 x 105 = 130 mW
(72)
AC rms current loss, PCIN, in the input capacitor:
PCIN = IIN(rms)2 x ESR = (0.134)2 x 0.006 = 0.1 mW (negligible)
(73)
DCR loss, PL, in the inductor
PL = IF2 x DCR = 0.352 x 1.1 = 135 mW
(74)
Recirculating diode loss, PD = 42 mW
Current Sense Resistor Loss, PSNS = 69 mW
Electrical efficiency, η = PO / (PO + Sum of all loss terms) = 17.76 / (17.76 + 0.62) = 96%
Temperature Rise in the LM3402HV IC is calculated as:
TLM3402 = (PC + PG + PS) x θJA = (0.16 + 0.084 + 0.13) x 200 = 74.8°C
26
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(75)
Copyright © 2006–2010, Texas Instruments Incorporated
Product Folder Links: LM3402 LM3402HV
LM3402, LM3402HV
www.ti.com
SNVS450D – SEPTEMBER 2006 – REVISED FEBRUARY 2010
Layout Considerations
The performance of any switching converter depends as much upon the layout of the PCB as the component
selection. The following guidelines will help the user design a circuit with maximum rejection of outside EMI and
minimum generation of unwanted EMI.
COMPACT LAYOUT
Parasitic inductance can be reduced by keeping the power path components close together and keeping the
area of the loops that high currents travel small. Short, thick traces or copper pours (shapes) are best. In
particular, the switch node (where L1, D1, and the SW pin connect) should be just large enough to connect all
three components without excessive heating from the current it carries. The LM3402/02HV operates in two
distinct cycles whose high current paths are shown in Figure 6:
+
-
Figure 31. Buck Converter Current Loops
The dark grey, inner loop represents the high current path during the MOSFET on-time. The light grey, outer loop
represents the high current path during the off-time.
GROUND PLANE AND SHAPE ROUTING
The diagram of Figure 6 is also useful for analyzing the flow of continuous current vs. the flow of pulsating
currents. The circuit paths with current flow during both the on-time and off-time are considered to be continuous
current, while those that carry current during the on-time or off-time only are pulsating currents. Preference in
routing should be given to the pulsating current paths, as these are the portions of the circuit most likely to emit
EMI. The ground plane of a PCB is a conductor and return path, and it is susceptible to noise injection just as
any other circuit path. The continuous current paths on the ground net can be routed on the system ground plane
with less risk of injecting noise into other circuits. The path between the input source and the input capacitor and
the path between the recirculating diode and the LEDs/current sense resistor are examples of continuous current
paths. In contrast, the path between the recirculating diode and the input capacitor carries a large pulsating
current. This path should be routed with a short, thick shape, preferably on the component side of the PCB.
Multiple vias in parallel should be used right at the pad of the input capacitor to connect the component side
shapes to the ground plane. A second pulsating current loop that is often ignored is the gate drive loop formed
by the SW and BOOT pins and capacitor CB. To minimize this loop at the EMI it generates, keep CB close to the
SW and BOOT pins.
CURRENT SENSING
The CS pin is a high-impedance input, and the loop created by RSNS, RZ (if used), the CS pin and ground should
be made as small as possible to maximize noise rejection. RSNS should therefore be placed as close as possible
to the CS and GND pins of the IC.
REMOTE LED ARRAYS
In some applications the LED or LED array can be far away (several inches or more) from the LM3402/02HV, or
on a separate PCB connected by a wiring harness. When an output capacitor is used and the LED array is large
or separated from the rest of the converter, the output capacitor should be placed close to the LEDs to reduce
the effects of parasitic inductance on the AC impedance of the capacitor. The current sense resistor should
remain on the same PCB, close to the LM3402/02HV.
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Product Folder Links: LM3402 LM3402HV
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LM3402, LM3402HV
SNVS450D – SEPTEMBER 2006 – REVISED FEBRUARY 2010
www.ti.com
Table 1. BOM for Design Example 1
ID
Part Number
Type
Size
Parameters
Qty
Vendor
U1
LM3402
LED Driver
VSSOP-8
40V, 0.5A
1
NSC
L1
SLF7045T-330MR82
Inductor
7.0x7.0 x4.5mm
33µH, 0.82A, 96mΩ
1
TDK
D1
CMHSH5-4
Schottky Diode
SOD-123
40V, 0.5A
1
Central Semi
Cf
VJ0805Y104KXXAT
Capacitor
0805
100nF 10%
1
Vishay
Cb
VJ0805Y103KXXAT
Capacitor
0805
10nF 10%
1
Vishay
Cin
C3216X7R1H105M
Capacitor
1206
1µF 50V
1
TDK
Co
C2012X7R1A225M
Capacitor
0805
2.2 µF 10V
1
TDK
Rsns
ERJ6BQFR75V
Resistor
0805
0.75Ω 1%
1
Panasonic
Ron
CRCW08055902F
Resistor
0805
59.0 kΩ 1%
1
Vishay
Table 2. BOM for Design Example 2
28
ID
Part Number
Type
Size
Parameters
Qty
Vendor
U1
LM3402HV
LED Driver
VSSOP-8
75V, 0.5A
1
NSC
L1
DO5022P-684
Inductor
18.5x15.2 x7.1mm
680µH, 1.2A, 1.1Ω
1
Coilcraft
D1
CMMSH1-100
Schottky Diode
SOD-123F
100V, 1A
1
Central Semi
Cf
VJ0805Y104KXXAT
Capacitor
0805
100nF 10%
1
Vishay
Cb
VJ0805Y103KXXAT
Capacitor
0805
10nF 10%
1
Vishay
Cin
C4532X7R2A225M
Capacitor
1812
2.2µF 100V
1
TDK
Rsns
ERJ6BQFR56V
Resistor
0805
0.56Ω 1%
1
Panasonic
Ron
CRCW08051214F
Resistor
0805
1.21MΩ 1%
1
Vishay
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PACKAGE OPTION ADDENDUM
www.ti.com
3-Mar-2013
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package Qty
Drawing
Eco Plan
Lead/Ball Finish
(2)
MSL Peak Temp
Op Temp (°C)
Top-Side Markings
(3)
(4)
LM3402HVMM/NOPB
ACTIVE
VSSOP
DGK
8
1000
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
SNFB
LM3402HVMMX/NOPB
ACTIVE
VSSOP
DGK
8
3500
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
SNFB
LM3402HVMR/NOPB
ACTIVE SO PowerPAD
DDA
8
95
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
L3402
HVMR
LM3402HVMRX/NOPB
ACTIVE SO PowerPAD
DDA
8
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
L3402
HVMR
TBD
Call TI
Call TI
-40 to 125
SNEB
1000
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
SNEB
LM3402MM
OBSOLETE
VSSOP
DGK
8
LM3402MM/NOPB
ACTIVE
VSSOP
DGK
8
LM3402MMX
OBSOLETE
VSSOP
DGK
8
TBD
Call TI
Call TI
-40 to 125
SNEB
LM3402MMX/NOPB
ACTIVE
VSSOP
DGK
8
3500
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
SNEB
LM3402MR/NOPB
ACTIVE SO PowerPAD
DDA
8
95
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
L3402
MR
LM3402MRX/NOPB
ACTIVE SO PowerPAD
DDA
8
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
L3402
MR
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
3-Mar-2013
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
Only one of markings shown within the brackets will appear on the physical device.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
5-Feb-2013
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
LM3402HVMM/NOPB
VSSOP
DGK
8
1000
178.0
12.4
5.3
3.4
1.4
8.0
12.0
Q1
LM3402HVMMX/NOPB
VSSOP
DGK
8
3500
330.0
12.4
5.3
3.4
1.4
8.0
12.0
Q1
LM3402HVMRX/NOPB
SO
Power
PAD
DDA
8
2500
330.0
12.4
6.5
5.4
2.0
8.0
12.0
Q1
LM3402MM/NOPB
VSSOP
DGK
8
1000
178.0
12.4
5.3
3.4
1.4
8.0
12.0
Q1
LM3402MMX/NOPB
VSSOP
DGK
8
3500
330.0
12.4
5.3
3.4
1.4
8.0
12.0
Q1
LM3402MRX/NOPB
SO
Power
PAD
DDA
8
2500
330.0
12.4
6.5
5.4
2.0
8.0
12.0
Q1
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
5-Feb-2013
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
LM3402HVMM/NOPB
VSSOP
DGK
8
1000
203.0
190.0
41.0
LM3402HVMMX/NOPB
VSSOP
DGK
8
3500
349.0
337.0
45.0
LM3402HVMRX/NOPB
SO PowerPAD
DDA
8
2500
358.0
343.0
63.0
LM3402MM/NOPB
VSSOP
DGK
8
1000
203.0
190.0
41.0
LM3402MMX/NOPB
VSSOP
DGK
8
3500
349.0
337.0
45.0
LM3402MRX/NOPB
SO PowerPAD
DDA
8
2500
358.0
343.0
63.0
Pack Materials-Page 2
MECHANICAL DATA
DDA0008B
MRA08B (Rev B)
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help enable customers to design and create their own end-product solutions that meet applicable functional safety standards and
requirements. Nonetheless, such components are subject to these terms.
No TI components are authorized for use in FDA Class III (or similar life-critical medical equipment) unless authorized officers of the parties
have executed a special agreement specifically governing such use.
Only those TI components which TI has specifically designated as military grade or “enhanced plastic” are designed and intended for use in
military/aerospace applications or environments. Buyer acknowledges and agrees that any military or aerospace use of TI components
which have not been so designated is solely at the Buyer's risk, and that Buyer is solely responsible for compliance with all legal and
regulatory requirements in connection with such use.
TI has specifically designated certain components as meeting ISO/TS16949 requirements, mainly for automotive use. In any case of use of
non-designated products, TI will not be responsible for any failure to meet ISO/TS16949.
Products
Applications
Audio
www.ti.com/audio
Automotive and Transportation
www.ti.com/automotive
Amplifiers
amplifier.ti.com
Communications and Telecom
www.ti.com/communications
Data Converters
dataconverter.ti.com
Computers and Peripherals
www.ti.com/computers
DLP® Products
www.dlp.com
Consumer Electronics
www.ti.com/consumer-apps
DSP
dsp.ti.com
Energy and Lighting
www.ti.com/energy
Clocks and Timers
www.ti.com/clocks
Industrial
www.ti.com/industrial
Interface
interface.ti.com
Medical
www.ti.com/medical
Logic
logic.ti.com
Security
www.ti.com/security
Power Mgmt
power.ti.com
Space, Avionics and Defense
www.ti.com/space-avionics-defense
Microcontrollers
microcontroller.ti.com
Video and Imaging
www.ti.com/video
RFID
www.ti-rfid.com
OMAP Applications Processors
www.ti.com/omap
TI E2E Community
e2e.ti.com
Wireless Connectivity
www.ti.com/wirelessconnectivity
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