Intersil ISL85402IRZ-TK 2.5a regulator with integrated high-side mosfet for synchronous buck or boost buck converter Datasheet

2.5A Regulator with Integrated High-Side MOSFET for
Synchronous Buck or Boost Buck Converter
ISL85402
Features
The ISL85402 is a synchronous buck controller with a 125mΩ
high-side MOSFET and low-side driver integrated. The ISL85402
supports a wide input range of 3V to 36V in buck mode. It
supports 2.5A continuous load under conditions of 5V VOUT, VIN
range of 8V to 36V, 500kHz and +105°C ambient temperature
with still air. For any specific application, the actual maximum
output current depends upon the die temperature not exceeding
+125°C with the power dissipated in the IC, which is related to input
voltage, output voltage, duty cycle, switching frequency, board layout
and ambient temperature, etc. Refer to “Output Current” on page 14
for more details.
• Buck Mode: Input Voltage Range 3V to 36V (Refer to “Input
Voltage” on page 13 for more details)
The ISL85402 has a flexible selection of operation modes of
forced PWM mode and PFM mode. In PFM mode, the
quiescent input current is as low as 180µA (AUXVCC connected
to VOUT). The load boundary between PFM and PWM can be
programmed to cover wide applications.
The low-side driver can be either used to drive an external low-side
MOSFET for a synchronous buck, or left unused for a standard
non-synchronous buck. The low-side driver can also be used to
drive a boost converter as a pre-regulator followed by a buck
controlled by the same IC, which greatly expands the operating
input voltage range down to 2.5V or lower (Refer to “Typical
Application Schematic III - Boost Buck Converter” on page 5).
The ISL85402 offers the most robust current protections. It
uses peak current mode control with cycle-by-cycle current
limiting. It is implemented with frequency foldback under
current limit condition; besides that, the hiccup overcurrent
mode is also implemented to guarantee reliable operations
under harsh short conditions.
The ISL85402 has comprehensive protections against various faults
including overvoltage and over-temperature protections, etc.
• Boost Mode Expands Operating Input Voltage Lower Than
2.5V (Refer to “Input Voltage” on page 13 for more details)
• Selectable Forced PWM Mode or PFM Mode
• 300µA IC Quiescent Current (PFM, No Load); 180µA Input
Quiescent Current (PFM, No Load, VOUT Connected to
AUXVCC)
• Less than 3µA Shut Down Input Current (IC Disabled)
• Operational Topologies
- Synchronous Buck
- Non-Synchronous Buck
- Two-Stage Boost Buck
• Programmable Frequency from 200kHz to 2.2MHz and
Frequency Synchronization Capability
• ±1% Tight Voltage Regulation Accuracy
• Reliable Overcurrent Protection
- Temperature Compensated Current Sense
- Cycle-by-Cycle Current Limiting with Frequency Foldback
- Hiccup Mode for Worst Case Short Condition
• 20 Ld 4x4 QFN Package
• Pb-Free (RoHS Compliant)
Applications
• General Purpose
• 24V Bus Power
• Battery Power
• Point of Load
• Embedded Processor and I/O Supplies
100
95
VIN
SYNC
AUXVCC
VIN
BOOT
ISL85402
PHASE
ILIMIT
LGATE
SS
EXT_BOOST
FS
SGND
VOUT
EFFICIENCY (%)
PGOOD
EN
MODE
VCC
6V VIN
90
85
80
36V VIN
75
24V VIN
70
65
PGND
60
FB
55
COMP
12V VIN
50
0.1m
1m
10m
100m
1.0
2.5
LOAD CURRENT (A)
FIGURE 1. TYPICAL APPLICATION
April 25, 2013
FN7640.1
1
FIGURE 2. EFFICIENCY, SYNCHRONOUS BUCK, PFM MODE,
VOUT 5V, TA = +25°C
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas LLC 2011, 2013. All Rights Reserved
Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries.
All other trademarks mentioned are the property of their respective owners.
ISL85402
Pin Configuration
AUXVCC
VCC
SGND
VIN
VIN
ISL85402
(20 LD QFN)
TOP VIEW
20
19
18
17
16
EN
1
15
BOOT
14
PGND
13
LGATE
FS
2
SS
3
FB
4
12
SYNC
COMP
5
11
EXT_BOOST
7
8
9
MODE
PGOOD
PHASE
10
PHASE
6
ILIMIT
21Pad
Thermal
21
PAD
Functional Pin Descriptions
PIN NAME
PIN #
DESCRIPTION
EN
1
The controller is enabled when this pin is left floating or pulled HIGH. The IC is disabled when this pin is pulled LOW.
Range: 0V to 5.5V.
FS
2
Connecting this pin to VCC, or GND, or leaving it open will force the IC to have 500kHz switching frequency. The oscillator
switching frequency can also be programmed by adjusting the resistor from this pin to GND.
SS
3
Connect a capacitor from this pin to ground. This capacitor, along with an internal 5µA current source, sets the soft-start
interval of the converter. Also, this pin can be used to track a ramp on this pin.
FB
4
This pin is the inverting input of the voltage feedback error amplifier. With a properly selected resistor divider connected from
VOUT to FB, the output voltage can be set to any voltage between the power rail (reduced by maximum duty cycle and voltage
drop) and the 0.8V reference. Loop compensation is achieved by connecting an RC network across COMP and FB. The FB pin
is also monitored for overvoltage events.
COMP
5
Output of the voltage feedback error amplifier.
ILIMIT
6
Programmable current limit pin. With this pin connected to the VCC pin, or to GND, or left open, the current limiting threshold
is set to default of 3.6A; the current limiting threshold can be programmed with a resistor from this pin to GND.
MODE
7
Mode selection pin. Pull this pin to GND for forced PWM mode; to have it floating or connected to VCC will enable PFM mode
when the peak inductor current is below the default threshold of 700mA. The current boundary threshold between PFM and
PWM can also be programmed with a resistor at this pin to ground. Check for more details in the “PFM Mode Operation” on
page 13.
PGOOD
8
PGOOD is an open drain output that will be pulled low immediately under the events when the output is out of regulation (OV
or UV) or when the EN pin is pulled low. PGOOD is equipped with a fixed delay of 1000 cycles upon output power-up (VO > 90%).
PHASE
9, 10
These pins are the PHASE nodes that should be connected to the output inductor. These pins are connected to the source of
the high-side N-channel MOSFET.
11
This pin is used to set boost mode and monitor the battery voltage that is the input of the boost converter. After VCC POR, the
controller will detect the voltage on this pin; if voltage on this pin is below 200mV, the controller is set in
synchronous/non-synchronous buck mode and will latch in this state unless VCC is below POR falling threshold; if the voltage
on this pin after VCC POR is above 200mV, the controller is set in boost mode and latch in this state. In boost mode, the
low-side driver output PWM with same duty cycle with upper-side driver to drive the boost switch.
In boost mode, this pin is used to monitor input voltage through a resistor divider. By setting the resistor divider, the high
threshold and hysteresis can be programmed. When voltage on this pin is above 0.8V, the PWM output (LGATE) for the boost
converter is disabled, and when voltage on this pin is below 0.8V minus the hysteresis, the boost PWM is enabled.
In boost mode operation, PFM is disabled when boost PWM is enabled. Check the “Boost Converter Operation” on page 14
for more details.
EXT_BOOST
2
FN7640.1
April 25, 2013
ISL85402
Functional Pin Descriptions
PIN NAME
PIN #
(Continued)
DESCRIPTION
SYNC
12
This pin can be used to synchronize two or more ISL85402 controllers. Multiple ISL85402s can be synchronized with their
SYNC pins connected together. 180 degree phase shift is automatically generated between the master and slave ICs.
The internal oscillator can also lock to an external frequency source applied on this pin with square pulse waveform (with
frequency 10% higher than the IC’s local frequency, and pulse width higher than 150ns). Range: 0V to 5.5V.
This pin should be left floating if not used.
LGATE
13
In synchronous buck mode, this pin is used to drive the lower side MOSFET to improve efficiency.
In non-synchronous buck when a diode is used as the bottom side power device, this pin should be connected to VCC before
VCC startup to have low-side driver (LGATE) disabled.
In boost mode, it can be used to drive the boost power MOSFET. The boost control PWM is same with the buck control PWM.
PGND
14
This pin is used as the ground connection of the power flow including driver. Connect it to large ground plane.
BOOT
15
This pin provides bias voltage to the high-side MOSFET driver. A bootstrap circuit is used to create a voltage suitable to drive
the internal N-channel MOSFET. The boot charge circuitries are integrated inside of the IC. No external boot diode is needed.
A 1µF ceramic capacitor is recommended to be used between BOOT and PHASE pin.
16, 17
Connect the input rail to these pins that are connected to the drain of the integrated high-side MOSFET as well as the source
for the internal linear regulator that provides the bias of the IC. Range: 3V to 36V.
With the part switching, the operating input voltage applied to the VIN pins must be under 36V. This recommendation allows
for short voltage ringing spikes (within a couple of ns time range) due to switching while not exceeding “Absolute Maximum
Ratings” on page 6.
VIN
SGND
18
This pin provides the return path for the control and monitor portions of the IC. Connect it to a quiet ground plane.
VCC
19
This pin is the output of the internal linear regulator that supplies the bias for the IC including the driver. A minimum 4.7µF
decoupling ceramic capacitor is recommended between VCC to ground.
20
This pin is the input of the auxiliary internal linear regulator, which can be supplied by the regulator output after power-up.
With such configuration, the power dissipation inside of the IC is reduced. The input range for this LDO is 3V to 20V.
In boost mode operation, this pin works as boost output overvoltage detection pin. It detects the boost output through a
resistor divider. When voltage on this pin is above 0.8V, the boost PWM is disabled; and when voltage on this pin is below 0.8V
minus the hysteresis, the boost PWM is enabled.
Range: 0V to 20V.
21
Bottom thermal pad. It is not connected to any electrical potential of the IC. In layout it must be connected to PCB ground
copper plane with area as large as possible to effectively reduce the thermal impedance.
AUXVCC
PAD
Ordering Information
PART NUMBER
(Notes 1, 2, 3)
PART
MARKING
ISL85402IRZ
85 402IRZ
ISL85402EVAL1Z
Evaluation Board
TEMP.
RANGE (°C)
-40 to +105
PACKAGE
(PB-Free)
20 Ld 4x4 QFN
PKG. DWG. #
L20.4x4C
NOTES:
1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte
tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil
Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page for ISL85402. For more information on MSL please see techbrief TB363.
3
FN7640.1
April 25, 2013
Block Diagram
AUXVCC
VCC
PGOOD
VIN (x2)
VIN
CURRENT
MONITOR
AUXILARY LDO
BIAS LDO
4
ILIMIT
POWER-ON
RESET
SGND
VCC
BOOT
OCP, OVP, OTP
PFM LOGIC
BOOST MODE CONTROL
EN
EXT_BOOST
MODE
PFM/FPWM
VOLTAGE
MONITOR
SYNC
FS
SLOPE
COMPENSATION
LGATE
OSCILLATOR
+
+
SOFT-START
LOGIC
VCC
0.8V
REFERENCE
5 µA
EA
SS
BOOT REFRESH
FB
COMPARATOR
COMP
PGND
ISL85402
PHASE (x2)
GATE DRIVE
FN7640.1
April 25, 2013
ISL85402
Typical Application Schematic I
PGOOD
EN
MODE
SYNC
AUXVCC
VCC
PGOOD
EN
MODE
VIN
VIN
SYNC
AUXVCC
BOOT
ISL85402
VIN
V OUT
PHASE
BOOT
VCC
ILIMIT
VIN
ISL85402
V OUT
PHASE
ILIMIT
LGATE
SS
LGATE
SS
PGND
PGND
EXT_BOOST
FS
SGND
EXT_BOOST
FS
SGND
FB
COMP
FB
COMP
(b) NON-SYNCHRONOUS BUCK
(a) SYNCHRONOUS BUCK
Typical Application Schematic II - VCC Switch-Over to VOUT
PGOOD
EN
MODE
SYNC
AUXVCC
VCC
PGOOD
EN
MODE
VIN
VIN
SYNC
AUXVCC
BOOT
ISL85402
VCC
VOUT
PHASE
EXT_BOOST
FS
SGND
VIN
BOOT
ISL85402
PHASE
V OUT
ILIMIT
ILIMIT
SS
VIN
LGATE
LGATE
SS
PGND
PGND
EXT_BOOST
FS
SGND
FB
COMP
(a) SYNCHRONOUS BUCK
FB
COMP
(b) NON-SYNCHRONOUS BUCK
Typical Application Schematic III - Boost Buck Converter
Battery
+
+
R1
R2
PGOOD
EN
MODE
EXT_BOOST
R3
LGATE
AUXVCC
SYNC
R4
VIN
VCC
ISL85402
ILIMIT
SS
FS
SGND
5
BOOT
PHASE
V OUT
PGND
COMP
FB
FN7640.1
April 25, 2013
ISL85402
Absolute Maximum Ratings
Thermal Information
VIN, PHASE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +44V
VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +6.0V
AUXVCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +22V
Absolute Boot Voltage, VBOOT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +50.0V
Upper Driver Supply Voltage, VBOOT - VPHASE . . . . . . . . . . . . . . . . . . . +6.0V
All Other Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to VCC + 0.3V
ESD Rating
Human Body Model (Tested per JESD22-A114F) . . . . . . . . . . . . . . 2000V
Machine Model (Tested per JESD22-A115C) . . . . . . . . . . . . . . . . . . 200V
Charged Device Model (Tested per JESD22-C101E). . . . . . . . . . . . 1000V
Latchup Rating (Tested per JESD78B; Class II, Level A) . . . . . . . . . 100mA
Thermal Resistance
θJA (°C/W) θJC (°C/W)
ISL85402 QFN 4x4 Package (Notes 4, 5). . . . . .
40
3.5
Maximum Junction Temperature (Plastic Package) . . . . . . . . . . . . . . +150°C
Maximum Storage Temperature Range. . . . . . . . . . . . . . . . . -65°C to +150°C
Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
Supply Voltage on VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3V to 36V
AUXVCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +20V
Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +105°C
Junction Temperature Range . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +125°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTES:
4. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech
Brief TB379.
5. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
Refer to “Block Diagram” on page 4 and “Typical Application Schematics” on page 5. Operating
Conditions Unless Otherwise Noted: VIN = 12V, or VCC = 4.5V ±10%, TA = -40°C to +105°C. Typicals are at TA = +25°C. Boldface limits apply
over the operating temperature range, -40°C to +105°C.
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6) UNITS
VIN PIN SUPPLY
VIN Pin Voltage Range
Operating Supply Current
IQ
Shut Down Supply Current
IIN_SD
VIN Pin
3.05
36
V
VIN Pin connected to VCC
3.05
5.5
V
MODE = VCC/FLOATING (PFM), no load at the
output
300
µA
MODE = GND (Forced PWM), VIN = 12V,
IC Operating, Not Including Driving Current
1.2
mA
EN connected to GND, VIN = 12V
1.8
3
µA
4.5
4.8
V
VIN = 4.2V, IVCC = 35mA
0.3
0.5
V
VIN = 3V, IVCC = 25mA
0.25
0.3
V
INTERNAL MAIN LINEAR REGULATOR
MAIN LDO VCC Voltage
VCC
MAIN LDO Dropout Voltage
VDROPOUT_MAIN
VIN > 5V
4.2
VCC Current Limit of MAIN LDO
60
mA
INTERNAL AUXILIARY LINEAR REGULATOR
AUXVCC Input Voltage Range
VAUXVCC
AUX LDO VCC Voltage
VCC
LDO Dropout Voltage
VDROPOUT_AUX
3
20
V
4.5
4.8
V
VAUXVCC = 4.2V, IVCC = 35mA
0.3
0.5
V
VAUXVCC = 3V, IVCC = 25mA
0.25
0.3
V
VAUXVCC > 5V
4.2
Current Limit of AUX LDO
60
AUX LDO Switch-over Rising Threshold
VAUXVCC_RISE
AUXVCC voltage rise; Switch to Auxiliary LDO
AUX LDO Switch-over Falling Threshold Voltage
VAUXVCC_FALL
AUXVCC voltage fall; Switch back to main BIAS
LDO
AUX LDO Switch-over Hysteresis
VAUXVCC_HYS
AUXVCC Switch-over Hysteresis
6
mA
3
3.1
3.2
V
2.73
2.87
2.97
V
0.2
V
FN7640.1
April 25, 2013
ISL85402
Electrical Specifications
Refer to “Block Diagram” on page 4 and “Typical Application Schematics” on page 5. Operating
Conditions Unless Otherwise Noted: VIN = 12V, or VCC = 4.5V ±10%, TA = -40°C to +105°C. Typicals are at TA = +25°C. Boldface limits apply
over the operating temperature range, -40°C to +105°C. (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6) UNITS
2.82
2.9
3.05
V
2.8
V
POWER-ON RESET
Rising VCC POR Threshold
VPORH_RISE
Falling VCC POR Threshold
VPORL_FALL
2.6
VCC POR Hysteresis
VPORL_HYS
0.3
V
ENABLE
Required Enable On Voltage
VENH
Required Enable Off Voltage
VENL
EN Pull-up Current
IEN_PULLUP
2
V
0.8
V
EN Left Floating, VIN = 24V
0.8
µA
EN Left Floating, VIN = 12V
0.5
µA
EN Left Floating, VIN = 5V
0.25
µA
OSCILLATOR
PWM Frequency
FOSC
RFS = 665kΩ
160
200
240
kHz
RFS = 51.1kΩ
1950
2200
2450
kHz
FS Pin Connected to VCC or Floating or GND
450
500
550
kHz
MIN ON Time
tMIN_ON
130
225
ns
MIN OFF Time
tMIN_OFF
210
325
ns
Input High Threshold
VIH
2
V
Input Low Threshold
VIL
0.5
V
SYNCHRONIZATION
Input Minimum Pulse Width
25
ns
Input Impedance
100
kΩ
Input Minimum Frequency Divided by Free
Running Frequency
1.1
Input Maximum Frequency Divided by Free
Running Frequency
1.6
Output Pulse Width
CSYNC = 100pF
100
ns
RLOAD = 1kΩ
VCC0.25
V
VOL
GND
V
VREF
0.8
V
Output Pulse High
VOH
Output Pulse Low
REFERENCE VOLTAGE
Reference Voltage
System Accuracy
-1.0
FB Pin Source Current
+1.0
5
%
nA
Soft-start
Soft-Start Current
ISS
3
5
7
µA
ERROR AMPLIFIER
Unity Gain-Bandwidth
CLOAD = 50pF
10
MHz
DC Gain
CLOAD = 50pF
88
dB
3.6
V
Maximum Output Voltage
7
FN7640.1
April 25, 2013
ISL85402
Electrical Specifications
Refer to “Block Diagram” on page 4 and “Typical Application Schematics” on page 5. Operating
Conditions Unless Otherwise Noted: VIN = 12V, or VCC = 4.5V ±10%, TA = -40°C to +105°C. Typicals are at TA = +25°C. Boldface limits apply
over the operating temperature range, -40°C to +105°C. (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
(Note 6)
Minimum Output Voltage
Slew Rate
SR
CLOAD = 50pF
TYP
MAX
(Note 6) UNITS
0.5
V
5
V/µs
700
mA
PFM MODE CONTROL
Default PFM Current Threshold
MODE = VCC or Floating
INTERNAL HIGH-SIDE MOSFET
Upper MOSFET rDS(ON)
rDS(ON)_UP
125
180
mΩ
LOW-SIDE MOSFET GATE DRIVER
LGate Source Resistance
100mA Source Current
3.5
Ω
LGATE Sink Resistance
100mA Sink Current
3.3
Ω
BOOST CONVERTER CONTROL
EXT_BOOST Boost_Off Threshold Voltage
EXT_BOOST Hysteresis Sink Current
IEXT_BOOST_HYS
AUXVCC Boost Turn-Off Threshold Voltage
AUXVCC Hysteresis Sink Current
IAUXVCC_HYS
0.74
0.8
0.86
V
2.4
3.2
3.8
µA
0.74
0.8
0.86
V
2.4
3.2
3.8
µA
104
110
116
%
84
90
96
%
POWER-GOOD MONITOR
Overvoltage Rising Trip Point
VFB/VREF
Percentage of Reference Point
Overvoltage Rising Hysteresis
VFB/VOVTRIP
Percentage Below OV Trip Point
Undervoltage Falling Trip Point
VFB/VREF
Percentage of Reference Point
Undervoltage Falling Hysteresis
VFB/VUVTRIP
Percentage Above UV Trip Point
PGOOD Rising Delay
tPGOOD_DELAY
PGOOD Leakage Current
PGOOD Low Voltage
3
%
3
%
fOSC = 500kHz
2
ms
PGOOD HIGH, VPGOOD = 4.5V
10
nA
0.10
V
VPGOOD
PGOOD LOW, IPGOOD = 0.2mA
Default Cycle-by-Cycle Current Limit Threshold
IOC_1
ILIMIT = GND or VCC or Floating
Hiccup Current Limit Threshold
IOC_2
Hiccup, IOC_2/IOC_1
115
%
OV Latching-off Trip Point
Percentage of Reference Point
LG = UG = LATCH LOW
120
%
OV Non-Latching-off Trip Point
Percentage of Reference Point
LG = UG = LOW
110
%
OV Non-Latching-off Release Point
Percentage of Reference Point
102.5
%
Over-Temperature Trip Point
155
°C
Over-Temperature Recovery Threshold
140
°C
OVERCURRENT PROTECTION
3
3.6
4.2
A
OVERVOLTAGE PROTECTION
OVER-TEMPERATURE PROTECTION
NOTE:
6. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization
and are not production tested.
8
FN7640.1
April 25, 2013
ISL85402
100
95
90
85
80
75
70
65
60
55
50
45
40
35
30
0.0
100
95
24V VIN
12V VIN
6V VIN
90
36V VIN
EFFICIENCY (%)
EFFICIENCY (%)
Performance Curves
6V VIN
12V VIN
85
80
36V VIN
75
24V VIN
70
65
60
55
0.5
1.0
1.5
2.0
50
0.1m
2.5
1m
FIGURE 3. EFFICIENCY, SYNCHRONOUS BUCK, FORCED PWM
MODE, 500kHz, VOUT 5V, TA = +25°C
4.970
4.968
4.968
4.966
4.966
1.0
2.5
4.964
IO = 0A
4.962
VOUT (V)
VOUT (V)
4.964
IO = 2A
4.960
4.958
4.956
IO = 1A
4.962
4.952
4.952
10
15
20
25
INPUT VOLTAGE (V)
30
12V VIN
4.950
0.0
36
0.5
1.0
1.5
LOAD CURRENT (A)
2.0
2.5
FIGURE 6. LOAD REGULATION, VOUT 5V, TA = +25°C
100
95
90
12V VIN
85
24V VIN
EFFICIENCY (%)
100
95
90
85
80
75
70
65
60
55
50
45
40
35
30
0.0
36V VIN
4.956
4.954
5
6V VIN
4.958
4.954
0
24V VIN
4.960
FIGURE 5. LINE REGULATION, VOUT 5V, TA = +25°C
EFFICIENCY (%)
100m
FIGURE 4. EFFICIENCY, SYNCHRONOUS BUCK, PFM MODE,
VOUT 5V, TA = +25°C
4.970
4.950
10m
LOAD CURRENT (A)
LOAD CURRENT (A)
36V VIN
6V VIN
6V VIN
12V VIN
80
36V VIN
75
24V VIN
70
65
60
55
50
45
0.5
1.0
1.5
2.0
2.5
LOAD CURRENT (A)
FIGURE 7. EFFICIENCY, SYNCHRONOUS BUCK, FORCED PWM
MODE, 500kHz, VOUT 3.3V, TA = +25°C
9
40
0.1m
1m
10m
100m
LOAD CURRENT (A)
1.0
2.5
FIGURE 8. EFFICIENCY, SYNCHRONOUS BUCK, PFM MODE,
VOUT 3.3V, TA = +25°C
FN7640.1
April 25, 2013
ISL85402
Performance Curves (Continued)
200
85
80
180
INPUT CURRENT (µA)
160
IC DIE TEMPERATURE (°C)
VIN = 12V
140
120
VIN = 24V
100
80
60
40
20
75
70
VO = 20V
65
60
55
50
45
VO = 12V
VO = 5V
40
35
30
0
-50
-25
0
25
50
75
100
125
25
0
5
10
15
AMBIENT TEMPERATURE (°C)
FIGURE 9. INPUT QUIESCENT CURRENT UNDER NO LOAD,
PFM MODE, VOUT = 5V
20
VIN (V)
25
30
35
40
FIGURE 10. IC DIE TEMPERATURE UNDER +25°C AMBIENT
TEMPERATURE, STILL AIR, 500kHz, IO = 2A
85
IC DIE TEMPERATURE (°C)
80
VO = 20V
75
VOUT 2V/DIV
70
65
VO = 12V
60
VO = 5V
55
50
PHASE 20V/DIV
45
40
35
30
25
0
5
10
15
20
25
30
35
VIN (V)
FIGURE 11. IC DIE TEMPERATURE UNDER +25°C AMBIENT
TEMPERATURE, STILL AIR, 500kHz, IO = 2.5A
40
2ms/DIV
FIGURE 12. SYNCHRONOUS BUCK MODE, VIN 36V, IO 2A,
ENABLE ON
VOUT 2V/DIV
VOUT 2V/DIV
PHASE 20V/DIV
PHASE 20V/DIV
2ms/DIV
FIGURE 13. SYNCHRONOUS BUCK MODE, VIN 36V, IO 2A,
ENABLE OFF
10
2ms/DIV
FIGURE 14. VIN 36V, PREBIASED START-UP
FN7640.1
April 25, 2013
ISL85402
Performance Curves (Continued)
VOUT 20mV/DIV (5V OFFSET)
VOUT 100mV/DIV (5V OFFSET)
IOUT 1A/DIV
PHASE 20V/DIV
PHASE 20V/DIV
5µs/DIV
FIGURE 15. SYNCHRONOUS BUCK WITH FORCE PWM MODE,
VIN 36V, IO 2A
1ms/DIV
FIGURE 16. VIN 24V, 0 TO 2A STEP LOAD, FORCE PWM MODE
VOUT 200mV/DIV (5V OFFSET)
VOUT 70mV/DIV (5V OFFSET)
VOUT 1V/DIV
LGATE 5V/DIV
LGATE 5V/DIV
IOUT 1A/DIV
PHASE 20V/DIV
PHASE 20V/DIV
100µs/DIV
1ms/DIV
FIGURE 17. VIN 24V, 80mA LOAD, PFM MODE
FIGURE 18. VIN 24V, 0 TO 2A STEP LOAD, PFM MODE
VOUT 10mV/DIV (5V OFFSET)
VOUT 10mV/DIV (5V OFFSET)
PHASE 5V/DIV
PHASE 10V/DIV
20µs/DIV
FIGURE 19. NON-SYNCHRONOUS BUCK, FORCE PWM MODE,
VIN 12V, NO LOAD
11
5µs/DIV
FIGURE 20. NON-SYNCHRONOUS BUCK, FORCE PWM MODE,
VIN 12V, 2A
FN7640.1
April 25, 2013
ISL85402
Performance Curves (Continued)
VOUT BUCK 100mV/DIV (5V OFFSET)
VOUT BUCK 100mV/DIV (5V OFFSET)
VIN_BOOST_INPUT 5V/DIV
VIN_BOOST_INPUT 5V/DIV
PHASE_BOOST 10V/DIV
PHASE_BUCK 10V/DIV
PHASE_BOOST 10V/DIV
PHASE_BUCK 10V/DIV
20ms/DIV
FIGURE 21. BOOST BUCK MODE, BOOST INPUT STEP FROM
36V TO 3V, VOUT BUCK = 5V, IOUT_BUCK = 1A
10ms/DIV
FIGURE 22. BOOST BUCK MODE, BOOST INPUT STEP FROM
3V TO 36V, VOUT BUCK = 5V, IOUT_BUCK = 1A
95
VOUT 5V/DIV
90
PHASE_BUCK 20V/DIV
30V VIN
85
EFFICIENCY (%)
IL_BOOST 2A/DIV
PHASE_BOOST 20V/DIV
15V VIN
80
5V VIN
75
6V VIN
70
65
60
9V VIN
55
50
0.0 0.2
10ms/DIV
FIGURE 23. BOOST BUCK MODE, VO = 9V, IO = 1.8A, BOOST INPUT
DROPS FROM 16V TO 9V DC
12
0.4 0.6
0.8 1.0 1.2
1.4 1.6
1.8 2.0
2.2 2.4
LOAD CURRENT (A)
FIGURE 24. EFFICIENCY, BOOST BUCK, 500kHz, VOUT 12V,
TA = +25°C
FN7640.1
April 25, 2013
ISL85402
Functional Description
default threshold is 700mA when there is no programming
resistor at the MODE pin.
Initialization
Initially the ISL85402 continually monitors the voltage at the EN
pin. When the voltage on the EN pin exceeds its rising ON
threshold, the internal LDO will start up to build up VCC. After
Power-On Reset (POR) circuits detect that VCC voltage has
exceeded the POR threshold, the soft-start will be initiated.
The current threshold for PWM/PFM boundary can be
programmed by choosing the MODE pin resistor value calculated
from Equation 2, where IPFM is the desired PWM/PFM boundary
current threshold and RMODE is the programming resistor.
118500
R MODE = ---------------------------------------IPFM + 0.2
Soft-Start
500
The soft-start (SS) ramp is built up in the external capacitor on
the SS pin that is charged by an internal 5µA current source.
400
C SS [ μF ] = 6.5 ⋅ t SS [ S ]
(EQ. 2)
The SS ramp starts from 0 to a voltage above 0.8V. Once SS
reaches 0.8V, the bandgap reference takes over and IC gets into
steady state operation.
The SS plays a vital role in the hiccup mode of operation. The IC
works as cycle-by-cycle peak current limiting at over load
condition. When a harsh conditon occurs and the current in the
upper side MOSFET reaches the second overcurrent threshold,
the SS pin is pulled to ground and a dummy soft-start cycle is
initiated. At dummy SS cycle, the current to charge soft-start cap
is cut down to 1/5 of its normal value. So a dummy SS cycle
takes 5x of the regular SS cycle. During the dummy SS period,
the control loop is disabled and no PWM output. At the end of
this cycle, it will start the normal SS. The hiccup mode persist
until the second overcurrent threshold is no longer reached.
The ISL85402 is capable of starting up with prebiased output.
PWM Control
Pulling the MODE pin to GND will set the IC in forced PWM mode.
The ISL85402 employs the peak current mode PWM control for
fast transient response and cycle-by-cycle current limiting. See
“Block Diagram” on page 4.
The PWM operation is initialized by the clock from the oscillator.
The upper MOSFET is turned on by the clock at the beginning of a
PWM cycle and the current in the MOSFET starts to ramp up.
When the sum of the current sense signal and the slope
compensation signal reaches the error amplifier output voltage
level, the PWM comparator is trigger to shut down the PWM logic
to turn off the high-side MOSFET. The high-side MOSFET stays off
until the next clock signal comes for next cycle.
The output voltage is sensed by a resistor divider from VOUT to
the FB pin. The difference between the FB voltage and 0.8V
reference is amplified and compensated to generate the error
voltage signal at the COMP pin. Then the COMP pin signal is
compared with the current ramp signal to shut down the PWM.
PFM Mode Operation
To pull the MODE pin HIGH (>2.5V) or leave the MODE pin floating
will set the IC to have PFM (Pulse Frequency Modulation)
operation in light load. In PFM mode, the switching frequency is
dramatically reduced to minimize the switching loss. The
ISL85402 enters PFM mode when the MOSFET peak current is
lower than the PWM/PFM boundary current threshold. The
13
RMODE (kΩ)
(EQ. 1)
300
200
100
0
0.0
0.2
0.4
0.6
0.8
IPFM (A)
1.0
1.2
1.4
FIGURE 25. RMODE vs IPFM
Synchronous and Non-Synchronous Buck
The ISL85402 supports both Synchronous and non-synchronous
buck operations. For a non-synchronous buck operation when a
power diode is used as the low-side power device, the LGATE
driver can be disabled with LGATE connected to VCC (before IC
start-up).
AUXVCC Switch-Over
The ISL85402 has an auxiliary LDO integrated as shown in the
“Block Diagram” on page 4. It is used to replace the internal
MAIN LDO function after the IC startup. “Typical Application
Schematic II - VCC Switch-Over to VOUT” on page 5 shows its
basic application setup with output voltage connected to
AUXVCC. After IC soft-start is done and the output voltage is built
up to steady state, and once the AUXVCC pin voltage is over the
AUX LDO Switch-over Rising Threshold, the MAIN LDO is shut off
and the AUXILIARY LDO is activated to bias VCC. Since the
AUXVCC pin voltage is lower than the input voltage VIN, the
internal LDO dropout voltage and the consequent power loss is
reduced. This feature brings substantial efficiency improvements
in light load range, especially at high input voltage applications.
When the voltage at AUXVCC falls below the AUX LDO Switch-over
Falling Threshold, the AUXILIARY LDO is shut off and the MAIN LDO
is re-activated to bias VCC. At the OV/UV fault events, the IC also
switches back over from AUXILIARY LDO to MAIN LDO.
The AUXVCC switchover function is offered in buck configuration.
It is not offered in boost configuration when the AUXVCC pin is
used to monitor the boost output voltage for OVP.
Input Voltage
With the part switching, the operating ISL85402 input voltage
must be under 36V. This recommendation allows for short
voltage ringing spikes (within a couple of ns time range) due to
FN7640.1
April 25, 2013
ISL85402
part switching while not exceeding the 44V, as stated in the
Absolute Maximum Ratings.
The lowest IC operating input voltage (VIN pin) depends on VCC
voltage and the Rising and Falling VCC POR Threshold in
Electrical Specifications table on page 7. At IC startup when VCC
is just over rising POR threshold, there is no switching before the
soft-start starts. Therefore, the IC minimum startup voltage on
the VIN pin is 3.05V (MAX of Rising VCC POR). When the soft-start
is initiated, the regulator is switching and the dropout voltage
across the internal LDO increases due to driving current. Thus,
the IC VIN pin shutdown voltage is related to driving current and
VCC POR falling threshold. The internal upper side MOSFET has
typical 10nC gate drive. For a typical example of synchronous
buck with 4nC lower MOSFET gate drive and 500kHz switching
frequency, the driving current is 7mA total causing 70mV drop
across internal LDO under 3V VIN. Then the IC shut down voltage
on the VIN pin is 2.87V (2.8V+0.07V). In practical design, extra
room should be taken into account with concern to voltage
spikes at VIN.
With boost buck configuration, the input voltage range can be
expanded further down to 2.5V or lower depending on the boost
stage voltage drop upon maximum duty cycle. Since the boost
output voltage is connected to the VIN pin as the buck inputs,
after the IC starts up, the IC will keep operating and switching as
long as the boost output voltage can keep the VCC voltage higher
than falling threshold. Refer to “Boost Converter Operation” on
page 14 for more details.
Output Voltage
The ISL85402 output voltage can be programmed down to 0.8V
by a resistor divider from VOUT to FB. The maximum achievable
voltage is (VIN*DMAX - VDROP), where VDROP is the voltage drop
in the power path including mainly the MOSFET rDS(ON) and
inductor DCR. The maximum duty cycle DMAX is decided by
(1 - Fs * tMIN(OFF)).
Output Current
With the high-side MOSFET integrated, the maximum output
current, which the ISL85402 can support is decided by the
package and many operating conditions. Thus, including input
voltage, output voltage, duty cycle, switching frequency and
temperature, etc. Note the following points.
• The maximum output current is limited by the maximum OC
threshold that is 4.18A (TYP).
• From the thermal perspective, the die temperature shouldn’t
exceed +125°C with the power loss dissipated inside of the IC.
Figures 10 and 11 show the thermal performance of this part
operating at different conditions.
Figures 10 and 11 show 2A and 2.5A applications under +25°C
still air conditions over VIN range. The temperature rise data in
these figures can be used to estimate the die temperature at
different ambient temperatures under various operating
conditions. Note that more temperature rise is expected at higher
ambient temperature due to more conduction loss caused by
rDS(ON) increase.
ambient conditions). For any other operating conditions, refer to
the previous mentioned thermal curves to estimate the
maximum output current. The output current should be derated
under any conditions causing the die temperature to exceed
+125°C.
Basically, the die temperature is equal to the sum of ambient
temperature and the temperature rise resulting from the power
dissipated by the IC package with a certain junction to ambient
thermal impedance θJA. The power dissipated in the IC is related
to the MOSFET switching loss, conduction loss and the internal
LDO loss. Besides the load, these losses are also related to input
voltage, output voltage, duty cycle, switching frequency and
temperature. With the exposed pad at the bottom, the heat of
the IC mainly goes through the bottom pad and θJA is greatly
reduced. The θJA is highly related to layout and air flow
conditions. In layout, multiple vias (≥9) are strongly
recommended in the IC bottom pad. The bottom pad with its vias
should be placed in the ground copper plane with an area as
large as possible across multiple layers. The θJA can be reduced
further with air flow. Refer to Figures 8 and 9 for the thermal
performance with 100 CFM air flow.
Boost Converter Operation
“Typical Application Schematic III - Boost Buck Converter” on
page 5, shows the circuits where the boost works as a pre-stage
to provide input to the following Buck stage. This is for
applications when the input voltage could drop to a very low
voltage in some constants (in some battery powered systems as
for example), causing the output voltage to drop out of
regulation. The boost converter can be enabled to boost the input
voltage up to keep the output voltage in regulation. When system
input voltage recovers back to normal, the boost stage is
disabled while only the buck stage is switching.
The EXT_BOOST pin is used to set boost mode and monitor the
boost input voltage. At IC start-up before soft-start, the controller
will be latched in boost mode when the voltage is at or above
200mV; it will latch in synchronous buck mode if voltage on this
pin is below 200mV. In boost mode the low-side driver output
PWM has the same PWM signal with the buck regulator.
In boost mode, the EXT_BOOST pin is used to monitor boost input
voltage to turn on and turn off the boost PWM. The AUXVCC pin is
used to monitor the boost output voltage to turn on and turn off
the boost PWM.
Referring to Figure 26 on page 15, a resistor divider from boost
input voltage to the EXT_BOOST pin is used to detect the boost
input voltage. When the voltage on EXT_BOOST pin is below 0.8V,
the boost PWM is enabled with a fixed 500µs soft-start and the
boost duty cycle increases linearly from tMIN(ON)*Fs to ~50%. A
3µA sinking current is enabled at the EXT_BOOST pin for
hysteresis purposes. When the voltage on the EXT_BOOST pin
recovers to be above 0.8V, the boost PWM is disabled
immediately. Use Equation 3 to calculate the upper resistor RUP
(R1 in Figure 26) for a desired hysteresis VHYS at boost input
voltage.
VHYS
R UP [ MΩ ] = ---------------------3 [ μA ]
(EQ. 3)
Generally, the part can output 2.5A in typical application
conditions (VIN 8~30V, VO 5V, 500kHz, still air and +105°C
14
FN7640.1
April 25, 2013
ISL85402
Use Equation 4 to calculate the lower resistor RLOW (R2 in Figure 26)
according to a desired boost enable threshold.
R UP ⋅ 0.8
R LOW = --------------------------------------VFTH – 0.8
(EQ. 4)
Where VFTH is the desired falling threshold on boost input
voltage to turn on the boost, 3µA is the hysteresis current, and
0.8V is the reference voltage to be compared with.
Note the boost start-up threshold has to be selected in a way that
the buck is operating working well and kept in close loop
regulation before boost start-up. Otherwise, large in-rush current
at boost start-up could occur at boost input due to the buck open
loop saturation.
Similarly, a resistor divider from the boost output voltage to the
AUXVCC pin is used to detect the boost output voltage. When the
voltage on the AUXVCC pin is below 0.8V, the boost PWM is
enabled with a fixed 500µs soft-start, and a 3µA sinking current
is enabled at AUXVCC pin for hysteresis purposes. When the
voltage on the AUXVCC pin recovers to be above 0.8V, the boost
PWM is disabled immediately. Use Equation 3 to calculate the
upper resistor RUP (R3 in Figure 26) according to a desired
hysteresis VHY at boost output voltage. Use Equation 4 to
calculate the lower resistor RLOW (R4 in Figure 26) according to a
desired boost enable threshold at boost output.
From Equations 5 and 6, Equation 7 can be derived to estimate
the steady state boost output voltage as function of VBAT and
VOUT:
(EQ. 7)
V OUTBST = V BAT + V OUT
After the IC starts up, the boost buck converters can keep
working when the battery voltage drops extremely low because
the IC’s bias (VCC) LDO is powered by the boost output. For
example, a 3.3V output application battery drops to 2V, and the
VIN pin voltage is powered by the boost output voltage that is
5.2V (Equation 7), meaning that the VIN pin (buck input) still sees
5.2V to keep the IC working.
Note that in the previously mentioned case, the boost input current
could be high because the input voltage is very low
(VIN*IIN = VOUT*IOUT/Efficiency). If the design is to achieve the low
input operation with full load, the inductor and MOSFET have to be
selected with enough current ratings to handle the high current
appearing at boost input. The boost inductor current are the same
with the boost input current, which can be estimated as Equation 8,
where POUT is the output power, VBAT is the boost input voltage, and
EFF is the estimated efficiency of the whole boost and buck stages.
P OUT
IL IN = ------------------------------------V BAT ⋅ EFF
(EQ. 8)
Assuming VBAT is the boost input voltage, VOUTBST is the boost
output voltage and VOUT is the buck output voltage, the steady
state transfer function are:
Based on the same concerns of the boost input current, the IC
should be disabled before the boost input voltage rises above a
certain level. PFM is not available in boost mode.
1
V OUTBST = ------------------ ⋅ V BAT
1–D
(EQ. 5)
Oscillator and Synchronization
D
V OUT = D ⋅ V OUTBST = ------------------ ⋅ V BAT
1–D
(EQ. 6)
The oscillator has a default frequency of 500kHz with the FS pin
connected to VCC, or ground, or floating. The frequency can be
programmed to any frequency between 200kHz and 2.2MHz with
a resistor from FS pin to GND.
145000 – 16 ⋅ FS [ kHz ]
R FS [ kΩ ] = -----------------------------------------------------------------------------------FS [ kHz ]
BATTERY
(EQ. 9)
VOUT_BST
+
+
R1
EXT_BOOST
0.8V
R2
I_HYS = 3µA
R3
LOGIC
LGATE
AUXVCC
R4
0.8V
PWM
LGATE
DRIVE
I_HYS = 3µA
FIGURE 26. BOOST CONVERTER CONTROL
15
FN7640.1
April 25, 2013
ISL85402
dummy soft-start duration equaling to 5 regular soft-start periods.
After this dummy soft-start cycle, the true soft-start cycle is
attempted again. The IOC2 offers a robust and reliable protections
against the worst case conditions.
1200
1000
The frequency foldback is implemented for the ISL85402. When
overcurrent limiting, the switching frequency is reduced to be
proportional to output voltage in order to keep the inductor
current under limit threshold during overload condition. The low
limit of frequency under frequency foldback operation is 40kHz.
RFS (kΩ)
800
600
400
370
200
320
0
500
1000
1500
FS (kHz)
2000
2500
270
FIGURE 27. RFS vs FREQUENCY
The SYNC pin is bi-directional and it outputs the IC’s default or
programmed local clock signal when it’s free running. The IC
locks to an external clock injected to the SYNC pin (external clock
frequency recommended to be 10% higher than the free running
frequency). The delay from the rising edge of the external clock
signal to the PHASE rising edge is half of the free running switching
period pulse 220ns, (0.5Tsw+220ns). The maximum external clock
frequency is recommended to be 1.6 of the free running frequency.
When the part enters PFM pulse skipping mode, the
synchronization function is shut off and also no clock signal
output in SYNC pin.
With the SYNC pins simply connected together, multiple
ISL85402s can be synchronized. The slave ICs automatically
have 180° phase shift with respective to the master IC.
Fault Protection
Overcurrent Protection
The overcurrent function protects against any overload condition
and output short at worst case, by monitoring the current flowing
through the upper MOSFET.
There are 2 current limiting thresholds. The first one IOC1 is to
limit the high-side MOSFET peak current cycle-by-cycle. The
current limit threshold is set to default at 3.6A with ILIMIT pin
connected to GND or VCC, or left open. The current limit threshold
can also be programmed by a resistor RLIM at ILIMIT pin to
ground. Use Equation 10 to calculate the resistor.
300000
R LIM = -----------------------------------------------------I OC [ A ] + 0.018
(EQ. 10)
Note that IOC1 is higher with lower RLIM. Considering the OC
programming circuit tolerances over the temperature range 40°C to 105°C, 71.5k is the lowest resistor value recommended
to be used for RLIM to achieve the highest OC threshold. With
71.5k RLIM, the OC limit is 4.18A (TYP). A resistor lower than
71.5k would result in a default 3.6A OC1 threshold.
The second current protection threshold IOC2 is 15% higher than
IOC1 mentioned previously. Instantly after the high-side MOSFET
current reaches IOC2, the PWM is shut off after 2-cycle delay and the
IC enters hiccup mode. In hiccup mode, the PWM is disabled for
16
RLIM (kΩ)
0
220
170
120
70
0.0
1.0
2.0
3.0
IOC1 (A)
4.0
5.0
6.0
FIGURE 28. RLIM vs IOC1
Overvoltage Protection
If the voltage detected on the FB pin is over 110% of reference,
the high-side and low-side driver shuts down immediately and
won’t be allowed on until FB voltage drops to 0.8V. When the FB
voltage drops to 0.8V, the drivers are released to ON. If the 120%
overvoltage threshold is reached, the high-side and low-side
driver shuts down immediately and the IC is latched off. The IC
has to be reset for restart.
Thermal Protection
The ISL85402 PWM will be disabled if the junction temperature
reaches +155°C. A +15°C hysteresis insures that the device will
not restart until the junction temperature drops below +140°C.
Component Selections
The ISL85402 iSim model, which is available on the internet can
be used to simulate the behaviors to, which will assist with the
design.
Output Capacitors
An output capacitor is required to filter the inductor current.
Output ripple voltage and transient response are 2 critical factors
when considering output capacitance choice. The current mode
control loop allows for the usage of low ESR ceramic capacitors
and thus smaller board layout. Electrolytic and polymer
capacitors may also be used.
Additional consideration applies to ceramic capacitors. While
they offer excellent overall performance and reliability, the actual
in-circuit capacitance must be considered. Ceramic capacitors
are rated using large peak-to-peak voltage swings with no DC
bias. In the DC/DC converter application, these conditions do not
FN7640.1
April 25, 2013
ISL85402
reflect reality. As a result, the actual capacitance may be
considerably lower than the advertised value. Consult the
manufacturers data sheet to determine the actual in-application
capacitance. Most manufacturers publish capacitance vs DC bias
so that this effect can be easily accommodated. The effects of
AC voltage are not frequently published, but an assumption of
~20% further reduction will generally suffice. The result of these
considerations can easily result in an effective capacitance 50%
lower than the rated value. Nonetheless, they are a very good
choice in many applications due to their reliability and extremely
low ESR.
The following equations allow calculation of the required
capacitance to meet a desired ripple voltage level. Additional
capacitance may be used.
For the ceramic capacitors (low ESR):
ΔI
V OUTripple = ----------------------------------8∗ F SW∗ C OUT
(EQ. 11)
where ΔI is the inductor’s peak to peak ripple current, FSW is the
switching frequency and COUT is the output capacitor.
If using electrolytic capacitors then:
V OUTripple = ΔI*ESR
(EQ. 12)
Regarding transient response needs, a good starting point is to
determine the allowable overshoot in VOUT if the load is suddenly
removed. In this case, energy stored in the inductor will be
transferred to COUT causing its voltage to rise. After calculating
capacitance required for both ripple and transient needs, choose
the larger of the calculated values. The following equation
determines the required output capacitor value in order to
achieve a desired overshoot relative to the regulated voltage.
I OUT 2 * L
C OUT = ------------------------------------------------------------------------------------V OUT 2 * ( V OUTMAX ⁄ V OUT ) 2 – 1 )
(EQ. 13)
Increasing the value of inductance reduces the ripple current and
thus ripple voltage. However, the larger inductance value may
reduce the converter’s response time to a load transient. The
inductor current rating should be such that it will not saturate in
overcurrent conditions.
Low-Side Power MOSFET
In synchronous buck application, a power N MOSFET is needed
as the synchronous low side MOSFET and a good one should
have low Qgd, low rDS(ON) and small Rg (Rg_typ < 1.5Ω
recommended). Vgth_min is recommended to be higher than
1.2V. A good example is SQS462EN.
Output Voltage Feedback Resistor Divider
The output voltage can be programmed down to 0.8V by a
resistor divider from VOUT to FB according to Equation 15.
R UP ⎞
⎛
V OUT = 0.8 ⋅ ⎜ 1 + --------------------⎟
R
⎝
LOW⎠
In an application requiring least input quiescent current, large
resistors should be used for the divider. 232k is recommended
for the upper resistor.
Loop Compensation Design
The ISL85402 uses constant frequency peak current mode
control architecture to achieve fast loop transient response. An
accurate current sensing pilot device in parallel with the upper
MOSFET is used for peak current control signal and overcurrent
protection. The inductor is not considered as a state variable
since its peak current is constant, and the system becomes
single order system. It is much easier to design the compensator
to stabilize the loop compared with voltage mode control. Peak
current mode control has inherent input voltage feed-forward
function to achieve good line regulation. Figure 29 shows the
small signal model of a buck regulator.
where VOUTMAX/VOUT is the relative maximum overshoot
allowed during the removal of the load.
Input Capacitors
1:D
The inductor value determines the converter’s ripple current.
Choosing an inductor current requires a somewhat arbitrary
choice of ripple current, ΔI. A reasonable starting point is 30% to
40% of total load current. The inductor value can then be
calculated using Equation 14:
LP
RLP
^
vo
Vin d^
RT
Rc
Ro
Co
T i(S)
d^
Fm
+
Buck Output Inductor
V IN – V OUT V OUT
L = ---------------------------- × ------------Fs × ΔI
V IN
ILd^
^
iL
+
GAIN (VLOOP (S(fi))
Ceramic capacitors must be used at VIN pin of the IC and
multiple capacitors including 1µF and 0.1µF are recommended.
Place these capacitors as closely as possible to the IC.
^
Vin
+
^
i in
Depending on the system input power rail conditions, the
aluminum electrolytic type capacitor is normally needed to
provide the stable input voltage. Thus, restrict the switching
frequency pulse current in a small area over the input traces for
better EMC performance. The input capacitor should be able to
handle the RMS current from the switching power devices.
(EQ. 15)
Tv (S)
He(S)
v^comp
-Av(S)
FIGURE 29. SMALL SIGNAL MODEL OF BUCK REGULATOR
(EQ. 14)
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FN7640.1
April 25, 2013
ISL85402
PWM Comparator Gain Fm:
If Ti(S)>>1, then Equation 23 can be simplified as Equation 24:
The PWM comparator gain Fm for peak current mode control is
given by Equation 16:
S
1 + -----------R o + R LP
ω esr A v ( S )
1
L v ( S ) = ----------------------- ---------------------- --------------- , ω p ≈ ------------Rt
Ro C o
S He ( S )
-----1+
ωp
(EQ. 16)
1
d̂
F m = ---------------- = -----------------------------( S e + S n )T s
v̂ comp
Where, Se is the slew rate of the slope compensation and Sn is
given by Equation 17:
V in – V o
S n = R t -------------------L
(EQ. 17)
P
(EQ. 24)
Equation 24 shows that the system is a single order system.
Therefore, a simple type II compensator can be easily used to
stabilize the system. A type III compensator is needed to expand
the bandwidth for current mode control in some cases.
where, Rt is the gain of the current amplifier.
C1
R2
R3
C3
CURRENT SAMPLING TRANSFER FUNCTION HE(S):
In current loop, the current signal is sampled every switching
cycle. It has the following transfer function in Equation 18:
2
VO
VCOMP
R1
VREF
RBIAS
(EQ. 18)
S
S
H e ( S ) = ------- + -------------- + 1
2 ω Q
n n
ωn
where, Qn and ωn are given by
2
Q n = – ---, ω n = πf s
π
FIGURE 30. TYPE III COMPENSATOR
Power Stage Transfer Functions
Transfer function F1(S) from control to output voltage is:
S
1 + -----------ω esr
v̂ o
F 1 ( S ) = ------ = V in -------------------------------------2
d̂
S
S
------- + -------------- + 1
2 ω Q
o
p
ωo
(EQ. 19)
C
LP
LP Co
Transfer function F2(S) from control to inductor current is given
by Equation 20:
S
1 + -----V in
ωz
Î o
F 2 ( S ) = ---- = ----------------------- -------------------------------------R o + R LP 2
d̂
S - ------------S -----+
+1
2 ω Q
o p
ωo
S ⎞⎛
S
⎛ 1 + -----------1 + ------------⎞
⎝
ω cz1⎠ ⎝
ω cz2⎠
v̂ comp
1
A v ( S ) = ---------------- = ------------------ --------------------------------------------------------SR 1 C
S ⎞
v̂ O
⎛ 1 + --------1
⎝
ω ⎠
(EQ. 25)
cp
1
1
Where, ω esr = ------------- ,Q p ≈ R o -----o- ,ω o = ----------------Rc Co
A type III compensator with 2 zeros and 1 pole is recommended
for this part, as shown in Figure 30. Its transfer function is
expressed as Equation 25:
(EQ. 20)
where,
1
1
1
ω cz1 = -------------- , ω cz2 = --------------------------------, ω cp = -------------( R 1 + R 3 )C 3
R2 C1
R3 C3
Compensator design goal:
⎛1
1⎞
- f
Loop bandwidth fc: ⎝ --4- to -----10⎠ s
Gain margin: >10dB
1
where ω z = ------------Ro Co .
Phase margin: 45°
Current loop gain Ti(S) is expressed as Equation 21:
T i ( S ) = R t F m F 2 ( S )H e ( S )
(EQ. 21)
The voltage loop gain with open current loop is expressed in
Equation 22:
T v ( S ) = KF m F 1 ( S )A v ( S )
(EQ. 22)
The Voltage loop gain with current loop closed is given by
Equation 23:
Tv ( S )
L v ( S ) = ----------------------1 + Ti ( S )
(EQ. 23)
The compensator design procedure is as follows:
1. Position ωCZ2 and ωCP to derive R3 and C3.
Put the compensator zero ωCZ2 at (1 to 3)/(RoCo)
(EQ. 26)
3
ω cz2 = ------------Ro Co
Put the compensator pole ωCP at ESR zero or 0.35 to 0.5 times
of switching frequency, whichever is lower. In all-ceramic-cap
design, the ESR zero is normally higher than half of the switching
frequency. R3 and C3 can be derived as following:
1
Case A: ESR zero --------------------- less than (0.35 to 0.5)fs
2πR c C o
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April 25, 2013
ISL85402
R o C o – 3R c C o
C 3 = ------------------------------------3R 1
(EQ. 27)
3R c R 1
R 3 = ----------------------R o – 3R c
(EQ. 28)
Loop Gain
80
60
1
Case B: ESR zero --------------------- larger than (0.35 to 0.5)fs
40
2πR c C o
(EQ. 29)
R1
R 3 = ---------------------------------------0.73R o C o f s – 1
(EQ. 30)
dB
20
0.33R o C o f s – 0.46
C 3 = ------------------------------------------------fs R
0
1
-20
-40
2. Derive R2 and C1.
-60
100
The loop gain Lv(S) at cross over frequency of fc has unity gain.
Therefore, C1 is determined by Equation 31.
( R 1 + R 3 )C 3
C 1 = --------------------------------2πf c R t R 1 C
1,000
10,000
100,000
1,000,000
Frequency
(EQ. 31)
Phase Margin
180
The compensator zero ωCZ1 can boost the phase margin and
bandwidth. To put ωCZ1 at 2 times of cross cover frequency fc is a
good start point. It can be adjusted according to specific design.
R1 can be derived from Equation 32.
(EQ. 32)
1
R 2 = ------------------4πf c C 1
160
140
120
100
D
Degree
o
80
60
Example: Vin = 12V, Vo = 5V, Io = 2A, fs = 500kHz,
Co = 60µF/3mΩ, L = 10µH, Rt = 0.20V/A, fc = 50kHz, R1=105k,
RBIAS = 20kΩ.
Select the crossover frequency to be 35kHz. Since the output
capacitors are all ceramic, use Equation 29 and 30 to derive R3
to be 20k and C3 to be 470pF.
Then use Equation 31 and 32 to calculate C1 to be 180pF and
R2 to be 12.7k. Select 150pF for C1 and 15k for R2.
There is approximately 30pF parasitic capacitance between
COMP to FB pins that contributes to a high frequency pole.
Figure 31 shows the simulated bode plot of the loop. It is shown
that it has 26kHz loop bandwidth with 70° phase margin and -28
dB gain margin.
Note in applications where the PFM mode is desired especially
when type III compensation network is used, the value of the
capacitor between the COMP pin and the FB pin (not the
capacitor in series with the resistor between COMP and FB)
should be minimal to reduce the noise coupling for proper PFM
operation. No external capacitor between COMP and FB is
recommended at PFM applications.
19
40
20
0
100
1,000
10,000
100,000
1,000,000
Frequency
FIGURE 31. SIMULATED LOOP BODE PLOT
Boost Inductor
Besides the need to sustain the current ripple to be within a
certain range (30% to 50%), the boost inductor current at its
soft-start is a more important perspective to be considered in
selection of the boost inductor. Each time the boost starts up,
there is a fixed 500µs soft-start time when the duty cycle
increases linearly from tMIN(ON)*Fs to ~50%. Before and after
boost start-up, the boost output voltage will jump from
VIN_BOOST to voltage (VIN_BOOST + VOUT_BUCK). The design target
in boost soft-start is to ensure the boost input current is
sustained to minimum but capable to charge the boost output
voltage to have a voltage step equaling to VOUT_BUCK. A big
inductor will block the inductor current to increase and not high
enough to be able to charge the output capacitor to the final
steady state value (VIN_BOOST + VOUT_BUCK) within 500µs. A
6.8µH inductor is a good starting point for its selection in design.
The boost inductor current at start-up must be checked by
oscilloscope to ensure it is under acceptable range. It is
suggested to run the iSim model, which is available on the
internet to assist in designing the proper inductor value.
FN7640.1
April 25, 2013
ISL85402
Boost Output Capacitor
Based on the same theory in boost start-up previously described
in the boost inductor selection, a large capacitor at boost output
will cause high in-rush current at boost PWM start-up. 22µF is a
good choice for applications with a buck output voltage less than
10V. Also some minimum amount of capacitance has to be used
in boost output to keep the system stable. It is suggested to run
the iSim model, which is available on the internet to assist in
designing the proper capacitor value.
Layout Suggestions
1. Place the input ceramic capacitors as closely as possible to
the IC VIN pin and power ground connecting to the power
MOSFET or Diode. Keep this loop (input ceramic capacitor, IC
VIN pin and MOSFET/Diode) as tiny as possible to achieve the
least voltage spikes induced by the trace parasitic
inductance.
as possible in multiple layers to effectively reduce the thermal
impedance.
6. Place the 4.7µF ceramic decoupling capacitor at the VCC pin
(the closest place to the IC). Put multiple vias (≥3) close to the
ground pad of this capacitor.
7. Keep the bootstrap capacitor close to the IC.
8. Keep the LGATE drive trace as short as possible and try to
avoid using via in the LGATE drive path to achieve the lowest
impedance.
9. Place the positive voltage sense trace close to the place to be
strictly regulated.
10. Place all the peripheral control components close to the IC.
2. Place the input aluminum capacitors closely as possible to
the IC VIN pin.
3. Keep the phase node copper area small but large enough to
handle the load current.
4. Place the output ceramic and aluminum capacitors close to
the power stage components as well.
FIGURE 32. PCB VIA PATTERN
5. Place vias (≥9) in the bottom pad of the IC. The bottom pad
should be placed in ground copper plane with an area as large
20
FN7640.1
April 25, 2013
ISL85402
Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make
sure you have the latest revision.
DATE
REVISION
CHANGE
April 25, 2012
FN7640.1
1. Expanded the maximum temperature from 85°C to 105°C for the electrical characteristics and Ordering
Information.
2. Added application design guide for selection of inductor and capacitor and loop compensation.
3. Added typical electrical specification of EN pull-up current, synchronization.
4. Added boost-buck efficiency curve and AUVVCC switchover description.
5. Under "Output Voltage" description, corrected "(1/Fs tMINOFF)" To " (1 - Fs * tMIN(OFF))".
6. Under "Boost Converter Operation", corrected "(VIN*IIN = VOUT*IOUT*Efficiency)" to "(VIN*IIN =
VOUT*IOUT/Efficiency)".
7. Added recommendation of the maximum programmable OC threshold to be 4.18A(TYP) with 71.5k RLIM.
8. Corrected sentence in first paragraph on page 1 from: “ The ISL85402 supports a wide input range of 3V to
40V in buck mode.“ to “ The ISL85402 supports a wide input range of 3V to 36V in buck mode.“
9. Removed following sentence from last paragraph of “Power Stage Transfer Functions” on page 19: “Deleted
following sentence from last paragraph of “Power Stage Transfer Functions” on page 19: “A capacitor (<1nF)
at the FB pin to ground also helps proper PFM mode operation".
September 29, 2011
FN7640.0
Initial Release
About Intersil
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21
FN7640.1
April 25, 2013
ISL85402
Package Outline Drawing
L20.4x4C
20 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 0, 11/06
4X
4.00
2.0
16X 0.50
A
B
16
6
PIN #1 INDEX AREA
20
6
PIN 1
INDEX AREA
1
4.00
15
2 .70 ± 0 . 15
11
(4X)
5
0.15
6
10
0.10 M C A B
4 20X 0.25 +0.05 / -0.07
20X 0.4 ± 0.10
TOP VIEW
BOTTOM VIEW
SEE DETAIL "X"
0.10 C
0 . 90 ± 0 . 1
C
BASE PLANE
( 3. 8 TYP )
(
SEATING PLANE
0.08 C
2. 70 )
( 20X 0 . 5 )
SIDE VIEW
( 20X 0 . 25 )
C
0 . 2 REF
5
( 20X 0 . 6)
0 . 00 MIN.
0 . 05 MAX.
DETAIL "X"
TYPICAL RECOMMENDED LAND PATTERN
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3. Unless otherwise specified, tolerance: Decimal ± 0.05
4. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
5. Tiebar shown (if present) is a non-functional feature.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 indentifier may be
either a mold or mark feature.
22
FN7640.1
April 25, 2013