AD AD600JN Dual, low noise, wideband variable gain amplifier Datasheet

Dual, Low Noise, Wideband
Variable Gain Amplifiers
AD600/AD602
FUNCTIONAL BLOCK DIAGRAM
FEATURES
APPLICATIONS
Ultrasound and sonar time-gain controls
High performance audio and RF AGC systems
Signal measurement
GENERAL DESCRIPTION
The AD600/AD602 1 dual channel, low noise, variable gain
amplifiers are optimized for use in ultrasound imaging systems
but are applicable to any application requiring precise gain, low
noise and distortion, and wide bandwidth. Each independent
channel provides a gain of 0 dB to +40 dB in the AD600 and
−10 dB to +30 dB in the AD602. The lower gain of the AD602
results in an improved signal-to-noise ratio (SNR) at the output.
However, both products have the same 1.4 nV/√Hz input noise
spectral density. The decibel gain is directly proportional to the
control voltage, accurately calibrated, and supply and
temperature stable.
To achieve the difficult performance objectives, a proprietary
circuit form, the X-AMP®, was developed. Each channel of the
X-AMP comprises a variable attenuator of 0 dB to −42.14 dB
followed by a high speed fixed gain amplifier. In this way, the
amplifier never has to cope with large inputs and can benefit
from the use of negative feedback to precisely define the gain
and dynamics. The attenuator is realized as a 7-stage R-2R
ladder network having an input resistance of 100 Ω, laser
trimmed to ±2%. The attenuation between tap points is 6.02 dB;
the gain-control circuit provides continuous interpolation between
these taps. The resulting control function is linear in dB.
GAT1
SCALING
REFERENCE
C1HI
PRECISION PASSIVE
INPUT ATTENUATOR
GATING
INTERFACE
VG
A1OP
C1LO
A1CM
GAIN CONTROL
INTERFACE
0dB
–12.04dB
–6.02dB
–18.06dB
–22.08dB
–36.12dB
–30.1dB
–42.14dB
A1HI
A1LO
500Ω
R-2R LADDER NETWORK
62.5Ω
RF2
2.24kΩ (AD600)
694Ω (AD602)
RF1
20Ω
FIXED-GAIN
AMPLIFIER
41.07dB (AD600)
31.07dB (AD602)
00538-001
2 channels with independent gain control
Linear in dB gain response
2 gain ranges
AD600: 0 dB to 40 dB
AD602: –10 dB to +30 dB
Accurate absolute gain: ±0.3 dB
Low input noise: 1.4 nV/√Hz
Low distortion: −60 dBc THD at ±1 V output
High bandwidth: dc to 35 MHz (−3 dB)
Stable group delay: ±2 ns
Low power: 125 mW (maximum) per amplifier
Signal gating function for each amplifier
Drives high speed ADCs
MIL-STD-883-compliant and DESC versions available
Figure 1.
The gain-control interfaces are fully differential, providing an
input resistance of ~15 MΩ and a scale factor of 32 dB/V (that
is, 31.25 mV/dB) defined by an internal voltage reference. The
response time of this interface is less than 1 μs. Each channel
also has an independent gating facility that optionally blocks
signal transmission and sets the dc output level to within a few
millivolts of the output ground. The gating control input is
TTL- and CMOS-compatible.
The maximum gain of the AD600 is 41.07 dB, and the
maximum gain of the AD602 is 31.07 dB; the −3 dB bandwidth
of both models is nominally 35 MHz, essentially independent of
the gain. The SNR for a 1 V rms output and a 1 MHz noise
bandwidth is typically 76 dB for the AD600 and 86 dB for the
AD602. The amplitude response is flat within ±0.5 dB from
100 kHz to 10 MHz; over this frequency range, the group delay
varies by less than ±2 ns at all gain settings.
Each amplifier channel can drive 100 Ω load impedances with
low distortion. For example, the peak specified output is ±2.5 V
minimum into a 500 Ω load or ±1 V into a 100 Ω load. For a
200 Ω load in shunt with 5 pF, the total harmonic distortion for
a ±1 V sinusoidal output at 10 MHz is typically −60 dBc.
The AD600J/AD602J are specified for operation from 0°C to 70°C
and are available in 16-lead PDIP (N) and 16-lead SOIC_W
packages. The AD600A/AD602A are specified for operation from
−40°C to +85°C and are available in 16-lead CERDIP (Q) and
16-lead SOIC_W packages. The AD600S/ AD602S are specified
for operation from −55°C to +125°C, are available in a 16-lead
CERDIP (Q) package, and are MIL-STD-883-compliant. The
AD600S/AD602S are also available under DESC SMD 5962-94572.
1
Patented.
Rev. E
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
©2006 Analog Devices, Inc. All rights reserved.
AD600/AD602
TABLE OF CONTENTS
Features .............................................................................................. 1
Applications..................................................................................... 15
Applications....................................................................................... 1
Time-Gain Control (TGC) and Time-Variable
Gain (TVG) ................................................................................. 15
General Description ......................................................................... 1
Functional Block Diagram .............................................................. 1
Revision History ............................................................................... 2
Specifications..................................................................................... 3
Absolute Maximum Ratings............................................................ 5
ESD Caution.................................................................................. 5
Pin Configuration and Function Descriptions............................. 6
Typical Performance Characteristics ............................................. 7
Theory of Operation ...................................................................... 10
Noise Performance ..................................................................... 10
The Gain-Control Interface ...................................................... 11
Signal-Gating Inputs .................................................................. 11
Increasing Output Drive............................................................ 15
Driving Capacitive Loads.......................................................... 15
Realizing Other Gain Ranges ................................................... 16
An Ultralow Noise VCA............................................................ 16
A Low Noise, 6 dB Preamplifier ............................................... 16
A Low Noise AGC Amplifier with 80 dB Gain Range .......... 17
A Wide Range, RMS-Linear dB Measurement System (2 MHz
AGC Amplifier with RMS Detector)....................................... 19
100 dB to 120 dB RMS Responding Constant Bandwidth
AGC Systems with High Accuracy dB Outputs ..................... 21
A 100 dB RMS/AGC System with Minimal Gain Error
(Parallel Gain with Offset) ........................................................ 22
Common-Mode Rejection ........................................................ 11
A 120 dB RMS/AGC System with Optimal SNR
(Sequential Gain) ....................................................................... 23
Achieving 80 dB Gain Range .................................................... 12
Outline Dimensions ....................................................................... 27
Sequential Mode (Maximum SNR) ......................................... 12
Ordering Guide .......................................................................... 28
Parallel Mode (Simplest Gain-Control Interface).................. 13
Low Ripple Mode (Minimum Gain Error) ............................. 13
REVISION HISTORY
1/06—Rev. D to Rev. E
Updated Format..................................................................Universal
Changes to Table 2............................................................................ 5
Changes to The Gain-Control Interface Section........................ 11
Updated Outline Dimensions ....................................................... 27
Changes to Ordering Guide .......................................................... 28
5/02—Rev. B to Rev. C
Changes to Specifications.................................................................2
Renumber Tables and TPCs...................................................Global
8/01—Rev. A to Rev. B
Changes to Accuracy Section of AD600A/AD602A column......2
3/04—Rev. C to Rev. D
Changes to Specifications ................................................................ 2
Changes to Ordering Guide ............................................................ 3
Changes to Figure 3.......................................................................... 8
Changes to Figure 29...................................................................... 18
Updated Outline Dimensions ....................................................... 20
Rev. E | Page 2 of 28
AD600/AD602
SPECIFICATIONS
Each amplifier section at TA = 25°C, VS = ±5 V, −625 mV ≤ VG ≤ +625 mV, RL = 500 Ω, and CL = 5 pF, unless otherwise noted.
Specifications for the AD600/AD602 are identical, unless otherwise noted.
Table 1.
Parameter
INPUT CHARACTERISTICS
Input Resistance
Input Capacitance
Input Noise Spectral Density 2
Noise Figure
Common-Mode Rejection Ratio
OUTPUT CHARACTERISTICS
−3 dB Bandwidth
Slew Rate
Peak Output 3
Output Impedance
Output Short-Circuit Current
Group Delay Change vs. Gain
Group Delay Change vs. Frequency
Total Harmonic Distortion
ACCURACY
AD600
Gain Error
Maximum Output Offset Voltage 4
Output Offset Variation
AD602
Gain Error
Maximum Output Offset Voltage4
Output Offset Variation
GAIN CONTROL INTERFACE
Gain Scaling Factor
Common-Mode Range
Input Bias Current
Input Offset Current
Differential Input Resistance
Response Rate
Conditions
AD600J/AD602J 1
Min
Typ Max
Pin 2 to Pin 3; Pin 6 to Pin 7
98
RS = 50 Ω, maximum gain
RS = 200 Ω, maximum gain
f = 100 kHz
VOUT = 100 mV rms
RL ≥ 500 Ω
f ≤ 10 MHz
±2.5
f = 3 MHz; full gain range
VG = 0 V, f = 1 MHz to 10 MHz
RL= 200 Ω, VOUT = ±1 V peak, RPD = 1 kΩ
100
2
1.4
5.3
2
30
102
35
275
±3
2
50
±2
±2
−60
AD600A/AD602A1
Min
Typ Max
95
±2.5
100
2
1.4
5.3
2
30
105
35
275
±3
2
50
±2
±2
−60
Unit
Ω
pF
nV/√Hz
dB
dB
dB
MHz
V/μs
V
Ω
mA
ns
ns
dBc
0 dB to 3 dB gain
3 dB to 37 dB gain
37 dB to 40 dB gain
VG = –625 mV to +625 mV
VG = –625 mV to +625 mV
0
−0.5
−1
+0.5
±0.2
−0.5
10
10
+1
+0.5
0
50
50
−0.5
−1.0
−1.5
+0.5
±0.2
−0.5
10
10
+1.5
+1.0
+0.5
65
65
dB
dB
dB
mV
mV
–10 dB to –7 dB gain
–7 dB to +27 dB gain
27 dB to 30 dB gain
VG = −625 mV to +625 mV
VG = −625 mV to +625 mV
0
−0.5
−1
+0.5
±0.2
−0.5
5
5
+1
+0.5
0
30
30
–0.5
−1.0
−1.5
+0.5
±0.2
−0.5
10
10
+1.5
+1.0
+0.5
45
45
dB
dB
dB
mV
mV
+3 dB to +37 dB (AD600);
−7 dB to +27 dB (AD602)
31.7
32
32.3
30.5
32
33.5
dB/V
+2.5
1
50
−0.75
+2.5
1
50
V
μA
nA
MΩ
dB/μs
−0.75
Pin 1 to Pin 16; Pin 8 to Pin 9
Full 40 dB gain change
Rev. E | Page 3 of 28
0.35
10
15
40
0.35
10
15
40
AD600/AD602
Parameter
SIGNAL GATING INTERFACE
Logic Input LO (Output On)
Logic Input HI (Output Off )
Response Time
Input Resistance
Output Gated Off
Output Offset Voltage
Output Noise Spectral Density
Signal Feedthrough @ 1 MHz
AD600
AD602
POWER SUPPLY
Specified Operating Range
Quiescent Current
AD600J/AD602J 1
Min
Typ Max
Conditions
AD600A/AD602A1
Min
Typ Max
0.8
2.4
On to off, off to on
Pin 4 to Pin 3; Pin 5 to Pin 6
±10
65
V
V
μs
kΩ
±400
mV
nV/√Hz
0.3
30
±10
65
±100
−80
−70
±4.75
11
1
0.8
2.4
0.3
30
−80
−70
±5.25
12.5
±4.75
11
Unit
dB
dB
±5.25
14
V
mA
Specifications shown in boldface are tested on all production units at final electrical test. Results from those tests are used to calculate outgoing quality levels. All
minimum and maximum specifications guaranteed, although only those shown in boldface are tested on all production units.
2
Typical open- or short-circuited input; noise is lower when the system is set to maximum gain and the input is short-circuited. This figure includes the effects of both
voltage and current noise sources.
3
With an additional 1 kΩ pull-down resistor, if RL < 500 Ω.
4
The dc gain of the main amplifier in the AD600 is × 113; therefore, an input offset of only 100 μV becomes an 11.3 mV output offset. In the AD602, the amplifier’s gain
is × 35.7; therefore, an input offset of 100 μV becomes a 3.57 mV output offset.
Rev. E | Page 4 of 28
AD600/AD602
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter
Supply Voltage ±VS
Input Voltages
Pin 1, Pin 8, Pin 9, Pin 16
Pin 2, Pin 3, Pin 6, Pin 7
Pin 4, Pin 5
Internal Power Dissipation
Operating Temperature Range
J Grade
A Grade
S Grade
Storage Temperature Range
Lead Temperature (Soldering 60 sec)
θJA
16-Lead PDIP
16-Lead SOIC_W
16-Lead CERDIP
Rating
±7.5 V
±VS
±2 V continuous
±VS for 10 ms
±VS
600 mW
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
0°C to 70°C
−40°C to +85°C
−55°C to +125°C
−65°C to +150°C
300°C
85°C/W
100°C/W
120°C/W
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. E | Page 5 of 28
AD600/AD602
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
C1LO
1
16 C1HI
A1HI
2
+
A1LO
3
–
GAT1
4
GAT2
5
A2LO
6
A1
15 A1CM
14 A1OP
REF
13 VPOS
12 VNEG
–
11 A2OP
A2
7
8
10 A2CM
+
AD600 /
AD602
9
C2HI
00538-002
A2HI
C2LO
Figure 2. Pin Configuration
Table 3. Pin Function Descriptions
Pin No.
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
Mnemonic
C1LO
A1HI
A1LO
GAT1
GAT2
A2LO
A2HI
C2LO
C2HI
A2CM
A2OP
VNEG
VPOS
A1OP
A1CM
C1HI
Description
CH1 Gain-Control Input LO (Positive Voltage Reduces CH1 Gain)
CH1 Signal Input HI (Positive Voltage Increases CH1 Output)
CH1 Signal Input LO (Usually Connected to CH1 Input Ground)
CH1 Gating Input (A Logic HI Shuts Off CH1 Signal Path)
CH2 Gating Input (A Logic HI Shuts Off CH2 Signal Path)
CH2 Signal Input LO (Usually Connected to CH2 Input Ground)
CH2 Signal Input HI (Positive Voltage Increases CH2 Output)
CH2 Gain-Control Input LO (Positive Voltage Reduces CH2 Gain)
CH2 Gain-Control Input HI (Positive Voltage Increases CH2 Gain)
CH2 Common (Usually Connected to CH2 Output Ground)
CH2 Output
Negative Supply for Both Amplifiers
Positive Supply for Both Amplifiers
CH1 Output
CH1 Common (Usually Connected to CH1 Output Ground)
CH1 Gain-Control Input HI (Positive Voltage Increases CH1 Gain)
Rev. E | Page 6 of 28
AD600/AD602
TYPICAL PERFORMANCE CHARACTERISTICS
0.45
10.0
0.35
9.8
9.6
GROUP DELAY (ns)
0.15
0.05
–0.05
–0.15
–0.25
9.2
9.0
8.8
8.6
8.4
00538-003
–0.35
–0.45
–0.7
9.4
–0.5
–0.3
–0.1
0.1
0.3
GAIN CONTROL VOLTAGE (V)
0.5
00538-006
GAIN ERROR (dB)
0.25
8.2
8.0
–0.7
0.7
Figure 3. Gain Error vs. Gain Control Voltage
–0.5
–0.3
–0.1
0.1
0.3
GAIN CONTROL VOLTAGE (V)
0.5
0.7
Figure 6. AD600 and AD602 Typical Group Delay vs. VC
VG = 0V
10dB/DIV
CENTER
FREQ 1MHz
10kHz/DIV
20dB
17dB
0°
–45°
00538-004
00538-007
–90°
100k
1M
10M
FREQUENCY (Hz)
100M
Figure 7. Third-Order Intermodulation Distortion, VOUT = 2 V p-p, RL = 500 Ω
Figure 4. AD600 Frequency and Phase Response vs. Gain
10dB
7dB
0°
–45°
00538-005
–90°
100k
1M
10M
FREQUENCY (Hz)
–1.2
–1.4
–1.6
–1.8
–2.0
–2.2
–2.4
–2.6
–2.8
–3.0
–3.2
–3.4
100M
00538-008
NEGATIVE OUTPUT VOLTAGE LIMIT (V)
–1.0
0
50
100
200
500
1000
LOAD RESISTANCE (Ω)
2000
Figure 8. Typical Output Voltage vs. Load Resistance
(Negative Output Swing Limits First)
Figure 5. AD602 Frequency and Phase Response vs. Gain
Rev. E | Page 7 of 28
AD600/AD602
102
101
50mV
GAIN = 40dB
OUTPUT
100
99
GAIN = 20dB
98
97
90
GAIN = 0dB
96
10
0%
00538-009
94
93
92
100k
1M
10M
FREQUENCY (Hz)
5V
100ns
00538-012
95
INPUT
INPUT IMPEDANCE (Ω)
100
100M
Figure 12. Gating Feedthrough to Output, Gating Off to On
Figure 9. Input Impedance vs. Frequency
6
50mV
4
100
AD600
90
2
OUTPUT
3
AD602
1
0
–2
–3
–4
–0.7
5V
–0.5
–0.3
–0.1
0.1
0.3
GAIN CONTROL VOLTAGE (V)
0.5
1V VOUT
100ns
0.7
Figure 10. Output Offset Voltage vs. Gain Control Voltage
(Control Channel Feedthrough)
Figure 13. Gating Feedthrough to Output, Gating On to Off
1µs
1V
100
100
90
90
10
0%
00538-011
1V VC
INPUT
10
0%
Figure 11. Gain Control Channel Response Time. Top: Output Voltage, 2 V
max, Bottom: Gain Control Voltage VC = ±625 mV
Rev. E | Page 8 of 28
100mV
500ns
Figure 14. Transient Response, Medium and High Gain
00538-014
OUTPUT
OUTPUT
INPUT
10
0%
00538-013
INPUT
–1
00538-010
OUTPUT OFFSET VOLTAGE (mV)
5
AD600/AD602
10
AD600: G = 20dB
AD602: G = 10dB
BOTH: VCM = 100mV rms
VS = ±5V
RL = 500Ω
TA = 25°C
5
0
100
90
–5
CMRR (dB)
–10
AD600
–15
–20
–25
10
0%
–30
200ns
AD602
00538-015
1V
00538-018
INPUT
OUTPUT
500mV
–35
–40
1k
10k
Figure 15. Input Stage Overload Recovery Time
100k
1M
FREQUENCY (Hz)
10M
100M
Figure 18. CMRR vs. Frequency
20
10
0
100
90
–20
–30
AD602
AD600: G = 40dB
AD602: G = 30dB
BOTH: RL = 500Ω
VIN = 0V
RS = 50Ω
–60
200mV
500ns
–70
–80
100k
1M
10M
FREQUENCY (Hz)
00538-019
–40
–50
10
0%
00538-016
INPUT
AD600
–10
PSRR (dB)
OUTPUT
1V
100M
Figure 19. PSRR vs. Frequency
Figure 16. Output Stage Overload Recovery Time
10
500mV
0
–10
100
90
CROSSTALK (dB)
–30
–40
AD600
–50
–60
10
0%
–70
500ns
AD602
00538-020
1V
00538-017
INPUT
OUTPUT
–20
AD600: CH1 G = 40dB, VIN = 0
CH2 G = 20dB, VIN = 100mV
AD602: CH1 G = 30dB, VIN = 0
CH2 G = 0dB, VIN = 316mV
BOTH: VOUT = 1V rms1, RS = 50Ω
RL = 500Ω
CH1 VOUT
CROSSTALK = 20log
CH2 V IN
–80
–90
100k
1M
10M
FREQUENCY (Hz)
Figure 17. Transient Response Minimum Gain
Figure 20. Crosstalk Between A1 and A2 vs. Frequency
Rev. E | Page 9 of 28
100M
AD600/AD602
THEORY OF OPERATION
The AD600/AD602 have the same general design and features.
They comprise two fixed gain amplifiers, each preceded by a
voltage-controlled attenuator of 0 dB to 42.14 dB with independent
control interfaces, each having a scaling factor of 32 dB per volt.
The AD600 amplifiers are laser trimmed to a gain of 41.07 dB
(×113), providing a control range of −1.07 dB to +41.07 dB
(0 dB to +40 dB with overlap). The AD602 amplifiers have a
gain of 31.07 dB (×35.8) and provide an overall gain of
−11.07 dB to +31.07 dB (−10 dB to +30 dB with overlap).
The advantage of this topology is that the amplifier can use
negative feedback to increase the accuracy of its gain. In
addition, because the amplifier does not have to handle large
signals at its input, the distortion can be very low. Another
feature of this approach is that the small-signal gain and phase
response, and thus the pulse response, are essentially
independent of gain.
Figure 21 is a simplified schematic of one channel. The input
attenuator is a 7-stage R-2R ladder network, using untrimmed
resistors of nominally R = 62.5 Ω, which results in a characteristic
resistance of 125 Ω ± 20%. A shunt resistor is included at the
input and laser trimmed to establish a more exact input
resistance of 100 Ω ± 2%, which ensures accurate operation
(gain and HP corner frequency) when used in conjunction with
external resistors or capacitors.
GAT1
SCALING
REFERENCE
C1HI
PRECISION PASSIVE
INPUT ATTENUATOR
GATING
INTERFACE
VG
A1OP
C1LO
A1CM
0dB
–12.04dB
–6.02dB
–18.06dB
–22.08dB
–36.12dB
–30.1dB
–42.14dB
A1HI
A1LO
500Ω
R-2R LADDER NETWORK
62.5Ω
RF2
2.24kΩ (AD600)
694Ω (AD602)
RF1
20Ω
FIXED-GAIN
AMPLIFIER
41.07dB (AD600)
31.07dB (AD602)
00538-021
GAIN CONTROL
INTERFACE
Figure 21. Simplified Block Diagram of Single Channel of the AD600/AD602
The nominal maximum signal at input A1HI is 1 V rms (±1.4 V
peak) when using the recommended ±5 V supplies; although,
operation to ±2 V peak is permissible with some increase in HF
distortion and feedthrough. Each attenuator is provided with a
separate signal LO connection for use in rejecting common
mode, the voltage between input and output grounds. Circuitry
is included to provide rejection of up to ±100 mV.
The signal applied at the input of the ladder network is
attenuated by 6.02 dB by each section; thus, the attenuation to
each of the taps is progressively 0 dB, 6.02 dB, 12.04 dB, 18.06 dB,
24.08 dB, 30.1 dB, 36.12 dB, and 42.14 dB. A unique circuit
technique is employed to interpolate between these tap points,
indicated by the slider in Figure 21, providing continuous
attenuation from 0 dB to 42.14 dB.
To understand the AD600, it helps to think in terms of a
mechanical means for moving this slider from left to right; in
fact, it is voltage controlled. The details of the control interface
are discussed later. Note that the gain is exactly determined at
all times, and a linear decibel relationship is guaranteed
automatically between the gain and the control parameter that
determines the position of the slider. In practice, the gain
deviates from the ideal law by about ±0.2 dB peak (see Figure 28).
Note that the signal inputs are not fully differential. A1LO,
A1CM (for CH1), A2LO, and A2CM (for CH2) provide
separate access to the input and output grounds. This recognizes
that even when using a ground plane, small differences arise in the
voltages at these nodes. It is important that A1LO and A2LO be
connected directly to the input ground(s). Significant impedance in
these connections reduces the gain accuracy. A1CM and A2CM
should be connected to the load ground(s).
NOISE PERFORMANCE
An important reason for using this approach is the superior
noise performance that can be achieved. The nominal resistance
seen at the inner tap points of the attenuator is 41.7 Ω (one third of
125 Ω), which, at 27°C, exhibits a Johnson noise spectral density
(NSD) of 0.84 nV/√Hz (that is, √4kTR), a large fraction of the
total input noise. The first stage of the amplifier contributes
another 1.12 nV/√Hz, for a total input noise of 1.4 nV/√Hz.
The noise at the 0 dB tap depends on whether the input is
short-circuited or open-circuited. When shorted, the minimum
NSD of 1.12 nV/√Hz is achieved. When open, the resistance of
100 Ω at the first tap generates 1.29 nV/√Hz, so the noise
increases to 1.71 nV/√Hz. This last calculation would be important
if the AD600 were preceded, for example, by a 900 Ω resistor to
allow operation from inputs up to ±10 V rms. However, in most
cases, the low impedance of the source limits the maximum
noise resistance.
Rev. E | Page 10 of 28
AD600/AD602
It is apparent from the foregoing that it is essential to use a low
resistance in the design of the ladder network to achieve low
noise. In some applications, this can be inconvenient, requiring
the use of an external buffer or preamplifier. However, very few
amplifiers combine the needed low noise with low distortion at
maximum input levels, and the power consumption required to
achieve this performance is quite high (due to the need to
maintain very low resistance values while also coping with large
inputs). On the other hand, there is little value in providing a
buffer with high input impedance, because the usual reason for
this—the minimization of loading of a high resistance source—
is not compatible with low noise.
Apart from the small variations just mentioned, the SNR at the
output is essentially independent of the attenuator setting,
because the maximum undistorted output is 1 V rms and the
NSD at the output of the AD600 is fixed at 113 × 114 nV/√Hz,
or 158 nV/√Hz. Therefore, in a 1 MHz bandwidth, the output
SNR is 76 dB. The input NSD of the AD600/AD602 are the
same, but because of the 10 dB lower gain in the AD602’s fixed
amplifier, its output SNR is 10 dB better, or 86 dB in a 1 MHz
bandwidth.
THE GAIN-CONTROL INTERFACE
The attenuation is controlled through a differential, high
impedance (15 MΩ) input, with a scaling factor that is laser
trimmed to 32 dB per volt, that is, 31.25 mV/dB. Each of the
two amplifiers has its own control interface. An internal band
gap reference ensures stability of the scaling with respect to
supply and temperature variations and is the only circuitry
common to both channels.
When the differential input voltage VG = 0 V, the attenuator
slider is centered, providing an attenuation of +21.07 dB,
resulting in an overall gain of +20 dB (= –21.07 dB + +41.07 dB).
When the control input is −625 mV, the gain is lowered by
+20 dB (= +0.625 × +32) to 0 dB; when set to +625 mV, the
gain is increased by +20 dB to +40 dB. When this interface is
overdriven in either direction, the gain approaches either
−1.07 dB (= −42.14 dB + +41.07 dB) or +41.07 dB (= 0 +
+41.07 dB), respectively.
The gain of the AD600 can be calculated by
Gain (dB) = 32 VG + 20
(1)
where VG is in volts.
For the AD602, the expression is
Gain (dB) = 32 VG + 10
For example, the gain-control input can be fed differentially to
the inputs or single-ended by simply grounding the unused
input. In another example, if the gain is controlled by a DAC
providing a positive-only, ground-referenced output, the gain
control LO pin (either C1LO or C2LO) should be biased to a
fixed offset of 625 mV to set the gain to 0 dB when gain control
HI (C1HI or C2HI) is at zero and to set the gain to 40 dB when
at 1.25 V.
It is a simple matter to include a voltage divider to achieve other
scaling factors. When using an 8-bit DAC with an FS output of
2.55 V (10 mV/bit), a 1.6 divider ratio (generating 6.25 mV/bit)
results in a gain setting resolution of 0.2 dB/bit. Later in this
data sheet, cascading the two sections of an AD600 or AD602
when various options exist for gain control is explained (see the
Achieving 80 DB Gain Range section.)
SIGNAL-GATING INPUTS
Each amplifier section of the AD600/AD602 is equipped with a
signal gating function, controlled by a TTL or CMOS logic
input (GAT1 or GAT2). The ground references for these inputs
are the signal input grounds A1LO and A2LO, respectively.
Operation of the channel is unaffected when this input is LO or
left open-circuited. Signal transmission is blocked when this
input is HI. The dc output level of the channel is set to within a
few millivolts of the output ground (A1CM or A2CM) and
simultaneously the noise level drops significantly. The reduction
in noise and spurious signal feedthrough is useful in ultrasound
beam-forming applications, where many amplifier outputs are
summed.
COMMON-MODE REJECTION
A special circuit technique provides rejection of voltages
appearing between input grounds (A1LO and A2LO) and
output grounds (A1CM and A2CM). This is necessary because
of the op amp form of the amplifier, as shown in Figure 21.
The feedback voltage is developed across the RF1 resistor
(which, to achieve low noise, has a value of only 20 Ω). The
voltage developed across this resistor is referenced to the input
common, so the output voltage is also referred to that node.
For zero differential signal input between A1HI and A1LO, the
output A1OP simply follows the voltage at A1CM. Note that the
range of voltage differences that can exist between A1LO and
A1CM (or A2LO and A2CM) is limited to about ±100 mV.
Figure 18 shows the typical common-mode rejection ratio vs.
frequency.
(2)
Operation is specified for VG in the range from −625 mV dc to
+625 mV dc. The high impedance gain-control input ensures
minimal loading when driving many amplifiers in multiplechannel applications. The differential input configuration
provides flexibility in choosing the appropriate signal levels
and polarities for various control schemes.
Rev. E | Page 11 of 28
AD600/AD602
ACHIEVING 80 dB GAIN RANGE
The two amplifier sections of the X-AMP can be connected in
series to achieve higher gain. In this mode, the output of A1
(A1OP and A1CM) drives the input of A2 via a high-pass
network (usually just a capacitor) that rejects the dc offset.
The nominal gain range is now –2 dB to +82 dB for the AD600
or −22 dB to +62 dB for the AD602.
85
80
75
70
SNR (dB)
65
There are several options in connecting the gain-control inputs.
The choice depends on the desired SNR and gain error (output
ripple). The following examples feature the AD600; the
arguments generally apply to the AD602, with appropriate
changes to the gain values.
60
55
50
45
00538-022
40
35
SEQUENTIAL MODE (MAXIMUM SNR)
30
–0.5
0
–41.07dB
41.07dB
–42.14dB
1.07dB
C1LO
C2HI
41.07dB
C2LO
VG2
VO1 = 0.592V
OUTPUT
0dB
00538-023
VG1
2.0
A2
–40.00dB
VC = 0V
1.5
VO2 = 1.908V
Figure 23. AD600 Gain Control Input Calculations for Sequential Control Operation (A)
A2
A1
–0.51dB
–0.51dB
C1HI
–1.07dB
41.07dB
C1LO
VG1
VC = 1.25V
–41.63dB
40.56dB
C2HI
41.07dB
C2LO
VG2
VO1 = 0.592V
OUTPUT
40dB
00538-055
INPUT
0dB
VO2 = 1.908V
Figure 24. AD600 Gain Control Input Calculations for Sequential Control Operation (B)
A2
A1
0dB
0dB
C1HI
VC = 2.5V
41.07dB
C1LO
VG1
38.93dB
41.07dB
–2.14dB
C2HI
41.07dB
C2LO
VG2
VO1 = 0.592V
VO2 = 1.908V
Figure 25. AD600 Gain Control Input Calculations for Sequential Control Operation (C)
Rev. E | Page 12 of 28
OUTPUT
80dB
00538-056
INPUT
0dB
2.5
3.0
An auxiliary amplifier that senses the voltage difference
between input and output commons is provided to reject this
common voltage.
A1
C1HI
1.0
Figure 22. SNR vs. Control Voltage Sequential Control (1 MHz Bandwidth)
–40.00dB
INPUT
0dB
0.5
VG
In the sequential mode of operation, the SNR is maintained at
its highest level for as much of the gain control range as
possible, as shown in Figure 22. Note here that the gain range is
0 dB to 80 dB. Figure 23, Figure 24, and Figure 25 show the
general connections to accomplish this. Both gain-control
inputs, C1HI and C2HI, are driven in parallel by a positive-only,
ground-referenced source with a range of 0 V to 2.5 V.
AD600/AD602
The gains are offset such that A2’s gain is increased only after
A1’s gain has reached its maximum value (see Figure 26). Note
that for a differential input of −700 mV or less, the gain of a
single amplifier (A1 or A2) is at its minimum value of −1.07 dB;
for a differential input of +700 mV or more, the gain is at its
maximum value of +41.07 dB. Control inputs beyond these
limits do not affect the gain and can be tolerated without damage or
foldover in the response. See the Specifications section for more
details on the allowable voltage range. The gain is now
(3)
Gain (dB) = 32 VC
As can be seen in Figure 28 and Figure 29, the output ripple is
periodic. By offsetting the gains of A1 and A2 by half the
period of the ripple, or 3 dB, the residual gain errors of the two
amplifiers can be made to cancel. Figure 31 shows the much
lower gain ripple when configured in this manner. Figure 32
plots the SNR as a function of gain; it is very similar to that in
the parallel mode.
+41.07dB
+40.56dB
+38.93dB
A2
+20dB
*
+1.07dB
–0.56dB
*
–1.07dB
0
0
0.625
20
1.908
1.25
40
1.875
60
2.5
80
VC (V)
82.14
*GAIN OFFSET OF 1.07dB, OR 33.44mV
00538-024
0.592
GAIN
–2.14
(dB)
In this mode, the gain-control voltage is applied to both inputs
in parallel—C1HI and C2HI are connected to the control
voltage, and C1LO and C2LO are optionally connected to an
offset voltage of 0.625 V. The gain scaling is then doubled to
64dB/V, requiring only 1.25 V for an 80 dB change of gain. In
this case, the amplitude of the gain ripple is also doubled, as is
shown in Figure 29, and the instantaneous SNR at the output of
A2 decreases linearly as the gain is increased (see Figure 30).
LOW RIPPLE MODE (MINIMUM GAIN ERROR)
where VC is the applied control voltage.
A1
PARALLEL MODE (SIMPLEST GAIN-CONTROL
INTERFACE)
Figure 26. Explanation of Offset Calibration for Sequential Control
When VC is set to zero, VG1 = −0.592 V and the gain of A1 is
1.07 dB (recall that the gain of each amplifier section is 0 dB for
VG = 625 mV); meanwhile, VG2 = −1.908 V, so the gain of A2 is
−1.07 dB. The overall gain is thus 0 dB (see Figure 23). When
VC = 1.25 V, VG1 = 1.25 V – 0.592 V = 0.658 V, which sets the
gain of A1 to 40.56 dB, while VG2 = 1.25 V – 1.908 V = −0.658 V,
which sets A2’s gain at −0.56 dB. The overall gain is now 40 dB
(see Figure 24). When VC = 2.5 V, the gain of A1 is 41.07 dB and
the gain of A2 is 38.93 dB, resulting in an overall gain of 80 dB
(see Figure 25). This mode of operation is further clarified by
Figure 27, which is a plot of the separate gains of A1 and A2 and
the overall gain vs. the control voltage. Figure 28 is a plot of the
gain error of the cascaded amplifiers vs. the control voltage.
Rev. E | Page 13 of 28
90
75
80
70
70
65
60
60
50
SNR (dB)
40
A1
30
COMBINED
40
10
35
00538-025
0
–10
–0.5
50
45
A2
20
55
0
0.5
1.0
1.5
2.0
2.5
30
3.0
00538-028
OVERALL GAIN (dB)
AD600/AD602
0
0.2
0.4
0.6
Figure 27. Plot of Separate and Overall Gains in Sequential Control
1.2
4
1.0
3
0.8
2
1.2
1.4
0.6
0
–1
–2
–3
–4
0.4
0.2
0.0
–0.2
–0.4
–5
–0.6
–6
–0.8
–7
0
0.5
1.0
1.5
2.0
2.5
00538-029
GAIN ERROR (dB)
1
00538-026
GAIN ERROR (dB)
1.0
Figure 30. SNR for Cascaded Stages—Parallel Control
5
–8
–0.5
0.8
VC
VC
–1.0
–1.2
3.0
0
VC
Figure 28. Gain Error for Cascaded Stages—Sequential Control
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2
VC
1.3
Figure 31. Gain Error for Cascaded Stages—Low Ripple Mode
5
80
4
75
3
70
65
SNR (dB)
1
0
–1
–2
60
55
50
–3
45
–4
00538-027
–5
–6
–0.1
0
0.2
0.4
0.6
0.8
1.0
40
35
1.2
00538-030
GAIN ERROR (dB)
2
0
0.2
0.4
0.6
0.8
1.0
1.2
VC
VC
Figure 29. Gain Error for Cascaded Stages—Parallel Control
Figure 32. SNR vs. Control Voltage—Low Ripple Mode
Rev. E | Page 14 of 28
1.4
AD600/AD602
APPLICATIONS
TIME-GAIN CONTROL (TGC) AND TIME-VARIABLE
GAIN (TVG)
Ultrasound and sonar systems share a similar requirement: both
need to provide an exponential increase in gain in response to a
linear control voltage, that is, a gain control that is linear in dB.
Figure 33 shows the AD600/AD602 configured for a control
voltage ramp starting at −625 mV and ending at +625 mV for a
gain-control range of 40 dB. The polarity of the gain-control
voltage can be reversed, and the control voltage inputs C1HI
and C1LO can be reversed to achieve the same effect. The gaincontrol voltage can be supplied by a voltage-output DAC, such
as the AD7244, which contains two complete DACs, operates
from ±5 V supplies, has an internal reference of +3 V, and
provides ±3 V of output swing. As such, it is well suited for use
with the AD600/AD602, needing only a few resistors to scale
the output voltage of the DACs to the levels needed by the
AD600/AD602.
+625mV
0dB
40dB
VOLTAGE-OUTPUT
DAC
VG
A1
GAIN
C1LO
–625mV
A1HI
A1LO
GAT1
1
16
2
+
3
–
15
A1
14
4
REF
GAT2
5
A2LO
6
–
A2HI
7
+
C2LO
8
13
12
C1HI
A1CM
A1OP
VPOS
VNEG
+5V
–5V
11 A2OP
A2
10 A2CM
AD600 OR
AD602
9
C2HI
00538-031
The following circuits show examples of time-gain control for
ultrasound and sonar, methods for increasing the output drive,
and AGC amplifiers for audio and RF/IF signal processing
using both peak and rms detectors. These circuits also illustrate
methods of cascading X-AMPs for either maintaining the
optimal SNR or maximizing the accuracy of the gain-control
voltage for use in signal measurement. These AGC circuits can
be modified for use as voltage-controlled amplifiers in sonar
and ultrasound applications by removing the detector and
substituting a DAC or other voltage source for supplying the
control voltage.
CONTROL VOLTAGE,
VG
Figure 33. The Simplest Application of the X-AMP Is as a TGC or TVG Amplifier
in Ultrasound or Sonar. Only the A1 connections are shown for simplicity.
INCREASING OUTPUT DRIVE
The AD600/AD602’s output stage has limited capability for
negative-load driving capability. For driving loads less than
500 Ω, the load drive can be increased by approximately 5 mA
by connecting a 1 kΩ pull-down resistor from the output to the
negative supply (see Figure 34).
DRIVING CAPACITIVE LOADS
For driving capacitive loads of greater than 5 pF, insert a 10 Ω
resistor between the output and the load. This lowers the
possibility of oscillation.
GAIN-CONTROL
VOLTAGE
C1LO
VIN
A1HI
A1LO
GAT1
1
16
2
+
3
–
A1
14
4
REF
GAT2
5
A2LO
6
–
A2HI
7
+
C2LO
8
15
13
12
C1HI
A1CM
A1OP
VPOS
+5V
1kΩ
VNEG
–5V
11 A2OP
ADDED
PULL-DOWN
RESISTOR
A2
10 A2CM
AD600/
AD602
9
C2HI
Figure 34. Adding a 1 kΩ Pull-Down Resistor Increases the X-AMP’s Output
Drive by About 5 mA. Only the A1 connections are shown for simplicity.
Rev. E | Page 15 of 28
00538-032
The full potential of any high performance amplifier can only
be realized by careful attention to details in its applications. The
following pages describe fully tested circuits in which many
such details have already been considered. However, as is always
true of high accuracy, high speed analog circuits, the schematic
is only part of the story; this is no less true for the AD600/
AD602. Appropriate choices in the overall board layout and the
type and placement of power supply decoupling components
are very important. As explained previously, the input grounds
A1LO and A2LO must use the shortest possible connections.
AD600/AD602
REALIZING OTHER GAIN RANGES
A LOW NOISE, 6 dB PREAMPLIFIER
Larger gain ranges can be accommodated by cascading
amplifiers. Combinations built by cascading two amplifiers
include −20 dB to +60 dB (using one AD602), −10 dB to +70 dB
(using ½ of an AD602 followed by ½ of an AD600), and 0 dB to
80 dB (using one AD600). In multiple-channel applications,
extra protection against oscillation can be provided by using
amplifier sections from different packages.
In some ultrasound applications, a high input impedance
preamplifier is needed to avoid the signal attenuation that
results from loading the transducer by the 100 Ω input resistance
of the X-AMP. High gain cannot be tolerated because the
peak transducer signal is typically ±0.5 V, while the peak input
capability of the AD600 or AD602 is only slightly more than
±1 V. A gain of 2 is a suitable choice. It can be shown that if the
preamplifier’s overall referred-to-input (RTI) noise is the same
as that due to the X-AMP alone (1.4 nV/√Hz), the input noise
of nX2 preamplifier must be √(3/4) times as large, that is,
1.2 nV/√Hz.
The two channels of the AD600 or AD602 can operate in
parallel to achieve a 3 dB improvement in noise level, providing
1 nV/√Hz without any loss of gain accuracy or bandwidth.
+5V
In the simplest case, as shown in Figure 35, the signal inputs
A1HI and A2HI are tied directly together. The outputs A1OP
and A2OP are summed via R1 and R2 (100 Ω each), and the
control inputs C1HI/C2HI and C1LO/C2LO operate in parallel.
Using these connections, both the input and output resistances
are 50 Ω. Thus, when driven from a 50 Ω source and terminated
in a 50 Ω load, the gain is reduced by 12 dB, so the gain range
becomes –12 dB to +28 dB for the AD600 and −22 dB to
+18 dB for the AD602. The peak input capability remains
unaffected (1 V rms at the IC pins, or 2 V rms from an
unloaded 50 Ω source). The loading on each output, with a
50 Ω load, is effectively 200 Ω, because the load current is
shared between the two channels, so the overall amplifier still
meets its specified maximum output and distortion levels for a
200 Ω load. This amplifier can deliver a maximum sine wave
power of 10 dBm to the load.
R1
49.9Ω
R2
174Ω
Q1
MRF904
R4
42.2Ω
VIN
R5
42.2Ω
A1LO
GAT1
VIN
GAT2
A2LO
A2HI
C2LO
1
2
3
A1
15
14
–
4
REF
5
13
12
6
–
7
+
11
A2
8
10
AD600 OR
AD602
INPUT
GROUND
+5V
0.1µF
R6
562Ω
1µF
Q2
MM4049
R7
174Ω
100Ω
RIN OF X-AMP
OUTPUT
GROUND
1µF
–5V
16
+
0.1µF
–5V
R8
49.9Ω
9
Figure 36. A Low Noise Preamplifier for the AD600/AD602
C1HI
A1CM
100Ω
A1OP
VPOS
VNEG
A2OP
VOUT
+5V
–5V
50Ω
100Ω
A2CM
C2HI
Figure 35. An Ultralow Noise VCA Using the AD600 or AD602
00538-033
A1HI
R3
562Ω
1µF
GAIN-CONTROL
VOLTAGE
VG
– +
C1LO
1µF
00538-034
AN ULTRALOW NOISE VCA
An inexpensive circuit using complementary transistor types
chosen for their low rbb is shown in Figure 36. The gain is
determined by the ratio of the net collector load resistance to
the net emitter resistance. It is an open-loop amplifier. The gain
is ×2 (6 dB) only into a 100 Ω load, assumed to be provided by
the input resistance of the X-AMP; R2 and R7 are in shunt
with this load, and their value is important in defining the gain.
For small-signal inputs, both transistors contribute an equal
transconductance that is rendered less sensitive to signal level
by the emitter resistors, R4 and R5. They also play a dominant
role in setting the gain.
Rev. E | Page 16 of 28
AD600/AD602
A LOW NOISE AGC AMPLIFIER WITH 80 dB GAIN
RANGE
This is a Class AB amplifier. As VIN increases in a positive
direction, Q1 conducts more heavily and its re becomes lower
while Q2 increases. Conversely, increasingly negative values of
VIN result in the re of Q2 decreasing, while the re of Q1 increases.
The design is chosen such that the net emitter resistance is
essentially independent of the instantaneous value of VIN,
resulting in moderately low distortion. Low values of resistance
and moderately high bias currents are important in achieving
the low noise, wide bandwidth, and low distortion of this
preamplifier. Heavy decoupling prevents noise on the power
supply lines from being conveyed to the input of the X-AMP.
Figure 37 provides an example of the ease with which the
AD600 can be connected as an AGC amplifier. A1 and A2 are
cascaded, with 6 dB of attenuation introduced by the 100 Ω
resistor R1, while a time constant of 5 ns is formed by C1 and
the 50 Ω of net resistance at the input of A2. This has the dual
effect of lowering the overall gain range from 0 dB to 80 dB to
−6 dB to 74 dB and introducing a single-pole, low-pass filter
with a −3 dB frequency of about 32 MHz. This ensures stability
at the maximum gain for a slight reduction in the overall
bandwidth. The Capacitor C4 blocks the small dc offset voltage
at the output of A1 (which may otherwise saturate A2 at its
maximum gain) and introduces a high-pass corner at about
8 kHz, useful in eliminating low frequency noise and spurious
signals that can be present at the input.
Table 4. Measured Preamplifier Performance
Measurement
Gain (f = 30 MHz)
Bandwidth (−3 dB)
Input Signal for 1 dB Compression
Distortion
VIN = 200 mV p-p
HD2
HD3
HD2
HD3
VIN = 500 mV p-p
System Input Noise
Spectral Density (NSD)
(Preamp plus X-AMP)
Input Resistance
Input Capacitance
Input Bias Current
Power Supply Voltage
Quiescent Current
Value
6
250
1
Unit
dB
MHz
V p-p
0.27
0.14
0.44
0.58
1.03
%
%
%
%
nV/√Hz
1.4
15
±150
±5
15
kΩ
pF
μA
V
mA
+5V
R3
46.4kΩ
+5V
R4
3.74kΩ
RF
INPUT
A1HI
A1LO
GAT1
GAT2
A2LO
A2HI
C2LO
1
16
2
+
3
–
15
A1
14
4
13
REF
5
6
12
–
11
A2
7
8
+
10
AD600
9
AD590
C1HI
A1CM
A1OP
VPOS
VNEG
C4
0.1µF
+5V DEC
R1
100Ω
C2
1µF
300µA
(AT 300K)
FB
0.1µF
–5V DEC
C3
15pF
A2OP
0.1µF
+5V DEC
Q1
2N3904
C1
100pF
–5V DEC
+5V
R2 +
806Ω VPTAT
1% –
A2CM
C2HI
FB
RF
OUTPUT
–5V
POWER SUPPLY
DECOUPLING NETWORK
00538-035
C1LO
V G´
Figure 37. This Accurate HF AGC Amplifier Uses Three Active Components
Rev. E | Page 17 of 28
AD600/AD602
To understand this, first note that the current in the AD590 is
closely proportional to absolute temperature (PTAT). In fact,
this IC is intended for use as a thermometer. For the moment,
assume that the signal is a square wave. When Q1 is conducting,
VOUT is the sum of VBE. VOUT is also a voltage that is PTAT and
that can be chosen to have a TC equal but opposite to the TC of
the base-to-emitter voltage. This is actually nothing more than
the band gap voltage reference principle in thinly veiled
disguise. When R2 is chosen so that the sum of the voltage
across it and the VBE of Q1 is close to the band gap voltage of
about 1.2 V, VOUT is stable over a wide range of temperatures,
provided Q1 and the AD590 share the same thermal environment.
In fact, the 6 dB interstage attenuator means that the overall
gain of this AGC system actually runs from –6 dB to +74 dB.
Thus, an input of 2 V rms would be required to produce a
1 V rms output at the minimum gain, which exceeds the 1 V rms
maximum input specification of the AD600. The available gain
range is therefore 0 dB to 74 dB (or X1 to X5000). Since the gain
scaling is 15.625 mV/dB (because of the cascaded stages), the
minimum value of VG´ is actually increased by 6 × +15.625 mV,
or about 94 mV, to −156 mV, so the risk of saturation in Q1 is
reduced.
The emitter circuit of Q1 is somewhat inductive (due its finite ft
and base resistance). Consequently, the effective value of R2
increases with frequency. This results in an increase in the
stabilized output amplitude at high frequencies, but for the
addition of C3, determined experimentally to be 15 pF for the
2N3904 for maximum response flatness. Alternatively, a faster
transistor can be used here to reduce HF peaking. Figure 38
shows the ac response at the stabilized output level of about
1.3 rms. Figure 39 demonstrates the output stabilization for the
sine wave inputs of 1 mV to 1 V rms at frequencies of 100 kHz,
1 MHz, and 10 MHz.
Since the average emitter current is 600 μA during each halfcycle of the square wave, a resistor of 833 Ω would add a PTAT
voltage of 500 mV at 300 K, increasing by 1.66 mV/°C. In
practice, the optimum value of R2 depends on the transistor
used and, to a lesser extent, on the waveform for which the
temperature stability is to be optimized; for the devices shown
and sine wave signals, the recommended value is 806 Ω. This
resistor also serves to lower the peak current in Q1 and the
200 Hz LP filter it forms with C2 helps to minimize distortion
due to ripple in VG. Note that the output amplitude under sine
wave conditions is higher than for a square wave because the
average value of the current for an ideal rectifier would be
0.637 times as large, causing the output amplitude to be 1.88 V
(= 1.2/0.637), or 1.33 V rms. In practice, the somewhat nonideal
rectifier results in the sine wave output being regulated to about
1.275 V rms.
Rev. E | Page 18 of 28
3dB
00538-036
First, consider the particular case where R2 is zero and the
output voltage VOUT is a square wave at, for example, 100 kHz,
well above the corner frequency of the control loop. During the
time VOUT is negative, Q1 conducts. When VOUT is positive, it is
cut off. Since the average collector current is forced to be
300 μA and the square wave has a 50% duty-cycle, the current
when conducting must be 600 μA. With R2 omitted, the peak
value of VOUT would be just the VBE of Q1 at 600 μA (typically
about 700 mV) or 2 VBE p-p. This voltage, thus the amplitude at
which the output stabilizes, has a strong negative temperature
coefficient (TC), typically –1.7 mV/°C. While this may not be
troublesome in some applications, the correct value of R2
renders the output stable with temperature.
An offset of 375 mV is applied to the inverting gain-control
inputs C1LO and C2LO. Therefore, the nominal –625 mV to
+625 mV range for VG is translated upwards (at VG´) to –0.25 V
for minimum gain to +1 V for maximum gain. This prevents
Q1 from going into heavy saturation at low gains and leaves
sufficient headroom of 4 V for the AD590 to operate correctly
at high gains when using a 5 V supply.
AGC OUTPUT CHANGE (dB)
A simple half-wave detector is used based on Q1 and R2. The
average current into Capacitor C2 is the difference between the
current provided by the AD590 (300 μA at 300 K, 27°C) and the
collector current of Q1. In turn, the control voltage VG is the
time integral of this error current. When VG (thus the gain) is
stable, the rectified current in Q1 must, on average, balance
exactly the current in the AD590. If the output of A2 is too
small to do this, VG ramps up, causing the gain to increase until
Q1 conducts sufficiently. The operation of this control system
follows.
0.1
1
10
FREQUENCY (MHz)
100
Figure 38. AC Response at the Stabilized Output Level of 1.3 V rms
These problems can be eliminated using an AD636 as the
detector element in an AGC loop, in which the difference
between the rms output of the amplifier and a fixed dc reference
are nulled in a loop integrator. The dynamic range and the
accuracy with which the signal can be determined are now
entirely dependent on the amplifier used in the AGC system.
Since the input to the rms-dc converter is forced to a constant
amplitude, close to its maximum input capability, the bandwidth is
no longer signal dependent. If the amplifier has an exactly
exponential (linear-dB) gain-control law, its control voltage VG
is forced by the AGC loop to have the general form
+0.2
100kHz
0
1MHz
–0.2
10MHz
–0.4
00538-037
RELATIVE OUTPUT (dB)
AD600/AD602
0.001
0.01
0.1
INPUT AMPLITUDE (V rms)
VOUT = VSCALE log 10
1
Figure 39. Output Stabilization vs. rms Input for
Sine Wave Inputs at 100 kHz, 1 MHz, and 10 MHz
While the band gap principle used here sets the output
amplitude to 1.2 V (for the square wave case), the stabilization
point can be set to any higher amplitude, up to the maximum
output of ±(VS − 2) V that the AD600 can support. It is only
necessary to split R2 into two components of appropriate ratio
whose parallel sum remains close to the zero-TC value of
806 Ω. Figure 40 shows this and how the output can be raised
without altering the temperature stability.
5V
AD590
300µA
(AT 300K)
VIN ( RMS )
VREF
(4)
Figure 41 shows a practical wide dynamic range rmsresponding measurement system using the AD600. Note that
the signal output of this system is available at A2OP, and the
circuit can be used as a wideband AGC amplifier with an rmsresponding detector. This circuit can handle inputs from
100 μV to 1 V rms with a constant measurement bandwidth of
20 Hz to 2 MHz, limited primarily by the AD636 rms converter.
Its logarithmic output is a loadable voltage accurately calibrated
to 100 mV/dB or 2 V per decade, which simplifies the
interpretation of the reading when using a DVM and is
arranged to be −4 V for an input of 100 μV rms input, zero for
10 mV, and +4 V for a 1 V rms input. In terms of Equation 4,
VREF is 10 mV and VSCALE is 2 V.
TO AD600 PIN 16
C2
1µF
Note that the peak log output of ±4 V requires the use of ±6 V
supplies for the dual op amp U3 (AD712) although lower
supplies would suffice for the AD600 and AD636. If only ±5 V
supplies are available, it is necessary to either use a reduced
value for VSCALE (say 1 V, in which case the peak output would
be only ±2 V) or restrict the dynamic range of the signal to
about 60 dB.
Q1
2N3904
R2B
TO AD600 PIN 11
R2A
R2 = R2A || R2B ≈ 806Ω
RF
OUTPUT
00538-038
C3
15pF
+
VPTAT
–
Figure 40. Modification in Detector to Raise Output to 2 V rms
A WIDE RANGE, RMS-LINEAR dB MEASUREMENT
SYSTEM (2 MHz AGC AMPLIFIER WITH RMS
DETECTOR)
Monolithic rms-dc converters provide an inexpensive means to
measure the rms value of a signal of arbitrary waveform; they
can also provide a low accuracy logarithmic (decibel-scaled)
output. However, they have certain shortcomings. The first of
these is their restricted dynamic range, typically only 50 dB.
More troublesome is that the bandwidth is roughly proportional
to the signal level; for example, the AD636 provides a 3 dB
bandwidth of 900 kHz for an input of 100 mV rms but has a
bandwidth of only 100 kHz for a 10 mV rms input. Its
logarithmic output is unbuffered, uncalibrated, and not stable
over temperature. Considerable support circuitry, including at
least two adjustments and a special high TC resistor, is required
to provide a useful output.
As in the previous case, the two amplifiers of the AD600 are
used in cascade. However, the 6 dB attenuator and low-pass
filter found in Figure 21 are replaced by a unity gain buffer
amplifier U3A, whose 4 MHz bandwidth eliminates the risk of
instability at the highest gains. The buffer also allows the use of
a high impedance coupling network (C1/R3) that introduces a
high-pass corner at about 12 Hz. An input attenuator of 10 dB
(X0.316) is now provided by R1 + R2 operating in conjunction
with the AD600’s input resistance of 100 Ω. The adjustment
provides exact calibration of the logarithmic intercept VREF in
critical applications, but R1 and R2 can be replaced by a fixed
resistor of 215 Ω if very close calibration is not needed, because
the input resistance of the AD600 (and all other key parameters
of it and the AD636) is already laser trimmed for accurate
operation. This attenuator allows inputs as large as ±4 V to be
accepted, that is, signals with an rms value of 1 V combined
with a crest factor of up to 4.
Rev. E | Page 19 of 28
AD600/AD602
Vrms
AF/RF
OUTPUT
C1
0.1µF
C4
4.7µF
+6V DEC
INPUT
1V rms
MAX
(SINE WAVE)
R1
115Ω
C1LO
A1HI
R2 200Ω
A1LO
GAT1
R3
133kΩ
U3A
1/2
AD712
VG
15.625mV/dB
GAT2
A2LO
A2HI
C2LO
R4
3.01kΩ
1
16
2
+
3
–
A1
14
4
REF
5
6
15
13
12
–
11
A2
7
8
10
+
U1
AD600
9
+6V
C1HI
1 VINP
A1CM
NC
A1OP
VPOS
VNEG
A2OP
–6V
DEC
+6V
DEC
–6V
DEC
C2
2µF
2
VPOS 14
U2
AD636
3 VNEG
12 NC
4 CAVG
11 NC
NC
5 VLOG
COMM 10
NC
6 BFOP
LDLO 9
7 BFIN
VRMS 8
A2CM
FB
13 NC
R7
56.2kΩ
R6
3.16kΩ
0.1µF
+6V
DEC
0.1µF
–6V
DEC
FB
+316.2mV
C2HI
1/2
AD712
U3B
C3
1µF
–6V
POWER SUPPLY
DECOUPLING
NETWORK
VOUT
+100mV/dB
0V = 0dB (AT 10mV rms)
R5
16.2kΩ
00538-039
CAL
0dB
NC = NO CONNECT
Figure 41. The Output of This Three-IC Circuit Is Proportional to the Decibel Value of the rms Input
Any difference in these voltages is integrated by the op amp
U3B, with a time constant of 3 ms formed by the parallel sum
of R6/R7 and C3. Now, if the output of the AD600 is too high,
V rms is greater than the setpoint of 316 mV, causing the output
of U3B—that is, VOUT—to ramp up (note that the integrator is
noninverting). A fraction of VOUT is connected to the inverting
gain-control inputs of the AD600, so causing the gain to be
reduced, as required, until V rms is exactly equal to 316 mV, at
which time the ac voltage at the output of A2 is forced to be
exactly 316 mV rms. This fraction is set by R4 and R5 such that
a 15.625 mV change in the control voltages of A1 and A2—
which would change the gain of the cascaded amplifiers by
1 dB—requires a change of 100 mV at VOUT. Notice here that
since A2 is forced to operate at an output level well below its
capacity, waveforms of high crest factor can be tolerated
throughout the amplifier.
To check the operation, assume an input of 10 mV rms is
applied to the input, which results in a voltage of 3.16 mV rms
at the input to A1, due to the 10 dB loss in the attenuator. If the
system operates as claimed, VOUT (and hence VG) should be 0.
This being the case, the gain of both A1 and A2 is 20 dB and the
output of the AD600 is therefore 100 times (40 dB) greater than
its input, which evaluates to 316 mV rms, the input required at
the AD636 to balance the loop. Finally, note that unlike most
AGC circuits that need strong temperature compensation for
the internal kT/q scaling, these voltages, and thus the output of
this measurement system, are temperature stable, arising
directly from the fundamental and exact exponential
attenuation of the ladder networks in the AD600.
Typical results are presented for a sine wave input at 100 kHz.
Figure 42 shows that the output is held close to the setpoint of
316 mV rms over an input range in excess of 80 dB.
450
425
400
375
350
VOUT (mV)
The output of A2 is ac-coupled via another 12 Hz high-pass
filter formed by C2 and the 6.7 kΩ input resistance of the
AD636. The averaging time constant for the rms-dc converter
is determined by C4. The unbuffered output of the AD636 (at
Pin 8) is compared with a fixed voltage of 316 mV set by the
positive supply voltage of 6 V and Resistors R6 and R7. VREF is
proportional to this voltage, and systems requiring greater
calibration accuracy should replace the supply dependent
reference with a more stable source.
325
300
275
250
225
00538-040
200
175
150
10µ
100µ
1m
10m
100m
INPUT SIGNAL (V rms)
1
10
Figure 42. RMS Output of A2 Held Close to the Setpoint 316 mV
for an Input Range of over 80 dB
Rev. E | Page 20 of 28
AD600/AD602
This ripple can be canceled whenever the X-AMP stages are
cascaded by introducing a 3 dB offset between the two pairs of
control voltages. A simple means to achieve this is shown in
Figure 45: the voltages at C1HI and C2HI are split by
±46.875 mV, or ±1.5 dB. Alternatively, either one of these pins
can be individually offset by 3 dB and a 1.5 dB gain adjustment
made at the input attenuator (R1 + R2).
C1HI
16
15
U1
AD600
5
1 VINP
A1CM
14 A1OP
VPOS
13
VNEG
12
A2OP
11
A2CM
10
C2HI
9
NC 2
+6V DEC
–6V DEC
1
–1
00538-041
–3
–4
100µ
1m
10m
100m
INPUT SIGNAL (V rms)
1
10
Figure 43. The dB Output of Figure 41’s Circuit is Linear over an 80 dB Range
3dB OFFSET
MODIFICATION
The error curve shown in Figure 46 demonstrates that over the
central portion of the range the output voltage can be maintained
close to the ideal value. The penalty for this modification is the
higher errors at the extremities of the range. The next two
applications show how three amplifier sections can be cascaded
to extend the nominal conversion range to 120 dB, with the
inclusion of simple LP filters of the type shown in Figure 37.
Very low errors can then be maintained over a 100 dB range.
2.5
2.0
2.0
1.5
1.5
1.0
1.0
OUTPUT ERROR (dB)
2.5
0.5
0
–0.5
–1.0
–1.5
0.5
0
–0.5
–1.0
00538-042
–1.5
–2.0
–2.5
10µ
+46.875mV
+6V
DEC
10kΩ
Figure 45. Reducing the Gain Error Ripple
0
–5
10µ
78.7Ω 78.7Ω
NC = NO CONNECT
–2
OUTPUT ERROR (dB)
10kΩ
U2
AD636
100µ
1m
10m
100m
INPUT SIGNAL (V rms)
1
00538-044
VOUT (V)
2
NC 5 VLOG
7 BFIN
–46.875mV
–6V
DEC
4 CAVG
C2
2µF
NC 6 BFOP
4
3
3 VNEG
–6V DEC
00538-043
This system can, of course, be used as an AGC amplifier in
which the rms value of the input is leveled. Figure 43 shows the
decibel output voltage. More revealing is Figure 44, which
shows that the deviation from the ideal output predicted by
Equation 1 over the input range 80 μV to 500 mV rms is within
±0.5 dB, and within ±1 dB for the 80 dB range from 80 μV to
800 mV. By suitable choice of the input attenuator R1 + R2, this
can be centered to cover any range from a low of 25 mV to
250 mV to a high of 1 mV to 10 V, with appropriate correction
to the value of VREF. Note that VSCALE is not affected by the
changes in the range. The gain ripple of ±0.2 dB seen in this
curve is the result of the finite interpolation error of the
X-AMP. Note that it occurs with a periodicity of 12 dB, twice
the separation between the tap points (because of the two
cascaded stages).
–2.0
–2.5
10µ
10
Figure 44. Data from Figure 42 Presented as the Deviation
from the Ideal Output Given in Equation 4
100µ
1m
10m
100m
INPUT SIGNAL (V rms)
1
10
Figure 46. Using a 3 dB Offset Network Reduces Ripple
100 dB TO 120 dB RMS RESPONDING CONSTANT
BANDWIDTH AGC SYSTEMS WITH HIGH
ACCURACY dB OUTPUTS
The next two applications double as both AGC amplifiers and
measurement systems. In both, precise gain offsets are used to
achieve either a high gain linearity of ±0.1 dB over the full
100 dB range or the optimal SNR at any gain.
Rev. E | Page 21 of 28
AD600/AD602
A1HI
A1LO
GAT1
GAT2
A2LO
A2HI
C2LO
1
2
3
16
+
A1
14
–
4
REF
5
6
15
13
12
–
11
A2
7
10
+
8
U1 AD600
9
C1LO
C1HI
A1CM
C1
0.1µF
A1OP
VPOS
VNEG
A2OP
+5V
DEC
–5V
DEC
U3A
–2dB
–62.5mV
–5V
R6
10kΩ
R7
127Ω
A1HI
A1LO
R3
200Ω
1/4
AD713
GAT1
GAT2
R5
1.58kΩ
A2LO
U3B
C3
220pF
A2HI
1/4
AD713
C2HI
0.1µF
+5V
DEC
C2
0.1µF
R4
133kΩ
A2CM
+5V
FB
R1
133kΩ
R2
487Ω
0dB
+2dB
+62.5mV
R8
127Ω
R9
10kΩ
2
3
16
+
15
A1
14
–
4
REF
5
13
12
6
–
7
+
A1CM
A1OP
VPOS
VNEG
C4
2µF
VOUT
+5V
DEC
–5V
DEC
11 A2OP
A2
8
C1HI
10
9
U2 AD600
A2CM
C2HI
+5V
C5
22µF
0.1µF
–5V
DEC
C2LO
1
+5V DEC
FB
1 VINP
NC
–5V
POWER SUPPLY
DECOUPLING
NETWORK
R14
301kΩ
Q1
2N3906
–5V
DEC
+5V DEC
R15
19.6kΩ
R16
6.65kΩ
VPOS 14
U4
AD636
2
13 NC
3 VNEG
12 NC
4 CAVG
11 NC
COMM 10
NC
5 VLOG
NC
6 BFOP
LDLO
9
7 BFIN
VRMS
8
R11
46.4kΩ
R10
3.16kΩ
C6
4.7µF
U3C
R12
11.3kΩ
+316.2mV
R13
3.01kΩ
VLOG
1/4
AD713
NC = NO CONNECT
00538-045
C1LO
INPUT
1V rms
MAX
(SINE WAVE)
Figure 47. RMS Responding AGC Circuit with 100 dB Dynamic Range
A 100 dB RMS/AGC SYSTEM WITH MINIMAL GAIN
ERROR (PARALLEL GAIN WITH OFFSET)
5
Figure 47 shows an rms-responding AGC circuit that can be
used equally well as an accurate measurement system. It accepts
inputs of 10 μV to 1 V rms (−100 dBV to 0 dBV) with generous
overrange. Figure 48 shows the logarithmic output, VLOG, which
is accurately scaled 1 V per decade, that is, 50 mV/dB, with an
intercept (VLOG = 0) at 3.16 mV rms (−50 dBV). Gain offsets of
±2 dB were introduced between the amplifiers, provided by the
±62.5 mV introduced by R6 to R9. These offsets cancel a small
gain ripple that arises in the X-AMP from its finite interpolation
error, which has a period of 18 dB in the individual VCA
sections. The gain ripple of all three amplifier sections without
this offset (in which case, the gain errors simply add) is shown
in Figure 49; it is still a remarkably low ±0.25 dB over the
108 dB range from 6 μV to 1.5 V rms. However, with the gain
offsets connected, the gain linearity remains under ±0.1 dB over
the specified 100 dB range (see Figure 50).
3
LOGARITHMIC OUTPUT (V)
4
2
1
0
–1
–2
00538-046
–3
–4
–5
1µ
10µ
100µ
1m
10m
100m
1
10
INPUT SIGNAL (V rms)
Figure 48. VLOG Plotted vs. VIN for Figure 47’s Circuit Showing 120 dB AGC Range
Rev. E | Page 22 of 28
AD600/AD602
2.0
The rms value of VLOG is generated at Pin 8 of the AD636; the
averaging time for this process is determined by C5, and the
value shown results in less than 1% rms error at 20 Hz. The
slowly varying V rms is compared with a fixed reference of
316 mV, derived from the positive supply by R10/R11. Any
difference between these two voltages is integrated in C6, in
conjunction with Op Amp U3C, the output of which is VLOG. A
fraction of this voltage, determined by R12 and R13, is returned
to the gain control inputs of all AD600 sections. An increase in
VLOG lowers the gain because this voltage is connected to the
inverting polarity control inputs.
1.5
GAIN ERROR (dB)
1.0
0.5
0.1
0
–0.1
–0.5
–1.0
–2.0
1µ
00538-047
–1.5
10µ
100µ
1m
10m
100m
INPUT SIGNAL (V rms)
1
10
Figure 49. Gain Error for Figure 41 Without the 2 dB Offset Modification
2.0
1.5
GAIN ERROR (dB)
1.0
Optionally, the scaling can be altered to 100 mV/dB, which
would be more easily interpreted when VLOG is displayed on a
DVM by increasing R12 to 25.5 kΩ. The full-scale output of
±5 V then requires the use of supply voltages of at least ±7.5 V.
0.5
0.1
0
–0.1
–0.5
A simple attenuator of 16.6 ± 1.25 dB is formed by R2/R3
and the 100 Ω input resistance of the AD600. This allows the
reference level of the decibel output to be precisely set to 0 for
an input of 3.16 mV rms, and thus center the 100 dB range
between 10 μV and 1 V. In many applications, R2/R3 can be
replaced by a fixed resistor of 590 Ω. For example, in AGC
applications, neither the slope nor the intercept of the
logarithmic output is important.
–1.0
00538-048
–1.5
–2.0
1µ
In this case, the gains of all three VCA sections are being varied
simultaneously, so the scaling is not 32 dB/V but 96 dB/V or
10.42 mV/dB. The fraction of VLOG required to set its scaling to
50 mV/dB is therefore 10.42/50 or 0.208. The resulting fullscale range of VLOG is nominally ±2.5 V. This scaling allows the
circuit to operate from ±5 V supplies.
10µ
100µ
1m
10m
100m
INPUT SIGNAL (V rms)
1
10
Figure 50. Adding the 2 dB Offsets Improves the Linearization
The maximum gain of this circuit is 120 dB. If no filtering was
used, the noise spectral density of the AD600 (1.4 nV/√Hz)
would amount to an input noise of 8.28 μV rms in the full
bandwidth (35 MHz). At a gain of one million, the output noise
would dominate. Consequently, some reduction of bandwidth is
mandatory, and in the circuit of Figure 47, it is due mostly to a
single-pole, low-pass filter R5/C3, which provides a −3 dB
frequency of 458 kHz, which reduces the worst-case output
noise (at VAGC) to about 100 mV rms at a gain of 100 dB. Of
course, the bandwidth (and therefore the output noise) could be
further reduced, for example, in audio applications, merely by
increasing C3. The value chosen for this application is optimal
in minimizing the error in the VLOG output for small input signals.
The AD600 is dc-coupled, but even miniscule offset voltages at
the input would overload the output at high gains; thus, highpass filtering is also needed. To provide operation at low
frequencies, two simple 0s at about 12 Hz are provided by
R1/C1 and R4/C2; op amp sections U3A and U3B (AD713)
are used to provide impedance buffering, because the input
resistance of the AD600 is only 100 Ω. A further 0 at 12 Hz is
provided by C4 and the 6.7 kΩ input resistance of the AD636
rms converter.
A few additional components (R14 to R16 and Q1) improve the
accuracy of VLOG at the top end of the signal range (that is, for
small gains). The gain starts rolling off when the input to the
first amplifier, U1A, reaches 0 dB. To compensate for this
nonlinearity, Q1 turns on at VLOG ~ 1.5 V and increases the
feedback to the control inputs of the AD600s, thereby needing a
smaller voltage at VLOG to maintain the input to the AD636 to
the setpoint of 316 mV rms.
A 120 dB RMS/AGC SYSTEM WITH OPTIMAL SNR
(SEQUENTIAL GAIN)
In the last case, all gains were adjusted simultaneously, resulting
in an output SNR that is always less than optimal. The use of
sequential gain control results in a major improvement in SNR,
with only a slight penalty in the accuracy of VLOG, and no
penalty in the stabilization accuracy of VAGC. The idea is to
increase the gain of the earlier stages first (as the signal level
decreases) and maintain the highest SNR throughout the
amplifier chain. This can be easily achieved with the AD600
because its gain is accurate even when the control input is
overdriven. That is, each gain control window of 1.25 V is
used fully before moving to the next amplifier to the right.
Rev. E | Page 23 of 28
AD600/AD602
Figure 51 shows the circuit for the sequential control scheme.
R6 to R9 with R16 provide offsets of 42.14 dB between the
individual amplifiers to ensure smooth transitions between the
gain of each successive X-AMP, with the sequence of gain
increase being U1A, then U1B, and then U2A. The adjustable
attenuator provided by R3 + R17 and the 100 Ω input resistance
of U1A, as well as the fixed 6 dB attenuation provided by R2
and the input resistance of U1B, are included both to set VLOG to
read 0 dB when VIN is 3.16 mV rms and to center the 100 dB
range between 10 μV rms and 1 V rms input. R5 and C3
provide a 3 dB noise bandwidth of 30 kHz. R12 to R15 change
the scaling from 625 mV/decade at the control inputs to
1 V/decade at the output. At the same time, R12 to R15 center
the dynamic range at 60 dB, which occurs if the VG of U1B is
equal to 0. These arrangements ensure that the VLOG still fits
within the ±6 V supplies.
0dB
ADJUST
C1LO
R3
R17 200Ω
115Ω
A1HI
A1LO
GAT1
GAT2
A2LO
A2HI
C2LO
1
2
3
16
+
A1
15
14
–
4
REF
5
13
12
6
–
7
+
11
A2
8
10
U1 AD600
9
C1LO
C1HI
A1CM
C1
0.1µF
A1OP
VPOS
VNEG
A2OP
R1
+6V
DEC 133kΩ
C2
–6V
R5
DEC 0.1µF 5.36kΩ
R4
133kΩ
A2CM
R2
100Ω
U3A
1/4
AD713
+6V
2
3
GAT1
15
A1
13
12
6
–
7
+
8
A1CM
A1OP
VPOS
VNEG
C4
2µF
VOUT
+5V
DEC
–5V
DEC
11 A2OP
A2
C2LO
C1HI
14
–
5
A2HI
1/4
AD713
+
REF
GAT2
U3B
16
4
A2LO
C3
0.001µF
R7
R8
1kΩ 294Ω
A1HI
A1LO
C2HI
R6
3.4kΩ
1
10
9
U2 AD600
A2CM
C2HI
R9 R16
1kΩ 287Ω
+6V
C5
22µF
FB
+6V DEC
0.1µF
+6V
DEC
1 VINP
0.1µF
–6V
DEC
NC
–6V
DEC
FB
–6V
POWER SUPPLY
DECOUPLING
NETWORK
R13
866Ω
VPOS 14
U4
AD636
13 NC
3 VNEG
12 NC
4 CAVG
11 NC
NC
5 VLOG
COMM 10
NC
6 BFOP
LDLO 9
7 BFIN
VRMS
+6V DEC
R15
5.11kΩ
2
R12
1kΩ
R11
56.2kΩ
R10
3.16kΩ
C6
4.7µF
U3C
8
+316.2mV
R14
7.32kΩ
NC = NO CONNECT
Figure 51. 120 dB Dynamic Range RMS Responding Circuit Optimized for SNR
Rev. E | Page 24 of 28
VLOG
1/4
AD713
00538-049
INPUT
AD600/AD602
400
5
4
350
2
GAIN ERROR (mV)
LOGARITHMIC OUTPUT (V)
3
1
0
–1
–2
300
250
–4
–5
1µ
10µ
100µ
1m
10m
100m
1
200
10
00538-052
00538-050
–3
1µ
10µ
100µ
1m
10m
100m
1
10
INPUT SIGNAL (V rms)
INPUT SIGNAL (V rms)
Figure 54. VAGC Remains Close to Its Setpoint of
316 mV rms over the Full 120 dB Range
Figure 52. VLOG Is Linear over the Full 120 dB Range
Figure 52 shows VLOG to be linear over a full 120 dB range.
Figure 53 shows the error ripple due to the individual gain
functions bounded by ±0.2 dB (dotted lines) from 6 μV to 2 V.
The small perturbations at about 200 μV and 20 mV, caused by
the impracticality of matching the gain functions perfectly, are
the only sign that the gains are now sequential. Figure 54 is a
plot of VAGC that remains very close to its set value of 316 mV
rms over the full 120 dB range.
90
VC SCALE = 10.417mV/dB
80
70
SNR (dB)
60
50
40
30
2.0
20
10
0
–833.2 –625.0 –416.6 –208.3
0
208.3
416.6
625.0
833.2
VC (mV)
Figure 55. SNR vs. Control Voltage for Parallel Gain Control (See Figure 47)
In contrast, the SNR for the sequential mode is shown in Figure 56.
U1A always acts as a fixed noise source; varying its gain has no
influence on the output noise. This is a feature of the X-AMP
technique. Therefore, for the first 40 dB of control range
(actually slightly more, as is explained later), when only this
VCA section has its gain varied, the SNR remains constant.
During this time, the gains of U1B and U2A are at their
minimum value of −1.07 dB.
1.5
90
VC SCALE = 31.25mV/dB
1.0
80
70
0.2
0
–0.2
60
SNR (dB)
0.5
–0.5
–1.0
50
40
30
–2.0
1µ
10µ
100µ
1m
10m
100m
1
20
10
10
00538-054
–1.5
00538-051
GAIN ERROR (dB)
00538-053
To compare the SNRs in the simultaneous and sequential
modes of operation more directly, all interstage attenuation was
eliminated (R2 and R3 in Figure 47 and R2 in Figure 51), the
input of U1A was shorted, R5 was selected to provide a 20 kHz
bandwidth (R5 = 7.87 kΩ), and only the gain control was
varied, using an external source. The rms value of the noise was
then measured at VOUT and expressed as an SNR relative to
0 dBV, which is almost the maximum output capability of the
AD600. Results for the simultaneous mode can be seen in
Figure 55. The SNR degrades uniformly as the gain is increased.
Note that since the inverting gain control was used, the gain in
this curve and in Figure 56 decreases for more positive values of
the gain-control voltage.
INPUT SIGNAL (V rms)
0
–1.183 –0.558
Figure 53. Error Ripple due to the Individual Gain Functions
0.067
0.692
1.317
1.942
2.567
3.192
3.817
VC (V)
Figure 56. SNR vs. Control Voltage for Sequential Gain Control (See Figure 51)
Rev. E | Page 25 of 28
AD600/AD602
For the next 40 dB of control range, the gain of U1A remains
fixed at its maximum value of 41.07 dB and only the gain of
U1B is varied, while that of U2A remains at its minimum value
of −1.07 dB. In this interval, the fixed output noise of U1A is
amplified by the increasing gain of U1B and the SNR
progressively decreases.
Once U1B reaches its maximum gain of 41.07 dB, its output
also becomes a gain independent noise source; this noise is
presented to U2A. As the control voltage is further increased,
the gains of both U1A and U1B remain fixed at their maximum
value of 41.07 dB, and the SNR continues to decrease. Figure 56
clearly shows this, because the maximum SNR of 90 dB is
extended for the first 40 dB of input signal before it starts to roll off.
This arrangement of staggered gains can be easily implemented
because when the control inputs of the AD600 are overdriven,
the gain limits to its maximum or minimum values without side
effects. This eliminates the need for awkward nonlinear shaping
circuits that have previously been used to break up the gain
range of multistage AGC amplifiers. The precise values of the
AD600’s maximum and minimum gain (not 0 dB and +40 dB
but −1.07 dB and +41.07 dB) explain the rather odd values of
the offset values that are used.
The optimization of the output SNR is of obvious value in AGC
systems. However, in applications where these circuits are
considered for their wide range logarithmic measurement
capabilities, the inevitable degradation of the SNR at high gains
need not seriously impair their utility. In fact, the bandwidth of
the circuit shown in Figure 47 was specifically chosen to improve
measurement accuracy by altering the shape of the log error
curve at low signal levels (see Figure 53).
Rev. E | Page 26 of 28
AD600/AD602
OUTLINE DIMENSIONS
0.800 (20.32)
0.790 (20.07)
0.780 (19.81)
16
9
1
8
0.280 (7.11)
0.250 (6.35)
0.240 (6.10)
0.325 (8.26)
0.310 (7.87)
0.300 (7.62)
PIN 1
0.100 (2.54)
BSC
0.060 (1.52)
MAX
0.210
(5.33)
MAX
0.195 (4.95)
0.130 (3.30)
0.115 (2.92)
0.015
(0.38)
MIN
0.150 (3.81)
0.130 (3.30)
0.115 (2.92)
0.015 (0.38)
GAUGE
PLANE
SEATING
PLANE
0.022 (0.56)
0.018 (0.46)
0.014 (0.36)
0.430 (10.92)
MAX
0.005 (0.13)
MIN
0.014 (0.36)
0.010 (0.25)
0.008 (0.20)
0.070 (1.78)
0.060 (1.52)
0.045 (1.14)
COMPLIANT TO JEDEC STANDARDS MS-001-AB
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
CORNER LEADS MAY BE CONFIGURED AS WHOLE OR HALF LEADS.
Figure 57. 16-Lead Plastic Dual In-Line Package [PDIP]
Narrow Body (N-16)
Dimensions shown in inches and (millimeters)
0.005 (0.13) MIN
0.098 (2.49) MAX
16
9
1
PIN 1
8
10.50 (0.4134)
10.10 (0.3976)
0.310 (7.87)
0.220 (5.59)
9
16
7.60 (0.2992)
7.40 (0.2913)
0.100 (2.54) BSC
0.840 (21.34) MAX
0.200 (5.08)
MAX
0.200 (5.08)
0.125 (3.18)
0.023 (0.58)
0.014 (0.36)
0.060 (1.52)
0.015 (0.38)
0.150
(3.81)
MIN
SEATING
0.070 (1.78) PLANE
0.030 (0.76)
8
1
0.320 (8.13)
0.290 (7.37)
1.27 (0.0500)
BSC
15°
0°
0.015 (0.38)
0.008 (0.20)
0.30 (0.0118)
0.10 (0.0039)
COPLANARITY
0.10
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
0.51 (0.0201)
0.31 (0.0122)
10.65 (0.4193)
10.00 (0.3937)
2.65 (0.1043)
2.35 (0.0925)
SEATING
PLANE
8°
0.33 (0.0130) 0°
0.20 (0.0079)
0.75 (0.0295)
× 45°
0.25 (0.0098)
1.27 (0.0500)
0.40 (0.0157)
COMPLIANT TO JEDEC STANDARDS MS-013-AA
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
Figure 59. 16-Lead Standard Small Outline Package [SOIC_W]
Wide Body (RW-16)
Dimensions shown in millimeters and (inches)
Figure 58. 16-Lead Ceramic Dual In-Line Package [CERDIP]
(Q-16)
Dimensions shown in inches and (millimeters)
Rev. E | Page 27 of 28
AD600/AD602
ORDERING GUIDE
Model
AD600AQ
AD600AR
AD600AR-REEL
AD600AR-REEL7
AD600ARZ 1
AD600ARZ-R71
AD600ARZ-RL1
AD600JN
AD600JNZ1
AD600JR
AD600JR-REEL
AD600JR-REEL7
AD600JRZ1
AD600JRZ-R71
AD600JRZ-RL1
AD600SQ/883B 2
AD602AQ
AD602AR
AD602AR-REEL
AD602AR-REEL7
AD602ARZ1
AD602ARZ-R71
AD602ARZ-RL1
AD602JCHIPS
AD602JN
AD602JNZ1
AD602JR
AD602JR-REEL
AD602JR-REEL7
AD602JRZ1
AD602JRZ-R71
AD602JRZ-RL1
AD602SQ/883B 3
1
2
3
Gain Range
0 dB to 40 dB
0 dB to 40 dB
0 dB to 40 dB
0 dB to 40 dB
0 dB to 40 dB
0 dB to 40 dB
0 dB to 40 dB
0 dB to 40 dB
0 dB to 40 dB
0 dB to 40 dB
0 dB to 40 dB
0 dB to 40 dB
0 dB to 40 dB
0 dB to 40 dB
0 dB to 40 dB
0 dB to 40 dB
−10 dB to +30 dB
−10 dB to +30 dB
−10 dB to +30 dB
−10 dB to +30 dB
−10 dB to +30 dB
−10 dB to +30 dB
−10 dB to +30 dB
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
−55°C to +125°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−10 dB to +30 dB
−10 dB to +30 dB
−10 dB to +30 dB
–10 dB to +30 dB
−10 dB to +30 dB
−10 dB to +30 dB
–10 dB to +30 dB
−10 dB to +30 dB
−10 dB to +30 dB
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
−55°C to +150°C
Z = Pb-free part.
Refer to AD600/AD602 military data sheet. Also available as 5962-9457201MEA.
Refer to AD600/AD602 military data sheet. Also available as 5962-9457202MEA.
©2006 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
C00538-0-1/06(E)
Rev. E | Page 28 of 28
Package Description
16-Lead CERDIP
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead PDIP
16-Lead PDIP
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead CERDIP
16-Lead CERDIP
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
DIE
16-Lead PDIP
16-Lead PDIP
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead SOIC_W
16-Lead CERDIP
Package Option
Q-16
RW-16
RW-16
RW-16
RW-16
RW-16
RW-16
N-16
N-16
RW-16
RW-16
RW-16
RW-16
RW-16
RW-16
Q-16
Q-16
RW-16
RW-16
RW-16
RW-16
RW-16
RW-16
N-16
N-16
RW-16
RW-16
RW-16
RW-16
RW-16
RW-16
Q-16