LINER LT1110-5 Micropower dc-dc converter adjustable and fixed 5v, 12v Datasheet

LT1110
Micropower
DC-DC Converter
Adjustable and Fixed 5V, 12V
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DESCRIPTIO
FEATURES
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Operates at Supply Voltages From 1.0V to 30V
Works in Step-Up or Step-Down Mode
Only Three External Off-the-Shelf Components
Required
Low-Battery Detector Comparator On-Chip
User-Adjustable Current Limit
Internal 1A Power Switch
Fixed or Adjustable Output Voltage Versions
Space-Saving 8-Pin MiniDIP or S8 Package
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APPLICATI
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Pagers
Cameras
Single-Cell to 5V Converters
Battery Backup Supplies
Laptop and Palmtop Computers
Cellular Telephones
Portable Instruments
Laser Diode Drivers
Hand-Held Inventory Computers
The 70kHz oscillator allows the use of surface mount
inductors and capacitors in many applications. Quiescent
current is just 300µA, making the device ideal in remote or
battery powered applications where current consumption
must be kept to a minimum.
The device can easily be configured as a step-up or
step-down converter, although for most step-down
applications or input sources greater than 3V, the LT1111
is recommended. Switch current limiting is user-adjustable
by adding a single external resistor. Unique reverse battery
protection circuitry limits reverse current to safe, nondestructive levels at reverse supply voltages up to 1.6V.
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The LT1110 is a versatile micropower DC-DC converter.
The device requires only three external components to
deliver a fixed output of 5V or 12V. The very low minimum
supply voltage of 1.0V allows the use of the LT1110 in
applications where the primary power source is a single
cell. An on-chip auxiliary gain block can function as a low
battery detector or linear post regulator.
TYPICAL APPLICATI
All Surface Mount
Single Cell to 5V Converter
SUMIDA
CD54-470K
47µH
Efficiency
90
85
MBRS120T3
5V
I LIM
1.5V
AA CELL*
2
V IN
SW1
3
LT1110-5
GND
5
SENSE
8
SW2
4
+
75
VIN = 1.25V
70
VIN = 1.00V
65
60
15µF
TANTALUM
55
50
OPERATES WITH CELL VOLTAGE ≥ 1.0V
*ADD 10µF DECOUPLING CAPACITOR IF BATTERY
IS MORE THAN 2" AWAY FROM LT1110.
VIN = 1.50V
80
EFFICIENCY (%)
1
0
LT1110 • TA01
5
10
15
20
25
30
35
40
LOAD CURRENT (mA)
LT1110 • TA02
1
LT1110
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RATI GS
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ABSOLUTE
PACKAGE/ORDER I FOR ATIO
Supply Voltage, Step-Up Mode ................................ 15V
Supply Voltage, Step-Down Mode ........................... 36V
SW1 Pin Voltage ...................................................... 50V
SW2 Pin Voltage ......................................... – 0.5V to VIN
Feedback Pin Voltage (LT1110) .............................. 5.5V
Switch Current ........................................................ 1.5A
Maximum Power Dissipation ............................. 500mW
Operating Temperature Range ..................... 0°C to 70°C
Storage Temperature Range .................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec.)................. 300°C
TOP VIEW
ILIM 1
8
FB (SENSE)*
VIN 2
7
SET
SW1 3
6
A0
SW2 4
5
GND
N8 PACKAGE
8-LEAD PLASTIC DIP
ORDER PART
NUMBER
LT1110CN8
LT1110CN8-5
LT1110CN8-12
*FIXED VERSIONS
TJMAX = 90°C, θJA = 130°C/W
TOP VIEW
ILIM 1
8
FB (SENSE)*
VIN 2
7
SET
SW1 3
6
A0
SW2 4
5
GND
LT1110CS8
LT1110CS8-5
LT1110CS8-12
S8 PART MARKING
1110
11105
11012
S8 PACKAGE
8-LEAD PLASTIC SOIC
*FIXED VERSIONS
TJMAX = 90°C, θJA = 150°C/W
Consult factory for Industrial and Military grade parts.
ELECTRICAL CHARACTERISTICS TA = 25°C, VIN = 1.5V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
IQ
Quiescent Current
Switch Off
●
VIN
Input Voltage
Step-Up Mode
●
Step-Down Mode
●
Comparator Trip Point Voltage
LT1110 (Note 1)
●
210
220
230
mV
Output Sense Voltage
LT1110-5 (Note 2)
●
4.75
5.00
5.25
V
LT1110-12 (Note 2)
●
11.4
12.00
12.6
V
Comparator Hysteresis
LT1110
●
4
8
mV
Output Hysteresis
LT1110-5
●
90
180
mV
LT1110-12
●
VOUT
MIN
1.15
1.0
UNITS
µA
12.6
12.6
V
V
30
V
200
400
mV
52
70
90
kHz
●
62
69
78
%
●
7.5
10
12.5
µs
70
150
nA
100
300
nA
0.15
0.4
V
●
0.35
1.0
%/V
●
0.05
0.1
%/V
Oscillator Frequency
DC
Duty Cycle
tON
Switch ON Time
IFB
Feedback Pin Bias Current
LT1110, VFB = 0V
●
ISET
Set Pin Bias Current
VSET = VREF
●
VAO
AO Output Low
IAO = –300µA, VSET = 150mV
●
Reference Line Regulation
1.0V ≤ VIN ≤ 1.5V
1.5V ≤ VIN ≤ 12V
2
MAX
●
fOSC
Full Load (VFB < VREF)
TYP
300
LT1110
ELECTRICAL CHARACTERISTICS TA = 25°C, VIN = 1.5V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
VCESAT
Switch Saturation Voltage
Step-Up Mode
VIN = 1.5V, ISW = 400mA
TYP
MAX
300
400
600
●
VIN = 1.5V, ISW = 500mA
400
●
VIN = 5V, ISW = 1A
700
AV
A2 Error Amp Gain
RL = 100kΩ (Note 3)
IREV
Reverse Battery Current
(Note 4)
ILIM
Current Limit
220Ω Between ILIM and VIN
●
1000
Current Limit Temperature
Coefficient
ILEAK
Switch OFF Leakage Current
Measured at SW1 Pin
VSW2
Maximum Excursion Below GND
ISW1 ≤ 10µA, Switch Off
The ● denotes the specifications which apply over the full operating
temperature range.
Note 1: This specification guarantees that both the high and low trip point
of the comparator fall within the 210mV to 230mV range.
UNITS
mV
mV
550
mV
750
mV
1000
mV
5000
V/V
750
mA
400
mA
– 0.3
%/°C
1
10
µA
– 400
– 350
mV
Note 3: 100kΩ resistor connected between a 5V source and the AO pin.
Note 4: The LT1110 is guaranteed to withstand continuous application of
+1.6V applied to the GND and SW2 pins while VIN, ILIM, and SW1 pins are
grounded.
Note 2: This specification guarantees that the output voltage of the fixed
versions will always fall within the specified range. The waveform at the
sense pin will exhibit a sawtooth shape due to the comparator hysteresis.
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TYPICAL PERFOR A CE CHARACTERISTICS
Oscillator Frequency
Oscillator Frequency
100
Switch On Time
14
80
13
76
80
70
60
12
74
ON TIME (µs)
FREQUENCY (KHz)
OSCILLATOR FREQUENCY (KHz)
78
90
72
70
68
66
10
9
64
50
8
62
40
–50
11
60
–25
0
25
50
75
100
TEMPERATURE (°C)
0
3
6
9
12 15 18 21 24 27 30
INPUT VOLTAGE (V)
LT1110 • TPC01
7
–50
–25
0
25
50
75
100
TEMPERATURE (°C)
LT1110 • TPC02
LT1110 • TPC03
3
LT1110
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TYPICAL PERFOR A CE CHARACTERISTICS
78
500
76
450
74
400
72
350
68
66
200
62
100
60
50
50
25
75
0
50
25
0
100
VIN = 12V
OSCILLATOR FREQUENCY (KHz)
1.0
0.8
0.6
0.4
0.2
100
400
95
90
380
85
80
0°C ≤ TA ≤ 70°C
75
70
65
60
55
50
1.0
8
9
10
11
450
1.3
350
300
250
200
0
25
320
300
280
260
240
0
50
75
100
TEMPERATURE (°C)
LT1110 • TPC10
3
6
9
12 15 18 21 24 27 30
INPUT VOLTAGE (V)
LT1110 • TPC09
Maximum Switch Current vs
RLIM Step-Up
1.5
Maximum Switch Current vs
RLIM Step-Down
1.3
1.1
STEP-UP MODE
VIN ≤ 5V
0.9
0.7
0.5
1.1
0.9
STEP-DOWN MODE
VIN = 12V
0.7
0.5
0.3
0.1
–25
1.6
360
13
0.3
150
4
12
SWITCH CURRENT (A)
1.5
SWITCH CURRENT (A)
QUIESCENT CURRENT (µA)
500
1.4
340
LT1110 • TPC08
Quiescent Current
1.2
Quiescent Current
SWITCH ON TIME (µs)
LT1110 • TPC07
400
0.8 1.0
200
7
ISWITCH (A)
100
–50
0.6
220
40
0.8
0.6
0.4
LT1110 • TPC06
45
0
0.4
0.2
ISWITCH (A)
Minimum/Maximum Frequency vs
On Time
1.4
ON VOLTAGE (V)
75
LT1110 • TPC05
Switch On Voltage
Step-Down Mode
0.2
VIN = 1.0V
TEMPERATURE (°C)
LT1110 • TPC04
1.2
VIN = 5.0V
0
– 25
TEMPERATURE (°C)
0
VIN = 1.2V
0.6
0.2
0
–50
100
0.8
0.4
QUIESCENT CURRENT (µA)
0
VIN = 1.5V
1.0
250
150
VIN = 3.0V
VIN= 2.0V
1.2
300
64
–25
1.4
VIN = 1.5V
ISW = 500mA
VCESAT (V)
70
58
–50
Saturation Voltage
Step-Up Mode
Switch Saturation Voltage
VCESAT (mV)
DUTY CYCLE (%)
Duty Cycle
0.1
10
100
RLIM (Ω)
1000
LT1110 • TPC11
10
100
RLIM (Ω)
1000
LT1110 • TPC12
LT1110
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TYPICAL PERFOR A CE CHARACTERISTICS
Set Pin Bias Current
120
224
100
120
90
BIAS CURRENT (nA)
BIAS CURRENT (nA)
Reference Voltage
226
110
140
100
80
60
40
222
80
70
60
50
40
20
–25
0
25
50
75
218
214
10
0
–50
100
220
216
30
20
0
–50
FB Pin Bias Current
VREF (mV)
160
–25
0
TEMPERATURE (°C)
25
50
75
212
–50
100
TEMPERATURE (°C)
LT1110 • TPC13
–25
0
25
50
75
100
TEMPERATURE (°C)
LT1110 • TPC15
LT1110 • TPC14
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ILIM (Pin 1): Connect this pin to VIN for normal use. Where
lower current limit is desired, connect a resistor between
ILIM and VIN. A 220Ω resistor will limit the switch current
to approximately 400mA.
VIN (Pin 2): Input supply voltage.
SW1 (Pin 3): Collector of power transistor. For step-up
mode connect to inductor/diode. For step-down mode
connect to VIN.
SW2 (Pin 4): Emitter of power transistor. For step-up
mode connect to ground. For step-down mode connect to
inductor/diode. This pin must never be allowed to go more
than a Schottky diode drop below ground.
GND (Pin 5): Ground.
AO (Pin 6): Auxiliary Gain Block (GB) output. Open collector,
can sink 300µA.
SET (Pin 7): GB input. GB is an op amp with positive input
connected to SET pin and negative input connected to
220mV reference.
FB/SENSE (Pin 8): On the LT1110 (adjustable) this pin
goes to the comparator input. On the LT1110-5 and
LT1110-12, this pin goes to the internal application resistor
that sets output voltage.
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LT1110 BLOCK DIAGRA
+
SET
A2
–
V IN
AO
GAIN BLOCK/ERROR AMP
I LIM
SW1
220mV
REFERENCE
A1
COMPARATOR
GND
FB
OSCILLATOR
Q1
DRIVER
SW2
LT1110 • BD01
5
LT1110
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LT1110 OPERATI
The LT1110 is a gated oscillator switcher. This type
architecture has very low supply current because the
switch is cycled only when the feedback pin voltage drops
below the reference voltage. Circuit operation can best be
understood by referring to the LT1110 block diagram
above. Comparator A1 compares the FB pin voltage with
the 220mV reference signal. When FB drops below
220mV, A1 switches on the 70kHz oscillator. The driver
amplifier boosts the signal level to drive the output NPN
power switch Q1. An adaptive base drive circuit senses
switch current and provides just enough base drive to
ensure switch saturation without overdriving the switch,
resulting in higher efficiency. The switch cycling action
raises the output voltage and FB pin voltage. When the FB
voltage is sufficient to trip A1, the oscillator is gated off. A
small amount of hysteresis built into A1 ensures loop
stability without external frequency compensation. When
the comparator is low the oscillator and all high current
circuitry is turned off, lowering device quiescent current to
just 300µA for the reference, A1 and A2.
The oscillator is set internally for 10µs ON time and 5µs
OFF time, optimizing the device for step-up circuits where
VOUT ≈ 3VIN, e.g., 1.5V to 5V. Other step-up ratios as well
as step-down (buck) converters are possible at slight
losses in maximum achievable power output.
A2 is a versatile gain block that can serve as a low battery
detector, a linear post regulator, or drive an under voltage
lockout circuit. The negative input of A2 is internally
connected to the 220mV reference. An external resistor
divider from VIN to GND provides the trip point for A2. The
AO output can sink 300µA (use a 47k resistor pull up to
+5V). This line can signal a microcontroller that the battery
voltage has dropped below the preset level. To prevent the
gain block from operating in its linear region, a 2MΩ
resistor can be connected from AO to SET. This provides
positive feedback.
A resistor connected between the ILIM pin and VIN adjusts
maximum switch current. When the switch current exceeds the set value, the switch is turned off. This feature
is especially useful when small inductance values are used
with high input voltages. If the internal current limit of 1.5A
is desired, ILIM should be tied directly to VIN. Propagation
delay through the current limit circuitry is about 700ns.
In step-up mode, SW2 is connected to ground and SW1
drives the inductor. In step-down mode, SW1 is connected to VIN and SW2 drives the inductor. Output voltage
is set by the following equation in either step-up or stepdown modes where R1 is connected from FB to GND and
R2 is connected from VOUT to FB.
 R2 
VOUT = 220mV 
+ 1 .
 R1 
(
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+
A2
AO
–
V IN
GAIN BLOCK/ERROR AMP
I LIM
SW1
220mV
REF
A1
COMPARATOR
R1
6
Q1
DRIVER
SW2
R2
300kΩ
SENSE
GND
OSCILLATOR
LT1110-5: R1 = 13.8kΩ
LT1110-12: R2 = 5.6kΩ
LT1110 • BD02
(01)
LT1110 --5, -12 OPERATI
LT1110-5, -12 BLOCK DIAGRA
SET
)
The LT1110-5 and LT1110-12 fixed output voltage versions have the gain setting resistors on-chip. Only three
external components are required to construct a 5V or 12V
output converter. 16µA flows through R1 and R2 in the
LT1110-5, and 39µA flows in the LT1110-12. This current
represents a load and the converter must cycle from time
to time to maintain the proper output voltage. Output
ripple, inherently present in gated oscillator designs, will
typically run around 90mV for the LT1110-5 and 200mV
for the LT1110-12 with the proper inductor/capacitor
selection. This output ripple can be reduced considerably
by using the gain block amp as a pre-amplifier in front of
the FB pin. See the Applications section for details.
LT1110
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APPLICATI
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Inductor Selection — General
A DC-DC converter operates by storing energy as magnetic flux in an inductor core, and then switching this
energy into the load. Since it is flux, not charge, that is
stored, the output voltage can be higher, lower, or opposite in polarity to the input voltage by choosing an appropriate switching topology. To operate as an efficient energy transfer element, the inductor must fulfill three requirements. First, the inductance must be low enough for
the inductor to store adequate energy under the worst
case condition of minimum input voltage and switch ON
time. The inductance must also be high enough so maximum current ratings of the LT1110 and inductor are not
exceeded at the other worst case condition of maximum
input voltage and ON time. Additionally, the inductor core
must be able to store the required flux; i.e., it must not
saturate. At power levels generally encountered with
LT1110 based designs, small surface mount ferrite core
units with saturation current ratings in the 300mA to 1A
range and DCR less than 0.4Ω (depending on application)
are adequate. Lastly, the inductor must have sufficiently
low DC resistance so excessive power is not lost as heat
in the windings. An additional consideration is ElectroMagnetic Interference (EMI). Toroid and pot core type
inductors are recommended in applications where EMI
must be kept to a minimum; for example, where there are
sensitive analog circuitry or transducers nearby. Rod core
types are a less expensive choice where EMI is not a
problem. Minimum and maximum input voltage, output
voltage and output current must be established before an
inductor can be selected.
Inductor Selection — Step-Up Converter
In a step-up, or boost converter (Figure 4), power generated by the inductor makes up the difference between
input and output. Power required from the inductor is
determined by
(
)(
PL = VOUT + V D – VIN MIN IOUT
)
(01)
where VD is the diode drop (0.5V for a 1N5818 Schottky).
Energy required by the inductor per cycle must be equal or
greater than
PL
(02)
fOSC
in order for the converter to regulate the output.
When the switch is closed, current in the inductor builds
according to
–R't 
V 
IL ( t) = IN  1– e L 
R' 

(03)
where R' is the sum of the switch equivalent resistance
(0.8Ω typical at 25°C) and the inductor DC resistance.
When the drop across the switch is small compared to VIN,
the simple lossless equation
()
V
I L t = IN t
(04)
L
can be used. These equations assume that at t = 0,
inductor current is zero. This situation is called “discontinuous mode operation” in switching regulator parlance.
Setting “t” to the switch ON time from the LT1110 specification table (typically 10µs) will yield IPEAK for a specific
“L” and VIN. Once IPEAK is known, energy in the inductor
at the end of the switch ON time can be calculated as
1 2
LI
(05)
2 PEAK
EL must be greater than PL/fOSC for the converter to deliver
the required power. For best efficiency IPEAK should be
kept to 1A or less. Higher switch currents will cause
excessive drop across the switch resulting in reduced
efficiency. In general, switch current should be held to as
low a value as possible in order to keep switch, diode and
inductor losses at a minimum.
EL =
As an example, suppose 12V at 120mA is to be generated
from a 4.5V to 8V input. Recalling equation (01),
(
)(
)
P L = 12 V + 0.5 V – 4.5 V 120mA = 960mW. (06)
Energy required from the inductor is
960mW
PL
=
= 13.7µJ.
f OSC 70kHz
(07)
7
LT1110
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APPLICATI
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VOUT = output voltage
VIN = minimum input voltage
Picking an inductor value of 47µH with 0.2Ω DCR results
in a peak switch current of
I PEAK =
−1.0 W•10ms 
4.5 V 
1 − e 47mH  = 862mA.
1.0 W 

(08)
Substituting IPEAK into Equation 05 results in
EL =
(
)(
)
1
47µH 0.862 A 2 = 17.5µJ.
2
Once IPEAK is known, inductor value can be derived from
(09)
Since 17.5µJ > 13.7µJ, the 47µH inductor will work. This
trial-and-error approach can be used to select the optimum inductor. Keep in mind the switch current maximum
rating of 1.5A. If the calculated peak current exceeds this,
an external power transistor can be used.
A resistor can be added in series with the ILIM pin to invoke
switch current limit. The resistor should be picked such
that the calculated IPEAK at minimum VIN is equal to the
Maximum Switch Current (from Typical Performance
Characteristic curves). Then, as VIN increases, switch
current is held constant, resulting in increasing efficiency.
Inductor Selection — Step-Down Converter
After establishing output voltage, output current and input
voltage range, peak switch current can be calculated by the
formula
2 IOUT  V OUT + V D 
DC  V IN – V SW + V D 
where DC = duty cycle (0.69)
VSW = switch drop in step-down mode
VD = diode drop (0.5V for a 1N5818)
IOUT = output current
8
L=
V IN MIN – V SW – V OUT
• tON
IPEAK
(11)
where tON = switch ON time (10µs).
Next, the current limit resistor RLIM is selected to give
IPEAK from the RLIM Step-Down Mode curve. The addition
of this resistor keeps maximum switch current constant as
the input voltage is increased.
As an example, suppose 5V at 250mA is to be generated
from a 9V to 18V input. Recalling Equation (10),
IPEAK =
(
)
2 250mA  5 + 0.5 
= 498mA . (12)
0.69  9 – 1.5 + 0.5 
Next, inductor value is calculated using Equation (11)
The step-down case (Figure 5) differs from the step-up in
that the inductor current flows through the load during
both the charge and discharge periods of the inductor.
Current through the switch should be limited to ~800mA
in this mode. Higher current can be obtained by using an
external switch (see Figure 6). The ILIM pin is the key to
successful operation over varying inputs.
IPEAK =
VSW is actually a function of switch current which is in turn
a function of VIN, L, time and VOUT. To simplify, 1.5V can
be used for VSW as a very conservative value.
(10)
L=
9 – 1.5 – 5
• 10µs = 50µH.
498mA
(13)
Use the next lowest standard value (47µH).
Then pick RLIM from the curve. For IPEAK = 500mA,
RLIM = 82Ω.
Inductor Selection — Positive-to-Negative Converter
Figure 7 shows hookup for positive-to-negative conversion. All of the output power must come from the inductor.
In this case,
(
)(
)
P L = | VOUT | + V D IOUT .
(14)
In this mode the switch is arranged in common collector
or step-down mode. The switch drop can be modeled as
a 0.75V source in series with a 0.65Ω resistor. When the
LT1110
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APPLICATI
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()
IL + =
VL
R'
–R't 

–
1
e
L 



(15)
where R' = 0.65Ω + DCRL
VL = VIN – 0.75V
As an example, suppose –5V at 75mA is to be generated
from a 4.5V to 5.5V input. Recalling Equation (14),
(
)(
)
P L = | −5 V | + 0.5 V 75mA = 413mW.
Energy required from the inductor is
413mW
PL
=
= 5.9µJ.
70kHz
fOSC
(16)
(17)
Picking an inductor value of 56µH with 0.2Ω DCR results
in a peak switch current of
IPEAK =
(4.5V – 0.75V) 1 – e –0.85Ω • 10µs  = 621mA .
56µH


(0.65Ω + 0.2Ω) 
(18)
capacitors provide still better performance at more expense. We recommend OS-CON capacitors from Sanyo
Corporation (San Diego, CA). These units are physically
quite small and have extremely low ESR. To illustrate,
Figures 1, 2 and 3 show the output voltage of an LT1110
based converter with three 100µF capacitors. The peak
switch current is 500mA in all cases. Figure 1 shows a
Sprague 501D, 25V aluminum capacitor. VOUT jumps by
over 120mV when the switch turns off, followed by a drop
in voltage as the inductor dumps into the capacitor. This
works out to be an ESR of over 240mΩ. Figure 2 shows the
same circuit, but with a Sprague 150D, 20V tantalum
capacitor replacing the aluminum unit. Output jump is
now about 35mV, corresponding to an ESR of 70mΩ.
Figure 3 shows the circuit with a 16V OS-CON unit. ESR is
now only 20mΩ.
50mV/DIV
switch closes, current in the inductor builds according to
Substituting IPEAK into Equation (04) results in
EL =
(
)(
)
1
56µH 0.621A 2 = 10.8µJ.
2
5µs/DIV
(19)
LT1110 • TA19
Figure 1. Aluminum
With this relatively small input range, RLIM is not usually
necessary and the ILIM pin can be tied directly to VIN. As in
the step-down case, peak switch current should be limited
to ~800mA.
50mV/DIV
Since 10.8µJ > 5.9µJ, the 56µH inductor will work.
Capacitor Selection
LT1110 • TA20
Figure 2. Tantalum
50mV/DIV
Selecting the right output capacitor is almost as important
as selecting the right inductor. A poor choice for a filter
capacitor can result in poor efficiency and/or high output
ripple. Ordinary aluminum electrolytics, while inexpensive
and readily available, may have unacceptably poor Equivalent Series Resistance (ESR) and ESL (inductance). There
are low ESR aluminum capacitors on the market specifically designed for switch mode DC-DC converters which
work much better than general-purpose units. Tantalum
5µs/DIV
5µs/DIV
LT1110 • TA21
Figure 3. OS-CON
9
LT1110
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Diode Selection
Speed, forward drop, and leakage current are the three
main considerations in selecting a catch diode for LT1110
converters. General purpose rectifiers such as the 1N4001
are unsuitable for use in any switching regulator application. Although they are rated at 1A, the switching time of
a 1N4001 is in the 10µs-50µs range. At best, efficiency will
be severely compromised when these diodes are used; at
worst, the circuit may not work at all. Most LT1110 circuits
will be well served by a 1N5818 Schottky diode, or its
surface mount equivalent, the MBRS130T3. The combination of 500mV forward drop at 1A current, fast turn ON and
turn OFF time, and 4µA to 10µA leakage current fit nicely
with LT1110 requirements. At peak switch currents of
100mA or less, a 1N4148 signal diode may be used. This
diode has leakage current in the 1nA-5nA range at 25°C
and lower cost than a 1N5818. (You can also use them to
get your circuit up and running, but beware of destroying
the diode at 1A switch currents.)
Immediately after switch turn off, the SW1 voltage pin
starts to rise because current cannot instantaneously stop
flowing in L1. When the voltage reaches VOUT + VD, the
inductor current flows through D1 into C1, increasing
VOUT. This action is repeated as needed by the LT1110 to
keep VFB at the internal reference voltage of 220mV. R1
and R2 set the output voltage according to the formula
 R2 
VOUT =  1 +  220mV .
R1

(
)
(21)
Step-Down (Buck Mode) Operation
A step-down DC-DC converter converts a higher voltage
to a lower voltage. The usual hookup for an LT1110 based
step-down converter is shown in Figure 5.
VIN
R3
220 Ω
+
C2
I LIM
V IN
SW1
FB
Step-Up (Boost Mode) Operation
LT1110
L1
A step-up DC-DC converter delivers an output voltage
higher than the input voltage. Step-up converters are not
short circuit protected since there is a DC path from input
to output.
The usual step-up configuration for the LT1110 is shown
in Figure 4. The LT1110 first pulls SW1 low causing VIN –
VCESAT to appear across L1. A current then builds up in L1.
At the end of the switch ON time the current in L1 is1:
VIN
IPEAK =
t ON
(20)
L
L1
V OUT
R3*
I LIM
V IN
SW1
LT1110
GND
R2
+
C1
FB
SW2
R1
* = OPTIONAL
LT1110 • TA14
Figure 4. Step-Up Mode Hookup.
10
GND
R2
D1
1N5818
+
C1
R1
LT1110 • TA15
Figure 5. Step-Down Mode Hookup
When the switch turns on, SW2 pulls up to VIN – VSW. This
puts a voltage across L1 equal to VIN – VSW – VOUT,
causing a current to build up in L1. At the end of the switch
ON time, the current in L1 is equal to
D1
V IN
VOUT
SW2
IPEAK =
VIN − VSW − VOUT
L
t ON .
(22)
When the switch turns off, the SW2 pin falls rapidly and
actually goes below ground. D1 turns on when SW2
reaches 0.4V below ground. D1 MUST BE A SCHOTTKY
DIODE. The voltage at SW2 must never be allowed to go
below –0.5V. A silicon diode such as the 1N4933 will allow
SW2 to go to –0.8V, causing potentially destructive power
Note 1: This simple expression neglects the effects of switch and coil
resistance. This is taken into account in the “Inductor Selection” section.
LT1110
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dissipation inside the LT1110. Output voltage is determined by
 R2 
VOUT =  1 +  220mV .
R1

(
)
(23)
R3 programs switch current limit. This is especially important in applications where the input varies over a wide
range. Without R3, the switch stays on for a fixed time
each cycle. Under certain conditions the current in L1 can
build up to excessive levels, exceeding the switch rating
and/or saturating the inductor. The 220Ω resistor programs the switch to turn off when the current reaches
approximately 800mA. When using the LT1110 in stepdown mode, output voltage should be limited to 6.2V or
less. Higher output voltages can be accommodated by
inserting a 1N5818 diode in series with the SW2 pin
(anode connected to SW2).
Converter” section with the following conservative expression for VSW:
V SW = V R1 + V SAT ≈ 0.9 V .
(24)
R2 provides a current path to turn off Q1. R3 provides base
drive to Q1. R4 and R5 set output voltage.
Inverting Configurations
The LT1110 can be configured as a positive-to-negative
converter (Figure 7), or a negative-to-positive converter
(Figure 8). In Figure 7, the arrangement is very similar to
a step-down, except that the high side of the feedback is
referred to ground. This level shifts the output negative. As
in the step-down mode, D1 must be a Schottky diode,
and VOUTshould be less than 6.2V. More negative output voltages can be accommodated as in the prior section.
+VIN
R3
Higher Current Step-Down Operation
Output current can be increased by using a discrete PNP
pass transistor as shown in Figure 6. R1 serves as a
current limit sense. When the voltage drop across R1
equals a VBE, the switch turns off. For temperature compensation a Schottky diode can be inserted in series with
the ILIM pin. This also lowers the maximum drop across R1
to VBE – VD, increasing efficiency. As shown, switch
current is limited to 2A. Inductor value can be calculated
based on formulas in the “Inductor Selection Step-Down
Q1
MJE210 OR
ZETEX ZTX789A
R1
0.3Ω
VIN
25V
MAX
L1
VOUT
R2
220
+
VIN
FB
+
C2
LT1110
L1
SW2
GND
R1
D1
1N5818
+
C1
R2
–VOUT
LT1110 • TA03
Figure 7. Positive-to-Negative Converter
In Figure 8, the input is negative while the output is
positive. In this configuration, the magnitude of the input
voltage can be higher or lower than the output voltage. A
level shift, provided by the PNP transistor, supplies proper
polarity feedback information to the regulator.
D1
+
+
C1
C1
I LIM
LT1110
R4
+
FB
GND
SW1
+VOUT
D1
1N5821
SW1
C2
V IN
L1
R3
330
IL
I LIM
C2
SW2
R5
VOUT = 220mV
(
R4
1 + R5
)
Figure 6. Q1 Permits Higher-Current Switching.
LT1110 Functions as Controller.
2N3906
LT1110
AO
GND
LT1110 • TA16
V IN
SW1
R1
FB
SW2
R2
VOUT =
220mV + 0.6V
( R1
R2 )
–VIN
LT1110 • TA04
Figure 8. Negative-to-Positive Converter
11
LT1110
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Using the ILIM Pin
The LT1110 switch can be programmed to turn off at a set
switch current, a feature not found on competing devices.
This enables the input to vary over a wide range without
exceeding the maximum switch rating or saturating the
inductor. Consider the case where analysis shows the
LT1110 must operate at an 800mA peak switch current
with a 2.0V input. If VIN rises to 4V, peak current will rise
to 1.6A, exceeding the maximum switch current rating.
With the proper resistor selected (see the “Maximum
Switch Current vs RLIM” characteristic), the switch current
will be limited to 800mA, even if the input voltage
increases.
switch ON times less than 3µs. Resistor values programming switch ON time for 800ns or less will cause spurious
response in the switch circuitry although the device will
still maintain output regulation.
IL
SWITCH
ON
OFF
LT1110 • TA05
Another situation where the ILIM feature is useful occurs
when the device goes into continuous mode operation.
This occurs in step-up mode when
VOUT + VDIODE
VI N − VSW
<
1
.
1 − DC
IL
(25)
When the input and output voltages satisfy this relationship, inductor current does not go to zero during the
switch OFF time. When the switch turns on again, the
current ramp starts from the non-zero current level in the
inductor just prior to switch turn on. As shown in Figure 9,
the inductor current increases to a high level before the
comparator turns off the oscillator. This high current can
cause excessive output ripple and requires oversizing the
output capacitor and inductor. With the ILIM feature,
however, the switch current turns off at a programmed
level as shown in Figure 10, keeping output ripple to a
minimum.
Figure 11 details current limit circuitry. Sense transistor
Q1, whose base and emitter are paralleled with power
switch Q2, is ratioed such that approximately 0.5% of Q2’s
collector current flows in Q1’s collector. This current is
passed through internal 80Ω resistor R1 and out through
the ILIM pin. The value of the external resistor connected
between ILIM and VIN set the current limit. When sufficient
switch current flows to develop a VBE across R1 + RLIM, Q3
turns on and injects current into the oscillator, turning off
the switch. Delay through this circuitry is approximately
800ns. The current trip point becomes less accurate for
12
Figure 9. No Current Limit Causes Large Inductor
Current Build-Up
SWITCH
PROGRAMMED CURRENT LIMIT
ON
OFF
LT1110 • TA06
Figure 10. Current Limit Keeps Inductor Current Under Control
RLIM
(EXTERNAL)
VIN
ILIM
R1
80Ω
(INTERNAL)
Q3
SW1
DRIVER
OSCILLATOR
Q1
Q2
SW2
LT1110 • TA17
Figure 11. LT1110 Current Limit Circuitry
Using the Gain Block
The gain block (GB) on the LT1110 can be used as an error
amplifier, low battery detector or linear post regulator. The
gain block itself is a very simple PNP input op amp with an
open collector NPN output. The negative input of the gain
block is tied internally to the 220mV reference. The positive input comes out on the SET pin.
LT1110
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Arrangement of the gain block as a low battery detector is
straightforward. Figure 12 shows hookup. R1 and R2 need
only be low enough in value so that the bias current of the
SET input does not cause large errors. 33kΩ for R2 is
adequate. R3 can be added to introduce a small amount of
hysteresis. This will cause the gain block to “snap” when
the trip point is reached. Values in the 1M-10M range are
optimal. The addition of R3 will change the trip point,
however.
+5V
VIN
R1
47k
L1
D1
LT1110
220mV
REF
VBAT
Output ripple of the LT1110, normally 90mV at 5VOUT can
be reduced significantly by placing the gain block in front
of the FB input as shown in Figure 13. This effectively
reduces the comparator hysteresis by the gain of the gain
block. Output ripple can be reduced to just a few millivolts
using this technique. Ripple reduction works with stepdown or inverting modes as well. For this technique to be
effective, output capacitor C1 must be large, so that each
switching cycle increases VOUT by only a few millivolts.
1000µF is a good starting value.
SET
V OUT
R3
270k
–
AO
TO
PROCESSOR
+
V IN
SW1
I LIM
AO
VBAT
+
C1
LT1110
GND
FB
GND
R2
SET
SW2
R1
R3
R1 =
R2
– 220mV
( VLB4.33µA
)
VLB = BATTERY TRIP POINT
R2 = 33kΩ
R3 = 2MΩ
(
)(
)
VOUT = R2 + 1 220mV
R1
LT1110 • TA08
LT1110 • TA07
Figure 12. Setting Low Battery Detector Trip Point
Table 1. Inductor Manufacturers
Figure 13. Output Ripple Reduction Using Gain Block
Table 2. Capacitor Manufacturers
MANUFACTURER
PART NUMBERS
MANUFACTURER
PART NUMBERS
Coiltronics International
984 S.W. 13th Court
Pompano Beach, FL 33069
305-781-8900
CTX100-4 Series
Surface Mount
Sanyo Video Components
2001 Sanyo Avenue
San Diego, CA 92173
619-661-6835
OS-CON Series
Sumida Electric Co. USA
708-956-0666
CD54
CDR74
CDR105
Surface Mount
Nichicon America Corporation
927 East State Parkway
Schaumberg, IL 60173
708-843-7500
PL Series
Sprague Electric Company
Lower Main Street
Sanford, ME 04073
207-324-4140
150D Solid Tantalums
550D Tantalex
Matsuo
714-969-2491
267 Series
Surface Mount
Table 3. Transistor Manufacturers
MANUFACTURER
PART NUMBERS
Zetex
Commack, NY
516-543-7100
ZTX Series
FZT Series
Surface Mount
13
LT1110
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TYPICAL APPLICATI
S
All Surface Mount
Flash Memory VPP Generator
L1*
47µH
+5V
±10%
MBRS12OT3
MMBT4403
10k
V IN
SW1
I LIM
+
22µF
LT1110CS8-12
1k
SENSE
SW2
GND
1 = PROGRAM
0 = SHUTDOWN
+
MMBF170
VPP
12V
120MA
47µF
20V
LT1110 • TA18
*L1= SUMIDA CD105-470M
1.5V Powered Laser Diode Driver
TOSHIBA
TOLD-9211
22nF
220Ω
4.7k
2N3906
1N4148
1
I LIM
6
AO
1.5V
2
V IN
3
SW1
+
10Ω
C1
100 µ F
OS-CON
0.22 µ F
CERAMIC
2Ω
LT1110
8
* ADJUST R1
MJE210
FB
GND
5
SET
SW2
4
1N5818
7
1k*
R1
L1✝
2.2 µ H
FOR CHANGE IN LASER OUTPUT POWER
✝ TOKO 262LYF-0076M
• LASER DIODE CASE COMMON TO +BATTERY TERMINAL
• 170mA CURRENT DRAIN FROM 1.5V CELL (50mA DIODE)
• NO OVERSHOOT
1.5V Powered Laser Diode Driver
14
LT1110 • TA13
LT1110
UO
TYPICAL APPLICATI
S
All Surface Mount
3V to 5V Step-Up Converter
All Surface Mount
9V to 5V Step-Down Converter
L1*
47µH
220
220
V IN
SW1
I LIM
I LIM
3V
2x
AA CELL
V IN
SW1
MBRL120
LT1110-5
9V
LT1110-5
GND
SENSE
SW2
5V
40mA
+
SENSE
SW2
GND
L1*
47µH
10µF
MBRL120
5V
40mA
+
10µF
*L1 = COILCRAFT 1812LS-473
LT1110 • TA09
*L1 = COILCRAFT 1812LS-473
LT1110 • TA10
All Surface Mount
1.5V to +10V, +5V Dual Output Step-Up Converter
L1*
82µH
I LIM
1.5V
AA OR
AAA
CELL
L1*
82µH
4.7µF
+
+10V
3mA
V IN
SW1
I LIM
490k
LT1110
GND
All Surface Mount
1.5V to ±5V Dual Output Step-Up Converter
+5V
3mA
FB
SW2
1.5V
AA OR
AAA
CELL
V IN
SW1
4.7µF
11k
4.7µF
LT1110
GND
+5V
4mA
SENSE
SW2
+
+
+
–5V
4mA
+
4.7µF
4.7µF
+
= MBRL120
= MBRL120
4.7µF
LT1110 • TA12
*L1 = COILCRAFT 1812LS-823
*L1 = COILCRAFT 1812LS-823
LT1110 • TA11
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LT1110
U
PACKAGE DESCRIPTIO
Dimensions in inches (millimeters) unless otherwise noted.
N8 Package
8-Lead Plastic DIP
0.300 – 0.320
(7.620 – 8.128)
0.009 – 0.015
(0.229 – 0.381)
(
+0.025
0.325 –0.015
+0.635
8.255
–0.381
)
0.130 ± 0.005
(3.302 ± 0.127)
0.045 – 0.065
(1.143 – 1.651)
0.400
(10.160)
MAX
8
7
6
5
0.065
(1.651)
TYP
0.250 ± 0.010
(6.350 ± 0.254)
0.125
(3.175)
MIN
0.045 ± 0.015
(1.143 ± 0.381)
0.020
(0.508)
MIN
1
2
4
3
0.018 ± 0.003
(0.457 ± 0.076)
0.100 ± 0.010
(2.540 ± 0.254)
S8 Package
8-Lead Plastic SOIC
0.189 – 0.197*
(4.801 – 5.004)
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
0.053 – 0.069
(1.346 – 1.752)
0°– 8° TYP
0.016 – 0.050
0.406 – 1.270
0.014 – 0.019
(0.355 – 0.483)
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006 INCH (0.15mm).
16
Linear Technology Corporation
8
7
6
5
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
BSC
0.150 – 0.157*
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
2
3
4
LT/GP 0594 2K REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7487
(408) 432-1900 ● FAX: (408) 434-0507 ● TELEX: 499-3977
 LINEAR TECHNOLOGY CORPORATION 1994