OPA861 SBOS338 – AUGUST 2005 Wide Bandwidth OPERATIONAL TRANSCONDUCTANCE AMPLIFIER (OTA) FEATURES DESCRIPTION • • • • • The OPA861 is a versatile monolithic component designed for wide-bandwidth systems, including high performance video, RF and IF circuitry. The OPA861 is a wideband, bipolar operational transconductance amplifier (OTA). Wide Bandwidth (80MHz, Open-Loop, G = +5) High Slew Rate (900V/µs) High Transconductance (95mA/V) External IQ-Control Low Quiescent Current (5.4mA) APPLICATIONS • • • • • • • Video/Broadcast Equipment Communications Equipment High-Speed Data Acquisition Wideband LED Drivers Control Loop Amplifiers Wideband Active Filters Line Drivers The OTA or voltage-controlled current source can be viewed as an ideal transistor. Like a transistor, it has three terminals—a high impedance input (base), a low-impedance input/output (emitter), and the current output (collector). The OPA861, however, is self-biased and bipolar. The output collector current is zero for a zero base-emitter voltage. AC inputs centered about zero produce an output current, which is bipolar and centered about zero. The transconductance of the OPA861 can be adjusted with an external resistor, allowing bandwidth, quiescent current, and gain trade-offs to be optimized. Used as a basic building block, the OPA861 simplifies the design of AGC amplifiers, LED driver circuits for fiber optic transmission, integrators for fast pulses, fast control loop amplifiers and control amplifiers for capacitive sensors and active filters. The OPA861 is available in SO-8 and SOT23-6 surface-mount packages. 0 −10 R C1 R V IN V OUT C2 Gain (dB) −20 −30 10MHz Low−Pass Filter −40 20kHz Low−Pass Filter −50 −60 −70 −80 1k 10k 100k 1M 10M 100M 1G Frequency (Hz) Low−Pass Negative Impedance Converter (NIC) Filter Frequency Response of 20kHz and 10MHz Low−Pass NIC Filters Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2005, Texas Instruments Incorporated OPA861 www.ti.com SBOS338 – AUGUST 2005 This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. ORDERING INFORMATION (1) (1) SPECIFIED TEMPERATURE RANGE PACKAGE MARKING PRODUCT PACKAGE PACKAGE DESIGNATOR OPA861 SO-8 D –45°C to +85°C OPA861 OPA861 SOT23-6 DBV –45°C to +85°C N5R ORDERING NUMBER TRANSPORT MEDIA, QUANTITY OPA861ID Rails, 75 OPA861IDR Tape and Reel, 2500 OPA861IDBVT Tape and Reel, 250 OPA861IDBVR Tape and Reel, 3000 For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI web site at www.ti.com. ABSOLUTE MAXIMUM RATINGS (1) ±6.5VDC Power Supply Internal Power Dissipation See Thermal Information ±1.2V Differential Input Voltage ±VS Input Common-Mode Voltage Range Storage Temperature Range: D –40°C to +125°C Lead Temperature (soldering, 10s) +260°C Junction Temperature (TJ) +150°C ESD Rating: (1) (2) Human Body Model (HBM) (2) 1500V Charge Device Model (CDM) 1000V Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. These are stress ratings only, and functional operations of the device at these and any other conditions beyond those specified is not supported. Pin 2 for the SO-8 package > 500V HBM. Pin 4 for the SOT23-6 package > 500V HBM. PIN CONFIGURATION Top View I Q Adjust 1 8 C E 2 7 V+ = +5V B 3 6 NC V− = −5V 4 5 NC SO−8 2 I Q Adjust 1 6 +VS −VS 2 5 C B 3 4 E SOT23−6 OPA861 www.ti.com SBOS338 – AUGUST 2005 ELECTRICAL CHARACTERISTICS: VS = ±5V RL = 500Ω and RADJ = 250Ω, unless otherwise noted. OPA861ID, IDBV TYP PARAMETER MIN/MAX OVER TEMPERATURE CONDITIONS +25°C +25°C (2) 0°C to 70°C (3) –40°C to +85°C (3) UNITS MIN/ MAX TEST LEVEL (1) G = +5, VO = 200mVPP, RL = 500Ω 80 77 75 74 MHz min B G = +5, VO = 1VPP 80 MHz typ C G = +5, VO = 5VPP 80 MHz typ C G = +5, VO = 5V Step 900 V/µs min B VO = 1V Step 4.4 ns typ C OTA — Open-Loop (see Figure 30) AC PERFORMANCE Bandwidth Slew Rate Rise Time and Fall Time Harmonic Distortion 860 850 840 G = +5, VO = 2VPP, 5MHz 2nd-Harmonic RL = 500Ω –68 –55 –54 –53 dB max B 3rd-Harmonic RL = 500Ω –57 –52 –51 –49 dB max B Base Input Voltage Noise f > 100kHz 2.4 3.0 3.3 3.4 nV/√Hz max B Base Input Current Noise f > 100kHz 1.7 2.4 2.45 2.5 pA/√Hz max B Emitter Input Current Noise f > 100kHz 5.2 15.3 16.6 17.5 pA/√Hz max B Minimum OTA Transconductance (gm) VO = ±10mV, RC = 50Ω, RE = 0Ω 95 80 77 75 mA/V min A Maximum OTA Transconductance (gm) VO = ±10mV, RC = 50Ω, RE = 0Ω 95 150 155 160 mA/V max A VB = 0V, RC = 0Ω, RE = 100Ω ±3 ±12 ±15 ±20 mV max A ±67 ±120 µV/°C max B ±6 ±6.6 µA max A ±20 ±25 nA/°C max B ±125 ±140 µA max A ±500 ±600 nA/°C max B ±30 ±38 µA max A ±250 ±300 nA/°C max B ±3.6 ±3.6 OTA DC PERFORMANCE (4) (see Figure 30) B-Input Offset Voltage Average B-Input Offset Voltage Drift B-Input Bias Current Average B-Input Bias Current Drift E-Input Bias Current Average E-Input Bias Current Drift C-Output Bias Current Average C-Output Bias Current Drift VB = 0V, RC = 0Ω, RE = 100Ω VB = 0V, RC = 0Ω, RE = 100Ω ±1 ±5 VB = 0V, RC = 0Ω, RE = 100Ω VB = 0V, VC = 0V ±30 ±100 VB = 0V, VC = 0V VB = 0V, VC = 0V ±5 ±18 VB = 0V, VC = 0V OTA INPUT (see Figure 30) ±4.2 B-Input Voltage Range B-Input Impedance ±3.7 455 || 2.1 V min B kΩ || pF typ C Min E-Input Resistance 10.5 12.5 13.0 13.3 Ω max B Max E-Input Resistance 10.5 6.7 6.5 6.3 Ω min B IE = ±1mA ±4.2 ±3.7 ±3.6 ±3.6 V min A VE = 0 ±15 ±10 ±9 ±9 mA min A IC = ±1mA ±4.7 ±4.0 ±3.9 ±3.9 V min A VC = 0 ±15 ±10 ±9 ±9 mA min A kΩ || pF typ C OTA OUTPUT E-Output Voltage Compliance E-Output Current, Sinking/Sourcing C-Output Voltage Compliance C-Output Current, Sinking/Sourcing C-Output Impedance (1) (2) (3) (4) 54 || 2 Test levels: (A) 100% tested at 25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. Junction temperature = ambient for 25°C specifications. Junction temperature = ambient at low temperature limit; junction temperature = ambient + 7°C at high temperature limit for over temperature specifications. Current is considered positive out of node. 3 OPA861 www.ti.com SBOS338 – AUGUST 2005 ELECTRICAL CHARACTERISTICS: VS = ±5V (continued) RL = 500Ω and RADJ = 250Ω, unless otherwise noted. OPA861ID, IDBV TYP MIN/MAX OVER TEMPERATURE +25°C (2) 0°C to 70°C (3) –40°C to +85°C (3) Maximum Operating Voltage ±6.3 ±6.3 ±6.3 Minimum Operating Voltage ±2.0 ±2.0 PARAMETER CONDITIONS +25°C MIN/ MAX TEST LEVEL (1) V typ C V max A ±2.0 V min B UNITS POWER SUPPLY ±5 Specified Operating Voltage Maximum Quiescent Current RADJ = 250Ω 5.4 5.9 7.0 7.4 mA max A Minimum Quiescent Current RADJ = 250Ω 5.4 4.9 4.3 3.4 mA min A ∆IC/∆VS ±20 ±50 ±60 ±65 µA/V max A –40 to +85 °C typ C OTA Power-Supply Rejection Ratio (+PSRR) THERMAL CHARACTERISTICS Specification: ID, IDBV Thermal Resistance θJA D SO-8 Junction-to-Ambient 125 °C/W typ C DBV SOT23-6 Junction-to-Ambient 150 °C/W typ C 4 OPA861 www.ti.com SBOS338 – AUGUST 2005 ELECTRICAL CHARACTERISTICS: VS = +5V RL = 500Ω to VS/2 and RADJ = 250Ω, unless otherwise noted. OPA861ID, IDBV TYP MIN/MAX OVER TEMPERATURE CONDITIONS +25°C +25°C (2) 0°C to 70°C (3) –40°C to +85°C (3) UNITS MIN/ MAX TEST LEVEL (1) Bandwidth G = +5, VO = 200mVPP, RL = 500Ω 73 72 72 70 MHz min B G = +5, VO = 1VPP 73 MHz typ C Slew Rate G = +5, VO = 2.5V Step 410 395 390 390 V/µs min B VO = 1V Step 4.4 ns typ C PARAMETER OTA—Open-Loop (see Figure 30) AC PERFORMANCE Rise Time and Fall Time Harmonic Distortion G = +5, VO = 2VPP, 5MHz 2nd-Harmonic RL = 500Ω –67 –55 –54 –54 dB max B 3rd-Harmonic RL = 500Ω –57 –50 –49 –48 dB max B Base Input Voltage Noise f > 100kHz 2.4 3.0 3.3 3.4 nV/√Hz max B Base Input Current Noise f > 100kHz 1.7 2.4 2.45 2.5 pA/√Hz max B Emitter Input Current Noise f > 100kHz 5.2 15.3 16.6 17.5 pA/√Hz max B Minimum OTA Transconductance (gm) VO = ±10mV, RC = 50Ω, RE = 0Ω 85 70 67 65 mA/V min A Maximum OTA Transconductance (gm) VO = ±10mV, RC = 50Ω, RE = 0Ω 85 140 145 150 mA/V max A VB = 0V, RC = 0Ω, RE = 100Ω ±3 ±12 ±15 ±20 mV max A ±67 ±120 µV/°C max B ±6 ±6.6 µA max A ±20 ±25 nA/°C max B ±125 ±140 µA max A ±500 ±600 nA/°C max B µA typ C B OTA DC PERFORMANCE (4) (see Figure 30) B-Input Offset Voltage Average B-Input Offset Voltage Drift B-Input Bias Current Average B-Input Bias Current Drift E-Input Bias Current Average E-Input Bias Current Drift C-Output Bias Current VB = 0V, RC = 0Ω, RE = 100Ω VB = 0V, RC = 0Ω, RE = 100Ω ±1 ±5 VB = 0V, RC = 0Ω, RE = 100Ω VB = 0V, VC = 0V ±30 ±100 VB = 0V, VC = 0V VB = 0V, VC = 0V ±15 OTA INPUT (see Figure 30) Most Positive B-Input Voltage 4.2 3.7 3.6 3.6 V min Least Positive B-Input Voltage 0.8 1.3 1.4 1.4 V max B kΩ || pF typ C B-Input Impedance 455 || 2.1 Min E-Input Resistance 11.8 14.4 14.9 15.4 Ω max B Max E-Input Resistance 11.8 7.1 6.9 6.7 Ω min B OTA OUTPUT Maximum E-Output Voltage Compliance IE = ±1mA 4.2 3.7 3.6 3.6 V min A Minimum E-Output Voltage Compliance IE = ±1mA 0.8 1.3 1.4 1.4 V max A VE = 0 ±8 ±7 ±6.5 ±6.5 mA min A Maximum C-Output Voltage Compliance IC = ±1mA 4.7 4.0 3.9 3.9 V min A Minimum C-Output Voltage Compliance IC = ±1mA 0.3 1.0 1.1 1.1 V max A VC = 0 ±8 ±7 ±6.5 ±6.5 mA min A kΩ || pF typ C E-Output Current, Sinking/Sourcing C-Output Current, Sinking/Sourcing C-Output Impedance (1) (2) (3) (4) 54 || 2 Test levels: (A) 100% tested at 25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. Junction temperature = ambient for 25°C specifications. Junction temperature = ambient at low temperature limit; junction temperature = ambient + 3°C at high temperature limit for over temperature specifications. Current is considered positive out of node. 5 OPA861 www.ti.com SBOS338 – AUGUST 2005 ELECTRICAL CHARACTERISTICS: VS = +5V (continued) RL = 500Ω to VS/2 and RADJ = 250Ω, unless otherwise noted. OPA861ID, IDBV TYP MIN/MAX OVER TEMPERATURE +25°C (2) 0°C to 70°C (3) –40°C to +85°C (3) Maximum Operating Voltage 12.6 12.6 12.6 Minimum Operating Voltage 4 4 PARAMETER CONDITIONS +25°C MIN/ MAX TEST LEVEL (1) V typ C V max A 4 V min B UNITS POWER SUPPLY Specified Operating Voltage 5 Maximum Quiescent Current RADJ = 250Ω 4.7 5.2 6.0 6.4 mA max A Minimum Quiescent Current RADJ = 250Ω 4.7 4.2 3.4 3.0 mA min A ∆IC/∆VS ±20 ±50 ±60 ±65 µA/V max A –40 to +85 °C typ C OTA Power-Supply Rejection Ratio (+PSRR) THERMAL CHARACTERISTICS Specification: ID, IDBV Thermal Resistance θJA D SO-8 Junction-to-Ambient 125 °C/W typ C DBV SOT23-6 Junction-to-Ambient 150 °C/W typ C 6 OPA861 www.ti.com SBOS338 – AUGUST 2005 TYPICAL CHARACTERISTICS: VS = ±5V At TA = +25°C, IQ = 5.4mA, and RL = 500Ω, unless otherwise noted. OTA TRANSCONDUCTANCE vs FREQUENCY OTA TRANSCONDUCTANCE vs QUIESCENT CURRENT 1000 150 IO UT VIN = 100mVPP RL = 50Ω VIN = 10mVPP 50Ω Transconductance (mA/V) Transconductance (mA/V) VI N 50Ω I Q = 5.4mA (102mA/V) IQ = 6.5mA (117mA/V) 100 IQ = 1.9mA (51mA/V) 120 90 I OUT 60 VIN 50Ω 30 50Ω IQ = 3.4mA (79mA/V) 10 0 1M 10M 100M 1G 6 140 6 IQ = 6.5mA IQ = 5.4mA 100 IQ = 3.4mA 80 60 IQ = 1.9mA 40 −30 −20 −10 0 IQ = 5.4mA 4 2 IQ = 3.4mA 0 IQ = 1.9mA −2 IOUT VIN −4 50Ω 50Ω 20 30 −70 −60 −50 −40 −30 −20 −10 40 20 30 Figure 3. Figure 4. 40 50 60 70 OTA LARGE-SIGNAL PULSE RESPONSE 3 0.2 0 G = +5V/V RL = 500Ω VIN = 0.25VPP f IN = 20MHz See Figure 48 Time (10ns/div) Figure 5. Output Voltage (V) 2 0.4 −0.8 10 OTA Input Voltage (mV) 0.6 −0.6 0 Input Voltage (mV) OTA SMALL-SIGNAL PULSE RESPONSE −0.4 13 −8 10 0.8 −0.2 12 IQ = 6.5mA −6 Small signal around input voltage. −40 11 OTA TRANSFER CHARACTERISTICS 8 OTA Output Current (mA) Transconductance (mA/V) 10 Figure 2. 20 Output Voltage (V) 9 Figure 1. OTA TRANSCONDUCTANCE vs INPUT VOLTAGE 0 8 Quiescent Current (mA) 160 120 7 Frequency (Hz) 1 0 −1 −2 G = +5V/V RL = 500Ω VIN = 1VPP fIN = 20MHz See Figure 48 −3 Time (10ns/div) Figure 6. 7 OPA861 www.ti.com SBOS338 – AUGUST 2005 TYPICAL CHARACTERISTICS: VS = ±5V (continued) At TA = +25°C, IQ = 5.4mA, and RL = 500Ω, unless otherwise noted. C-OUTPUT RESISTANCE vs QUIESCENT CURRENT 120 490 110 OTA C−Output Resistance (kΩ ) OTA B−Input Resistance (kΩ ) B-INPUT RESISTANCE vs QUIESCENT CURRENT 500 480 470 460 450 440 100 430 80 70 60 50 40 7 8 9 10 11 12 13 7 10 11 Figure 7. Figure 8. 12 13 INPUT VOLTAGE AND CURRENT NOISE DENSITY Input Voltage Noise Density (nV/√Hz) Input Current Noise Density (pA/√Hz) 100 50 40 30 20 10 0 E−Input Current Noise (5.2pA/√Hz) 10 B−Input Voltage Noise (2.4nV/√Hz) B−Input Current Noise (1.65pA/√Hz) 1 7 8 9 10 11 12 100 13 Quiescent Current (mA) 1k 10k 100k 1M 10M Frequency (Hz) Figure 9. Figure 10. QUIESCENT CURRENT vs RADJ 1MHz OTA VOLTAGE AND CURRENT NOISE DENSITY vs QUIESCENT CURRENT ADJUST RESISTOR 16 Input Voltage Noise Density (nV/√Hz) Input Current Noise Density (pA/√Hz) 8 7 Quiescent Current (mA) 9 Quiescent Current (mA) E-OUTPUT RESISTANCE vs QUIESCENT CURRENT 6 5 4 3 2 1 E−Input Current Noise (pA/√Hz) 14 12 10 8 B−Input Voltage Noise (nV/√Hz) 6 B−Input Current Noise (pA/√Hz) 4 2 0 0 0.1 1 10 100 1k 10k Quiescent Current Adjust Resistor (Ω) Figure 11. 8 8 Quiescent Current (mA) 60 OTA E−Output Resistance (Ω) 90 100k 0 200 400 600 800 1000 1200 1400 1600 1800 2000 Quiescent Current Adjust Resistor (Ω ) Figure 12. OPA861 www.ti.com SBOS338 – AUGUST 2005 TYPICAL CHARACTERISTICS: VS = ±5V (continued) At TA = +25°C, IQ = 5.4mA, and RL = 500Ω, unless otherwise noted. QUIESCENT CURRENT vs TEMPERATURE 3 9 4 2 8 1 B−Input Offset Voltage 0 0 −2 −4 −6 −40 −1 B−Input Bias Current −20 0 20 40 60 80 100 Input Bias Current (µA) 2 Quiescent Current (mA) 6 7 6 5 −2 4 −3 3 −40 120 −20 0 Ambient Temperature ( C) 20 40 60 80 100 120 Ambient Temperature ( C) Figure 13. Figure 14. C-OUTPUT BIAS CURRENT vs TEMPERATURE 40 OTA C−Output Bias Current (µA) Offset Voltage (mV) B-INPUT OFFSET VOLTAGE AND BIAS CURRENT vs TEMPERATURE Five Representative Units 30 20 10 0 −10 −20 −30 −40 −40 −20 0 20 40 60 80 100 120 Ambient Temperature ( C) Figure 15. 9 OPA861 www.ti.com SBOS338 – AUGUST 2005 TYPICAL CHARACTERISTICS: VS = +5V At TA = +25°C, IQ = 4.7mA, and RL = 500Ω to VS/2, unless otherwise noted. OTA TRANSCONDUCTANCE vs FREQUENCY 100 OTA TRANSCONDUCTANCE vs IQ 150 OTA Transconductance (mA/V) IQ = 5.8mA (93mA/V) Transconductance (mA/V) IQ = 4.7mA (80mA/V) IQ = 3.1mA (60mA/V) IQ = 1.65mA (37mA/V) I OU T VIN 50Ω 50Ω 90 60 30 VIN = 100mVPP 10 100 1k 0 1 2 3 4 5 Frequency (Hz) Quiescent Current (mA) Figure 16. Figure 17. OTA TRANSCONDUCTANCE vs INPUT VOLTAGE 6 7 OTA TRANSFER CHARACTERISTICS 120 6 IQ = 5.8mA IQ = 5.8mA IQ = 4.7mA 100 80 OTA Output Current (mA) Transconductance (mA/V) 50Ω 50Ω 0 1 IQ = 3.1mA 60 IQ = 1.65mA 40 20 Small−signal around input voltage. 0 −30 −20 −10 0 10 20 4 IQ = 3.1mA 2 IQ = 4.7mA I OUT −2 V IN −4 −6 −50 −40 −30 −20 −10 30 20 Figure 18. Figure 19. 1.0 −0.05 G = +5V/V R L = 500Ω VIN = 0.07VPP f IN = 20MHz Time (10ns/div) Figure 20. Output Voltage (V) 1.5 0.10 0 30 40 OTA LARGE-SIGNAL PULSE RESPONSE 0.15 −0.20 10 OTA Input Voltage (mV) 2.0 −0.15 0 Input Voltage (mV) 0.05 50 Ω 50 Ω 0.20 −0.10 IQ = 1.65mA 0 OTA SMALL-SIGNAL PULSE RESPONSE Output Voltage (V) VIN 120 RL = 50Ω VIN = 10mVPP 10 10 IOUT 0.5 0 −0.5 −1.0 −1.5 −2.0 G = +5V/V R L = 500Ω VIN = 0.7VPP fIN = 20MHz Time (10ns/div) Figure 21. 50 OPA861 www.ti.com SBOS338 – AUGUST 2005 TYPICAL CHARACTERISTICS: VS = +5V (continued) At TA = +25°C, IQ = 4.7mA, and RL = 500Ω to VS/2, unless otherwise noted. C-OUTPUT RESISTANCE vs QUIESCENT CURRENT 120 490 110 OTA C−Output Resistance (kΩ ) OTA B−Input Resistance (kΩ ) B-INPUT RESISTANCE vs QUIESCENT CURRENT 500 480 470 460 450 440 430 100 90 80 70 60 50 40 420 0 1 2 3 4 5 6 0 7 1 3 4 5 Quiescent Current (mA) Figure 22. Figure 23. E-OUTPUT RESISTANCE vs QUIESCENT CURRENT 7 50 6 40 30 20 10 6 7 QUIESCENT CURRENT vs RADJ 60 Quiescent Current (mA) OTA E−Output Resistance (Ω ) 2 Quiescent Current (mA) 5 4 3 2 1 0 0 0 1 2 3 4 Quiescent Current (mA) Figure 24. 5 6 7 0.1 1 10 100 1k 10k 100k Quiescent Current Adjust Resistor (Ω) Figure 25. 11 OPA861 www.ti.com SBOS338 – AUGUST 2005 APPLICATION INFORMATION The OPA861 is a versatile monolithic transconductance amplifier designed for wide-bandwidth systems, including high-performance video, RF, and IF circuitry. The operation of the OPA861 is discussed in the OTA (Operational Transconductance Amplifier) section of this data sheet. Over the years and depending on the writer, the OTA section of an op amp has been referred to as a Diamond Transistor, Voltage-Controlled Current source, Transconductor, Macro Transistor, or positive second-generation current conveyor (CCII+). Corresponding symbols for these terms are shown in Figure 26. C 3 VIN1 B 1 IOUT 2 VIN2 E Diamond Transistor Transconductor (used here) Voltage−Controlled Current Source C VIN1 VIN2 Z CCII+ IOUT B The symbol for the OTA section is similar to a transistor (see Figure 26). Applications circuits for the OTA look and operate much like transistor circuits—the transistor is also a voltage-controlled current source. Not only does this characteristic simplify the understanding of application circuits, it aids the circuit optimization process as well. Many of the same intuitive techniques used with transistor designs apply to OTA circuits. The three terminals of the OTA are labeled B, E, and C. This labeling calls attention to its similarity to a transistor, yet draws distinction for clarity. While the OTA is similar to a transistor, one essential difference is the sense of the C-output current: it flows out the C terminal for positive B-to-E input voltage and in the C terminal for negative B-to-E input voltage. The OTA offers many advantages over a discrete transistor. The OTA is self-biased, simplifying the design process and reducing component count. In addition, the OTA is far more linear than a transistor. Transconductance of the OTA is constant over a wide range of collector currents—this feature implies a fundamental improvement of linearity. BASIC CONNECTIONS E Current Conveyor II+ TRANSCONDUCTANCE (OTA) SECTION—AN OVERVIEW Macro Transistor Figure 26. Symbols and Terms Regardless of its depiction, the OTA section has a high-input impedance (B-input), a low-input/output impedance (E-input), and a high-impedance current source output (C-output). Figure 27 shows basic connections required for operation. These connections are not shown in subsequent circuit diagrams. Power-supply bypass capacitors should be located as close as possible to the device pins. Solid tantalum capacitors are generally best. RQ = 250Ω, roughly sets I Q = 5.4mA. 1 RS (25Ωto 200Ω) RADJ 250Ω 2 7 +5V(1) 0.1µF +VS + VIN −5V(1) RC 8 −VS 3 6 4 5 2.2µF Solid Tantalum 0.1µF + 2.2µF Solid Tantalum NOTE: (1) VS = ±6.5V absolute maximum. Figure 27. Basic Connections 12 OPA861 www.ti.com SBOS338 – AUGUST 2005 QUIESCENT CURRENT CONTROL PIN The quiescent current of the transconductance portion of the OPA861 is set with a resistor, RADJ, connected from pin 1 to –VS. The maximum quiescent current is 6mA. RADJ should be set between 50Ω and 1kΩ for optimal performance of the OTA section. This range corresponds to the 5mA quiescent current for RADJ = 50Ω, and 1mA for RADJ = 1kΩ. If the IQ adjust pin is connected to the negative supply, the quiescent current will be set by the 250Ω internal resistor. Reducing or increasing the quiescent current for the OTA section controls the bandwidth and AC behavior as well as the transconductance. With RADJ = 250Ω, this sets approximately 5.4mA total quiescent current at 25°C. It may be appropriate in some applications to trim this resistor to achieve the desired quiescent current or AC performance. Applications circuits generally do not show the resistor RQ, but it is required for proper operation. With a fixed RADJ resistor, quiescent current increases with temperature (see Figure 11 in the Typical Characteristics section). This variation of current with temperature holds the transconductance, gm, of the OTA relatively constant with temperature (another advantage over a transistor). It is also possible to vary the quiescent current with a control signal. The control loop in Figure 28 shows 1/2 of a REF200 current source used to develop 100mV on R1. The loop forces 125mV to appear on R2. Total quiescent current of the OPA861 is approximately 37 × I1, where I1 is the current made to flow out of pin 1. V+ OPA861 1/2 REF200 100µA R1 1.25kΩ With this control loop, quiescent current will be nearly constant with temperature. Since this method differs from the temperature-dependent behavior of the internal current source, other temperature-dependent behavior may differ from that shown in the Typical Characteristics. The circuit of Figure 28 will control the IQ of the OPA861 somewhat more accurately than with a fixed external resistor, RQ. Otherwise, there is no fundamental advantage to using this more complex biasing circuitry. It does, however, demonstrate the possibility of signal-controlled quiescent current. This capability may suggest other possibilities such as AGC, dynamic control of AC behavior, or VCO. BASIC APPLICATIONS CIRCUITS Most applications circuits for the OTA section consist of a few basic types, which are best understood by analogy to a transistor. Used in voltage-mode, the OTA section can operate in three basic operating states—common emitter, common base, and common collector. In the current-mode, the OTA can be useful for analog computation such as current amplifier, current differentiator, current integrator, and current summer. Common-E Amplifier or Forward Amplifier Figure 29 compares the common-emitter configuration for a BJT with the common-E amplifier for the OTA section. There are several advantages in using the OTA section in place of a BJT in this configuration. Notably, the OTA does not require any biasing, and the transconductance gain remains constant over temperature. The output offset voltage is close to 0, compared with several volts for the common-emitter amplifier. The gain is set in a similar manner as for the BJT equivalent with Equation 1: R G 1 L RE gm (1) IQ Adjust 1 I1 R2 425Ω TLV2262 Just as transistor circuits often use emitter degeneration, OTA circuits may also use degeneration. This option can be used to reduce the effects that offset voltage and offset current might otherwise have on the DC operating point of the OTA. The E-degeneration resistor may be bypassed with a large capacitor to maintain high AC gain. Other circumstances may suggest a smaller value capacitor used to extend or optimize high-frequency performance. Figure 28. Optional Control Loop for Setting Quiescent Current 13 OPA861 www.ti.com SBOS338 – AUGUST 2005 The forward amplifier shown in Figure 30 and Figure 31 corresponds to one of the basic circuits used to characterize the OPA861. Extended characterization of this topology appears in the Typical Characteristics section of this datasheet. V+ RS RL VO VO VI Inverting Gain VOS = Several Volts RS R1 160Ω RE VI 8 C 3 B OPA861 E 2 V− VI 8 C 3 B VO OPA861 Figure 30. Forward Amplifier Configuration and Test Circuit RL E 2 RE G = 5V/V IQ = 5.4mA RE 78Ω (a) Transistor Common−Emitter Amplifier Transconductance varies over temperature. 100Ω RC 500Ω RL1 Noninverting Gain VOS = 0V VO Network Analyzer 8 3 (b) OTA Common−E Amplifier Transconductance remains constant over temperature. Figure 29. Common-Emitter vs Common-E Amplifier (2) A positive voltage at the B-input, pin 3, causes a positive current to flow out of the C-input, pin 8. This gives a noninverting gain where the circuit of Figure 29a is inverting. Figure 29b shows an amplifier connection of the OPA861, the equivalent of a common-emitter transistor amplifier. Input and output can be ground-referenced without any biasing. The amplifier is non-inverting because of the sense of the output current. 14 RL2 rE 2 VI The transconductance of the OTA with degeneration can be calculated by Equation 2: g m_deg 1 1 gm R E RIN 50Ω OPA861 R1 100Ω RL = RL1 + RL2 || RIN RE G RL r E g1 m RE rE At I Q 5.4mA G rE 1 10.5 95mAV RL at I Q 5.4mA R E 10.5 Figure 31. Forward Amplifier Design Equations OPA861 www.ti.com SBOS338 – AUGUST 2005 Common-C Amplifier Current-Mode Analog Computations Figure 32b shows the OPA861 connected as an E-follower—a voltage buffer. It is interesting to notice that the larger the RE resistor, the closer to unity gain the buffer will be. If the OPA861 is to be used as a buffer, use RE ≥ 500Ω for best results. For the OPA861 used as a buffer, the gain is given by Equation 3: 1 G 1 1 1 g R m As mentioned earlier, the OPA861 can be used advantageously for analog computation. Among the application possibilities are functionality as a current amplifier, current differentiator, current integrator, current summer, and weighted current summer. Table 1 lists these different uses with the associated transfer functions. (3) E These functions can easily be combined to form active filters. Some examples using these current-mode functions are shown later in this document. V+ V+ G=1 VOS = 0.7V VO VI RL VO RE Noninverting Gain VOS = Several Volts V− (a) Transistor Common−Collector Amplifier (Emitter Follower) G 100Ω VI RE OPA861 1g 1 1 V− mR E R O g1 R E m 8 C 3 B RE 1 (a) Transistor Common−Base Amplifier G G=1 VOS = 0V E 2 VO 100Ω (b) OTA Common−C Amplifier (Buffer) RL R L 1 R R E gm E 8 C 3 B OPA861 E 2 VO Inverting Gain VOS = 0V RL RE Figure 32. Common-Collector vs Common-C Amplifier V− A low value resistor in series with the B-input is recommended. This resistor helps isolate trace parasitic from the inputs, reduces any tendency to oscillate, and controls frequency response peaking. Typical resistor values are from 25Ω to 200Ω. (b) OTA Common−B Amplifier Figure 33. Common-Base Transistor vs Common-B OTA Common-B Amplifier Figure 33 shows the Common-B amplifier. This configuration produces an inverting gain and a low impedance input. Equation 4 shows the gain for this configuration. RL R G L 1 R R E gm E (4) This low impedance can be converted to a high impedance by inserting the buffer amplifier in series. 15 OPA861 www.ti.com SBOS338 – AUGUST 2005 Table 1. Current-Mode Analog Computation Using the OTA Section FUNCTIONAL ELEMENT TRANSFER FUNCTION Current Amplifier R I OUT 1 I IN R2 IMPLEMENTATION WITH THE OTA SECTION IOUT IIN R1 R2 IOUT 1 I OUT Current Integrator CR IIN I dt C IN R IOUT n I OUT 1 I j j1 Current Summer I1 I2 In I OUT Rj I OUT 1 I j j1 R n Weighted Current Summer R R1 I1 R Rn In OPA861 APPLICATIONS Control-Loop Amplifier DC-Restore Circuit A new type of control loop amplifier for fast and precise control circuits can be designed with the OPA861. The circuit of Figure 34 illustrates a series connection of two voltage control current sources that have an integral (and at higher frequencies, a proportional) behavior versus frequency. The control loop amplifiers show an integrator behavior from DC to the frequency represented by the RC time constant of the network from the C-output to GND. Above this frequency, they operate as an amp with constant gain. The series connection increases the overall gain to about 110dB and thus minimizes the control loop deviation. The differential configuration at the inputs enables one to apply the measured output signal and the reference voltage to two identical high-impedance inputs. The output buffer decouples the C-output of the second OTA in order to insure the AC performance and to drive subsequent output stages. The OPA861 can be used advantageously with an operational amplifier, here the OPA656, as a DC-restore circuit. Figure 35 illustrates this design. Depending on the collector current of the transconductance amplifier (OTA) of the OPA861, a switching function is realized with the diodes D1 and D2. 16 When the C-output is sourcing current, the capacitor C1 is being charged. When the C-output is sinking current, D1 is turned off and D2 is turned on, letting the voltage across C1 be discharged through R2. The condition to charge C1 is set by the voltage difference between VREF and VOUT. For the OTA C-output to source current, VREF has to be greater than VOUT. The rate of charge of C1 is set by both R1 and C1. The discharge rate is given by R2 and C1. OPA861 www.ti.com SBOS338 – AUGUST 2005 8 6 5 BUF602 VOUT 3 8 180Ω 2 10pF VREF 10pF 3 2 10Ω 180Ω 33Ω 10Ω 33Ω 6 VIN Figure 34. Control-Loop Amplifier Using Three OPA861s C1 100pF 20Ω JFET−Input, Wideband VIN D 1, D2 = 1N4148 RQ = 1kΩ OPA656 R2 100kΩ D1 VOUT 20Ω D2 CCII 8 C The OTA amplifier works as a current conveyor (CCII) in this circuit, with a current gain of 1. R1 and C1 set the DC restoration time constant. D1 adds a propagation delay to the DC restoration. R2 and C1 set the decay time constant. E 2 R1 40.2Ω B 3 R2 100Ω VREF Figure 35. DC Restorer Circuit Negative Impedance Converter Filter: Low-Pass Filter The OPA861 can be used as a negative impedance converter to realize the low-pass filer shown in Figure 36. R VIN VOUT C2 (5) with: 0 R C1 The transfer function is shown in Equation 5: V OUT 1 V IN 1 sC 2R s 2C1C 2R 2 Q 1 C1C 2 R C1 C2 Figure 36. Low-Pass Negative Impedance Converter Filter 17 OPA861 www.ti.com SBOS338 – AUGUST 2005 Differential Line Driver/Receiver The input impedance is shown in Equation 6: Z IN 1 R 1 sRC 2sC 1 2sRC (6) Figure 37 shows the frequency responses for low-pass, Butterworth filters set at 20kHz and 10MHz. For the 20kHz filter, set R to 1KΩ and C 1 1 C 2 5.6F 2 . For the 10MHz filter, the parasitic capacitance at the output pin needs to be taken into consideration. In the example of Figure 37, the parasitic is 3pF, which gives us the settings of R = 1.13kΩ, C1 = 10pF, and C2 = 17pF. The wide bandwidth and high slew rate of the OPA861 current-mode amplifier make it an ideal line driver. The circuit in Figure 39 makes use of two OPA861s to realize a single-ended to differential conversion. The high-impedance current source output of the OPA861 allows it to drive low-impedance or capacitive loads without series resistances and avoids any attenuation that would have otherwise occured in the resistive network. The OPA861 used as a differential receiver exhibits excellent common-mode rejection ratio, as can be seen in Figure 38. Common−Mode Rejection Ratio (dBc) 0 −10 Gain (dB) −20 −30 −40 −50 −60 −70 −80 1k 10k 100k 1M 10M 100M 1G 0 −10 −20 −30 −40 −50 −60 −70 −80 −90 −100 0.001 0.01 Frequency (Hz) 0.1 1 Figure 37. Small-Signal Frequency Response for a Low-Pass Negative Impedance Converter Filter 50Ω VIN 10Ω 50Ω 100Ω 50Ω 10Ω 50Ω Figure 39. Twisted-Pair Differential Driver and Receiver with the OPA861 18 100 Figure 38. Differential Driver Common-Mode Rejection Ratio for 2VPP Input Signals To 50Ω Load 50Ω 10 Frequency (MHz) OPA861 www.ti.com SBOS338 – AUGUST 2005 ACTIVE FILTERS USING THE OPA861 IN CURRENT CONVEYOR STRUCTURE One further example of the versatility of the Diamond Transistor and Buffer is the construction of high-frequency (> 10MHz) active filters. Here, the Current Conveyor structure, shown in Figure 40, is used with the Diamond Transistor as a Current Conveyor. The method of converting RC circuit loops with operational amplifiers in Current Conveyor structures is based upon the adjoint network concept. A network is reversible or reciprocal when the transfer function does not change even when the input and output have been exchanged. Most networks, of course, are nonreciprocal. The networks of Figure 41, perform interreciprocally when the input and output are exchanged, while the original network, N, is exchanged for a new network NA. In this case, the transfer function remains the same, and NA is the adjoing network. It is easy to construct an adjoint network for any given circuit, and these networks are the base for circuits in Current-Conveyor structure. Individual elements can be interchanged according to the list in Figure 42. Voltage sources at the input become short circuits, and the current flowing there becomes the output variable. In contrast, the voltage output becomes the input, which is excitated by a current source. The following equation describes the interreciprocal features of the circuit: VOUT/VIN = IOUT/IIN. Resistances and capacitances remain unchanged. In the final step, the operational amplifier with infinite input impedance and 0Ω output impedance is transformed into a current amplifier with 0Ω input impedance and infinite output impedance. A Diamond Transistor with the base at ground comes quite close to an ideal current amplifier. The well-known Sallen-Key low-pass filter with positive feedback, is an example of conversion into Current-Conveyor structure, see Figure 43. The positive gain of the operational amplifier becomes a negative second type of Current Conveyor (CCII), as shown in Figure 40. Both arrangements have identical transfer functions and the same level of sensitivity to deviations. The most recent implementation of active filters in a Current-Conveyor structure produced a second-order Bi-Quad filter. The value of the resistance in the emitter of the Diamond Transistor controls the filter characteristic. For more information, refer to application note SBOS047, New Ultra High-Speed Circuit Techniques with Analog ICs. VOUT +1 C E CCII− B C VIN IOUT C R R R R IIN C/2 T(s) = C/2 VOUT VIN = IOUT IIN = 4KQ2/R2C2 s2 + 2/RC[2Q(1 − K) + 1]s + 4KQ2/R2C2 Figure 40. Current Conveyor 19 OPA861 www.ti.com SBOS338 – AUGUST 2005 Reciprocal Networks + VIN VOUT N I OUT N I IN NA I IN − VOUT = I OUT VIN IIN Interreciprocal Networks + VIN VOUT N I OUT − Figure 41. Networks Element VIN 1 Signal Sources 1 − VOUT + R 1 Passive Elements C 1 1 Controlled Sources 1 Adjoint + V − 2 IOUT 1 2 1 2 1 2 1 2 IIN R C 3 µV 2 2 3 µI 4 2 I 4 Figure 42. Individual Elements in the Current Conveyor 20 OPA861 www.ti.com SBOS338 – AUGUST 2005 R3 R2 VIN BUF602 C1 R1 RB1 RB2 R1S C2 R1M R2M RB3 R2S VOUT R3S Figure 43. Universal Active Filter Transfer Function The transfer function of the universal active filter of Figure 43 is shown in Equation 7. R 1 1M R 3S 1R 2S R (7) 1S Filter Characteristics Five filter types can be made with this structure: • For a low-pass filter, set R2 = R3 = ∞, • For a high-pass filter, set R1 = R2 = ∞, • For a bandpass filter, set R1 = R3 = ∞, • For a band rejection filter, set R2 = ∞; R1 = R3, • For an all-pass filter, set R1 = R15; R2 = R25; and R3 = R35. A few designs for a low-pass filter are shown in Figure 44 and Table 2. Table 2. Component Values for Filters Shown In Figure 44 fO R RO CO 1MHz 150 100 2nF 20MHz 150 100 112.5pF 50MHz 150 100 55pF Gain (dB) R s 2C1R 1M R2M sC 1 R1M R1 V OUT 3 2 1 F(p) R2M R 1M 1 VIN 2 sCR sC 3 0 50MHz Filter −3 −6 −9 −12 −15 1MHz Filter −18 20MHz Filter −21 −24 −27 −30 For All Filters: −33 R = R = ∞ 2 3 −36 −39 R1 = R15 = R25 = 1/2 R35 = R −42 R1M = R2M = R0 −45 C1 = C2 = C0 −48 10k 100k 1M 10M 100M 1G Frequency (Hz) Figure 44. Butterworth Low-Pass Filter with the Universal Active Filter The advantages of building active filters using a Current Conveyor structure are: • The increase in output resistance of operational amplifiers at high frequencies makes it difficult to construct feedback filter structures (decrease in stop-band attenuation). • All filter coefficients are represented by resistances, making it possible to adjust the filter frequency response without affecting the filter coefficients. 21 OPA861 www.ti.com SBOS338 – AUGUST 2005 • The capacitors which determine the frequency are located between the ground and the current source outputs and are thus grounded on one side. Therefore, all parasitic capacitances can be viewed as part of these capacitors, making them easier to comprehend. The features which determine the frequency characteristics are currents, which charge the integration capacitors. This situation is similar to the transfer characteristic of the Diamond Transistor. 6 5.6dB 3 Gain (dB) • OPA861 VIN1 600Ω RE 600Ω VCM −9 1M 1G 75 Input−Referred 70 65 60 55 50 45 40 35 30 25 20 10k 100k 1M 10M 100M Frequency (Hz) 600Ω Figure 45. High CMRR, Moderate Precision, Differential I/O ADC Driver 22 100M Figure 46. ADC Driver, Small-Signal Frequency Response 1k OPA861 VIN2 10M Frequency (Hz) Common−Mode Rejection Ratio(dB) ADS5272 −3 −6 High-CMRR, Moderate Precision, Differential I/O ADC Driver The circuit shown in Figure 45 depicts an ADC driver implemented with two OPA861s. Since the gain is set here by the ratio of the internal 600Ω resistors and RE, its accuracy will only be as good as the input resistor of the ADS5272. The small-signal frequency response for this circuit has 150MHz at –3dB bandwidth for a gain of approximately 5.6dB, as shown in Figure 46. The advantage of this circuit lies in its high CMRR to 100kHz, as shown in Figure 47. This circuit also has more than 10 bits of linearity. 0 Figure 47. CMRR of the ADC Driver 1G OPA861 www.ti.com SBOS338 – AUGUST 2005 DESIGN-IN TOOLS DEMONSTRATION BOARDS A printed circuit board (PCB) is available to assist in the initial evaluation of circuit performance using the OPA861. This module is available free, as an unpopulated PCB delivered with descriptive documentation. The summary information for the board is shown below: PRODUCT PACKAGE BOARD PART NUMBER OPA861ID SO-8 DEM-OPA86xD LITERATURE REQUEST NUMBER SBOU035 The board can be requested on Texas Instruments web site (www.ti.com). MACROMODELS AND APPLICATIONS SUPPORT Computer simulation of circuit performance using SPICE is often useful when analyzing the performance of analog circuits and systems. This principle is particularly true for Video and RF amplifier circuits where parasitic capacitance and inductance can have a major effect on circuit performance. A SPICE model for the OPA861 is available through the Texas Instruments web page (www.ti.com). These models do a good job of predicting small-signal AC and transient performance under a wide variety of operating conditions. They do not do as well in predicting the harmonic distortion. These models do not attempt to distinguish between the package types in their small-signal AC performance. NOISE PERFORMANCE The OTA noise model consists of three elements: a voltage noise on the B-input; a current noise on the B-input; and a current noise on the E-input. Figure 48 shows the OTA noise analysis model with all the noise terms included. In this model, all noise terms are taken to be noise voltage or current density terms in either nV/√Hz or pA/√Hz. en VO RL RS √4kTRS ibn RG The total output spot noise voltage can be computed as the square root of the sum of all squared output noise voltage contributors. Equation 8 shows the general form for the output noise voltage using the terms shown in Figure 48. eO 2 e RSi bn 4kTR S 2 n RL RG g1m 2 2 R R Gibi 4kTR G 1L gm (8) THERMAL ANALYSIS Maximum desired junction temperature will set the maximum allowed internal power dissipation as described below. In no case should the maximum junction temperature be allowed to exceed 150°C. Operating junction temperature (TJ) is given by TA + PD × θJA. The total internal power dissipation (PD) is the sum of quiescent power (PDQ) and additional power dissipated in the output stage (PDL) to deliver output current. Quiescent power is simply the specified no-load supply current times the total supply voltage across the part. PDL will depend on the required output signal and load but would, for the OPA861 be at a maximum when the maximum IO is being driven into a voltage source that puts the maximum voltage across the output stage. Maximum IO is 15mA times a 9V maximum across the output. Note that it is the power in the output stage and not into the load that determines internal power dissipation. As a worst-case example, compute the maximum TJ using an OPA861IDBV in the circuit of Figure 29b operating at the maximum specified ambient temperature of +85°C and driving a –1V voltage reference. PD = 10V × 5.4mA + (15mA × 9V) = 185mW Maximum TJ = +85°C + (0.19W × 150°C/W) = 114°C. Although this is still well below the specified maximum junction temperature, system reliability considerations may require lower tested junction temperatures. The highest possible internal dissipation will occur if the load requires current to be forced into the output for positive output voltages or sourced from the output for negative output voltages. This puts a high current through a large internal voltage drop in the output transistors. ibi √4kTRS Figure 48. OTA Noise Analysis Model BOARD LAYOUT GUIDELINES Achieving optimum performance with a high-frequency amplifier like the OPA861 requires careful attention to board layout parasitics and external component types. Recommendations that will optimize performance include: 23 OPA861 www.ti.com SBOS338 – AUGUST 2005 a) Minimize parasitic capacitance to any AC ground for all of the signal I/O pins. Parasitic capacitance on the inverting input pin can cause instability: on the noninverting input, it can react with the source impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the signal I/O pins should be opened in all of the ground and power planes around those pins. Otherwise, ground and power planes should be unbroken elsewhere on the board. b) Minimize the distance (< 0.25") from the power-supply pins to high-frequency 0.1µF decoupling capacitors. At the device pins, the ground and power-plane layout should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. The power-supply connections should always be decoupled with these capacitors. An optional supply decoupling capacitor (0.1µF) across the two power supplies (for bipolar operation) will improve 2nd-harmonic distortion performance. Larger (2.2µF to 6.8µF) decoupling capacitors, effective at lower frequency, should also be used on the main supply pins. These may be placed somewhat farther from the device and may be shared among several devices in the same area of the PC board. c) Careful selection and placement of external components will preserve the high-frequency performance of the OPA861. Resistors should be a very low reactance type. Surface-mount resistors work best and allow a tighter overall layout. Metal film or carbon composition, axially-leaded resistors can also provide good high-frequency performance. Again, keep their leads and PC board traces as short as possible. Never use wirewound type resistors in a high-frequency application. d) Connections to other wideband devices on the board may be made with short, direct traces or through onboard transmission lines. For short con- 24 nections, consider the trace and the input to the next device as a lumped capacitive load. Relatively wide traces (50mils to 100mils) should be used, preferably with ground and power planes opened up around them. e) Socketing a high-speed part like the OPA861 is not recommended. The additional lead length and pin-to-pin capacitance introduced by the socket can create an extremely troublesome parasitic network that makes it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering the OPA861 onto the board. INPUT AND ESD PROTECTION The OPA861 is built using a very high-speed complementary bipolar process. The internal junction breakdown voltages are relatively low for these very small geometry devices. These breakdowns are reflected in the Absolute Maximum Ratings table. All device pins are protected with internal ESD protection diodes to the power supplies as shown in Figure 49. +VCC External Pin Internal Circuitry −VCC Figure 49. Internal ESD Protection These diodes provide moderate protection to input overdrive voltages above the supplies as well. The protection diodes can typically support 30mA continuous current. Where higher currents are possible (for example, in systems with ±15V supply parts driving into the OPA861), current-limiting series resistors should be added into the two inputs. Keep these resistor values as low as possible since high values degrade both noise performance and frequency response. PACKAGE OPTION ADDENDUM www.ti.com 27-Sep-2005 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Eco Plan (2) Qty Lead/Ball Finish MSL Peak Temp (3) OPA861ID ACTIVE SOIC D 8 75 TBD Call TI Call TI OPA861IDBVR ACTIVE SOT-23 DBV 6 3000 TBD Call TI Call TI OPA861IDBVT ACTIVE SOT-23 DBV 6 250 TBD Call TI Call TI OPA861IDR ACTIVE SOIC D 8 2500 TBD Call TI Call TI (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS) or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. 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