LT1611 Inverting 1.4MHz Switching Regulator in SOT-23 U DESCRIPTIO FEATURES The LT®1611 is the industry’s first inverting 5-lead SOT-23 current mode DC/DC converter. Intended for use in small, low power applications, it operates from an input voltage as low as 1.1V and switches at 1.4MHz, allowing the use of tiny, low cost capacitors and inductors 2mm or less in height. Its small size and high switching frequency enable the complete DC/DC converter function to consume less than 0.25 square inches of PC board area. Capable of generating – 5V at 150mA from a 5V supply or – 5V at 100mA from a 3V supply, the LT1611 replaces nonregulated “charge pump” solutions in many applications. Very Low Noise: 1mVP–P Output Ripple – 5V at 150mA from a 5V Input Better Regulation Than a Charge Pump Effective Output Impedance: 0.14Ω Uses Tiny Capacitors and Inductors Internally Compensated Fixed Frequency 1.4MHz Operation Low Shutdown Current: <1µA Low VCESAT Switch: 300mV at 300mA Tiny 5-Lead SOT-23 Package ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ U APPLICATIO S The LT1611 operates in a dual inductor inverting topology which filters the input side as well as the output side of the DC/DC converter. Fixed frequency switching ensures a clean output free from low frequency noise typically present with charge pump solutions. No load quiescent current of the LT1611 is 3mA, while in shutdown quiescent current drops to 0.5µA. The 36V switch allows VIN to VOUT differential of up to 33V. MR Head Bias Digital Camera CCD Bias LCD Bias GaAs FET Bias Positive-to-Negative Conversion ■ ■ ■ ■ ■ The LT1611 is available in the 5-lead SOT-23 package. , LTC and LT are registered trademarks of Linear Technology Corporation. U TYPICAL APPLICATIO C2 1µF L1A 22µH VIN 5V D1 VIN + SW SHDN C1 22µF R1 29.4k LT1611 1200pF NFB GND Transient Response L1B 22µH R2 10k VOUT –5V 150mA C3 22µF VOUT 20mV/DIV AC COUPLED LOAD CURRENT C1: AVX TAJB226M010 C2: TAIYO YUDEN LMK212BJ105MG C3: TAIYO YUDEN JMK325BJ226MM (1210 SIZE) D1: MBR0520 L1: SUMIDA CLS62-220 OR 2× MURATA LQH3C220 (UNCOUPLED) 1611 TA01 150mA 50mA 100µs/DIV 1611 F10 Figure 1. 5V to – 5V, 150mA Low Noise Inverting DC/DC Converter 1 LT1611 W U PACKAGE/ORDER INFORMATION U W W W (Note 1) VIN Voltage .............................................................. 10V SW Voltage ................................................– 0.4V to 36V NFB Voltage ............................................................. – 3V Current into NFB Pin ............................................. ±1mA SHDN Voltage .......................................................... 10V Maximum Junction Temperature .......................... 125°C Operating Temperature Range Commercial ............................................. 0°C to 70°C Extended Commercial (Note 2) ........... – 40°C to 85°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C U ABSOLUTE MAXIMUM RATINGS ORDER PART NUMBER TOP VIEW SW 1 5 VIN LT1611CS5 GND 2 4 SHDN NFB 3 S5 PACKAGE 5-LEAD PLASTIC SOT-23 S5 PART MARKING LTES TJMAX = 125°C, θJA = 256°C/W Consult factory for Industrial and Military grade parts. ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 1.5V, VSHDN = VIN unless otherwise noted. PARAMETER CONDITIONS MIN Minimum Operating Voltage TYP MAX 0.9 1.1 Maximum Operating Voltage NFB Pin Bias Current VNFB = –1.23V Feedback Voltage Quiescent Current VSHDN = 1.5V, Not Switching Quiescent Current in Shutdown VSHDN = 0V, VIN = 2V VSHDN = 0V, VIN = 5V Reference Line Regulation 1.5V ≤ VIN ≤ 10V UNITS V 10 V ● – 2.7 – 4.7 – 6.7 µA ● – 1.205 – 1.23 – 1.255 V 3 4.5 mA 0.01 0.01 0.5 1.0 µA µA 0.02 0.2 %/V Switching Frequency ● 1.0 1.4 1.8 MHz Maximum Duty Cycle ● 82 86 550 800 % Switch Current Limit (Note 3) Switch VCESAT ISW = 300mA 300 350 mV Switch Leakage Current VSW = 5V 0.01 1 µA SHDN Input Voltage High 1 V SHDN Input Voltage Low SHDN Pin Bias Current VSHDN = 3V VSHDN = 0V Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: C grade device specifications are guaranteed over the 0°C to 70°C temperature range. In addition, C grade device specifications are assured over the – 40°C to 85°C temperature range by design or correlation, but are not production tested. 2 mA 25 0 0.3 V 50 0.1 µA µA Note 3: Current limit guaranteed by design and/or correlation to static test. Slope compensation reduces current limit at higher duty cycle. LT1611 U W TYPICAL PERFOR A CE CHARACTERISTICS Efficiency, VOUT = – 5V 80 –1.245 6 –1.240 5 NFB PIN BIAS CURRENT (µA) 85 VIN = 5V 75 –1.235 70 VIN = 3V VNFB (V) EFFICIENCY (%) NFB Pin Bias Current vs Temperature VNFB vs Temperature 65 –1.230 –1.225 60 –1.220 55 –1.215 50 0 25 50 75 100 LOAD CURRENT (mA) 125 150 –1.210 –50 –25 0 25 50 TEMPERATURE (°C) 75 Switch VCESAT vs Switch Current SHDN Pin Bias Current vs VSHDN 200 100 SWITCH CURRENT LIMIT (mA) 300 0 25 50 TEMPERATURE (°C) 75 100 Switch Current Limit vs Duty Cycle TA = 25°C 800 400 –25 900 TA = 25°C SHDN PIN BIAS CURRENT (µA) VCESAT (mV) 1 1611 G03 50 500 40 30 20 10 700 600 500 400 300 200 100 0 0 0 100 200 300 400 500 SWITCH CURRENT (mA) 600 700 0 0 1 2 3 4 SHDN PIN VOLTAGE (V) 5 1611 G04 VIN = 5V 0.75 0.50 0.25 0 –50 5.0 4.5 4.0 3.5 3.0 0 25 50 TEMPERATURE (°C) 75 100 2.0 –50 1611 G07 700 600 500 400 300 200 100 2.5 –25 80 800 SWITCH CURRENT LIMIT (mA) OPERATING CURRENT (mA) 1.00 70 900 5.5 VIN = 1.5V 1.25 30 40 50 60 DUTY CYCLE (%) Switch Current Limit vs Temperature (Duty Cycle = 30%) 6.0 1.50 20 1611 G06 No-Load Operating Quiescent Current vs Temperature* 2.00 1.75 10 1611 G05 Oscillator Frequency vs Temperature SWITCHING FREQUENCY (MHz) 2 1611 G02 700 600 3 0 –50 100 1611 G01 4 –25 0 25 50 TEMPERATURE (°C) 75 100 1611 G08 0 –50 –25 0 25 50 TEMPERATURE (°C) 75 100 1611 G09 * Includes bias current through R1, R2 and Schottky leakage current at T ≥ 75°C 3 LT1611 U U U PIN FUNCTIONS SW (Pin 1): Switch Pin. Minimize trace area at this pin to keep EMI down. GND (Pin 2): Ground. Tie directly to local ground plane. NFB (Pin 3): Negative Feedback Pin. Minimize trace area. Reference voltage is –1.23V. Connect resistive divider tap here. The suggested value for R2 is 10k. Set R1 and R2 according to: VOUT − 1.23 R1 = 1.23 + 4.5 • 10− 6 R2 SHDN (Pin 4): Shutdown Pin. Tie to 1V or more to enable device. Ground to shut the device down. VIN (Pin 5): Input Supply Pin. Must be locally bypassed. W BLOCK DIAGRAM VIN 5 VIN R5 40k R6 40k 1 SW + – Q1 VOUT CPL (OPTIONAL) Q2 x10 – A1 gm RC Σ RAMP GENERATOR + COMPARATOR A2 NFB R2 (EXTERNAL) FF S DRIVER Q3 Q + CC R3 30k R1 (EXTERNAL) R 0.15Ω A=3 – 1.4MHz OSCILLATOR R4 140k SHDN 3 NFB 4 SHUTDOWN 2 GND 1611 BD Figure 2 U OPERATIO The LT1611 combines a current mode, fixed frequency PWM architecture with a –1.23V reference to directly regulate negative outputs. Operation can be best understood by referring to the block diagram of Figure 2. Q1 and Q2 form a bandgap reference core whose loop is closed around the output of the converter. The driven reference point is the lower end of resistor R4, which normally sits at a voltage of –1.23V. As the load current changes, the NFB pin voltage also changes slightly, driving the output of gm amplifier A1. Switch current is regulated directly on a cycle-to-cycle basis by A1’s output. The flip-flop is set at the beginning of each cycle, turning on the switch. When the summation of a signal representing switch current and a ramp generator (introduced to avoid subharmonic oscillations at duty factors greater than 50%) exceeds the VC signal, comparator A2 changes stage, resetting the flip- 4 flop and turning off the switch. Output voltage decreases (the magnitude increases) as switch current is increased. The output, attenuated by external resistor divider R1 and R2, appears at the NFB pin, closing the overall loop. Frequency compensation is provided internally by RC and CC. Transient response can be optimized by the addition of a phase lead capacitor, CPL, in parallel with R1 in applications where large value or low ESR output capacitors are used. As load current is decreased, the switch turns on for a shorter period each cycle. If the load current is further decreased, the converter will skip cycles to maintain output voltage regulation. The LT1611 can work in either of two topologies. The simpler topology appends a capacitive level shift to a LT1611 U OPERATIO boost converter, generating a negative output voltage, which is directly regulated. The circuit schematic is detailed in Figure 3. Only one inductor is required, and the two diodes can be in a single SOT-23 package. Output noise is the same as in a boost converter, because current is delivered to the output only during the time when the LT1611’s internal switch is off. When Q1 turns off during the second phase of switching, the SW node voltage abruptly increases to (VIN + |VOUT|). The SWX node voltage increases to VD (about 350mV). Now current in the first loop, begining at C1, flows through L1, C2, D1 and back to C1. Current in the second loop flows from C3 through L2, D1 and back to C3. Load current continues to be supplied by L2 and C3. If D2 is replaced by an inductor, as shown in Figure 4, a higher performance solution results. This converter topology was developed by Professor S. Cuk of the California Institute of Technology in the 1970s. A low ripple voltage results with this topology due to inductor L2 in series with the output. Abrupt changes in output capacitor current are eliminated because the output inductor delivers current to the output during both the off-time and the on-time of the LT1611 switch. With proper layout and high quality output capacitors, output ripple can be as low as 1mVP–P. An important layout issue arises due to the chopped nature of the currents flowing in Q1 and D1. If they are both tied directly to the ground plane before being combined, switching noise will be introduced into the ground plane. It is almost impossible to get rid of this noise, once present in the ground plane. The solution is to tie D1’s cathode to the ground pin of the LT1611 before the combined currents are dumped into the ground plane as drawn in Figures 4, 5 and 6. This single layout technique can virtually eliminate high frequency “spike” noise so often present on switching regulator outputs. The operation of Cuk’s topology is shown in Figures 5 and␣ 6. During the first switching phase, the LT1611’s switch, represented by Q1, is on. There are two current loops in operation. The first loop begins at input capacitor C1, flows through L1, Q1 and back to C1. The second loop flows from output capacitor C3, through L2, C2, Q1 and back to C3. The output current from RLOAD is supplied by L2 and C3. The voltage at node SW is VCESAT and at node SWX the voltage is –(VIN + |VOUT|). Q1 must conduct both L1 and L2 current. C2 functions as a voltage level shifter, with an approximately constant voltage of (VIN + |VOUT|) across it. C2 1µF L1 D2 C2 1µF L1 L2 VIN D1 + VIN SW –VOUT C1 LT1611 R1 NFB GND VIN SW –VOUT C1 LT1611 GND 1611 F03 Figure 3. Direct Regulation of Negative Output Using Boost Converter with Charge Pump R1 NFB C3 R2 10k + SHDN D1 + C3 R2 10k + VIN SHUTDOWN Output ripple voltage appears as a triangular waveform riding on VOUT. Ripple magnitude equals the ripple current of L2 multiplied by the equivalent series resistance (ESR) of output capacitor C3. Increasing the inductance of L1 and L2 lowers the ripple current, which leads to lower output voltage ripple. Decreasing the ESR of C3, by using ceramic or other low ESR type capacitors, lowers output ripple voltage. Output ripple voltage can be reduced to arbitrarily low levels by using large value inductors and low ESR, high value capacitors. 1611 F04 Figure 4. L2 Replaces D2 to Make Low Output Ripple Inverting Topology. Coupled or Uncoupled Inductors Can Be Used. Follow Phasing If Coupled for Best Results 5 LT1611 U OPERATIO –(VIN + VOUT) VCESAT L1 SW C2 L2 SWX VIN –VOUT D1 Q1 + C1 C3 RLOAD + 1611 F05 Figure 5. Switch-On Phase of Inverting Converter. L1 and L2 Current Have Positive dI/dt VIN + VOUT+ VD L1 SW VD C2 L2 SWX VIN –VOUT Q1 + D1 C1 C3 RLOAD + 1611 F06 Figure 6. Switch-Off Phase of Inverting Converter. L1 and L2 Current Have Negative dI/dt Transient Response The inverting architecture of the LT1611 can generate a very low ripple output voltage. Recently available high value ceramic capacitors can be used successfully in LT1611 designs with the addition of a phase lead capacitor, CPL (see Figure 7). Connected in parallel with feedback resistor R1, this capacitor reduces both output perturba- 6 tions due to load steps and output ripple voltage to very low levels. To illustrate, Figure 7 shows an LT1611 inverting converter with resistor loads RL1 and RL2. RL1 is connected across the output, while RL2 is switched in externally via a pulse generator. Output voltage waveforms are pictured in subsequent figures, illustrating the performance of output capacitor type and the effect of CPL connected across R1. LT1611 U OPERATIO C2 1µF L1A 22µH VIN 5V L1B 22µH D1 VIN –VOUT SW SHDN + C1 R1 LT1611 CPL NFB C3 RL2 50Ω + GND RL1 100Ω R2 10k C1: AVX TAJB226M010 C2: TAIYO YUDEN LMK212BJ105MG C3: SEE TEXT D1: MBR0520 L1A, L1B: SUMIDA CLS62-220 1611 F07 Figure 7. Switching RL2 Provides 50mA to 150mA Load Step for LT1611 5V to – 5V Converter Figure 8 shows the output voltage with a 50mA to 150mA load step, using an AVX TAJ “B” case 22µF tantalum capacitor at the output. Output perturbation is approximately 100mV as the load changes from 50mA to 150mA. Steady-state ripple voltage is 20mVP–P, due to L1’s ripple current and C3’s ESR. Step response can be improved by adding a 3.3nF capacitor (CPL) as shown in Figure 9. Settling time improves from 150µs to 40µs, although steady-state ripple voltage does not improve. Figure 10 pictures the output voltage and switch pin voltage at 200ns per division. Note the absence of high frequency spikes at the output. This is easily repeatable with proper layout, described in the next section. VOUT 50mV/DIV AC COUPLED LOAD CURRENT VOUT 20mV/DIV AC COUPLED 150mA 50mA LOAD CURRENT 100µs/DIV 150mA 50mA 20µs/DIV 1611 F08 Figure 8. Load Step Response of LT1611 with 22µF Tantalum Output Capacitor 1611 F09 Figure 9. Addition of CPL to Figure 7’s Circuit Improves Load Step Response. CPL = 3.3nF VOUT 10mV/DIV SWITCH VOLTAGE 5V/DIV LOAD = 150mA 200ns/DIV 1611 F10 Figure 10. 22µF “B” Case Tantalum Capacitor (AVX TAJ “B” Series) Has ESR Resulting in 20mVP–P Voltage Ripple at Output 7 LT1611 U OPERATIO In Figure 11 (also shown on the first page), output capacitor C3 is replaced by a ceramic unit. These large value ceramic capacitors have ESR of about 2mΩ and result in very low output ripple. At the 20mV/division scale, output voltage ripple cannot be seen. Figure 12 pictures the output and switch nodes at 200ns per division. The output voltage ripple is approximately 1mVP–P. Again, good layout is mandatory to achieve this level of performance. Layout The LT1611 switches current at high speed, mandating careful attention to layout for best performance. You will not get advertised performance with careless layout. Figure␣ 13 shows recommended component placement. Follow this closely in your printed circuit layout. The cut ground copper at D1’s cathode is essential to obtain the low noise achieved in Figures 11 and 12’s oscillographs. Input bypass capacitor C1 should be placed close to the LT1611 as shown. The load should connect directly to output capacitor C2 for best load regulation. You can tie the local ground into the system ground plane at C3’s ground terminal. VOUT 5mV/DIV AC COUPLED VOUT 20mV/DIV AC COUPLED SWITCH VOLTAGE 5V/DIV 150mA LOAD CURRENT 50mA 100µs/DIV LOAD = 150mA 1611 F11 Figure 11. Replacing C3 with 22µF Ceramic Capacitor (Taiyo Yuden JMK325BJ226MM) Improves Output Noise. CPL = 1200pF Results in Best Phase Margin L1B Figure 12. 22µF Ceramic Capacitor at Output Reduces Ripple to 1mVP–P. Proper Layout Is Essential to Achieve Low Noise L1A C1 –VOUT D1 200ns/DIV + C2 VIN C3 5 + 1 2 3 4 SHUTDOWN R2 GND R1 1611 F13 Figure 13. Suggested Component Placement. Note Cut in Ground Copper at D1’s Cathode 8 1611 F12 LT1611 U OPERATIO Start-Up/Soft-Start measured at VIN, is limited to a peak value of 450mA as the time required to reach final value increases to 700µs. In Figure 16, CSS is increased to 0.1µF, resulting in a lower peak input current of 240mA with a VOUT ramp time of 2.1ms. CSS can be increased further for an even slower ramp, if desired. Diode D2 serves to quickly discharge CSS when VSS is driven low to shut down the device. D2 can be omitted, resulting in a “soft-stop” slow discharge of the output capacitor. The LT1611, starting from VOUT = 0V, reaches final voltage in approximately 450µs after SHDN is pulled high, with COUT = 22µF, VIN = 5V and VOUT = – 5V. Charging the output capacitor at this speed requires an inrush current of over 1A. If a longer start-up time is acceptable, a soft-start circuit consisting of RSS and CSS, as shown in Figure 14, can be used to limit inrush current to a lower value. Figure 15 pictures VOUT and input current, starting into a 33Ω load, with RSS of 33kΩ and CSS of 33nF. Input current, CURRENT PROBE VIN 5V + C2 1µF L1A 22µH D1 C1 22µF VIN RSS 33k SW R1 29.4k LT1611 VSS SHDN VOUT CP 1200pF VOUT –5V C3 22µF NFB GND D2 1N4148 CSS 33nF/0.1µF L1B 22µH R2 10k C1: AVX TAJB226M010 C2: TAIYO YUDEN LMK212BJ105MG C3: TAIYO YUDEN JMK325BJ226MM (1210 SIZE) D1: MBR0520 L1: SUMIDA CLS62-220 OR 2× MURATA LQH3C220 (UNCOUPLED) 1611 F14 Figure 14. RSS and CSS at SHDN Pin Provide Soft-Start to LT1611 Inverting Converter VOUT 2V/DIV VOUT 2V/DIV IIN 200mA/DIV IIN 200mA/DIV VS 5V/DIV VS 5V/DIV LOAD = 150mA 500µs/DIV 1611 F15 Figure 15. RSS = 33k, CSS = 33nF; VOUT Reaches – 5V in 750µs; Input Current Peaks at 450mA LOAD = 150mA 500µs/DIV 1611 F16 Figure 16. RSS = 33k, CSS = 0.1µF; VOUT Reaches – 5V in 2.1ms; Input Current Peaks at 240mA 9 LT1611 U OPERATIO COMPONENT SELECTION Output Current The LT1611 will deliver 150mA at – 5V from a 5V ±10% input supply. If a higher voltage supply is available, more output current can be obtained. Figure 17’s schematic shows how to get more current. Although the LT1611’s maximum voltage allowed at VIN is 10V, the SW pin can handle higher voltage (up to 36V). In Figure 17, the VIN pin of the LT1611 is driven from a 5V supply, while input inductor L1A is driven from a separate 12V supply. Figure 18’s graph shows maximum recommended output current as the voltage on L1A is varied. Up to 300mA can be delivered when driving L1A from a 12V supply. Inductors Each of the two inductors used with the LT1611 should have a saturation current rating (where inductance is approximately 70% of zero current inductance) of approximately 0.25A or greater. If the device is used in “charge pump” mode, where there is only one inductor, then its rating should be 0.5A or greater. DCR of the inductors should be 0.5Ω or less. A value of 22µH is suitable if using a coupled inductor such as Sumida CLS62-220 or Coiltronics CTX20-1. If using two separate inductors, increasing the value to 47µH will result in the same ripple current. Inductance can be reduced if operating from a supply voltage below 3V. Table 1 lists several inductors that will work with the LT1611, although this is not an exhaustive list. There are many magnetics vendors whose components are suitable. VL (SEE TEXT) 350 C2 1µF L1B 22µH 5V D1 VIN SW SHDN C1 1µF 29.4k LT1611 C3 22µF NFB GND 1200pF VOUT –5V UP TO 300mA 10k C1, C2: TAIYO YUDEN LMK212BJ105MG C3: TAIYO YUDEN JMK325BJ226MM D1: MBR0520 L1A, L1B: SUMIDA CLS62-220 300 250 200 150 100 1611 F17 Figure 17. Increase Output Current By Driving L1A from a Higher Voltage 10 MAXIMUM RECOMMENDED OUTPUT CURRENT (mA) L1A 22µH 3 4 5 6 7 8 VL (V) 9 10 11 12 1611 F18 Figure 18. Output Current Increases to 300mA When Driving VL from 12V Supply LT1611 U OPERATIO Capacitors ceramic can be used with little trade-off in circuit performance. Some capacitor types appropriate for use with the LT1611 are listed in Table 2. As described previously, ceramic capacitors can be used with the LT1611 provided loop stability is considered. For lower cost applications, small tantalum units can be used. A value of 22µF is acceptable, although larger capacitance values can be used. ESR is the most important parameter in selecting an output capacitor. The “flying” capacitor (C2 in the schematic figures) should be a 1µF ceramic type. An X5R or X7R dielectric should be used to avoid capacitance decreasing severely with applied voltage. The input bypass capacitor is less critical, and either tantalum or Diodes A Schottky diode is recommended for use with the LT1611. The Motorola MBR0520 is a very good choice. Where the input to output voltage differential exceeds 20V, use the MBR0530 ( a 30V diode). If cost is more important than efficiency, a 1N4148 can be used, but only at low current loads. Table 1. Inductor Vendors VENDOR PHONE URL PART COMMENT Sumida (847) 956-0666 www.sumida.com CLS62-22022 CD43-470 22µH Coupled 47µH Murata (404) 436-1300 www.murata.com LQH3C-220 22µH, 2mm Height Coiltronics (407) 241-7876 www.coiltronics.com CTX20-1 20µH Coupled, Low DCR Table 2. Capacitor Vendors VENDOR PHONE URL PART COMMENT Taiyo Yuden (408) 573-4150 www.t-yuden.com Ceramic Caps X5R Dielectric AVX (803) 448-9411 www.avxcorp.com Ceramic Caps Tantalum Caps Murata (404) 436-1300 www.murata.com Ceramic Caps U TYPICAL APPLICATIO S “Charge Pump” Inverting DC/DC Converter C2 1µF L1 10µH 3.3V D2 D1 VIN SW –5V 70mA SHDN C1 1µF 29.4k LT1611 C3 22µF NFB GND 10k C1, C2: TAIYO YUDEN LMK212BJ105MG C3: TAIYO YUDEN JMK325BJ226MM D1, D2: MBR0520 L1: MURATA LQH3C-100 1611 TA02 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 11 LT1611 U TYPICAL APPLICATIO S 4-Cell to –10V Inverting Converter C2 1µF L1A 15µH 4-Cell to –10V Inverting Converter Efficiency 85 L1B 15µH VIN 80 D1 SHUTDOWN VIN SW LT1611 SHDN VOUT –10V/60mA 68.1k NFB GND EFFICIENCY (%) C1 22µF + + VIN = 6.5V 75 10k C3 6.8µF VIN = 5V VIN = 3.6V 70 65 60 55 C1: AVX TAJB226M010 (803) 946-0362 C2: TAIYO YUDEN LMK212BJ105MG C3: AVX TAJA685M016 D1: MOTOROLA MBR0520 (800) 441-2447 L1: SUMIDA CL562-150 (847) 956-0666 U PACKAGE DESCRIPTION 1611 TA03 50 0 25 50 75 100 LOAD CURRENT (mA) 125 150 1611 TA04 Dimensions in inches (millimeters) unless otherwise noted. S5 Package 5-Lead Plastic SOT-23 (LTC DWG # 05-08-1633) 2.60 – 3.00 (0.102 – 0.118) 1.50 – 1.75 (0.059 – 0.069) 0.35 – 0.55 (0.014 – 0.022) 0.00 – 0.15 (0.00 – 0.006) 0.09 – 0.20 (0.004 – 0.008) (NOTE 2) 0.90 – 1.45 (0.035 – 0.057) 2.80 – 3.00 (0.110 – 0.118) (NOTE 3) 0.35 – 0.50 0.90 – 1.30 (0.014 – 0.020) (0.035 – 0.051) FIVE PLACES (NOTE 2) NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DIMENSIONS ARE INCLUSIVE OF PLATING 3. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 4. MOLD FLASH SHALL NOT EXCEED 0.254mm 5. PACKAGE EIAJ REFERENCE IS SC-74A (EIAJ) 0.95 (0.037) REF 1.90 (0.074) REF S5 SOT-23 0599 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1307 Single Cell Micropower DC/DC with Low Battery Detector 3.3V/75mA from 1V, 600kHz Fixed Frequency TM LT1316 Burst Mode Operation DC/DC with Programmable Current Limit 1.5V Minimum VIN, Precise Control of Peak Switch Current LT1317 2-Cell Micropower DC/DC with Low Battery Detector 3.3V/200mA from Two Cells, 600kHz Fixed Frequency LT1370/LT1371 500kHz High Efficiency DC/DC Converter 42V, 6A/3A Internal Switch, Negative Feedback Regulation LT1610 Single Cell Micropower DC/DC 3V/30mA from 1V, 1.7MHz Fixed Frequency, 30µA IQ LT1613 1.4MHz SOT-23 Step-Up DC/DC Converter 5V at 200mA from 3.3V Input LT1614 Inverting Mode Switching Regulator with Low-Battery Detector – 5V at 200mA from 5V Input in MSOP LT1615 Micropower SOT-23 Step-Up DC/DC Converter 20µA Quiescent Current, VOUT Up to 34V LT1617 Micropower SOT-23 Inverting Regulator VOUT Up to –34V, 20µA Quiescent Current Burst Mode is a trademark of Linear Technology Corporation. 12 Linear Technology Corporation 1611f LT/TP 0999 4K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com LINEAR TECHNOLOGY CORPORATION 1998