Linear Dimensions LT8697EUDD Usb 5v 2.5a output, 42v input synchronous buck with cable drop compensation Datasheet

Electrical Specifications Subject to Change
LT8697
USB 5V 2.5A Output,
42V Input Synchronous Buck with
Cable Drop Compensation
Description
Features
n
n
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n
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Wide Input Range: 5V to 42V
Low Dropout Under All Conditions: 450mV at 2.1A
Accurate 5V Output: ±1.3% Over Full Temperature
Range
Programmable Cable Drop Compensation
Programmable Output Current Limit
Output Current Monitor
Dual Input Feedback Permits Regulation on Output
of USB Switch
Forced Continuous Mode for Fast Load Step
Response
High Efficiency Synchronous Operation at 2MHz:
93% Efficiency at 2.1A, 5VOUT from 12VIN
95% Efficiency at 0.9A, 5VOUT from 12VIN
Fast Minimum Switch-On Time: 70ns
Adjustable Output from 5.0V to 5.25V
Adjustable and Synchronizable: 300kHz to 2.2MHz
Small Thermally Enhanced 3mm × 5mm 24-Lead
QFN Package
The LT®8697 is a compact, high efficiency, high speed
synchronous monolithic step-down switching regulator
designed to power 5V USB applications. A precise output
voltage and programmable cable drop compensation
maintain accurate 5V regulation at the USB socket at the
end of a long cable. Forced continuous operation allows
the LT8697 to sink current, further enhancing accurate
5V regulation during load transients.
Accurate and programmable output current limit, a power
good indicator pin and an output current monitor pin
improve system reliability and safety, allow the user to
implement latch-off or auto-retry functionality and can
eliminate the need for a USB switch IC. Dual feedback allows regulation on the output of a USB switch and limits
cable drop compensation to a maximum of 5.8V output,
protecting USB devices during fault conditions. Thermal
shutdown provides additional protection by limiting power
dissipation in the IC during an overtemperature fault.
The LT8697 is available in a small 24-lead 3mm × 5mm
package with an exposed pad for low thermal resistance.
Applications
n
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L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
of Linear Technology Corporation. All other trademarks are the property of their respective owners.
Automotive and Industrial USB
Precision 5V Supply
Typical Application
2MHz 5V Step-Down Converter with Cable Drop Compensation
5.50
0.1µF
4.7µF
VIN
VOUT
BST
3.3µH
EN/UV
1µF
10nF
16.5k
SYS
LT8697
ICTRL
ISP
SYNC
ISN
TR/SS
USB5V
RT
47µF
×2
1nF
+
10k
LOAD
INTVCC
PGND
VLOAD
5V, 2.4A
–
RCBL
GND
5.25
0.1Ω
0.018Ω
18.2k
0.1Ω
5.00
3
VLOAD
4.75
4.50
4.25
8697 TA01a
4
VOUT
2
ILOAD
50mA/µs
100µs/DIV
CURRENT (A)
PG
SW
5
3 METERS AWG 20
TWISTED PAIR CABLE
VOLTAGE (V)
VIN
6V TO 42V
Transient Response Through 3 Meters
AWG 20 Twisted-Pair Cable
1
0
8697 TA01b
8697p
For more information www.linear.com/LT8697
1
LT8697
Pin Configuration
ISP
ISN
ICTRL
RCBL
TOP VIEW
VIN, EN/UV, PG, ISP, ISN............................................42V
SYS............................................................................30V
USB5V......................................................................3mA
BST Above SW.............................................................4V
SW Above VIN...........................................................0.3V
TR/SS, ICTRL..............................................................4V
RT, RCBL......................................................................2V
SYNC...........................................................................6V
Operating Junction Temperature Range (Notes 2, 3)
LT8697E.................................................. –40°C to 125°C
LT8697I................................................... –40°C to 125°C
24 23 22 21
SYNC 1
20 USB5V
TR/SS 2
19 PG
RT 3
18 SYS
EN/UV 4
17 INTVCC
25
GND
VIN 5
16 BST
VIN 6
15 SW
PGND 7
14 SW
PGND 8
13 SW
NC
NC
9 10 11 12
NC
(Note 1)
NC
Absolute Maximum Ratings
UDD PACKAGE
24-LEAD (3mm × 5mm) PLASTIC QFN
θJA = 46°C/W, θJC = 5°C/W
EXPOSED PAD (PIN 25) IS GND, MUST BE SOLDERED TO PCB
Order Information
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT8697EUDD#PBF
LT8697EUDD#TRPBF
LGGW
24-Lead (3mm × 5mm) Plastic QFN
–40°C to 125°C
LT8697IUDD#PBF
LT8697IUDD#TRPBF
LGGW
24-Lead (3mm × 5mm) Plastic QFN
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
Electrical Characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. (Notes 2, 4)
PARAMETER
CONDITIONS
VIN Undervoltage Lockout
MIN
l
VIN Shutdown Current
VEN/UV = 0.3V
VIN Current in Regulation
VIN = 12V, ILOAD = 100µA, RT = 56.2k
l
l
TYP
MAX
2.9
3.4
V
1
3
8
µA
µA
9
12
mA
l
VIN to Disable Forced Continuous Mode
VIN Rising
Output Sink Current in Forced Continuous Mode
VUSB5V = 5.5V, L = 6.8µH, RT = 56.2k
USB5V Voltage
VIN = 12V
l
USB5V Voltage Line Regulation
VIN = 6V to 42V
l
Regulated Load Voltage Through 0.3Ω
VIN = 12V, ILOAD = 2.1A, Voltage at Point of Load (End
of Cable), RCBL = 13.7k, RCDC = 10k, RSENSE = 20mΩ
l
USB5V Clamp Voltage
IUSB5V = 3mA
UNIT
27
29
31
V
0.6
1
1.7
A
4.97
4.91
4.99
4.99
5.01
5.04
V
V
6
35
4.925
5
5.075
9
mV
V
V
8697p
2
For more information www.linear.com/LT8697
LT8697
Electrical Characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. (Notes 2, 4)
PARAMETER
CONDITIONS
MIN
TYP
MAX
USB5V Current
VISN = 5V, VISP – VISN = 40mV, RCBL = 13.7k
VISN = 5V, VISP – VISN = 10mV, RCBL = 13.7k
VISN = 5V, VISP – VISN = 0V, RCBL = 13.7k
RCBL = Open
l
l
l
l
58
12
0
0
60
15
2
1
62
23
8
2
µA
µA
µA
µA
Current Sense Voltage (VISP – VISN)
VCTRL = 1.5V, VISN = 5V
VCTRL = 1.5V, VISN = 0V
VCTRL = 800mV, VISN = 5V
VCTRL = 800mV, VISN = 0V
VCTRL = 200mV, VISN = 5V
VCTRL = 200mV, VISN = 0V
l
l
l
l
l
l
45.5
44.5
37.5
36
6
5
48
48.5
39.5
40
9.5
10
50
53
42
44.5
13
15
mV
mV
mV
mV
mV
mV
RCBL Monitor Voltage
VISP – VISN = 40mV, RCBL = 13.7k
VISP – VISN = 10mV, RCBL = 13.7k
l
l
720
130
800
205
880
280
mV
mV
RCBL Output Current Limit
VISP – VISN = 50mV, VRCBL = 0V
–2
–3
–4
mA
20
µA
–3
µA
ISP, ISN Bias Current
VISP = VISN = 0V, 5V
–20
ICTRL Current
VICTRL = 1.5V
–0.5
INTVCC Voltage
VSYS = 0V, 5V
3.4
INTVCC Undervoltage Lockout
SYS Voltage
VIN = 12V
–2
l
V
2.6
2.9
3.15
V
5.63
5.8
5.92
V
SYS Voltage to Disable Forced Continuous Mode
7.5
SYS Current in Regulation
VSYS = 5V, RT = 56.2k
Dropout Voltage (VIN – VSYS)
VSYS = 5V, ILOAD = 2.1A
l
Minimum On-Time
ILOAD = 1A
Minimum Off-Time
ILOAD = 0.5A
3
4
96
97.5
30
50
V
5
450
Maximum Duty Cycle in Dropout
l
UNIT
mA
mV
99
%
45
70
ns
80
110
ns
Minimum VIN for SYS Regulation at Full Frequency RT = 16.5k, VUSB5V = 0V, ILOAD = 0.5A
l
6.2
7
7.9
V
Oscillator Frequency
RT = 140k
RT = 56.2k
RT = 16.5k
l
l
l
250
620
1.9
300
700
2.00
340
750
2.05
kHz
kHz
MHz
Top Power NMOS On-Resistance
ISW = 1A
l
3.2
4.8
3.5
4.5
0.94
Top Power NMOS Current Limit
Bottom Power NMOS On-Resistance
VINTVCC = 3.4V, ISW = 1A
Bottom Power NMOS Current Limit
VINTVCC = 3.4V
SW Leakage Current
VIN = 42V, VSW = 0V, 42V
EN/UV Threshold
VEN/UV Rising
120
mΩ
6
65
l
EN/UV Hysteresis
A
mΩ
5.8
A
0.1
5
µA
1.0
1.06
V
40
mV
EN/UV Bias Current
VEN/UV = 2V
PG Upper Threshold Offset from VUSB5V
VUSB5V Falling
l
6
9
12
%
PG Lower Threshold Offset from VUSB5V
VUSB5V Rising
l
–6
–9
–12
%
VPG = 0.1V
l
–20
PG Hysteresis
PG Pull-Down Resistance
20
1.3
680
PG Transition Delay
VUSB5V from 5V to 4V
SYNC Threshold
VSYNC Falling
VSYNC Rising
0.8
1.6
SYNC Current
VSYNC = 2V
–40
TR/SS Current
VTR/SS = 0V
–1.2
TR/SS Pull-Down Resistance
Fault Condition, VTR/SS = 0.1V
%
2000
40
1.1
2.0
–2.2
230
nA
Ω
µs
1.4
2.4
V
V
40
nA
–3.2
µA
Ω
8697p
For more information www.linear.com/LT8697
3
LT8697
Electrical Characteristics
Note 3: This IC includes overtemperature protection that is intended to
protect the device during overload conditions. Junction temperature will
exceed 150°C when overtemperature protection is active. Continuous
operation above the specified maximum operating junction temperature
will reduce lifetime.
Note 4: Polarity specification for current into a pin is positive and out of
a pin is negative. All voltages are referenced to GND unless otherwise
specified. MAX and MIN refer to absolute values.
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT8697E is guaranteed to meet performance specifications
from 0°C to 125°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design,
characterization, and correlation with statistical process controls. The
LT8697I is guaranteed over the full –40°C to 125°C operating junction
temperature range. High junction temperatures degrade operating
lifetimes.
Typical Performance Characteristics
5.00
VUSB5V vs Temperature
5.02
VIN = 12V
4.99
TA = 25°C, unless otherwise noted.
VUSB5V vs VIN
70
IUSB5V vs Temperature
RCBL = 13.7k
60
5.01
VSENSE = 40mV
4.97
5.00
IUSB5V (µA)
4.98
VUSB5V (V)
VUSB5V (V)
50
4.99
40
30
VSENSE = 10mV
20
4.96
4.98
4.95
–55
–25
5
65
35
TEMPERATURE (°C)
95
4.97
125
10
6
10
14
18
22 26
VIN (V)
30
34
8697 G01
225
200
62
IUSB5V (µA)
IUSB5V (µA)
175
125
100
75
VIN = 12V
RCDC = 10k
RCBL = 13.3k
ILOAD = 2A
RSENSE = 20mΩ
0
10
30
20
VSENSE (mV)
40
50
60
58
54
0.2
155
VSENSE vs VCTRL
40
30
20
10
0.7
1.2
1.7
2.2
FREQUENCY (MHz)
8697 G04
125
50
56
RCBL = 40.2k
25
0
60
RCBL = 13.7k
50
5
65
35
95
TEMPERATURE (°C)
8697 G03
IUSB5V vs Frequency
64
RCBL = 4.12k
–25
8697 G02
IUSB5V vs VSENSE
150
0
–55
42
VSENSE (mV)
250
38
VSENSE = 0mV
8697 G05
0
0
0.2
0.4
0.8
0.6
VCTRL (V)
1
1.2
8697 G06
8697p
4
For more information www.linear.com/LT8697
LT8697
Typical Performance Characteristics
VEN vs Temperature
100
1.01
EFFICIENCY (%)
EN/UV THRESHOLD (V)
1.02
EN/UV RISING
1.00
0.99
0.98
0.97
Efficiency vs ILOAD
100
95
90
90
80
85
70
80
75
fSW = 2MHz
RSENSE = 18mΩ
DCRL = 20mΩ
RCBL = OPEN
70
65
60
EN/UV FALLING
0.96
–25
35
5
65
TEMPERATURE (°C)
95
50
125
0
0.5
1.5
1
LOAD CURRENT (A)
Efficiency vs Frequency
100
16
70
60
0.2
VIN = 8V
VIN = 12V
VIN = 24V
0.7
VIN = 8V
VIN = 12V
VIN = 24V
0
0.001
00.01
1
0.1
LOAD CURRENT (A)
fSW = 700kHz
12
10
8
6
4
0
2.2
6
12
24
30
18
INPUT VOLTAGE (V)
36
8697 G10
8697 G09
10.0
fSW = 700kHz
VIN = 12V
9.5
9.0
8.5
8.0
7.5
7.0
–55
42
–25
5
35
65
95
TEMPERATURE (°C)
Top FET Current Limit
vs Temperature
155
Bottom FET Current Limit
vs Temperature
5.0
5.0
5.0
125
8697 G12
8697 G11
Top FET Current Limit
vs Duty Cycle
10
No Load Supply Current
vs Temperature
2
1.7
1.2
FREQUENCY (MHz)
20
NO LOAD INPUT CURRENT (mA)
INPUT CURRENT (mA)
EFFICIENCY (%)
L = 15µH
RSENSE = 18mΩ
DCRL = 40mΩ
RCBL = OPEN
ILOAD = 0.9A
30
10
No Load Supply Current vs VIN
14
80
fSW = 2MHz
RSENSE = 18mΩ
DCRL = 20mΩ
RCBL = OPEN
40
8697 G08
8697 G07
90
50
2.5
2
Efficiency vs ILOAD
60
VIN = 8V
VIN = 12V
VIN = 24V
55
0.95
–55
EFFICIENCY (%)
1.03
TA = 25°C, unless otherwise noted.
30% DC
4.1
CURRENT LIMIT (A)
4.4
70% DC
4.0
3.5
3.8
3.5
4.6
4.5
CURRENT LIMIT (A)
CURRENT LIMIT (A)
4.7
0
0.2
0.4
0.6
DUTY CYCLE (%)
0.8
1.0
8697 G13
3.0
–55
4.2
3.8
3.4
–25
5
35
65
TEMPERATURE (°C)
95
125
8697 G13
3.0
–55
–25
5
35
65
95
TEMPERATURE (°C)
125
155
8697 G15
8697p
For more information www.linear.com/LT8697
5
LT8697
Typical Performance Characteristics
Switch Drop vs Temperature
350
SWITCH CURRENT = 1A
100
300
SWITCH DROP (mV)
200
SWITCH DROP (mV)
Minimum On-Time
vs Temperature
Switch Drop vs ISW
TOP SW
150
100
BOTTOM SW
90
250
200
TOP SW
150
100
BOTTOM SW
50
–25
5
35
65
95
TEMPERATURE (°C)
125
0
155
0
0.5
1.5
2
1
SWITCH CURRENT (A)
95
ILOAD = 2.1A
95
90
ILOAD = 0.9A
85
ILOAD = 0A
80
75
–55
–25
5
35
65
95
TEMPERATURE (°C)
90
85
ILOAD = 2.1A
80
75
ILOAD = 0.9A
70
ILOAD = 0A
60
–55 –25
155
2.10
2.05
2.00
1.95
1.90
300
200
100
95
65
35
TEMPERATURE (°C)
125
5
0
155
0
0.5
1.5
1
LOAD CURRENT (A)
2
Switching Frequency vs ILOAD
2500
RT = 16.5k
2050
2000
1950
2.5
8697 G21
SWITCHING FREQUENCY (kHz)
RT = 16.5k
SWITCHING FREQUENCY (kHz)
SWITCHING FREQUENCY (MHz)
2100
155
400
8697 G20
Switching Frequency
vs Temperature
2.15
125
DCRL = 20mΩ
RSENSE = 18mΩ
500
8697 G19
2.20
65
35
95
5
TEMPERATURE (°C)
Dropout Voltage vs ILOAD
600
RT = 30.1k
SYNC = 1.2MHz
65
125
ILOAD = 2.1A
8697 G18
DROPOUT VOLTAGE (mV)
MINIMUM OFF-TIME (ns)
MINIMUM OFF-TIME (ns)
100
105
ILOAD = 0.9A
50
Minimum Off-Time
vs Temperature
RT = 30.1k
SYNC = 3V
100
60
8697 G17
Minimum Off-Time
vs Temperature
110
ILOAD = 0A
70
30
–55 –25
2.5
8697 G16
115
80
40
50
0
–55
120
MINIMUM ON-TIME (ns)
250
TA = 25°C, unless otherwise noted.
Switching Frequency vs VSYS
RT = 16.5k
2000
1500
1000
500
1.85
1.80
–55
–25
5
35
65
95
TEMPERATURE (°C)
125
155
8697 G22
1900
0.0001
0.001
0.1
0.01
LOAD CURRENT (A)
1
10
8697 G23
0
0
1
3
2
VSYS (V)
4
5
8697 G24
8697p
6
For more information www.linear.com/LT8697
LT8697
Typical Performance Characteristics
VUSB5V vs VTR/SS
2.2
11.0
VTR/SS = 0.5V
2.1
5
PG High Thresholds
ITR/SS vs Temperature
PG THRESHOLD OFFSET FROM VUSB5V (%)
6
TA = 25°C, unless otherwise noted.
2.0
ITR/SS (µA)
VUSB5V (V)
4
3
2
1.9
1.8
1.7
1.6
1
0
1.5
0
0.2
0.4
0.6
0.8
1.0
1.4
–55
1.2
5
35
65
95
TEMPERATURE (°C)
–25
VTR/SS (V)
9.5
USB5V RISING
9.0
8.5
USB5V FALLING
8.0
7.5
7.0
6.5
6.0
–55
–25
35
65
95
5
TEMPERATURE (°C)
13
12
fSW = 2MHz
–8.5
12
10
11
8
VIN = 24V
USB5V FALLING
–10.5
–11.0
–11.5
–12.0
VIN = 12V
ISYS (mA)
ISYS (mA)
–10.0
155
ISYS vs Switching Frequency
USB5V RISING
–9.5
125
8697 G27
ISYS vs VIN
PG Low Thresholds
–8.0
10
6
9
4
8
2
–12.5
–25
35
65
95
5
TEMPERATURE (°C)
125
155
7
5
10
15
20
25 30
VIN (V)
35
8697 G28
Transient Response
0A to 1A Load Step
5.50
FRONT PAGE
APPLICATION CIRCUIT
5.50
4
5.25
3
VLOAD
2
0.5
1
1.5
2
SWITCHING FREQUENCY (MHz)
2.5
8697 G30
5
VOUT
4
5.00
3
VLOAD
4.75
2
ILOAD
25mA/µs
ILOAD
25mA/µs
100µs/DIV
0
CURRENT (A)
4.25
5
CURRENT (A)
VOUT
4.75
4.50
0
45
Transient Response
1A to 2A Load Step
5.25
5.00
40
8697 G29
VOLTAGE (V)
–13.0
–55
VOLTAGE (V)
PG THRESHOLD OFFSET FROM VUSB5V (%)
155
10.0
8697 G26
8697 G25
–9.0
125
10.5
1
0
4.50
4.25
8697 G31
1
FRONT PAGE
APPLICATION CIRCUIT
100µs/DIV
0
8697 G32
8697p
For more information www.linear.com/LT8697
7
LT8697
Typical Performance Characteristics
Transient Response Output
Current Limit
10
TA = 25°C, unless otherwise noted.
Transient Response USB5V
Shorted to GND
FRONT PAGE
APPLICATION CIRCUIT
5
12
6.5
9
6.0
ILOAD = 0A
FRONT PAGE
APPLICATION CIRCUT
6
VPG
–5
3
ILOAD
50mA/µs
–10
–15
100µs/DIV
VOUT (V)
0
CURRENT (A)
VOLTAGE (V)
VLOAD
5.5
0
4.5
–3
4.0
8697 G34
Start-Up Dropout Performance
RLOAD = ∞
RLOAD = 5.6Ω
VOUT
2V/DIV
100µs/DIV
8697 G33
Start-Up Dropout Performance
VIN
2V/DIV
USB5V
SHORTED
TO GND
5.0
VIN
VIN
2V/DIV
VOUT
100ms/DIV
VOUT
2V/DIV
VIN
VOUT
100ms/DIV
8697 G35
8697 G36
Pin Functions
SYNC (Pin 1): External Clock Synchronization Input.
Tie to a clock source for synchronization to an external
frequency and forced continuous mode. Tie to INTVCC if
not used. Do not float.
TR/SS (Pin 2): Output Tracking and Soft-Start Pin. This
pin allows user control of output voltage ramp rate during
start-up. A TR/SS voltage below 0.97V forces the LT8697
to regulate VUSB5V to 5 times the TR/SS voltage. When
TR/SS is above 0.97V, the tracking function is disabled
and the internal reference resumes control of the error
amplifier. An internal 2.2µA pull-up current from INTVCC
on this pin allows a capacitor to program output voltage
slew rate. This pin is pulled to ground when EN/UV is low,
during thermal shutdown and when VIN below its undervoltage lockout threshold; use a series resistor of at least
10k if driving from a low impedance output.
RT (Pin 3): Tie a resistor between RT and ground to set
the switching frequency.
EN/UV (Pin 4): The LT8697 is shut down when this pin
is low and active when this pin is high. The hysteretic
threshold voltage is 1.00V going up and 0.96V when going
down. Tie to VIN if the shutdown feature is not used. An
external resistor divider from VIN can be used to program
a VIN threshold below which the LT8697 will shut down.
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LT8697
Pin Functions
VIN (Pins 5, 6): The VIN pins supply current to the LT8697
internal circuitry and to the internal topside power switch.
These pins must be tied together and be locally bypassed.
Place the positive terminal of the input capacitor as close
as possible to the VIN pins, and the negative terminal as
close as possible to the PGND pins.
PGND (Pins 7, 8): Power Switch Ground. These pins are
the return path of the internal bottom side power switch
and must be tied together. Place the negative terminal of
the input capacitor as close as possible to the PGND pins.
NC (Pins 9-12): No Connect. These pins are floating and
are not connected to the LT8697. Tie these pins to the
same copper as the exposed pad. See Figure 8.
SW (Pins 13, 14, 15): The SW pins are the outputs of the
internal power switches. Tie these pins together and connect them to the inductor and boost capacitor. This node
should be kept small on the PCB for good performance.
BST (Pin 16): This pin is used to provide a drive voltage,
higher than the input voltage, to the topside power switch.
Place a 0.1µF boost capacitor between this pin and SW as
close as possible to the LT8697 IC.
INTVCC (Pin 17): Internal 3.4V Regulator Bypass Pin. The
internal power drivers and control circuits are powered
from this voltage. The INTVCC maximum output current
is 20mA. INTVCC current will be supplied from SYS if
VSYS > 3.1V, otherwise current will be drawn from VIN.
Decouple this pin to power ground with at least a 1µF low
ESR ceramic capacitor. Do not load the INTVCC pin with
external circuitry.
SYS (Pin 18): The internal regulator will draw current
from SYS instead of VIN when SYS is tied to a voltage
higher than 3.3V. The SYS pin must be tied to the side of
the inductor opposite the SW pin and must be bypassed
by the output capacitor. SYS is also the secondary input
to the error amp and regulates to a maximum of 5.8V.
USB5V (Pin 20): The LT8697 regulates the USB5V pin to
5V. For cable drop compensation, the USB5V pin input
current is proportional to the sensed output current. The
USB5V ESD cell clamps to 9V. To allow the LT8697 output
to survive a short to 30V, the 10k RCDC resistor must be
in place between the USB5V pin and the output to limit
the current into this pin.
ISP (Pin 21): Current Sense (+) Pin. This is the non-inverting
input to the current sense amplifier.
ISN (Pin 22): Current Sense (–) Pin. This is the inverting
input to the current sense amplifier.
RCBL (Pin 23): Cable Drop Compensation Program Pin.
A resistor RCBL tied from RCBL to ground programs cable
drop compensation by setting the USB5V input current.
RCBL can source 1mA. Excessive capacitive loading on
RCBL can degrade load transient response. Isolate load
capacitance on this pin by tying a 100k resistor between
RCBL and the capacitive load. The RCBL load monitor
output is valid when the LT8697 is enabled, otherwise the
output is zero. Float RCBL if neither the current monitor
nor the cable drop compensation feature is desired.
ICTRL (PIN 24): Current Adjustment Pin. ICTRL adjusts
the maximum VISP – VISN drop before the LT8697 limits
the output current. Connect directly to INTVCC or float for
a full scale VISP – VISN threshold of 48mV or apply values
between ground and 1V to modulate the output current
limit. There is an internal 2µA pull-up current on this pin.
Float or tie to INTVCC when unused.
GND (Exposed Pad Pin 25): Ground. The exposed pad
must be connected to the negative terminal of the input
capacitor and soldered to the PCB for proper operation
and in order to lower the thermal resistance.
PG (Pin 19): The PG pin is the open-drain output of an
internal window comparator. PG remains low until the
USB5V pin is within ±9% of the final regulation voltage
and there are no fault conditions. The PG transition delay
is approximately 40µs. PG is valid when VIN is above 3.4V
regardless of the EN/UV state.
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9
10
R4
OPT
R3
OPT
VIN
RT
CSS
CIN
For more information www.linear.com/LT8697
1
3
2
19
4
VIN
SYNC
RT
TR/SS
PG
EN/UV
5, 6
1V
SHDN
2.2µA
SHDN
TSD
INTVCC UVLO
VIN UVLO
FB
SYS
+
+
–
1V
ERROR
AMP
INTERNAL 1V REF
±9% WINDOW
COMPARATOR
+
–
–
+
VC
SLOPE COMP
25
SHDN
TSD
VIN UVLO
GND
OSCILLATOR
300kHz TO 2.2MHz
–
+
2µA
–
+
+
24
ICTRL
1V
SWITCH
LOGIC
AND
ANTISHOOT
THROUGH
20R
3.4V
REG
+
–
+
–
23
RCBL
SW
20
22
21
6
17
18
8697 BD
1M
4M
USB5V
ISN
ISP
7,8
PGND
13,14,15
RCBL
FB
R
R
M2
M1
BST
INTVCC
SYS
RCDC
RSENSE
L
CBST
CVCC
VOUT
CCDC
COUT
CABLE
RCABLE /2
RCABLE /2
–
VLOAD
+
LT8697
Block Diagram
8697p
LOAD
LT8697
Operation
The LT8697 is a monolithic, constant frequency, current
mode step-down DC/DC converter. An oscillator, with
frequency set using a resistor on the RT pin, turns on the
internal top power switch at the beginning of each clock
cycle. Current in the inductor then increases until the top
switch current comparator trips and turns off the switch.
The peak inductor current is controlled by the voltage on
the internal VC node. When the top power switch turns off,
the synchronous power switch turns on until the next clock
cycle begins or inductor current falls to zero. If overload
conditions result in more than 4.2A flowing through the
bottom switch, the next clock cycle will be delayed until
switch current returns to a safe level.
the output voltage to 5.8V. When regulation is determined
by either the output current limit or the SYS pin, USB5V
is not regulated to 5V and the output voltage falls below
its programmed value.
To control the output voltage, the LT8697’s error amplifier servos the VC node by comparing the voltage on the
USB5V pin, divided down about 5:1, with an internal 0.97V
reference. When the load current increases, it causes a
reduction in the feedback voltage relative to the reference.
This differential error makes the error amplifier raise the
VC voltage which raises the top switch peak current limit.
The feedback process continues until the average inductor current matches the new load current and the output
voltage is in regulation.
To improve efficiency across all loads, supply current
to internal circuitry is sourced from the SYS pin when
biased at 3.3V or above. Else, the internal circuitry will
draw current from VIN.
To implement cable drop compensation, the LT8697 drives
the RCBL pin to 20(VISP – VISN). Current sourced from
the RCBL pin is derived from the USB5V pin, creating an
output offset above the 5V USB5V pin voltage through RCDC
that is proportional to the load current and the RCDC/RCBL
resistor ratio. This negative output impedance compensates
for resistive drops in wiring for remote loads.
The LT8697 error amp has two additional feedback paths
that can override the USB5V pin control of the VC node.
For output current limit, the voltage VISP – VISN across
the output current sense resistor is not allowed to exceed
the lower of 48mV or VICTRL/20. Also, the SYS pin limits
If the EN/UV pin is low, the LT8697 is shut down and
draws 1μA from the input. When the EN/UV pin is above
1V, the switching regulator will become active.
The LT8697 operates in forced continuous mode (FCM) for
fast transient response and full frequency operation over
a wide load range. If a clock is applied to the SYNC pin
the part will synchronize to the external clock frequency
and operate in FCM.
When in FCM the oscillator operates continuously and
positive SW transitions are aligned to the clock. Negative
inductor current is allowed. The LT8697 can sink current
from the output and return this charge to the input in this
mode, improving load step transient response. FCM is
disabled if the VIN pin is held above 29V or if the SYS pin
is held above 7.5V. When FCM is disabled in these ways,
negative inductor current is not allowed and the LT8697
skips SW cycles in light load conditions.
Comparators monitoring the USB5V pin voltage will pull
the PG pin low if the output voltage varies more than ±9%
(typical) from the set point, or if a fault condition is present.
The oscillator reduces the LT8697’s operating frequency
when the voltage at the SYS pin is below 4V. This frequency
foldback helps to control the inductor current when the
output voltage is lower than the programmed value during
start-up or overcurrent conditions.
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LT8697
Applications Information
The LT8697 includes the necessary circuitry to implement
cable drop compensation. Cable drop compensation allows
the regulator to maintain 5V regulation on the USB VLOAD
despite high cable resistance. The LT8697 increases its
local output voltage VOUT above 5V as the load increases
to keep VLOAD regulated to 5V. This compensation does
not require running an additional pair of Kelvin sense
wires from the regulator to the load, but does require the
system designer to know the cable resistance RCABLE as
the LT8697 does not sense this value.
Program the cable drop compensation using the following ratio:
RCBL = 20.55 •
RSENSE • RCDC
RCABLE
where RCDC is a resistor tied between the regulator output
and the USB5V pin, RCBL is a resistor tied between the
RCBL pin and GND, RSENSE is the sense resistor tied between the ISP and ISN pins in series between the regulator
output and the load, and RCABLE is the cable resistance.
RSENSE is typically chosen based on the desired current
limit and is typically 20mΩ for 2.1A systems and 50mΩ
for 0.9A. Please see the Setting the Current Limit section
for more information.
The current flowing into the USB5V pin through RCDC is
identical to the current flowing out of the RCBL resistor.
While the ratio of these two resistors should be chosen
per the equation above, choose the absolute values of
these resistors to keep this current between about 30µA
and 200µA at full load current. This restriction results in
RCBL and RCDC values between 5k and 33k. If IUSB5V is
too low, capacitive loading on the USB5V and RCBL pins
will degrade the load step transient performance of the
regulator. If IUSB5V is too high, the RCBL pin will go into
current limit and the cable drop compensation feature
will not work.
Capacitance across the remote load to ground downstream
of RSENSE forms a zero in the LT8697’s feedback loop
due to cable drop compensation. CCDC reduces the cable
drop compensation gain at high frequency. The 1nF CCDC
capacitor tied across the 10k RCDC is required for stability
of the LT8697’s output. If RCDC is changed, CCDC should
also be changed to maintain roughly the same 10µs RC
time constant. If the capacitance across the remote load
is large compared to the LT8697 output capacitor tied to
the SYS pin, a longer RCDC • CCDC time constant may be
necessary for stability depending on the amount of cable
drop compensation used. Output stability should always
be verified in the end application circuit.
The LT8697 limits the maximum voltage of VOUT by
limiting the voltage on the SYS pin VSYS to 5.8V. If the
cable drop compensation is programmed to compensate
for more than 0.8V of cable drop at the maximum ILOAD,
this VSYS maximum will prevent VOUT from rising higher
and the voltage at the point of load will drop below 5V.
The following equation shows how to derive the LT8697
output voltage VOUT:
VOUT = 5V +
20.55 •ILOAD • RSENSE • RCDC
RCBL
As stated earlier, the LT8697’s cable drop compensation
feature does not allow VOUT to exceed the SYS regulation point of 5.8V. If additional impedance is placed in
between the SYS pin and the OUT node such as RSENSE
or a USB Switch, the voltage drop through these impedances at the maximum ILOAD must also be factored in to
this maximum allowable VOUT value. Refer to Figure 1
for load lines of VOUT and VLOAD to see how cable drop
compensation works.
6.0
RCABLE = 0.3Ω
RSENSE = 20mΩ
RCDC = 10kΩ
RCBL = 13.7kΩ
5.8
VOLTAGE (V)
Cable Drop Compensation
5.6
VOUT
5.4
5.2
VLOAD
5.0
4.8
0
0.5
1
2
1.5
LOAD CURRENT (A)
2.5
3
8697 F01
Figure 1. Cable Drop Compensation Load Line
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LT8697
Applications Information
Cable Drop Compensation Over a Wide Temperature
Range
Cable drop compensation with zero temperature variation
may be used in many applications. However, matching
the cable drop compensation temperature variation to the
cable resistance temperature variation may result in better overall output voltage accuracy over a wide operating
temperature range. For example, in an application with
0.26Ω of wire resistance and a maximum output current
of 2.1A, cable drop compensation adds 0.55V at 25°C to
the output at max load for a fully compensated wire resistance. If the wire in this example is copper, the copper
resistance temperature coefficient of about 4000ppm/°C
results in an output voltage error of –130mV at 85°C and
55mV at 0°C. Figure 2a shows this behavior.
5.8
5.6
ILOAD = 2.1A
CABLE = 4 METERS AWG
20 TWISTED-PAIR COPPER
VOLTAGE (V)
VOUT
5.4
5.2
VLOAD
5.0
4.8
–20
0
20
40
60
TEMPERATURE (°C)
80
100
8697 F02a
Figure 2a. Cable Drop Compensation Through 4m of AWG 20
Twisted-Pair Cable (260mΩ) without Temperature Compensation
RCBL
10k
1%
MURATA
NCP21XV103J03RA
10k THERMISTOR
10k
1%
8697 F02b
Figure 2b. RCBL Resistor Network for Matching
Copper Wire Temperature Coefficient
See Table 1 for a list of copper wire resistances vs gauge.
Table 1. Copper Wire Resistance vs Wire Gauge
AWG
RESISTANCE OF Cu WIRE AT 20°C (mΩ/m)
15
10.4
16
13.2
17
16.6
18
21.0
19
26.4
20
33.3
21
42.0
22
53.0
23
66.8
24
84.2
25
106
26
134
27
169
28
213
29
268
30
339
31
427
32
538
33
679
34
856
35
1080
36
1360
37
1720
38
2160
39
2730
40
3440
Cable drop compensation can be made to vary positively
versus temperature with the addition of a negative temperature coefficient (NTC) resistor as a part of the RCBL
resistance. This circuit idea assumes the NTC resistor is
at the same temperature as the cable. Figure 2b shows
an example resistor network for RCBL that matches copper resistance variation over a wide –40°C to 125°C
temperature range. Figure 2c shows the resultant cable
drop compensation output at several temperatures using
RCBL with negative temperature variation.
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LT8697
Applications Information
5.8
5.8
VOUT
VOUT
5.6
5.4
VOLTAGE (V)
VOLTAGE (V)
5.6
ILOAD = 2.1A
CABLE = 4 METERS AWG
20 TWISTED-PAIR COPPER
5.2
5.4
ILOAD = 2.1A
CABLE = 4 METERS AWG
20 TWISTED-PAIR COPPER
5.2
VLOAD
VLOAD
5.0
4.8
–20
5.0
0
60
20
40
TEMPERATURE (°C)
80
4.8
–20
100
0
60
20
40
TEMPERATURE (°C)
8697 F02c
Figure 2c. Cable Drop Compensation Through 4m of AWG 20
Twisted-Pair Cable (260mΩ) with Temperature Compensation
Using NTC RCBL
The NTC resistor does not give a perfectly linear transfer
function versus temperature. Here, for typical component
values, the worse case error is <10% of the cable compensation output, or <1% of the total output voltage accuracy.
Better output voltage accuracy versus temperature can be
achieved if RCBL resistor values are optimized for a narrower temperature range. Contact LTC for help designing
an RCBL resistor network.
Choosing an RSENSE resistor with a temperature coefficient
that matches the cable resistance temperature coefficient
can reduce this output voltage error overtemperature if the
sense resistor is at roughly the same ambient temperature
as RSENSE. Small value copper wire inductors can be used
in this way if the inductor resistance is well specified.
Figure 2d shows the resultant cable drop compensation
output at several temperatures using a copper RSENSE.
Use of an RSENSE that varies over temperature will make
the LT8697 output current limit vary over temperature. To
achieve the rated output current over the full operating temperature range, a higher room temperature output current
limit may be necessary. Table 2 shows the manufacturer
specified DCR of several copper wire inductors that may
be used for RSENSE.
80
100
8697 F02d
Figure 2d. Cable Drop Compensation Through 4m of AWG 20
Twisted-Pair Cable (260mΩ) with Temperature Compensation
Using Copper RSENSE
Table 2. Copper Wire Inductors for Use as Sense Resistors
VENDOR
PART NUMBER
DC RESISTANCE (mΩ)
Coilcraft
NA5931-AL
15.7 ±5%
Coilcraft
NA5932-AL
21.8 ±5%
Coilcraft
NA5933-AL
32.4 ±5%
Coilcraft
NA5934-AL
34.3 ±5%
Coilcraft
NA5935-AL
44.1 ±5%
Coilcraft
NA5936-AL
47.2 ±5%
Effect of Cable Inductance on Load Step Transient
Response
The inductance of long cabling limits the peak-to-peak
transient performance of a 2-wire sense regulator to fast
load steps. Since a 2-wire sense regulator like the LT8697
detects the output voltage at its local output and not at
the point of load, the load step response degradation due
to cable inductance is present even with cable resistance
compensation. The local regulator output capacitor and
the input capacitor of the remote load form a LC tank
circuit through the inductive cabling between them. Fast
load steps through long cabling show a large peak-to-peak
transient response and ringing at the resonant frequency of
the circuit. This ringing is a property of the LC tank circuit
and does not indicate regulator instability.
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LT8697
Applications Information
Figure 3 shows the LT8697 load step transient response
to a 50mA/µs, 0.5A load step. Two cable impedances are
compared: resistive only and then resistive plus inductive.
First, a surface mount 0.2Ω resistor is tied between the
LT8697 output and the load step generator. This resistor
stands in for a purely resistive “cable”. Second, actual AWG
20 twisted-pair cabling 3 meters long with 0.2Ω of total
resistance and about 2.3µH of inductance is connected
between the LT8697 output and the load step generator.
Even though the resistance in these two circuits is the
same, the transient load step response in the cable is
worse due to the inductance.
The degree that cable inductance degrades LT8697 load
transient response performance depends on the inductance
of the cable and on the load step rate. Long cables have
higher inductance than short cables. Cables with less
separation between supply and return conductor pairs
show lower inductance per unit length than those with
separated conductors. Faster load step rate exacerbates
the effect of inductance on load step response.
5.50
5
Since the local ground at the LT8697 is separated by
a current carrying cable from the remote ground at the
point of load, the ground reference points for these two
locations are different.
Use a differential probe across the remote output at the
end of the cable to measure output voltage at that point.
Do not simultaneously tie an oscilloscope’s probe ground
leads to both the local LT8697 ground and the remote
point of load ground. Doing so will result in high current
flow in the probe ground lines and a strange and incorrect measurement. Figure 4 shows this behavior. A 1A/µs,
0.5A load step is applied to the LT8697 output through
3 meters of AWG 20 twisted-pair cable. On one curve,
the resultant output voltage is measured correctly using
a differential probe tied across the point of load. On the
other curve, the oscilloscope ground lead is tied to the
remote ground. This poor probing causes both a DC error
due to the lower ground return resistance and an AC error
showing increased overshoot and ringing. Do not add your
oscilloscope, lab bench, and input power supply ground
lines into your measurement of the LT8697 remote output.
5.6
VLOAD
THROUGH
0.2Ω
4
5.00
VLOAD
THROUGH
0.2Ω CABLE
2
ILOAD
50mA/µs
4.50
5
VLOAD
INCORRECTLY
PROBED
5.0
4.7
1
3
VLOAD
CORRECTLY
PROBED
100µs/DIV
2
ILOAD
1A/µs
4.4
4.25
4
0
CURRENT (A)
4.75
3
VOLTAGE (V)
5.3
CURRENT (A)
VOLTAGE (V)
5.25
1
8697 F03
4.1
Figure 3. Effect of Cable Inductance on Load Step
Transient Response
100µs/DIV
0
8697 F04
Figure 4. Effect of Probing Remote Output Incorrectly
Probing a Remote Output Correctly
Reducing Output Overshoot
Take care when probing the LT8697’s remote output to
obtain correct results. The whole point of cable drop
compensation is that the local regulator output has a
different voltage than the remote output at the end of a
cable due to the cable resistance and high load current.
The same is true for the ground return line which also has
resistance and carries the same current as the output.
A consequence of the use of cable drop compensation
is that the local output voltage at the LT8697 SYS pin is
regulated to a voltage that is higher than the remote output voltage at the point of load. Several hundred mΩ of
cable resistance can separate these two outputs, so at 2A
of load current, the SYS pin voltage may be significantly
higher than the nominal 5V output at the point of load.
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15
LT8697
Applications Information
Ensure that any components tied to the LT8697 output
can withstand this increased voltage.
The LT8697 has several features designed to mitigate
any effects of higher output voltage due to cable drop
compensation. First, the LT8697 error amplifier, in addition to regulating the voltage on the USB5V pin to 5V for
the primary output, also regulates the SYS pin voltage to
less than 5.8V. For VSYS < 5.8V, the USB5V feedback input
runs the LT8697 control loop, and for VSYS > 5.8V, the
SYS feedback input runs the LT8697 control loop. This
5.8V upper limit on the maximum SYS voltage protects
components tied to the LT8697 output, such as a USB
device or a USB Switch, from an overvoltage condition,
but limits the possible amount of cable drop compensation to 0.8V.
Additionally, the LT8697 can sink current from the output
and return the charge to the input when in forced continuous mode (FCM). This feature improves the step response
for a load step from high to low. Cable drop compensation adds voltage to the output to compensate for voltage
drop across the line resistance at high load. Since most
DC/DC convertors can only source current, a load step
from high to near zero current leaves the output voltage
high and out of regulation.
VLOAD
WITHOUT
FCM
5.00
4.75
Using SYS as a Secondary Output
2
ILOAD
25mA/µs
4.50
4.25
400µs/DIV
The LT8697 has output current limit. Many USB Switches
implement current limit as well. For well controlled and
predicable behavior, ensure that only one chip sets the
output current limit, and the other chip has current limit
that exceeds the desired current limit over all operating
conditions.
6
4
VLOAD
WITH FCM
A USB or similar electronic switch can be tied between
the LT8697 output and the point of load. The switch on
resistance can be included in the cable drop compensation
calculation. Alternately, to improve load regulation, tie the
USB5V feedback input through RCDC to the output of the
USB Switch so the USB Switch impedance is removed from
the DC load response. Tie the output to the USB Switch
input. The SYS pin regulates to a maximum of 5.8V, so
the USB Switch should be chosen accordingly.
8
0
–2
8697 F05
Figure 5. Load Step Response with and
without Forced Continuous Mode
CURRENT (A)
VOLTAGE (V)
5.25
Interfacing with a USB Switch
The LT8697 has many of the features of USB Switches:
programmable output current limit, filtered fault reporting and on/off functionality. In addition, unlike many
USB Switches, the LT8697 output can survive shorts to
30V, enhancing system robustness. Therefore, in many
cases a USB Switch is not necessary and the LT8697 can
provide both the functionality of a voltage regulator and
a USB Switch.
The LT8697 fixes this problem by allowing the regulator
to sink current from the output when USB5V is too high
using FCM. Figure 5 shows the output voltage of the front
page application circuit with and without FCM.
5.50
The load step response from high current to zero without
the FCM is extremely slow and is limited by the SYS pin
bias current. However, with FCM enabled, the output slews
quickly back into regulation. If VIN is above 29V or VSYS
is above 7.5V, FCM is disabled.
For some applications, the SYS pin can be used as a secondary voltage output in addition to the primary voltage
output regulated by the USB5V pin. The SYS pin voltage
varies between 5V and 5.8V depending on the load current if cable drop compensation is used on the primary
output. A 3.3V low dropout regulator can be tied to SYS to
provide a secondary regulated output such as to power a
USB μcontroller. The SYS output will not have cable drop
compensation, but will rise above 5V depending on the
USB output load current. The load on the SYS pin should
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LT8697
Applications Information
be designed to limit load current. Also, an electronic switch
may be necessary to prevent an output overcurrent condition on the USB5V output from bringing down the SYS
output. See the Inductor Selection and Maximum Output
Current discussion below to determine how much total
load current can be drawn from the outputs for a given
LT8697 application.
above VIN = 29V the LT8697 disables forced continuous
mode so the part can pulse skip to maintain regulation at
any low VOUT to VIN ratio. For VIN < 29V, use the following
equation to find the minimum output voltage (VOUT(MIN))
where the LT8697 can regulate the output current limit:
Setting the Current Limit
where fSW is the switching frequency, tON(MIN) is the
minimum on-time, VSW(TOP) and VSW(BOT) are the internal switch drops (~0.3V and ~0.15V) respectively at
maximum load), VSENSE is voltage across the RSENSE at
the programmed output current and VL is the resistive
drop across the inductor ESR at the programmed output
current. If the calculated VOUT(MIN) is negative or is less
than the IR drop across the resistive short on the output
at the programmed current limit, then the LT8697 can
regulate the output current limit.
In addition to regulating the output voltage, the LT8697
includes a current regulation loop for setting the average
output current limit. The LT8697 measures the voltage drop
across an external current sense resistor RSENSE using
the ISP and ISN pins. This resistor should be connected
in series with the load current after the output capacitor.
The current loop modulates the cycle-by-cycle top switch
switch current limit such that the average voltage across
the ISP–ISN pins does not exceed its regulation point.
The LT8697 current limit can be programmed by forcing
a voltage on the ICTRL pin between 0V and 1V. Program
the current limit using the following equation:
ILIM =
VCTRL
RSENSE • 20.3
The preceding ILIM equation is valid for VISP – VISN <
48mV. At 48mV VSENSE, the internal current limit loop
takes over output current regulation from the ICTRL pin.
The maximum programmable output current (ILIM(MAX))
is therefore found by the following equation:
VOUT(MIN) = 0.1 • fSW • tON(MIN) • (VIN – VSW(TOP) +
VSW(BOT)) – VSW(BOT) – VSENSE – VL
In practical applications, the resistances of the cable,
inductor and sense resistor are more than adequate to
allow the LT8697 to regulate to the output current limit for
any switching frequency and input voltage. For a 400kHz
application in a worst-case condition, the programmed
output current can be regulated into VOUT = 0V for any
input voltage up to 42V. For a 2MHz application in a worstcase condition, the programmed output current can be
regulated into VOUT = 0.3V or higher. Refer to Figure 6
to see how the front page application circuit responds to
a short directly on the regulator output without a cable.
1.0
48mV
ILIM(MAX) =
RSENSE
VCTRL = OPEN, VIN = 27V
VCTRL = 0.5V, VIN = 16V
VCTRL = OPEN, VIN = 16V
VCTRL = 0.5V, VIN = 27V
The internal 2μA pull-up on the ICTRL pin allows this pin
to be floated if unused, in which case the ILIM(MAX) would
be the output current limit.
When in forced continuous mode, the LT8697’s ability to
regulate the output current is limited by its tON(MIN). In this
scenario, at very low output voltage the output current can
exceed the programmed output current limit and is limited
by the bottom switch current limit of 4.5A plus 1/2 the
ripple current. To help mitigate this effect, at low output
voltage the LT8697 folds back the switching frequency
by 10:1 to allow regulation at very low duty cycle. Also,
VOUT (V)
0.8
0.6
0.4
0.2
0
1
1.5
2
2.5
3
IOUT (A)
8697 F06
Figure 6. Output Current Regulation
vs VOUT at fSW = 2MHz, RSENSE = 18mΩ
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LT8697
Applications Information
Using RCBL as an Output Current Monitor
The primary function of the RCBL pin is to set the cable
drop compensation as discussed in the cable drop compensation section earlier. However, the RCBL pin produces
an output voltage that is proportional to the output load
current. The RCBL pin can therefore be used as an output
load monitor. The voltage on the RCBL pin obeys the following relation to USB load current:
VCBL = ILOAD • RSENSE • 20.55
VCBL is valid when the LT8697 is switching.
Since the RCBL pin current is part of the cable drop compensation control loop, excessive capacitive loading on the
RCBL pin can cause USB output voltage overshoot during
load steps. Keep the capacitive loading on the RCBL pin
below 100pF or isolate the load capacitance with 100kΩ
in series between the RCBL pin and the input it is driving,
as shown in Figure 7.
RCBL
100k
ADC
RCBL
8697 F07
Figure 7. Using the RCBL Pin as Output Current Monitor
fSW (MHz)
RT (kΩ)
0.3
140
0.4
102
0.5
80.6
0.6
66.5
0.7
56.2
0.8
47.5
1.0
37.4
1.2
30.1
1.4
25.5
1.6
21.5
1.8
18.7
2.0
16.5
2.2
14.7
Operating Frequency Selection and Trade-Offs
Selection of the operating frequency is a trade-off between
efficiency, component size, and input voltage range. The
advantage of high frequency operation is that smaller
inductor and capacitor values may be used. The disadvantages are lower efficiency and a reduced input voltage
range with constant frequency operation.
The highest switching frequency (fSW(MAX)) for a given
application can be calculated as follows:
fSW(MAX) =
Setting the Switching Frequency
The LT8697 uses a constant frequency PWM architecture
that can be programmed to switch from 300kHz to 2.2MHz
by using a resistor tied from the RT pin to ground. A table
showing the necessary RT value for a desired switching
frequency is in Table 3. The RT resistor required for a
desired switching frequency can be calculated using the
following equation:
RT =
Table 3. SW Frequency vs RT Value
43
− 5.2
fSW
where RT is in kΩ and fSW is the desired switching frequency in MHz.
(
5V + VSW(BOT)
tON(MIN) • VIN − VSW(TOP) + VSW(BOT)
)
where VIN is the typical input voltage, VSW(TOP) and
VSW(BOT) are the internal switch drops (~0.3V and ~0.15V
respectively, at maximum load) and tON(MIN) is the minimum top switch on-time (see the Electrical Characteristics
section). This equation shows that a slower switching
frequency is necessary to accommodate a high VIN/VOUT
ratio.
For transient operation, VIN may go as high as the absolute maximum rating of 42V regardless of the RT value.
However, the LT8697 will reduce switching frequency as
necessary to maintain control of inductor current to assure safe operation.
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LT8697
Applications Information
The LT8697 can operate at very high duty cycle, thus
maintaining the output voltage in regulation with the input
voltage only several hundred mV higher. This dropout voltage depends on load current and the RDS(ON) of the top
switch. However, the LT8697 skips off-times in very high
duty cycle conditions, reducing the switching frequency
below that programmed by RT. In this dropout mode, the
maximum allowable on-time is about 18µs. If this 18µs
on-time threshold is reached, the LT8697 enforces a 400ns
off-time to keep the BST capacitor charged at light loads.
This behavior limits the maximum duty cycle to 97.5%,
but guarantees good dropout performance across all loads
and any start-up condition.
For applications that cannot allow deviation from the programmed switching frequency at low VIN /VOUT ratios, use
the following formula to set switching frequency:
fSW(MAX) =
 VIN(MIN) − 5.8 − VSW(TOP) 


tOFF(MIN)  VIN(MIN) + VSW(BOT) − VSW(TOP) 
1
where VIN(MIN) is the minimum input voltage without
skipped cycles, VSW(TOP) and VSW(BOT) are the internal
switch drops (~0.3V, ~0.15V, respectively at maximum
load), fSW is the switching frequency (set by RT), and
tOFF(MIN) is the minimum switch off-time. Note that higher
switching frequency will increase the minimum input
voltage below which cycles will be dropped to achieve
higher duty cycle.
Inductor Selection and Maximum Output Current
The LT8697 is designed to minimize solution size by
allowing the inductor to be chosen based on the output
load requirements of the application. During overload or
short-circuit conditions the LT8697 safely tolerates operation with a saturated inductor through the use of a high
speed peak-current mode architecture.
A good first choice for the inductor value is as follows:
L=
5.8V + VSW(BOT)
fSW
where fSW is the switching frequency in MHz, VSW(BOT)
is the bottom switch drop (~0.15V) and L is the inductor
value in μH.
To avoid overheating and poor efficiency, an inductor must
be chosen with an RMS current rating that is greater than
the maximum expected output load of the application. In
addition, the saturation current (typically labeled ISAT) rating of the inductor must be higher than the load current
plus 1/2 of the inductor ripple current:
IL(PEAK) = IOUT(MAX) +
∆IL
2
where ΔIL is the inductor ripple current as calculated
below and IOUT(MAX) is the maximum output load for a
given application.
As a quick example, an application requiring 1A output
should use an inductor with an RMS rating of greater than
1A and an ISAT of greater than 1.3A. During long duration
overload or short-circuit conditions, the inductor RMS
current rating requirement is greater to avoid overheating of the inductor. To keep the efficiency high, the series
resistance (DCR) should be less than 0.04Ω, and the core
material should be intended for high frequency applications.
The LT8697 limits the peak switch current in order to
protect the switches and the system from overload faults.
The top switch current limit (ILIM) is at least 4.8A at low
duty cycles and decreases linearly to 4A at DC = 0.8. The
inductor value must then be sufficient to supply the desired
maximum output current (IOUT(MAX)), which is a function
of the switch current limit (ILIM) and the ripple current.
IOUT(MAX) = ILIM −
∆IL
2
The peak-to-peak ripple current in the inductor can be
calculated as follows:
∆IL =
5V
L • fSW

5V 
• 1−

 VIN(MAX) 
where fSW is the switching frequency of the LT8697 and
L is the value of the inductor. Therefore, the maximum
output current that the LT8697 will deliver depends on
the switch current limit, the inductor value, and the input
and output voltages. The inductor value may have to be
increased if the inductor ripple current does not allow
sufficient maximum output current (IOUT(MAX)) given the
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LT8697
Applications Information
switching frequency and maximum input voltage used in
the desired application.
The optimum inductor for a given application may differ
from the one indicated by this design guide. A larger value
inductor provides a higher maximum load current and
reduces the output voltage ripple. For applications requiring smaller load currents, the value of the inductor may
be lower and the LT8697 may operate with higher ripple
current. This allows use of a physically smaller inductor,
or one with a lower DCR resulting in higher efficiency.
For more information about maximum output current
and discontinuous operation, see Linear Technology’s
Application Note 44.
Finally, for duty cycles greater than 50% (VOUT/VIN >
0.5), a minimum inductance LMIN is required to avoid
sub-harmonic oscillation:
L MIN =
5.8V + VSW(BOT)
fSW
• 0.8
Input Capacitor
Bypass the input of the LT8697 circuit with a ceramic capacitor of X7R or X5R type placed as close as possible to
the VIN and PGND pins. Y5V types have poor performance
over temperature and applied voltage, and should not be
used. A 4.7μF to 10μF ceramic capacitor is adequate to
bypass the LT8697 and will easily handle the ripple current.
Note that larger input capacitance is required when a lower
switching frequency is used. If the input power source has
high impedance, or there is significant inductance due to
long wires or cables, additional bulk capacitance may be
necessary. This can be provided with a low performance
electrolytic capacitor.
Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage
ripple at the LT8697 and to force this very high frequency
switching current into a tight local loop, minimizing EMI.
A 4.7μF capacitor is capable of this task, but only if it is
placed close to the LT8697 (see the PCB Layout section).
A second precaution regarding the ceramic input capacitor
concerns the maximum input voltage rating of the LT8697.
A ceramic input capacitor combined with trace or cable
inductance forms a high quality (under damped) tank circuit. If the LT8697 circuit is plugged into a live supply, the
input voltage can ring to twice its nominal value, possibly
exceeding the LT8697’s voltage rating. This situation is
easily avoided (see Linear Technology Application Note 88).
Output Capacitor and Output Ripple
The output capacitor has two essential functions. Along
with the inductor, it filters the square wave generated
by the LT8697 to produce the DC output. In this role it
determines the output ripple, thus low impedance at the
switching frequency is important. The second function
is to store energy in order to satisfy transient loads and
stabilize the LT8697’s control loop. Ceramic capacitors
have very low equivalent series resistance (ESR) and
provide the best ripple performance. For good starting
values, see the Typical Applications section.
Use X5R or X7R types. This choice will provide low output
ripple and good transient response. Increasing the output
capacitance will also decrease the output voltage ripple. A
lower value of output capacitor can be used to save space
and cost but this may cause loop instability if the output
capacitor is too small. Since cable drop compensation
slews the voltage across the output capacitor in response
to transient load steps, a smaller output capacitor can give
faster response time. See the Typical Applications in this
data sheet for suggested capacitor values.
When choosing a capacitor, special attention should be
given to the data sheet to calculate the effective capacitance
under the relevant operating conditions of voltage bias and
temperature. A physically larger capacitor or one with a
higher voltage rating may be required.
Enable Pin
The LT8697 is in shutdown when the EN/UV pin is low and
active when the pin is high. The rising threshold of the EN
comparator is 1.0V, with 40mV of hysteresis. The EN/UV
pin can be tied to VIN if the shutdown feature is not used,
or tied to a logic level if shutdown control is required.
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LT8697
Applications Information
Adding a resistor divider from VIN to EN/UV programs the
LT8697 to regulate the output only when VIN is above a
desired voltage (see the Block Diagram). Typically, this
threshold, VIN(EN), is used in situations where the input
supply is current limited, or has a relatively high source
resistance. A switching regulator draws constant power
from the source, so source current increases as source
voltage drops. This looks like a negative resistance load
to the source and can cause the source to current limit or
latch low under low source voltage conditions. The VIN(EN)
threshold prevents the regulator from operating at source
voltages where the problems might occur. This threshold
can be adjusted by setting the values R3 and R4 such that
they satisfy the following equation:
 R3 
VIN(EN) =  + 1 • 1.0V
 R4 
pin voltage. For output tracking applications, TR/SS can
be externally driven by another voltage source. From 0V to
0.97V, the TR/SS voltage will override the internal 0.97V
reference input to the error amplifier, thus regulating the
USB5V pin voltage to 5× that of TR/SS pin. When TR/SS
is above 0.97V, tracking is disabled and USB5V will regulate
to 5V. The TR/SS pin may be left floating if the function
is not needed.
An active pull-down circuit is connected to the TR/SS pin
which will discharge the external soft-start capacitor in
the case of fault conditions and restart the ramp when the
faults are cleared. Fault conditions that clear the soft-start
capacitor are the EN/UV pin transitioning low, VIN voltage
falling too low or thermal shutdown.
Output Power Good
where the LT8697 will remain off until VIN is above VIN(EN).
Due to the comparator’s hysteresis, switching will not stop
until the input falls slightly below VIN(EN).
INTVCC Regulator
An internal low dropout (LDO) regulator produces the 3.4V
supply from VIN that powers the drivers and the internal
bias circuitry. The INTVCC can supply enough current for
the LT8697’s circuitry and must be bypassed to ground
with a minimum capacitance of 1μF. Use an X5R or an
X7R ceramic capacitor. Good bypassing is necessary to
supply the high transient currents required by the power
MOSFET gate drivers. To improve efficiency the internal
LDO can also draw current from the SYS pin when the SYS
pin is at 3.3V or higher. SYS must be tied to the LT8697
output capacitor. If the SYS pin is below 3.3V, the internal
LDO will consume current from VIN. Do not load INTVCC
with more than 100µA.
Output Voltage Tracking and Soft-Start
The LT8697 allows the user to program its output voltage
ramp rate by means of the TR/SS pin. An internal 2.2μA
pulls up the TR/SS pin to INTVCC. Putting an external capacitor on TR/SS enables soft starting the output to prevent
current surge on the input supply. During the soft-start
ramp the output voltage will proportionally track the TR/SS
When the LT8697’s output voltage is within the ±9%
window of the regulation point, which is VUSB5V in the
range of 4.55V to 5.45V (typical), the output voltage is
considered good and the open-drain PG pin goes high
impedance and is typically pulled high with an external
resistor. Otherwise, the internal pull-down device will pull
the PG pin low. To prevent glitching, both the upper and
lower thresholds include 1.3% of hysteresis.
The PG pin is also actively pulled low during several fault
conditions: EN/UV pin is below 1V, INTVCC has fallen too
low, VIN is too low, or thermal shutdown.
Synchronization
To synchronize the LT8697 oscillator to an external frequency connect a square wave (with 20% to 80% duty
cycle) to the SYNC pin. The square wave amplitude should
have valleys that are below 0.4V and peaks above 2.4V
(up to 6V).
The LT8697 may be synchronized over a 300kHz to 2.2MHz
range. The RT resistor should be chosen to set the LT8697
switching frequency equal to or below the lowest synchronization input. For example, if the synchronization signal
will be 500kHz and higher, the RT should be selected for
500kHz. The slope compensation is set by the RT value,
while the minimum slope compensation required to avoid
subharmonic oscillations is established by the inductor
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LT8697
Applications Information
size, input voltage, and output voltage. Since the synchronization frequency will not change the slopes of the
inductor current waveform, if the inductor is large enough
to avoid subharmonic oscillations at the frequency set by
RT, then the slope compensation will be sufficient for all
synchronization frequencies.
current when held above 9V. A minimum 10k RCDC resistor
must be tied from USB5V to VOUT for robust operation with
VOUT above its regulation point. The remaining pins SW,
ISP, ISN, PG and SYS tied at or near the output voltage
have at least a 30V maximum rating. The output capacitor
COUT absorbs ESD events on the LT8697 output.
Output Short Protection
If VIN is held low or floated while VOUT is held high, the
body diode of the LT8697 internal top power switch will
conduct high current from the SW pin to the VIN pin,
regardless of the state of the EN/UV pin, causing damage
to the LT8697. VOUT must remain equal to or lower than
VIN to avoid this damage to the LT8697.
The LT8697 will tolerate a shorted output. Several features
are used for protection during output short-circuit and
brownout conditions. The first is the switching frequency
will be folded back while the output is lower than the set
point to maintain inductor current control. Second, the
bottom switch current is monitored such that if inductor
current is beyond safe levels, switching of the top switch
will be delayed until the inductor current falls to safe levels.
The LT8697 withstands a short between its output and 12V
or 24V automotive battery voltage. The USB5V pin draws
PCB Layout
For proper operation and minimum EMI, care must be
taken during printed circuit board layout. Figure 8 shows
the recommended component placement with trace,
ground plane and via locations. Note that large, switched
Figure 8. Recommended PCB Layout for the LT8697
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LT8697
Applications Information
currents flow in the LT8697’s VIN pins, PGND pins, and the
input capacitors (CIN1 and CIN2). The loop formed by the
input capacitor should be as small as possible by placing
the capacitor adjacent to the VIN and PGND pins. When
using a physically large input capacitor the resulting loop
may become too large in which case using a small case/
value capacitor placed close to the VIN and PGND pins
plus a larger capacitor further away is preferred. These
components, along with the inductor and output capacitor,
should be placed on the same side of the circuit board,
and their connections should be made on that layer. Place
a local ground plane under the application circuit on the
layer closest to the surface layer. The SW and BST nodes
should be as small as possible. Finally, keep the USB5V
and RT nodes small so that the ground traces will shield
them from the SW and BST nodes. The exposed pad on
the bottom of the package must be soldered to ground so
that the pad is connected to ground electrically and also
acts as a heat sink thermally. To keep thermal resistance
low, extend the ground plane as much as possible, and
add thermal vias under and near the LT8697 to additional
ground planes within the circuit board and on the bottom
side.
High Temperature Considerations
For applications with higher ambient temperatures, lay
out the PCB to ensure good heat sinking of the LT8697.
The exposed pad on the bottom of the package must be
soldered to a ground plane. This ground should be tied
to large copper layers below with thermal vias; these
layers will spread heat dissipated by the LT8697. Placing
additional vias can reduce thermal resistance further. The
maximum load current should be derated as the ambient
temperature approaches the maximum junction rating.
Power dissipation within the LT8697 can be estimated
by calculating the total power loss from an efficiency
measurement and subtracting the inductor loss. The
die temperature is calculated by multiplying the LT8697
power dissipation by the thermal resistance from junction
to ambient. The LT8697 will stop switching and indicate
a fault condition if safe junction temperature is exceeded.
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LT8697
Typical Applications
5V Step-Down Converter with Cable Drop Compensation for
Copper Cabling Over Wide Temperature Range
CIN
10µF
CBST
0.1µF
L
10µH
BST
VIN
EN/UV
SW
PG
CVCC
1µF
CSS
10nF
RT
56.2k
RSENSE
0.02Ω
INTVCC
SYS
LT8697
ICTRL
ISP
SYNC
ISN
TR/SS
USB5V
RT
4 METERS AWG20
TWISTED PAIR CABLE
0.13Ω
+
CCDC
1nF
COUT
100µF
RCDC
10k
0.13Ω
–
VLOAD
5V
2.1A
8697 TA02
RCBL
RCBL1
10k
PGND
GND
VOUT
LOAD
VIN
RCBL2
10k
RNTC
10k
fSW = 700kHz FOR
VIN = 7V TO 27V
CIN : X7R OR X5R
COUT: 1210 CASE SIZE, X7R OR X5R
CVCC: X7R OR X5R
CBST: X7R OR X5R
L: WÜRTH 74437368100
RNTC: MURATA NCP21XV103J03RA
VIN(MAX) = 42V
VIN(MIN) = 5.8V AT 1A ILOAD
6.6V AT 2.1A ILOAD
Temperature Correction for Cable Drop Compensation
Through Copper Cabling
5.8
VOUT
VOLTAGE (V)
5.6
5.4
ILOAD = 2.1A
CABLE = 4 METERS AWG
20 TWISTED-PAIR COPPER
5.2
VLOAD
5.0
4.8
–50
0
50
100
TEMPERATURE (°C)
150
8697 TA02b
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LT8697
Typical Applications
Transient Response Through 3 Meters AWG
20 Twisted-Pair Cable
2MHz 5V Step-Down Converter with Cable Drop Compensation
BST
VIN
L
3.3µH
EN/UV
SW
CVCC
1µF
CSS
10nF
SYS
LT8697
INTVCC
ICTRL
ISP
SYNC
ISN
TR/SS
USB5V
RT
RT
16.5k
0.1Ω
CCDC
1nF
RCDC
10k
COUT
47µF
×2
–
fSW = 2MHz FOR
VIN = 8V TO 27V
RCBL
18.2k
PGND
0.1Ω
5
5.25
+
RCBL
GND
5.50
3 METERS AWG 20
TWISTED PAIR CABLE
VLOAD
5V
2.4A
5.00
4.75
2
ILOAD
50mA/µs
4.25
100µs/DIV
0
8697 TA03b
Transient Response
0.5A to 1.5A Load Step
VIN
BST
EN/UV
SW
PG
SYS
INTVCC
LT8697
ISP
SYNC
ISN
TR/SS
USB5V
fSW = 2MHz FOR
VIN = 8V TO 27V
RCBL
GND
PGND
RSENSE VOUT
0.02Ω
100k
RCBL
15.4k
CCDC
1nF
VMON
0.41V/A
CBST: X7R OR X5R
CIN : X7R OR X5R
COUT: 1210 CASE SIZE, X7R OR X5R
CVCC: X7R OR X5R
5.00
0.13Ω
+
RCDC
10k
COUT
47µF
×2
VLOAD
4 METERS AWG20
TWISTED PAIR CABLE
0.13Ω
–
VLOAD
5.1V
2.1A
8697 TA04a
R5V1
500k
4.75
1.00
ILOAD
10mA/µs
2
0.50
VMON
0.41V/A
1
0
400µs/DIV
8697 TA04b
CURRENT (A)
ICTRL
RT
5.25
CBST
0.1µF
L
3.3µH
VOLTAGE (V)
CIN
4.7µF
LOAD
VIN
CSS
10nF
RT
16.5k
1
L: COILCRAFT XAL7070-332
CBST: X7R OR X5R
VIN(MAX) = 42V
CIN: X7R OR X5R
COUT: 1210 CASE SIZE, X7R OR X5R VIN(MIN) = 5.7V AT 1A ILOAD
6.3V AT 2.4A ILOAD
CVCC: X7R OR X5R
2MHz 5.1V Step-Down Converter with Cable Drop Compensation
and Output Current Monitor
CVCC
1µF
3
VLOAD
4.50
8697 TA03a
4
VOUT
CURRENT (A)
PG
RSENSE
0.018Ω
VOUT
VOLTAGE (V)
CBST
0.1µF
CIN
4.7µF
LOAD
VIN
0
L1: SUMIDA CDRR105NP-3R3NC
VIN(MAX) = 42V
VIN(MIN) = 5.8V AT 1A ILOAD
6.5V AT 2.1ILOAD
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LT8697
Typical Applications
2.2MHz, 5.05V Step-Down Converter with Negative Output Resistance
CBST
0.1µF
CIN
4.7µF
BST
VIN
L
2.7µH
EN/UV
CVCC
1µF
CSS
10nF
SYS
LT8697
INTVCC
ICTRL
ISP
SYNC
ISN
TR/SS
USB5V
RT
RT
14.7k
fSW = 2.2MHz FOR
VIN = 8V TO 27V
RSENSE
0.018Ω
SW
PG
3 METERS AWG 20
TWISTED PAIR CABLE
0.1Ω
COUT
47µF
×2
RCDC
1nV
RCDC2
9.09k
RCBL
15.4k
PGND
–
R5VO5
100k
RCBL
GND
+
RCDC1
1k
LOAD
VIN
0.1Ω
VLOAD
5.05V
2.4A
8697 TA07a
L: COILCRAFT XAL7030-272
CBST: X7R OR X5R
VIN(MAX) = 42V
CIN: X7R OR X5R
COUT: 1210 CASE SIZE, X7R OR X5R VIN(MIN) = 5.8V AT 1A ILOAD
6.5V AT 2.4A ILOAD
CVCC: X7R OR X5R
Output Voltage vs Load Current
5.8
VOLTAGE (V)
5.6
VOUT
5.4
5.2
5.0
VLOAD
0
0.5
1.0
1.5
2.0
ILOAD (A)
2.5
3.0
8697 TA07b
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LT8697
Package Description
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
UDD Package
24-Lead Plastic QFN (3mm × 5mm)
UDD Package
(Reference LTC DWG # 05-08-1833 Rev Ø)
24-Lead Plastic QFN (3mm × 5mm)
(Reference LTC DWG # 05-08-1833 Rev Ø)
0.70 ±0.05
3.50 ±0.05
2.10 ±0.05
3.65 ±0.05
1.50 REF
1.65 ±0.05
PACKAGE OUTLINE
0.25 ±0.05
0.50 BSC
3.50 REF
4.10 ±0.05
5.50 ±0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
3.00 ±0.10
0.75 ±0.05
1.50 REF
23
R = 0.05 TYP
PIN 1 NOTCH
R = 0.20 OR 0.25
× 45° CHAMFER
24
0.40 ±0.10
PIN 1
TOP MARK
(NOTE 6)
5.00 ±0.10
1
2
3.65 ±0.10
3.50 REF
1.65 ±0.10
(UDD24) QFN 0808 REV Ø
0.200 REF
0.00 – 0.05
R = 0.115
TYP
0.25 ±0.05
0.50 BSC
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
8697p
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection
of its circuits
as described
herein will not infringe on existing patent rights.
For more
information
www.linear.com/LT8697
27
LT8697
Typical Application
High Efficiency 5.2V Step-Down Converter with
Programmable Output Current Limit and SYNC
CIN
10µF
BST
ICTRL
CVCC
1µF 100k
CSS
10nF
RT
140k
INTVCC
SYS
LT8697
PG
ISP
SYNC
ISN
TR/SS
USB5V
RT
RCBL
GND
PGND
100
95
90
RSENSE
0.018Ω
SW
EN/UV
CURRENT LIMIT
2.67A/V FOR
VCTRL = 0.2V TO 1V
300kHz TO
500kHz
VIN
CBST
0.1µF
L
22µH
COUT
100µF
VOUT
5.2V
0.5A TO 2.4A
RCDC
10k
85
80
fSW = 300kHz
RSENSE = 18mΩ
DCRL = 20mΩ
RCBL = OPEN
VOUT = 5.2V
L = 22µH SUMIDA
75
70
65
60
8697 TA06
249k
EFFICIENCY (%)
VIN
6V TO 42V
Efficiency vs Load
CIN : X7R OR X5R
COUT: 1210 CASE SIZE, X7R OR X5R
CVCC: X7R OR X5R
CBST: X7R OR X5R
L: SUMIDA CRH15D78/ANP-220MC
fSW = 300kHz TO 500kHz
FOR VIN = 6.5V TO 27V
VIN = 8V
VIN = 12V
VIN = 24V
55
50
0
0.5
1.5
2.0
1.0
LOAD CURRENT (A)
2.5
8697 TA06b
Related Parts
PART NUMBER
DESCRIPTION
COMMENTS
LT8610
42V, 2.5A, 96% Efficiency, 2.2MHz Synchronous Micropower
Step-Down DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.985V, IQ = 2.5µA,
ISD <1µA, MSOP-16E Package
LT8611
42V, 2.5A, 96% Efficiency, 2.2MHz Synchronous Micropower
Step-Down DC/DC Converter with IQ = 2.5µA and Input/Output
Current Limit/Monitor
VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.985V, IQ = 2.5µA,
ISD <1µA, 3mm × 5mm QFN-24
LT3690
36V with 60V Transient Protection, 4A, 92% Efficiency, 1.5MHz
Synchronous Micropower Step-Down DC/DC Converter with
IQ = 70µA
VIN(MIN) = 3.9V, VIN(MAX) = 36V, VOUT(MIN) = 0.985V, IQ = 70µA,
ISD <1µA, 4mm × 6mm QFN-26
LT3971A-5
38V, 1.2A, 2.2MHz High Efficiency Micropower Step-Down
DC/DC Converter with IQ = 2.8µA
VIN(MIN) = 4.2V, VIN(MIN) = 40V, VOUT(MIN) = 1.21V, IQ = 2.8µA,
ISD <1µA, MSOP-10E
LT3991
55V, 1.2A, 2.2MHz High Efficiency Micropower Step-Down
DC/DC Converter with IQ = 2.8µA
VIN(MIN) = 4.2V, VIN(MAX) = 62V, VOUT(MIN) = 1.21V, IQ = 2.8µA,
ISD <1µA, 3mm × 3mm DFN-10, MSOP-10E
LT3970
40V, 350mA, 2.2MHz High Efficiency Micropower Step-Down
DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 4.2V, VIN(MAX) = 40V, VOUT(MIN) = 1.21V, IQ = 2.5µA,
ISD <1µA, 3mm × 2mm DFN-10, MSOP-10
LT3990
62V, 350mA, 2.2MHz High Efficiency Micropower Step-Down
DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 4.2V, VIN(MAX) = 62V, VOUT(MIN) = 1.21V, IQ = 2.5µA,
ISD <1µA, 3mm × 2mm DFN-10, MSOP-10
LT3480
VIN(MIN) = 3.6V, VIN(MAX) = 36V, Transient to 60V, VOUT(MIN) =
36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High
Efficiency Step-Down DC/DC Converter with Burst Mode® Operation 0.78V, IQ = 70µA, ISD <1µA, 3mm × 3mm DFN-10, MSOP-10E
LT3980
58V with Transient Protection to 80V, 2A (IOUT), 2.4MHz, High
Efficiency Step-Down DC/DC Converter with Burst Mode Operation
VIN(MIN) = 3.6V, VIN(MAX) = 58V, Transient to 80V, VOUT(MIN) =
0.78V, IQ = 85µA, ISD <1µA, 3mm × 4mm DFN-16, MSOP-16E
8697p
28 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
For more information www.linear.com/LT8697
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com/LT8697
LT 0713 • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 2013