LINER LTC1538-AUX Dual high efficiency, low noise, synchronous step-down switching regulator Datasheet

LTC1538-AUX/LTC1539
Dual High Efficiency,
Low Noise, Synchronous
Step-Down Switching Regulators
DESCRIPTION
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FEATURES
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Maintains Constant Frequency at Low Output Currents
Dual N-Channel MOSFET Synchronous Drive
Programmable Fixed Frequency (PLL Lockable)
Wide VIN Range: 3.5V to 36V Operation
Ultrahigh Efficiency
Very Low Dropout Operation: 99% Duty Cycle
Low Dropout, 0.5A Linear Regulator for VPP
Generation or Low Noise Audio Supply
Built-In Power-On Reset Timer
Programmable Soft Start
Low-Battery Detector
Remote Output Voltage Sense
Foldback Current Limiting (Optional)
Pin Selectable Output Voltage
5V Standby Regulator Active in Shutdown: IQ < 200µA
Output Voltages from 1.19V to 9V
Available in 28- and 36-Lead SSOP Packages
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APPLICATIONS
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Notebook and Palmtop Computers, PDAs
Portable Instruments
Battery-Operated Devices
DC Power Distribution Systems
The LTC ®1538-AUX/LTC1539 are dual, synchronous stepdown switching regulator controllers which drive external
N-channel power MOSFETs in a phase-lockable fixed
frequency architecture. The Adaptive PowerTM output stage
selectively drives two N-channel MOSFETs at frequencies
up to 400kHz while reducing switching losses to maintain
high efficiencies at low output currents.
An auxiliary 0.5A linear regulator using an external PNP
pass device provides a low noise, low dropout voltage
source. A secondary winding feedback control pin (SFB1)
guarantees regulation regardless of load on the main
output by forcing continuous operation.
A 5V/20mA regulator, internal 1.19V reference and an
uncommitted comparator remain active when both controllers are shut down. A power-on reset timer (POR) is
included which generates a signal delayed by 65536/fCLK
(typ 300ms) after the controller’s output is within 5% of
the regulated first voltage. Internal resistive dividers provide pin selectable output voltages with remote sense
capability on one of the two outputs.
The operating current levels are user-programmable via
external current sense resistors. Wide input supply range
allows operation from 3.5V to 30V (36V maximum).
, LTC and LT are registered trademarks of Linear Technology Corporation.
Adaptive Power is a trademark of Linear Technology Corporation.
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TYPICAL APPLICATION
5V STANDBY
DB1, CMDSH-3
DB2, CMDSH-3
VPROG1
BOOST 1
M1
L1
10µH
M3*
CB1
0.1µF
INTVCC
VIN
TGL2
TGS1
TGS2
SW2
VOUT1
5V
3.5A
+
COUT1
220µF
10V
M1, M2, M4, M5: Si4412DY
CB2, 0.1µF
M5
LTC1539
L2
10µH
D2
MBR140T3
SENSE + 2
SENSE + 1
RSENSE1
0.03Ω
VIN
CIN 5.2V TO 28V
22µF
35V
×4
M4
M6*
BG2
BG1
M2
+
4.7µF
16V
BOOST 2
TGL1
SW1
D1
MBR140T3
+
SENSE – 2
1000pF
RSENSE2
0.03Ω
1000pF
CC1
1000pF
CC1A
220pF
VOSENSE2
SENSE – 1
ITH1
ITH2
RUN/SS1
RC1
10k
M3, M6: IRLML2803
CSS1
0.1µF
CDSC
VPROG2
SGND
COSC
56pF
*NOT REQUIRED FOR LTC1538-AUX
PGND RUN/SS2
CSS2
0.1µF
VOUT2
3.3V
3.5A
CC2
1000pF
RC2
10k
CC2A
470pF
BOLD LINES INDICATE HIGH CURRENT PATHS
+
COUT
220µF
10V
1538 F01
Figure 1. High Efficiency Dual 5V/3V Step-Down Converter
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LTC1538-AUX/LTC1539
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ABSOLUTE MAXIMUM RATINGS
Input Supply Voltage (VIN)....................... 36V to – 0.3V
Topside Driver Voltage (BOOST 1, 2) ...... 42V to – 0.3V
Peak Switch Voltage > 10µs (SW 1, 2) ... VIN + 5V to – 5V
EXTVCC Voltage........................................ 10V to – 0.3V
POR1, LBO Voltages ................................ 12V to – 0.3V
AUXFB Voltage ........................................ 20V to – 0.3V
AUXDR Voltage ........................................ 28V to – 0.3V
SENSE+ 1, SENSE+ 2, SENSE– 1, SENSE– 2,
VOSENSE2 Voltages ................... INTVCC + 0.3V to – 0.3V
VPROG1, VPROG2 Voltages .................... INTVCC to – 0.3V
PLL LPF, ITH1, ITH2 Voltages ................... 2.7V to – 0.3V
AUXON, PLLIN, SFB1,
RUN/SS1, RUN/SS2, LBI, Voltages ......... 10V to – 0.3V
Peak Output Current < 10µs (TGL1, 2, BG1, 2) ......... 2A
Peak Output Current < 10µs (TGS1, 2) .............. 250mA
INTVCC Output Current ........................................ 50mA
Operating Temperature Range
LTC1538-AUXCG/LTC1539CGW............ O°C to 70°C
LTC1538-AUXIG/LTC1539IGW .......... – 40°C to 85°C
Junction Temperature (Note 1) ............................ 125°C
Storage Temperature Range ................ – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
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PACKAGE/ORDER INFORMATION
ORDER
PART NUMBER
TOP VIEW
BOOST 1
RUN/SS1
SENSE +
1
2
RUN/SS1
1
36 PLL LPF
28 TGL1
SENSE + 1
2
35 PLLIN
27 SW1
SENSE – 1
3
34 BOOST 1
VPROG1
4
33 TGL1
1
3
26 VIN
SENSE – 1
4
25 BG1
VPROG1
ITH1
5
6
ITH1
5
32 SW1
24 INTVCC /5V
6
31 TGS1
23 PGND
COSC
7
30 VIN
SGND
8
29 BG1
LBI
9
28 INTVCC /5V
7
22 BG2
SGND
8
21 EXTVCC
9
ITH2 10
20 SW2
LBO 10
19 TGL2
SFB1 11
ITH2 12
27 PGND
26 BG2
25 EXTVCC
VOSENSE2 11
18 BOOST 2
SENSE – 2 12
17 AUXON
VPROG2 13
24 TGS2
16 AUXFB
VOSENSE2 14
23 SW2
15 AUXDR
SENSE – 2 15
22 TGL2
SENSE+ 2
21 BOOST 2
SENSE + 2
13
RUN/SS2 14
G PACKAGE
28-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 95°C/ W
Consult factory for Military grade parts.
2
LTC1538CG-AUX
LTC1538IG-AUX
POR1
COSC
SFB1
TOP VIEW
16
RUN/SS2 17
20 AUXON
AUXDR 18
19 AUXFB
GW PACKAGE
36-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 85°C/ W
ORDER
PART NUMBER
LTC1539CGW
LTC1539IGW
LTC1538-AUX/LTC1539
ELECTRICAL CHARACTERISTICS
SYMBOL
PARAMETER
Main Control Loops
IIN VOSENSE2 Feedback Current
Regulated Output Voltage
VOUT1,2
1.19V (Adjustable) Selected
3.3V Selected
5V Selected
VLINEREG1,2
Reference Voltage Line Regulation
VLOADREG1,2 Output Voltage Load Regulation
VSFB1
ISFB1
VOVL
IPROG1,2
Secondary Feedback Threshold
Secondary Feedback Current
Output Overvoltage Lockout
VPROG1,2 Input Current
Input DC Supply Current
Normal Mode
Shutdown
VRUN/SS1,2
Run Pin Threshold
IRUN/SS1,2
Soft Start Current Source
∆VSENSE(MAX) Maximum Current Sense Threshold
TGL1, 2 t r, t f TGL1, TGL2 Transition Time
Rise Time
Fall Time
TGS1, 2 t r, t f TGS1, TGS2 Transition Time
Rise Time
Fall Time
BG1, BG2 Transition Time
BG1, 2 t r, t f
Rise Time
Fall Time
Internal VCC Regulator – 5V Standby
VINTVCC
Internal VCC Voltage
VLDO INT
INTVCC Load Regulation
VLDO EXT
EXTVCC Voltage Drop
VEXTVCC
EXTVCC Switchover Voltage
Oscillator and Phase-Locked Loop
Oscillator Frequency
fOSC
VCO High
RPLLIN
PLLIN Input Resistance
Phase Detector Output Current
IPLLLPF
Sinking Capability
Sourcing Capability
Power-On Reset
POR1 Saturation Voltage
VSATPOR1
IQ
ILPOR1
VTHPOR1
POR1 Leakage
POR1 Trip Voltage
tDPOR1
POR1 Delay
TA = 25°C, VIN = 15V, VRUN/SS1,2 = 5V unless otherwise noted.
CONDITIONS
TYP
MAX
10
50
nA
1.19
3.30
5.00
0.002
0.5
– 0.5
1.19
–1
1.28
–3
3
1.202
3.380
5.100
0.01
0.8
– 0.8
1.22
–2
1.32
–6
6
V
V
V
%/V
%
%
V
µA
V
µA
µA
320
70
1.3
3
150
200
2
4.5
180
µA
µA
V
µA
mV
CLOAD = 3000pF
CLOAD = 3000pF
50
50
150
150
ns
ns
CLOAD = 500pF
CLOAD = 500pF
100
50
150
150
ns
ns
CLOAD = 3000pF
CLOAD = 3000pF
50
50
150
150
ns
ns
5.0
– 0.2
170
4.7
5.2
–1
300
V
%
mV
V
VPROG1, VPROG2 Pins Open (Note 2)
(Note 2)
VPROG1, VPROG2 Pins Open
VPROG1, VPROG2 = 0V
VPROG1, VPROG2 = INTVCC
VIN = 3.6V to 20V (Note 2), VPROG1,2 Pins Open
ITH1,2 Sinking 5µA (Note 2)
ITH1,2 Sourcing 5µA
VSFB1 Ramping Negative
VSFB1 = 1.5V
VPROG1,2 Pin Open, SENSE – 1 and VOSENSE2 Pins
0.5V > VPROG1,2
INTVCC – 0.5V < VPROG1,2 < INTVCC
EXTVCC = 5V (Note 3)
3.6V < VIN < 30V, VAUXON = 0V
VRUN/SS1,2 = 0V, 3.6V < VIN < 15V
MIN
●
●
●
●
●
●
1.16
1.24
●
VRUN/SS1,2 = 0V
VOSENSE1,2 = 0V, 5V VPROG1,2 = Pins Open
6V < VIN < 30V, VEXTVCC = 4V
INTVCC = 20mA, VEXTVCC = 4V
INTVCC = 20mA, VEXTVCC = 5V
INTVCC = 20mA, EXTVCC Ramping Positive
1.178
3.220
4.900
0.8
1.5
130
●
4.8
●
4.5
UNITS
COSC = 100pF, LTC1539: PLL LPF = 0V (Note 4)
LTC1539, VPLLLPF = 2.4V
112
200
125
240
50
138
kHz
kHz
kΩ
LTC1539
fPLLIN < fOSC
fPLLIN > fOSC
10
10
15
15
20
20
µA
µA
0.6
1
V
0.2
1
µA
– 7.5
65536
–4
%
Cycles
IPOR1 = 1.6mA, VOSENSE1 = 1V,
VPROG1 Pins Open
VPOR1 = 12V, VOSENSE1 = 1.19V, VPROG1 Pin Open
VPROG1 Pin Open % of VREF
VOSENSE1 Ramping Negative
VPROG1 Pin Open
– 11
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LTC1538-AUX/LTC1539
ELECTRICAL CHARACTERISTICS
SYMBOL
PARAMETER
Low-Battery Comparator
VSATLBO
LBO Saturation Voltage
ILLBO
LBO Leakage
VTHLB1
LBI Trip Voltage
IINLB1
LBI Input Current
VHYSLBO
LBO Hysteresis
Auxiliary Regulator/Comparator
AUXDR Current
IAUXDR
Max Current Sinking Capability
Control Current
Leakage when OFF
IINAUXFB
AUXFB Input Current
IINAUXON
AUXON Input Current
VTHAUXON
AUXON Trip Voltage
VSATAUXDR
AUXDR Saturation Voltage
VAUXFB
AUXFB Voltage
VTHAUXDR
AUXFB Divider Disconnect Voltage
TA = 25°C, VIN = 15V, VRUN/SS1,2 = 5V unless otherwise noted.
CONDITIONS
ILBO = 1.6mA, VLBI = 1.1V
VLBO = 12V, VLBI = 1.4V
High to Low Transition on LBO
VLBI = 1.19V
VEXTVCC = 0V
VAUXDR = 4V, VAUXFB = 1.0V, VAUXON = 5V
VAUXDR = 5V, VAUXFB = 1.5V, VAUXON = 5V
VAUXDR = 24V, VAUXFB = 1.5V, VAUXON = 0V
VAUXFB = 1.19V, VAUXON = 5V
VAUXON = 5V
VAUXDR = 4V, VAUXFB = 1V
IAUXDR = 1.6mA, VAUXFB = 1V, VAUXON = 5V
VAUXON = 5V, 11V < VAUXDR < 24V (Note 5)
VAUXON = 5V, 3V < VAUXDR < 7V
VAUXON = 5V (Note 5); Ramping Negative
The ● denotes specifications which apply over the full operating
temperature range.
Note 1: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formulas:
LTC1538CG-AUX: TJ = TA + (PD)(95°C/W)
LTC1539CGW: TJ = TA + (PD)(85°C/W)
Note 2: The LTC1538-AUX and LTC1539 are tested in a feedback loop
which servos VOSENSE1,2 to the balance point for the error amplifier
(VITH1,2 = 1.19V).
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MIN
TYP
MAX
UNITS
1.16
0.6
0.01
1.19
1
20
1
1
1.22
50
V
µA
V
nA
mV
●
●
●
10
1.0
●
●
11.5
1.14
7.5
15
1
0.01
0.01
0.01
1.19
0.4
12.00
1.19
8.5
5
1
1
1
1.4
0.8
12.5
1.24
9.5
mA
µA
µA
µA
µA
V
V
V
V
V
Note 3: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
Note 4: Oscillator frequency is tested by measuring the COSC charge and
discharge current (IOSC) and applying the formula:
fOSC (kHz) = 8.4(108)[COSC (pF) + 11]-1 (1/ICHG + 1/IDISC) –1
Note 5: The auxiliary regulator is tested in a feedback loop which servos
VAUXFB to the balance point for the error amplifier. For applications with
VAUXDR > 9.5V, VAUXFB uses an internal resistive divider. See Applications
Information section.
LTC1538-AUX/LTC1539
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TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Input Voltage:
VOUT = 3.3V
Efficiency vs Input Voltage:
VOUT = 5V
VOUT = 3.3V
VOUT = 5V
90
ILOAD = 1A
ILOAD = 1A
85
ILOAD = 100mA
80
EFFICIENCY (%)
90
EFFICIENCY (%)
90
ILOAD = 100mA
85
80
CONTINUOUS
MODE
85
80
Burst ModeTM
OPERATION
75
70
65
Adaptive PowerTM
MODE
60
75
75
VIN = 10V
VOUT = 5V
RSENSE = 0.33Ω
95
95
95
EFFICIENCY (%)
Efficiency vs Load Current
100
100
100
55
70
70
0
5
10
15
20
INPUT VOLTAGE (V)
25
0
30
10
15
20
INPUT VOLTAGE (V)
5
VIN – VOUT Dropout Voltage vs
Load Current
3.0
RSENSE = 0.033Ω
∆VOUT (%)
0.3
0.2
0.1
0
– 0.25
2.5
– 0.50
2.0
VITH (V)
0.4
– 0.75
1.0
1.5
2.0
LOAD CURRENT (A)
2.5
–1.25
0.5
3.0
0
0.5
1.0
1.5
2.0
LOAD CURRENT (A)
1538/39 • G04
80
5V, 3.3V OFF
5V STANDBY
1.5
60
5V, 3.3V ON
40
5V OFF, 3.3V ON
0.5
20
5V ON, 3.3V OFF
0
10
15
20
INPUT VOLTAGE (V)
25
30
5V STANDBY CURRENT (µA)
2.0
INTVCC PERCENT CHANGE, NORMALIZED (V)
100
5
0
3.0
0
10 20 30 40 50 60 70 80 90 100
OUTPUT CURRENT (%)
1538/39 • G06
INTVCC Regulation
vs INTVCC Load Current
2.5
0
2.5
CONTINUOUS/
Adaptive Power
MODE
1538/39 • G05
Input Supply Current
vs Input Voltage
1.0
Burst Mode
OPERATION
1.0
–1.50
0.5
1.5
–1.00
EXTVCC Switch Drop
vs INTVCC Load Current
300
2
EXTVCC = 0V
1
EXTVCC – INTVCC (mV)
VIN – VOUT (V)
VITH Pin Voltage vs Output Current
Load Regulation
RSENSE = 0.033Ω
VOUT DROP OF 5%
0
10
1
0.01
0.1
LOAD CURRENT (A)
1538/39 • G03
0
0.5
SUPPLY CURRENT (mA)
50
0.001
30
1538/39 • G02
1538/39 • G01
0
25
70°C
0
25°C
–1
–2
70°C
200
25°C
– 45°C
100
0
0
LTC1538/39 • TPC07
20
30
40
10
INTVCC LOAD CURRENT (mA)
50
1538/39 • G08
0
5
15
20
25
10
INTVCC LOAD CURRENT (mA)
30
1538/39 • G09
Adaptive Power and Burst Mode are trademarks of Linear Technology Corporation.
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LTC1538-AUX/LTC1539
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TYPICAL PERFORMANCE CHARACTERISTICS
Normalized Oscillator Frequency
vs Temperature
RUN/SS Pin Current vs
Temperature
10
4
5
3
SFB1 Pin Current vs
Temperature
0
fO
–5
SFB CURRENT (µA)
RUN/SS CURRENT (µA)
FREQUENCY (%)
– 0.25
2
–1.50
– 0.75
–1.00
1
–1.25
–10
– 40 –15
60
35
85
10
TEMPERATURE (°C)
110
135
0
– 40 –15
85
10
35
60
TEMPERATURE (°C)
110
135
–1.50
– 40 –15
60
35
85
10
TEMPERATURE (°C)
1538/39 • G11
1538/39 • G10
Maximum Current Comparator
Threshold Voltage vs Temp
110
135
1538/39 • G12
Transient Response
Transient Response
CURRENT SENSE THRESHOLD (mV)
154
152
VOUT
50mV/DIV
VOUT
50mV/DIV
150
148
ILOAD = 50mA to 1A
146
– 40 –15
85
10
35
60
TEMPERATURE (°C)
110
ILOAD = 1A to 3A
1538/39 • G14
1538/39 • G15
135
1538/39 • G13
Burst Mode Operation
Auxiliary Regulator Load
Regulation
Soft Start: Load Current vs Time
VOUT
20mV/DIV
RUN/SS
5V/DIV
VOUT
200mV/DIV
INDUCTOR
CURRENT
1A/DIV
ILOAD = 50mA
1538/39 • G16
1538/39 • G17
AUXILIARY OUTPUT VOLTAGE (V)
12.2
EXTERNAL PNP: 2N2907A
12.1
12.0
11.9
11.8
11.7
0
40
120
160
80
AUXILIARY LOAD CURRENT (mA)
200
1538/39 • G18
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LTC1538-AUX/LTC1539
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TYPICAL PERFORMANCE CHARACTERISTICS
Auxiliary Regulator Sink
Current Available
Auxiliary Regulator PSRR
100
20
IL = 10mA
80
15
70
PSRR (dB)
AUX DR CURRENT (mA)
90
10
IL = 100mA
60
50
40
30
5
20
10
0
0
2
4
10 12
6
8
AUX DR VOLTAGE (V)
14
0
10
16
100
FREQUENCY (kHz)
1000
1538/39 • G20
1538/39 • G19
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PIN FUNCTIONS
VIN: Main Supply Pin. Must be closely decoupled to the
IC’s signal ground pin.
INTVCC/5V STANDBY: Output of the Internal 5V Regulator
and the EXTVCC Switch. The driver and control circuits are
powered from this voltage. Must be closely decoupled to
power ground with a minimum of 2.2µF tantalum or
electrolytic capacitor. The INTVCC regulator remains on
when both RUN/SS1 and RUN/SS2 are low. Refer to the
LTC1438/LTC1439 for applications which do not require a
5V standby regulator.
EXTVCC: External Power Input to an Internal Switch. This
switch closes and supplies INTVCC, bypassing the internal
low dropout regulator whenever EXTVCC is higher than 4.8V.
Connect this pin to VOUT of the controller with the higher
output voltage. Do not exceed 10V on this pin. See EXTVCC
connection in Applications Information section.
BOOST 1, BOOST 2: Supplies to the Topside Floating Drivers.
The bootstrap capacitors are returned to these pins. Voltage
swing at these pins is from INTVCC to VIN + INTVCC.
SGND: Small Signal Ground. Common to both controllers,
must be routed separately from high current grounds to
the (–) terminals of the COUT capacitors.
PGND: Driver Power Ground. Connects to sources of
bottom N-channel MOSFETs and the (–) terminals of CIN.
SENSE – 1, SENSE – 2: Connects to the (–) input for the
current comparators. SENSE– 1 is internally connected to
the first controllers VOUT sensing point preventing true
remote output voltage sensing operation. The first controller can only be used as a 3.3V or 5.0V regulator
controlled by the VPROG1 pin. The second controller can be
set to a 3.3V, 5.0V or an adjustable regulator controlled by
the VPROG2 pin (see Table 1).
Table 1. Output Voltage Table
CONTROLLER 1
CONTROLLER 2
LTC1538-AUX
LTC1539
5V or 3.3V Only, Secondary Feedback Loop
Adjustable Only
Remote Sensing
5V/3.3V/Adjustable
Remote Sensing
POR1 Output
SW1, SW2: Switch Node Connections to Inductors. Voltage swing at these pins is from a Schottky diode (external)
voltage drop below ground to VIN.
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LTC1538-AUX/LTC1539
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PIN FUNCTIONS
SENSE + 1, SENSE + 2: The (+) Input to Each Current
Comparator. Built-in offsets between SENSE– 1 and
SENSE+ 1 pins in conjunction with RSENSE1 set the current
trip threshold (same for second controller).
VOSENSE2: Receives the remotely sensed feedback voltage for
the second controller either from the output directly or from
an external resistive divider across the output . The VPROG2
pin determines which point. VOSENSE2 must connect to.
VPROG1, VPROG2: Programs Internal Voltage Attenuators
for Output Voltage Sensing. The voltage sensing for the
first controller is internally connected to SENSE– 1 while
the VOSENSE2 pin allows for remote sensing for the second
controller. For VPROG1, VPROG2 < VINTVCC /3, the divider is
set for an output voltage of 3.3V. With V PROG1 ,
VPROG2 > VINTVCC /1.5 the divider is set for an output
voltage of 5V. Leaving VPROG2 open (DC) allows the output
voltage of the second controller to be set by an external
resistive divider connected to VOSENSE2.
COSC: External capacitor COSC from this pin to ground sets
the operating frequency.
ITH1, ITH2: Error Amplifier Compensation Point. Each associated current comparator threshold increases with this
control voltage.
RUN/SS1, RUN/SS2: Combination of Soft Start and RUN
Control Inputs. A capacitor to ground at each of these pins
sets the ramp time to full current output. The time is
approximately 0.5s/µF. Forcing either of these pins below
1.3V causes the IC to shut down the circuitry required for
that particular controller. Forcing both of these pins below
1.3V causes the device to shut down both controllers,
leaving the 5V standby regulator, internal reference and a
comparator active. Refer to the LTC1438/LTC1439 for applications which do not require a 5V standby regulator.
TGL1, TGL2: High Current Gate Drives for Main Top
N-Channel MOSFET. These are the outputs of floating
drivers with a voltage swing equal to INTVCC superimposed on the switch node voltage SW1 and SW2.
TGS1, TGS2: Gate Drives for Small Top N-Channel
MOSFET. These are the outputs of floating drivers with a
voltage swing equal to INTVCC superimposed on the
switch node voltage SW. Leaving TGS1 or TGS2 open
invokes Burst Mode operation for that controller.
8
BG1, BG2: High Current Gate Drive Outputs for Bottom NChannel MOSFETs. Voltage swing at these pins is from
ground to INTVCC.
SFB1: Secondary Winding Feedback Input. This input acts
only on the first controller and is normally connected to a
feedback resistive divider from the secondary winding.
Pulling this pin below 1.19V will force continuous synchronous operation for the first controller. This pin should
be tied to: ground to force continuous operation; INTVCC
in applications that don’t use a secondary winding; and a
resistive divider from the output in applications using a
secondary winding.
POR1: This output is a drain of an N-channel pull-down.
This pin sinks current when the output voltage of the first
controller drops 7.5% below its regulated voltage and releases 65536 oscillator cycles after the output voltage of the
first controller rises to within –5% value of its regulated value.
The POR1 output is asserted when RUN/SS1 and RUN/SS2
are both low, independent of the VOUT1.
LBO: This output is a drain of an N-channel pull-down. This
pin will sink current when the LBI pin goes below 1.19V
irrespective of the RUN/SS pin voltage.
LBI: The (+) input of a comparator which can be used as
a low-battery voltage detector irrespective of the RUN/SS
pin voltage. The (–) input is connected to the 1.19V
internal reference.
PLLIN: External Synchronizing Input to Phase Detector.
This pin is internally terminated to SGND with 50kΩ. Tie
this pin to SGND in applications which do not use the
phase-locked loop.
PLL LPF: Output of Phase Detector and Control Input of
Oscillator. Normally a series RC lowpass filter network is
connected from this pin to ground. Tie this pin to SGND in
applications which do not use the phase-locked loop. Can
be driven by a 0V to 2.4V logic signal for a frequency
shifting option.
AUXFB: Feedback Input to the Auxiliary Regulator/Comparator. When used as a linear regulator, this input can
either be connected to an external resistive divider or
directly to the collector of the external PNP pass device for
12V operation. When used as a comparator, this is the
LTC1538-AUX/LTC1539
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PIN FUNCTIONS
noninverting input of a comparator whose inverting input
is tied to the internal 1.19V reference. See Auxiliary Regulator Application section.
AUXDR: Open Drain Output of the Auxiliary Regulator/
Comparator. The base of an external PNP device is connected to this pin when used as a linear regulator. An
external pull-up resistor is required for use as a comparator. A voltage > 9.5V on AUXDR causes the internal 12V
resistive divider to be connected in series with the AUXFB pin.
AUXON: Pulling this pin high turns on the auxiliary regulator/comparator. The threshold is 1.19V. This is a convenient linear power supply logic-controlled on/off input.
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FUNCTIONAL DIAGRA
LTC1539 shown, see specific package pinout for availability of specific functions.
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2.4V
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PLLIN**
BOOST
DUPLICATE FOR SECOND CONTROLLER CHANNEL
PHASE
DETECTOR
fIN
VIN
INTVCC
DB
50k
RLP PLL LPF**
TGL
DROPOUT
DETECTOR
COSC
SFB
OSCILLATOR
COSC
CB
S Q
R Q
POR1**
VFB1
POWER-ON
RESET
LBI**
TGS**
0.6V
1.11V
BATTERY
SENSE
+
SHUTDOWN
–
LBO**
SWITCH
LOGIC
SW
•
INTVCC
–
CLP
BG
–
I1
+
+
–
COUT
INTVCC
RSENSE
I2
–
AUXDR
–
+
–
9V
CIN
PGND
+
AUXON
VSEC
•
+
+
+
8k
30k
SENSE –
VLDO
+
4k
90.8k
+
+
VFB
EA
SFB
VIN
1.19V
REF
–
1µA
gm = 1m
320k
VOSENSE*
CSEC
VPROG*
119k
VREF
+
OV
VIN
DFB†
–
4.8V
+
5V LDO
REGULATOR
EXTVCC
VOUT
61k
–
Ω
+
SFB1*
SENSE +
180k
10k
+
AUXFB
1.28V
3µA
1.19V
–
ITH
SHUTDOWN
CC
RC
RUN
SOFT START
INTVCC
+
6V
RUN/SS
SGND
CSS
INTERNAL
SUPPLY
*NOT AVAILABLE ON BOTH CHANNELS
**NOT AVAILABLE ON LTC1538-AUX
†
FOLDBACK CURRENT LIMITING OPTION
BOLD LINES INDICATE HIGH CURRENT PATHS
1438 FD
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LTC1538-AUX/LTC1539
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OPERATION (Refer to Functional Diagram)
Main Control Loop
The LTC1538-AUX/LTC1539 use a constant frequency,
current mode step-down architecture. During normal operation, the top MOSFET is turned on each cycle when the
oscillator sets the RS latch and turned off when the main
current comparator I1 resets the RS latch. The peak
inductor current at which I1 resets the RS latch is controlled by the voltage on the ITH1 (ITH2) pin, which is the
output of each error amplifier (EA). The VPROG1 pin,
described in the Pin Functions, allows the EA to receive a
selectively attenuated output feedback voltage VFB1 from
the SENSE– 1 pin while VPROG2 and VOSENSE2 allow EA to
receive an output feedback voltage VFB2 from either internal or external resistive dividers on the second controller.
When the load current increases, it causes a slight decrease in VFB relative to the 1.19V reference, which in turn
causes the ITH1 (ITH2) voltage to increase until the average
inductor current matches the new load current. After the
large top MOSFET has turned off, the bottom MOSFET is
turned on until either the inductor current starts to reverse,
as indicated by current comparator I2, or the beginning of
the next cycle.
The top MOSFET drivers are biased from floating boot
strap capacitor CB, which normally is recharged during
each Off cycle. When VIN decreases to a voltage close to
VOUT, however, the loop may enter dropout and attempt to
turn on the top MOSFET continuously. The dropout detector counts the number of oscillator cycles that the top
MOSFET remains on and periodically forces a brief off
period to allow CB to recharge.
The main control loop is shut down by pulling the RUN/
SS1 (RUN/SS2) pin low. Releasing RUN/SS1 (RUN/SS2)
allows an internal 3µA current source to charge soft start
capacitor CSS. When CSS reaches 1.3V, the main control
loop is enabled with the ITH1 (ITH2) voltage clamped at
approximately 30% of its maximum value. As CSS continues to charge, ITH1 (ITH2) is gradually released allowing
normal operation to resume. When both RUN/SS1 and
RUN/SS2 are low, all LTC1538-AUX/LTC1539 functions
are shut down except for the 5V standby regulator, internal
reference and a comparator. Refer to the LTC1438/LTC1439
for applications which do not require a 5V standby regulator.
10
Comparator OV guards against transient overshoots
> 7.5% by turning off the top MOSFET and keeping it off
until the fault is removed.
Low Current Operation
Adaptive Power Mode allows the LTC1539 to automatically change between two output stages sized for different
load currents. The TGL1 (TGL2) and BG1 (BG2) pins drive
large synchronous N-channel MOSFETs for operation at
high currents, while the TGS1 (TGS2) pin drives a much
smaller N-channel MOSFET used in conjunction with a
Schottky diode for operation at low currents. This allows
the loop to continue to operate at normal operating frequency as the load current decreases without incurring the
large MOSFET gate charge losses. If the TGS1 (TGS2) pin
is left open, the loop defaults to Burst Mode operation in
which the large MOSFETs operate intermittently based on
load demand. Adaptive Power mode provides constant
frequency operation down to approximately 1% of rated
load current. This results in an order of magnitude reduction of load current before Burst Mode operation commences. Without the small MOSFET (ie: no Adaptive
Power mode) the transition to Burst Mode operation is
approximately 10% of rated load current. The transition to
low current operation begins when comparator I2 detects
current reversal and turns off the bottom MOSFET. If the
voltage across RSENSE does not exceed the hysteresis of
I2 (approximately 20mV) for one full cycle, then on following cycles the top drive is routed to the small MOSFET at
the TGS1 (TGS2) pin and the BG1 (BG2) pin is disabled.
This continues until an inductor current peak exceeds
20mV/RSENSE or the ITH1 (ITH2) voltage exceeds 0.6V,
either of which causes drive to be returned to the TGL1
(TGL2) pin on the next cycle.
Two conditions can force continuous synchronous operation, even when the load current would otherwise dictate
low current operation. One is when the common mode
voltage of the SENSE+ 1 (SENSE+ 2) and SENSE– 1
(SENSE– 2) pins are below 1.4V, and the other is when the
SFB1 pin is below 1.19V. The latter condition is used to
assist in secondary winding regulation, as described in the
Applications Information section.
LTC1538-AUX/LTC1539
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OPERATION
(Refer to Functional Diagram)
Frequency Synchronization
A Phase-Locked Loop (PLL) is available on the LTC1539
to allow the oscillator to be synchronized to an external
source connected to the PLLIN pin. The output of the
phase detector at the PLL LPF pin is also the control input
of the oscillator, which operates over a 0V to 2.4V range
corresponding to – 30% to 30% in frequency. When
locked, the PLL aligns the turn-on of the top MOSFET to
the rising edge of the synchronizing signal. When PLLIN
is left open, PLL LPF goes low, forcing the oscillator to
minimum frequency.
Power-On Reset
The POR1 pin is an open drain output which pulls low
when the main regulator output voltage of the LTC1539
first controller is out of regulation. When the output
voltage rises to within 5% of regulation, a timer is started
which releases POR1 after 216 (65536) oscillator cycles.
Auxiliary Linear Regulator
The auxiliary linear regulator in the LTC1538-AUX and
LTC1539 controls an external PNP transistor for operation
up to 500mA. A precise internal AUXFB resistive divider is
invoked when the AUXDR pin is above 9.5V to allow
regulated 12V VPP supplies to be easily implemented.
When AUXDR is below 8.5V an external feedback divider
may be used to set other output voltages. Taking the
AUXON pin low shuts down the auxiliary regulator providing a convenient logic-controlled power supply.
The AUX block can be used as a comparator having its
inverting input tied to the internal 1.19V reference. The
AUXDR pin is used as the output and requires an external
pull-up to a supply of less than 8.5V in order to inhibit the
invoking of the internal resistive divider.
INTVCC / EXTVCC Power
Power for the top and bottom MOSFET drivers and most
of the other LTC1538-AUX/LTC1539 circuitry is derived
from the INTVCC pin. The bottom MOSFET driver supply is
also connected to INTVCC. When the EXTVCC pin is left
open, an internal 5V low dropout regulator supplies INTVCC
power. If EXTVCC is taken above 4.8V, the 5V regulator is
turned off and an internal switch is turned on to connect
EXTVCC to INTVCC. This allows the INTVCC power to be
derived from a high efficiency external source such as the
output of the regulator itself or a secondary winding, as
described in the Applications Information section.
The 5V/20mA INTVCC regulator can be used as a standby
regulator when the two controllers are in shutdown or
when either or both controllers are on. Irrespective of the
signals on the RUN/SS pins, the INTVCC pin will follow the
voltage applied to the EXTVCC pin when the voltage applied
to the EXTVCC pin is taken above 4.8V. The externally
applied voltage is required to be less than the voltage
applied to the VIN pin at all times, even when both controllers are shut down. This prevents a voltage backfeed
situation from the source applied to the EXTVCC pin to the
VIN pin. If the EXTVCC pin is tied to the first controller’s 5V
output, the nominal INTVCC pin voltage will stay in the
guaranteed range of 4.7V to 5.2V.
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APPLICATIONS INFORMATION
The basic LTC1539 application circuit is shown in Figure 1. External component selection is driven by the load
requirement and begins with the selection of RSENSE. Once
RSENSE is known, COSC and L can be chosen. Next, the
power MOSFETs and D1 are selected. Finally, CIN and COUT
are selected. The circuit shown in Figure 1 can be configured for operation up to an input voltage of 28V (limited by
the external MOSFETs).
RSENSE Selection for Output Current
RSENSE is chosen based on the required output current.
The LTC1538-AUX/LTC1539 current comparator has a
maximum threshold of 150mV/RSENSE and an input common mode range of SGND to INTVCC. The current comparator threshold sets the peak of the inductor current,
yielding a maximum average output current IMAX equal to
the peak value less half the peak-to-peak ripple current, ∆IL.
11
LTC1538-AUX/LTC1539
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APPLICATIONS INFORMATION
Allowing some margin for variations in the LTC1538-AUX/
LTC1539 and external component values yield:
100mV
IMAX
The LTC1538-AUX/LTC1539 work well with values of
RSENSE from 0.005Ω to 0.2Ω.
RSENSE =
COSC Selection for Operating Frequency
The LTC1538-AUX/LTC1539 use a constant frequency
architecture with the frequency determined by an external
oscillator capacitor on COSC. Each time the topside MOSFET
turns on, the voltage on COSC is reset to ground. During the
on-time, COSC is charged by a fixed current plus an
additional current which is proportional to the output
voltage of the phase detector (VPLLLPF)(LTC1539 only).
When the voltage on the capacitor reaches 1.19V, COSC is
reset to ground. The process then repeats.
The value of COSC is calculated from the desired operating
frequency. Assuming the phase-locked loop has no external oscillator input (VPLLLPF = 0V):
C OSC (pF ) =
( )
1.37 104
Frequency (kHz )
– 11
A graph for selecting COSC vs frequency is given in Figure
2. As the operating frequency is increased the gate charge
losses will be higher, reducing efficiency (see Efficiency
Considerations). The maximum recommended switching
frequency is 400kHz. When using Figure 2 for
synchronizable applications, choose COSC corresponding
to a frequency approximately 30% below your center
frequency. (See Phase-Locked Loop and Frequency
Sychronization).
Inductor Value Calculation
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because of
MOSFET gate charge losses. In addition to this basic trade
off, the effect of inductor value on ripple current and low
current operation must also be considered.
The inductor value has a direct effect on ripple current. The
inductor ripple current ∆IL decreases with higher inductance or frequency and generally increases with higher VIN
or VOUT:
∆IL =
 V

1
VOUT  1 – OUT 
(f)(L)
VIN 

Accepting larger values of ∆IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆IL = 0.4(IMAX). Remember, the
maximum ∆IL occurs at the maximum input voltage.
The inductor value also has an effect on low current
operation. The transition to low current operation begins
60
300
VOUT = 5.0V
VOUT = 3.3V
VOUT = 2.5V
VPLLLPF = 0V
50
INDUCTOR VALUE (µH)
COSC VALUE (pF)
250
200
150
100
20
0
0
100
200
300
400
OPERATING FREQUENCY (kHz)
500
LTC1538 • F02
Figure 2. Timing Capacitor Value
12
30
10
50
0
40
0
100
150
200
250
50
OPERATING FREQUENCY (kHz)
300
1538 F03
Figure 3. Recommended Inductor Values
LTC1538-AUX/LTC1539
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APPLICATIONS INFORMATION
when the inductor current reaches zero while the bottom
MOSFET is on. Lower inductor values (higher ∆IL) will cause
this to occur at higher load currents, which can cause a dip
in efficiency in the upper range of low current operation. In
Burst Mode operation (TGS1, 2 pins open), lower inductance
values will cause the burst frequency to decrease.
The Figure 3 graph gives a range of recommended inductor values vs operating frequency and VOUT.
Inductor Core Selection
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ® cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires more
turns of wire and therefore copper losses will increase.
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
ferrite. A reasonable compromise from the same manufacturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more
difficult. However, designs for surface mount are available
which do not increase the height significantly.
Power MOSFET and D1 Selection
Three external power MOSFETs must be selected for each
controller with the LTC1539: a pair of N-channel MOSFETs
for the top (main) switch and an N-channel MOSFET for
the bottom (synchronous) switch. Only one top MOSFET
is required for each LTC1538-AUX controller.
To take advantage of the Adaptive Power output stage, two
topside MOSFETs must be selected. A large [low RSD(ON)]
MOSFET and a small [higher RDS(ON)] MOSFET are required. The large MOSFET is used as the main switch and
works in conjunction with the synchronous switch. The
smaller MOSFET is only enabled under low load current
conditions. The benefit of this is to boost low to midcurrent
efficiencies while continuing to operate at constant frequency. Also, by using the small MOSFET the circuit will
keep switching at a constant frequency down to lower
currents and delay skipping cycles.
The RDS(ON) recommended for the small MOSFET is
around 0.5Ω. Be careful not to use a MOSFET with an
RDS(ON) that is too low; remember, we want to conserve
gate charge. (A higher RDS(ON) MOSFET has a smaller gate
capacitance and thus requires less current to charge its
gate). For all LTC1538-AUX and cost sensitive LTC1539
applications, the small MOSFET is not required. The circuit
then begins Burst Mode operation as the load current
drops.
The peak-to-peak drive levels are set by the INTVCC voltage. This voltage is typically 5V during start-up (see
EXTVCC Pin Connection). Consequently, logic level threshold MOSFETs must be used in most LTC1538-AUX/
LTC1539 applications. The only exception is applications
in which EXTVCC is powered from an external supply
greater than 8V (must be less than 10V), in which standard
threshold MOSFETs (VGS(TH) < 4V) may be used. Pay close
attention to the BVDSS specification for the MOSFETs as well;
many of the logic level MOSFETs are limited to 30V or less.
Selection criteria for the power MOSFETs include the "ON"
resistance RSD(ON), reverse transfer capacitance CRSS,
input voltage and maximum output current. When the
LTC1538-AUX/LTC1539 are operating in continuous mode
the duty cycles for the top and bottom MOSFETs are given
by:
V
Main Switch Duty Cycle = OUT
VIN
Synchronous Switch Duty Cycle =
(VIN – VOUT)
VIN
Kool Mµ is a registered trademark of Magnetics, Inc.
13
LTC1538-AUX/LTC1539
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APPLICATIONS INFORMATION
The MOSFET power dissipations at maximum output
current are given by:
V
2
PMAIN = OUT (IMAX ) (1 + δ )RDS(ON) +
VIN
k (VIN)
1.85
(IMAX)(CRSS )( f)
V –V
2
PSYNC = IN OUT (IMAX ) (1 + δ ) RDS(ON)
VIN
where δ is the temperature dependency of RDS(ON) and k
is a constant inversely related to the gate drive current.
I2R
losses while the topside
Both MOSFETs have
N-channel equation includes an additional term for transition losses, which are highest at high input voltages. For
VIN < 20V the high current efficiency generally improves
with larger MOSFETs, while for VIN > 20V the transition
losses rapidly increase to the point that the use of a higher
RDS(ON) device with lower CRSS actual provides higher
efficiency. The synchronous MOSFET losses are greatest
at high input voltage or during a short circuit when the duty
cycle in this switch is nearly 100%. Refer to the Foldback
Current Limiting section for further applications information.
The term (1 + δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs. CRSS is usually specified in the MOSFET
characteristics. The constant k = 2.5 can be used to
estimate the contributions of the two terms in the main
switch dissipation equation.
The Schottky diode D1 shown in Figure 1 serves two
purposes. During continuous synchronous operation, D1
conducts during the dead-time between the conduction of
the two large power MOSFETs. This prevents the body
diode of the bottom MOSFET from turning on and storing
charge during the dead-time, which could cost as much as
1% in efficiency. During low current operation, D1 operates in conjunction with the small top MOSFET to provide
an efficient low current output stage. A 1A Schottky is
14
generally a good compromise for both regions of operation due to the relatively small average current.
CIN and COUT Selection
In continuous mode, the source current of the top
N-channel MOSFET is a square wave of duty cycle VOUT/
VIN. To prevent large voltage transients, a low ESR input
capacitor sized for the maximum RMS current must be
used. The maximum RMS capacitor current is given by:
CIN Required IRMS ≈ IMAX
[VOUT (VIN – VOUT)]1/ 2
VIN
This formula has a maximum at VIN = 2VOUT, where IRMS =
IOUT/2. This simple worst-case condition is commonly used
for design because even significant deviations do not offer
much relief. Note that capacitor manufacturer’s ripple current
ratings are often based on only 2000 hours of life. This makes
it advisable to further derate the capacitor or to choose a
capacitor rated at a higher temperature than required. Several
capacitors may also be paralleled to meet size or height
requirements in the design. Always consult the manufacturer
if there is any question.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR requirement is satisfied the capacitance is adequate for filtering.
The output ripple (∆VOUT) is approximated by:

1 
∆VOUT ≈ ∆IL  ESR +

4 fC OUT 

where f = operating frequency, COUT = output capacitance
and ∆IL = ripple current in the inductor. The output ripple
is highest at maximum input voltage since ∆IL increases
with input voltage. With ∆IL = 0.4IOUT(MAX) the output
ripple will be less than 100mV at max VIN assuming:
COUT Required ESR < 2RSENSE
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest (ESR size)
product of any aluminum electrolytic at a somewhat
LTC1538-AUX/LTC1539
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higher price. Once the ESR requirement for COUT has been
met, the RMS current rating generally far exceeds the
IRIPPLE(P-P) requirement.
In surface mount applications multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in
surface mount configurations. In the case of tantalum, it is
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS series of surface mount tantalums, available in case
heights ranging from 2mm to 4mm. Other capacitor types
include Sanyo OS-CON, Nichicon PL series and Sprague
593D and 595D series. Consult the manufacturer for other
specific recommendations.
INTVCC / 5V Standby Regulator
An internal P-channel low dropout regulator produces 5V
at the INTVCC pin from the VIN supply pin. INTVCC powers
the drivers and internal circuitry within the LTC1538-AUX/
LTC1539, as well as any “wake-up” circuitry tied externally
to the INTVCC pin. The INTVCC pin regulator can supply
40mA and must be bypassed to ground with a minimum
of 2.2µF tantalum or low ESR electrolytic capacitor. Good
bypassing is necessary to supply the high transient currents required by the MOSFET gate drivers.
To prevent any interaction due to the high transient gate
currents being drawn from the external capacitor an
additional series filter of 10Ω and 10µF to SGND can be
added.
High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the LTC1538-AUX/
LTC1539 to be exceeded. The IC supply current is dominated by the gate charge supply current when not using an
output derived EXTVCC source. The gate charge is dependent on operating frequency as discussed in the Efficiency
Considerations section. The junction temperature can be
estimated by using the equations given in Note 1 of the
Electrical Characteristics. For example, the LTC1539 is
limited to less than 21mA from a 30V supply:
TJ = 70°C + (21mA)(30V)(85°C/W) = 124°C
To prevent maximum junction temperature from being
exceeded, the input supply current must be checked while
operating in continuous mode at maximum VIN.
EXTVCC Connection
The LTC1538-AUX/LTC1539 contain an internal P-channel MOSFET switch connected between the EXTVCC and
INTVCC pins. When the voltage applied to EXTVCC rises
above 4.8V, the internal regulator is turned off and an
internal switch closes, connecting the EXTVCC pin to the
INTVCC pin thereby supplying internal power to the IC. The
switch remains closed as long as the voltage applied to
EXTVCC remains above 4.5V. This allows the MOSFET
driver and control power to be derived from the output
during normal operation (4.8V < VOUT < 9V) and from the
internal regulator when the output is out of regulation
(start-up, short circuit). Do not apply greater than 10V to
the EXTVCC pin and ensure that EXTVCC ≤ VIN.
Significant efficiency gains can be realized by powering
INTVCC from the output, since the VIN current resulting
from the driver and control currents will be scaled by a
factor of Duty Cycle/Efficiency. For 5V regulators this
supply means connecting the EXTVCC pin directly to VOUT.
However, for 3.3V and other lower voltage regulators,
additional circuitry is required to derive INTVCC power
from the output.
The following list summarizes the four possible connections for EXTVCC:
1. EXTVCC left open (or grounded). This will cause INTVCC
to be powered from the internal 5V regulator resulting
in an efficiency penalty of up to 10% at high input
voltages.
2. EXTVCC connected directly to VOUT. This is the normal
connection for a 5V regulator and provides the highest
efficiency.
3. EXTVCC connected to an output-derived boost network.
For 3.3V and other low voltage regulators, efficiency
gains can still be realized by connecting EXTVCC to an
output-derived voltage which has been boosted to
15
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LTC1538-AUX
LTC1539*
+
+
VIN
1N4148
CIN
VIN
EXTVCC
TGL1
TGS1*
R6
SFB1
SGND
•
LTC1538-AUX
LTC1539*
+
L1
1:1
1µF
N-CH
N-CH
COUT
PGND
OPTIONAL EXTVCC
CONNECTION
5V ≤ VSEC ≤ 9V
N-CH
*TGS1 ONLY AVAILABLE ON THE LTC1539
Figure 4a. Secondary Output Loop and EXTVCC Connection
greater than 4.8V. This can be done with either the
inductive boost winding as shown in Figure 4a or the
capacitive charge pump shown in Figure 4b. The charge
pump has the advantage of simple magnetics.
4. EXTVCC connected to an external supply. If an external
supply is available in the 5V to 10V range (EXTVCC ≤ VIN)
it may be used to power EXTVCC providing it is compatible with the MOSFET gate drive requirements. When
driving standard threshold MOSFETs, the external supply must be always present during operation to prevent
MOSFET failure due to insufficient gate drive. Note: care
must be taken when using the connections in items 3 or
4. These connections will effect the INTVCC voltage
when either or both controllers are on.
External bootstrap capacitors CB connected to the BOOST
1 and BOOST 2 pins supply the gate drive voltages for the
topside MOSFETs. Capacitor CB in the Functional Diagram
is charged through diode DB from INTVCC when the
SW1(SW2) pin is low. When one of the topside MOSFETs
is to be turned on, the driver places the CB voltage across
the gate source of the desired MOSFET. This enhances the
MOSFET and turns on the topside switch. The switch node
voltage SW1(SW2) rises to VIN and the BOOST 1(BOOST
2) pin follows. With the topside MOSFET on, the boost
voltage is above the input supply: VBOOST = VIN + VINTVCC.
The value of the boost capacitor CB needs to be 100 times
that of the total input capacitance of the topside MOSFET(s).
The reverse breakdown on DB must be greater than
VIN(MAX).
16
VN2222LL
BAT85
RSENSE
VOUT
SW1
+
N-CH
COUT
PGND
1538 F04b
*TGS1 ONLY AVAILABLE ON THE LTC1539
Figure 4b. Capacitive Charge Pump for EXTVCC
Output Voltage Programming
The LTC1538-AUX/LTC1539 have pin selectable output
voltage programming. The output voltage is selected by
the VPROG1 (VPROG2) pin as follows:
VPROG1,2 = 0V
VPROG1,2 = INTVCC
VPROG2 = Open (DC)
VOUT1,2 = 3.3V
VOUT1,2 = 5V
VOUT2 = Adjustable
The top of an internal resistive divider is connected to
SENSE– 1 pin in Controller 1. For fixed output voltage
applications the SENSE– 1 pin is connected to the output
voltage as shown in Figure 5a. When using an external
resistive divider for Controller 2, the VPROG2 pin is left open
VPROG1
Topside MOSFET Driver Supply (CB,DB)
BAT85
L1
N-CH
BG1
1538 F04a
0.22µF
BAT85
CIN
TGS1*
EXTVCC
+
VIN
TGL1
VOUT
•
+
VIN
RSENSE
SW1
BG1
R5
N-CH
1µF
VSEC
–
GND: VOUT = 3.3V
INTVCC: VOUT = 5V
SENSE 1
LTC1538-AUX
LTC1539
VOUT
+
COUT
SGND
1538 F05a
Figure 5a. LTC1538-AUX/LTC1539 Fixed Output Applications
1.19V ≤ VOUT ≤ 9V
VPROG2*
R2
OPEN (DC)
VOSENSE2
LTC1538-AUX
LTC1539
SGND
*LTC1539 ONLY
R1
100pF
( )
VOUT = 1.19V 1 +
1538 F05b
R2
R1
Figure 5b. LTC1538-AUX/LTC1539 Adjustable Applications
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(DC) and the VOSENSE2 pin is connected to the feedback
resistors as shown in Figure 5b. Controller 2 will force the
externally attenuated output voltage to 1.19V.
Power-On Reset Function (POR)
The power-on reset function monitors the output voltage
of the first controller and turns on an open drain device
when it is below its properly regulated voltage. An external
pull-up resistor is required on the POR1 pin.
When power is first applied or when coming out of
shutdown, the POR1 output is held at ground. When the
output voltage rises above a level which is 5% below the final
regulated output value, an internal counter starts. After this
counter counts 216 (65536) clock cycles, the POR1 pulldown device turns off. The POR1 output is active when both
controllers are shut down as long as VIN is powered.
The POR1 output will go low whenever the output voltage
of the first controller drops below 7.5% of its regulated
value for longer than approximately 30µs, signaling an
out-of-regulation condition. In shutdown, when RUN/SS1
and RUN/SS2 are both below 1.3V, the POR1 output is
pulled low even if the regulator’s output is held up by an
external source.
RUN/ Soft Start Function
The RUN/SS1 and RUN/SS2 pins each serve two functions. Each pin provides the soft start function and a
means to shut down each controller. Soft start reduces
surge currents from VIN by providing a gradual ramp-up of
the internal current limit. Power supply sequencing can
also be accomplished using this pin.
An internal 3µA current source charges up an external
capacitor CSS. When the voltage on RUN/SS1 (RUN/SS2)
reaches 1.3V the particular controller is permitted to start
operating. As the voltage on the pin continues to ramp
from 1.3V to 2.4V, the internal current limit is also ramped
at a proportional linear rate. The current limit begins at
approximately 50mV/RSENSE (at VRUN/SS = 1.3V) and ends
at 150mV/RSENSE (VRUN/SS ≥ 2.7V). The output current
thus ramps up slowly, reducing the starting surge current
required from the input power supply. If RUN/SS has been
pulled all the way to ground there is a delay before starting
of approximately 500ms/µF, followed by a similar time to
reach full current on that controller.
By pulling both RUN/SS controller pins below 1.3V, the
LTC1538-AUX/LTC1539 are put into shutdown
(IQ < 200µA). These pins can be driven directly from logic
as shown in Figure 6. Diode D1 in Figure 6 reduces the start
3.3V
OR 5V
RUN/SS1
(RUN/SS2)
RUN/SS1
(RUN/SS2)
D1
CSS
CSS
1538 F06
Figure 6. RUN/SS Pin Interfacing
delay but allows CSS to ramp up slowly providing the soft
start function; this diode and CSS can be deleted if soft start
is not needed. Each RUN/SS pin has an internal 6V Zener
clamp (See Functional Diagram).
Foldback Current Limiting
As described in Power MOSFET and D1 Selection, the
worst-case dissipation for either MOSFET occurs with a
short-circuited output, when the synchronous MOSFET
conducts the current limit value almost continuously. In
most applications this will not cause excessive heating,
even for extended fault intervals. However, when heat
sinking is at a premium or higher RDS(ON) MOSFETs are
being used, foldback current limiting should be added to
reduce the current in proportion to the severity of the fault.
Foldback current limiting is implemented by adding diode
DFB between the output and the ITH pin as shown in the
Functional Diagram. In a hard short (VOUT = 0V) the
current will be reduced to approximately 25% of the
maximum output current. This technique may be used for
all applications with regulated output voltages of 1.8V or
greater.
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EXTERNAL
FREQUENCY
RLP
NORMALIZED FREQUENCY
2.4V
PHASE
DETECTOR*
1.3
*PLL LPF
*PLLIN
0.7
SGND
50k
0.5
1.0
1.5
VPLLLPF (V)
2.0
DIGITAL
PHASE/
FREQUENCY
DETECTOR
*LTC1539 ONLY
Figure 7. Operating Frequency vs VPLLLPF
Phase-Locked Loop and Frequency Synchronization
The LTC1539 has an internal voltage-controlled oscillator
and phase detector comprising a phase-locked loop. This
allows the top MOSFET turn-on to be locked to the rising
edge of an external source. The frequency range of the
voltage-controlled oscillator is ±30% around the center
frequency fO.
The value of COSC is calculated from the desired operating
frequency (fO). Assuming the phase-locked loop is locked
(VPLLLPF = 1.19V):
( )
4
C OSC (pF ) =
2.1 10
Frequency (kHz)
– 11
Stating the frequency as a function of VPLLLPF and COSC:
Frequency (kHz) =
OSC
2.5
1538 F07
( )
8.4 108




1
C OSC (pF ) + 11 
+ 2000
17 µA + 18µA VPLLLPF 





 2.4V 


]
The phase detector used is an edge sensitive digital type
which provides zero degrees phase shift between the
18
COSC
fO
0
[
COSC
CLP
1538 F08
Figure 8. Phase-Locked Loop Block Diagram
external and internal oscillators. This type of phase detector will not lock up on input frequencies close to the
harmonics of the VCO center frequency. The PLL hold-in
range, ∆fH, is equal to the capture range, ∆fC:
∆fH = ∆fC = ±0.3 fO.
The output of the phase detector is a complementary pair
of current sources charging or discharging the external
filter network on the PLL LPF pin. A simplified block
diagram is shown in Figure 8.
If the external frequency fPLLIN is greater than the oscillator frequency fOSC, current is sourced continuously, pulling up the PLL LPF pin. When the external frequency is less
than f0SC, current is sunk continuously, pulling down the
PLL LPF pin. If the external and internal frequencies are the
same but exhibit a phase difference, the current sources
turn on for an amount of time corresponding to the phase
difference. Thus the voltage on the PLL LPF pin is adjusted
until the phase and frequency of the external and internal
oscillators are identical. At this stable operating point the
phase comparator output is open and the filter capacitor
CLP holds the voltage. The LTC1539 PLLIN pin must be
driven from a low impedance such as a logic gate located
close to the pin. Any external attenuator used needs to be
referenced to SGND.
The loop filter components CLP, RLP smooth out the
current pulses from the phase detector and provide a
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stable input to the voltage-controlled oscillator. The filter
components CLP and RLP determine how fast the loop
acquires lock. Typically, RLP = 10k and CLP is 0.01µF to 0.1µF.
The low side of the filter needs to be connected to SGND.
The PLL LPF pin can be driven with external logic to obtain
a 1:1.9 frequency shift. The circuit shown in Figure 9 will
provide a frequency shift from fO to 1.9fO as the voltage on
VPLLLPF increases from 0V to 2.4V. Do not exceed 2.4V on
VPLLLPF.
3.3V OR 5V
2.4V
MAX
PLL LPF
18k
LTC1538 • F09
Figure 9. Directly Driving PLL LPF Pin
Low Battery Comparator
The LTC1539 has an on-chip low battery comparator
which can be used to sense a low battery condition when
implemented as shown in Figure 10. This comparator is
active during shutdown allowing battery charge level
interrogation prior to and after powering up part or all of
the system. The resistor divider R3/R4 sets the comparator trip point as follows:
 R4 
VLBITRIP = 1.19V 1 + 
 R3 
SFB1 Pin Operation
When the SFB1 pin drops below its ground referenced
1.19V threshold, continuous mode operation is forced. In
continuous mode, the large N-channel main and synchronous switches are used regardless of the load on the main
output.
In addition to providing a logic input to force continuous
synchronous operation, the SFB1 pin provides a means to
regulate a flyback winding output. The use of a synchronous switch removes the requirement that power must be
drawn from the inductor primary in order to extract power
from the auxiliary winding. With the loop in continuous
mode, the auxiliary output may be loaded without regard
to the primary output load. The SFB1 pin provides a way
to force continuous synchronous operation as needed by
the flyback winding.
The secondary output voltage is set by the turns ratio of the
transformer in conjunction with a pair of external resistors
returned to the SFB1 pin as shown in Figure 4a. The
secondary regulated voltage VSEC in Figure 4a is given by:
 R6
VSEC ≈ (N + 1)VOUT > 1.19V  1 + 
 R5
where N is the turns ratio of the transformer, and VOUT is
the main output voltage sensed by SENSE– 1.
Auxiliary Regulator/Comparator
The divided down voltage at the negative (–) input to the
comparator is compared to an internal 1.19V reference. A
20mV hysteresis is built in to assure rapid switching. The
output is an open drain MOSFET and requires a pull-up
resistor. This comparator is active when both the RUN/
SS1 and RUN/SS2 pins are low. The low side of the resistive
divider needs to be connected to SGND.
The auxiliary regulator/comparator can be used as a
comparator or low dropout regulator (by adding an external PNP pass device).
When the voltage present at the AUXON pin is greater than
1.19V the regulator/comparator is on. The amplifier is
stable when operating as a low dropout regulator. This
same amplifier can be used as a comparator whose
inverting input is tied to the 1.19V reference.
VIN
R4
LTC1539
LBI
R3
SGND
LBO
–
+
1.19V REFERENCE
Figure 10. Low Battery Comparator
1538 F10
The AUXDR pin is internally connected to an open drain
MOSFET which can sink up to 10mA. The voltage on
AUXDR determines whether or not an internal 12V resistive divider is connected to AUXFB as described below. A
pull-up resistor is required on AUXDR and the voltage
must not exceed 28V.
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With the addition of an external PNP pass device, a linear
regulator capable of supplying up to 0.5A is created. As
shown in Figure 11a, the base of the external PNP connects to the AUXDR pin together with a pull-up resistor.
The output voltage VOAUX at the collector of the external
PNP is sensed by the AUXFB pin.
The input voltage to the auxiliary regulator can be taken
from a secondary winding on the primary inductor as
shown in Figure 11a. In this application, the SFB1 pin
regulates the input voltage to the PNP regulator (see SFB1
Pin Operation) and should be set to approximately 1V to
2V above the required output voltage of the auxiliary
regulator. A Zener clamp diode may be required to keep the
secondary winding resultant output voltage under the 28V
AUXDR pin specification when the primary is heavily
loaded and the secondary is not.
The AUXFB pin is the feedback point of the regulator. An
internal resistor divider is available to provide a 12V output
by simply connecting AUXFB directly to the collector of the
external PNP. The internal resistive divider is switched in
when the voltage at AUXFB goes above 9.5V with 1V builtSECONDARY
WINDING
1:N
in hysteresis. For other output voltages, an external resistive divider is fed back to AUXFB as shown in Figure 11b.
The output voltage VOAUX is set as follows:
 R8 
VOAUX = 1.19V  1 +  < 8V AUX DR < 8.5V
 R7 
VOAUX = 12V
AUX DR ≥ 12V
When used as a voltage comparator as shown in Figure
11c, the auxiliary block has a noninverting characteristic.
When AUXFB drops below 1.19V, the AUXDR pin will be
pulled low. A minimum current of 5µA is required to pull up
the AUXDR pin to 5V when used as a comparator output in
order to counteract a 1.5µA internal pull-down current source.
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
SECONDARY
WINDING
1:N
( )
VSEC = 1.19V 1 +
R6
> 13V
R5
VSEC
+
( )
VSEC = 1.19V 1 +
R6
> (VOAUX + 1V)
R5
VSEC
VOAUX
AUXDR
R6
SFB1
AUXFB
R5 LTC1538-AUX/
LTC1539
AUXON
+
AUXDR
R6
+
VOAUX
12V
SFB1
AUXFB
R5 LTC1538-AUX/
LTC1539
AUXON
10µF
ON/OFF
R8
+
10µF
R7
ON/OFF
1538 F11a
1538 F11b
Figure 11a. 12V Output Auxiliary Regulator
Using Internal Feedback Resistors
Figure 11b. 5V Output Auxiliary Regulator Using
External Feedback Resistors
VPULL-UP < 7.5V
LTC1538-AUX/LTC1539
ON/OFF
INPUT
AUXON
AUXFB
AUXDR
–
OUTPUT
+
1.19V REFERENCE
1538 F11c
Figure 11c. Auxiliary Comparator Configuration
20
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where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1538-AUX/LTC1539 circuits. LTC1538-AUX/
LTC1539 VIN current, INTVCC current, I2R losses and
topside MOSFET transition losses.
1. The VIN current is the DC supply current given in the
Electrical Characteristics which excludes MOSFET driver
and control currents. VIN current typically results in a
small (<< 1%) loss which increases with VIN.
2. INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge dQ moves
from INTVCC to ground. The resulting dQ/dt is a current
out of INTVCC which is typically much larger than the
control circuit current. In continuous mode, IGATECHG =
f(QT + QB), where QT and QB are the gate charges of the
topside and bottom side MOSFETs. It is for this reason
that the large topside and synchronous MOSFETs are
turned off during low current operation in favor of the
small topside MOSFET and external Schottky diode,
allowing efficient, constant-frequency operation at low
output currents.
By powering EXTVCC from an output-derived source,
the additional VIN current resulting from the driver and
control currents will be scaled by a factor of Duty Cycle/
Efficiency. For example, in a 20V to 5V application,
10mA of INTVCC current results in approximately 3mA
of VIN current. This reduces the midcurrent loss from
10% or more (if the driver was powered directly from
VIN) to only a few percent.
3. I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current sense R. In continuous
mode the average output current flows through L and
RSENSE, but is “chopped” between the topside main
MOSFET and the synchronous MOSFET. If the two
MOSFETs have approximately the same RDS(ON), then
the resistance of one MOSFET can simply be summed
with the resistances of L and RSENSE to obtain I2R
losses. For example, if each RDS(ON) = 0.05Ω, RL =
0.15Ω and RSENSE = 0.05Ω, then the total resistance is
0.25Ω. This results in losses ranging from 3% to 10%
as the output current increases from 0.5A to 2A. I2R
losses cause the efficiency to roll off at high output
currents.
4. Transition losses apply only to the topside MOSFET(s)
and only when operating at high input voltages (typically
20V or greater). Transition losses can be estimated from:
Transition Loss ≈ 2.5(VIN)1.85(IMAX)(CRSS)(f)
Other losses including CIN and COUT ESR dissipative
losses, Schottky conduction losses during dead-time,
and inductor core losses, generally account for less
than 2% total additional loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in DC (resistive) load
current. When a load step occurs, VOUT shifts by an
amount equal to (∆ILOAD)(ESR) where ESR is the effective
series resistance of COUT. ∆ILOAD also begins to charge or
discharge COUT generating the feedback error signal which
forces the regulator loop to adapt to the current change
and return VOUT to its steady-state value. During this
recovery time VOUT can be monitored for overshoot or
ringing which would indicate a stability problem. The ITH
external components shown in Figure 1 will prove adequate compensation for most applications.
A second, more severe transient is caused by switching in
loads with large (> 1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive so that
the load rise time is limited to approximately (25)(CLOAD).
Thus a 10µF capacitor would require a 250µs rise time,
limiting the charging current to about 200mA.
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Automotive Considerations: Plugging into the
Cigarette Lighter
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserve or even recharge battery packs during operation.
But before you connect, be advised: you are plugging into
the supply from hell. The main battery line in an automobile is the source of a number of nasty potential transients,
including load dump, reverse battery and double battery.
Load dump is the result of a loose battery cable. When the
cable breaks connection, the field collapse in the alternator
can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse battery is
just what it says, while double battery is a consequence of
tow-truck operators finding that a 24V jump start cranks
cold engines faster than 12V.
The network shown in Figure 12 is the most straightforward approach to protect a DC/DC converter from the
ravages of an automotive battery line. The series diode
prevents current from flowing during reverse battery,
while the transient suppressor clamps the input voltage
during load dump. Note that the transient suppressor
should not conduct during double battery operation, but
must still clamp the input voltage below breakdown of the
converter. Although the LTC1538-AUX/LTC1539 has a
maximum input voltage of 36V, most applications will be
limited to 30V by the MOSFET BVDSS.
Design Example
and COSC can immediately be calculated:
RSENSE = 100mV/3A = 0.033Ω
COSC = (1.37(104 )/250) – 11 ≈ 43pF
Referring to Figure 3, a 10µH inductor falls within the
recommended range. To check the actual value of the
ripple current the following equation is used :
 V

V
∆IL = OUT  1 – OUT 
(f)(L) 
VIN 
The highest value of the ripple current occurs at the
maximum input voltage:
∆IL =
3.3V
 3.3V 
1–
 = 1.12A
250kHz(10µH) 
22V 
The power dissipation on the topside MOSFET can be
easily estimated. Using a Siliconix Si4412DY for example;
RDS(ON) = 0.042Ω, CRSS = 100pF. At maximum input
voltage with T(estimated) = 50°C:
PMAIN =
+ 2.5(22V )
1.85
]
(3A)(100pF )(250kHz) = 122mW
The most stringent requirement for the synchronous
N-channel MOSFET is with VOUT = 0V (i.e. short circuit).
During a continuous short circuit, the worst-case dissipation rises to:
PSYNC = [ISC(AVG)]2(1 + δ)RDS(ON)
As a design example, assume VIN = 12V(nominal), VIN =
22V(max), VOUT = 3.3V, IMAX = 3A and f = 250kHz, RSENSE
12V
50A IPK RATING
VIN
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
LTC1538-AUX/
LTC1539
1538 F12
Figure 12. Automotive Application Protection
22
[
3.3V 2
(3) 1+ (0.005)(50°C − 25°C ) (0.042Ω)
22V
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With the 0.033Ω sense resistor ISC(AVG) = 4A will result,
increasing the Si4412DY dissipation to 950mW at a die
temperature of 105°C.
4. Do the (+) plates of CIN connect to the drains of the
topside MOSFETs as closely as possible? This capacitor provides the AC current to the MOSFETs.
CIN will require an RMS current rating of at least 1.5A at
temperature and COUT will require an ESR of 0.03Ω for low
output ripple. The output ripple in continuous mode will be
highest at the maximum input voltage. The output voltage
ripple due to ESR is approximately:
5. Is the INTVCC decoupling capacitor connected closely
between INTVCC and the power ground pin? This capacitor carries the MOSFET driver peak currents.
VORIPPLE = RESR(∆IL) = 0.03Ω(1.12A) = 34mVP-P
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1538-AUX/LTC1539. These items are also illustrated
graphically in the layout diagram of Figure 13. Check the
following in your layout:
1. Are the high current power ground current paths using
or running through any part of signal ground? The
LTC1438/LTC1438X/LTC1439 IC’s have their sensitive
pins on one side of the package. These pins include the
signal ground for the reference, the oscillator input, the
voltage and current sensing for both controllers and the
low battery/comparator input. The signal ground area
used on this side of the IC must return to the bottom
plates of all of the output capacitors. The high current
power loops formed by the input capacitors and the
ground returns to the sources of the bottom
N-channel MOSFETs, anodes of the Schottky diodes,
and (-) plates of CIN, should be as short as possible and
tied through a low resistance path to the bottom plates
of the output capacitors for the ground return.
2. Do the LTC1538-AUX/LTC1539 SENSE– 1 and VOSENSE2
pins connect to the (+) plates of COUT? In adjustable
applications, the resistive divider R1/R2 must be connected between the (+) plate of COUT and signal ground
and the HF decoupling capacitor should be as close as
possible to the LTC1538-AUX/LTC1539.
3. Are the SENSE– and SENSE+ leads routed together with
minimum PC trace spacing? The filter capacitors between SENSE+ 1 (SENSE+ 2) and SENSE– 1 (SENSE– 2)
should be as close as possible to the LTC1538-AUX/
LTC1539.
6. Keep the switching nodes, SW1 (SW2), away from
sensitive small-signal nodes. Ideally the switch nodes
should be placed at the furthest point from the
LTC1538-AUX/LTC1539.
7. Use a low impedance source such as a logic gate to drive
the PLLIN pin and keep the lead as short as possible.
PC BOARD LAYOUT SUGGESTIONS
Switching power supply printed circuit layouts are certainly among the most difficult analog circuits to design.
The following suggestions will help to get a reasonably
close solution on the first try.
The output circuits, including the external switching
MOSFETs, inductor, secondary windings, sense resistor,
input capacitors and output capacitors all have very large
voltage and/or current levels associated with them. These
components and the radiated fields (electrostatic and/or
electromagnetic) must be kept away from the very sensitive control circuitry and loop compensation components
required for a current mode switching regulator.
The electrostatic or capacitive coupling problems can be
reduced by increasing the distance from the radiator,
typically a very large or very fast moving voltage signal.
The signal points that cause problems generally include:
the “switch” node, any secondary flyback winding voltage
and any nodes which also move with these nodes. The
switch, MOSFET gate, and boost nodes move between VIN
and Pgnd each cycle with less than a 100ns transition time.
The secondary flyback winding output has an AC signal
component of – VIN times the turns ratio of the transformer, and also has a similar < 100ns transition time. The
feedback control input signals need to have less than a few
millivolts of noise in order for the regulator to perform
properly. A rough calculation shows that 80dB of isolation
at 2MHz is required from the switch node for low noise
switcher operation. The situation is worse by a factor of the
23
LTC1538-AUX/LTC1539
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APPLICATIONS INFORMATION
turns ratio for the secondary flyback winding. Keep these
switch-node-related PC traces small and away from the
“quiet” side of the IC (not just above and below each other
on the opposite side of the board).
The electromagnetic or current-loop induced feedback
problems can be minimized by keeping the high ACcurrent (transmitter) paths and the feedback circuit (receiver) path small and/or short. Maxwell’s equations are at
work here, trying to disrupt our clean flow of current and
voltage information from the output back to the controller
input. It is crucial to understand and minimize the susceptibility of the control input stage as well as the more
obvious reduction of radiation from the high-current output stage(s). An inductive transmitter depends upon the
frequency, current amplitude and the size of the current
loop to determine the radiation characteristic of the generated field. The current levels are set in the output stage
once the input voltage, output voltage and inductor value(s)
have been selected. The frequency is set by the outputstage transition times. The only parameter over which we
have some control is the size of the antenna we create on
the PC board, i.e., the loop. A loop is formed with the input
capacitance, the top MOSFET, the Schottky diode, and the
path from the Schottky diode’s ground connection and the
input capacitor’s ground connection. A second path is
formed when a secondary winding is used comprising the
secondary output capacitor, the secondary winding and
the rectifier diode or switching MOSFET (in the case of a
synchronous approach). These “loops” should be kept as
small and tightly packed as possible in order to minimize
their “far field” radiation effects. The radiated field pro-
24
duced is picked up by the current comparator input filter
circuit(s), as well as by the voltage feedback circuit(s). The
current comparator’s filter capacitor placed across the
sense pins attenuates the radiated current signal. It is
important to place this capacitor immediately adjacent to
the IC sense pins. The voltage sensing input(s) minimizes
the inductive pickup component by using an input capacitance filter to SGND. The capacitors in both case serve to
integrate the induced current, reducing the susceptibility
to both the “loop” radiated magnetic fields and the transformer or inductor leakage fields.
The capacitor on INTVCC acts as a reservoir to supply the
high transient currents to the bottom gates and to recharge the boost capacitor. This capacitor should be a
4.7µF tantalum capacitor placed as close as possible to the
INTVCC and PGND pins of the IC. Peak current driving the
MOSFET gates exceeds 1A. The power ground pin of the
IC, connected to this capacitor, should connect directly to
the lower plates of the output capacitors to minimize the
AC ripple on the INTVCC IC power supply.
The previous instructions will yield a PC layout which has
three separate ground regions returning separately to the
bottom plates of the output capacitors: a signal ground, a
MOSFET gate/INTVCC ground and the ground from the
input capacitors, Schottky diode and synchronous
MOSFET. In practice, this may produce a long power
ground path from the input and output capacitors. A long,
low resistance path between the input and output capacitor power grounds will not upset the operation of the
switching controllers as long as the signal and power
grounds from the IC pins does not “tap in” along this path.
LTC1538-AUX/LTC1539
U
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APPLICATIONS INFORMATION
CLP
0.01µF
RLP
10k
CSS
0.1µF
1
1000pF
2
1000pF
CC1B
220pF
CC1A
1000pF
3
INTVCC
4
5
RC1
10k
COSC
100k 6
VIN
7
8
9
CC2B
470pF
10
CC2A
1000pF
RC2
10k
INT VCC
11
12
100pF
13
56pF
14
22pF
OUTPUT DIVIDER
REQUIRED WITH
VPROG OPEN
1000pF
10Ω
220pF
PLL LPF
+1
PLLIN
SENSE
SENSE – 1 BOOST 1
VPROG1
TGL1
ITH1
SW1
POR2
TGS1
LTC1539
COSC
VIN
SGND
BG1
LBI
INTVCC
LBO
PGND
SFB1
ITH2
BG2
EXTVCC
VPROG2
TGS2
VOSENSE2
SW2
15
SENSE – 2
TGL2
16
SENSE+ 2 BOOST 2
17
CSS
0.1µF
RUN/SS1
18
RUN/SS2
AUXDR
AUXON
AUXFB
36
35
EXT
CLOCK
CB1
0.1µF
34
33
M1
32
+
CIN1
L1
31
M3
RSENSE1
30
DB1
29
28
M2
+
D1
COUT1
GROUND PLANE
27
26
25
+
4.7µF
D2
M5
+
VOUT1
VIN
–
–
–
VOUT2
RSENSE2
L2
M6
+
COUT2
DB2
24
+
+
23
+
22
M4
21
20
19
CIN2
CB2
0.1µF
10Ω
1538 F13
NOT ALL PINS CONNECTED FOR CLARITY
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 13. LTC1539 Physical Layout Diagram
25
220pF
0.1µF
56pF
221k, 1%
10k
1k
220pF
392k, 1%
1000pF
1000pF
100Ω
1N4148
22pF
56pF
10Ω
1000pF
1000pF
100Ω
VPROG1
SENSE – 1
SENSE + 1
RUN/SS1
BOOST1
INTVCC
BG1
VIN
SW1
TGL1
10Ω
ITH1
PGND
LTC1538-AUX
7
COSC
BG2
8
SGND
EXTVCC
9
SFB1
SW2
10
ITH2
TGL2
11
VOSENSE2 BOOST 2
12
SENSE – 2 AUXON
13
SENSE + 2
AUXFB
14
RUN/SS2
AUXDR
6
5
4
3
2
1
15
16
17
18
19
20
21
22
23
24
25
26
27
28
5V STANDBY
VOUT1
0.1µF
0.1µF
CMDSH-3
M2
M3
0.033Ω
MBRS140T3
MBRS140T3
T1 1: 1.8
10µH
SUMIDA
CDRH125-100MC
M1B
M1A
100µF
10V, × 2
0.033Ω
MBRS1100T3
+
+
4.7µF
47k
22µF
35V
×2
100µF
10V
×2
10µF
25V
+
2N2905
R5
90.9k
1%
R6
1M
1%
VOUT3
12V
VOUT2
3.3V/3.5A
GND
VOUT1
5V/3A
VIN
5.2V TO 28V
1538 TA01
VIN 5.2 TO 28V: SWITCHING FREQUENCY = 200kHz
T1: DALE LPE6562-A262 GAPPED E-CORE OR BH ELECTRONICS #501-0657 GAPPED TOROID
M1A, M1B = SILICONIX Si4936DY
M2, M3 = SILICONIX Si4412DY
ALL INPUT AND OUTPUT CAPACITORS ARE AVX-TPS SERIES
35V
×2
+ 22µF
4.7µF 16V
CMDSH-3
+
470pF
10k
1000pF
1N4148
0.1µF
0.1µF
10Ω
+
26
+
LTC1538-AUX 5V/3A, 3.3V/3.5A, 12V/0.2A Regulator
LTC1538-AUX/LTC1539
TYPICAL APPLICATIONS
U
RC1
10k
CC2A
470pF
100pF
10Ω
D6
1N4148
1k
RC
10k
CSS2
0.1µF
100k
100k
10Ω
1000pF
LBO
100Ω
INTVCC
1000pF
100Ω
POR2
CC2
1000pF
390k, 1%
COSC
56pF
D5
1N4148
110k, 1%
CC1
1000pF
1000pF
220pF
CC1A
220pF
CSS1
0.1µF
CLP
0.01µF
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
TGL2
SW2
AUXDR
RUN/SS2
AUXFB
AUXON
SENSE+ 2 BOOST 2
SENSE – 2
VOSENSE2
TGS2
EXTVCC
BG2
PGND
INTVCC
BG1
VIN
TGS1
SW1
TGL1
BOOST 1
PLLIN
PLL LPF
LTC1539
VPROG2
ITH2
SFB1
LBO
LBI
SGND
COSC
POR2
ITH1
VPROG1
SENSE – 1
SENSE + 1
RUN/SS1
5V
STANDBY
19
20
21
22
23
24
25
26
27
28
29
30
31
32
33
34
35
36
+
M3
CIN1
22µF
35V
×2
M6
0.1µF
VOUT1
D4
CMDSH-3
4.7µF 16V
D2
CMDSH-3
0.1µF
EXT
CLOCK
M4
M5
M2
M1
RSENSE2
0.03Ω
COUT2
220µF
10V
COUT1
220µF
10V
4.7µF
25V
+
47k
Q1
MMBT
2907
CIN2
22µF
35V
×2
VOUT2
3.3V
3.5A
VOUT1
5V/3A
VOUT2
12V
200mA
* T1 = DALE LPE6562-A262 OR BH ELECTRONICS #501-0657
M1, M2, M4, M5 = IRF7403
M3, M6 = IRLML2803
L2 = SUMIDA CDRH125-100MC
ALL INPUT CAPACITORS ARE AVX-TPS SERIES
ALL OUTPUT CAPACITORS ARE AVX-TPSV LEVEL II SERIES
L2
10µH
D3
MBRS140T3
+
10µF +
25V
RSENSE1
0.03Ω
D1
MBRS140T3
T1*
1: 1.8
D7
MBRS1100T3
+
+
+
RLP
10k
LTC1539 High Efficiency Low Noise 5V/20mA Standby, 5V/3A, 3.3V/3.5A and 12V/200mA Regulator
1538 TA02
R5
90.9k
1%
R6
1M
1%
VIN
6V TO 28V
LTC1538-AUX/LTC1539
TYPICAL APPLICATIONS
27
U
CSS1
0.1µF
CC2A
470pF
RC
10k
390k, 1%
220pF
OUTPUT DIVIDER 121k
REQUIRED WITH 1%
VPROG OPEN DC
56pF
100pF
COSC
56pF
100pF
100k
100k
10Ω
CSS2
0.1µF
10Ω
1000pF
22pF
CC2
1000pF
LBO
POR2
1000pF
100Ω
* T1 = DALE LPE-6562-A214
M1, M2, M4, M5 = Si4412DY
M3, M6 = IRLML2803
L2 = SUMIDA CDRH127-100MC
INPUT CAPACITORS ARE AVX-TPS SERIES
OUTPUT CAPACITORS ARE AVX-TPSV LEVEL II SERIES
110k
1%
RC1 10k
CC1, 1000pF
110k, 1%
1000pF
CC1A, 220pF
RLP
10k
100Ω
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
AUXDR
TGL2
SW2
AUXFB
AUXON
2 BOOST 2
RUN/SS2
SENSE+
SENSE – 2
VOSENSE2
TGS2
EXTVCC
BG2
PGND
INTVCC
BG1
VIN
TGS1
SW1
TGL1
BOOST 1
PLLIN
PLL LPF
LTC1539
VPROG2
ITH2
SFB1
LBO
LBI
SGND
COSC
POR2
ITH1
VPROG1
SENSE – 1
SENSE
+1
RUN/SS1
19
20
21
22
23
24
25
26
27
28
29
30
31
32
33
34
35
36
5V STANDBY
0.1µF
10Ω
+
M3
CIN1
22µF
35V
×2
0.1µF
D4
CMDSH-3
VOUT1
M6
4.7µF 16V
D2
CMDSH-3
0.1µF
EXT
CLOCK
M4
M5
M2
M1
RSENSE1
0.025Ω
3.3µF +
35V
L2
10µH
RSENSE2
0.02Ω
D3
MBRS140T3
D1
MBRS140T3
T1*
9µH
1:3.74
MBRS1100T3
COUT2
470µF
10V
+
COUT1
330µF
10V
4.7µF
25V
+
47k
MMBT2907
ALTI
VOUT2
12V
200mA
CIN2
22µF
35V
×2
VOUT2
2.5V
5A
VOUT1
3.3V/4A
24V
+
+
28
+
CLP
0.01µF
LTC1539 High Efficiency 5V/20mA Standby, 3.3V/2.5V Regulator with Low Noise 12V Linear Regulator
1538 TA03
R5
90.9k
1%
R6
1M
1%
VIN
4V TO 28V
LTC1538-AUX/LTC1539
TYPICAL APPLICATIONS
U
R4
11.3k
1%
R3
100k
1%
R12
1k
C6,
1000pF
INTVCC
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
PLLIN
PLL LPF
TGL2
SW2
TGS2
EXTVCC
BG2
PGND
AUXDR
RUN/SS2
19
20
21
22
23
24
25
26
27
28
29
30
31
32
M1B
M1A
C1, C21
C22
22µF
35V
C17, 22pF
M4
M5
C18, 0.01µF
L2
10µH
D3
MBRS140
C28, C29
100µF
10V
R5
4.7k
C16, C19
100µF
10V
R12
0.02Ω
1W
D6
CMDSH-3
R10
D1
MBRS140
T1*
10µH
1:1.42
ALL INPUT AND OUTPUT CAPACITORS
ARE AVX-TPS SERIES
C27, 0.1µF
D4
CMDSH-3
C24, 4.7µF, 16V
D2
CMDSH-3
C20
0.1µF
+
R8, 316k,1%
5V STANDBY
(≤ 20mA)
VOUT1
C23, 0.1µF
R22
10Ω
Q1 = MOTOROLA, MMBT2907ALT1
Q2 = ZETEX, FZT849
T1 = DALE, LPE-6562-A236
L2 = SUMIDA, CDRH127-100MC
AUXFB
AUXON
SENSE+ 2 BOOST 2
SENSE 2
–
VOSENSE2
BG1
VIN
TGS1
SW1
TGL1
INTVCC
LTC1539
VPROG2
ITH2
SFB1
LBO
LBI
SGND
COSC
POR2
ITH1
VPROG1
33
34
35
36
R19, 100Ω
SENSE – 1 BOOST 1
SENSE + 1
RUN/SS1
C13, 1000pF
+
C12
6.8nF
+
+
R2
100Ω
R9
47k
C25, C26
22µF
35V
M7
C5
330µF
6.3V
R11
10Ω
Q1
2N2907
R1
27Ω
C4
3.3µF
25V
+
1538 TA04
OPTIONAL
330µF
6.3V
Q2 ZETEX
FZT849
U
VLDO
2.9V/1A
2.6A PEAK
5V/20mA
STANDBY
VOUT2
3.3V
3A
VOUT1
5V
3A
VOUT3
12V
120mA
VIN
(28V MAX)
TYPICAL APPLICATIONS
VIN 5.2V TO 28V: SWITCHING FREQUENCY = 200kHz
5V/3A, 3.3V/3A, 2.9V/1A, 2.6A PEAK, LINEAR 12V/200mA
M1 = SILICONIX, Si4936DY
M4, M5 = SILICONIX, Si4412DY
M3, M6 = IRLML2803
M7 = INTERNATIONAL RECTIFIER, IRLL014
R20
10Ω
LBO
LB1
POR2
R21
10Ω
C10, 1000pF
R7
221k
1%
C3
56pF
C2
1000pF
D7
MMBD914L
C11
0.1µF
C9
220pF
C7,
470pF
R15
10k
R13, 10k
C8
220pF
D5
MMBD914L
C14, 0.1µF
C15
1000pF
+
+
+
R18, 100Ω
LTC1539 5-Output High Efficiency Low Noise 5V/3A, 3.3V/3A, 2.9V/2.6A, 12V/200mA, 5V/20mA Notebook Computer Power Supply
(See PCB LAYOUT AND FILM for Layout of Schematic)
LTC1538-AUX/LTC1539
29
LTC1538-AUX/LTC1539
U
W
PCB LAYOUT A D FIL (Gerber files for this circuit board are available. Call the LTC factory.)
30
Silkscreen Top
Silkscreen Bottom
Copper Layer 1
Copper Layer 2 Ground Plane
Copper Layer 3
Copper Layer 4
LTC1538-AUX/LTC1539
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
G Package
28-Lead Plastic SSOP (0.209)
(LTC DWG # 05-08-1640)
0.397 – 0.407*
(10.07 – 10.33)
28 27 26 25 24 23 22 21 20 19 18 17 16 15
0.301 – 0.311
(7.65 – 7.90)
1 2 3 4 5 6 7 8 9 10 11 12 13 14
0.205 – 0.212**
(5.20 – 5.38)
0.068 – 0.078
(1.73 – 1.99)
0° – 8°
0.005 – 0.009
(0.13 – 0.22)
0.0256
(0.65)
BSC
0.022 – 0.037
(0.55 – 0.95)
0.010 – 0.015
(0.25 – 0.38)
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
0.002 – 0.008
(0.05 – 0.21)
G28 SSOP 0694
GW Package
36-Lead Plastic SSOP (Wide 0.300)
(LTC DWG # 05-08-1642)
0.602 – 0.612*
(15.290 – 15.544)
36 35 34 33 32 31 30 29 28 27 26 25 24 23 22 21 20 19
0.400 – 0.410
(10.160 – 10.414)
0.292 – 0.299**
(7.417 – 7.595)
1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18
0.097 – 0.104
(2.463 – 2.641)
0.010 – 0.016 × 45°
(0.254 – 0.406)
0.090 – 0.094
(2.286 – 2.387)
0° – 8° TYP
0.031
0.012 – 0.017
(0.800)
(0.304 – 0.431)
TYP
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
0.009 – 0.012
(0.231 – 0.305)
0.024 – 0.040
(0.610 – 1.016)
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
0.005 – 0.012
(0.127 – 0.305)
GW36 SSOP 0795
31
LTC1538-AUX/LTC1539
U
TYPICAL APPLICATION
3.3V to 2.9V at 3A Low Noise Linear Regulator
5V
6.8nF
47k
27Ω
3.3V
Q1
MMBT2907ALTI
100Ω
ZETEX
FZT849
(SURFACE MOUNT)
10Ω
2.9V
3A
AUXDR
LTC1539
22pF
316k
1%
AUXFB
2.9V
ON/OFF
AUXON
+
330µF
×2
221k
1%
1538 TA05
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1142/LTC1142HV
Dual High Efficiency Synchronous Step-Down Switching Regulators
Dual Synchronous, VIN ≤ 20V
LTC1148/LTC1148HV
High Efficiency Step-Down Switching Regulator Controllers
Synchronous, VIN ≤ 20V
LTC1159
High Efficiency Step-Down Switching Regulator Controller
Synchronous, VIN ≤ 40V, For Logic Threshold FETs
LT 1375/LT1376
1.5A, 500kHz Step-Down Switching Regulators
High Frequency, Small Inductor, High Efficiency
Switchers, 1.5A Switch
LTC1430
High Power Step-Down Switching Regulator Controller
High Efficiency 5V to 3.3V Conversion at Up to 15A
LTC1435
Single High Efficiency Low Noise Switching Regulator Controller
16-Pin Narrow SO and SSOP Packages
®
LTC1436/LTC1436-PLL/ High Efficiency Low Noise Synchronous Step-Down
LTC1437
Switching Regulator Controllers
Full-Featured Single Controllers
LTC1438
Dual, Synchronous Controller with Power-On Reset
and an Extra Comparator
Shutdown Current < 30µA
LTC1439/LTC1438X
Dual Synchronous Controller with Power-On Reset, Extra Linear
Controller, Adaptive Power, Synchronization, Auxiliary Regulator and
an Extra Uncommited Comparators
Shutdown Current < 30µA, Power-On Reset Eliminated
LT1510
Constant-Voltage/Constant-Current Battery Charger
1.3A, Li-Ion, NiCd, NiMH, Pb-Acid Charger
32
Linear Technology Corporation
LT/GP 0896 7K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● TELEX: 499-3977
 LINEAR TECHNOLOGY CORPORATION 1996
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