LMZ12003EXT 3A SIMPLE SWITCHER® Power Module with 20V Maximum Input Voltage for Military and Rugged Applications Easy To Use 7 Pin Package Performance Benefits ● Low radiated emissions / High radiated immunity ● Passes vibration standard MIL-STD-883 Method 2007.2 Condition A JESD22–B103B Condition 1 ● Passes drop standard MIL-STD-883 Method 2002.3 Condition B JESD22–B110 Condition B System Performance 30117586 TO-PMOD 7 Pin Package 10.16 x 13.77 x 4.57 mm (0.4 x 0.542 x 0.18 in) θJA = 20°C/W, θJC = 1.9°C/W RoHS Compliant Efficiency VIN = 12V VOUT = 5.0V Electrical Specifications ● ● ● ● ● 18W maximum total power output Up to 3A output current Input voltage range 4.5V to 20V Output voltage range 0.8V to 6V Efficiency up to 92% Key Features ● ● ● ● ● ● ● ● ● ● -55°C to 125°C junction temperature range Integrated shielded inductor Simple PCB layout Flexible startup sequencing using external soft-start capacitor and precision enable Protection against inrush currents and faults such as input UVLO and output short circuit Single exposed pad and standard pinout for easy mounting and manufacturing Fast transient response for FPGAs and ASICs Low output voltage ripple Pin-to-pin compatible family: LMZ14203EXT/2EXT/1EXT (42V max 3A, 2A, 1A) LMZ14203/2/1 (42V max 3A, 2A, 1A) LMZ12003/2/1 (20V max 3A, 2A, 1A) Fully Webench® Power Designer enabled 30117518 Thermal Derating Curve VIN = 12V VOUT = 5.0V 30117519 Applications ● ● ● ● Point of load conversions from 5V and 12V input rail Time critical projects Space constrained high thermal requirement applications Negative output voltage applications (See AN-2027) PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. 301175 SNVS663E Copyright © 1999-2012, Texas Instruments Incorporated LMZ12003EXT Radiated Emissions (EN 55022 Class B) from Evaluation Board 30117550 Simplified Application Schematic 30117501 Connection Diagram 30117509 Top View 7-Lead TO-PMOD Ordering Information 2 Order Number Package Type NSC Package Drawing Supplied As LMZ12003EXTTZ TO-PMOD-7 TZA07A 250 Units on Tape and Reel LMZ12003EXTTZX TO-PMOD-7 TZA07A 500 Units on Tape and Reel LMZ12003EXTTZE TO-PMOD-7 TZA07A 45 Units in a Rail Copyright © 1999-2012, Texas Instruments Incorporated LMZ12003EXT Pin Descriptions Pin Name Description 1 VIN Supply input — Nominal operating range is 4.5V to 20V . A small amount of internal capacitance is contained within the package assembly. Additional external input capacitance is required between this pin and exposed pad. 2 RON On Time Resistor — An external resistor from VIN to this pin sets the on-time of the application. Typical values range from 25k to 124k ohms. 3 EN 4 GND 5 SS Soft-Start — An internal 8 µA current source charges an external capacitor to produce the soft-start function. This node is discharged at 200 µA during disable, over-current, thermal shutdown and internal UVLO conditions. 6 FB Feedback — Internally connected to the regulation, over-voltage, and short-circuit comparators. The regulation reference point is 0.8V at this input pin. Connected the feedback resistor divider between the output and ground to set the output voltage. 7 EP Enable — Input to the precision enable comparator. Rising threshold is 1.18V nominal; 90 mV hysteresis nominal. Maximum recommended input level is 6.5V. Ground — Reference point for all stated voltages. Must be externally connected to EP. VOUT Output Voltage — Output from the internal inductor. Connect the output capacitor between this pin and exposed pad. EP Exposed Pad — Internally connected to pin 4. Used to dissipate heat from the package during operation. Must be electrically connected to pin 4 external to the package. Copyright © 1999-2012, Texas Instruments Incorporated 3 LMZ12003EXT Absolute Maximum Ratings (Note 1) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and specifications. VIN, RON to GND -0.3V to 25V EN, FB, SS to GND -0.3V to 7V Junction Temperature 150°C Storage Temperature Range -65°C to 150°C ESD Susceptibility(Note 2) ± 2 kV Peak Reflow Case Temperature 245°C (30 sec) For soldering specifications, refer to the following document: www.ti.com/lit/snoa549c Operating Ratings (Note 1) VIN EN Operation Junction Temperature 4.5V to 20V 0V to 6.5V −55°C to 125°C Electrical Characteristics Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature (TJ) range of -55°C to +125°C. Minimum and Maximum limits are guaranteed through test, design or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated the following conditions apply: VIN = 12V, Vout = 1.8V Symbol Parameter Conditions Min (Note 3) Typ (Note 4) Max (Note 3) 1.1 1.18 1.26 Units SYSTEM PARAMETERS Enable Control VEN VEN-HYS EN threshold trip point VEN rising EN threshold hysteresis VEN falling SS source current VSS = 0V 90 V mV Soft-Start ISS ISS-DIS 4.9 SS discharge current 8 11 -200 µA µA Current Limit ICL Current limit threshold d.c. average VIN= 12V to 20V 3.15 4.2 5.3 A ON/OFF Timer tON-MIN tOFF ON timer minimum pulse width 150 ns OFF timer pulse width 260 ns Regulation and Over-Voltage Comparator VFB VFB-OV In-regulation feedback voltage VSS >+ 0.8V TJ = -55°C to 125°C IO = 3A 0.773 0.793 0.813 VSS >+ 0.8V TJ = 25°C IO = 10 mA 0.784 0.800 0.816 Feedback over-voltage protection threshold V 0.92 V 5 nA IFB Feedback input bias current IQ Non Switching Input Current VFB= 0.86V 1 mA ISD Shut Down Quiescent Current VEN= 0V 25 μA Thermal Shutdown Rising 165 °C Thermal shutdown hysteresis Falling 15 °C Thermal Characteristics TSD TSD-HYST 4 Copyright © 1999-2012, Texas Instruments Incorporated LMZ12003EXT Symbol θJA θJC Min (Note 3) Typ (Note 4) Max (Note 3) Parameter Conditions Junction to Ambient 4 layer JEDEC Printed Circuit Board, 100 vias, No air flow 19.3 °C/W 2 layer JEDEC Printed Circuit Board, No air flow 21.5 °C/W No air flow 1.9 °C/W 8 mV PP Junction to Case Units PERFORMANCE PARAMETERS ΔVO Output Voltage Ripple ΔVO/ΔVIN Line Regulation VIN = 8V to 20V, IO= 3A .01 % ΔVO/ΔVIN Load Regulation VIN = 12V 1.5 mV/A η Efficiency VIN = 12V VO = 1.8V IO = 1A 87 % η Efficiency VIN = 12V VO = 1.8V IO = 3A 77 % Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics. Note 2: The human body model is a 100pF capacitor discharged through a 1.5 kΩ resistor into each pin. Test method is per JESD-22-114. Note 3: Min and Max limits are 100% production tested at 25°C. Limits over the operating temperature range are guaranteed through correlation using Statistical Quality Control (SQC) methods. Limits are used to calculate National’s Average Outgoing Quality Level (AOQL). Note 4: Typical numbers are at 25°C and represent the most likely parametric norm. Note 5: EN 55022:2006, +A1:2007, FCC Part 15 Subpart B: 2007. See AN-2024 and layout for information on device under test. Note 6: Theta JA measured on a 1.705” x 3.0” four layer board, with one ounce copper, thirty five 12 mil thermal vias, no air flow, and 1W power dissipation. Refer to PCB layout diagrams Copyright © 1999-2012, Texas Instruments Incorporated 5 LMZ12003EXT Typical Performance Characteristics Unless otherwise specified, the following conditions apply: VIN = 12V; Cin = 10uF X7R Ceramic; CO = 100uF X7R Ceramic; Tambient = 25 C for efficiency curves and waveforms. Efficiency 6V Input @ 25°C Dissipation 6V Input @ 25°C 30117521 Efficiency 12V Input @ 25°C 30117522 Dissipation 12V Input @ 25°C 30117503 Efficiency 6V Input @ 85°C Dissipation 6V Input @ 85°C 30117533 6 30117504 30117534 Copyright © 1999-2012, Texas Instruments Incorporated LMZ12003EXT Efficiency 8V Input @ 85°C Dissipation 8V Input @ 85°C 30117541 30117540 Efficiency 12V Input @ 85°C Dissipation 12V Input @ 85°C 30117542 Line and Load Regulation @ 25°C 30117548 Copyright © 1999-2012, Texas Instruments Incorporated 30117543 Line and Load Regulation @ 85C 30117556 7 LMZ12003EXT Line and Load Regulation @ –55°C Output Ripple 12VIN 3.3VO 3A 20mV/div 1μSec/div 30117505 30117557 Transient Response 12VIN 3.3VO 0.6A to 3A Step Thermal Derating VOUT = 1.8V 30117506 30117551 Current Limit 1.8VOUT @ 25°C Current Limit 1.8VOUT @ 85°C 30117558 30117560 8 Copyright © 1999-2012, Texas Instruments Incorporated LMZ12003EXT Current Limit 1.8VOUT @ -55°C 30117559 Application Block Diagram 30117508 General Description The LMZ12003EXT SIMPLE SWITCHER power module is an easy-to-use step-down DC-DC solution capable of driving up to 3A load with exceptional power conversion efficiency, line and load regulation, and output accuracy. The LMZ12003EXT is available in an innovative package that enhances thermal performance and allows for hand or machine soldering. The LMZ12003EXT can accept an input voltage rail between 4.5V and 20V and deliver an adjustable and highly accurate output voltage as low as 0.8V. The LMZ12003EXT only requires three external resistors and four external capacitors to complete the power solution. The LMZ12003EXT is a reliable and robust design with the following protection features: thermal shutdown, input under-voltage lockout, output over-voltage protection, short-circuit protection, output current limit, and allows startup into a prebiased output. A single resistor adjusts the switching frequency up to 1 MHz. COT Control Circuit Overview Constant On Time control is based on a comparator and an on-time one shot, with the output voltage feedback compared with an internal 0.8V reference. If the feedback voltage is below the reference, the main MOSFET is turned on for a fixed on-time determined by a programming resistor RON. RON is connected to VIN such that on-time is reduced with increasing input supply voltage. Following this on-time, the main MOSFET remains off for a minimum of 260 ns. If the voltage on the feedback pin falls below the reference level again the on-time cycle is repeated. Regulation is achieved in this manner. Copyright © 1999-2012, Texas Instruments Incorporated 9 LMZ12003EXT Design Steps for the LMZ12003EXT Application The LMZ12003EXT is fully supported by Webench® and offers the following: Component selection, electrical and thermal simulations as well as the build-it board for a reduction in design time. The following list of steps can be used to manually design the LMZ12003EXT application. • Select minimum operating VIN with enable divider resistors • Program VO with divider resistor selection • Program turn-on time with soft-start capacitor selection • Select CO • Select CIN • Set operating frequency with RON • Determine module dissipation • Layout PCB for required thermal performance ENABLE DIVIDER, RENT AND RENB SELECTION The enable input provides a precise 1.18V band-gap rising threshold to allow direct logic drive or connection to a voltage divider from a higher enable voltage such as Vin. The enable input also incorporates 90 mV (typ) of hysteresis resulting in a falling threshold of 1.09V. The maximum recommended voltage into the EN pin is 6.5V. For applications where the midpoint of the enable divider exceeds 6.5V, a small zener can be added to limit this voltage. The function of this resistive divider is to allow the designer to choose an input voltage below which the circuit will be disabled. This implements the feature of programmable under voltage lockout. This is often used in battery powered systems to prevent deep discharge of the system battery. It is also useful in system designs for sequencing of output rails or to prevent early turn-on of the supply as the main input voltage rail rises at power-up. Applying the enable divider to the main input rail is often done in the case of higher input voltage systems where a lower boundary of operation should be established. In the case of sequencing supplies, the divider is connected to a rail that becomes active earlier in the power-up cycle than the LMZ12003EXT output rail. The two resistors should be chosen based on the following ratio: RENT / RENB = (VIN UVLO / 1.18V) – 1 (1) The LMZ12003EXT demonstration and evaluation boards use 11.8kΩ for RENB and 32.4kΩ for RENT resulting in a rising UVLO of 4.5V. This divider presents 5.34V to the EN input when the divider input is raised to 20V. OUTPUT VOLTAGE SELECTION Output voltage is determined by a divider of two resistors connected between VO and ground. The midpoint of the divider is connected to the FB input. The voltage at FB is compared to a 0.8V internal reference. In normal operation an on-time cycle is initiated when the voltage on the FB pin falls below 0.8V. The main MOSFET on-time cycle causes the output voltage to rise and the voltage at the FB to exceed 0.8V. As long as the voltage at FB is above 0.8V, on-time cycles will not occur. The regulated output voltage determined by the external divider resistors RFBT and RFBB is: VO = 0.8V * (1 + RFBT / RFBB) (2) Rearranging terms; the ratio of the feedback resistors for a desired output voltage is: RFBT / RFBB = (VO / 0.8V) - 1 (3) These resistors should be chosen from values in the range of 1.0 kohm to 10.0 kohm. For VO = 0.8V the FB pin can be connected to the output directly so long as an output preload resistor remains that draws more than 20uA. Converter operation requires this minimum load to create a small inductor ripple current and maintain proper regulation when no load is present. A feed-forward capacitor is placed in parallel with RFBT to improve load step transient response. Its value is usually determined experimentally by load stepping between DCM and CCM conduction modes and adjusting for best transient response and minimum output ripple. A table of values for RFBT , RFBB , CFF and RON is included in the applications schematic. SOFT-START CAPACITOR SELECTION Programmable soft-start permits the regulator to slowly ramp to its steady state operating point after being enabled, thereby reducing current inrush from the input supply and slowing the output voltage rise-time to prevent overshoot. Upon turn-on, after all UVLO conditions have been passed, an internal 8uA current source begins charging the external soft-start capacitor. The soft-start time duration to reach steady state operation is given by the formula: tSS = VREF * CSS / Iss = 0.8V * CSS / 8uA (4) This equation can be rearranged as follows: CSS = tSS * 8 μA / 0.8V (5) Use of a 0.022μF capacitor results in 2.2 msec soft-start duration. This is recommended as a minimum value. As the soft-start input exceeds 0.8V the output of the power stage will be in regulation. The soft-start capacitor continues charging until it reaches approximately 3.8V on the SS pin. Voltage levels between 0.8V and 3.8V have no effect on other circuit operation. 10 Copyright © 1999-2012, Texas Instruments Incorporated LMZ12003EXT Note that the following conditions will reset the soft-start capacitor by discharging the SS input to ground with an internal 200 μA current sink. • The enable input being “pulled low” • Thermal shutdown condition • Over-current fault • Internal Vcc UVLO (Approx 4V input to VIN) CO SELECTION None of the required CO output capacitance is contained within the module. At a minimum, the output capacitor must meet the worst case minimum ripple current rating of 0.5 * ILR P-P, as calculated in equation (19) below. Beyond that, additional capacitance will reduce output ripple so long as the ESR is low enough to permit it. A minimum value of 10 μF is generally required. Experimentation will be required if attempting to operate with a minimum value. Ceramic capacitors or other low ESR types are recommended. See AN-2024 for more detail. The following equation provides a good first pass approximation of CO for load transient requirements: CO≥ISTEP*VFB*L*VIN/ (4*VO*(VIN—VO)*VOUT-TRAN)(6) Solving: CO≥ 3A*0.8V*6.8μH*12V / (4*3.3V*( 12V — 3.3V)*33mV) ≥ 52μF (7) The LMZ12003EXT demonstration and evaluation boards are populated with a 100 uF 6.3V X5R output capacitor. Locations for extra output capacitors are provided. CIN SELECTION The LMZ12003EXT module contains an internal 0.47 µF input ceramic capacitor. Additional input capacitance is required external to the module to handle the input ripple current of the application. This input capacitance should be located in very close proximity to the module. Input capacitor selection is generally directed to satisfy the input ripple current requirements rather than by capacitance value. Worst case input ripple current rating is dictated by the equation: I(CIN(RMS)) ≊ 1 /2 * IO * √ (D / 1-D) (8) where D ≊ VO / VIN (As a point of reference, the worst case ripple current will occur when the module is presented with full load current and when VIN = 2 * VO). Recommended minimum input capacitance is 10uF X7R ceramic with a voltage rating at least 25% higher than the maximum applied input voltage for the application. It is also recommended that attention be paid to the voltage and temperature deratings of the capacitor selected. It should be noted that ripple current rating of ceramic capacitors may be missing from the capacitor data sheet and you may have to contact the capacitor manufacturer for this rating. If the system design requires a certain minimum value of input ripple voltage ΔVIN be maintained then the following equation may be used. CIN ≥ IO * D * (1–D) / fSW-CCM * ΔVIN(9) If ΔVIN is 1% of VIN for a 20V input to 3.3V output application this equals 200 mV and fSW = 400 kHz. CIN≥ 3A * 3.3V/20V * (1– 3.3V/20V) / (400000 * 0.200 V) ≥ 5.2μF Additional bulk capacitance with higher ESR may be required to damp any resonant effects of the input capacitance and parasitic inductance of the incoming supply lines. RON RESISTOR SELECTION Many designs will begin with a desired switching frequency in mind. For that purpose the following equation can be used. fSW(CCM) ≊ VO / (1.3 * 10-10 * RON) (10) This can be rearranged as RON ≊ VO / (1.3 * 10 -10 * fSW(CCM)) (11) The selection of RON and fSW(CCM) must be confined by limitations in the on-time and off-time for the COT control section. The on-time of the LMZ12003EXT timer is determined by the resistor RON and the input voltage VIN. It is calculated as follows: tON = (1.3 * 10-10 * RON) / VIN (12) The inverse relationship of tON and VIN gives a nearly constant switching frequency as VIN is varied. RON should be selected such that the on-time at maximum VIN is greater than 150 ns. The on-timer has a limiter to ensure a minimum of 150 ns for tON. This limits the maximum operating frequency, which is governed by the following equation: fSW(MAX) = VO / (VIN(MAX) * 150 nsec) (13) This equation can be used to select RON if a certain operating frequency is desired so long as the minimum on-time of 150 ns is observed. The limit for RON can be calculated as follows: RON ≥ VIN(MAX) * 150 nsec / (1.3 * 10 -10) (14) Copyright © 1999-2012, Texas Instruments Incorporated 11 LMZ12003EXT If RON calculated in (11) is less than the minimum value determined in (14) a lower frequency should be selected. Alternatively, VIN(MAX) can also be limited in order to keep the frequency unchanged. Additionally note, the minimum off-time of 260 ns limits the maximum duty ratio. Larger RON (lower FSW) should be selected in any application requiring large duty ratio. Discontinuous Conduction and Continuous Conduction Modes At light load the regulator will operate in discontinuous conduction mode (DCM). With load currents above the critical conduction point, it will operate in continuous conduction mode (CCM). When operating in DCM the switching cycle begins at zero amps inductor current; increases up to a peak value, and then recedes back to zero before the end of the off-time. Note that during the period of time that inductor current is zero, all load current is supplied by the output capacitor. The next on-time period starts when the voltage on the at the FB pin falls below the internal reference. The switching frequency is lower in DCM and varies more with load current as compared to CCM. Conversion efficiency in DCM is maintained since conduction and switching losses are reduced with the smaller load and lower switching frequency. Operating frequency in DCM can be calculated as follows: fSW(DCM)≊VO*(VIN-1)*6.8μH*1.18*1020*IO/(VIN–VO)*RON2 (15) In CCM, current flows through the inductor through the entire switching cycle and never falls to zero during the off-time. The switching frequency remains relatively constant with load current and line voltage variations. The CCM operating frequency can be calculated using equation 7 above. Following is a comparison pair of waveforms of the showing both CCM (upper) and DCM operating modes. CCM and DCM Operating Modes VIN = 12V, VO = 3.3V, IO = 3A/0.4A 2 μsec/div 30117512 The approximate formula for determining the DCM/CCM boundary is as follows: IDCB≊VO*(VIN–VO)/(2*6.8 μH*fSW(CCM)*VIN) (16) Following is a typical waveform showing the boundary condition. Transition Mode Operation VIN = 12V, VO = 3.3V, IO = 0.5 A 2 μsec/div 30117514 The inductor internal to the module is 6.8 μH. This value was chosen as a good balance between low and high input voltage applications. The main parameter affected by the inductor is the amplitude of the inductor ripple current (ILR). ILR can be calculated with: ILR P-P=VO*(VIN- VO)/(6.8µH*fSW*VIN) (17) Where VIN is the maximum input voltage and fSW is determined from equation 10. 12 Copyright © 1999-2012, Texas Instruments Incorporated LMZ12003EXT If the output current IO is determined by assuming that IO = IL, the higher and lower peak of ILR can be determined. Be aware that the lower peak of ILR must be positive if CCM operation is required. POWER DISSIPATION AND BOARD THERMAL REQUIREMENTS For the design case of VIN = 12V, VO = 3.3V, IO = 3A, TAMB(MAX) = 85°C , and TJUNCTION = 125°C, the device must see a thermal resistance from case to ambient of less than: θCA< (TJ-MAX — TAMB(MAX)) / PIC-LOSS - θJC (18) Given the typical thermal resistance from junction to case to be 1.9 °C/W .Use the 85°C power dissipation curves in the Typical Performance Characteristics section to estimate the PIC-LOSS for the application being designed. In this application it is 2.25W θCA< (125 — 85) / 2.25W —1.9 = 15.8 To reach θCA = 15.8, the PCB is required to dissipate heat effectively. With no airflow and no external heat, a good estimate of the required board area covered by 1 oz. copper on both the top and bottom metal layers is: Board Area_cm2 > 500°C x cm2/W / θJC (19) As a result, approximately 31 square cm of 1 oz copper on top and bottom layers is required for the PCB design. The PCB copper heat sink must be connected to the exposed pad. Approximately thirty six, 10 mils (254 μm) thermal vias spaced 59 mils (1.5 mm) apart must connect the top copper to the bottom copper. For an example of a high thermal performance PCB layout, refer to the demo board application note AN-2024. PC BOARD LAYOUT GUIDELINES PC board layout is an important part of DC-DC converter design. Poor board layout can disrupt the performance of a DC-DC converter and surrounding circuitry by contributing to EMI, ground bounce and resistive voltage drop in the traces. These can send erroneous signals to the DC-DC converter resulting in poor regulation or instability. Good layout can be implemented by following a few simple design rules. 30117511 1. Minimize area of switched current loops. From an EMI reduction standpoint, it is imperative to minimize the high di/dt current paths during PC board layout. The high current loops that do not overlap have high di/dt content that will cause observable high frequency noise on the output pin if the input capacitor CIN1 is placed a distance away for the LMZ12003. Therefore physically place CIN1 asa close as possible to the LMZ12003EXT VIN and GND exposed pad. This will minimize the high di/dt area and reduce radiated EMI. Additionally, grounding for both the input and output capacitor should consist of a localized top side plane that connects to the GND exposed pad (EP). 2. Have a single point ground. The ground connections for the feedback, soft-start, and enable components should be routed to the GND pin of the device. This prevents any switched or load currents from flowing in the analog ground traces. If not properly handled, poor grounding can result in degraded load regulation or erratic output voltage ripple behavior. Provide the single point ground connection from pin 4 to EP. 3. Minimize trace length to the FB pin. Both feedback resistors, R FBT and RFBB, and the feed forward capacitor CFF, should be located close to the FB pin. Since the FB node is high impedance, maintain the copper area as small as possible. The trace are from RFBT, RFBB, and CFF should be routed away from the body of the LMZ12003EXT to minimize noise. 4. Make input and output bus connections as wide as possible. This reduces any voltage drops on the input or output of the converter and maximizes efficiency. To optimize voltage accuracy at the load, ensure that a separate feedback voltage sense trace is made to the load. Doing so will correct for voltage drops and provide optimum output accuracy. 5. Provide adequate device heat-sinking. Use an array of heat-sinking vias to connect the exposed pad to the ground plane on the bottom PCB layer. If the PCB has a plurality of copper layers, these thermal vias can also be employed to make connection to inner layer heat-spreading ground planes. For best results use a 6 x 6 via array with minimum via diameter of 10mils (254 μm) thermal vias spaced 59mils (1.5 mm). Ensure enough copper area is used for heat-sinking to keep the junction temperature below 125°C. Copyright © 1999-2012, Texas Instruments Incorporated 13 LMZ12003EXT Additional Features OUTPUT OVER-VOLTAGE COMPARATOR The voltage at FB is compared to a 0.92V internal reference. If FB rises above 0.92V the on-time is immediately terminated. This condition is known as over-voltage protection (OVP). It can occur if the input voltage is increased very suddenly or if the output load is decreased very suddenly. Once OVP is activated, the top MOSFET on-times will be inhibited until the condition clears. Additionally, the synchronous MOSFET will remain on until inductor current falls to zero. CURRENT LIMIT Current limit detection is carried out during the off-time by monitoring the current in the synchronous MOSFET. Referring to the Functional Block Diagram, when the top MOSFET is turned off, the inductor current flows through the load, the PGND pin and the internal synchronous MOSFET. If this current exceeds 4.2A (typical) the current limit comparator disables the start of the next ontime period. The next switching cycle will occur only if the FB input is less than 0.8V and the inductor current has decreased below 4.2A. Inductor current is monitored during the period of time the synchronous MOSFET is conducting. So long as inductor current exceeds 4.2A, further on-time intervals for the top MOSFET will not occur. Switching frequency is lower during current limit due to the longer off-time. It should also be noted that current limit is dependent on both duty cycle and temperature as illustrated in the graphs in the typical performance section. THERMAL PROTECTION The junction temperature of the LMZ12003EXT should not be allowed to exceed its maximum ratings. Thermal protection is implemented by an internal Thermal Shutdown circuit which activates at 165 °C (typ) causing the device to enter a low power standby state. In this state the main MOSFET remains off causing VO to fall, and additionally the CSS capacitor is discharged to ground. Thermal protection helps prevent catastrophic failures for accidental device overheating. When the junction temperature falls back below 145 °C (typ Hyst = 20 °C) the SS pin is released, VO rises smoothly, and normal operation resumes. Applications requiring maximum output current especially those at high input voltage may require application derating at elevated temperatures. ZERO COIL CURRENT DETECTION The current of the lower (synchronous) MOSFET is monitored by a zero coil current detection circuit which inhibits the synchronous MOSFET when its current reaches zero until the next on-time. This circuit enables the DCM operating mode, which improves efficiency at light loads. PRE-BIASED STARTUP The LMZ12003EXT will properly start up into a pre-biased output. This startup situation is common in multiple rail logic applications where current paths may exist between different power rails during the startup sequence. The following scope capture shows proper behavior during this event. Pre-Biased Startup 30117525 14 Copyright © 1999-2012, Texas Instruments Incorporated LMZ12003EXT Evaluation Board Schematic Diagram 30117507 Ref Des Description Case Size Case Size Manufacturer P/N U1 SIMPLE SWITCHER ® TO-PMOD-7 National Semiconductor LMZ12003EXTTZ-ADJ Cin1 1 µF, 50V, X7R 1206 Taiyo Yuden UMK316B7105KL-T Cin2 10 µF, 50V, X7R 1210 Taiyo Yuden UMK325BJ106MM-T CO1 1 µF, 50V, X7R 1206 Taiyo Yuden UMK316B7105KL-T CO2 100 µF, 6.3V, X7R 1210 Taiyo Yuden JMK325BJ10CR7MM-T RFBT 1.37 kΩ 0603 Vishay Dale CRCW06031K37FKEA RFBB 1.07 kΩ 0603 Vishay Dale CRCW06031K07FKEA RON 32.4 kΩ 0603 Vishay Dale CRCW060332K4FKEA 32.4 kΩ 0603 Vishay Dale CRCW060332K4FKEA RENB 11.8 kΩ 0603 Vishay Dale CRCW060311k8FKEA CFF 22 nF, ±10%, X7R, 16V 0603 TDK C1608X7R1H223K CSS 22 nF, ±10%, X7R, 16V 0603 TDK C1608X7R1H223K RENT Copyright © 1999-2012, Texas Instruments Incorporated 15 LMZ12003EXT 30117516 30117517 16 Copyright © 1999-2012, Texas Instruments Incorporated LMZ12003EXT Physical Dimensions inches (millimeters) unless otherwise noted 7-Lead TZA Package NS Package Number TZA07A Copyright © 1999-2012, Texas Instruments Incorporated 17 Notes Copyright © 1999-2012, Texas Instruments Incorporated IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other changes to its semiconductor products and services per JESD46C and to discontinue any product or service per JESD48B. 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