Intersil ISL6251HAZ-T Low cost multi-chemistry battery charger controller Datasheet

ISL6251, ISL6251A
®
Data Sheet
June 17, 2005
Low Cost Multi-Chemistry Battery
Charger Controller
Features
• ±0.5% Charge Voltage Accuracy (-10°C to 100°C)
The ISL6251, ISL6251A is a highly integrated battery
charger controller for Li-Ion/Li-Ion polymer batteries and
NiMH batteries. High Efficiency is achieved by a
synchronous buck topology and the use of a MOSFET,
instead of a diode, for selecting power from the adapter or
battery. The low side MOSFET emulates a diode at light
loads to improve the light load efficiency and prevent system
bus boosting.
The constant output voltage can be selected for 2, 3 and 4
series Li-Ion cells with 0.5% accuracy over temperature. It
can be also programmed between 4.2V+5%/cell and
4.2V-5%/cell to optimize battery capacity. When supplying
the load and battery charger simultaneously, the input
current limit for the AC adapter is programmable to within
3% accuracy to avoid overloading the AC adapter, and to
allow the system to make efficient use of available adapter
power for charging. It also has a wide range of
programmable charging current. The ISL6251, ISL6251A
provides outputs that are used to monitor the current drawn
from the AC adapter, and monitor for the presence of an AC
adapter. The ISL6251, ISL6251A automatically transitions
from regulating current mode to regulating voltage mode.
TEMP
RANGE (°C)
• ±3% Accurate Input Current Limit
• ±5% Accurate Battery Charge Current Limit
• ±25% Accurate Battery Trickle Charge Current Limit
(ISL6251A)
• Programmable Charge Current Limit, Adapter Current
Limit and Charge Voltage
• Fixed 300kHz PWM Synchronous Buck Controller with
Diode Emulation at Light Load
• Output for Current Drawn from AC Adapter
• AC Adapter Present Indicator
• Fast Input Current Limit Response
• Input Voltage Range 7V to 25V
• Support 2, 3 and 4 Cells Battery Pack
• Up to 17.64V Battery-Voltage Set Point
• Thermal Shutdown
• Support Pulse Charging
• Less than 10µA Battery Leakage Current
• Charge Any Battery Chemistry: Li-Ion, NiCd, NiMH, etc.
Ordering Information
PART
NUMBER
FN9202.1
PACKAGE
PKG.
DWG. #
ISL6251HRZ
(Notes 1, 2)
-10 to 100
28 Ld 5x5 QFN
(Pb-free)
L28.5×5
ISL6251HAZ
(Notes 1, 2)
-10 to 100
24 Ld QSOP
(Pb-free)
M24.15
ISL6251AHRZ
(Notes 1, 2)
-10 to 100
28 Ld 5x5 QFN
(Pb-free)
L28.5×5
ISL6251AHAZ
(Notes 1, 2)
-10 to 100
24 Ld QSOP
(Pb-free)
M24.15
• Pb-Free Plus Anneal Available (RoHS Compliant)
Applications
• Notebook, Desknote and Sub-notebook Computers
• Personal Digital Assistant
NOTES:
1. Intersil Pb-free plus anneal products employ special Pb-free
material sets; molding compounds/die attach materials and
100% matte tin plate termination finish, which are RoHS
compliant and compatible with both SnPb and Pb-free soldering
operations. Intersil Pb-free products are MSL classified at
Pb-free peak reflow temperatures that meet or exceed the
Pb-free requirements of IPC/JEDEC J STD-020.
2. Add “-T” for Tape and Reel.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2005. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6251, ISL6251A
Pinouts
ISL6251, ISL6251A
(24 LD QSOP)
TOP VIEW
EN
CELLS
NA
ACSET
VDD
DCIN
NA
ACPRN
CSON
ISL6251, ISL6251A
(28 LD QFN)
TOP VIEW
28
27
26
25
24
23
22
1
21
2
20
VDD
1
24
DCIN
ACSET
2
23
ACPRN
EN
3
22
CSON
CSOP
CELLS
4
21
CSOP
CSIN
ICOMP
5
20
CSIN
VCOMP
6
19
CSIP
ICOMP
3
19
CSIP
ICM
7
18
PHASE
VCOMP
4
18
NA
VREF
8
17
UGATE
CHLIM
9
16
BOOT
ACLIM
10
15
VDDP
VADJ
11
14
LGATE
GND
12
13
PGND
ICM
5
17
NA
VREF
6
16
PHASE
CHLIM
7
8
9
10
11
12
13
14
ACLIM
VADJ
GND
PGND
LGATE
VDDP
BOOT
15
2
UGATE
FN9202.1
June 17, 2005
ISL6251, ISL6251A
Absolute Maximum Ratings
Thermal Information
DCIN, CSIP, CSON to GND. . . . . . . . . . . . . . . . . . . . . -0.3V to +28V
CSIP-CSIN, CSOP-CSON . . . . . . . . . . . . . . . . . . . . . -0.3V to +0.3V
PHASE to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -7V to 30V
BOOT to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +35V
BOOT-PHASE, VDD-GND, VDDP-PGND,
ACPRN to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to 7V
ACSET to GND (Note 3) . . . . . . . . . . . . . . . . . . -0.3V to VDD+0.3V
ICM, ICOMP, VCOMP to GND. . . . . . . . . . . . . . -0.3V to VDD+0.3V
ACLIM, CHLIM, VREF, CELLS to GND . . . . . . . -0.3V to VDD+0.3V
EN, VADJ, PGND to GND . . . . . . . . . . . . . . . . . .-0.3V to VDD+0.3V
UGATE. . . . . . . . . . . . . . . . . . . . . . . . . PHASE-0.3V to BOOT+0.3V
LGATE . . . . . . . . . . . . . . . . . . . . . . . . . . PGND-0.3V to VDDP+0.3V
Thermal Resistance
θJA (°C/W)
θJC (°C/W)
QFN Package (Notes 4, 6). . . . . . . . . .
39
9.5
QSOP Package (Note 5) . . . . . . . . . . .
88
N/A
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Level 2
Junction Temperature Range. . . . . . . . . . . . . . . . . .-10°C to +150°C
Operating Temperature Range . . . . . . . . . . . . . . . .-10°C to +100°C
Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C
Lead Temperature (soldering, 10s) . . . . . . . . . . . . . . . . . . . . +300°C
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTES:
3. When the voltage across ACSET is below 0V, the current through ACSET should be limited to less than 1mA.
4. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech
Brief TB379.
5. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
6. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
DCIN=CSIP=CSIN=18V, CSOP=CSON=12V, ACSET=1.5V, ACLIM=VREF, VADJ=Floating, EN=VDD=5V,
BOOT-PHASE=5.0V, GND=PGND=0V, CVDD=1µF, IVDD=0mA, TA=-10°C to +100°C, TJ ≤ 125°C, unless
otherwise noted.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
25
V
1.4
3
mA
3
10
µA
4.925
5.075
5.225
V
SUPPLY AND BIAS REGULATOR
DCIN Input Voltage Range
7
DCIN Quiescent Current
EN=VDD or GND, 7V ≤ DCIN ≤ 25V
Battery Leakage Current (Note 7)
DCIN=0, no load
VDD Output Voltage/Regulation
7V ≤ DCIN ≤ 25V, 0 ≤ IVDD ≤ 30mA
VDD Undervoltage Lockout Trip Point
VDD Rising
4.0
4.4
4.6
V
Hysteresis
200
250
400
mV
2.365
2.39
2.415
V
Reference Output Voltage VREF
0 ≤ IVREF ≤ 300µA
Battery Charge Voltage Accuracy
CSON=16.8V, CELLS=VDD, VADJ=Float
-0.5
0.5
%
CSON=12.6V, CELLS=GND, VADJ=Float
-0.5
0.5
%
CSON=8.4V, CELLS=Float, VADJ=Float
-0.5
0.5
%
CSON=17.64V, CELLS=VDD, VADJ=VREF
-0.5
0.5
%
CSON=13.23V, CELLS=GND, VADJ=VREF
-0.5
0.5
%
CSON=8.82V, CELLS=Float, VADJ=VREF
-0.5
0.5
%
CSON=15.96V, CELLS=VDD, VADJ=GND
-0.5
0.5
%
CSON=11.97V, CELLS=GND, VADJ=GND
-0.5
0.5
%
CSON=7.98V, CELLS=Float, VADJ=GND
-0.5
0.5
%
TRIP POINTS
ACSET Threshold
1.24
1.26
1.28
V
ACSET Input Bias Current Hysteresis
2.2
3.4
4.4
µA
ACSET Input Bias Current
ACSET ≥ 1.26V
2.2
3.4
4.4
µA
ACSET Input Bias Current
ACSET < 1.26V
-1
0
1
µA
3
FN9202.1
June 17, 2005
ISL6251, ISL6251A
Electrical Specifications
DCIN=CSIP=CSIN=18V, CSOP=CSON=12V, ACSET=1.5V, ACLIM=VREF, VADJ=Floating, EN=VDD=5V,
BOOT-PHASE=5.0V, GND=PGND=0V, CVDD=1µF, IVDD=0mA, TA=-10°C to +100°C, TJ ≤ 125°C, unless
otherwise noted. (Continued)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
245
300
355
kHz
OSCILLATOR
Frequency
PWM Ramp Voltage (peak-peak)
CSIP=18V
1.6
V
CSIP=11V
1
V
SYNCHRONOUS BUCK REGULATOR
Maximum Duty Cycle
97
99
99.6
%
3.0
Ω
UGATE Pull-up Resistance
BOOT-PHASE=5V, 500mA source current
1.8
UGATE Source Current
BOOT-PHASE=5V, BOOT-UGATE=2.5V
1.0
UGATE Pull-down Resistance
BOOT-PHASE=5V, 500mA sink current
1.0
UGATE Sink Current
BOOT-PHASE=5V, UGATE-PHASE=2.5V
1.8
LGATE Pull-up Resistance
VDDP-PGND=5V, 500mA source current
1.8
LGATE Source Current
VDDP-PGND=5V, VDDP-LGATE=2.5V
1.0
LGATE Pull-down Resistance
VDDP-PGND=5V, 500mA sink current
1.0
LGATE Sink Current
VDDP-PGND=5V, LGATE=2.5V
1.8
A
1.8
Ω
A
3.0
Ω
A
1.8
Ω
A
CHARGING CURRENT SENSING AMPLIFIER
Input Common-Mode Range
0
V
0
2.5
mV
Input Offset Voltage
Guarantee by design
Input Bias Current at CSOP
0 < CSOP < 18V
0.25
2
µA
Input Bias Current at CSON
0 < CSON < 18V
75
100
µA
3.6
V
CHLIM Input Voltage Range
-2.5
18
0
CSOP to CSON Full-Scale Current
Sense Voltage
ISL6251: CHLIM=3.3V
157
165
173
mV
ISL6251A, CHLIM=3.3V
160
165
170
mV
ISL6251: CHLIM=2.0V
95
100
105
mV
ISL6251A: CHLIM=2.0V
97
100
103
mV
ISL6251: CHLIM=0.2V
5.0
10
15.0
mV
ISL6251A: CHLIM=0.2V
7.5
10
12.5
mV
CHLIM Input Bias Current
CHLIM=GND or 3.3V, DCIN=0V
-1
1
µA
CHLIM Power-Down Mode Threshold
Voltage
CHLIM rising
80
88
95
mV
15
25
40
mV
7
25
V
-2
2
mV
CHLIM Power-Down Mode Hysteresis
Voltage
ADAPTER CURRENT SENSING AMPLIFIER
Input Common-Mode Range
Input Offset Voltage
Guarantee by design
Input Bias Current at CSIP & CSIN
Combined
CSIP=CSIN=25V
100
130
µA
Input Bias Current at CSIN
0 < CSIN < DCIN, Guaranteed by design
0.10
1
µA
4
FN9202.1
June 17, 2005
ISL6251, ISL6251A
Electrical Specifications
DCIN=CSIP=CSIN=18V, CSOP=CSON=12V, ACSET=1.5V, ACLIM=VREF, VADJ=Floating, EN=VDD=5V,
BOOT-PHASE=5.0V, GND=PGND=0V, CVDD=1µF, IVDD=0mA, TA=-10°C to +100°C, TJ ≤ 125°C, unless
otherwise noted. (Continued)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
97
100
103
mV
72
75
78
mV
ACLIM=GND
47
50
53
mV
ACLIM=VREF
10
16
20
µA
ACLIM=GND
-20
-16
-10
µA
ADAPTER CURRENT LIMIT THRESHOLD
CSIP to CSIN Full-Scale Current Sense ACLIM=VREF
Voltage
ACLIM=Float
ACLIM Input Bias Current
VOLTAGE REGULATION ERROR AMPLIFIER
Error Amplifier Transconductance from
CSON to VCOMP
CELLS=VDD
30
µA/V
Charging Current Error Amplifier
Transconductance
50
µA/V
Adapter Current Error Amplifier
Transconductance
50
µA/V
CURRENT REGULATION ERROR AMPLIFIER
BATTERY CELL SELECTOR
CELLS Input Voltage for 4 Cell Select
4.3
V
CELLS Input Voltage for 3 Cell Select
CELLS Input Voltage for 2 Cell Select
2
V
2.1
4.2
V
0
VDD
V
LOGIC INTERFACE
EN Input Voltage Range
EN Threshold Voltage
Rising
1.030
1.06
1.100
V
Falling
0.985
1.000
1.025
V
Hysteresis
30
60
90
mV
EN Input Bias Current
EN=2.5V
1.8
2.0
2.2
µA
ACPRN Sink Current
ACPRN=0.4V
3
8
11
mA
ACPRN Leakage Current
ACPRN=5V
0.5
µA
ICM Output Accuracy
(Vicm=19.9 x (Vcsip-Vcsin))
CSIP-CSIN=100mV
-3
0
+3
%
CSIP-CSIN=75mV
-4
0
+4
%
CSIP-CSIN=50mV
-5
0
+5
%
-0.5
Thermal Shutdown Temperature
150
°C
Thermal Shutdown Temperature
Hysteresis
25
°C
NOTE:
7. This is the sum of currents in these pins (CSIP, CSIN, BOOT, UGATE, PHASE, CSOP, CSON) all tied to 16.8V. No current in pins EN, ACSET,
VADJ, CELLS, ACLIM, CHLIM.
5
FN9202.1
June 17, 2005
ISL6251, ISL6251A
Typical Operating Performance
DCIN=20V, 4S2P Li-Battery, TA=25°C, unless otherwise noted.
0.1
VREF LOAD REGULATION ACCURACY (%)
VDD LOAD REGULATION ACCURACY (%)
0.6
VDD=5.075V
EN=0
0.3
0
-0.3
-0.6
0
8
16
24
32
VREF=2.390V
0.08
0.06
0.04
0.02
0
0
40
100
300
400
FIGURE 2. VREF LOAD REGULATION
FIGURE 1. VDD LOAD REGULATION
1
10
9
8
7
6
5
4
3
2
1
0
0 .9 6
VCSON=12.6V
(3 CELLS)
0 .9 2
EFFICIENCY (%)
| ICM ACCURACY | (%)
200
LOAD CURRENT (µA)
LOAD CURRENT (mA)
VCSON=8.4V
2 CELLS
VCSON=16.8V
4 CELLS
0 .8 8
0 .8 4
0 .8
0 .76
10 20 30 40 50 60 70 80 90 100
CSIP-CSIN (mV)
FIGURE 3. ICM ACCURACY vs AC ADAPTER CURRENT
6
1
1.5
2
2.5
3
3 .5
4
CHARGE CURRENT (A)
FIGURE 4. SYSTEM EFFICIENCY vs CHARGE CURRENT
CSON
5V/div
ADAPTER
CURRENT
5A/div
EN
5V/div
BATTERY
VOLTAGE
2V/div
FIGURE 5. LOAD TRANSIENT RESPONSE
0 .5
LOAD
CURRENT
5A/div
CHARGE
CURRENT
2A/div
LOAD STEP: 0-4A
CHARGE CURRENT: 3A
AC ADAPTER CURRENT LIMIT: 5.15A
0
INDUCTOR
CURRENT
2A/div
CHARGE
CURRENT
2A/div
FIGURE 6. CHARGER ENABLE & SHUTDOWN
FN9202.1
June 17, 2005
ISL6251, ISL6251A
Typical Operating Performance
DCIN=20V, 4S2P Li-Battery, TA=25°C, unless otherwise noted. (Continued)
INDUCTOR
CURRENT
2A/div
PHASE
10V/div
BATTERY
REMOVAL
BATTERY
INSERTION
CHLIM=0.2V
CSON=8V
CSON
10V/div
INDUCTOR
CURRENT
1A/div
VCOMP
2V/div
VCOMP
ICOMP
ICOMP
2V/div
FIGURE 7. BATTERY INSERTION AND REMOVAL
PHASE
10V/div
UGATE
5V/div
FIGURE 8. SWITCHING WAVEFORMS AT DIODE EMULATION
CHARGE
CURRENT
1A/div
UGATE
2V/div
CHLIM
1V/div
LGATE
2V/div
FIGURE 9. SWITCHING WAVEFORMS IN CC MODE
7
FIGURE 10. TRICKLE TO FULL-SCALE CHARGING
FN9202.1
June 17, 2005
ISL6251, ISL6251A
Functional Pin Descriptions
ICM
BOOT
ICM is the adapter current output. The output of this pin
produces a voltage proportional to the adapter current.
Connect BOOT to a 0.1µF ceramic capacitor to PHASE pin
and connect to the cathode of the bootstrap schottky diode.
UGATE
UGATE is the high side MOSFET gate drive output.
LGATE
LGATE is the low side MOSFET gate drive output; swing
between 0V and VDDP.
PHASE
The Phase connection pin connects to the high side
MOSFET source, output inductor, and low side MOSFET
drain.
CSOP/CSON
CSOP/CSON is the battery charging current sensing
positive/negative input. The differential voltage across CSOP
and CSON is used to sense the battery charging current,
and is compared with the charging current limit threshold to
regulate the charging current. The CSON pin is also used as
the battery feedback voltage to perform voltage regulation.
CSIP/CSIN
CSIP/CSIN is the AC adapter current sensing
positive/negative input. The differential voltage across CSIP
and CSIN is used to sense the AC adapter current, and is
compared with the AC adapter current limit to regulate the
AC adapter current.
GND
PGND
PGND is the power ground. Connect PGND to the source of
the low side MOSFET for the low side MOSFET gate driver.
VDD
VDD is an internal LDO output to supply IC analog circuit.
Connect a 1µF ceramic capacitor to ground.
VDDP
VDDP is the supply voltage for the low-side MOSFET gate
driver. Connect a 4.7Ω resistor to VDD and a 1µF ceramic
capacitor to power ground.
ICOMP
ICOMP is a current loop error amplifier output.
VCOMP
VCOMP is a voltage loop amplifier output.
CELLS
This pin is used to select the battery voltage. CELLS=VDD
for a 4S battery pack, CELLS=GND for a 3S battery pack,
CELLS=Float for a 2S battery pack.
VADJ
VADJ adjusts battery regulation voltage. VADJ=VREF for
4.2V+5%/cell; VADJ=Floating for 4.2V/cell; VADJ=GND for
4.2V-5%/cell. Connect to a resistor divider to program the
desired battery cell voltage between 4.2V-5% and 4.2V+5%.
CHLIM
GND is an analog ground.
DCIN
The DCIN pin is the input of the internal 5V LDO. Connect it
to the AC adapter output. Connect DCIN to a 0.1µF ceramic
capacitor.
ACSET
ACSET is an AC adapter detection input. Connect to a
resistor divider from the adapter input.
ACPRN
ACPRN is an AC adapter present open drain output.
ACPRN is active low when ACSET is higher than typically
1.26V, and active high when ACSET is lower than typically
1.26V.
CHLIM is the battery charge current limit set pin. CHLIM
input voltage range is 0.1V to 3.6V. When CHLIM=3.3V, the
set point for CSOP-CSON is 165mV. The charger shuts
down if CHLIM is forced below 88mV.
ACLIM
ACLIM is the adapter current limit set pin. ACLIM=VREF for
100mV, ACLIM=Floating for 75mV, and ACLIM=GND for
50mV. Connect a resistor divider to program the adapter
current limit threshold between 50mV and 100mV.
VREF
VREF is a 2.39V reference output pin. It is internally
compensated. Do not connect a decoupling capacitor.
EN
EN is the Charge Enable input. Connecting EN to high
enables the charge control function, connecting EN to low
disables charging functions. Use with a thermistor to detect
a hot battery and suspend charging.
8
FN9202.1
June 17, 2005
ISL6251, ISL6251A
CSIN
CSIP
ICM
+
+ CA1 ×CA1
19.9
DCIN
ACSET
ACPRN
LDO
Regulator
++
--
1.27V
1.26V
2.1V
2.1V
ICOMP
+
gm1
-
VCOMP
LDO
Regulator
gm3
+
Adapter
Current Limit
Limit Set
Set
Current
ACLIM
BOOT
Min
Current
Buffer
Min
Voltage
Buffer
0.25 VCA2
0.25V
CA2
UGATE
++
gm2
-
PGND
1.06V
- 1.065V
VDD
+
-
CSON
+
+
GND
×
20
CA2
- CA2
LGATE
+
VCA2
Reference
VDDP
VDDP
CELLS
VREF
PHASE
PWM
+
Voltage
Selector
VADJ
VDD
CSOP
EN
CHLIM
FIGURE 11. FUNCTIONAL BLOCK DIAGRAM
9
FN9202.1
June 17, 2005
ISL6251, ISL6251A
D4
AC ADAPTER
R8
130K
1%
D3
R9
10.2K
1%
C8
0.1µF
DCIN
CSIP
CSIP
ACSET
ACSET
C2
0.1µF
ISL6251
ISL6251
ISL6251A
ISL6251A
C7
1µF
VDDP
VDDP
SYSTEM LOAD
CSIN
CSIN
R10
4.7Ω
3.3V
R3
18Ω
To Host
Controller
D2
ACPRN
ACPRN
UGATE
UGATE
ICOMP
ICOMP
PHASE
PHASE
VCOMP
VCOMP
LGATE
LGATE
C6:6.8nF
FLOATING
4.2V/CELL
CHARGE
ENABLE
VREF
BOOT
BOOT
C9
1µF
R6:10K C5:10nF
VADJ
VADJ
PGND
PGND
EN
EN
CSOP
CSOP
R12
2.6A CHARGE LIMIT
20K 1% 253mA Trickle Charge
R13
1.87K
1%
Trickle Charge
Enable
Q3
R11
130K
1%
C1
10µF
VDDP
VDD
VDD
R5
100K
R2
20mΩ
Q1
C4
0.1µF
Q2
D1
Optional
C3
1µF
R4
2.2Ω
L
10µH
R1
40mΩ
BAT+
CSON
CSON
CHLIM
CHLIM
ACLIM
ACLIM
VREF
VREF
CELLS
CELLS
VDD
4 CELLS
C10
10µF
Battery
Pack
BAT-
ICM
ICM
R7: 100Ω
GND
GND
C11
3300pF
FIGURE 12. ISL6251, ISL6251A TYPICAL APPLICATION CIRCUIT WITH FIXED CHARGING PARAMETERS
10
FN9202.1
June 17, 2005
ISL6251, ISL6251A
D4
AC ADAPTER
C8
0.1µF
D3
R8
130K
1%
R9
10.2K,1%
DCIN
DCIN
CSIP
CSIP
ACSET
ACSET
C2
0.1µF
VDDP
VDDP
C7
1µF
R10
4.7Ω
VCC
R5
100K
DIGITAL
INPUT
C9
1µF
D/A OUTPUT
OUTPUT
ISL6251
ISL6251
CSIN
CSIN
ISL6251A
ISL6251A
VDD
VDD
BOOT
BOOT
C11
3300pF 5.15A INPUT
CURRENT LIMIT
C6
6.8nF
HOST
UGATE
UGATE
CHLIM
CHLIM
PHASE
PHASE
EN
EN
LGATE
LGATE
ICM
ICM
PGND
PGND
ACLIM
ACLIM
CSOP
CSOP
VCOMP
VCOMP
AVDD/VREF
C5
10nF
VADJ
VADJ
R11, R12, R13
10k
FLOATING
4.2V/CELL
SCL
SDL
A/D INPUT
GND
C1
10µF
VDDP
Q1
C4
0.1µF
D1
Optional
Q2
C3
1µF
VREF
VREF
ICOMP
ICOMP
R6
10K
R3: 18Ω
SYSTEM LOAD
D2
ACPRN
ACPRN
R7: 100Ω
A/D INPUT
R2
20mΩ
R1
40mΩ
R4
2.2Ω
CSON
CSON
CELLS
CELLS
L
10µH
BAT+
3 CELLS
C10
10µF
Battery
Pack
GND
GND
SCL
SDL
TEMP
BAT-
FIGURE 13. ISL6251, ISL6251A TYPICAL APPLICATION CIRCUIT WITH MICRO-CONTROLLER
11
FN9202.1
June 17, 2005
ISL6251, ISL6251A
Theory of Operation
Introduction
The ISL6251, ISL6251A includes all of the functions
necessary to charge 2 to 4 cell Li-Ion and Li-polymer
batteries. A high efficiency synchronous buck converter is
used to control the charging voltage and charging current up
to 10A. The ISL6251, ISL6251A has input current limiting
and analog inputs for setting the charge current and charge
voltage; CHLIM inputs are used to control charge current
and VADJ inputs are used to control charge voltage.
The ISL6251, ISL6251A charges the battery with constant
charge current, set by CHLIM input, until the battery voltage
rises up to a programmed charge voltage set by VADJ input;
then the charger begins to operate at a constant voltage
charge mode.
The EN input allows shutdown of the charger through a
command from a micro-controller. It also uses EN to safely
shutdown the charger when the battery is in extremely hot
conditions. The amount of adapter current is reported on the
ICM output. Figure 11 shows the IC functional block
diagram.
The synchronous buck converter uses external N-channel
MOSFETs to convert the input voltage to the required
charging current and charging voltage. Figure 12 shows the
ISL6251, ISL6251A typical application circuit with charging
current and charging voltage fixed at specific values. The
typical application circuit shown in Figure 13 shows the
ISL6251, ISL6251A typical application circuit which uses a
micro-controller to adjust the charging current set by CHLIM
input. The voltage at CHLIM and the value of R1 sets the
charging current. The DC/DC converter generates the
control signals to drive two external N-channel MOSFETs to
regulate the voltage and current set by the ACLIM, CHLIM,
VADJ and CELLS inputs.
The ISL6251, ISL6251A features a voltage regulation loop
(VCOMP) and two current regulation loops (ICOMP). The
VCOMP voltage regulation loop monitors CSON to ensure
that its voltage never exceeds the voltage and regulates the
battery charge voltage set by VADJ. The ICOMP current
regulation loops regulate the battery charging current
delivered to the battery to ensure that it never exceeds the
charging current limit set by CHLIM; and the ICOMP current
regulation loops also regulate the input current drawn from
the AC adapter to ensure that it never exceeds the input
current limit set by ACLIM, and to prevent a system crash
and AC adapter overload.
PWM Control
The ISL6251, ISL6251A employs a fixed frequency PWM
current mode control architecture with a feed forward
function. The feed-forward function maintains a constant
modulator gain of 11 to achieve fast line regulation as the
buck input voltage changes. When the battery charge
12
voltage approaches the input voltage, the DC/DC converter
operates in dropout mode, where there is a timer to prevent
the frequency from dropping into the audible frequency
range. It can achieve duty cycle of up to 99.6%.
To prevent boosting of the system bus voltage, the battery
charger operates in standard-buck mode when CSOPCSON drops below 4.25mV. Once in standard-buck mode,
hysteresis does not allow synchronous operation of the
DC/DC converter until CSOP-CSON rises above 12.5mV.
An adaptive gate drive scheme is used to control the dead
time between two switches. The dead time control circuit
monitors the LGATE output and prevents the upper side
MOSFET from turning on until LGATE is fully off, preventing
cross-conduction and shoot-through. In order for the dead
time circuit to work properly, there must be a low resistance,
low inductance path from the LGATE driver to MOSFET
gate, and from the source of MOSFET to PGND. The
external Schottky diode is between the VDDP pin and BOOT
pin to keep the bootstrap capacitor charged.
Setting the Battery Regulation Voltage
The ISL6251, ISL6251A uses a high-accuracy trimmed
band-gap voltage reference to regulate the battery charging
voltage. The VADJ input adjusts the charger output voltage,
and the VADJ control voltage can vary from 0 to VREF,
providing a 10% adjustment range (from 4.2V-5% to
4.2V+5%) on CSON regulation voltage. An overall voltage
accuracy of better than 0.5% is achieved.
The per-cell battery termination voltage is a function of the
battery chemistry. Consult the battery manufacturers to
determine this voltage.
• Float VADJ to set the battery voltage VCSON=4.2V ×
number of the cells,
• Connect VADJ to VREF to set 4.41V × number of cells,
• Connect VADJ to ground to set 3.99V × number of the
cells.
So, the maximum battery voltage of 17.6V can be achieved.
Note that other battery charge voltages can be set by
connecting a resistor divider from VREF to ground. The
resistor divider should be sized to draw no more than 100µA
from VREF; or connect a low impedance voltage source like
the D/A converter in the micro-controller. The programmed
battery voltage per cell can be determined by the following
equation:
VCELL = 0.175 VVADJ + 3.99 V
Connect CELLS as shown in Table 1 to charge 2, 3 or 4 Li+
cells. When charging other cell chemistries, use CELLS to
select an output voltage range for the charger. The internal
error amplifier gm1 maintains voltage regulation. The voltage
error amplifier is compensated at VCOMP. The component
values shown in Figure 12 provide suitable performance for
most applications. Individual compensation of the voltage
FN9202.1
June 17, 2005
ISL6251, ISL6251A
regulation and current-regulation loops allows for optimal
compensation.
TABLE 1. CELL NUMBER PROGRAMMING
CELLS
CELL NUMBER
VDD
4
GND
3
Float
2
Setting the Battery Charge Current Limit
The CHLIM input sets the maximum charging current. The
current set by the current sense-resistor connects between
CSOP and CSON. The full-scale differential voltage between
CSOP and CSON is 165mV for CHLIM=3.3V, so the
maximum charging current is 4.125A for a 40mΩ sensing
resistor. Other battery charge current-sense threshold
values can be set by connecting a resistor divider from
VREF or 3.3V to ground, or by connecting a low impedance
voltage source like a D/A converter in the micro-controller.
The charge current limit threshold is given by:
165mV V CHLIM
I CHG = ------------------- ---------------------3.3V
R1
To set the trickle charge current for the dumb charger, a
resistor in series with a switch Q3 (Figure 12) controlled by
the micro-controller is connected from CHLIM pin to ground.
The trickle charge current is determined by:
165mV V CHLIM ,trickle
I CHG = ------------------- ---------------------------------------3.3V
R1
When the CHLIM voltage is below 88mV (typical), it will
disable the battery charger. When choosing the current
sensing resistor, note that the voltage drop across the
sensing resistor causes further power dissipation, reducing
efficiency. However, adjusting CHLIM voltage to reduce the
voltage across the current sense resistor R1 will degrade
accuracy due to the smaller signal to the input of the current
sense amplifier. There is a trade-off between accuracy and
power dissipation. A low pass filter is recommended to
eliminate switching noise. Connect the resistor to the CSOP
pin instead of the CSON pin, as the CSOP pin has lower
bias current and less influence on current-sense accuracy
and voltage regulation accuracy.
Setting the Input Current Limit
The total input current from an AC adapter, or other DC
source, is a function of the system supply current and the
battery-charging current. The input current regulator limits
the input current by reducing the charging current, when the
input current exceeds the input current limit set point.
System current normally fluctuates as portions of the system
are powered up or down. Without input current regulation,
the source must be able to supply the maximum system
13
current and the maximum charger input current
simultaneously. By using the input current limiter, the current
capability of the AC adapter can be lowered, reducing
system cost.
The ISL6251, ISL6251A limits the battery charge current
when the input current-limit threshold is exceeded, ensuring
the battery charger does not load down the AC adapter
voltage. This constant input current regulation allows the
adapter to fully power the system and prevent the AC
adapter from overloading and crashing the system bus.
An internal amplifier gm3 compares the voltage between
CSIP and CSIN to the input current limit threshold voltage
set by ACLIM. Connect ACLIM to REF, Float and GND for
the full-scale input current limit threshold voltage of 100mV,
75mV and 50mV, respectively, or use a resistor divider from
VREF to ground to set the input current limit as the following
equation:
IINPUT =
1  0.05

VACLIM + 0.050 

R2  VREF

When choosing the current sense resistor, note that the
voltage drop across this resistor causes further power
dissipation, reducing efficiency. The AC adapter current
sense accuracy is very important. Use a 1% tolerance
current-sense resistor. The highest accuracy of ±3% is
achieved with 100mV current-sense threshold voltage for
ACLIM=VREF, but it has the highest power dissipation. For
example, it has 400mW power dissipation for rated 4A AC
adapter and 1W sensing resistor may have to be used. ±4%
and ±6% accuracy can be achieved with 75mV and 50mV
current-sense threshold voltage for ACLIM=Floating and
ACLIM=GND, respectively.
A low pass filter is suggested to eliminate the switching
noise. Connect the resistor to CSIN pin instead of CSIP pin
because CSIN pin has lower bias current and less influence
on the current-sense accuracy.
AC Adapter Detection
Connect the AC adapter voltage through a resistor divider to
ACSET to detect when AC power is available, as shown in
Figure 12. ACPRN is an open-drain output and is high when
ACSET is less than Vth,rise, and active low when ACSET is
above Vth,fall. Vth,rise and Vth,fall are given by:

R
Vth ,rise =  8 + 1  • VACSET
R

 9
R

Vth,fall =  8 + 1  • V ACSET − I hys R8
 R9

Where Ihys is the ACSET input bias current hysteresis and
VACSET = 1.24V (min), 1.26V (typ.) and 1.28V (max.). The
hysteresis is IhysR8, where Ihys=2.2µA (min.), 3.4µA (typ.)
and 4.4µA (max.).
FN9202.1
June 17, 2005
ISL6251, ISL6251A
Current Measurement
Application Information
Use ICM to monitor the input current being sensed across
CSIP and CSIN. The output voltage range is 0 to 2.5V. The
voltage of ICM is proportional to the voltage drop across
CSIP and CSIN, and is given by the following equation:
The following battery charger design refers to the typical
application circuit in Figure 12, where typical battery
configuration of 4S2P is used. This section describes how to
select the external components including the inductor, input
and output capacitors, switching MOSFETs, and current
sensing resistors.
ICM = 19.9 • I INPUT • R 2
where IINPUT is the DC current drawn from the AC adapter.
ICM has ±3% accuracy.
A low pass filter connected to ICM output is used to filter the
switching noise.
LDO Regulator
VDD provides a 5.075V supply voltage from the internal LDO
regulator from DCIN and can deliver up to 30mA of current.
The MOSFET drivers are powered by VDDP, which must be
connected to VDDP as shown in Figure 12. VDDP connects
to VDD through an external resistor. Bypass VDDP and VDD
with a 1µF capacitor.
Shutdown
The ISL6251, ISL6251A features a low-power shutdown
mode. Driving EN low shuts down the charger. In shutdown,
the DC/DC converter is disabled, and VCOMP and ICOMP
are pulled to ground. The ICM, ACPRN outputs continue to
function.
EN can be driven by a thermistor to allow automatic
shutdown when the battery pack is hot. Often a NTC
thermistor is included inside the battery pack to measure its
temperature. When connected to the charger, the thermistor
forms a voltage divider with a resistive pull-up to the VREF.
The threshold voltage of EN is 1.06V with 60mV hysteresis.
The thermistor can be selected to have a resistance vs
temperature characteristic that abruptly decreases above a
critical temperature. This arrangement automatically shuts
down the charger when the battery pack is above a critical
temperature.
Inductor Selection
The inductor selection has trade-offs between cost, size and
efficiency. For example, the lower the inductance, the
smaller the size, but ripple current is higher. This also results
in higher AC losses in the magnetic core and the windings,
which decrease the system efficiency. On the other hand,
the higher inductance results in lower ripple current and
smaller output filter capacitors, but it has higher DCR (DC
resistance of the inductor) loss, and has slower transient
response. So, the practical inductor design is based on the
inductor ripple current being ±(15-20)% of the maximum
operating DC current at maximum input voltage. The
required inductance can be calculated from:
L=
VIN ,MAX − VBAT
∆ IL
VBAT
VIN ,MAX fs
Where VIN,MAX, VBAT, and fs are the maximum input
voltage, battery voltage and switching frequency,
respectively. The inductor ripple current ∆I is found from:
∆ I L = 30% ⋅ I BAT,MAX
where the maximum peak-to-peak ripple current is 30% of
the maximum charge current is used.
For VIN,MAX=19V, VBAT=16.8V, IBAT,MAX=2.6A, and
fs=300kHz, the calculated inductance is 8.3µH. Choosing
the closest standard value gives L=10µH. Ferrite cores are
often the best choice since they are optimized at 300kHz to
600kHz operation with low core loss. The core must be large
enough not to saturate at the peak inductor current IPeak:
Another method for inhibiting charging is to force CHLIM
below 88mV (Typ.).
I Peak = I BAT ,MAX +
Short Circuit Protection and 0V Battery Charging
Output Capacitor Selection
Since the battery charger will regulate the charge current to
the limit set by CHLIM, it automatically has short circuit
protection and is able to provide the charge current to wake
up an extremely discharged battery.
The output capacitor in parallel with the battery is used to
absorb the high frequency switching ripple current and
smooth the output voltage. The RMS value of the output
ripple current Irms is given by:
Over Temperature Protection
If the die temp exceeds 150°C, it stops charging. Once the
die temp drops below 125°C, charging will start up again.
14
IRMS =
VIN ,MAX
12 L fs
1
∆ IL
2
D (1 − D )
where the duty cycle D is the ratio of the output voltage
(battery voltage) over the input voltage for continuous
conduction mode which is typical operation for the battery
charger. During the battery charge period, the output voltage
varies from its initial battery voltage to the rated battery
voltage. So, the duty cycle change can be in the range of
FN9202.1
June 17, 2005
ISL6251, ISL6251A
between 0.53 and 0.88 for the minimum battery voltage of
10V (2.5V/Cell) and the maximum battery voltage of 16.8V.
resistance of the gate driver. The following switching loss
calculation provides a rough estimate.
For VIN,MAX=19V, VBAT=16.8V, L=10µH, and fs=300kHz,
the maximum RMS current is 0.19A. A typical 10F ceramic
capacitor is a good choice to absorb this current and also
has very small size. The tantalum capacitor has a known
failure mechanism when subjected to high surge current.
PQ1,Switching =
EMI considerations usually make it desirable to minimize
ripple current in the battery leads. Beads may be added in
series with the battery pack to increase the battery
impedance at 300kHz switching frequency. Switching ripple
current splits between the battery and the output capacitor
depending on the ESR of the output capacitor and battery
impedance. If the ESR of the output capacitor is 10mΩ and
battery impedance is raised to 2Ω with a bead, then only
0.5% of the ripple current will flow in the battery.
MOSFET Selection
The Notebook battery charger synchronous buck converter
has the input voltage from the AC adapter output. The
maximum AC adapter output voltage does not exceed 25V.
Therefore, 30V logic MOSFET should be used.
The high side MOSFET must be able to dissipate the
conduction losses plus the switching losses. For the battery
charger application, the input voltage of the synchronous
buck converter is equal to the AC adapter output voltage,
which is relatively constant. The maximum efficiency is
achieved by selecting a high side MOSFET that has the
conduction losses equal to the switching losses. Ensure that
ISL6251 LGATE gate driver can supply sufficient gate
current to prevent it from conduction, which is due to the
injected current into the drain-to-source parasitic capacitor
(Miller capacitor Cgd), and caused by the voltage rising rate
at phase node at the time instant of the high-side MOSFET
turning on; otherwise, cross-conduction problems may
occur. Reasonably slowing turn-on speed of the high-side
MOSFET by connecting a resistor between the BOOT pin
and gate drive supply source, and the high sink current
capability of the low-side MOSFET gate driver help reduce
the possibility of cross-conduction.
For the high-side MOSFET, the worst-case conduction
losses occur at the minimum input voltage:
PQ1,Conduction =
VOUT 2
I BAT R DSON
VIN
The optimum efficiency occurs when the switching losses
equal the conduction losses. However, it is difficult to
calculate the switching losses in the high-side MOSFET
since it must allow for difficult-to-quantify factors that
influence the turn-on and turn-off times. These factors
include the MOSFET internal gate resistance, gate charge,
threshold voltage, stray inductance, pull-up and pull-down
15
Qgd
Qgd
1
1
+ VIN ILP fs
+ QrrVIN fs
VIN ILV fs
2
Ig ,source 2
Ig ,sin k
Where Qgd: drain-to-gate charge, Qrr: total reverse recovery
charge of the body-diode in low side MOSFET, ILV: inductor
valley current, ILP: Inductor peak current, Ig,sink and
Ig,source are the peak gate-drive source/sink current of Q1,
respectively.
To achieve low switching losses, it requires low drain-to-gate
charge Qgd. Generally, the lower the drain-to-gate charge,
the higher the on-resistance. Therefore, there is a trade-off
between the on-resistance and drain-to-gate charge. Good
MOSFET selection is based on the Figure of Merit (FOM),
which is a product of the total gate charge and onresistance. Usually, the smaller the value of FOM, the higher
the efficiency for the same application.
For the low-side MOSFET, the worst-case power dissipation
occurs at minimum battery voltage and maximum input
voltage:

V
PQ2 = 1 − OUT
VIN

 2
 I BAT R DSON


Choose a low-side MOSFET that has the lowest possible
on-resistance with a moderate-sized package like the SO-8
and is reasonably priced. The switching losses are not an
issue for the low side MOSFET because it operates at zerovoltage-switching.
Choose a Schottky diode in parallel with low-side MOSFET
Q2 with a forward voltage drop low enough to prevent the
low-side MOSFET Q2 body-diode from turning on during the
dead time. This also reduces the power loss in the high-side
MOSFET associated with the reverse recovery of the lowside MOSFET Q2 body diode.
As a general rule, select a diode with DC current rating equal
to one-third of the load current. One option is to choose a
combined MOSFET with the Schottky diode in a single
package. The integrated packages may work better in
practice because there is less stray inductance due to a
short connection. This Schottky diode is optional and may be
removed if efficiency loss can be tolerated. In addition,
ensure that the required total gate drive current for the
selected MOSFETs should be less than 24mA. So, the total
gate charge for the high-side and low-side MOSFETs is
limited by the following equation:
QGATE ≤
I GATE
fs
Where IGATE is the total gate drive current and should be
less than 24mA. Substituting IGATE=24mA and fs=300kHz
into the above equation yields that the total gate charge
FN9202.1
June 17, 2005
ISL6251, ISL6251A
should be less than 80nC. Therefore, the ISL6251 easily
drives the battery charge current up to 10A.
Input Capacitor Selection
The input capacitor absorbs the ripple current from the
synchronous buck converter, which is given by:
Irms = IBAT
VOUT (VIN − VOUT )
VIN
This RMS ripple current must be smaller than the rated RMS
current in the capacitor datasheet. Non-tantalum chemistries
(ceramic, aluminum, or OSCON) are preferred due to their
resistance to power-up surge currents when the AC adapter
is plugged into the battery charger. For Notebook battery
charger applications, it is recommend that ceramic
capacitors or polymer capacitors from Sanyo be used due to
their small size and reasonable cost.
Table 2 shows the component lists for the typical application
circuit in Figure 12.
TABLE 2. COMPONENT LIST
PARTS
C1, C10
PART NUMBERS AND MANUFACTURER
10µF/25V ceramic capacitor, Taiyo Yuden
TMK325 MJ106MY X5R (3.2x2.5x1.9mm)
C2, C4, C8 0.1µF/50V ceramic capacitor
C3, C7, C9 1µF/10V ceramic capacitor, Taiyo Yuden
LMK212BJ105MG
Loop Compensation Design
ISL6251 uses constant frequency current mode control
architecture to achieve fast loop transient response.
Accurate current sensing resistors in series with the output
inductor is used to regulate the charge current, and the
sensed current signal is injected into the voltage loop to
achieve current mode control to simplify the loop
compensation design. The inductor is not considered as a
state variable for current mode control and the system
becomes single order system. It is much easier to design a
compensator to stabilize the voltage loop than voltage mode
control. Figure 14 shows the small signal model of the
synchronous buck regulator.
PWM Comparator Gain Fm:
The PWM comparator gain Fm for peak current mode
control is given by:
Fm =
d̂
v̂ comp
=
1
VPWM
Where VPWM is the peak-peak voltage of the PWM ramp
signal.
Current Sampling Transfer Function He(S):
In current loop, the current signal is sampled every switching
cycle. It has the following transfer function:
He (S ) =
S2
ωn2
+
S
ω nQn
+1
C5
10nF ceramic capacitor
C6
6.8nF ceramic capacitor
C11
3300pF ceramic capacitor
where Qn and ωn are given by Qn = − 2 , ωn=π fs,
π
respectively.
D1
30V/3A Schottky diode, EC31QS03L (optional)
Power Stage Transfer Functions
100mA/30V Schottky Diode, Central Semiconductor
Transfer function F1(S) from control to output voltage is:
D2, D3
D4
L
Q1, Q2
8A/30V Schottky rectifier, STPS8L30B (optional)
10µH/3.8A/26mΩ, Sumida, CDRH104R-100
30V/35mΩ, FDS6912A, Fairchild.
F1 (S ) =
v̂ o
d̂
1+
= Vin
S2
ω o2
+
S
ω esr
S
ω oQ p
+1
Q3
Signal N-channel MOSFET, 2N7002
R1
40mΩ, ±1%, LRC-LR2512-01-R040-F, IRC
Where ω esr =
R2
20mΩ, ±1%, LRC-LR2010-01-R020-F, IRC
Transfer function F2(S) from control to inductor current is:
R3
18Ω, ±5%, (0805)
R4
2.2Ω, ±5%, (0805)
R5
100kΩ, ±5%, (0805)
R6
10k, ±5%, (0805)
R7
100Ω, ±5%, (0805)
R8, R11
130k, ±1%, (0805)
R9
10.2kΩ, ±1%, (0805)
R10
4.7Ω, ±5%, (0805)
R12
20kΩ, ±1%, (0805)
R13
1.87kΩ, ±1%, (0805)
16
1
Co ω o =
, Q p ≈ Ro
,
Rc Co
L
Vin
=
F2 (S ) =
Ro + RL S 2
d̂
î L
1+
1
LC o
S
ωz
S
+
+1
2
ω
oQ p
ωo
, where ω z ≈
1
.
Ro Co
Current loop gain Ti(S) is expressed as the following
equation:
Ti ( S ) = RT Fm F2 (S )H e (S )
where RT is the trans-resistance in current loop. RT is
usually equal to the product of the current sensing resistance
of the current amplifier. For ISL6251, RT=20R1.
FN9202.1
June 17, 2005
ISL6251, ISL6251A
The voltage gain with open current loop is:
Tv ( S ) = KFm F1 (S )Av (S )
V
Where K = FB , VFB is the feedback voltage of the voltage
Vo
Av (S ) =
1+
v̂ comp
v̂ FB
where ω cz =
= gm
1
,
R1C1
S
ω cz
SC1
error amplifier. The Voltage loop gain with current loop
Compensator design goal:
closed is given by:
• High DC gain
Tv (S )
Lv ( S ) =
1 + Ti (S )
1 
1
−
 fs
 5 20 
• Loop bandwidth fc: 
• Gain margin: >10dB
If Ti(S)>>1, then it can be simplified as follows:
• Phase margin: 40°
S
The compensator design procedure is as follows:
1+
R + RL
ω esr Av (S )
V
1
Lv ( S ) = FB o
ω ≈
S H e (S ) , p R o C o
Vo
RT
1+
1. Put compensator zero at:
ωp
From the above equation, it is shown that the system is a
single order system, which has a single pole located at ω p
before the half switching frequency. Therefore, simple type II
compensator can be easily used to stabilize the system.
Figure 15 shows the voltage loop compensator, and its
transfer function is expressed as follows:
îL
v̂ in
L
1:D
ILd̂
2. Put one compensator pole at zero frequency to achieve
high DC gain, and put another compensator pole at either
ESR zero frequency or half switching frequency,
whichever is lower.
R1 =
Vind̂
+
RT
Rc
Ro
C1 =
Ti(S)
K
Fm
Tv(S)
He(S)
+
v̂ comp
-Av(S)
FIGURE 14. SMALL SIGNAL MODEL OF SYNCHRONOUS
BUCK REGULATOR
VREF
1
R1 ω cz
Example: Vin=19V, Vo=16.8V, Io=2.6A, fs=300kHz,
Co=10µF/10mΩ, L=10µH, gm=250µs, RT=0.2Ω, VFB=2.1V,
VPWM=VIN/11, fc=20kHz, then compensator resistance
R1=8.0kΩ. Choose R1=10kΩ. Put the compensator zero at
1.5kHz. The compensator capacitor is C1=10nF. Therefore,
choose voltage loop compensator: R1=10K, C1=10nF.
PCB Layout Considerations
Power and Signal Layers Placement on the PCB
As a general rule, power layers should be close together,
either on the top or bottom of the board, with signal layers on
the opposite side of the board. As an example, layer
arrangement on a 4-layer board is shown below:
Vo
VFB
2π fcVo Co RT
g mVFB
where gm is the trans-conductance of the voltage loop error
amplifier. Compensator capacitor C1 is then given by:
Co
d̂
1
RoCo
The loop gain Tv(S) at cross over frequency of fc has unity
gain. Therefore, the compensator resistance R1 is
determined by:
v̂ o
+
î in
ωcz = (1 − 3 )
+
gm
VCOMP
R1
1. Top Layer: signal lines, or half board for signal lines and
the other half board for power lines
2. Signal Ground
3. Power Layers: Power Ground
C1
FIGURE 15. VOLTAGE LOOP COMPENSATOR
17
4. Bottom Layer: Power MOSFET, Inductors and other
Power traces
Separate the power voltage and current flowing path from
the control and logic level signal path. The controller IC will
FN9202.1
June 17, 2005
ISL6251, ISL6251A
stay on the signal layer, which is isolated by the signal
ground to the power signal traces.
Component Placement
The power MOSFET should be close to the IC so that the
gate drive signal, the LGATE, UGATE, PHASE, and BOOT,
traces can be short.
UGATE Pin
This pin has a square shape waveform with high dv/dt. It
provides the gate drive current to charge and discharge the
top MOSFET with high di/dt. This trace should be wide,
short, and away from other traces similar to the LGATE.
BOOT Pin
Place the components in such a way that the area under the
IC has less noise traces with high dv/dt and di/dt, such as
gate signals and phase node signals.
This pin’s di/dt is as high as the UGATE; therefore, this trace
should be as short as possible.
Signal Ground and Power Ground Connection.
At minimum, a reasonably large area of copper, which will
shield other noise couplings through the IC, should be used
as signal ground beneath the IC. The best tie-point between
the signal ground and the power ground is at the negative
side of the output capacitor on each side, where there is little
noise; a noisy trace beneath the IC is not recommended.
The current sense resistor connects to the CSON and the
CSOP pins through a low pass filter. The CSON pin is also
used as the battery voltage feedback. The traces should be
away from the high dv/dt and di/di pins like PHASE, BOOT
pins. In general, the current sense resistor should be close
to the IC. Other layout arrangements should be adjusted
accordingly.
GND and VDD Pin
EN Pin
At least one high quality ceramic decoupling cap should be
used to cross these two pins. The decoupling cap can be put
close to the IC.
This pin stays high at enable mode and low at idle mode and
is relatively robust. Enable signals should refer to the signal
ground.
LGATE Pin
DCIN Pin
This is the gate drive signal for the bottom MOSFET of the
buck converter. The signal going through this trace has both
high dv/dt and high di/dt, and the peak charging and
discharging current is very high. These two traces should be
short, wide, and away from other traces. There should be no
other traces in parallel with these traces on any layer.
This pin connects to AC adapter output voltage, and should
be less noise sensitive.
PGND Pin
PGND pin should be laid out to the negative side of the
relevant output cap with separate traces. The negative side
of the output capacitor must be close to the source node of
the bottom MOSFET. This trace is the return path of LGATE.
PHASE Pin
This trace should be short, and positioned away from other
weak signal traces. This node has a very high dv/dt with a
voltage swing from the input voltage to ground. No trace
should be in parallel with it. This trace is also the return path
for UGATE. Connect this pin to the high-side MOSFET
source.
18
CSOP, CSON Pins
Copper Size for the Phase Node
The capacitance of PHASE should be kept very low to
minimize ringing. It would be best to limit the size of the
PHASE node copper in strict accordance with the current
and thermal management of the application.
Identify the Power and Signal Ground
The input and output capacitors of the converters, the source
terminal of the bottom switching MOSFET PGND should
connect to the power ground. The other components should
connect to signal ground. Signal and power ground are tied
together at one point.
Clamping Capacitor for Switching MOSFET
It is recommended that ceramic caps be used closely
connected to the drain of the high-side MOSFET, and the
source of the low-side MOSFET. This capacitor reduces the
noise and the power loss of the MOSFET.
FN9202.1
June 17, 2005
ISL6251, ISL6251A
Quad Flat No-Lead Plastic Package (QFN)
Micro Lead Frame Plastic Package (MLFP)
2X
9
MILLIMETERS
D/2
D1
D1/2
2X
N
6
INDEX
AREA
28 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
(COMPLIANT TO JEDEC MO-220VHHD-1 ISSUE I)
0.15 C A
D
A
L28.5x5
0.15 C B
SYMBOL
MIN
NOMINAL
MAX
NOTES
A
0.80
0.90
1.00
-
A1
-
0.02
0.05
-
A2
-
0.65
1.00
9
0.30
5,8
A3
1
2
3
E1/2
E/2
E1
b
E
9
2X
2X
0.15 C A
5.00 BSC
-
4.75 BSC
9
0
4X
E2
A
A1
A3
SIDE VIEW
D2
(DATUM B)
4.75 BSC
2.95
3.10
9
3.25
7,8
0.20
-
-
-
L
0.50
0.60
0.75
8
9
NX k
D2
2 N
-
k
8
7
7,8
e
0.10 M C A B
4X P
3.25
0.08 C
5
NX b
3.10
5.00 BSC
/ / 0.10 C
C
SEATING PLANE
2.95
E1
A2
9
D
E
B
TOP VIEW
0.25
D1
D2
0.15 C B
0.20 REF
0.18
0.50 BSC
-
N
28
2
Nd
7
3
Ne
7
3
P
-
-
0.60
9
θ
-
-
12
9
4X P
Rev. 1 11/04
1
(DATUM A)
NOTES:
2
3
6
INDEX
AREA
1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
(Ne-1)Xe
REF.
E2
2. N is the number of terminals.
7
E2/2
NX L
4. All dimensions are in millimeters. Angles are in degrees.
N e
8
3. Nd and Ne refer to the number of terminals on each D and E.
8
9
CORNER
OPTION 4X
(Nd-1)Xe
REF.
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
BOTTOM VIEW
A1
NX b
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
5
C
L
C
L
L1
10
L
L1
e
C C
8. Nominal dimensions are provided to assist with PCB Land Pattern
Design efforts, see Intersil Technical Brief TB389.
SECTION "C-C"
10
L
9. Features and dimensions A2, A3, D1, E1, P & θ are present when
Anvil singulation method is used and not present for saw
singulation.
e
TERMINAL TIP
FOR ODD TERMINAL/SIDE
FOR EVEN TERMINAL/SIDE
19
FN9202.1
June 17, 2005
ISL6251, ISL6251A
Shrink Small Outline Plastic Packages (SSOP)
Quarter Size Outline Plastic Packages (QSOP)
M24.15
N
INDEX
AREA
H
0.25(0.010) M
E
2
SYMBOL
3
0.25
0.010
SEATING PLANE
-A-
INCHES
GAUGE
PLANE
-B1
24 LEAD SHRINK SMALL OUTLINE PLASTIC PACKAGE
(0.150” WIDE BODY)
B M
A
D
L
h x 45°
-C-
α
e
A2
A1
B
C
0.10(0.004)
0.17(0.007) M
C A M
B S
NOTES:
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2
of Publication Number 95.
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension “D” does not include mold flash, protrusions or gate
burrs. Mold flash, protrusion and gate burrs shall not exceed
0.15mm (0.006 inch) per side.
4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch)
per side.
5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. Dimension “B” does not include dambar protrusion. Allowable dambar protrusion shall be 0.10mm (0.004 inch) total in excess of “B”
dimension at maximum material condition.
10. Controlling dimension: INCHES. Converted millimeter dimensions
are not necessarily exact.
MIN
MAX
MILLIMETERS
MIN
MAX
NOTES
A
0.053
0.069
1.35
1.75
-
A1
0.004
0.010
0.10
0.25
-
A2
-
0.061
-
1.54
-
B
0.008
0.012
0.20
0.30
9
C
0.007
0.010
0.18
0.25
-
D
0.337
0.344
8.55
8.74
3
E
0.150
0.157
3.81
3.98
4
e
0.025 BSC
0.635 BSC
-
H
0.228
0.244
5.80
6.19
-
h
0.0099
0.0196
0.26
0.49
5
L
0.016
0.050
0.41
1.27
6
N
α
24
0°
24
8°
0°
7
8°
Rev. 2 6/04
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Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
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20
FN9202.1
June 17, 2005