19-4722; Rev 2; 11/09 TION KIT EVALUA BLE A IL A AV Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller Features The MAX16814 high-efficiency, high-brightness LED (HB LED) driver provides up to four integrated LED currentsink channels. An integrated current-mode switching DC-DC controller drives a DC-DC converter that provides the necessary voltage to multiple strings of HB LEDs. The MAX16814 accepts a wide 4.75V to 40V input voltage range and withstands direct automotive loaddump events. The wide input range allows powering HB LEDs for small to medium-sized LCD displays in automotive and general lighting applications. S 4-Channel Linear LED Current Sinks with Internal MOSFETs Full-Scale LED Current Adjustable from 20mA to 150mA Drives One to Four LED Strings An internal current-mode switching DC-DC controller supports the boost, coupled-inductor boost-buck, or SEPIC topologies and operates in an adjustable frequency range between 200kHz and 2MHz. It can also be used for single-inductor boost-buck topology in conjunction with the MAX15054 and an additional MOSFET. The current-mode control with programmable slope compensation provides fast response and simplifies loop compensation. The MAX16814 also features an adaptive output-voltage control scheme that minimizes the power dissipation in the LED current-sink paths. The MAX16814 consists of four identical linear current source channels to drive four strings of HB LEDs. The channel current is adjustable from 20mA to 150mA with an accuracy of ±3% using an external resistor. The external resistor sets all 4-channel currents to the same value. The device allows connecting multiple channels in parallel to achieve higher current per LED string. The MAX16814 also features pulsed dimming control, on all four channels through a logic input (DIM). In addition,the MAX16814A_ _ and MAX16814U_ _ include a unique feature that allows a very short minimum pulse width as low as 1µs. The MAX16814 includes an output overvoltage, openLED detection and protection, programmable shorted LED detection and protection, and overtemperature protection. The device operates over the -40NC to +125NC automotive temperature range. The MAX16814 is available in the 6.5mm x 4.4mm, 20-pin TSSOP and 4mm x 4mm, 20-pin TQFN packages. S Boost, SEPIC, or Coupled-Inductor Boost-Buck Current-Mode DC-DC Controller 200kHz to 2MHz Programmable Switching Frequency External Switching Frequency Synchronization S Adaptive Output-Voltage Optimization to Minimize Power Dissipation S 4.75V to 40V Operating Input Voltage Range S Less than 40µA Shutdown Current S 5000:1 PWM Dimming at 200Hz (MAX16814A _ _ and MAX16814U_ _ Only) S Open-Drain Fault Indicator Output S Open-LED and LED Short Detection and Protection S Overtemperature Protection S Thermally Enhanced, 20-Pin TQFN and TSSOP Packages Ordering Information TEMP RANGE PIN-PACKAGE MAX16814ATP+ PART -40°C to +125°C 20 TQFN-EP* MAX16814ATP/V+ -40°C to +125°C 20 TQFN-EP* MAX16814AUP+ -40°C to +125°C 20 TSSOP-EP* MAX16814BUTP+ 0°C to +85°C 20 TQFN-EP* MAX16814BUUP+ 0°C to +85°C 20 TSSOP-EP* MAX16814UTP+ 0°C to +85°C 20 TQFN-EP* 0°C to +85°C 20 TSSOP-EP* MAX16814UUP+ +Denotes a lead(Pb)-free/RoHS-compliant package. *EP = Exposed pad. /V Denotes an automotive qualified part. Applications Automotive Displays LED Backlights Automotive RCL, DRL, Front Position, and Fog Lights Typical Operating Circuit and Pin Configurations appear at end of data sheet. LCD TV and Desktop Display LED Backlights Architectural, Industrial, and Ambient Lighting ________________________________________________________________ Maxim Integrated Products 1 For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. MAX16814 General Description MAX16814 Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller ABSOLUTE MAXIMUM RATINGS IN to SGND.............................................................-0.3V to +45V EN to SGND................................................-0.3V to (VIN + 0.3V) PGND to SGND.....................................................-0.3V to +0.3V LEDGND to SGND................................................-0.3V to +0.3V OUT_ to LEDGND..................................................-0.3V to +45V VCC to SGND........... -0.3V to the lower of (VIN + 0.3V) and +6V DRV, FLT, DIM, RSDT, OVP to SGND......................-0.3V to +6V CS, RT, COMP, SETI to SGND.................. -0.3V to (VCC + 0.3V) NDRV to PGND........................................-0.3V to (VDRV + 0.3V) NDRV Peak Current (< 100ns).............................................. Q3A NDRV Continuous Current............................................. Q100mA OUT_ Continuous Current.............................................. Q175mA VCC Short-Circuit Duration.........................................Continuous Continuous Power Dissipation (TA = +70NC) (Note 1) 20-Pin TQFN (derate 25.6mW/NC above +70NC)........2051mW Junction-to-Case Thermal Resistance (BJC)...................6NC/W Junction-to-Ambient Thermal Resistance (BJA)............39NC/W 20-Pin TSSOP (derate 26.5mW/NC above +70NC)......2122mW Junction-to-Case Thermal Resistance (BJC)................2.0NC/W Junction-to-Ambient Thermal Resistance (BJA).........37.7NC/W Operating Temperature Range MAX16814A_ _............................................... -40NC to +125NC MAX16814U_ _and MAX16814BU_ _.................0NC to +85NC Junction Temperature......................................................+150NC Storage Temperature Range............................. -65NC to +150NC Lead Temperature (soldering, 10s).................................+300NC Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer board. For detailed information on package thermal considerations, refer to http://www.maxim-ic.com/thermal-tutorial. Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (VIN = VEN = 12V, RRT = 12.25kI, RSETI = 15kI, CVCC = 1FF, VCC = VDRV, NDRV = COMP = OUT_ = unconnected, VRSDT = VDIM = VCC, VOVP = VCS = VLEDGND = VPGND = VSGND = 0V, TA = TJ = -40NC to +125NC for MAX16814A_ _ and TA = TJ = 0NC to +85NC for MAX16814U_ _ and MAX16814BU_ _, unless otherwise noted. Typical values are at TA = +25NC.) (Note 2) PARAMETER SYMBOL Operating Voltage Range VIN Active Supply Current IIN CONDITIONS MIN TYP 4.75 MAX UNITS 40 V MAX16814A_ _ and MAX16814U_ _ 2.5 5 MAX16814BU_ _ only 2.75 5.5 Standby Supply Current VEN = 0V IN Undervoltage Lockout VIN rising 3.975 IN UVLO Hysteresis mA 15 40 μA 4.3 4.625 V 170 mV VCC REGULATOR Regulator Output Voltage VCC Dropout Voltage 6.5V < VIN < 10V, 1mA < ILOAD < 50mA 10V < VIN < 40V, 1mA < ILOAD < 10mA 4.75 VIN - VCC, VIN = 4.75V, ILOAD = 50mA VCC shorted to SGND Short-Circuit Current Limit VCC Undervoltage Lockout Threshold VCC rising VCC UVLO Hysteresis 5.0 5.25 V 200 500 mV 100 mA 4 V 100 mV RT OSCILLATOR Switching Frequency Range fSW 200 2 _______________________________________________________________________________________ 2000 kHz Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller (VIN = VEN = 12V, RRT = 12.25kI, RSETI = 15kI, CVCC = 1FF, VCC = VDRV, NDRV = COMP = OUT_ = unconnected, VRSDT = VDIM = VCC, VOVP = VCS = VLEDGND = VPGND = VSGND = 0V, TA = TJ = -40NC to +125NC for MAX16814A_ _ and TA = TJ = 0NC to +85NC for MAX16814U_ _ and MAX16814BU_ _ , unless otherwise noted. Typical values are at TA = +25NC.) (Note 2) PARAMETER SYMBOL Maximum Duty Cycle Oscillator Frequency Accuracy CONDITIONS MIN TYP MAX fSW = 200kHz to 600kHz, MAX16814A_ _ and MAX16814U_ _ 85 89 93 fSW = 600kHz to 2000kHz, MAX16814A_ _ and MAX16814U_ _ 82 86 90 fSW = 200kHz to 600kHz, MAX16814BU_ 90 94 98 fSW = 600kHz to 2000kHz, MAX16814BU_ _ 86 90 94 fSW = 200kHz to 2MHz, MAX16814A_ _ and MAX16814U_ _ -7.5 +7.5 -7 +7 fSW = 200kHz to 2MHz, MAX16814BU_ _ Sync Rising Threshold Minimum Sync Frequency UNITS % % 4 V 1.1fSW Hz PWM COMPARATOR PWM Comparator Leading-Edge Blanking Time PWM to NDRV Propagation Delay Including leading-edge blanking time 60 ns 90 ns SLOPE COMPENSATION Current ramp added to the CS input, MAX16814A_ _ only 44 Current ramp added to the CS input, MAX16814U_ _ and MAX16814BU_ _ 45 50 55 Current-Limit Threshold (Note 3) 396 416 437 CS Limit Comparator to NDRV Propagation Delay 10mV overdrive, excluding leading-edge blanking time Peak Slope Compensation Current Ramp Magnitude 49 54 μA x fSW CS LIMIT COMPARATOR 10 mV ns ERROR AMPLIFIER OUT_ Regulation Voltage Transconductance 1 gM No-Load Gain (Note 4) COMP Sink Current VOUT_ = 5V, VCOMP = 2.5V VOUT_ = 0V, VCOMP = 2.5V COMP Source Current V 340 600 880 160 375 800 μA 160 375 800 μA 75 μS dB MOSFET DRIVER ISINK = 100mA (nMOS) 0.9 ISOURCE = 100mA (pMOS) 1.1 Peak Sink Current VNDRV = 5V 2.0 Peak Source Current VNDRV = 0V 2.0 A Rise Time CLOAD = 1nF 6 ns Fall Time CLOAD = 1nF 6 ns NDRV On-Resistance ω A _______________________________________________________________________________________ 3 MAX16814 ELECTRICAL CHARACTERISTICS (continued) MAX16814 Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller ELECTRICAL CHARACTERISTICS (continued) (VIN = VEN = 12V, RRT = 12.25kI, RSETI = 15kI, CVCC = 1FF, VCC = VDRV, NDRV = COMP = OUT_ = unconnected, VRSDT = VDIM = VCC, VOVP = VCS = VLEDGND = VPGND = VSGND = 0V, TA = TJ = -40NC to +125NC for MAX16814A_ _ and TA = TJ = 0NC to +85NC for MAX16814U_ _ and MAX16814BU_ _, unless otherwise noted. Typical values are at TA = +25NC.) (Note 2) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS 150 mA ±2 % LED CURRENT SOURCES OUT_ Current-Sink Range VOUT_ = VREF Channel-to-Channel Matching IOUT_ = 100mA IOUT_ = 100mA Output Current Accuracy OUT_ Leakage Current 20 TA = +125°C, MAX16814A_ _ only ±3 TA = -40°C to +125°C, MAX16814A_ _ only ±5 mA TA = +25°C, MAX16814U_ _ and IOUT_ = MAX16814BU_ _ 50mA to 150mA TA = 0°C to +85°C, MAX16814U_ _ and MAX16814BU_ _ VDIM = 0V, VOUT_ = 40V ±2.75 mA ±4 1 μA LOGIC INPUTS/OUTPUTS EN Reference Voltage VEN rising, MAX16814A_ _ only 1.125 1.23 1.335 VEN rising, MAX16814U_ _ and MAX16814BU_ _ 1.144 1.23 1.316 EN Hysteresis EN Input Current 50 mV VEN = 40V, MAX16814A_ _ only ±200 VEN = 40V, MAX16814U_ _ and MAX16814BU_ _ ±250 DIM Input High Voltage 2.1 nA V DIM Input Low Voltage 0.8 DIM Hysteresis V 250 DIM Input Current V mV ±2 μA DIM to LED Turn-On Delay DIM rising edge to 10% rise in IOUT_ 100 ns DIM to LED Turn-Off Delay DIM falling edge to 10% fall in IOUT_ 100 ns IOUT_ Rise and Fall Times 200 ns FLT Output Low Voltage VIN = 4.75V and ISINK = 5mA 0.4 V FLT Output Leakage Current LED Short Detection Threshold VFLT = 5.5V Gain = 3 1.0 μA 2.25 V ±600 nA 1.75 Short Detection Comparator Delay 2.0 6.5 RSDT Leakage Current OVP Trip Threshold Output rising OVP Hysteresis 1.228 1.266 70 OVP Leakage Current VOVP = 1.25V Thermal-Shutdown Threshold Temperature rising Thermal-Shutdown Hysteresis 1.19 μs V mV ±200 nA 165 °C 15 °C Note 2: All MAX16814A_ _ are 100% tested at TA = +125NC, while alll MAX16814U_ _ and MAX16814BU_ _ are 100% tested at TA = +25°C. All limits overtemperature are guaranteed by design , not production tested. Note 3: CS threshold includes slope compensation ramp magnitude. Note 4: Gain = δVCOMP/δVCS, 0.05V < VCS < 0.15V. 4 _______________________________________________________________________________________ Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage Boost and SEPIC Controller SWITCHING WAVEFORM AT 5kHz (50% DUTY CYCLE) DIMMING SUPPLY CURRENT vs. SUPPLY VOLTAGE MAX16814 toc01 CNDRV = 13pF 3.4 0V TA = +25NC 3.2 IIN (mA) IOUT1 100mA/div 3.0 0A FIGURE 2 TA = +125NC 3.6 VLX 10V/div MAX16814 toc02 3.8 VOUT 10V/div 2.8 0V 2.4 TA = -40NC 2.6 5 40Fs/div 10 15 20 25 30 35 40 45 VIN (V) SWITCHING FREQUENCY vs. TEMPERATURE 3.6 3.4 3.2 306 304 302 300 298 1.232 1.228 296 1.224 294 292 1.220 290 3.0 -50 200 400 600 800 1000 1200 1400 1600 1800 2000 -25 0 25 50 75 100 -25 0 25 50 75 TEMPERATURE (NC) TEMPERATURE (NC) VSETI vs. PROGRAMMED CURRENT EN THRESHOLD VOLTAGE vs. TEMPERATURE EN LEAKAGE CURRENT vs. TEMPERATURE 1.231 1.230 VEN RISING 1.20 VEN FALLING 1.15 46 72 98 124 LED STRING CURRENT (mA) 150 100 125 VEN = 2.5V 120 90 60 30 1.229 1.228 125 MAX16814 toc08 1.25 100 150 EN LEAKAGE CURRENT (nA) 1.232 1.30 EN THRESHOLD VOLTAGE (V) MAX16814 toc06 1.233 20 -50 125 fSW (kHz) 1.234 VSETI (V) 1.236 VSETI (V) 3.8 308 MAX16814 toc07 IIN (mA) 4.0 1.240 MAX16814 toc04 CNDRV = 13pF 4.2 VSETI vs. TEMPERATURE 310 SWITCHING FREQUENCY (kHz) MAX16814 toc03 4.4 MAX16814 toc05 SUPPLY CURRENT vs. SWITCHING FREQUENCY 1.10 0 -50 -25 0 25 50 75 TEMPERATURE (NC) 100 125 -50 -25 0 25 50 75 TEMPERATURE (NC) _______________________________________________________________________________________ 5 MAX16814 Typical Operating Characteristics (VIN = VEN = 12V, fSW = 300kHz, RSETI = 15kI, CVCC = 1FF, VCC = VDRV, NDRV = COMP = OUT_ = unconnected, VOVP = VCS = VLEDGND = VDIM = VPGND = VSGND = 0V, load = four strings of seven white LEDs, TA = +25NC, unless otherwise noted.) Typical Operating Characteristics (continued) (VIN = VEN = 12V, fSW = 300kHz, RSETI = 15kI, CVCC = 1FF, VCC = VDRV, NDRV = COMP = OUT_ = unconnected, VOVP = VCS = VLEDGND = VDIM = VPGND = VSGND = 0V, load = four strings of seven white LEDs, TA = +25NC, unless otherwise noted.) VCC (V) 5.02 TA = +125NC 5.04 TA = +25NC 5.00 4.98 TA = -40NC 5.00 5.02 4.96 TA = -40NC 4.94 4.98 1.80 10 15 20 25 30 35 0 40 1.40 1.20 1.00 0.80 0.60 0.20 4.90 5 1.60 0.40 4.92 4.96 MAX16814 toc11 5.06 2.00 SWITCHING FREQUENCY (MHz) 5.08 TA = +125NC TA = +25NC MAX16814 toc10 5.10 MAX16814 toc09 5.06 5.04 SWITCHING FREQUENCY vs. 1/RT VCC LOAD REGULATION VCC LINE REGULATION 5.08 VCC (V) 20 VIN (V) STARTUP WAVEFORM WITH DIM ON PULSE WIDTH < tSW MAX16814 toc12 40 IVCC (mA) 80 60 0.02 0.06 0.10 0.14 VIN 20V/div 0V MAX16814 toc13 VDIM 5V/div 0V IOUT_ 100mA/div 0A IOUT1 100mA/div 0A VLED 10V/div FIGURE 2 0V 40ms/div 40ms/div STARTUP WAVEFORM WITH DIM CONTINOUSLY ON MOSFET DRIVER ON-RESISTANCE vs. TEMPERATURE IOUT1 100mA/div 0A VLED 10V/div 40ms/div 0V 0V MAX16814 toc15 1.5 VIN 20V/div 0V VDIM 5V/div 0V FIGURE 2 VIN 20V/div 0V VDIM 5V/div 0V VLED 20V/div MAX16814 toc14 0.18 1/RT (mS) STARTUP WAVEFORM WITH DIM ON PULSE WIDTH = 10tSW 1.3 ON-RESISTANCE (I) MAX16814 Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage Boost and SEPIC Controller pMOS 1.1 0.9 nMOS 0.7 0.5 -50 -25 0 25 50 75 100 125 TEMPERATURE (NC) 6 _______________________________________________________________________________________ 0.22 0.26 0.30 Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller LED CURRENT RISING AND FALLING WAVEFORM LED CURRENT SWITCHING WITH DIM AT 5kHz AND 50% DUTY CYCLE MAX16814 toc17 MAX16814 toc16 FIGURE 2 IOUT1 100mA/div 0A VDIM 5V/div 0V IOUT2 100mA/div 0A IOUT3 100mA/div 0A ILED 50mA/div 0A IOUT4 100mA/div 0A 100Fs/div 4Fs/div OUT_ CURRENT vs. 1/RSETI COMP LEAKAGE CURRENT vs. TEMPERATURE VDIM = 0V COMP LEAKAGE CURRENT (nA) 140 120 100 80 60 40 0.040 0.055 0.070 0.085 0.6 VCOMP = 4.5V 0.4 VCOMP = 0.5V 0.2 0 0.100 -25 -50 0 1/RSETI (mS) OUT_ LEAKAGE CURRENT vs. TEMPERATURE 75 10 1 100 125 RSDT LEAKAGE CURRENT vs. TEMPERATURE MAX16814 toc21 2.0 VOVP = 1.25V 1.8 OVP LEAKAGE CURRENT (nA) VDIM = 0V VOUT = 40V 50 OVP LEAKAGE CURRENT vs. TEMPERATURE MAX16814 toc20 OUT_ LEAKAGE CURRENT (nA) 100 25 TEMPERATURE (NC) 1.6 1.4 1.2 1.0 0.8 0.6 0.4 250 MAX16814 toc22 0.025 0.8 RSDT LEAKAGE CURRENT (nA) 20 0.010 MAX16814 toc19 1.0 MAX16814 toc18 160 IOUT_ (mA) FIGURE 2 200 VRSDT = 0.5V 150 100 VRSDT = 2.5V 0.2 0.1 -50 -25 0 25 50 75 TEMPERATURE (NC) 100 125 50 0 -50 -25 0 25 50 75 TEMPERATURE (NC) 100 125 -50 -25 0 25 50 75 100 125 TEMPERATURE (NC) _______________________________________________________________________________________ 7 MAX16814 Typical Operating Characteristics (continued) (VIN = VEN = 12V, fSW = 300kHz, RSETI = 15kI, CVCC = 1FF, VCC = VDRV, NDRV = COMP = OUT_ = unconnected, VOVP = VCS = VLEDGND = VDIM = VPGND = VSGND = 0V, load = four strings of seven white LEDs, TA = +25NC, unless otherwise noted.) MAX16814 Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller Pin Description PIN NAME FUNCTION TQFN TSSOP 1 4 IN Bias Supply Input. Connect a 4.75V to 40V supply to IN. Bypass IN to SGND with a ceramic capacitor. 2 5 EN Enable Input. Connect EN to logic-low to shut down the device. Connect EN to logic-high or IN for normal operation. The EN logic threshold is internally set to 1.23V. 3 6 COMP Switching Converter Compensation Input. Connect the compensation network from COMP to SGND for current-mode control (see the Feedback Compensation section). 4 7 RT Oscillator Timing Resistor Connection. Connect a timing resistor (RT) from RT to SGND to program the switching frequency according to the formula RT = 7.350 x 109/fsw (for the MAX16814A_ _ and the MAX16814U_ _) or to the formula RT = 7.72 x 109/fsw (for the MAX16814BU_ _). Apply an AC-coupled external clock at RT to synchronize the switching frequency with an external clock. 5 8 FLT Open-Drain Fault Output. FLT asserts low when an open LED, short LED, or thermal shutdown is detected. Connect a 10kω pullup resistor from FLT to VCC. 6 9 OVP Overvoltage Threshold Adjust Input. Connect a resistor-divider from the switching converter output to OVP and SGND. The OVP comparator reference is internally set to 1.23V. 7 10 SETI LED Current Adjust Input. Connect a resistor (RSETI) from SETI to SGND to set the current through each LED string (ILED) according to the formula ILED = 1500/RSETI. 8 11 RSDT LED Short Detection Threshold Adjust Input. Connect a resistive divider from VCC to RSDT and SGND to program the LED short detection threshold. Connect RSDT directly to VCC to disable LED short detection. The LED short detection comparator is internally referenced to 2V. 9 12 SGND Signal Ground. SGND is the current return path connection for the low-noise analog signals. Connect SGND, LEDGND, and PGND at a single point. 10 13 DIM Digital PWM Dimming Input. Apply a PWM signal to DIM for LED dimming control. Connect DIM to VCC if dimming control is not used. 11 14 OUT1 LED String Cathode Connection 1. OUT1 is the open-drain output of the linear current sink that controls the current through the LED string connected to OUT1. OUT1 sinks up to 150mA. If unused, connect OUT1 to LEDGND. 8 _______________________________________________________________________________________ Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller PIN NAME FUNCTION 15 OUT2 LED String Cathode Connection 2. OUT2 is the open-drain output of the linear current sink that controls the current through the LED string connected to OUT2. OUT2 sinks up to 150mA. If unused, connect OUT2 to LEDGND. 13 16 LEDGND 14 17 OUT3 LED String Cathode Connection 3. OUT3 is the open-drain output of the linear current sink that controls the current through the LED string connected to OUT3. OUT3 sinks up to 150mA. If unused, connect OUT3 to LEDGND. 15 18 OUT4 LED String Cathode Connection 4. OUT4 is the open-drain output of the linear current sink that controls the current through the LED string connected to OUT4. OUT4 sinks up to 150mA. If unused, connect OUT4 to LEDGND. Current-Sense Input. CS is the current-sense input for the switching regulator. A sense resistor connected from the source of the external power MOSFET to PGND sets the switching current limit. A resistor connected between the source of the power MOSFET and CS sets the slope compensation ramp rate (see the Slope Compensation section). TQFN TSSOP 12 LED Ground. LEDGND is the return path connection for the linear current sinks. Connect SGND, LEDGND, and PGND at a single point. 16 19 CS 17 20 PGND Power Ground. PGND is the switching current return path connection. Connect SGND, LEDGND, and PGND at a single point. 18 1 NDRV Switching n-MOSFET Gate-Driver Output. Connect NDRV to the gate of the external switching power MOSFET. 19 2 DRV MOSFET Gate-Driver Supply Input. Connect a resistor between VCC and DRV to power the MOSFET driver with the internal 5V regulator. Bypass DRV to PGND with a minimum of 0.1μF ceramic capacitor. 20 3 VCC 5V Regulator Output. Bypass VCC to SGND with a minimum of 1μF ceramic capacitor as close as possible to the device. — — EP Exposed Pad. Connect EP to a large-area contiguous copper ground plane for effective power dissipation. Do not use as the main IC ground connection. EP must be connected to SGND. _______________________________________________________________________________________ 9 MAX16814 Pin Description (continued) MAX16814 Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage Boost and SEPIC Controller FLT RSDT VREF MAX16814 SHORT LED DETECTOR FAULT FLAG LOGIC POKD UNUSED STRING DETECTOR OPEN-LED DETECTOR SHDN DRV TSHDN PWM LOGIC NDRV PGND CLK SLOPE COMPENSATION RAMP/RT OSC RT MIN STRING VOLTAGE ILIM OUT_ 0.425V 1.8V di ( dt = 50FA x fsw) CS BLANKING CS COMP OVP COMP THERMAL SHUTDOWN R LOGIC gM TSHDN REF FB VBG SHDN BANDGAP IN UVLO LEDGND LOGIC (REF/FB SELECTOR) VBG = 1.235V DIM 5V LDO REGULATOR VCC SS_DONE SS_REF UVLO VREF TSHDN POK SOFT-START 100ms SHDN POKD VBG P EN SHDN 1.23V TSHDN SGND SGND OVP SETI Figure 1. Simplified Functional Diagram 10 ������������������������������������������������������������������������������������� Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller L2 10FH C6 10FF D1 B160B-1 L1 22FH C5 47FF 7 HBLEDS PER STRING C2 1FF C1 1FF M1 IRF7468 R1 261kI C8 1nF R2 24kI Q10% D2 IN R7 22I NDRV RCOMP 24kI C7 33FF RCS 0.1I CS OVP EN OUT1 VCC C3 1FF OUT2 OUT3 R5 4.7I MAX16814 VDRV OUT4 RSETI 15kI C4 0.1FF SETI R6 50kI DIM VCC FLT COMP R3 30.1kI RSDT RCOMP 14kI R4 20kI RT RT 21kI CCOMP 0.022FF SGND PGND LEDGND Figure 2. Circuit Used for Typical Operating Characteristics ______________________________________________________________________________________ 11 MAX16814 VIN MAX16814 Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller Detailed Description The MAX16814 high-efficiency HB LED driver integrates all the necessary features to implement a highperformance backlight driver to power LEDs in small to medium-sized displays for automotive as well as general applications. The device provides load-dump voltage protection up to 40V in automotive applications. The MAX16814 incorporates two major blocks: a DC-DC controller with peak current-mode control to implement a boost, coupled-inductor boost-buck, or a SEPIC-type switched-mode power supply and a 4-channel LED driver with 20mA to 150mA constant current-sink capability per channel. Figure 1 is the simplified functional diagram and Figure 2 shows the circuit used for typical operating characteristics. The MAX16814 features a constant-frequency peak current-mode control with programmable slope compensation to control the duty cycle of the PWM controller. The high-current FET driver can provide up to 2A of current to the external n-channel MOSFET. The DC-DC converter implemented using the controller generates the required supply voltage for the LED strings from a wide input supply range. Connect LED strings from the DC-DC converter output to the 4-channel constant current-sink drivers that control the current through the LED strings. A single resistor connected from the SETI input to ground adjusts the forward current through all four LED strings. The MAX16814 features adaptive voltage control that adjusts the converter output voltage depending on the forward voltage of the LED strings. This feature minimizes the voltage drop across the constant current-sink drivers and reduces power dissipation in the device. A logic input (EN) shuts down the device when pulled low. The device includes an internal 5V LDO capable of powering additional external circuitry. All the versions of the MAX16814 include PWM dimming. The MAX16814A_ and the MAX16814U_ versions, in particular, provide very wide (5000:1) PWM dimming range where a dimming pulse as narrow as 1µs is possible at a 200Hz dimming frequency. This is made possible by a unique feature that detects short PWM dimming input pulses and adjusts the converter feedback accordingly. Advanced features include detection and string-disconnect for open-LED strings, partial or fully shorted strings and unused strings. Overvoltage protection clamps the converter output voltage to the programmed OVP threshold in the event of an open-LED condition. Shorted LED string detection and overvoltage protection thresholds are programmable using RSDT and OVP inputs, respectively. An open-drain FLT signal asserts to indicate open-LED, shorted LED, and overtemperature conditions. Disable individual current-sink channels by connecting the corresponding OUT_ to LEDGND. In this case, FLT does not assert indicating an open-LED condition for the disabled channel. The device also features an overtemperature protection that shuts down the controller if the die temperature exceeds +165NC. Current-Mode DC-DC Controller The peak current-mode controller allows boost, coupledinductor buck-boost, or SEPIC-type converters to generate the required bias voltage for the LED strings. The switching frequency can be programmed over the 200kHz to 2MHz range using a resistor connected from RT to SGND. Programmable slope compensation is available to compensate for subharmonic oscillations that occur at above 50% duty cycles in continuous conduction mode. The external MOSFET is turned on at the beginning of every switching cycle. The inductor current ramps up linearly until it is turned off at the peak current level set by the feedback loop. The peak inductor current is sensed from the voltage across the current-sense resistor RCS connected from the source of the external MOSFET to PGND. The MAX16814 features leading-edge blanking to suppress the external MOSFET switching noise. A PWM comparator compares the current-sense voltage plus the slope compensation signal with the output of the transconductance error amplifier. The controller turns off the external MOSFET when the voltage at CS exceeds the error amplifier’s output voltage. This process repeats every switching cycle to achieve peak current-mode control. Error Amplifier The internal error amplifier compares an internal feedback (FB) with an internal reference (REF) and regulates its output to adjust the inductor current. An internal minimum string detector measures the minimum currentsink voltage with respect to SGND out of the 4 constantcurrent-sink channels. During normal operation, this minimum OUT_ voltage is regulated to 1V through feedback. The error amplifier takes 1V as the REF and the minimum OUT_ voltage as the FB input. The amplified error at the COMP output controls the inductor peak current to regulate the minimum OUT_ voltage at 1V. The resulting DC-DC converter output voltage is the highest LED string voltage plus 1V. 12 ������������������������������������������������������������������������������������� Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller For the MAX16814A_ _ and the MAX16814U_ _, if the PWM dimming on-pulse is less than or equal to five switching cycles, the feedback controls the voltage on OVP so that the converter output voltage is regulated at 95% of the OVP threshold. This mode ensures that narrow PWM dimming pulses are not affected by the response time of the converter. During this mode, the error amplifier remains connected to the COMP output continuously and the DC-DC converter continues switching. Undervoltage Lockout (UVLO) The MAX16814 features two undervoltage lockouts that monitor the input voltage at IN and the output of the internal LDO regulator at VCC. The device turns on after both VIN and VCC exceed their respective UVLO thresholds. The UVLO threshold at IN is 4.3V when VIN is rising and 4.15V when VIN is falling. The UVLO threshold at VCC is 4V when VCC is rising and 3.9V when VCC is falling. Enable EN is a logic input that completely shuts down the device when connected to logic-low, reducing the current consumption of the device to less than 40FA. The logic threshold at EN is 1.23V (typ). The voltage at EN must exceed 1.23V before any operation can commence. There is a 50mV hysteresis on EN. The EN input also allows programming the supply input UVLO threshold using an external voltage-divider to sense the input voltage as shown below. Use the following equation to calculate the value of R1 and R2 in Figure 3: V R1 = UVLO − 1 × R2 1.23V where VUVLO is the desired undervoltage lockout level and 1.23V is the EN input reference. Connect EN to IN if not used. Soft-Start The MAX16814 provides soft-start with internally set timing. At power-up, the MAX16814 enters soft-start once unused LED strings are detected and disconnected (see the Open-LED Management and Overvoltage Protection section). During soft-start, the DC-DC converter output ramps towards 95% of the OVP voltage and uses feedback from the OVP input. Soft-start terminates when the minimum current-sink voltage reaches 1V or when the converter output reaches 95% OVP. The typical soft-start period is 100ms. The 1V minimum OUT_ voltage is detected only when the LED strings are enabled by PWM dimming. Connect OVP to the boost converter output through a resistive divider network (see the Typical Operating Circuit). When there is an open-LED condition, the converter output hits the OVP threshold. After the OVP is triggered, openLED strings are disconnected and, at the beginning of the dimming PWM pulse, control is transferred to the adaptive voltage control. The converter output discharges to a level where the new minimum OUT_ voltage is 1V. Oscillator Frequency/External Synchronization The internal oscillator frequency is programmable between 200kHz and 2MHz using a resistor (RT) connected from the RT input to SGND. Use the equation below to calculate the value of RT for the desired switching frequency, fSW. RT = 7.35 × 10 9 Hz fSW (for the MAX16814A_ _ and the MAX16814U_ _). RT = 7.72 × 10 9 fSW (for the MAX16814BU_ _). Synchronize the oscillator with an external clock by AC-coupling the external clock to the RT input. The capacitor used for the AC-coupling should satisfy the following relation: 9862 C SYNC ≤ −0.144×10 −3 (µF) RT where RT is in Ω. VIN MAX16814 R1 EN R2 1.23V Figure 3. Setting the MAX16814 Undervoltage Lockout Threshold ______________________________________________________________________________________ 13 MAX16814 The converter stops switching when the LED strings are turned off during PWM dimming. The error amplifier is disconnected from the COMP output to retain the compensation capacitor charge. This allows the converter to settle to steady-state level almost immediately when the LED strings are turned on again. This unique feature provides fast dimming response, without having to use large output capacitors. MAX16814 Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller The pulse width for the synchronization pulse should satisfy the following relations: t PW VS < 0.5 t CLK t PW VS + VS > 3.4 0.8 − t CLK t CLK t PW < (t CI − 1.05 ×t CLK ) t CI where tPW is the synchronization source pulse width, tCLK is the synchronization clock time period, tCI is the programmed clock period, and VS is the synchronization pulse voltage level. 5V LDO Regulator (VCC) The internal LDO regulator converts the input voltage at IN to a 5V output voltage at VCC. The LDO regulator supplies up to 50mA current to provide power to internal control circuitry and the gate driver. Connect a resistor between VCC and DRV to power the gate-drive circuitry; the recommended value is 4.7I. Bypass DRV with a capacitor to PGND. The external resistor and bypass capacitor provide noise filtering. Bypass VCC to SGND with a minimum of 1FF ceramic capacitor as close to the device as possible. PWM MOSFET Driver The NDRV output is a push-pull output with the on-resistance of the pMOS typically 1.1I and the on-resistance of the nMOS typically 0.9I. NDRV swings from PGND to DRV to drive an external n-channel MOSFET. The driver typically sources 2.0A and sinks 2.0A allowing for fast turn-on and turn-off of high gate-charge MOSFETs. The power dissipation in the MAX16814 is mainly a function of the average current sourced to drive the external MOSFET (IDRV) if there are no additional loads on VCC. IDRV depends on the total gate charge (QG) and operating frequency of the converter. Connect DRV to VCC with a 4.7I resistor to power the gate driver with the internal 5V regulator. where IOUT_ is the desired output current for each of the four channels. If more than 150mA is required in an LED string, use two or more of the current source outputs (OUT_) connected together to drive the string as shown in Figure 4. LED Dimming Control The MAX16814 features LED brightness control using an external PWM signal applied at DIM. A logic-high signal on the DIM input enables all four LED current sources and a logic-low signal disables them. For the MAX16814A_ _ and the MAX16814U_ _, the duty cycle of the PWM signal applied to DIM also controls the DC-DC converter’s output voltage. If the turn-on duration of the PWM signal is less than 5 oscillator clock cycles (DIM pulse width decreasing) then the boost converter regulates its output based on feedback from the OVP input. During this mode, the converter output voltage is regulated to 95% of the OVP threshold voltage. If the turn-on duration of the PWM signal is greater than or equal to 6 oscillator clock cycles (DIM pulse width increasing), then the converter regulates its output so that the minimum voltage at OUT_ is 1V. Fault Protections Fault protections in the MAX16814 include cycle-bycycle current limiting using the PWM controller, DC-DC converter output overvoltage protection, open-LED detection, short LED detection and protection, and overtemperature shutdown. An open-drain LED fault flag output (FLT) goes low when an open-LED string is detected, a shorted LED string is detected, and during BOOST CONVERTER OUTPUT 40mA TO 300mA PER STRING LED Current Control The MAX16814 features four identical constant-current sources used to drive multiple HB LED strings. The current through each one of the four channels is adjustable between 20mA and 150mA using an external resistor (RSETI) connected between SETI and SGND. Select RSETI using the following formula: R SETI = 1500 IOUT_ OUT1 MAX16814 OUT2 OUT3 OUT4 Figure 4. Configuration for Higher LED String Current 14 ������������������������������������������������������������������������������������� Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller Open-LED Management and Overvoltage Protection On power-up, the MAX16814 detects and disconnects any unused current-sink channels before entering softstart. Disable the unused current-sink channels by connecting the corresponding OUT_ to LEDGND. This avoids asserting the FLT output for the unused channels. After soft-start, the MAX16814 detects open LED and disconnects any strings with an open LED from the internal minimum OUT_ voltage detector. This keeps the DC-DC converter output voltage within safe limits and maintains high efficiency. During normal operation, the DC-DC converter output regulation loop uses the minimum OUT_ voltage as the feedback input. If any LED string is open, the voltage at the opened OUT_ goes to VLEDGND. The DC-DC converter output voltage then increases to the overvoltage protection threshold set by the voltage-divider network connected between the converter output, OVP input, SGND. The overvoltage protection threshold at the DC-DC converter output (VOVP) is determined using the following formula: R1 (see the Typical Operating Circuit) VOVP = 1.23 × 1 + R2 where 1.23V (typ) is the OVP threshold. Select R1 and R2 such that the voltage at OUT_ does not exceed the absolute maximum rating. As soon as the DC-DC converter output reaches the overvoltage protection threshold, the PWM controller is switched off setting NDRV low. Any current-sink output with VOUT_ < 300mV (typ) is disconnected from the minimum voltage detector. Connect the OUT_ of all channels without LED connections to LEDGND before power-up to avoid OVP triggering at startup. When an open-LED overvoltage condition occurs, FLT is latched low. Short LED Detection The MAX16814 checks for shorted LEDs at each rising edge of DIM. An LED short is detected at OUT_ if the following condition is met: VOUT_ > VMINSTR + 3 x VRSDT where VOUT_ is the voltage at OUT_, VMINSTR is the minimum current-sink voltage, and VRSDT is the programmable LED short detection threshold set at the RSDT input. Adjust VRSDT using a voltage-divider resistive network connected at the VCC output, RSDT input, and SGND. Once a short is detected on any of the strings, the LED strings with the short are disconnected and the FLT output flag asserts until the device detects that the shorts are removed on any of the following rising edges of DIM. Connect RSDT directly to VCC to always disable LED short detection. Applications Information DC-DC Converter Three different converter topologies are possible with the DC-DC controller in the MAX16814, which has the ground-referenced outputs necessary to use the constant current-sink drivers. If the LED string forward voltage is always more than the input supply voltage range, use the boost converter topology. If the LED string forward voltage falls within the supply voltage range, use the boost-buck converter topology. Boost-buck topology is implemented using either a conventional SEPIC configuration or a coupled-inductor boost-buck configuration. The latter is basically a flyback converter with 1:1 turns ratio. 1:1 coupled inductors are available with tight coupling suitable for this application. Figure 6 shows the coupled-inductor boost-buck configuration. It is also possible to implement a single inductor boost-buck converter using the MAX15054 high-side FET driver. The boost converter topology provides the highest efficiency among the above mentioned topologies. The coupled-inductor boost-buck topology has the advantage of not using a coupling capacitor over the SEPIC configuration. Also, the feedback loop compensation for SEPIC becomes complex if the coupling capacitor is not large enough. A coupled-inductor boost-buck is not suitable for cases where the coupled-inductor windings are not tightly coupled. Considerable leakage inductance requires additional snubber components and degrades the efficiency. ______________________________________________________________________________________ 15 MAX16814 thermal shutdown. FLT is cleared when the fault condition is removed during thermal shutdown and shorted LEDs. FLT is latched low for an open-LED condition and can be reset by cycling power or toggling the EN pin. The thermal shutdown threshold is +165NC and has 15NC hysteresis. MAX16814 Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller Power-Circuit Design First select a converter topology based on the above factors. Determine the required input supply voltage range, the maximum voltage needed to drive the LED strings including the minimum 1V across the constant LED current sink (VLED), and the total output current needed to drive the LED strings (ILED) as follows: ILED = I STRING × N STRING Use the following equations to calculate the maximum average inductor current (ILAVG) and peak inductor current (ILP) in amperes: IL AVG = Allowing the peak-to-peak inductor ripple DIL to be +30% of the average inductor current: where ISTRING is the LED current per string in amperes and NSTRING is the number of strings used. Calculate the maximum duty cycle (DMAX) using the following equations: ∆IL = IL AVG × 0.3 × 2 and: For boost configuration: D MAX = ILED 1 − D MAX IL P = IL AVG + (VLED + VD1 − VIN_MIN ) (VLED + VD1 − VDS − 0.3V) For SEPIC and coupled-inductor boost-buck-configurations: (VLED + VD1) D MAX = (VIN_MIN − VDS − 0.3V + VLED + VD1) where VD1 is the forward drop of the rectifier diode in volts (approximately 0.6V), VIN_MIN is the minimum input supply voltage in volts, and VDS is the drain-to-source voltage of the external MOSFET in volts when it is on, and 0.3V is the peak current-sense voltage. Initially, use an approximate value of 0.2V for VDS to calculate DMAX. Calculate a more accurate value of DMAX after the power MOSFET is selected based on the maximum inductor current. Select the switching frequency (fSW) depending on the space, noise, and efficiency constraints. Inductor Selection Boost and Coupled-Inductor Boost-Buck Configurations In all the three converter configurations, the average inductor current varies with the line voltage and the maximum average current occurs at the lowest line voltage. For the boost converter, the average inductor current is equal to the input current. Select the maximum peak-to-peak ripple on the inductor current (DIL). The recommended peak-to-peak ripple is 60% of the average inductor current. ∆IL 2 Calculate the minimum inductance value, LMIN, in henries with the inductor current ripple set to the maximum value: L MIN = (VINMIN − VDS − 0.3V) × D MAX fSW × ∆IL where 0.3V is the peak current-sense voltage. Choose an inductor that has a minimum inductance greater than the calculated LMIN and current rating greater than ILP. The recommended saturation current limit of the selected inductor is 10% higher than the inductor peak current for boost configuration. For the coupled-inductor boost-buck, the saturation limit of the inductor with only one winding conducting should be 10% higher than ILP. SEPIC Configuration Power circuit design for the SEPIC configuration is very similar to a conventional boost-buck design with the output voltage referenced to the input supply voltage. For SEPIC, the output is referenced to ground and the inductor is split into two parts (see Figure 5 for the SEPIC configuration). One of the inductors (L2) takes LED current as the average current and the other (L1) takes input current as the average current. Use the following equations to calculate the average inductor currents (IL1AVG, IL2AVG) and peak inductor currents (IL1P, IL2P) in amperes: I × D MAX × 1.1 IL1AVG = LED 1 − D MAX 16 ������������������������������������������������������������������������������������� Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller IL2 AVG = ILED For simplifying further calculations, consider L1 and L2 as a single inductor with L1 and L2 connected in parallel. The combined inductance value and current is calculated as follows: Assuming the peak-to-peak inductor ripple DIL is Q30% of the average inductor current: ∆IL1 = IL1AVG × 0.3 × 2 and: L MIN = L1MIN × L2 MIN L1MIN + L2 MIN and: IL AVG = IL1AVG + IL2 AVG IL1P = IL1AVG + ∆IL1 2 ∆IL2 = IL2 AVG × 0.3 × 2 and: IL2 P = IL2 AVG + ∆IL2 2 Calculate the minimum inductance values L1MIN and L2MIN in henries with the inductor current ripples set to the maximum value as follows: L1MIN = (VINMIN − VDS − 0.3V) × D MAX fSW × ∆IL1 L2 MIN = (VINMIN − VDS − 0.3V) × D MAX fSW × ∆IL2 where 0.3V is the peak current-sense voltage. Choose inductors that have a minimum inductance greater than the calculated L1MIN and L2MIN and current rating greater than IL1P and IL2P, respectively. The recommended saturation current limit of the selected inductor is 10% higher than the inductor peak current: where ILAVG represents the total average current through both the inductors together for SEPIC configuration. Use these values in the calculations for SEPIC configuration in the following sections. Select coupling capacitor CS so that the peak-to-peak ripple on it is less than 2% of the minimum input supply voltage. This ensures that the second-order effects created by the series resonant circuit comprising L1, CS, and L2 does not affect the normal operation of the converter. Use the following equation to calculate the minimum value of CS: CS ≥ ILED × D MAX VIN_MIN × 0.02 × fSW where CS is the minimum value of the coupling capacitor in farads, ILED is the LED current in amperes, and the factor 0.02 accounts for 2% ripple. Slope Compensation The MAX16814 generates a current ramp for slope compensation. This ramp current is in sync with the switching frequency and starts from zero at the beginning of every clock cycle and rises linearly to reach 50FA at the end of the clock cycle. The slope-compensating resistor, RSCOMP, is connected between the CS input and the source of the external MOSFET. This adds a programmable ramp voltage to the CS input voltage to provide slope compensation. ______________________________________________________________________________________ 17 MAX16814 The factor 1.1 provides a 10% margin to account for the converter losses: MAX16814 Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller Use the following equation to calculate the value of slope compensation resistance, RSCOMP. For boost configuration: R SCOMP = (VLED − 2VIN_MIN ) × R CS × 3 L MIN × 50FA × fSW × 4 For SEPIC and coupled-inductor boost-buck: R SCOMP = ( VLED − VIN_MIN ) × R CS × 3 L MIN × 50FA × fSW × 4 where VLED and VIN_MIN are in volts, RSCOMP and RCS are in ohms, LMIN is in henries and fSW is in hertz. The value of the switch current-sense resistor, RCS, can be calculated as follows: PWM dimming, the amount of ceramic capacitors on the output are usually minimized. In this case, an additional electrolytic or tantalum capacitor provides most of the bulk capacitance. External MOSFET Selection The external MOSFET should have a voltage rating sufficient to withstand the maximum output voltage together with the rectifier diode drop and any possible overshoot due to ringing caused by parasitic inductances and capacitances. The recommended MOSFET VDS voltage rating is 30% higher than the sum of the maximum output voltage and the rectifier diode drop. The recommended continuous drain current rating of the MOSFET (ID), when the case temperature is at +70NC, is greater than that calculated below: ID RMS = IL AVG 2 × D MAX × 1.3 For boost: 0.396 × 0.9 = ILP ×RCS + (DMAX × (VLED − 2VIN_MIN)×RCS × 3) 4 ×L MN × fSW The MOSFET dissipates power due to both switching losses and conduction losses. Use the following equation to calculate the conduction losses in the MOSFET: For SEPIC and boost-buck: 0.396 × 0.9 = ILP ×RCS + (DMAX × (VLED − VIN_MIN ) ×RCS × 3) 4 ×L MN × fSW where 0.396 is the minimum value of the peak current-sense threshold. The current-sense threshold also includes the slope compensation component. The minimum current-sense threshold of 0.396 is multiplied by 0.9 to take tolerances into account. Output Capacitor Selection For all the three converter topologies, the output capacitor supplies the load current when the main switch is on. The function of the output capacitor is to reduce the converter output ripple to acceptable levels. The entire output-voltage ripple appears across constant currentsink outputs because the LED string voltages are stable due to the constant current. For the MAX16814, limit the peak-to-peak output voltage ripple to 200mV to get stable output current. The ESR, ESL, and the bulk capacitance of the output capacitor contribute to the output ripple. In most of the applications, using low-ESR ceramic capacitors can dramatically reduce the output ESR and ESL effects. To reduce the ESL and ESR effects, connect multiple ceramic capacitors in parallel to achieve the required bulk capacitance. To minimize audible noise during PCOND = IL AVG 2 × D MAX × RDS (ON) where RDS(ON) is the on-state drain-to-source resistance of the MOSFET. Use the following equation to calculate the switching losses in the MOSFET: PSW = IL AVG × VLED 2 × C GD × fSW 1 1 × + 2 I I GON GOFF where IGON and IGOFF are the gate currents of the MOSFET in amperes, with VGS at the threshold voltage in volts, when it is turned on and turned off, respectively. CGD is the gate-to-drain MOSFET capacitance in farads. Rectifier Diode Selection Using a Schottky rectifier diode produces less forward drop and puts the least burden on the MOSFET during reverse recovery. A diode with considerable reverserecovery time increases the MOSFET switching loss. Select a Schottky diode with a voltage rating 20% higher than the maximum boost-converter output voltage and current rating greater than that calculated in the following equation: ID = 1.2 × IL AVG 1 − D MAX 18 ������������������������������������������������������������������������������������� Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller The worst-case condition for the feedback loop is when the LED driver is in normal mode regulating the minimum OUT_ voltage to 1V. The switching converter small-signal transfer function has a right-half plane (RHP) zero for boost configuration if the inductor current is in continuous conduction mode. The RHP zero adds a 20dB/decade gain together with a 90N-phase lag, which is difficult to compensate. The worst-case RHP zero frequency (fZRHP) is calculated as follows: For boost configuration: fZRHP = VLED (1 − D MAX ) 2 2π × L × ILED For SEPIC and coupled-inductor boost-buck configurations: fZRHP = VLED (1 − D MAX ) 2 2π × L × ILED × D MAX where fZRHP is in hertz, VLED is in volts, L is the inductance value of L1 in henries, and ILED is in amperes. A simple way to avoid this zero is to roll off the loop gain to 0dB at a frequency less than one fifth of the RHP zero frequency with a -20dB/decade slope. The switching converter small-signal transfer function also has an output pole. The effective output impedance together with the output filter capacitance determines the output pole frequency fP1 that is calculated as follows: For boost configuration: fP1 = ILED 2 × π × VLED × C OUT For SEPIC and coupled-inductor boost-buck configurations: fP1 = ILED × D MAX 2 × π × VLED × C OUT where fP1 is in hertz, VLED is in volts, ILED is in amperes, and COUT is in farads. Compensation components, RCOMP and CCOMP, perform two functions. CCOMP introduces a low-frequency pole that presents a -20dB/decade slope to the loop gain. RCOMP flattens the gain of the error amplifier for frequencies above the zero formed by RCOMP and CCOMP. For compensation, this zero is placed at the output pole frequency fP1 so that it provides a -20dB/ decade slope for frequencies above fP1 to the combined modulator and compensator response. The value of RCOMP needed to fix the total loop gain at fP1 so that the total loop gain crosses 0dB with -20dB/ decade slope at 1/5 the RHP zero frequency is calculated as follows: For boost configuration: R COMP = fZRHP × R CS × ILED 5 × fP1 × GM COMP × VLED × (1 − D MAX ) For SEPIC and coupled-inductor boost-buck configurations: R COMP = fZRHP × R CS × ILED × D MAX 5 × fP1 × GM COMP × VLED × (1 − D MAX ) where RCOMP is the compensation resistor in ohms, fZRHP and fP2 are in hertz, RCS is the switch currentsense resistor in ohms, and GMCOMP is the transconductance of the error amplifier (600FS). The value of CCOMP is calculated as follows: C COMP = 1 2π × R COMP × fP1 If the output capacitors do not have low ESR, the ESR zero frequency may fall within the 0dB crossover frequency. An additional pole may be required to cancel out this pole placed at the same frequency. This is usually implemented by connecting a capacitor in parallel with CCOMP and RCOMP. Figure 5 shows the SEPIC configuration and Figure 6 shows the coupled-inductor boost-buck configuration. ______________________________________________________________________________________ 19 MAX16814 Feedback Compensation During normal operation, the feedback control loop regulates the minimum OUT_ voltage to 1V when LED string currents are enabled during PWM dimming. When LED currents are off during PWM dimming, the control loop turns off the converter and stores the steady-state condition in the form of capacitor voltages, mainly the output filter capacitor voltage and compensation capacitor voltage. For the MAX16814A_ _ and the MAX16814U_ _, when the PWM dimming pulses are less than five switching clock cycles, the feedback loop regulates the converter output voltage to 95% of the OVP threshold. MAX16814 Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller Analog Dimming Using External Control Voltage SETI through the resistor RSETI2. The resulting change in the LED current with the control voltage is linear and inversely proportional. The LED current control range remains between 20mA to 150mA. Connect a resistor RSETI2 to the SETI input as shown in Figure 7 for controlling the LED string current using an external control voltage. The MAX16814 applies a fixed 1.23V bandgap reference voltage at SETI and measures the current through SETI. This measured current multiplied by a factor of 1220 is the current through each one of the four constant current-sink channels. Adjust the current through SETI to get analog dimming functionality by connecting the external control voltage to Use the following equation to calculate the LED current set by the control voltage applied: I OUT = 1500 (1.23 − VC ) + × 1220 R SETI R SETI2 VIN 4.75V TO 40V CS L1 C1 N NDRV UP TO 40V CS R2 OVP EN OUT1 VCC OUT2 C3 MAX16814 R5 C2 R1 L2 RCS RSCOMP IN D1 OUT3 OUT4 DRV RSETI C4 SETI DIM VCC FLT R3 COMP RSDT RCOMP RT SGND CCOMP PGND LEDGND R4 RT Figure 5. SEPIC Configuration 20 ������������������������������������������������������������������������������������� Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller LED driver circuits based on the MAX16814 device use a high-frequency switching converter to generate the voltage for LED strings. Take proper care while laying out the circuit to ensure proper operation. The switchingconverter part of the circuit has nodes with very fast voltage changes that could lead to undesirable effects on the sensitive parts of the circuit. Follow the guidelines below to reduce noise as much as possible: 1) Connect the bypass capacitor on VCC and DRV as close to the device as possible and connect the capacitor ground to the analog ground plane using vias close to the capacitor terminal. Connect SGND of the device to the analog ground plane using a via close to SGND. Lay the analog ground plane on the inner layer, preferably next to the top layer. Use the analog ground plane to cover the entire area under critical signal components for the power converter. 2) Have a power ground plane for the switching-converter power circuit under the power components (input filter capacitor, output filter capacitor, inductor, MOSFET, rectifier diode, and current-sense resistor). Connect PGND to the power ground plane as close to PGND as possible. Connect all other ground connections to the power ground plane using vias close to the terminals. 3) There are two loops in the power circuit that carry high-frequency switching currents. One loop is when the MOSFET is on (from the input filter capacitor positive terminal, through the inductor, the internal MOSFET, and the current-sense resistor, to the input capacitor negative terminal). The other loop is when the MOSFET is off (from the input capacitor positive terminal, through the inductor, the rectifier diode, output filter capacitor, to the input capacitor negative terminal). Analyze these two loops and make the loop areas as small as possible. Wherever possible, have a return path on the power ground plane for the switching currents on the top layer copper traces, or through power components. This reduces the loop area considerably and provides a low-inductance path for the switching currents. Reducing the loop area also reduces radiation during switching. 4) Connect the power ground plane for the constantcurrent LED driver part of the circuit to LEDGND as close to the device as possible. Connect SGND to PGND at the same point. ______________________________________________________________________________________ 21 MAX16814 PCB Layout Considerations MAX16814 Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller VIN 4.75V TO 40V T1 (1:1) D1 C1 UP TO 40V IN NDRV R2 RCS RSCOMP CS OVP EN OUT1 VCC OUT2 C3 MAX16814 R5 C2 R1 N OUT3 OUT4 DRV RSETI C4 SETI DIM VCC FLT R3 COMP RSDT RCOMP RT SGND PGND CCOMP LEDGND R4 RT Figure 6. Coupled-Inductor Boost-Buck Configuration MAX16814 1.23V SETI RSETI2 RSETI VC Figure 7. Analog Dimming with External Control Voltage 22 ������������������������������������������������������������������������������������� Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller MAX16814 IN 4 EN 5 17 OUT3 14 OUT1 FLT 8 13 DIM OVP 9 EP* SETI 10 OUT2 OUT1 CS 16 10 DIM PGND 17 9 SGND 8 RSDT 7 SETI 6 OVP MAX16814 DRV 19 EP* VCC 20 12 SGND 1 2 11 RSDT EN RT 7 11 IN 15 OUT2 12 NDRV 18 16 LEDGND COMP 6 13 3 4 5 FLT 18 OUT4 14 RT 19 CS VCC 3 15 COMP DRV 2 LEDGND 20 PGND OUT3 + NDRV 1 OUT4 TOP VIEW TOP VIEW THIN QFN TSSOP *EXPOSED PAD. Typical Operating Circuit VIN 4.75V TO 40V D1 L C1 UP TO 40V IN NDRV R2 RCS RSCOMP CS C2 R1 N OVP EN OUT1 VCC OUT2 C3 MAX16814 R5 OUT3 OUT4 DRV RSETI C4 SETI DIM VCC FLT R3 COMP RSDT RCOMP RT SGND CCOMP PGND LEDGND R4 RT ______________________________________________________________________________________ 23 MAX16814 Pin Configurations MAX16814 Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller Chip Information PROCESS: BiCMOS DMOS Package Information For the latest package outline information and land patterns, go to www.maxim-ic.com/packages. PACKAGE TYPE PACKAGE CODE DOCUMENT NO. 20 TSSOP-EP U20E+1 21-0108 20 TQFN-EP T2044+3 21-0139 24 ������������������������������������������������������������������������������������� Integrated, 4-Channel, High-Brightness LED Driver with High-Voltage DC-DC Controller REVISION NUMBER REVISION DATE 0 7/09 Initial release 1 9/09 Correction to slope compensation description and block diagram 2 11/09 Correction to synchronization description frequency and minor edits DESCRIPTION PAGES CHANGED — 10, 18 1–4, 8, 12–20, 22, 25 Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 2009 Maxim Integrated Products 25 Maxim is a registered trademark of Maxim Integrated Products, Inc. MAX16814 Revision History