MCP651/2/5 50 MHz, 6 mA Op Amps with mCal Features Description • • • • • • • • • The Microchip Technology, Inc. MCP651/2/5 family of operational amplifiers features low offset. At power up, these op amps are self-calibrated using mCal. Some package options also provide a calibration/chip select pin (CAL/CS) that supports a low power mode of operation, with offset calibration at the time normal operation is re-started. These amplifiers are optimized for high speed, low noise and distortion, single-supply operation with rail-to-rail output and an input that includes the negative rail. Gain Bandwidth Product: 50 MHz (typical) Short Circuit Current: 100 mA (typical) Noise: 7.5 nV/√Hz (typical, at 1 MHz) Calibrated Input Offset: ±200 µV (maximum) Rail-to-Rail Output Slew Rate: 30 V/µs (typical) Supply Current: 6.0 mA (typical) Power Supply: 2.5V to 5.5V Extended Temperature Range: -40°C to +125°C This family is offered in single with CAL/CS pin (MCP651), dual (MCP652) and dual with CAL/CS pins (MCP655). All devices are fully specified from -40°C to +125°C. Typical Applications • • • • Driving A/D Converters Power Amplifier Control Loops Barcode Scanners Optical Detector Amplifier Typical Application Circuit VDD/2 Design Aids • • • • • • R1 R2 R3 SPICE Macro Models FilterLab® Software Mindi™ Circuit Designer & Simulator Microchip Advanced Part Selector (MAPS) Analog Demonstration and Evaluation Boards Application Notes VOUT RL MCP65X VIN Power Driver with High Gain Package Types MCP651 SOIC NC 1 MCP652 3×3 DFN * MCP655 3×3 DFN * 8 CAL/CS VOUTA 1 8 VDD VOUTA 1 VIN– 2 7 VDD VINA– 2 7 VOUTB VINA– 2 VIN+ 3 6 VOUT VSS 4 5 VCAL VINA+ 3 6 VINB– VINA+ 3 5 VINB+ VSS 4 EP 9 VSS 4 10 VDD 9 VOUTB EP 11 8 VINB– 7 VINB+ CALA/CSA 5 MCP655 MSOP MCP652 SOIC VOUTA 1 8 VDD VINA– 2 7 VOUTB VINA+ 3 6 VINB– 5 VINB+ VSS 4 6 CALB/CSB VOUTA 1 VINA– 2 VINA+ 3 VSS 4 CALA/CSA 5 10 VDD 9 VOUTB 8 VINB– 7 VINB+ 6 CALB/CSB * Includes Exposed Thermal Pad (EP); see Table 3-1. © 2009 Microchip Technology Inc. DS22146A-page 1 MCP651/2/5 NOTES: DS22146A-page 2 © 2009 Microchip Technology Inc. MCP651/2/5 1.0 ELECTRICAL CHARACTERISTICS 1.1 Absolute Maximum Ratings † † Notice: Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation of the device at those or any other conditions above those indicated in the operational listings of this specification is not implied. Exposure to maximum rating conditions for extended periods may affect device reliability. VDD – VSS .......................................................................6.5V Current at Input Pins ....................................................±2 mA Analog Inputs (VIN+ and VIN–) †† . VSS – 1.0V to VDD + 1.0V All other Inputs and Outputs .......... VSS – 0.3V to VDD + 0.3V Difference Input voltage ...................................... |VDD – VSS| Output Short Circuit Current ................................ Continuous Current at Output and Supply Pins ..........................±150 mA Storage Temperature ...................................-65°C to +150°C Max. Junction Temperature ........................................ +150°C ESD protection on all pins (HBM, MM) ................≥ 1 kV, 200V 1.2 †† See Section 4.2.2 “Input Voltage and Current Limits”. Specifications TABLE 1-1: DC ELECTRICAL SPECIFICATIONS Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to +5.5V, VSS = GND, VCM = VDD/3, VOUT ≈ VDD/2, VL = VDD/2, RL = 1 kΩ to VL and CAL/CS = VSS (refer to Figure 1-2). Parameters Sym Min Typ Max Units Conditions Input Offset Input Offset Voltage Input Offset Voltage Trim Step Input Offset Voltage Drift Power Supply Rejection Ratio VOS -200 — +200 µV VOSTRM — 37 200 µV After calibration (Note 1) ΔVOS/ΔTA — ±2.5 — PSRR 61 76 — IB — 6 — pA IB — 130 — pA TA= +85°C TA= +125°C µV/°C TA= -40°C to +125°C dB Input Current and Impedance Input Bias Current Across Temperature IB — 1700 5,000 pA Input Offset Current IOS — ±1 — pA Common Mode Input Impedance ZCM — 1013||9 — Ω||pF Differential Input Impedance ZDIFF — 1013||2 — Ω||pF Common-Mode Input Voltage Range VCMR VSS − 0.3 — VDD − 1.3 V (Note 2) Common-Mode Rejection Ratio CMRR 65 81 — dB VDD = 2.5V, VCM = -0.3 to 1.2V CMRR 68 84 — dB VDD = 5.5V, VCM = -0.3 to 4.2V AOL 88 114 — dB VDD = 2.5V, VOUT = 0.3V to 2.2V AOL 94 123 — dB VDD = 5.5V, VOUT = 0.3V to 5.2V VOL, VOH VSS + 25 — VDD − 25 mV VDD = 2.5V, G = +2, 0.5V Input Overdrive VOL, VOH VSS + 50 — VDD − 50 mV VDD = 5.5V, G = +2, 0.5V Input Overdrive ISC ±50 ±95 ±145 mA VDD = 2.5V (Note 3) ISC ±50 ±100 ±150 mA VDD = 5.5V (Note 3) Across Temperature Common Mode Open-Loop Gain DC Open-Loop Gain (large signal) Output Maximum Output Voltage Swing Output Short Circuit Current Note 1: 2: 3: Describes the offset (under the specified conditions) right after power up, or just after the CAL/CS pin is toggled. Thus, 1/f noise effects (an apparent wander in VOS; see Figure 2-35) are not included. See Figure 2-6 and Figure 2-7 for temperature effects. The ISC specifications are for design guidance only; they are not tested. © 2009 Microchip Technology Inc. DS22146A-page 3 MCP651/2/5 TABLE 1-1: DC ELECTRICAL SPECIFICATIONS (CONTINUED) Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to +5.5V, VSS = GND, VCM = VDD/3, VOUT ≈ VDD/2, VL = VDD/2, RL = 1 kΩ to VL and CAL/CS = VSS (refer to Figure 1-2). Parameters Sym Min Typ Max Units mV Conditions Calibration Input VCALRNG VSS + 0.1 — VDD – 1.4 Internal Calibration Voltage Calibration Input Voltage Range VCAL 0.31VDD 0.33VDD 0.35VDD Input Impedance ZCAL — 100 || 5 — VDD 2.5 — 5.5 V IQ 3 6 9 mA POR Input Threshold, Low VPRL 1.15 1.40 — V POR Input Threshold, High VPRH — 1.40 1.65 V VCAL pin externally driven VCAL pin open kΩ||pF Power Supply Supply Voltage Quiescent Current per Amplifier Note 1: 2: 3: IO = 0 Describes the offset (under the specified conditions) right after power up, or just after the CAL/CS pin is toggled. Thus, 1/f noise effects (an apparent wander in VOS; see Figure 2-35) are not included. See Figure 2-6 and Figure 2-7 for temperature effects. The ISC specifications are for design guidance only; they are not tested. TABLE 1-2: AC ELECTRICAL SPECIFICATIONS Electrical Characteristics: Unless otherwise indicated, TA = 25°C, VDD = +2.5V to +5.5V, VSS = GND, VCM = VDD/2, VOUT ≈ VDD/2, VL = VDD/2, RL = 1 kΩ to VL, CL = 20 pF and CAL/CS = VSS (refer to Figure 1-2). Parameters Sym Min Typ Max Units Conditions GBWP — 50 — MHz PM — 65 — ° ROUT — 20 — Ω THD+N — 0.0012 — % G = +1, VOUT = 4VP-P, f = 1 kHz, VDD = 5.5V, BW = 80 kHz G = +1, VOUT = 100 mVP-P AC Response Gain Bandwidth Product Phase Margin Open-Loop Output Impedance G = +1 AC Distortion Total Harmonic Distortion plus Noise Step Response tr — 6 — ns SR — 30 — V/µs G = +1 Input Noise Voltage Eni — 17 — µVP-P f = 0.1 Hz to 10 Hz Input Noise Voltage Density eni — 7.5 — nV/√Hz f = 1 MHz Input Noise Current Density ini 4 — fA/√Hz Rise Time, 10% to 90% Slew Rate Noise DS22146A-page 4 f = 1 kHz © 2009 Microchip Technology Inc. MCP651/2/5 TABLE 1-3: DIGITAL ELECTRICAL SPECIFICATIONS Electrical Characteristics: Unless otherwise indicated, TA = 25°C, VDD = +2.5V to +5.5V, VSS = GND, VCM = VDD/2, VOUT ≈ VDD/2, VL = VDD/2, RL = 1 kΩ to VL, CL = 20 pF and CAL/CS = VSS (refer to Figure 1-1 and Figure 1-2). Parameters Sym Min Typ Max Units Conditions CAL/CS Logic Threshold, Low VIL VSS — 0.2VDD V CAL/CS Input Current, Low ICSL — 0 — nA CAL/CS Logic Threshold, High VIH 0.8VDD VDD V CAL/CS Input Current, High ICSH — 0.7 — µA CAL/CS = VDD ISS -3.5 -1.8 — µA Single, CAL/CS = VDD = 2.5V ISS -8 -4 — µA Single, CAL/CS = VDD = 5.5V ISS -5 -2.5 — µA Dual, CAL/CS = VDD = 2.5V Dual, CAL/CS = VDD = 5.5V CAL/CS Low Specifications CAL/CS = 0V CAL/CS High Specifications GND Current ISS -10 -5 — µA RPD — 5 — MΩ IO(LEAK) — 50 — nA CAL/CS = VDD VDD Low to Amplifier Off Time (output goes High-Z) tPOFF — 200 — ns G = +1 V/V, VL = VSS, VDD = 2.5V to 0V step to VOUT = 0.1 (2.5V) VDD High to Amplifier On Time (including calibration) tPON 100 200 300 ms G = +1 V/V, VL = VSS, VDD = 0V to 2.5V step to VOUT = 0.9 (2.5V) CAL/CS Input Hysteresis VHYST — 0.25 — V CAL/CS Setup Time (between CAL/CS edges) tCSU 1 — — µs G = +1 V/V, VL = VSS (Notes 2, 3) CAL/CS = 0.8VDD to VOUT = 0.1 (VDD/2) CAL/CS High to Amplifier Off Time (output goes High-Z) tCOFF — 200 — ns G = +1 V/V, VL = VSS, CAL/CS = 0.8VDD to VOUT = 0.1 (VDD/2) CAL/CS Low to Amplifier On Time (including calibration) tCON — 3 4 ms G = +1 V/V, VL = VSS, CAL/CS = 0.2VDD to VOUT = 0.9 (VDD/2) CAL/CS Internal Pull Down Resistor Amplifier Output Leakage POR Dynamic Specifications CAL/CS Dynamic Specifications Note 1: 2: 3: The MCP652 has its CAL/CS input internally pulled down to VSS (0V). This time ensures that the internal logic recognizes the edge. However, for the rising edge case, if CAL/CS is raised before the calibration is complete, the calibration will be aborted and the part will return to low power mode. For the MCP655 dual, there is an additional constraint. CALA/CSA and CALB/CSB can be toggled simultaneously (within a time much smaller than tCSU) to make both op amps perform the same function simultaneously. If they are toggled independently, then CALA/CSA (CALB/CSB) cannot be allowed to toggle while op amp B (op amp A) is in calibration mode; allow more than the maximum tCON time (4 ms) before the other side is toggled. TABLE 1-4: TEMPERATURE SPECIFICATIONS Electrical Characteristics: Unless otherwise indicated, all limits are specified for: VDD = +2.5V to +5.5V, VSS = GND. Parameters Sym Min Typ Max Units Specified Temperature Range TA -40 — +125 °C Operating Temperature Range TA -40 — +125 °C Storage Temperature Range TA -65 — +150 °C Thermal Resistance, 8L-3×3 DFN θJA — 63 — °C/W Thermal Resistance, 8L-SOIC θJA — 163 — °C/W Conditions Temperature Ranges (Note 1) Thermal Package Resistances Note 1: 2: (Note 2) Operation must not cause TJ to exceed Maximum Junction Temperature specification (150°C). Measured on a standard JC51-7, four layer printed circuit board with ground plane and vias. © 2009 Microchip Technology Inc. DS22146A-page 5 MCP651/2/5 TABLE 1-4: TEMPERATURE SPECIFICATIONS Electrical Characteristics: Unless otherwise indicated, all limits are specified for: VDD = +2.5V to +5.5V, VSS = GND. Parameters Sym Min Thermal Resistance, 10L-3×3 DFN θJA — 71 — °C/W Thermal Resistance, 10L-MSOP θJA — 202 — °C/W Note 1: 2: 1.3 Typ Units Conditions (Note 2) Operation must not cause TJ to exceed Maximum Junction Temperature specification (150°C). Measured on a standard JC51-7, four layer printed circuit board with ground plane and vias. Timing Diagram CAL/CS VIL VIH VDD VPRH tPON High-Z On ISS -3 µA (typical) -6 mA (typical) VPRL tCON tPOFF High-Z On -6 mA (typical) -3 µA (typical) ICS 0 nA (typical) FIGURE 1-1: tCSU tCOFF VOUT High-Z 1.4 Max -3 µA (typical) 0 nA (typical) 0.7 µA (typical) Timing Diagram. Test Circuits The circuit used for most DC and AC tests is shown in Figure 1-2. This circuit can independently set VCM and VOUT; see Equation 1-1. Note that VCM is not the circuit’s common mode voltage ((VP + VM)/2), and that VOST includes VOS plus the effects (on the input offset error, VOST) of temperature, CMRR, PSRR and AOL. CF 6.8 pF RG 10 kΩ VP EQUATION 1-1: CB1 100 nF MCP65X V CM = ( V P + V DD ⁄ 2 ) ⁄ 2 V OST = V IN– – V IN+ V OUT = ( V DD ⁄ 2 ) + ( V P – V M ) + V OST ( 1 + G DM ) Where: GDM = Differential Mode Gain (V/V) VCM = Op Amp’s Common Mode Input Voltage (V) DS22146A-page 6 VDD VIN+ G DM = R F ⁄ R G VOST = Op Amp’s Total Input Offset Voltage RF 10 kΩ (mV) VDD/2 CB2 2.2 µF VIN– VM RG 10 kΩ RL 1 kΩ RF 10 kΩ CF 6.8 pF VOUT CL 20 pF VL FIGURE 1-2: AC and DC Test Circuit for Most Specifications. © 2009 Microchip Technology Inc. MCP651/2/5 2.0 TYPICAL PERFORMANCE CURVES Note: The graphs and tables provided following this note are a statistical summary based on a limited number of samples and are provided for informational purposes only. The performance characteristics listed herein are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified operating range (e.g., outside specified power supply range) and therefore outside the warranted range. Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kΩ to VL, CL = 20 pF, and CAL/CS = VSS. DC Signal Inputs Percentage of Occurrences 35% 30% 25% 700 80 Samples TA = +25°C VDD = 2.5V and 5.5V Calibrated at +25°C Input Offset Voltage (µV) 2.1 20% 15% 10% 5% 0% FIGURE 2-1: 400 300 Input Offset Voltage. 100 0 -8 -6 -4 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 Power Supply Voltage (V) FIGURE 2-4: Input Offset Voltage vs. Power Supply Voltage. 80 Samples VDD = 2.5V and 5.5V TA = -40°C to +125°C Calibrated at +25°C -10 -2 0 2 4 6 8 50 40 30 20 10 0 -10 -20 -30 -40 -50 10 Input Offset Voltage Drift. VDD = 5.5V 0.0 80 Samples TA = +25°C VDD = 2.5V and 5.5V No Change (includes noise) Calibration Changed VDD = 2.5V FIGURE 2-5: Output Voltage. Calibration Changed Low Input Common Mode Headroom (V) Percentage of Occurrences FIGURE 2-2: Representative Part 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Output Voltage (V) Input Offset Voltage Drift (µV/°C) 55% 50% 45% 40% 35% 30% 25% 20% 15% 10% 5% 0% +125°C +85°C +25°C -40°C 200 80 100 Input Offset Voltage (µV) Percentage of Occurrences 500 -100 -100 -80 -60 -40 -20 0 20 40 60 Input Offset Voltage (µV) 20% 18% 16% 14% 12% 10% 8% 6% 4% 2% 0% Representative Part Calibrated at VDD = 6.5V 600 Input Offset Voltage vs. 1 Lot Low (VCMR_L – VSS) -0.1 -0.2 VDD = 2.5V -0.3 VDD = 5.5V -0.4 -0.5 -100 -80 -60 -40 -20 0 20 40 60 80 100 Input Offset Voltage Repeatability (µV) FIGURE 2-3: Input Offset Voltage Repeatability (repeated calibration). © 2009 Microchip Technology Inc. -50 -25 0 25 50 75 100 Ambient Temperature (°C) 125 FIGURE 2-6: Low Input Common Mode Voltage Headroom vs. Ambient Temperature. DS22146A-page 7 MCP651/2/5 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kΩ to VL, CL = 20 pF, and CAL/CS = VSS. 1 Lot High (VDD – VCMR_H) CMRR, PSRR (dB) High Input Common Mode Headroom (V) 1.4 1.3 VDD = 2.5V 1.2 1.1 VDD = 5.5V 1.0 -50 -25 0 25 50 75 100 Ambient Temperature (°C) 125 CMRR, VDD = 5.5V CMRR, VDD = 2.5V -50 -25 120 115 VDD = 2.5V 110 105 100 95 -50 10,000 Input Bias, Offset Currents (pA) Input Common Mode Voltage (V) FIGURE 2-9: Input Offset Voltage vs. Common Mode Voltage with VDD = 5.5V. 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 -40°C +25°C +85°C +125°C 0.0 125 -25 0 25 50 75 Ambient Temperature (°C) 100 125 FIGURE 2-11: DC Open-Loop Gain vs. Ambient Temperature. VDD = 5.5V Representative Part -0.5 Input Offset Voltage (µV) FIGURE 2-8: Input Offset Voltage vs. Common Mode Voltage with VDD = 2.5V. DS22146A-page 8 100 VDD = 5.5V 125 Input Common Mode Voltage (V) 1000 800 600 400 200 0 -200 -400 -600 -800 -1000 0 25 50 75 Ambient Temperature (°C) FIGURE 2-10: CMRR and PSRR vs. Ambient Temperature. DC Open-Loop Gain (dB) 2.0 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0.0 -0.2 -40°C +25°C +85°C +125°C -0.4 PSRR 130 VDD = 2.5V Representative Part -0.6 Input Offset Voltage (µV) FIGURE 2-7: High Input Common Mode Voltage Headroom vs. Ambient Temperature. 1000 800 600 400 200 0 -200 -400 -600 -800 -1000 110 105 100 95 90 85 80 75 70 65 60 VDD = 5.5V VCM = VCMR_H 1,000 IB 100 10 -IOS 1 25 45 65 85 105 Ambient Temperature (°C) 125 FIGURE 2-12: Input Bias and Offset Currents vs. Ambient Temperature with VDD = +5.5V. © 2009 Microchip Technology Inc. MCP651/2/5 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, 160 140 120 100 80 60 40 20 0 -20 -40 -60 1.E-03 1m TA = +85°C VDD = 5.5V 100µ 1.E-04 10µ 1.E-05 IB 1µ 1.E-06 100n 1.E-07 IOS 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Common Mode Input Voltage (V) FIGURE 2-13: Input Bias and Offset Currents vs. Common Mode Input Voltage with TA = +85°C. 2000 Input Bias, Offset Currents (pA) Input Current Magnitude (A) Input Bias, Offset Currents (pA) VL = VDD/2, RL = 1 kΩ to VL, CL = 20 pF, and CAL/CS = VSS. 1500 10n 1.E-08 1n 1.E-09 100p 1.E-10 10p 1.E-11 +125°C +85°C +25°C -40°C 1p 1.E-12 -1.0 -0.9 -0.8 -0.7 -0.6 -0.5 -0.4 -0.3 -0.2 -0.1 0.0 Input Voltage (V) FIGURE 2-15: Input Bias Current vs. Input Voltage (below VSS). TA = +125°C VDD = 5.5V IB 1000 500 0 IOS -500 -1000 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Common Mode Input Voltage (V) FIGURE 2-14: Input Bias and Offset Currents vs. Common Mode Input Voltage with TA = +125°C. © 2009 Microchip Technology Inc. DS22146A-page 9 MCP651/2/5 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kΩ to VL, CL = 20 pF, and CAL/CS = VSS. Other DC Voltages and Currents 8 VDD = 5.5V 12 7 VOL – VSS -IOUT Supply Current (mA/amplifier) 10 8 6 4 VDD – VOH IOUT VDD = 2.5V 2 2 RL = 1 kΩ 6.5 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 FIGURE 2-19: Supply Voltage. Supply Current vs. Power 9 VOL – VSS 8 VDD = 5.5V 8 6 VDD – VOH VDD = 2.5V 1.5 Power Supply Voltage (V) 12 10 1.0 100 Supply Current (mA/amplifier) 2 7 VDD = 5.5V 6 5 VDD = 2.5V 4 3 2 1 5.5 5.0 4.5 4.0 3.5 125 3.0 100 2.5 0 25 50 75 Ambient Temperature (°C) 2.0 -25 0.0 -50 1.5 0 0 1.0 Output Headroom (mV) +125°C +85°C +25°C -40°C 3 0.0 10 Output Current Magnitude (mA) FIGURE 2-16: Ratio of Output Voltage Headroom to Output Current. Common Mode Input Voltage (V) FIGURE 2-17: Output Voltage Headroom vs. Ambient Temperature. FIGURE 2-20: Supply Current vs. Common Mode Input Voltage. POR Trip Voltages (V) 1.8 100 80 60 40 20 0 -20 -40 -60 -80 -100 +125°C +85°C +25°C -40°C 1.6 1.4 VPRH 1.2 1.0 VPRL 0.8 0.6 0.4 0.2 6.5 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 0.0 Output Short Circuit Current (mA) 4 0 1 4 5 1 0 14 6 0.5 Ratio of Output Headroom to Output Current (mV/mA) 14 0.5 2.2 Power Supply Voltage (V) FIGURE 2-18: Output Short Circuit Current vs. Power Supply Voltage. DS22146A-page 10 -50 -25 0 25 50 75 Ambient Temperature (°C) 100 125 FIGURE 2-21: Power On Reset Voltages vs. Ambient Temperature. © 2009 Microchip Technology Inc. MCP651/2/5 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, 25% 144 Samples VDD = 2.5V and 5.5V 20% 15% 10% 5% Normalized Internal Calibration Voltage; VCAL/VDD FIGURE 2-22: Normalized Internal Calibration Voltage. © 2009 Microchip Technology Inc. 33.52% 33.48% 33.44% 33.40% 33.36% 33.32% 33.28% 33.24% 0% Internal V CAL Resistance (kΩ) 30% 33.20% Percentage of Occurrences VL = VDD/2, RL = 1 kΩ to VL, CL = 20 pF, and CAL/CS = VSS. 140 120 100 80 60 40 20 0 -50 -25 FIGURE 2-23: Temperature. 0 25 50 75 Ambient Temperature (°C) 100 125 VCAL Input Resistance vs. DS22146A-page 11 MCP651/2/5 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kΩ to VL, CL = 20 pF, and CAL/CS = VSS. Frequency Response 90 0 100 -30 80 -60 ∠AOL -90 -120 | AOL | 20 -150 0 -180 -20 -210 50 40 40 50 30 40 20 30 -50 -25 0 25 50 75 100 Ambient Temperature (°C) 10 125 FIGURE 2-26: Gain Bandwidth Product and Phase Margin vs. Ambient Temperature. DS22146A-page 12 Open-Loop Output Impedance (Ω) GBWP VDD = 5.5V VDD = 2.5V 50 Phase Margin (°) Gain Bandwidth Product (MHz) 60 70 GBWP 30 60 VDD = 2.5V 20 50 PM 10 Phase Margin (°) 40 30 FIGURE 2-28: Gain Bandwidth Product and Phase Margin vs. Output Voltage. 60 70 80 VDD = 5.5V 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Output Voltage (V) Open-Loop Gain vs. PM 90 0 70 80 6.0 60 Frequency (Hz) 90 5.5 FIGURE 2-27: Gain Bandwidth Product and Phase Margin vs. Common Mode Input Voltage. 1.E+1 1.E+5 1.E+6 1.E+7 100M 1.E+8 1.E+9 10 1.E+2 100 1.E+3 1k 1.E+4 10k 100k 1M 10M 1G FIGURE 2-25: Frequency. 5.0 10 Common Mode Input Voltage (V) 120 40 30 10M 1.E+7 CMRR and PSRR vs. 60 20 4.5 1M 1.E+6 40 Phase Margin (°) Open-Loop Gain (dB) FIGURE 2-24: Frequency. 10k 100k 1.E+4 1.E+5 Frequency (Hz) 30 GBWP 4.0 1k 1.E+3 Open-Loop Phase (°) 0 100 1.E+2 50 3.5 10 50 40 3.0 20 60 2.5 30 VDD = 5.5V VDD = 2.5V 1.5 40 70 1.0 50 60 0.5 60 80 0.0 PSRR+ PSRR- CMRR 70 PM -0.5 70 Gain Bandwidth Product (MHz) CMRR, PSRR (dB) 80 Gain Bandwidth Product (MHz) 90 2.0 2.3 1000 100 10 G = 101 V/V G = 11 V/V G = 1 V/V 1 0.1 1k 10k 100k 1.0E+06 1M 10M 1.0E+08 100M 1.0E+03 1.0E+04 1.0E+05 1.0E+07 Frequency (Hz) FIGURE 2-29: Closed-Loop Output Impedance vs. Frequency. © 2009 Microchip Technology Inc. MCP651/2/5 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, 10 9 8 7 6 5 G = 1 V/V G = 2 V/V 4 G 4 V/V 3 2 1 0 10p 100p 1.0E-10 1n 10n 1.0E-11 1.0E-09 Normalized Capacitive Load; CL/G (F) FIGURE 2-30: Gain Peaking vs. Normalized Capacitive Load. © 2009 Microchip Technology Inc. Channel-to-Channel Separation (dB) Gain Peaking (dB) VL = VDD/2, RL = 1 kΩ to VL, CL = 20 pF, and CAL/CS = VSS. 150 140 130 120 110 100 90 80 70 RS = 10 kΩ 60 RS = 100 kΩ 50 1k 10k 1.E+03 1.E+04 RS = 0Ω RS = 1 kΩ RTI VCM = VDD/2 G = +1 V/V 100k 1M 1.E+05 1.E+06 Frequency (Hz) 10M 1.E+07 FIGURE 2-31: Channel-to-Channel Separation vs. Frequency. DS22146A-page 13 MCP651/2/5 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kΩ to VL, CL = 20 pF, and CAL/CS = VSS. Input Noise and Distortion 1.E+4 10μ Input Offset + Noise; VOS + e ni(t) (µV) 20 1.E+3 1μ 1.E+2 100n 1.E+1 10n 1 1.E+0 10 1.E+1 160 Representative Part NPBW = 0.1 Hz 10 5 0 -5 -10 -15 0 100 1.E+3 1k 1.E+4 10k 100k 1M 1.E+7 10M 1.E+2 1.E+5 1.E+6 Frequency (Hz) Input Noise Voltage Density 5 20 25 30 Time (min) THD + Noise (%) 100 VDD = 5.5V 80 60 40 35 40 45 50 VDD = 5.0V 0.1 BW = 22 Hz to > 500 kHz 0.01 0.001 G = 1 V/V G = 11 V/V BW = 22 Hz to 80 kHz f = 100 Hz 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.0 0.5 0.0 0 0.0001 1.E+2 100 Common Mode Input Voltage (V) FIGURE 2-33: Input Noise Voltage Density vs. Input Common Mode Voltage with f = 100 Hz. FIGURE 2-36: 1.E+3 1k 1.E+4 10k Frequency (Hz) 1.E+5 100k THD+N vs. Frequency. VDD = 2.5V VDD = 5.5V 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 f = 1 MHz -0.5 Input Noise Voltage Density (nV/Hz) 15 1 120 12 11 10 9 8 7 6 5 4 3 2 1 0 10 FIGURE 2-35: Input Noise plus Offset vs. Time with 0.1 Hz Filter. VDD = 2.5V 140 -0.5 Input Noise Voltage Density (nV/√Hz) FIGURE 2-32: vs. Frequency. 20 15 -20 1n 1.E+0 0.1 1.E-1 1.5 Input Noise Voltage Density (nV/Hz) 2.4 Common Mode Input Voltage (V) FIGURE 2-34: Input Noise Voltage Density vs. Input Common Mode Voltage with f = 1 MHz. DS22146A-page 14 © 2009 Microchip Technology Inc. MCP651/2/5 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kΩ to VL, CL = 20 pF, and CAL/CS = VSS. 2.5 Time Response VIN 0 Output Voltage (V) Output Voltage (10 mV/div) VDD = 5.5V G=1 VOUT 20 40 60 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 80 100 120 140 160 180 200 Time (ns) Non-inverting Small Signal VIN VOUT 100 200 FIGURE 2-40: Response. 300 400 500 Time (ns) 600 700 Input, Output Voltages (V) VIN VOUT 800 Inverting Large Signal Step 7 VDD = 5.5V G=1 VDD = 5.5V G=2 VIN 6 5 VOUT 4 3 2 1 0 -1 0 100 200 Output Voltage (10 mV/div) FIGURE 2-38: Step Response. 300 400 500 Time (ns) 600 700 0 800 Non-inverting Large Signal VDD = 5.5V G = -1 RF = 499Ω VOUT 50 FIGURE 2-39: Response. 100 150 200 250 Time (ns) 300 350 400 Inverting Small Signal Step © 2009 Microchip Technology Inc. 1 2 3 4 5 6 Time (ms) 7 8 9 10 FIGURE 2-41: The MCP651/2/5 family shows no input phase reversal with overdrive. VIN 0 VDD = 5.5V G = -1 RF = 499Ω 0 Slew Rate (V/µs) Output Voltage (V) FIGURE 2-37: Step Response. 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 60 55 50 45 40 35 30 25 20 15 10 5 0 Falling Edge VDD = 5.5V VDD = 2.5V Rising Edge -50 -25 FIGURE 2-42: Temperature. 0 25 50 75 Ambient Temperature (°C) 100 125 Slew Rate vs. Ambient DS22146A-page 15 MCP651/2/5 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kΩ to VL, CL = 20 pF, and CAL/CS = VSS. Maximum Output Voltage Swing (VP-P) 10 VDD = 5.5V VDD = 2.5V 1 0.1 100k 1.E+05 1M 10M 1.E+06 1.E+07 Frequency (Hz) 100M 1.E+08 FIGURE 2-43: Maximum Output Voltage Swing vs. Frequency. DS22146A-page 16 © 2009 Microchip Technology Inc. MCP651/2/5 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kΩ to VL, CL = 20 pF, and CAL/CS = VSS. Calibration and Chip Select Response 1.1 1.0 0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0.0 0.40 CAL/CS = VDD CAL/CS Hysteresis (V) CAL/CS Current (µA) 2.6 0.35 0.30 0.20 0.10 0.05 0.00 -50 IDD Op Amp turns off Op Amp turns on Calibration starts CAL/CS VOUT 0 1 2 3 4 5 6 Time (ms) 7 8 9 8 6 4 2 0 -2 -4 -6 -8 -10 -12 Calibration starts Op Amp turns off Op Amp turns on CAL/CS VOUT 0 1 2 3 4 5 6 Time (ms) 7 8 9 Power Supply Current; IDD (mA) IDD 10 FIGURE 2-46: CAL/CS Voltage, Output Voltage and Supply Current (for Side A) vs. Time with VDD = 5.5V. © 2009 Microchip Technology Inc. 100 125 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 -50 10 8 6 4 2 0 -2 -4 -6 -8 -10 -12 -14 VDD = 5.5V G=1 VL = 0V 0 25 50 75 Ambient Temperature (°C) 4.0 10 FIGURE 2-45: CAL/CS Voltage, Output Voltage and Supply Current (for Side A) vs. Time with VDD = 2.5V. 11 10 9 8 7 6 5 4 3 2 1 0 -1 CAL/CS Turn On Time (ms) VDD = 2.5V G=1 VL = 0V -25 FIGURE 2-47: CAL/CS Hysteresis vs. Ambient Temperature. -25 0 25 50 75 Ambient Temperature (°C) 100 125 FIGURE 2-48: CAL/CS Turn On Time vs. Ambient Temperature. 8 CAL/CS Pull-down Resistor (MΩ) 9 8 7 6 5 4 3 2 1 0 -1 CAL/CS Current vs. Power Power Supply Current; IDD (mA) FIGURE 2-44: Supply Voltage. CAL/CS, V OUT (V) VDD = 2.5V 0.15 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Power Supply Voltage (V) CAL/CS, V OUT (V) VDD = 5.5V 0.25 Representative Part 7 6 5 4 3 2 1 0 -50 -25 0 25 50 75 Ambient Temperature (°C) 100 125 FIGURE 2-49: CAL/CS’s Pull-down Resistor (RPD) vs. Ambient Temperature. DS22146A-page 17 MCP651/2/5 Note: Unless otherwise indicated, TA = +25°C, VDD = +2.5V to 5.5V, VSS = GND, VCM = VDD/3, VOUT = VDD/2, VL = VDD/2, RL = 1 kΩ to VL, CL = 20 pF, and CAL/CS = VSS. 1.E-06 CAL/CS = VDD -1 Output Leakage Current (A) Negative Power Supply Current; ISS (µA) 0 -2 -3 -4 +125°C +85°C +25°C -40°C -5 -6 Power Supply Voltage (V) FIGURE 2-50: Quiescent Current in Shutdown vs. Power Supply Voltage. DS22146A-page 18 6.5 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 -7 CAL/CS = VDD = 5.5V 1.E-07 +125°C 1.E-08 +85°C 1.E-09 1.E-10 +25°C 1.E-11 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 Output Voltage (V) FIGURE 2-51: Output Voltage. Output Leakage Current vs. © 2009 Microchip Technology Inc. MCP651/2/5 3.0 PIN DESCRIPTIONS Descriptions of the pins are listed in Table 3-1. TABLE 3-1: MCP651 PIN FUNCTION TABLE MCP652 MCP655 Symbol Description SOIC SOIC DFN MSOP DFN 6 2 3 4 1 2 3 4 1 2 3 4 1 2 3 4 1 2 3 4 VOUT, VOUTA VIN–, VINA– VIN+, VINA+ VSS Output (op amp A) Inverting Input (op amp A) Non-inverting Input (op amp A) Negative Power Supply 8 — — 5 5 CAL/CS, CALA/CSA Calibrate/Chip Select Digital Input (op amp A) — — — — 7 5 1 — — 5 6 7 8 — — — — 5 6 7 8 — — 9 6 7 8 9 10 — — — 6 7 8 9 10 — — 11 CALB/CSB VINB+ VINB– VOUTB VDD VCAL NC EP Calibrate/Chip Select Digital Input (op amp B) Non-inverting Input (op amp B) Inverting Input (op amp B) Output (op amp B) Positive Power Supply Calibration Common Mode Voltage Input No Internal Connection Exposed Thermal Pad (EP); must be connected to VSS 3.1 Analog Outputs The analog output pins (VOUT) are low-impedance voltage sources. 3.2 Analog Inputs The non-inverting and inverting inputs (VIN+, VIN–, …) are high-impedance CMOS inputs with low bias currents. 3.3 Power Supply Pins The positive power supply (VDD) is 2.5V to 5.5V higher than the negative power supply (VSS). For normal operation, the other pins are between VSS and VDD. Typically, these parts are used in a single (positive) supply configuration. In this case, VSS is connected to ground and VDD is connected to the supply. VDD will need bypass capacitors. 3.4 Calibration Common Mode Voltage Input A low impedance voltage placed at this input (VCAL) analog input will set the op amps’ common mode input voltage during calibration. If this pin is left open, the common mode input voltage during calibration is approximately VDD/3. The internal resistor divider is disconnected from the supplies whenever the part is not in calibration. © 2009 Microchip Technology Inc. 3.5 Calibrate/Chip Select Digital Input This input (CAL/CS, …) is a CMOS, Schmitt-triggered input that affects the calibration and low power modes of operation. When this pin goes high, the part is placed into a low power mode and the output is high-Z. When this pin goes low, a calibration sequence is started (which corrects VOS). At the end of the calibration sequence, the output becomes low impedance and the part resumes normal operation. An internal POR triggers a calibration event when the part is powered on, or when the supply voltage drops too low. Thus, the MCP652 parts are calibrated, even though they do not have a CAL/CS pin. 3.6 Exposed Thermal Pad (EP) There is an internal connection between the Exposed Thermal Pad (EP) and the VSS pin; they must be connected to the same potential on the Printed Circuit Board (PCB). This pad can be connected to a PCB ground plane to provide a larger heat sink. This improves the package thermal resistance (θJA). DS22146A-page 19 MCP651/2/5 NOTES: DS22146A-page 20 © 2009 Microchip Technology Inc. MCP651/2/5 4.0 APPLICATIONS The MCP651/2/5 family of self-zeroed op amps is manufactured using Microchip’s state of the art CMOS process. It is designed for low cost, low power and high precision applications. Its low supply voltage, low quiescent current and wide bandwidth makes the MCP651/2/5 ideal for battery-powered applications. 4.1 Calibration and Chip Select These op amps include circuitry for dynamic calibration of the offset voltage (VOS). 4.1.1 mCal CALIBRATION CIRCUITRY The internal mCal circuitry, when activated, starts a delay timer (to wait for the op amp to settle to its new bias point), then calibrates the input offset voltage (VOS). The mCal circuitry is triggered at power-up (and after some power brown out events) by the internal POR, and by the memory’s Parity Detector. The power up time, when the mCal circuitry triggers the calibration sequence, is 200 ms (typical). 4.1.2 CAL/CS PIN The CAL/CS pin gives the user a means to externally demand a low power mode of operation, then to calibrate VOS. Using the CAL/CS pin makes it possible to correct VOS as it drifts over time (1/f noise and aging; see Figure 2-35) and across temperature. The CAL/CS pin performs two functions: it places the op amp(s) in a low power mode when it is held high, and starts a calibration event (correction of VOS) after a rising edge. 4.1.3 This part includes an internal Power On Reset (POR) to protect the internal calibration memory cells. The POR monitors the power supply voltage (VDD). When the POR detects a low VDD event, it places the part into the low power mode of operation. When the POR detects a normal VDD event, it starts a delay counter, then triggers an calibration event. The additional delay gives a total POR turn on time of 200 ms (typical); this is also the power up time (since the POR is triggered at power up). 4.1.4 Once the calibration event is completed, the amplifier is reconnected to the external circuitry. The turn on time, when calibration is started with the CAL/CS pin, is 3 ms (typical). There is an internal 5 MΩ pull-down resistor tied to the CAL/CS pin. If the CAL/CS pin is left floating, the amplifier operates normally. © 2009 Microchip Technology Inc. PARITY DETECTOR A parity error detector monitors the memory contents for any corruption. In the rare event that a parity error is detected (e.g., corruption from an alpha particle), a POR event is automatically triggered. This will cause the input offset voltage to be re-corrected, and the op amp will not return to normal operation for a period of time (the POR turn on time, tPON). 4.1.5 CALIBRATION INPUT PIN A VCAL pin is available in some options (e.g., the single MCP651) for those applications that need the calibration to occur at an internally driven common mode voltage other than VDD/3. Figure 4-1 shows the reference circuit that internally sets the op amp’s common mode reference voltage (VCM_INT) during calibration (the resistors are disconnected from the supplies at other times). The 5 kΩ resistor provides over-current protection for the buffer. To op amp during calibration VDD While in the low power mode, the quiescent current is quite small (ISS = -3 µA, typical). The output is also is in a High-Z state. During the calibration event, the quiescent current is near, but smaller than, the specified quiescent current (6 mA, typical). The output continues in the High-Z state, and the inputs are disconnected from the external circuit, to prevent internal signals from affecting circuit operation. The op amp inputs are internally connected to a common mode voltage buffer and feedback resistors. The offset is corrected (using a digital state machine, logic and memory), and the calibration constants are stored in memory. INTERNAL POR 300 kΩ VCAL 150 kΩ VCM_INT 5 kΩ BUFFER VSS FIGURE 4-1: Input Circuitry. Common-Mode Reference’s When the VCAL pin is left open, the internal resistor divider generates a VCM_INT of approximately VDD/3, which is near the center of the input common mode voltage range. It is recommended that an external capacitor from VCAL to ground be added to improve noise immunity. When the VCAL pin is driven by an external voltage source, which is within its specified range, the op amp will have its input offset voltage calibrated at that common mode input voltage. Make sure that VCAL is within its specified range. DS22146A-page 21 MCP651/2/5 It is possible to use an external resistor voltage divider to modify VCM_INT; see Figure 4-2. The internal circuitry at the VCAL pin looks like 100 kΩ tied to VDD/3. The parallel equivalent of R1 and R2 should be much smaller than 100 kΩ to minimize differences in matching and temperature drift between the internal and external resistors. Again, make sure that VCAL is within its specified range. VDD MCP65X R1 VCAL C1 R2 VSS FIGURE 4-2: Resistors. Setting VCM with External For instance, a design goal to set VCM_INT = 0.1V when VDD = 2.5V could be met with: R1 = 24.3 kΩ, R2 = 1.00 kΩ and C1 = 100 nF. This will keep VCAL within its range for any VDD, and should be close enough to 0V for ground based applications. 4.2 4.2.1 VIN+ Bond Pad VSS Bond Pad FIGURE 4-3: Structures. Simplified Analog Input ESD In order to prevent damage and/or improper operation of these amplifiers, the circuit must limit the currents (and voltages) at the input pins (see Section 1.1 “Absolute Maximum Ratings †”). Figure 4-4 shows the recommended approach to protecting these inputs. The internal ESD diodes prevent the input pins (VIN+ and VIN–) from going too far below ground, and the resistors R1 and R2 limit the possible current drawn out of the input pins. Diodes D1 and D2 prevent the input pins (VIN+ and VIN–) from going too far above VDD, and dump any currents onto VDD. When implemented as shown, resistors R1 and R2 also limit the current through D1 and D2. VDD PHASE REVERSAL INPUT VOLTAGE AND CURRENT LIMITS The ESD protection on the inputs can be depicted as shown in Figure 4-3. This structure was chosen to protect the input transistors, and to minimize input bias current (IB). The input ESD diodes clamp the inputs when they try to go more than one diode drop below VSS. They also clamp any voltages that go too far above VDD; their breakdown voltage is high enough to allow normal operation, and low enough to bypass quick ESD events within the specified limits. Bond V – IN Pad Input Stage Input The input devices are designed to not exhibit phase inversion when the input pins exceed the supply voltages. Figure 2-41 shows an input voltage exceeding both supplies with no phase inversion. 4.2.2 VDD Bond Pad V1 V2 D1 R1 D2 MCP65X VOUT R2 VSS – (minimum expected V1) 2 mA VSS – (minimum expected V2) R2 > 2 mA R1 > FIGURE 4-4: Inputs. Protecting the Analog It is also possible to connect the diodes to the left of the resistor R1 and R2. In this case, the currents through the diodes D1 and D2 need to be limited by some other mechanism. The resistors then serve as in-rush current limiters; the DC current into the input pins (VIN+ and VIN–) should be very small. A significant amount of current can flow out of the inputs (through the ESD diodes) when the common mode voltage (VCM) is below ground (VSS); see Figure 2-15. Applications that are high impedance may need to limit the usable voltage range. DS22146A-page 22 © 2009 Microchip Technology Inc. MCP651/2/5 4.2.3 NORMAL OPERATION 4.3.0.3 The input stage of the MCP651/2/5 op amps uses a differential PMOS input stage. It operates at low common mode input voltage (VCM), with VCM up to VDD – 1.3V and down to VSS – 0.3V. The input offset voltage (VOS) is measured at VCM = VSS – 0.3V and VDD – 1.3V to ensure proper operation. See Figure 2-6 and Figure 2-7 for temperature effects. When operating at very low non-inverting gains, the output voltage is limited at the top by the VCM range (< VDD – 1.3V); see Figure 4-5 Power Dissipation Since the output short circuit current (ISC) is specified at ±100 mA (typical), these op amps are capable of both delivering and dissipating significant power. Two common loads, and their impact on the op amp’s power dissipation, will be discussed. Figure 4-7 shows a resistive load (RL) with a DC output voltage (VOUT). VL is RL’s ground point, VSS is usually ground (0V) and IOUT is the output current. The input currents are assumed to be negligible. VDD VDD MCP65X VIN IDD VOUT VSS VL FIGURE 4-7: Diagram for Resistive Load Power Calculations. Rail-to-Rail Output The DC currents are: Maximum Output Voltage The Maximum Output Voltage (see Figure 2-16 and Figure 2-17) describes the output range for a given load. For instance, the output voltage swings to within 15 mV of the negative rail with a 1 kΩ load tied to VDD/2. 4.3.0.2 RL ISS FIGURE 4-5: Unity Gain Voltage Limitations for Linear Operation. 4.3.0.1 VOUT MCP65X V SS < V IN, V OUT ≤ V DD – 1.3V 4.3 IOUT EQUATION 4-1: V OUT – V L I OUT = ------------------------RL I DD ≈ I Q + max ( 0, I OUT ) I SS ≈ – I Q + min ( 0, I OUT ) Where: Output Current Figure 4-6 shows the possible combinations of output voltage (VOUT) and output current (IOUT). IOUT is positive when it flows out of the op amp into the external circuit. IQ = Quiescent supply current for one op amp (mA/amplifier) VOUT = A DC value (V) EQUATION 4-2: VOH Limited (VDD = 5.5V) P OA = I DD ( V DD – V OUT ) + I SS ( V SS – V OUT ) RL = 100Ω -ISC Limited RL = 1 kΩ RL = 10Ω +ISC Limited 6.0 5.5 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 -0.5 The maximum op amp power, for resistive loads at DC, occurs when VOUT is halfway between VDD and VL or halfway between VSS and VL: IOUT (mA) 120 80 100 60 40 20 0 -20 EQUATION 4-3: -40 -60 -80 -100 VOL Limited -120 VOUT (V) The DC op amp power is: max ( P OA ) = I DD ( V DD – V SS ) 2 FIGURE 4-6: Output Current. © 2009 Microchip Technology Inc. max ( V DD – V L, V L – V SS ) + ----------------------------------------------------------------4R L DS22146A-page 23 MCP651/2/5 Figure 4-7 shows a capacitive load (CL), which is driven by a sine wave with DC offset. The capacitive load causes the op amp to output higher currents at higher frequencies. Because the output rectifies IOUT, the op amp’s dissipated power increases (even though the capacitor does not dissipate power). EQUATION 4-7: n P PKG = ∑ POA k=1 Where: n = Number of op amps in package (1 or 2) VDD IDD IOUT VOUT MCP65X ISS CL The maximum ambient to junction temperature rise (ΔTJA) and junction temperature (TJ) can be calculated using the maximum expected package power (PPKG), ambient temperature (TA) and the package thermal resistance (θJA) found in Table 1-4: EQUATION 4-8: VSS ΔT JA = P PKG θ JA FIGURE 4-8: Diagram for Capacitive Load Power Calculations. The output voltage is assumed to be: T J = T A + ΔT JA The worst case power de-rating for the op amps in a particular package can be easily calculated: EQUATION 4-4: V OUT = V DC + V AC sin ( ω t ) EQUATION 4-9: Where: T Jmax – T A P PKG ≤ -------------------------θ JA VDC = DC offset (V) VAC = Peak output swing (VPK) ω = Radian frequency (2π f) (rad/s) Where: TJmax = Absolute maximum junction temperature (°C) TA = Ambient temperature (°C) The op amp’s currents are: EQUATION 4-5: dV OUT I OUT = C L ⋅ ----------------- = V AC ω C L cos ( ω t ) dt I DD ≈ I Q + max ( 0, I OUT ) I SS ≈ – I Q + min ( 0, I OUT ) Where: IQ = Quiescent supply current for one op amp (mA/amplifier) The op amp’s instantaneous power, average power and peak power are: Several techniques are available to reduce ΔTJA for a given package: • Reduce θJA - Use another package - Improve the PCB layout (ground plane, etc.) - Add heat sinks and air flow • Reduce max(PPKG) - Increase RL - Decrease CL - Limit IOUT using RISO (see Figure 4-9) - Decrease VDD EQUATION 4-6: P OA = I DD ( V DD – V OUT ) + I SS ( V SS – V OUT ) 4V AC fC L⎞ -⎠ ave ( P OA ) = ( V DD – V SS ) ⎛⎝ I Q + ----------------------- π max ( P OA ) = ( V DD – V SS ) ( I Q + 2V AC fC L ) The power dissipated in a package depends on the powers dissipated by each op amp in that package: DS22146A-page 24 © 2009 Microchip Technology Inc. MCP651/2/5 4.4 Improving Stability 4.4.1 4.4.2 CAPACITIVE LOADS Driving large capacitive loads can cause stability problems for voltage feedback op amps. As the load capacitance increases, the feedback loop’s phase margin decreases and the closed-loop bandwidth is reduced. This produces gain peaking in the frequency response, with overshoot and ringing in the step response. See Figure 2-30. A unity gain buffer (G = +1) is the most sensitive to capacitive loads, though all gains show the same general behavior. Figure 4-11 shows an op amp circuit that represents non-inverting amplifiers (VM is a DC voltage and VP is the input) or inverting amplifiers (VP is a DC voltage and VM is the input). The capacitances CN and CG represent the total capacitance at the input pins; they include the op amp’s common mode input capacitance (CCM), board parasitic capacitance and any capacitor placed in parallel. When driving large capacitive loads with these op amps (e.g., > 20 pF when G = +1), a small series resistor at the output (RISO in Figure 4-9) improves the feedback loop’s phase margin (stability) by making the output load resistive at higher frequencies. The bandwidth will be generally lower than the bandwidth with no capacitive load. RG RF GAIN PEAKING RN VP MCP65X VOUT VM RG FIGURE 4-11: Capacitance. RISO CN CG RF Amplifier with Parasitic VOUT CG acts in parallel with RG (except for a gain of +1 V/V), which causes an increase in gain at high frequencies. CG also reduces the phase margin of the feedback loop, which becomes less stable. This effect can be reduced by either reducing CG or RF. FIGURE 4-9: Output Resistor, RISO Stabilizes Large Capacitive Loads. CN and RN form a low-pass filter that affects the signal at VP. This filter has a single real pole at 1/(2πRNCN). Figure 4-10 gives recommended RISO values for different capacitive loads and gains. The x-axis is the normalized load capacitance (CL/GN), where GN is the circuit’s noise gain. For non-inverting gains, GN and the Signal Gain are equal. For inverting gains, GN is 1+|Signal Gain| (e.g., -1 V/V gives GN = +2 V/V). The largest value of RF that should be used depends on noise gain (see GN in Section 4.4.1 “Capacitive Loads”) and CG. Figure 4-12 shows the maximum recommended RF for several CG values. CL MCP65X 100 Recommended R ISO (Ω) 1.E+05 100k Maximum Recommended RF (Ω) RN GN > +1 V/V CG = 10 pF CG = 32 pF CG = 100 pF CG = 320 pF CG = 1 nF 1.E+04 10k 1.E+03 1k 10 GN = +1 GN ≥ +2 1.E+02 100 1 1 10p 1.E-11 100p 1n 1.E-10 1.E-09 Normalized Capacitance; CL/GN (F) 10n 1.E-08 FIGURE 4-10: Recommended RISO Values for Capacitive Loads. After selecting RISO for your circuit, double check the resulting frequency response peaking and step response overshoot. Modify RISO’s value until the response is reasonable. Bench evaluation and simulations with the MCP651/2/5 SPICE macro model are helpful. © 2009 Microchip Technology Inc. FIGURE 4-12: RF vs. Gain. 10 Noise Gain; GN (V/V) 100 Maximum Recommended Figure 2-37 and Figure 2-38 show the small signal and large signal step responses at G = +1 V/V. The unity gain buffer usually has RF = 0Ω and RG open. Figure 2-39 and Figure 2-40 show the small signal and large signal step responses at G = -1 V/V. Since the noise gain is 2 V/V and CG ≈ 10 pF, the resistors were chosen to be RF = RG = 499Ω and RN = 249Ω. DS22146A-page 25 MCP651/2/5 It is also possible to add a capacitor (CF) in parallel with RF to compensate for the de-stabilizing effect of CG. This makes it possible to use larger values of RF. The conditions for stability are summarized in Equation 4-10. EQUATION 4-10: Given: G N1 = 1 + R F ⁄ R G G N2 = 1 + C G ⁄ C F fF = 1 ⁄ ( 2 π RF CF ) f Z = f F ( G N1 ⁄ G N2 ) We need: f F ≤ f GBWP ⁄ ( 2G N2 ) , G N1 < G N2 Use coax cables, or low inductance wiring, to route signal and power to and from the PCB. Mutual and self inductance of power wires is often a cause of crosstalk and unusual behavior. 4.7 Typical Applications 4.7.1 POWER DRIVER WITH HIGH GAIN Figure 4-13 shows a power driver with high gain (1 + R2/R1). The MCP651/2/5 op amp’s short circuit current makes it possible to drive significant loads. The calibrated input offset voltage supports accurate response at high gains. R3 should be small, and equal to R1||R2, in order to minimize the bias current induced offset. f F ≤ f GBWP ⁄ ( 4G N1 ) , G N1 > G N2 4.5 Power Supply With this family of operational amplifiers, the power supply pin (VDD for single supply) should have a local bypass capacitor (i.e., 0.01 µF to 0.1 µF) within 2 mm for good high frequency performance. Surface mount, multilayer ceramic capacitors, or their equivalent, should be used. These op amps require a bulk capacitor (i.e., 2.2 µF or larger) within 50 mm to provide large, slow currents. Tantalum capacitors, or their equivalent, may be a good choice. This bulk capacitor can be shared with other nearby analog parts as long as crosstalk through the supplies does not prove to be a problem. 4.6 High Speed PCB Layout These op amps are fast enough that a little extra care in the PCB (Printed Circuit Board) layout can make a significant difference in performance. Good PC board layout techniques will help you achieve the performance shown in the specifications and Typical Performance Curves; it will also help you minimize EMC (Electro-Magnetic Compatibility) issues. Use a solid ground plane. Connect the bypass local capacitor(s) to this plane with minimal length traces. This cuts down inductive and capacitive crosstalk. Separate digital from analog, low speed from high speed, and low power from high power. This will reduce interference. Keep sensitive traces short and straight. Separate them from interfering components and traces. This is especially important for high frequency (low rise time) signals. R1 VDD/2 R2 VOUT RL R3 VIN MCP65X FIGURE 4-13: 4.7.2 Power Driver. OPTICAL DETECTOR AMPLIFIER Figure 4-14 shows a transimpedance amplifier, using the MCP651 op amp, in a photo detector circuit. The photo detector is a capacitive current source. The op amp’s input common mode capacitance (5 pF, typical) acts in parallel with CD. RF provides enough gain to produce 10 mV at VOUT. CF stabilizes the gain and limits the transimpedance bandwidth to about 1.1 MHz. RF’s parasitic capacitance (e.g., 0.2 pF for a 0805 SMD) acts in parallel with CF. CF 1.5 pF Photo Detector ID 100 nA RF 100 kΩ CD 30pF VOUT MCP651 VDD/2 FIGURE 4-14: Transimpedance Amplifier for an Optical Detector. Sometimes, it helps to place guard traces next to victim traces. They should be on both sides of the victim trace, and as close as possible. Connect guard traces to ground plane at both ends, and in the middle for long traces. DS22146A-page 26 © 2009 Microchip Technology Inc. MCP651/2/5 4.7.3 H-BRIDGE DRIVER Figure 4-15 shows the MCP652 dual op amp used as a H-bridge driver. The load could be a speaker or a DC motor. ½ MCP652 VIN RF RL RGT RGB VDD/2 VOT RF RF VOB ½ MCP652 FIGURE 4-15: H-Bridge Driver. This circuit automatically makes the noise gains (GN) equal, when the gains are set properly, so that the frequency responses match well (in magnitude and in phase). Equation 4-11 shows how to calculate RGT and RGB so that both op amps have the same DC gains; GDM needs to be selected first. EQUATION 4-11: V OT – V OB G DM ≡ -------------------------------- ≥ 2 V/V V IN – V DD ⁄ 2 RF R GT = -------------------------------( G DM ⁄ 2 ) – 1 RF R GB = -----------------G DM ⁄ 2 Equation 4-12 gives the resulting common mode and differential mode output voltages. EQUATION 4-12: V OT + V OB V DD -------------------------- = ---------2 2 V DD⎞ V OT – V OB = G DM ⎛⎝ V IN – ---------2 ⎠ © 2009 Microchip Technology Inc. DS22146A-page 27 MCP651/2/5 NOTES: DS22146A-page 28 © 2009 Microchip Technology Inc. MCP651/2/5 5.0 DESIGN AIDS Microchip provides the basic design aids needed for the MCP651/2/5 family of op amps. 5.1 SPICE Macro Model The latest SPICE macro model for the MCP651/2/5 op amps is available on the Microchip web site at www.microchip.com. This model is intended to be an initial design tool that works well in the op amp’s linear region of operation over the temperature range. See the model file for information on its capabilities. Bench testing is a very important part of any design and cannot be replaced with simulations. Also, simulation results using this macro model need to be validated by comparing them to the data sheet specifications and characteristic curves. 5.2 FilterLab® Software Microchip’s FilterLab® software is an innovative software tool that simplifies analog active filter (using op amps) design. Available at no cost from the Microchip web site at www.microchip.com/filterlab, the Filter-Lab design tool provides full schematic diagrams of the filter circuit with component values. It also outputs the filter circuit in SPICE format, which can be used with the macro model to simulate actual filter performance. 5.3 Mindi™ Circuit Designer & Simulator Microchip’s Mindi™ Circuit Designer & Simulator aids in the design of various circuits useful for active filter, amplifier and power management applications. It is a free online circuit designer & simulator available from the Microchip web site at www.microchip.com/mindi. This interactive circuit designer & simulator enables designers to quickly generate circuit diagrams, and simulate circuits. Circuits developed using the Mindi Circuit Designer & Simulator can be downloaded to a personal computer or workstation. 5.4 Microchip Advanced Part Selector (MAPS) 5.5 Analog Demonstration and Evaluation Boards Microchip offers a broad spectrum of Analog Demonstration and Evaluation Boards that are designed to help customers achieve faster time to market. For a complete listing of these boards and their corresponding user’s guides and technical information, visit the Microchip web site at www.microchip.com/analog tools. Some boards that are especially useful are: • • • • • • MCP6XXX Amplifier Evaluation Board 1 MCP6XXX Amplifier Evaluation Board 2 MCP6XXX Amplifier Evaluation Board 3 MCP6XXX Amplifier Evaluation Board 4 Active Filter Demo Board Kit 8-Pin SOIC/MSOP/TSSOP/DIP Evaluation Board, P/N SOIC8EV 5.6 Application Notes The following Microchip Application Notes are available on the Microchip web site at www.microchip. com/appnotes and are recommended as supplemental reference resources. • ADN003: “Select the Right Operational Amplifier for your Filtering Circuits”, DS21821 • AN722: “Operational Amplifier Topologies and DC Specifications”, DS00722 • AN723: “Operational Amplifier AC Specifications and Applications”, DS00723 • AN884: “Driving Capacitive Loads With Op Amps”, DS00884 • AN990: “Analog Sensor Conditioning Circuits – An Overview”, DS00990 • AN1177: “Op Amp Precision Design: DC Errors”, DS01177 • AN1228: “Op Amp Precision Design: Random Noise”, DS01228 Some of these application notes, and others, are listed in the design guide: • “Signal Chain Design Guide”, DS21825 MAPS is a software tool that helps efficiently identify Microchip devices that fit a particular design requirement. Available at no cost from the Microchip website at www.microchip.com/maps, the MAPS is an overall selection tool for Microchip’s product portfolio that includes Analog, Memory, MCUs and DSCs. Using this tool, a customer can define a filter to sort features for a parametric search of devices and export side-by-side technical comparison reports. Helpful links are also provided for Data sheets, Purchase and Sampling of Microchip parts. © 2009 Microchip Technology Inc. DS22146A-page 29 MCP651/2/5 NOTES: DS22146A-page 30 © 2009 Microchip Technology Inc. MCP651/2/5 6.0 PACKAGING INFORMATION 6.1 Package Marking Information Example: 8-Lead DFN (3x3) (MCP652) Device XXXX YYWW NNN MCP652 Note: Applies to 8-Lead 3x3 DFN 8-Lead SOIC (150 mil) (MCP651, MCP652) XXXXXXXX XXXXYYWW NNN XXXX YYWW NNN e3 * Note: Example: Example: BAFC 0910 256 10-Lead MSOP (MCP655) Legend: XX...X Y YY WW NNN DABP 0913 256 MCP651E SN e3 0910 256 10-Lead DFN (3x3) (MCP655) XXXXXX YWWNNN Code DABP Example: 655EUN 910256 Customer-specific information Year code (last digit of calendar year) Year code (last 2 digits of calendar year) Week code (week of January 1 is week ‘01’) Alphanumeric traceability code Pb-free JEDEC designator for Matte Tin (Sn) This package is Pb-free. The Pb-free JEDEC designator ( e3 ) can be found on the outer packaging for this package. In the event the full Microchip part number cannot be marked on one line, it will be carried over to the next line, thus limiting the number of available characters for customer-specific information. © 2009 Microchip Technology Inc. DS22146A-page 31 MCP651/2/5 % 2% & %! % * %% 133)))& &3 " ) * ' !"#$ % * D e b N $ % % " % N L EXPOSED PAD E E2 K NOTE 1 1 2 D2 2 NOTE 1 1 BOTTOM VIEW TOP VIEW A NOTE 2 A3 A1 4% & 5&% 6!&( $ 55,, 6 6 67 8 9 % :./0 7 ; % 9 % "$$ . 0% %* + 7 5 % ,# , " "<"% ,2 +/0 = : 7 <"% , ,# = ( . + +. 5 + .. > = = " "5 % 0% %<"% 0% %5 % 0% % % ,# " " +/0 % ! " #$ %! & '(!%&! %( % ")%% % " * & & # "% ( % " + * ) ! % " & "% ,-. /01 / & % # % ! ))%!%% ,21 $ & '! ! )%!%% '$$& % ! ) 0 :/ DS22146A-page 32 © 2009 Microchip Technology Inc. MCP651/2/5 % 2% & %! % * %% 133)))& &3 © 2009 Microchip Technology Inc. " ) * ' !"#$ % * $ % % " % DS22146A-page 33 MCP651/2/5 % & ' (&))* 2% & %! % * %% 133)))& &3 " ) * ' !"#&'+,$ % * $ % % " % D e N E E1 NOTE 1 1 2 3 α h b h A2 A c φ L A1 L1 4% & 5&% 6!&( $ 55,, 6 6 8 /0 7 ; % * % "$$? 7 <"% " " * 67 9 % " " * β = = . = = = . , <"% 7 5 % . :/0 , +/0 /0 0 &$ @ % A . = . 2%5 % 5 = 2% % 5 2% I B = 9B 5 "* = . 5 "<"% ,2 ( + = . " $% D .B = .B " $% /%%& E .B = .B % ! " #$ %! & '(!%&! %( % ")%% % " ? $ %0 % % + & ","%!" &"$ %! "$ %! % # ".&& & "% ,-. /01 / & % # % ! ))%!%% ,21 $ & '! ! )%!%% '$$& % ! " ) 0 ./ DS22146A-page 34 © 2009 Microchip Technology Inc. MCP651/2/5 % & ' (&))* 2% & %! % * %% 133)))& &3 © 2009 Microchip Technology Inc. " ) * ' !"#&'+,$ % * $ % % " % DS22146A-page 35 MCP651/2/5 - % 2% & %! % * %% 133)))& &3 " ) * ' !"#$ % * D $ % % " % e b N N L K E E2 EXPOSED PAD NOTE 1 1 2 2 1 NOTE 1 D2 BOTTOM VIEW TOP VIEW A A1 A3 NOTE 2 4% & 5&% 6!&( $ 55,, 6 6 67 8 % ./0 7 ; % 9 % "$$ . 0% %* + 7 5 % ,# " "5 % ,2 +/0 +. 9 7 <"% , ,# , .9 . ( 9 . + 5 + . > = = " "<"% 0% %<"% 0% %5 % 0% % % ,# " " +/0 % ! " #$ %! & '(!%&! %( % ")%% % " * & & # "% ( % " + * ) ! % " & "% ,-. /01 / & % # % ! ))%!%% ,21 $ & '! ! )%!%% '$$& % ! ) 0 :+/ DS22146A-page 36 © 2009 Microchip Technology Inc. MCP651/2/5 - % 2% & %! % * %% 133)))& &3 © 2009 Microchip Technology Inc. " ) * ' !"#$ % * $ % % " % DS22146A-page 37 MCP651/2/5 - % )& ' ( .#&' $ 2% & %! % * %% 133)))& &3 " ) * ' % * $ % % " % D N E E1 NOTE 1 1 2 b e A A2 c φ L A1 L1 4% & 5&% 6!&( $ 55,, 6 6 ./0 7 ; % * % "$$ 7 <"% " " * 7 5 % 8 % " " * 67 = = . 9. . = . , <"% /0 , +/0 +/0 2%5 % 5 2% % 5 2% I B = 9B 5 9 = + "* : 9 .,2 5 "<"% ( . = ++ % ! " #$ %! & '(!%&! %( % ")%% % " & ","%!" &"$ %! "$ %! % # ".&& + & "% ,-. /01 / & % # % ! ))%!%% ,21 $ & '! ! )%!%% '$$& % ! " ) 0 / DS22146A-page 38 © 2009 Microchip Technology Inc. MCP651/2/5 APPENDIX A: REVISION HISTORY Revision A (April 2009) • Original Release of this Document. © 2009 Microchip Technology Inc. DS22146A-page 39 MCP651/2/5 NOTES: DS22146A-page 40 © 2009 Microchip Technology Inc. MCP651/2/5 PRODUCT IDENTIFICATION SYSTEM To order or obtain information, e.g., on pricing or delivery, refer to the factory or the listed sales office. PART NO. X /XX Device Temperature Range Package Device: MCP651: MCP651T: MCP652: MCP652T: MCP655: MCP655T: Single Op Amp Single Op Amp (Tape and Reel) (DFN and SOIC) Dual Op Amp Dual Op Amp (Tape and Reel) (DFN and SOIC) Dual Op Amp Dual Op Amp (Tape and Reel) (DFN and MSOP) Temperature Range: E = -40°C to +125°C Package: MF = Plastic Dual Flat, No Lead (3x3 DFN), 8-lead, 10-lead SN = Plastic Small Outline, (3.90 mm), 8-lead UN = Plastic Micro Small Outline, (MSOP), 10-lead © 2009 Microchip Technology Inc. Examples: a) MCP651T-E/SN: Tape and Reel, Extended Temperature, 8LD SOIC package. a) MCP652T-E/MF: Tape and Reel, Extended Temperature, 8LD DFN package. Tape and Reel, Extended Temperature, 8LD SOIC package. b) MCP652T-E/SN: a) MCP655T-E/MF: b) MCP655T-E/UN: Tape and Reel, Extended Temperature, 10LD DFN package. Tape and Reel, Extended Temperature, 10LD MSOP package. DS22146A-page 41 MCP651/2/5 NOTES: DS22146A-page 42 © 2009 Microchip Technology Inc. Note the following details of the code protection feature on Microchip devices: • Microchip products meet the specification contained in their particular Microchip Data Sheet. • Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the intended manner and under normal conditions. • There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data Sheets. Most likely, the person doing so is engaged in theft of intellectual property. • Microchip is willing to work with the customer who is concerned about the integrity of their code. • Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not mean that we are guaranteeing the product as “unbreakable.” Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act. Information contained in this publication regarding device applications and the like is provided only for your convenience and may be superseded by updates. It is your responsibility to ensure that your application meets with your specifications. MICROCHIP MAKES NO REPRESENTATIONS OR WARRANTIES OF ANY KIND WHETHER EXPRESS OR IMPLIED, WRITTEN OR ORAL, STATUTORY OR OTHERWISE, RELATED TO THE INFORMATION, INCLUDING BUT NOT LIMITED TO ITS CONDITION, QUALITY, PERFORMANCE, MERCHANTABILITY OR FITNESS FOR PURPOSE. Microchip disclaims all liability arising from this information and its use. Use of Microchip devices in life support and/or safety applications is entirely at the buyer’s risk, and the buyer agrees to defend, indemnify and hold harmless Microchip from any and all damages, claims, suits, or expenses resulting from such use. No licenses are conveyed, implicitly or otherwise, under any Microchip intellectual property rights. Trademarks The Microchip name and logo, the Microchip logo, Accuron, dsPIC, KEELOQ, KEELOQ logo, MPLAB, PIC, PICmicro, PICSTART, rfPIC, SmartShunt and UNI/O are registered trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. FilterLab, Linear Active Thermistor, MXDEV, MXLAB, SEEVAL, SmartSensor and The Embedded Control Solutions Company are registered trademarks of Microchip Technology Incorporated in the U.S.A. Analog-for-the-Digital Age, Application Maestro, CodeGuard, dsPICDEM, dsPICDEM.net, dsPICworks, dsSPEAK, ECAN, ECONOMONITOR, FanSense, In-Circuit Serial Programming, ICSP, ICEPIC, Mindi, MiWi, MPASM, MPLAB Certified logo, MPLIB, MPLINK, mTouch, PICkit, PICDEM, PICDEM.net, PICtail, PIC32 logo, PowerCal, PowerInfo, PowerMate, PowerTool, REAL ICE, rfLAB, Select Mode, Total Endurance, WiperLock and ZENA are trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. SQTP is a service mark of Microchip Technology Incorporated in the U.S.A. All other trademarks mentioned herein are property of their respective companies. © 2009, Microchip Technology Incorporated, Printed in the U.S.A., All Rights Reserved. Printed on recycled paper. Microchip received ISO/TS-16949:2002 certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona; Gresham, Oregon and design centers in California and India. The Company’s quality system processes and procedures are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping devices, Serial EEPROMs, microperipherals, nonvolatile memory and analog products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001:2000 certified. © 2009 Microchip Technology Inc. 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