AD AD8010AN 200 ma output current high speed amplifier Datasheet

a
200 mA Output Current
High Speed Amplifier
AD8010
FEATURES
200 mA of Output Current
9 ⍀ Load
SFDR –54 dBc @ 1 MHz
Differential Gain Error 0.04%, f = 4.43 MHz
Differential Phase Error 0.06ⴗ, f = 4.43 MHz
Maintains Video Specifications Driving Eight Parallel
75 ⍀ Loads
0.02% Differential Gain
0.03ⴗ Differential Phase
0.1 dB Gain Flatness to 60 MHz
THD –72 dBc @ 1 MHz, RL = 18.75 ⍀
IP3 42 dBm @ 5 MHz, RL = 18.75 ⍀
1 dB Gain Compression 21 dBm @ 5 MHz, RL = 100 ⍀
230 MHz –3 dB Bandwidth, G = +1, RL = 18.75 ⍀
800 V/␮s Slew Rate, R L = 18.75 ⍀
25 ns Settling Time to 0.1%
Available in 8-Lead DIP, 16-Lead Wide Body SOIC and
Thermally Enhanced 8-Lead SOIC
CONNECTION DIAGRAMS
8-Lead DIP and SOIC
NC 1
AD8010
8
NC
–IN 2
7
+VS
+IN 3
6
OUT
–VS 4
5
NC
NC = NO CONNECT
16-Lead Wide Body SOIC
NC 1
16
NC
2
15
NC
–IN 3
14
+VS
NC
APPLICATIONS
Video Distribution Amplifier
VDSL, xDSL Line Driver
Communications
ATE
Instrumentation
AD8010
NC 4
13
NC
+IN 5
12
OUT
NC 6
11
NC
–VS 7
10
NC
NC 8
9
NC
NC = NO CONNECT
PRODUCT DESCRIPTION
The AD8010 is a low power, high current amplifier capable of
delivering a minimum load drive of 175 mA. Signal performance
such as 0.02% and 0.03° differential gain and phase error is
maintained while driving eight 75 Ω back terminated video lines.
The current feedback amplifier features gain flatness to 60 MHz
and –3 dB (G = +1) signal bandwidth of 230 MHz and only
requires a typical of 15.5 mA supply current from ±5 V supplies.
These features make the AD8010 an ideal component for Video
Distribution Amplifiers or as the drive amplifier within high data
rate Digital Subscriber Line (VDSL and xDSL) systems.
75V
VOUT1
RF
RG
VOUT2
VOUT3
+5V
VOUT4
RS
VIN
AD8010
VOUT5
RT
–5V
VOUT6
The AD8010 is an ideal component choice for any application
that needs a driver that will maintain signal quality when driving
low impedance loads.
The AD8010 is offered in three package options: an 8-lead DIP,
16-lead wide body SOIC and a low thermal resistance 8-lead
SOIC, and operates over the industrial temperature range of
–40°C to +85°C.
VOUT7
VOUT8
75V
Figure 1. Video Distribution Amplifier
REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 1998
AD8010–SPECIFICATIONS
(@ +25ⴗC, VS = ⴞ5 V, G = +2, RL = 18.75 ⍀, RS+ = 150 ⍀, R F = RG = 604 ⍀ (R-16),
RF = RG = 562 ⍀ (N-8), RF = R G = 499 ⍀ (R-8). TMIN = –40ⴗC, TMAX = +85ⴗC unless otherwise noted)
Model
DYNAMIC PERFORMANCE
–3 dB Bandwidth
0.1 dB Bandwidth
Large Signal Bandwidth
Peaking
Slew
Rise and Fall Time
Settling Time
NOISE/HARMONIC PERFORMANCE
Distortion
2nd Harmonic
3rd Harmonic
IMD
IP3
1 dB Gain Compression
Input Noise Voltage
Input Noise Current
Differential Gain
Differential Phase
Conditions
Min
Typ
G = +1, V OUT = 0.2 V p-p
G = +2, V OUT = 0.2 V p-p
VOUT = 0.2 V p-p
VOUT = 4 V p-p
VOUT = 0.2 V p-p, < 5 MHz
VOUT = 2 V p-p
VOUT = 2 V p-p
0.1%, VOUT = 2 V p-p
180
130
30
230
190
60
90
0.02
800
2.0
25
MHz
MHz
MHz
MHz
dB
V/µs
ns
ns
–73
–58
–53
–67
–44
–77
–63
–57
–63
–50
–73
42
21
2
3
20
0.02
0.02
0.02
0.03
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBm
dBm
nV√Hz
pA√Hz
pA√Hz
%
%
Degrees
Degrees
VOUT = 2 V p-p
1 MHz
5 MHz
10 MHz
10 MHz, RL = 39 Ω
20 MHz
1 MHz
5 MHz
10 MHz
10 MHz, RL = 39 Ω
20 MHz
5 MHz ∆f = 10 kHz
5 MHz
5 MHz
f = 10 kHz
f = 10 kHz, +In
f = 20 kHz, –In
f = 4.43 MHz, RL = 150 Ω
f = 4.43 MHz, RL = 18.75 Ω
f = 4.43 MHz, RL = 150 Ω
f = 4.43 MHz, RL =18.75 Ω
DC PERFORMANCE
Input Offset Voltage
5
TMIN–TMAX
Offset Drift
Input Bias Current (–)
10
10
TMIN–TMAX
Input Bias Current (+)
6
TMIN–TMAX
INPUT CHARACTERISTICS
Input Resistance
Input Capacitance
Common-Mode Rejection Ratio
Input Common-Mode Voltage Range
Open Loop Transresistance
OUTPUT CHARACTERISTICS
Output Voltage Swing
RL = 18.75 Ω
RL = 150 Ω
Output Current
Short-Circuit Current
Capacitive Load Drive
+Input
–Input
VCM = ± 2.5 V
50
VOUT = ± 2.5 V
TMIN–TMAX
300
250
RL = 9 Ω
± 2.1
± 2.7
175
POWER SUPPLY
Operating Range
Quiescent Current
Power Supply Rejection Ratio
135
200
12
20
mV
mV
µV/°C
µA
µA
µA
µA
kΩ
Ω
pF
dB
V
kΩ
kΩ
± 2.5
± 3.0
200
240
40
V
V
mA
mA
pF
15.5
60
50
12
15
Units
125
12.5
2.75
54
± 2.5
500
± 4.5
TMIN to TMAX
+VS = +4 V to +6 V, –VS = +5 V
+VS = +5 V, –VS = –4 V to –6 V
Max
66
56
± 6.0
17
20
V
mA
mA
dB
dB
Specifications subject to change without notice.
–2–
REV. A
AD8010
ABSOLUTE MAXIMUM RATINGS 1
MAXIMUM POWER DISSIPATION
Supply␣ Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .12.6␣ V
Internal␣ Power␣ Dissipation2
Plastic␣ Package (N) . . . . . . . Observe Power Derating Curves
Small␣ Outline␣ Package (R) . . Observe Power Derating Curves
Wide Body SOIC (R-16) . . . . Observe Power Derating Curves
Input Voltage (Common-Mode) . . . . . . . . . . . . . . . . . . . . ± VS
Differential␣ Input␣ Voltage . . . . . . . . . . . . . . . . . . . . . . ± 1.2␣ V
Output Short Circuit Duration
.␣ . . . . . . . . . . . . . . . . . . . . . . Observe Power Derating Curves
Storage Temperature Range N, R . . . . . . . . –65°C to +125°C
Operating Temperature Range (A Grade) . . . –40°C to +85°C
Lead Temperature Range (Soldering␣ 10␣ sec) . . . . . . . . +300°C
The maximum power that can be safely dissipated by the
AD8010 is limited by the associated rise in junction temperature. The maximum safe junction temperature for plastic
encapsulated devices is determined by the glass transition temperature of the plastic, approximately +150°C. Temporarily
exceeding this limit may cause a shift in parametric performance due to a change in the stresses exerted on the die by the
package. Exceeding a junction temperature of +175°C for an
extended period can result in device failure.
NOTES
1Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2Specification is for device in free air:
8-Lead Plastic Package: θJA = 90°C/Watt
8-Lead SOIC Package: θJA = 122°C/Watt
16-Lead SOIC Package: θJA = 73°C/Watt
While the AD8010 is internally short circuit protected, this
may not be sufficient to guarantee that the maximum junction
temperature (+150°C) is not exceeded under all conditions. To
ensure proper operation, it is necessary to observe the maximum
power derating curves.
MAXIMUM POWER DISSIPATION – Watts
3.0
TJ = +1508C
2.5
8-LEAD MINI-DIP PACKAGE
2.0
16-LEAD SOIC
PACKAGE (WIDEBODY)
1.5
1.0
8-LEAD SOIC PACKAGE
0.5
0
–50 –40 –30 –20 –10 0 10 20 30 40 50 60 70 80 90
AMBIENT TEMPERATURE – 8C
Figure 2. Plot of Maximum Power Dissipation vs. Temperature
ORDERING GUIDE
Model
Temperature Range
Package Description
Package Options
AD8010AN
AD8010AR
AD8010AR-16
AD8010AR-REEL
AD8010AR-REEL7
AD8010AR-16-REEL
AD8010AR-16-REEL7
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
8-Lead Plastic DIP
8-Lead Plastic SOIC
16-Lead Wide Body SOIC
REEL SOIC
REEL SOIC
REEL SOIC
REEL SOIC
N-8
SO-8
R-16
13" REEL
7" REEL
13" REEL
7" REEL
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD8010 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
REV. A
–3–
WARNING!
ESD SENSITIVE DEVICE
60
dG
PERCENTAGE OF UNITS
50
DIFFERENTIAL GAIN – %
SAMPLE SIZE = 300
G = +2
f = 4.43MHz (PAL)
RL = 18.75V
40
dG
30
DIFFERENTIAL GAIN dG IN %
DIFFERENTIAL PHASE df IN Degrees
df
df
df
20
dG
df
10
df df
dG
0.05
0.10
0.04
0.08
DIFFERENTIAL GAIN
DIFFERENTIAL PHASE
0.02
0.04
0.01
0.02
df df df df df
0
0
0 0.01 0.02 0.03 0.04 0.05 0.06 0.07 0.08 0.09 0.10 0.11 0.12 0.13
dG (%)/df – Degrees
Figure 3. Distribution of Differential Gain (dG) and
Differential Phase (dφ); R L = 18.75 Ω
1
2
4
6
8
10
12
NUMBER OF VIDEO LOADS
16
0
45
G = +2
VO = 2V p-p
RL AS SHOWN
–55
40
INTERCEPT POINT – dBm
–50
2ND
–60
RL = 18.75V
–65
3RD
–70
–75
–80
3RD
2ND
RL = 100V
–85
35
30
G = +2
RL = 18.75V
25
20
15
10
–90
–95
1
2
3
4
5 6 7 8 9 10
FREQUENCY – MHz
5
20
1
Figure 4. Harmonic Distortion vs. Frequency; G = +2
10
FREQUENCY – MHz
6.20
6.15
G = +2
RL = 18.75V
VO = 0.2V p-p
+858C
4
G = +2
VO = 0.2V p-p
NUMBER OF VIDEO
LOADS AS SHOWN
6.4
6.3
GAIN FLATNESS – dB
+258C
6.10
6.05
6.0
–408C
5.95
100
Figure 7. Two-Tone, 3rd Order IMD Intercept vs.
Frequency; G = +2, RL = 18.75 Ω
6.5
GAIN FLATNESS – dB
14
Figure 6. Differential Gain and Phase vs. Number of Video
Loads Over Temperature (–40 °C to +85 °C); f = 4.43 MHz
–45
HARMONIC DISTORTION – dBc
0.06
0.03
DIFFERENTIAL PHASE – Degrees
AD8010–Typical Performance Characteristics
5.90
2
6
6.2
8
6.1
6.0
5.9
1
5.8
5.85
5.7
5.80
5.6
10
12
14
5.5
0.1
1
10
FREQUENCY – MHz
100
1
500
10
100
1000
FREQUENCY – MHz
Figure 5. Gain Flatness vs. Frequency Over Temperature
(–40 °C to +85 °C)
Figure 8. Gain Flatness vs. Frequency vs. Number of
Video Loads
–4–
REV. A
AD8010
0
5
4dBm
4dBm
–5
G = +2
RL = 18.75V
fO = 5MHz
Df = 10kHz
–15
–25
–35
–45
–55
–65
GAIN = 6.6
RF
–20
–69dBm
RG
–30
–40
50V
50V
150V
50V
–50
500kHz TONE SPACING FROM
500kHz TO 5.5MHz WITH
4 MISSING TONES
–60
–70
–69dBm
–75
–80
–90
–85
4.965
4.985
5.0
5.015
FREQUENCY – MHz
5.035
0
–35
RL = 18.75V
–65
–75
RF
RL = 100V
–85
RG
50V P
OUT
150V
–105
–10
3
4
5
6
7
FREQUENCY – MHz
8
9
10
P
RL1 = FOR RL = 100V IN
RL1 = 23.1 FOR RL = 18.75V
–8
–6
–4
–2
RL1
50V
0
2
POUT – dBm
G = +2
VO = 2V p-p
f = 5MHz
–60
–55
–95
2
–55
FREQUENCY = 5MHz
G = +2
RL = AS SHOWN (SEE SCHEMATIC)
HARMONIC DISTORTION – dBc
–45
1
Figure 12. Multitone Distortion; RL = 100 Ω
Figure 9. Intermodulation Distortion
TOTAL HARMONIC DISTORTION – dBc
PMEASURE = 10dBm
(FULL SCALE)
–10
PMEASURE – dBm
INTERMODULATION DISTORTION – dBm
POUT
50V
–65
–70
–75
–80
2ND
–85
3RD
–90
4
6
8
10
15
12
Figure 10. Total Harmonic Distortion vs. P OUT; G = +2
100
200
300
LOAD – V
400
500
Figure 13. Harmonic Distortion vs. Load
2
G = +1
G = +2
7.0
–2
G = +3
–3
–5
4.0
12
3.0
2.0
1.0
0.0
1
10
FREQUENCY – MHz
100
1
1000
Figure 11. Small Signal Closed-Loop Frequency
Response; RL = 18.75 Ω
REV. A
5.0
8
GAIN AS SHOWN
VO = 0.2V p-p
RL = 18.75V
–6
–7
0.1
4
1
6.0
–1
GAIN – dB
NORMALIZED GAIN – dB
0
–4
G = +2
VO = 0.2V p-p
NUMBER OF VIDEO LOADS AS SHOWN
8.0
1
10
100
FREQUENCY – MHz
1000
Figure 14. Closed-Loop Frequency Response vs.
Number of Video Loads
–5–
AD8010
–10
0
–10
–20
–20
–30
CMRR – dB
PSRR – dB
–30
–40
–50
–PSRR
–40
–50
–60
–70
–60
+PSRR
–80
–70
–90
–80
0.03
0.1
1
10
100
–100
0.1
500
1
FREQUENCY – MHz
10
FREQUENCY – MHz
100
500
Figure 18. CMRR vs. Frequency
Figure 15. PSRR vs. Frequency
310
G = +2
316
0
100
45
31
10
3.1
1
0.31
PHASE
90
31.6
135
10
TRANSRESISTANCE
3.16
180
1
225
PHASE – Degrees
100
TRANSRESISTANCE – kV
CLOSED-LOOP OUTPUT RESISTANCE – V
1000
0.1
0.031
0.1
1
10
FREQUENCY – MHz
100
0.316
10k
500
Figure 16. Closed-Loop Output Resistance vs. Frequency
G = +2
1.0
GAIN AS SHOWN
VO = 2V p-p
RL = 18.75V
NORMALIZED GAIN – dB
NORMALIZED GAIN – dB
1G
2.0
0
G = +2
–3
G = +10
–4
–5
0.0
–1.0
–2.0
–3.0
GAIN AS SHOWN
VO = 4V p-p
RL = 18.75V
G = +10
–4.0
–5.0
–6
–7
0.1
100M
3.0
G = +1
1
–2
1M
10M
FREQUENCY – Hz
Figure 19. Transresistance and Phase vs. Frequency;
RL = 18.75 Ω
2
–1
100k
–6.0
1
10
FREQUENCY – MHz
100
–7.0
0.1
1000
Figure 17. Large Signal Frequency Response; VO = 2 V p-p
0
10
FREQUENCY – MHz
100
1000
Figure 20. Large Signal Frequency Response; VO = 4 V p-p
–6–
REV. A
AD8010
5
G = +1
RL = 18.75V
VO = 0.2V p-p
0.2
3
0.1
2
0.05
1
VOLTS
VOLTS
0.15
G = +1
RL = 18.75V
VO = 4V p-p
4
0
–0.05
0
–1
–0.1
–2
–0.15
–3
–4
–0.2
50mV
20ns
1V
20ns
–5
Figure 21. Small-Signal Pulse Response; G = +1
G = +2, –1
RL = 18.75V
VO = 0.2V p-p
0.2
0.15
3
0.1
2
0.05
1
0
–1
–0.1
–2
–0.15
–3
–4
1V
20ns
50mV
20ns
Figure 25. Large-Signal Pulse Response; G = +2, –1
Figure 22. Small-Signal Pulse Response; G = +2, –1
1000
INPUT CURRENT NOISE – PA/ Hz
100
INPUT VOLTAGE NOISE – nV/ Hz
0
–0.05
–0.2
100
10
1
10
100
1k
10k
100k
1M
INVERTING CURRENT
10
NONINVERTING CURRENT
1
10
10M
100
1k
10k
100k
1M
10M
FREQUENCY – Hz
FREQUENCY – Hz
Figure 26. Input Current Noise vs. Frequency
Figure 23. Input Voltage Noise vs. Frequency
REV. A
G = +2, –1
RL = 18.75V
VO = 4V p-p
4
VOLTS
VOLTS
Figure 24. Large-Signal Pulse Response; G = +1
–7–
AD8010
Driving Capacitance Loads
G = +6
RF = 604V
RL = 18.75V
VOLTS
INPUT
The AD8010 was designed primarily to drive nonreactive loads.
If driving loads with a capacitive component is desired, best
frequency response is obtained by the addition of a small series
resistance as shown in Figure 28. The inset figure shows the
optimum value for RSERIES vs. capacitive load. It is worth noting
that the frequency response of the circuit when driving large
capacitive loads will be dominated by the passive roll-off of
RSERIES and CL.
OUTPUT
0
INPUT (500mV/DIV)
OUTPUT (1V/DIV)
LAYOUT CONSIDERATIONS
100ns
The specified high speed performance of the AD8010 requires
careful attention to board layout and component selection.
Proper RF design techniques and low-pass parasitic component
selection are necessary.
Figure 27. Overdrive Recovery; G = +6
OVERDRIVE RECOVERY
The PCB should have a ground plane covering all unused portions of the component side of the board to provide low impedance path. The ground plane should be removed from the area
near the input pins to reduce the parasitic capacitance.
Overdrive of an amplifier occurs when the output and/or input
range are exceeded. The amplifier must recover from this overdrive condition. As shown in Figure 27, the AD8010 recovers
within 35 ns from negative overdrive and within 75 ns from
positive overdrive.
–VS
+VS
THEORY OF OPERATION
The AD8010 is a current feedback amplifier optimized for high
current output while maintaining excellent performance with
respect to flatness, distortion and differential gain/phase. As a
video distribution amplifier, the AD8010 will drive up to 12
parallel video loads (12.5 Ω) from a single output with 0.04%
differential gain and 0.04° differential phase errors. This means
that, unlike designs with one driver per output, any output is a
true reflection of the signal on all other outputs.
FB
C1
+
150V
VIN
RBT
RT
RF
The high output current capability of the AD8010 also make it
useful in xDSL applications. The AD8010 can drive a 12.5 Ω
single-ended or 25 Ω differential load with low harmonic distortion. This makes it useful in designs that utilize a step-up transformer to drive a twisted-pair transmission line.
The standard noninverting closed-loop configuration with the
recommended power supply bypassing technique is shown in
Figure 29. Ferrite beads (Amidon Associates, Torrance CA,
Part Number 43101) are used to suppress high frequency power
supply energy on the DUT supply lines at the DUT. C1 and C2
each represent the parallel combination of a 47 µF (16 V) tantalum electrolytic capacitor, a 10 µF (10 V) tantalum electrolytic
capacitor and a 0.1 µF ceramic chip capacitor. Connect C1
from the +VS pin to the –VS pin. Connect C2 from the –VS pin
to signal ground.
GAIN AS SHOWN
VO = 0.2V p-p
w/# 30% OVERSHOOT
CAPACITIVE LOAD – pF
RF
RG
RS
150V
VIN
10
50V
VOUT
CL
The feedback resistor should be located close to the inverting
input pin in order to keep the parasitic capacitance at this node
to a minimum. Parasitic capacitances of less than 1 pF at the
inverting input can significantly affect high speed performance.
G = +2
Stripline design techniques should be used for long traces
(greater than about 3 cm). These should be designed with a
characteristic impedance (ZO) of 50 Ω or 75 Ω and be properly
terminated at each end.
G = +1
1
0
5
10
RL
Figure 29. Standard Noninverting Closed-Loop Configuration with Recommended Bypassing Technique
200
G = +5
C2
+
RG
To achieve these levels of performance special precautions with
respect to supply bypassing are recommended (Figure 29). This
configuration minimizes the contribution from high frequency
supply rejection to differential gain and phase errors as well as
reducing distortion due to harmonic energy in the power
supplies.
100
ZO
AD8010
15
20
RS – V
Figure 28. Capacitive Load Drive vs. Series Resistor for
Various Gains
–8–
REV. A
AD8010
The bypassing scheme that is used for the AD8010 requires
special attention. It was found that the conventional technique
of bypassing each power pin individually to ground can have an
adverse effect on the differential phase error of the circuit. The
cause of this is attributed to the fact that there is an internal
compensation capacitor in the AD8010 that is referenced to the
negative supply.
APPLICATIONS
Video Distribution Amplifier
The AD8010 is optimized for the specific function of providing
excellent video performance when driving multiple video loads
in parallel. Significant power is saved and heat sinking is greatly
simplified because of the ability of the AD8010 to obtain this
performance when running on a ± 5 V supply. However, due to
the high currents that flow when driving many parallel video
loads, special layout and bypassing techniques are required to
assure optimal performance.
The recommended technique is to connect parallel bypass
capacitors from the positive supply to the negative supply and
then to bypass the negative supply to ground. For high frequency bypassing, 0.1 µF ceramic capacitors are recommended.
These should be placed within a few millimeters of the power
pins and should preferably be chip type capacitors.
When designing a video distribution amplifier with the AD8010, it
is very important to keep in mind where the high (ac) currents
will flow. These paths include the power supply pins of the chip
along with the bypass capacitors and the return path for these
capacitors, the output circuits and the return path of the output
current from the loads.
The high currents that can potentially flow through the power
supply pins require large bypassing capacitors. These should be
low inductance tantalum types and at least 47 µF. The ground
side of the capacitor that bypasses the negative supply should be
brought to a single point ground that is the common for the
returns of the outputs.
In general, any loops that are formed by any of the above paths
should be made as small as possible. Large loops are both generators and receivers of magnetic fields and can cause undesired
coupling of signals that lowers the performance of the amplifier.
Effects that have not been seen before in other op amp circuits
might arise because of the high currents. Most op amp circuits
output, at most, tens of milliamps and do not require extremely
tight video specifications, while a video distribution amplifier
can output hundreds of milliamps and require extremely low
differential gain and phase errors.
Figure 30 shows a circuit for making an N-channel video distribution amplifier. As a practical matter, the AD8010 can readily
drive eight standard 150 Ω video loads. When driving up to 12
video loads, there is minimal degradation in video performance.
Another important consideration when driving multiple cables
is the high frequency isolation between the outputs of the
cables. Due to its low output impedance, the AD8010 achieves
better than 46 dB of output-to-output isolation at 5 MHz driving back terminated 75 Ω cables.
+5V
FB
499V
499V
75V
AD8010
150V
VIN
C1
75V
RL1
75V
RL2
75V
RLN
75V
75V
C2
FB
–5V
75V
Figure 30. An N-Channel Video Distribution Amplifier Using An AD8010.
NOTE: Please see Figure 29 for Recommended Bypassing Technique.
REV. A
–9–
AD8010
Differential Line Driver
In the circuit in Figure 31 the A device is configured as a gainof-two follower, while the B device is a gain-of-two inverter.
These will produce a differential output signal whose maximum
value is twice the peak-to-peak value of the maximum output of
one device. For this circuit a 12 V peak-to-peak output can be
obtained.
Twisted pair transmission lines are more often being used for
high frequency analog and digital signals. Over long distances,
however, the attenuation characteristics of these lines can
degrade the performance of the transmission system. To compensate for this, larger signals are transmitted, which after the
attenuation, will still have useful signal strength.
The high output current of two AD8010s can be used along
with a transformer to create a high power differential line driver.
The differential configuration effectively doubles the output
swing, while the step-up transformer further increases the output voltage.
The op amps drive a 1:2 step-up transformer that drives a
100 Ω transmission line. Since the impedance reflected back to
the primary varies as the square of the turns ratio, it will appear
as 25 Ω at the primary. This source terminating resistor is split
as a 12.4 Ω resistor at the output of each device.
The circuit shown is capable of delivering 12 V p-p to the line
and operates with a –3 dB bandwidth of 40 MHz. The peak
current output of either op amp is 100 mA.
499V
499V
AD8010
12.4V
+6
150V
VIN
402V
806V
AD8010
100V
12.4V
1:2 –6
150V
Figure 31. High Output Differential Line Driver Using Two AD8010s.
NOTE: Please see Figure 29 for Recommended Bypassing Technique.
–10–
REV. A
AD8010
Closed-Loop Gain and Bandwidth
The AD8010 is a current feedback amplifier optimized for use
in high performance video and data acquisition applications.
Since it uses a current feedback architecture, its closed-loop
–3 dB bandwidth is dependent on the magnitude of the feedback resistor. The desired closed-loop bandwidth and gain are
obtained by varying the feedback resistor (RF) to set the bandwidth, and varying the gain resistor (RG) to set the desired gain.
The characteristic curves and specifications for this data sheet
reflect the performance of the AD8010 using the values of RF
noted at the top of the specifications table. If a greater –3 dB bandwidth and/or slew rate is required (at the expense of video performance), Table I provides the recommended resistor values.
Figure 32 shows the test circuit and conditions used to produce
Table I.
Effect of Feedback Resistor Tolerance on Gain Flatness
Because of the relationship between the 3 dB bandwidth and the
feedback resistor, the fine scale gain flatness will, to some extent, vary with feedback resistor tolerance. It is therefore recommended that resistors with a 1% tolerance be used if it is desired
to maintain flatness over a wide range of production lots. In
addition, resistors of different construction have different associated parasitic capacitance and inductance. Metal-film resistors
were used for the bulk of the characterization for this data sheet.
It is possible that values other than those indicated will be optimal for other resistor types.
Quality of Coaxial Cable
Optimum flatness when driving a coax cable is possible only
when the driven cable is terminated at each end with a resistor
matching its characteristic impedance. If the coax was ideal,
then the resulting flatness would not be affected by the length of
the cable. While outstanding results can be achieved using inexpensive cables, it should be noted that some variation in flatness
due to varying cable lengths may be experienced.
Table I. –3 dB Bandwidth and Slew Rate vs. Closed-Loop
Gain and Resistor Values
Package: N-8
Closed-Loop
Gain
RF (⍀)
RG (⍀)
–3 dB BW
(MHz)
Slew Rate
(V/␮s)
+1
+2
+5
+10
453
374
348
562
374
86.6
61.9
285
255
200
120
900
900
800
550
Closed-Loop
Gain
RF (⍀)
RG (⍀)
–3 dB BW
(MHz)
Slew Rate
(V/␮s)
+1
+2
+5
+10
412
392
392
604
392
97.6
66.5
245
220
160
95
900
900
800
550
Closed-Loop
Gain
RF (⍀)
RG (⍀)
–3 dB BW
(MHz)
Slew Rate
(V/␮s)
+1
+2
+5
+10
392
374
348
499
374
86.6
54.9
345
305
220
135
950
1000
1000
650
∞
Package: R-16
∞
Package: SO-8
VIN
∞
150V
VOUT
50V
RF
18.75V
RG
Figure 32. Test Circuit for Table I
NOTES
1
VO = 0.2 V p-p for –3 dB Bandwidth.
2
VO = 2 V p-p for Slew Rate.
3
Bypassing per Figure 29.
REV. A
–11–
AD8010
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
C3208a–0–10/98
8-Lead Plastic Mini-DIP
(N-8)
0.430 (10.92)
0.348 (8.84)
8
5
0.280 (7.11)
0.240 (6.10)
1
4
0.060 (1.52)
0.015 (0.38)
PIN 1
0.210 (5.33)
MAX
0.325 (8.25)
0.300 (7.62)
0.195 (4.95)
0.115 (2.93)
0.130
(3.30)
MIN
0.160 (4.06)
0.115 (2.93)
0.022 (0.558) 0.100 0.070 (1.77)
0.014 (0.356) (2.54) 0.045 (1.15)
BSC
0.015 (0.381)
0.008 (0.204)
SEATING
PLANE
8-Lead SOIC
(SO-8)
0.1968 (5.00)
0.1890 (4.80)
0.1574 (4.00)
0.1497 (3.80)
PIN 1
0.0098 (0.25)
0.0040 (0.10)
8
5
1
4
0.2440 (6.20)
0.2284 (5.80)
0.0688 (1.75)
0.0532 (1.35)
0.0196 (0.50)
3 458
0.0099 (0.25)
0.0500 0.0192 (0.49)
SEATING (1.27)
0.0098 (0.25)
PLANE BSC 0.0138 (0.35) 0.0075 (0.19)
88
08
0.0500 (1.27)
0.0160 (0.41)
16-Lead Wide Body SOIC
(R-16)
1
8
PIN 1
0.0118 (0.30)
0.0040 (0.10)
0.0500
(1.27)
BSC
0.1043 (2.65)
0.0926 (2.35)
0.0291 (0.74)
x 45°
0.0098 (0.25)
8°
0.0192 (0.49)
0°
SEATING 0.0125 (0.32)
0.0138 (0.35) PLANE
0.0091 (0.23)
–12–
PRINTED IN U.S.A.
9
0.4193 (10.65)
0.3937 (10.00)
16
0.2992 (7.60)
0.2914 (7.40)
0.4133 (10.50)
0.3977 (10.00)
0.0500 (1.27)
0.0157 (0.40)
REV. A