Intersil ISL6328CRZ Dual pwm controller for powering amd svi split-plane processor Datasheet

Dual PWM Controller For Powering AMD SVI Split-Plane
Processors
ISL6328
Features
The ISL6328 dual PWM controller delivers high efficiency and
tight regulation from two synchronous buck DC/DC converters.
The ISL6328 supports power control of AMD processors, which
operate from a serial VID interface (SVI). The dual output
ISL6328 features a multi-phase controller to support the Core
voltage (VDD) and a single phase controller to power the
Northbridge (VDDNB).
• Processor Core Voltage Via Integrated Multi-Phase Power Conversion
• Configuration Flexibility
- 1 or 2-Phase Operation with Internal Drivers
- 3 or 4-Phase Operation with External PWM Drivers
A precision core voltage regulation system is provided by a
one-to-four-phase PWM voltage regulator (VR) controller. The
integration of two power MOSFET drivers adds flexibility in layout
and reduces the number of external components in the
multi-phase section. A single phase PWM controller with
integrated driver provides a second precision voltage regulation
system for the Northbridge portion of the processor. This
monolithic, dual controller with an integrated driver solution
provides a cost and space saving power management solution.
• PSI_L Support
- Phase Shedding for Improved Efficiency at Light Load
- Diode Emulation in PSI mode
- Gate Voltage Optimization
• Precision Core Voltage Regulation
- Differential Remote Voltage Sensing
- ±0.6% System Accuracy Over-Temperature
• Optimal Processor Core Voltage Transient Response
- Adaptive Phase Alignment (APA)
- Active Pulse Positioning Modulation
For applications that benefit from load line programming to reduce
bulk output capacitors, the ISL6328 features temperature
compensated output voltage droop. The multi-phase portion also
includes advanced control loop features for optimal transient
response to load application and removal. One of these features
is highly accurate, fully differential, continuous DCR current
sensing for load line programming and channel current balance.
Dual edge modulation is another unique feature, allowing for
quicker initial response to high di/dt load transients.
• Fully Differential, Continuous DCR Current Sensing
- Accurate Load Line Programming
- Precision Channel Current Balancing
- Temperature Compensated
The ISL6328 supports Power Savings Mode by dropping the
number of phases when the PSI_L bit is set.
• Multi-tiered Overvoltage Protection
• Serial VID interface Handles up to 3.4MHz Clock Rates
• Two Level Overcurrent Protection Allows for High Current
Throttling (IDD_SPIKE)
• Selectable Switching Frequency up to 1MHz
• Simultaneous Digital Soft-Start of Both Outputs
June 7, 2011
FN7621.1
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas Inc. 2011. All Rights Reserved
Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries.
All other trademarks mentioned are the property of their respective owners.
ISL6328
Integrated Driver Block Diagram
Channels 1 and 2 Gate Drive
PVCC
GVOT
BOOTn
UGATEn
PWM
20kΩ
SOFT-START
AND
FAULT LOGIC
GATE
CONTROL
LOGIC
SHOOTTHROUGH
PROTECTION
PHASEn
10kΩ
LGATEn
Northbridge Gate Drive
PVCC
BOOT_NB
UGATE_NB
PWM
20kΩ
SOFT-START
AND
FAULT LOGIC
GATE
CONTROL
LOGIC
SHOOTTHROUGH
PROTECTION
PHASE_NB
10kΩ
LGATE_NB
2
FN7621.1
June 7, 2011
ISL6328
Controller Block Diagram
COMP_NB
FB_NB
RGND
VSEN_NB
BOOT_NB
E/A
CURRENT
SENSE
ISEN_NB-
DRPCTRL
∑
NB_REF
ISEN_NB+
UV
LOGIC
OV
LOGIC
MOSFET
DRIVER
RAMP
DROOP
CONTROL
EN_12V
OFFSET
COMP
CH3_OFF
PSI
SVC
SVD
OCP
VCC
GVOT
SOFT-START
+
FAULT LOGIC
E/A
∑
+
BOOT1
NB_REF
SVI
SLAVE
BUS
MOSFET
DRIVER
LOAD APPLY
TRANSIENT
ENHANCEMENT
LGATE1
CLOCK AND
TRIANGLE WAVE
GENERATOR
UV
LOGIC
APA
FS
OC
PWM1
I_TRIP
DUAL
OCP
∑
BOOT2
8
N
PWM2
MOSFET
DRIVER
∑
PWM3
TCOMP1
TCOMP2
ISEN1+
ISEN1ISEN2+
ISEN2ISEN3+
ISEN3-
ISEN4-
UGATE2
PHASE2
LGATE2
∑
TEMPERATURE
COMPENSATION
PH3/PH4
POR
PWM4
I_TC_IN
∑
CH1
CURRENT
SENSE
EN_12V
CHANNEL
DETECT
CH2
CURRENT
SENSE
CHANNEL
CURRENT
BALANCE
I_AVG
CH3
CURRENT
SENSE
CH4
CURRENT
SENSE
∑
ISEN2ISEN3ISEN4-
1
N
I_TC_IN
ISEN3ISEN4+
UGATE1
PHASE1
OV
LOGIC
VSEN
APA
EN
AND
RGND
PWROK
ENABLE
LOGIC
POWER-ON
RESET
FB
RGND
PVCC
LDO
NB
FAULT
LOGIC
FB_PSI
PHASE_NB
LGATE_NB
VDDPWRGD
OFS
UGATE_NB
1
8
PWM3
SIGNAL
LOGIC
PWM4
SIGNAL
LOGIC
PWM3
PWM4
ISEN4GND
3
FN7621.1
June 7, 2011
ISL6328
Typical Application
VCC
+5V
VCC RSVD
TCOMP1
CS1-
ISEN1-
CS1+
ISEN1+
TCOMP2
CS2-
ISEN2-
PWM3
PWM4
CS2+
ISEN2+
CS3-
ISEN3-
PVCC
ISEN3+
GVOT
CS3+
PWM3
PWM4
+12V
ISL6328
CS4-
ISEN4-
CS4+
ISEN4+
CS_NB-
ISEN_NB-
CS_NB+
ISEN_NB+
+12V
BOOT1
UGATE1
PHASE1
LGATE1
CS1CS1+
BOOT2
DRPCTRL
CS3CS3+
+12V
OFS
LGATE2
OCP
+12V
SVC
SVD
PWROK
VDDPWRGD
EN
APA
LGATE1
PVCC
VCC
BOOT2
+12V
GND
UGATE2
PHASE2
FS
ENABLE
ISL6614
BOOT1 PWM1
PWM3
PWM2
PWM4
UGATE1
PHASE1
+12V
+12V
UGATE2
PHASE2
CS2CS2+
CS4CS4+
PGND
LGATE2
VSEN
RGND
CORE_FB
BOOT_NB
+12V
CPU
UGATE_NB
PHASE_NB
LGATE_NB
CORE
CS_NBCS_NB+
NORTHBRIDGE
VSEN_NB
CORE_FB
FB_PSI
FB
COMP
4
FB_NB
COMP_NB
GND
FN7621.1
June 7, 2011
ISL6328
Pin Configuration
ISEN_NB+
ISEN_NB-
ISEN4+
ISEN4-
ISEN3+
ISEN3-
PVCC
LGATE_NB
BOOT_NB
UGATE_NB
PHASE_NB
PWM3
ISL6328
(48 LD QFN)
TOP VIEW
48
47
46
45
44
43
42
41
40
39
38
37
COMP_NB
1
36
PWM4
FB_NB
2
35
PWROK
VSEN_NB
3
34
VDDPWRGD
DRPCTRL
4
33
PHASE1
SVC
5
32
UGATE1
SVD
6
31
BOOT1
VCC
7
30
LGATE1
RSVD
8
29
GVOT
OFS
9
28
LGATE2
OCP
10
27
BOOT2
TCOMP1
11
26
UGATE2
TCOMP2
12
25
EN
18
19
VSEN
FB_PSI
FB
COMP
FS
APA
20
21
22
23
24
PHASE2
17
ISEN2-
16
ISEN2+
15
ISEN1-
14
ISEN1+
13
RGND
49
GND
Functional Pin Descriptions
PIN NAME
PIN NUMBER
COMP_NB
1
Output of the internal error amplifier for the Northbridge regulator.
FB_NB
2
Inverting input to the internal error amplifier for the Northbridge regulator.
VSEN_NB
3
Non-inverting input to the Northbridge regulator precision differential remote-sense amplifier. This pin
should be connected to the remote Northbridge sense pin of the processor, VDDNB_SENSE.
DRPCTRL
4
Droop Control for Core and Northbridge. This pin is used to set up one of four user programmable selections
via a resistor: Core Droop On and Northbridge Droop On; Core Droop Off and Northbridge Droop On, Core
Droop On and Northbridge Droop Off; Core Droop Off and Northbridge Droop Off.
If the resistor is tied to ground, the number of active phases in PSI mode is 1. If the resistor is tied to
VCC, the number of active phases in PSI mode is 2.
SVC
5
Serial VID clock input from the AMD processor.
SVD
6
Serial VID data bi-directional signal to and from the master device on AMD processor.
VCC
7
VCC is the bias supply for the ICs small-signal circuitry. Connect this pin to a +5V supply and decouple
using a quality 0.1µF ceramic capacitor.
RSVD
8
RESERVED. Connect this pin directly to the VCC pin.
OFS
9
The OFS pin provides a means to program a DC current for generating an offset voltage across the resistor
between FB and VSEN. The offset current is generated via an external resistor and precision internal
voltage references. The polarity of the offset is selected by connecting the resistor to GND or VCC. For no
offset, the OFS pin should be left unconnected.
5
DESCRIPTION
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ISL6328
Functional Pin Descriptions (Continued)
PIN NAME
PIN NUMBER
DESCRIPTION
OCP
10
A capacitor from this pin to ground determines the time that the regulator is allowed to service a load
current spike that exceeds the internal OCP trip point.
TCOMP1, TCOMP2
11, 12
RGND
13
Inverting input to the Core and Northbridge regulator precision differential remote-sense amplifiers. This
pin should be connected to the remote ground sense pin of the processor core, VSS_SENSE.
VSEN
14
Non-inverting input to the Core regulator precision differential remote-sense amplifier. This pin should be
connected to the remote Core sense pin of the processor, VDD_SENSE.
FB_PSI
15
In PSI mode this pin is internally shorted to the FB pin to augment the feedback compensation network
for the lower phase count.
FB
16
Inverting input to the internal error amplifier for the Core regulator.
COMP
17
Output of the internal error amplifier for the Core regulator.
FS
18
This is a dual function pin. A resistor, placed from FS to either Ground or VCC sets the switching
frequency of both controllers. Refer to Equation 1 for proper resistor calculation.
These two pins are used to compensate the inductor current sensing for fluctuations due to
temperature.
R T = 10
[ 10.61 – 1.035 log ( f s ) ]
(EQ. 1)
This pin also controls the SVID high and low trip thresholds.
APA
19
Allows for programming of the Auto Phase Alignment threshold. A resistor in parallel with a capacitor
to ground is used to set this threshold.
ISENn+, ISENn-,
ISEN_NB+, ISEN_NB-
20, 21, 22,
23, 43, 44,
45, 46, 47,
48
These pins are used for differentially sensing the corresponding channel output currents. The sensed
currents are used for channel balancing, protection, and core load line regulation.
Connect ISEN- to the node between the RC sense elements surrounding the inductor of the respective
channel. Tie the ISEN+ pin to the other end of the sense capacitor through a resistor, RISEN. The voltage
across the sense capacitor is proportional to the inductor current. The sense current, therefore, is
proportional to the inductor current and scaled by the DCR of the inductor and RISEN.
PHASE1,
PHASE2
33,
24
Connect these pins to the sources of the corresponding upper MOSFETs. These pins are the return path
for the upper MOSFET drives.
GND
49
Bias and reference ground for the IC. The GND connection for the ISL6328 is made with three pins and
through the thermal pad on the bottom of the package.
EN
25
This pin is a threshold-sensitive (approximately 0.85V) system enable input for the controller. Held low, this
pin disables both CORE and NB controller operation. Pulled high, the pin enables both controllers for
operation.
A second function of this pin is to provide driver bias monitor for external drivers. A resistor divider with the
center tap connected to this pin from the drive bias supply prevents enabling the controller before
insufficient bias is provided to external driver. The resistors should be selected such that when the POR-trip
point of the external driver is reached, the voltage at this pin meets the above mentioned threshold level.
UGATE1, UGATE2
32, 26
Connect this pin to the corresponding upper MOSFET gate. This pin provides the PWM-controlled gate
drive for the upper MOSFET and is monitored for shoot-through prevention purposes.
BOOT1, BOOT2
31, 27
This pin provides the bias voltage for the corresponding upper MOSFET drive. Connect this pin to
appropriately-chosen external bootstrap capacitor. The internal bootstrap diode connected to the PVCC
pin provides the necessary bootstrap charge.
LGATE1, LGATE2
30, 28
Connect this pin to the corresponding MOSFET’s gate. This pin provides the PWM-controlled gate drive
for the lower MOSFET. This pin is also monitored by the adaptive shoot-through protection circuitry to
determine when the lower MOSFET has turned off.
GVOT
29
The power supply pin for the multi-phase internal MOSFET drivers. In normal operation, this pin is
shorted to the PVCC pin. While in PSI mode, this pin is tied to the output of the internal LDO for Gate
Drive Voltage Optimization. Decouple this pin with a quality 2.2μF ceramic capacitor.
VDDPWRGD
34
During normal operation this pin indicates whether both output voltages are within specified overvoltage
and undervoltage limits. If either output voltage exceeds these limits or a reset event occurs (such as an
overcurrent event), the pin is pulled low. This pin is always low prior to the end of soft-start.
6
FN7621.1
June 7, 2011
ISL6328
Functional Pin Descriptions (Continued)
PIN NAME
PIN NUMBER
DESCRIPTION
PWROK
35
System wide Power Good input signal. If this pin is low, the two SVI bits are decoded to determine the
“metal VID”. When pin is high, the SVI is actively running its protocol.
PWM3, PWM4
37, 36
Pulse-width modulation outputs. Connect these pins to the PWM input pins of an Intersil driver IC if
3- or 4-phase operation is desired. Connect the ISEN- pins of the channels not desired to +5V to disable
them. Channels must be disabled in decremental order.
PHASE_NB
38
Connect this pin to the source of the corresponding upper MOSFET. This pin is the return path for the
upper MOSFET drive. This pin is used to monitor the voltage drop across the upper MOSFET for
overcurrent protection.
UGATE_NB
39
Connect this pin to the corresponding upper MOSFET gate. This pin provides the PWM-controlled gate
drive for the upper MOSFET and is monitored for shoot-through prevention purposes.
BOOT_NB
40
This pin provides the bias voltage for the corresponding upper MOSFET drive. Connect this pin to
appropriately-chosen external bootstrap capacitor. The internal bootstrap diode connected to the PVCC
pin provides the necessary bootstrap charge.
LGATE_NB
41
Connect this pin to the corresponding MOSFET’s gate. This pin provides the PWM-controlled gate drive
for the lower MOSFET. This pin is also monitored by the adaptive shoot-through protection circuitry to
determine when the lower MOSFET has turned off.
PVCC
42
The power supply pin for the internal MOSFET drivers. Connect this pin to +12V. This pin is the input to
the internal LDO for GVOT. Decouple this pin with a quality 1.0µF ceramic capacitor.
Ordering Information
PART NUMBER
(Notes 1, 2, 3)
PART
MARKING
TEMP. RANGE
(°C)
PACKAGE
(Pb-free)
PKG.
DWG. #
ISL6328CRZ
ISL6328 CRZ
0 to +70
48 Ld 6x6 QFN
L48.6x6B
ISL6328IRZ
ISL6328 IRZ
-40 to +85
48 Ld 6x6 QFN
L48.6x6B
NOTES:
1. Please refer to TB347 for details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate
plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pb-free products are
MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page for ISL6328. For more information on MSL please see techbrief TB363.
7
FN7621.1
June 7, 2011
ISL6328
Table of Contents
Integrated Driver Block Diagram . . . . . . . . . . . . . . . . . . . . . 2
Controller Block Diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Typical Application. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Functional Pin Descriptions . . . . . . . . . . . . . . . . . . . . . . . . . 5
Absolute Maximum Ratings . . . . . . . . . . . . . . . . . . . . . . . . . 9
Thermal Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Recommended Operating Conditions. . . . . . . . . . . . . . . . . 9
Timing Diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Multi-phase Power Conversion . . . . . . . . . . . . . . . . . . . 12
Interleaving . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Active Pulse Positioning Modulated
PWM Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Adaptive Phase Alignment (APA) . . . . . . . . . . . . . . . . . 13
PWM Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Continuous Current Sampling . . . . . . . . . . . . . . . . . . . 14
Temperature Compensated Current Sensing . . . . . . . 15
Channel-Current Balance . . . . . . . . . . . . . . . . . . . . . . . 15
Serial VID Interface (SVI) . . . . . . . . . . . . . . . . . . . . . . . . .
Pre-PWROK METAL VID . . . . . . . . . . . . . . . . . . . . . . . . .
SVI Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Power Savings Mode: PSI_L . . . . . . . . . . . . . . . . . . . . .
Voltage Regulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Load-Line (Droop) Regulation . . . . . . . . . . . . . . . . . . .
Droop Control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Output-Voltage Offset Programming . . . . . . . . . . . . . .
Dynamic VID . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Advanced Adaptive Zero Shoot-Through
Deadtime Control (Patent Pending) . . . . . . . . . . . . .
15
16
16
17
18
18
18
19
19
Soft-Start Output Voltage Targets . . . . . . . . . . . . . . . . .20
Soft-Start. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Pre-Biased Soft-Start. . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Fault Monitoring and Protection . . . . . . . . . . . . . . . . . . . . 21
Power-Good Signal . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Overvoltage Protection . . . . . . . . . . . . . . . . . . . . . . . . . .21
Pre-POR Overvoltage Protection . . . . . . . . . . . . . . . . . .21
Undervoltage Detection . . . . . . . . . . . . . . . . . . . . . . . . 22
Open Sense Line Protection . . . . . . . . . . . . . . . . . . . . .22
Overcurrent Protection . . . . . . . . . . . . . . . . . . . . . . . . . .22
Individual Channel Overcurrent Limiting . . . . . . . . . . 23
General Design Guide . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Power Stages . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .23
Internal Bootstrap Device . . . . . . . . . . . . . . . . . . . . . . . 24
Gate Drive Voltage Versatility. . . . . . . . . . . . . . . . . . . . 24
Package Power Dissipation . . . . . . . . . . . . . . . . . . . . . 24
Inductor DCR Current Sensing Component
Fine Tuning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Loadline Regulation Resistor . . . . . . . . . . . . . . . . . . . . 26
Compensation With Loadline Regulation. . . . . . . . . . 26
Compensation Without Loadline Regulation . . . . . . . 26
Output Filter Design. . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Switching Frequency . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Input Capacitor Selection . . . . . . . . . . . . . . . . . . . . . . .28
Layout Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Routing UGATE, LGATE, and PHASE Traces . . . . . . . . .30
Current Sense Component Placement and
Trace Routing. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
Thermal Management . . . . . . . . . . . . . . . . . . . . . . . . . 30
19
Revision History . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Initialization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Power-On Reset . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Enable Comparator . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Phase Detection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Products . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Package Outline Drawing . . . . . . . . . . . . . . . . . . . . . . . . . 33
8
FN7621.1
June 7, 2011
ISL6328
Absolute Maximum Ratings
Thermal Information
Supply Voltage (VCC) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6V
Supply Voltage (PVCC) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +15V
Absolute Boot Voltage (VBOOT) . . . . . . . . . . . . . . . GND - 0.3V to GND + 36V
Phase Voltage (VPHASE). . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to 24V
(PVCC = 12V) GND - 8V (<400ns, 20µJ) to 31V (<200ns, 20µJ,
VBOOT-PHASE = 5V)
Upper Gate Voltage (VUGATE). . . . . . . . . . . . VPHASE - 0.3V to VBOOT + 0.3V
VPHASE - 3.5V (<100ns Pulse Width, 2µJ) to VBOOT + 0.3V
Lower Gate Voltage (VLGATE). . . . . . . . . . . . . . . .GND - 0.3V to PVCC + 0.3V
GND - 5V (<100ns Pulse Width, 2µJ) to PVCC+ 0.3V
Input, Output, or I/O Voltage . . . . . . . . . . . . . . . . . GND - 0.3V to VCC + 0.3V
ESD Rating
Human Body Model (Tested per JESD22-A114F) . . . . . . . . . . . . . . 2000V
Machine Model (Tested per JESD22-A115C) . . . . . . . . . . . . . . . . . 200V
Charge Device Model (Tested per JESD22-C101D) . . . . . . . . . . . . 2000V
Latch Up (Tested per JESD78C; Class II, Level A) . . . . . . . . . . . . . . . 100mA
Thermal Resistance
θJA (°C/W) θJC (°C/W)
QFN Package (Notes 4, 5) . . . . . . . . . . . . . .
27
1
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . .+150°C
Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°C
Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
VCC Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V ±5%
PVCC Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V to 12V ±5%
Ambient Temperature
ISL6328CRZ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to +70°C
ISL6328IRZ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -40°C to +85°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTES:
4. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech
Brief TB379.
5. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
Boldface limits apply over the operating temperature range.
PARAMETER
TEST CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6)
UNITS
BIAS SUPPLIES
Input Bias Supply Current
IVCC; EN = high
26
35
mA
Gate Drive Bias Current - PVCC Pin
IPVCC; EN = high
4
8
mA
VCC POR (Power-On Reset) Threshold
VCC Rising
4.35
4.7
V
VCC Falling
PVCC POR (Power-On Reset) Threshold
PVCC Rising
PVCC Falling
GVOT POR (Power-On Reset) Threshold
3.6
3.85
4.45
3.6
GVOT Rising
V
4.7
3.95
4.45
V
V
4.7
V
GVOT Falling
3.6
3.95
V
Oscillator Frequency Accuracy, FSW
RT = 100kΩ to Ground
230
250
265
kHz
Oscillator Frequency Accuracy, FSW
RT = 100kΩ to VCC
225
255
287
kHz
Typical Adjustment Range of Switching Frequency
(Note 7)
1.5
MHz
Oscillator Ramp Amplitude, VPP
(Note 7)
PWM MODULATOR
0.150
1.5
V
CONTROL THRESHOLDS
EN Rising Threshold
0.8
0.87
0.92
V
EN Hysteresis
70
100
190
mV
PWROK Input HIGH Threshold
0.9
1.05
1.2
V
PWROK Input LOW Threshold
0.8
0.95
1.1
V
VDDPWRGD
Open drain, VDDPWRGD = 1.24kΩ to 5V
0.5
V
PWM Channel Disable Threshold
VISEN2-, VISEN3-, VISEN4-
4.4
V
9
FN7621.1
June 7, 2011
ISL6328
Electrical Specifications
Boldface limits apply over the operating temperature range. (Continued)
PARAMETER
TEST CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6)
UNITS
PIN ADJUSTABLE OFFSET
OFS Source Current Accuracy (Positive Offset)
0.29
0.32
V
OFS Sink Current Accuracy (Negative Offset)
VCC-1.7
VCC-1.56
V
System Accuracy (VDAC > 1.000V)
-0.6
+0.6
%
System Accuracy (0.600V < VDAC < 1.000V)
-1.0
+1.0
%
System Accuracy (VDAC < 0.600V)
-2.0
+2.0
%
REFERENCE AND DAC
GATE VOLTAGE OPTIMIZATION
GVOT Voltage
PVCC = 12V, IGVOT = 50mA
5.75
V
DC Gain
RL = 10k to ground, (Note 7)
96
dB
Gain-Bandwidth Product (Note 7)
CL = 100pF, RL = 10k to ground, (Note 7)
20
MHz
Slew Rate (Note 7)
CL = 100pF, Load = ±400µA, (Note 7)
8
V/µs
Maximum Output Voltage
Load = 500µA
4.2
V
Minimum Output Voltage
Load = -500µA
1.4
FB_PSI Impedance
Impedance Between FB and FB_PSI
60
ERROR AMPLIFIER
3.8
1.65
V
Ω
SOFT-START RAMP
Soft-Start Slew Rate (SRSS) and VID-On-The-Fly Slew
Rate (SRVOF)
2.5
3
3.8
mV/µs
0.35
V
PWM OUTPUTS
PWM Output Voltage LOW
ILOAD = ±500µA
PWM Output Voltage HIGH
ILOAD = ±500µA
4.0
PWM Tri-State Output Voltage
ILOAD = 100µA
1.5
2
2.7
V
Core Tolerance
4 Phases Active, RISENn = 100Ω, ISENn = 80μA
67
75
85
µA
North Bridge Tolerance
RISEN_NB = 100Ω, IISEN_NB = 100μA
67
75
83
µA
Instant Overcurrent Trip Level - IDROOP
115
141
166
µA
Instant Overcurrent Trip Level - ISEN_AVG
131
150
178
µA
V
DROOP CURRENT
CORE OVERCURRENT PROTECTION
Delayed Overcurrent Trip Level - ISEN_AVG
(Note 7)
100
µA
Overcurrent Trip Level - Individual Channel
(Note 7)
170
µA
OCP Pin Current
Delayed OCP Level Tripped
20
µA
OCP Pin Voltage Trip
Delayed OCP Level Tripped
1.95
2.05
2.15
V
82
141
µA
NORTH BRIDGE OVERCURRENT PROTECTION
Overcurrent Trip Level - ISEN_NB
Overcurrent Trip Level - IDROOP_NB
ISL6328CRZ
80
128
µA
Overcurrent Trip Level - IDROOP_NB
ISL6328IRZ
76
128
µA
10
FN7621.1
June 7, 2011
ISL6328
Electrical Specifications
Boldface limits apply over the operating temperature range. (Continued)
PARAMETER
TEST CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6)
UNITS
POWER GOOD
Overvoltage Threshold
VSEN Rising
VDAC +
220mV
VDAC +
325mV
V
Undervoltage Threshold
VSEN Falling
VDAC 345mV
VDAC 190mV
mV
Power-Good Hysteresis
50
mV
OVERVOLTAGE PROTECTION
OVP Trip Level
VDAC = 1.1V
OVP Lower Gate Release Threshold
1.75
1.79
1.85
0.35
V
V
SWITCHING TIME (Note 7) [See “Timing Diagram” on page 12]
UGATE Rise Time
tRUGATE; VPVCC = 8V, 3nF Load, 10% to 90%
26
ns
LGATE Rise Time
tRLGATE; VPVCC = 8V, 3nF Load, 10% to 90%
18
ns
UGATE Fall Time
tFUGATE; VPVCC = 12V, 3nF Load, 90% to 10%
18
ns
LGATE Fall Time
tFLGATE; VPVCC = 12V, 3nF Load, 90% to 10%
12
ns
UGATE Turn-On Non-overlap
tPDHUGATE; VPVCC = 12V, 3nF Load, Adaptive
10
ns
LGATE Turn-On Non-overlap
tPDHLGATE; VPVCC = 12V, 3nF Load, Adaptive
10
ns
Upper Drive Source Resistance
VPVCC = 12V, 15mA Source Current
2.5
Ω
Upper Drive Sink Resistance
VPVCC = 12V, 15mA Sink Current
2.0
Ω
Lower Drive Source Resistance
VPVCC = 12V, 15mA Source Current
1.6
Ω
Lower Drive Sink Resistance
VPVCC = 12V, 15mA Sink Current
1.1
Ω
GATE DRIVE RESISTANCE
SVI INTERFACE
SVC, SVD Input HIGH (VIH)
FS resistor tied to GND
SVC, SVD Input LOW (VIL)
FS resistor tied to GND
SVC, SVD Input HIGH (VIH)
FS resistor tied to VCC
SVC, SVD Input LOW (VIL)
FS resistor tied to VCC
0.55
V
SVD Low Level Output Voltage
510Ω Resistor to 1.8V
0.4
V
±5
µA
SVC, SVD Leakage (Note 7)
0.85
V
0.45
1.05
V
V
NOTES:
6. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization
and are not production tested.
7. Limits should be considered typical and are not production tested.
11
FN7621.1
June 7, 2011
ISL6328
Timing Diagram
tPDHUGATE
tRUGATE
tFUGATE
UGATE
LGATE
tFLGATE
tRLGATE
tPDHLGATE
Operation
The ISL6328 utilizes a multi-phase architecture to provide a low
cost, space saving power conversion solution for the processor
core voltage. The controller also implements a simple single
phase architecture to provide the Northbridge voltage on the
same chip.
individual channel current. Each PWM pulse is terminated 1/3 of
a cycle after the PWM pulse of the previous phase. The
peak-to-peak current for each phase is about 7A, and the DC
components of the inductor currents combine to feed the load.
IL1 + IL2 + IL3, 7A/DIV
NOTE: All references to VCC refer to the VCC pin or the node that
is tied to the VCC pin. This should not be confused with the bias
voltage as the bias rail can be separated from the VCC node by
an RC filter resistor.
IL3, 7A/DIV
PWM3, 5V/DIV
IL2, 7A/DIV
Multi-phase Power Conversion
Microprocessor load current profiles have changed to the point
that the advantages of multi-phase power conversion are
impossible to ignore. The technical challenges associated with
producing a single-phase converter that is both cost-effective and
thermally viable have forced a change to the cost-saving
approach of multi-phase. The ISL6328 controller helps simplify
implementation by integrating vital functions and requiring
minimal external components. The “Controller Block Diagram”
on page 3 provides a top level view of the multi-phase power
conversion using the ISL6328 controller.
Interleaving
The switching of each channel in a multi-phase converter is
timed to be symmetrically out-of-phase with each of the other
channels. In a 3-phase converter, each channel switches 1/3
cycle after the previous channel and 1/3 cycle before the
following channel. As a result, the three-phase converter has a
combined ripple frequency three times greater than the ripple
frequency of any one phase. In addition, the peak-to-peak
amplitude of the combined inductor currents is reduced in
proportion to the number of phases (Equations 2 and 3).
Increased ripple frequency and lower ripple amplitude mean that
the designer can use less per-channel inductance and lower total
output capacitance for any performance specification.
Figure 1 illustrates the multiplicative effect on output ripple
frequency. The three channel currents (IL1, IL2, and IL3) combine
to form the AC ripple current and the DC load current. The ripple
component has three times the ripple frequency of each
PWM2, 5V/DIV
IL1, 7A/DIV
PWM1, 5V/DIV
1µs/DIV
FIGURE 1. PWM AND INDUCTOR-CURRENT WAVEFORMS FOR
3-PHASE CONVERTER
To understand the reduction of ripple current amplitude in the
multi-phase circuit, examine Equation 2, which represents an
individual channel peak-to-peak inductor current.
( V IN – V OUT ) V OUT
I PP = ---------------------------------------------L fS V
(EQ. 2)
IN
In Equation 2, VIN and VOUT are the input and output voltages
respectively, L is the single-channel inductor value, and fS is the
switching frequency.
The output capacitors conduct the ripple component of the
inductor current. In the case of multi-phase converters, the
capacitor current is the sum of the ripple currents from each of
the individual channels. Compare Equation 2 to the expression
for the peak-to-peak current after the summation of N
symmetrically phase-shifted inductor currents in Equation 3.
Peak-to-peak ripple current decreases by an amount proportional
to the number of channels. Output-voltage ripple is a function of
capacitance, capacitor equivalent series resistance (ESR), and
inductor ripple current. Reducing the inductor ripple current
allows the designer to use fewer or less costly output capacitors.
( V IN – N V OUT ) V OUT
I C, PP = ----------------------------------------------------L fS V
(EQ. 3)
IN
12
FN7621.1
June 7, 2011
ISL6328
Another benefit of interleaving is to reduce input ripple current.
Input capacitance is determined in part by the maximum input
ripple current. Multi-phase topologies can improve overall system
cost and size by lowering input ripple current and allowing the
designer to reduce the cost of input capacitance. The example in
Figure 2 illustrates input currents from a 3-phase converter
combining to reduce the total input ripple current.
The converter depicted in Figure 2 delivers 1.5V to a 36A load from
a 12V input. The RMS input capacitor current is 5.9A. Compare
this to a single-phase converter also stepping down 12V to 1.5V at
36A. The single-phase converter has 11.9ARMS input capacitor
current. The single-phase converter must use an input capacitor
bank with twice the RMS current capacity as the equivalent
three-phase converter.
INPUT-CAPACITOR CURRENT, 10A/DIV
CHANNEL 3
INPUT CURRENT
10A/DIV
CHANNEL 2
INPUT CURRENT
10A/DIV
CHANNEL 1
INPUT CURRENT
10A/DIV
1μs/DIV
FIGURE 2. CHANNEL INPUT CURRENTS AND INPUT-CAPACITOR
RMS CURRENT FOR 3-PHASE CONVERTER
Figures 23, 24 and 25 in the section entitled “Input Capacitor
Selection” on page 28 can be used to determine the inputcapacitor RMS current based on load current, duty cycle, and the
number of channels. They are provided as aids in determining
the optimal input capacitor solution.
Active Pulse Positioning Modulated PWM
Operation
The ISL6328 uses a proprietary Active Pulse Positioning (APP)
modulation scheme to control the internal PWM signals that
command each channel’s driver to turn their upper and lower
MOSFETs on and off. The time interval in which a PWM signal can
occur is generated by an internal clock, whose cycle time is the
inverse of the switching frequency set by the resistor between the
FS pin and ground. The advantage of Intersil’s proprietary Active
Pulse Positioning (APP) modulator is that the PWM signal has
the ability to turn on at any point during this PWM time interval,
and turn off immediately after the PWM signal has transitioned
high. This is important because it allows the controller to quickly
respond to output voltage drops associated with current load
spikes, while avoiding the ring back affects associated with other
modulation schemes.
The PWM output state is driven by the position of the error
amplifier output signal, VCOMP, minus the current correction
signal relative to the proprietary modulator ramp waveform as
13
illustrated in Figure 3. At the beginning of each PWM time
interval, this modified VCOMP signal is compared to the internal
modulator waveform. As long as the modified VCOMP voltage is
lower then the modulator waveform voltage, the PWM signal is
commanded low. The internal MOSFET driver detects the low
state of the PWM signal and turns off the upper MOSFET and
turns on the lower synchronous MOSFET. When the modified
VCOMP voltage crosses the modulator ramp, the PWM output
transitions high, turning off the synchronous MOSFET and turning
on the upper MOSFET. The PWM signal will remain high until the
modified VCOMP voltage crosses the modulator ramp again.
When this occurs the PWM signal will transition low again.
During each PWM time interval, the PWM signal can only
transition high once. Once PWM transitions high, it can not
transition high again until the beginning of the next PWM time
interval. This prevents the occurrence of double PWM pulses
occurring during a single period.
To further improve the transient response, ISL6328 also
implements Intersil’s proprietary Adaptive Phase Alignment
(APA) technique, which turns on all phases together under
transient events with large step current. With both APP and APA
control, ISL6328 can achieve excellent transient performance
and reduce the demand on the output capacitors.
Adaptive Phase Alignment (APA)
When a load is applied, the output will fall in direct relation to the
amount of load being applied and the speed at which the load is
being applied. The ISL6329 monitors the output differentially
through the VSEN pin. If the sensed voltage drops quickly by a
user programmable magnitude (VAPATRIP), all of the upper
MOSFETs will immediately be turned on simultaneously. The trip
level is relative, not absolute, and can be programmed through a
resistor and capacitor tied in parallel from the APA pin to ground.
V APATRIP
R APA = ------------------------1.75μA
(EQ. 4)
A 3900pF, X7R capacitor is required to be placed in parallel to
the APA resistor.
PWM Operation
The timing of each core channel is set by the number of active
channels. Channel detection on the ISEN2-, ISEN3- and ISEN4-,
pins selects 1-channel to 4-channel operation for the ISL6328.
The switching cycle is defined as the time between PWM pulse
termination signals of each channel. The cycle time of the pulse
signal is the inverse of the switching frequency set by the resistor
between the FS pin and ground (or VCC). The PWM signals
command the MOSFET driver to turn on/off the channel
MOSFETs.
The channel firing order for 4-channel operation, the channel
firing order is 1-2-3-4. For 3-channel operation, the channel firing
order is 1-2-3.
Connecting ISEN4- to VCC selects three channel operation. To set
2-channel operation, both ISEN4- and ISEN3- must be tied to
VCC. Similarly, to set single channel operation, ISEN4-, ISEN3and ISEN2- must be tied to VCC.
FN7621.1
June 7, 2011
ISL6328
Continuous Current Sampling
I
VIN
In order to realize proper current-balance, the currents in each
channel are sampled continuously every switching cycle. During
this time, the current-sense amplifier uses the ISEN inputs to
reproduce a signal proportional to the inductor current, IL. This
sensed current, ISEN, is simply a scaled version of the inductor
current.
L
L
MOSFET
LGATE(n)
DCR
+
VL(s)
ISL6328 INTERNAL
CIRCUIT
COUT
VC(s)
C
R2
ISENnISENn+
CISEN
IL
VC(s)
RISEN
-
-
+
+
In
SWITCHING PERIOD
VOUT
-
+
INDUCTOR
-
DRIVER
R1
PWM
n
UGATE(n)
ISEN
FIGURE 4. INDUCTOR DCR CURRENT SENSING
CONFIGURATION
ISEN
multiplied by the ratio of the resistor divider, K. If a resistor
divider is not being used, the value for K is 1.
R1 ⋅ R2
L
-⋅C
------------ = ------------------R1 + R2
DCR
TIME
FIGURE 3. CONTINUOUS CURRENT SAMPLING
The ISL6328 supports Inductor DCR current sensing to continuously
sample each channel’s current for channel-current balance. The
internal circuitry shown in Figure 4 represents Channel n of an
n-Channel converter. This circuitry is repeated for each channel in
the converter, but may not be active depending on how many
channels are operating.
Inductor windings have a characteristic distributed resistance or
DCR (Direct Current Resistance). For simplicity, the inductor DCR
is considered as a separate lumped quantity, as shown in
Figure 4. The channel current ILn, flowing through the inductor,
passes through the DCR. Equation 5 shows the S-domain
equivalent voltage, VL, across the inductor.
(EQ. 5)
V L ( s ) = I L ⋅ ( s ⋅ L + DCR )
n
A simple R-C network across the inductor (R1, R2 and C) extracts
the DCR voltage, as shown in Figure 6. The voltage across the
sense capacitor, VC, can be shown to be proportional to the
channel current ILn, shown in Equation 6.
s⋅L
⎛ ----------- + 1⎞
⎝ DCR
⎠
V C ( s ) = ------------------------------------------------------ ⋅ K ⋅ DCR ⋅ I L
n
⎛ ( R1 ⋅ R2 )
⎞
⎜ s ⋅ ----------------------- ⋅ C + 1⎟
R1 + R2
⎝
⎠
(EQ. 6)
Where:
R2
K = -------------------R2 + R1
(EQ. 7)
If the R-C network components are selected such that the RC
time constant matches the inductor L/DCR time constant (see
Equation 8), then VC is equal to the voltage drop across the DCR
14
(EQ. 8)
The capacitor voltage VC, is then replicated across the effective
internal sense resistor, RISEN. This develops a current through
RISEN which is proportional to the inductor current. This current,
ISEN, is continuously sensed and is then used by the controller for
load-line regulation, channel-current balancing, and overcurrent
detection and limiting. Equation 9 shows that the proportion
between the channel current, IL, and the sensed current, ISEN, is
driven by the value of the effective sense resistance, RISEN, and
the DCR of the inductor.
DCR
I SEN = I L ⋅ --------------R
(EQ. 9)
ISEN
The Northbridge regulator samples the load current in the same
manner as the Core regulator does.
The sampled currents, In, from each active channel are summed
together and divided by the number of active channels. this
current is then gained by 30%. The resulting cycle average
current, IAVG, provides a measure of the total load-current
demand on the converter during each switching cycle. Assuming
that the current in all the active channels is balanced, the
average sensed current can be calculated from Equation 10.
I Load DCR
I AVG = -------------- ⋅ --------------N
R ISEN
(EQ. 10)
In the ISL6328, the average scaled version of the load current,
IAVG, has a 100µA range. At 100µA, the Overcurrent Protection
circuitry is enabled (refer to the “Overvoltage Protection” on
page 21 for detailed information). It is recommended that the
maximum load current correlate to an average sensed current,
IAVG, of 80µA.
FN7621.1
June 7, 2011
ISL6328
A capacitor, CISEN, should be placed between the ISENn+ pin and
ground. The value of the capacitor can be calculated using
Equation 11.
9.5ns
C ISEN = --------------R ISEN
(EQ. 11)
representation of the load current. The load current is given by
Equation 14.
V TCOMP2 R ISEN
I LOAD = 8 ⋅ ------------------------ ⋅ --------------R TC
DCR
(EQ. 14)
Where RISEN is the current sense resistor value and DCR is the
DC resistance of the output inductors. It is recommended that a
high impedance buffer be used when monitoring the voltage on
the TCOMP2 pin.
Channel-Current Balance
ITC_IN
IITC_OUT
TCOMP1
One important benefit of multi-phase operation is the thermal
advantage gained by distributing the dissipated heat over
multiple devices and greater area. By doing this the designer
avoids the complexity of driving parallel MOSFETs and the
expense of using expensive heat sinks and exotic magnetic
materials.
TCOMP2
R1
R2
RNTC
RT(T)
RTC = RT(25°C)
R3
FIGURE 5. AVERAGE CURRENT TEMPERATURE
COMPENSATION
Temperature Compensated Current Sensing
As the load increases, the conduction losses in the output
inductors will cause the temperature of the inductors to rise. As
the inductor temperature rises, the DCR of the output inductors
will also rise. An increase in the DCR will result in an increase in
the sensed current even if the load current remains constant. To
counteract this error in the sensed current, the ISL6328 features
a temperature compensating circuit that utilizes an NTC resistor
to adjust the average current as inductor temperature increases.
Figure 5 shows the implementation of the ISL6328 average
current temperature compensation. The temperature dependent
resistor, RT(T), is a combination of resistors and an NTC which
create an approximate linearization of the NTC resistor (refer to
Equation 12). Resistors R1, R2 and R3 should be adjusted so that
Equation 13 is satisfied.
1
R T ( T ) = R 3 + ----------------------------------------1
1
--------------------------- + ------R 1 + R NTC R 2
(EQ. 12)
DCR ( +25 ° C )
R T ( T ) = ---------------------------------- ⋅ R TC
DCR ( T )
(EQ. 13)
In order to realize the thermal advantage, it is important that
each channel in a multi-phase converter be controlled to carry
about the same amount of current at any load level. To achieve
this, the currents through each channel must be sampled every
switching cycle. The sampled currents, In, from each active
channel are summed together and divided by the number of
active channels. The resulting cycle average current, IAVG,
provides a measure of the total load-current demand on the
converter during each switching cycle. Channel-current balance is
achieved by comparing the sampled current of each channel to
the cycle average current, and making the proper adjustment to
each channel pulse width based on the error. Intersil’s patented
current-balance method is illustrated in Figure 6, with error
correction for Channel 1 represented. In the figure, the cycle
average current, IAVG, is compared with the Channel 1 sample,
I1, to create an error signal IER.
VCOMP
+
MODULATOR
RAMP
WAVEFORM
FILTER
+
PWM1
-
TO GATE
CONTROL
LOGIC
f(s)
I4
IER
IAVG
-
+
1.3
N
Σ
I3
I2
I1
NOTE: CHANNELS 2, 3 AND 4 ARE OPTIONAL.
Where RTC = RT (+25°C)
LOAD MONITORING
The TCOMP2 pin can be utilized to monitor the load current. The
voltage across the RTC resistor is a temperature compensated
FIGURE 6. CHANNEL-1 PWM FUNCTION AND CURRENTBALANCE ADJUSTMENT
The filtered error signal modifies the pulse width commanded by
VCOMP to correct any unbalance and force IER toward zero. The
same method for error signal correction is applied to each active
channel.
Serial VID Interface (SVI)
The on-board Serial VID interface (SVI) circuitry allows the
processor to directly drive the core voltage and Northbridge
voltage reference level within the ISL6328. The SVC and SVD
15
FN7621.1
June 7, 2011
ISL6328
states are decoded with direction from the PWROK input as
described in the sections that follow. The ISL6328 uses a digital
to analog converter (DAC) to generate a reference voltage based
on the decoded SVI value. See Figure 7 for a simple SVI interface
timing diagram.
The Pre-PWROK metal VID code is decoded and latched on the
rising edge of the enable signal. Once enabled, the ISL6328
passes the Pre-PWROK metal VID code on to internal DAC
circuitry. The internal DAC circuitry begins to ramp both the VDD
and VDDNB planes to the decoded Pre-PWROK metal VID output
level. The digital soft-start circuitry actually stair steps the
internal reference to the target gradually over a fixed interval. The
controlled ramp of both output voltage planes reduces in-rush
current during the soft-start interval. At the end of the soft-start
interval, the VDDPWRGD output transitions high indicating both
output planes are within regulation limits.
The upper and lower threshold levels for the SVI inputs are
programmable through the FS pin. The FS resistor can be tied to
either ground or to VCC. This option allows for selection of the
SVID threshold levels.
TABLE 1. SVID THRESHOLDS
FS RESISTOR TIED TO
SVI VIL (V)
SVI VIH (V)
VCC
0.55
1.05
Ground
0.45
0.85
SVI Mode
Once the controller has successfully soft-started and VDDPWRGD
transitions high, the Northbridge SVI interface can assert PWROK
to signal the ISL6328 to prepare for SVI commands. The
controller actively monitors the SVI interface for set VID
commands to move the plane voltages to start-up VID values.
Details of the SVI Bus protocol are provided in the AMD Design
Guide for Voltage Regulator Controllers Accepting Serial VID
Codes specification.
Pre-PWROK METAL VID
At start-up, the controller decodes the SVC and SVD inputs to
determine the Pre-PWROK metal VID setting. Once the POR
circuitry is satisfied, the ISL6328 begins decoding the inputs per
Table 2. Once the EN input exceeds the rising enable threshold,
the ISL6328 saves the Pre-PWROK metal VID value in an
on-board holding register and passes this target to the internal
DAC circuitry.
Once the set VID command is received, the ISL6328 decodes the
information to determine which plane and the VID target
required. See Table 3. The internal DAC circuitry steps the
required output plane voltage to the new VID level. During this
time one or both of the planes could be targeted. In the event the
core voltage plane, VDD, is commanded to power off by serial VID
commands, the VDDPWRGD signal remains asserted. The
Northbridge voltage plane must remain active during this time.
TABLE 2. PRE-PWROK METAL VID CODES
SVC
SVD
OUTPUT VOLTAGE
(V)
0
0
1.1
0
1
1.0
1
0
0.9
1
1
0.8
1
2
3
If the PWROK input is de-asserted, then the controller steps both
VDD and VDDNB planes back to the stored Pre-PWROK metal VID
level in the holding register from initial soft-start. No attempt is
made to read the SVC and SVD inputs during this time. If PWROK
is reasserted, then the on-board SVI interface waits for a set VID
command.
4
5
6
7
8
V_SVI
metal_VID
9
10
11
12
VCC
SVC
SVD
ENABLE
PWROK
metal_VID
VDD and VDDNB
V_SVI
VDDPWRGD
FIGURE 7. SVI INTERFACE TIMING DIAGRAM: TYPICAL PRE-PWROK METAL VID START-UP
16
FN7621.1
June 7, 2011
ISL6328
TABLE 3. SERIAL VID CODES
SVID[6:0]
VOLTAGE (V)
SVID[6:0]
VOLTAGE (V)
SVID[6:0]
VOLTAGE (V)
SVID[6:0]
VOLTAGE (V)
000_0000b
1.5500
010_0000b
1.1500
100_0000b
0.7500
110_0000b
0.3500*
000_0001b
1.5375
010_0001b
1.1375
100_0001b
0.7375
110_0001b
0.3375*
000_0010b
1.5250
010_0010b
1.1250
100_0010b
0.7250
110_0010b
0.3250*
000_0011b
1.5125
010_0011b
1.1125
100_0011b
0.7125
110_0011b
0.3125*
000_0100b
1.5000
010_0100b
1.1000
100_0100b
0.7000
110_0100b
0.3000*
000_0101b
1.4875
010_0101b
1.0875
100_0101b
0.6875
110_0101b
0.2875*
000_0110b
1.4750
010_0110b
1.0750
100_0110b
0.6750
110_0110b
0.2750*
000_0111b
1.4625
010_0111b
1.0625
100_0111b
0.6625
110_0111b
0.2625*
000_1000b
1.4500
010_1000b
1.0500
100_1000b
0.6500
110_1000b
0.2500*
000_1001b
1.4375
010_1001b
1.0375
100_1001b
0.6375
110_1001b
0.2375*
000_1010b
1.4250
010_1010b
1.0250
100_1010b
0.6250
110_1010b
0.2250*
000_1011b
1.4125
010_1011b
1.0125
100_1011b
0.6125
110_1011b
0.2125*
000_1100b
1.4000
010_1100b
1.0000
100_1100b
0.6000
110_1100b
0.2000*
000_1101b
1.3875
010_1101b
0.9875
100_1101b
0.5875
110_1101b
0.1875*
000_1110b
1.3750
010_1110b
0.9750
100_1110b
0.5750
110_1110b
0.1750*
000_1111b
1.3625
010_1111b
0.9625
100_1111b
0.5625
110_1111b
0.1625*
001_0000b
1.3500
011_0000b
0.9500
101_0000b
0.5500
111_0000b
0.1500*
001_0001b
1.3375
011_0001b
0.9375
101_0001b
0.5375
111_0001b
0.1375*
001_0010b
1.3250
011_0010b
0.9250
101_0010b
0.5250
111_0010b
0.1250*
001_0011b
1.3125
011_0011b
0.9125
101_0011b
0.5125
111_0011b
0.1125*
001_0100b
1.3000
011_0100b
0.9000
101_0100b
0.5000
111_0100b
0.1000*
001_0101b
1.2875
011_0101b
0.8875
101_0101b
0.4875*
111_0101b
0.0875*
001_0110b
1.2750
011_0110b
0.8750
101_0110b
0.4750*
111_0110b
0.0750*
001_0111b
1.2625
011_0111b
0.8625
101_0111b
0.4625*
111_0111b
0.0625*
001_1000b
1.2500
011_1000b
0.8500
101_1000b
0.4500*
111_1000b
0.0500*
001_1001b
1.2375
011_1001b
0.8375
101_1001b
0.4375*
111_1001b
0.0375*
001_1010b
1.2250
011_1010b
0.8250
101_1010b
0.4250*
111_1010b
0.0250*
001_1011b
1.2125
011_1011b
0.8125
101_1011b
0.4125*
111_1011b
0.0125*
001_1100b
1.2000
011_1100b
0.8000
101_1100b
0.4000*
111_1100b
OFF
001_1101b
1.1875
011_1101b
0.7875
101_1101b
0.3875*
111_1101b
OFF
001_1110b
1.1750
011_1110b
0.7750
101_1110b
0.3750*
111_1110b
OFF
001_1111b
1.1625
011_1111b
0.7625
101_1111b
0.3625*
111_1111b
OFF
NOTE: * Indicates a VID not required for AMD Family 10h processors.
Power Savings Mode: PSI_L
Bit 7 of the Serial VID codes transmitted as part of the 8-bit data
stream over the SVI bus is allocated for the PSI_L. If bit 7 is 0,
then the processor is at an optimal load for the regulator to enter
power savings mode. If bit 7 is 1, then the regulator should not
be in power savings mode.
With the ISL6328, Power Savings mode is realized through
phase shedding, Gate Voltage Optimization and Diode
Emulation. Once a Serial VID command with Bit 7 set to 0 is
received, the ISL6328 will shed phases in a sequential manner
until then minimum phase count for PSI is reached. The
minimum phase count for PSI can be programmed via the
DRPCTRL pin to be either 1 phase or 2 phases. If the DRPCTRL
resistor is tied to ground, then the minimum phase count in PSI is
17
1 phase. If the DRPCTRL resistor is tied to VCC then the
minimum phase count in PSI is 2 phases. Channels are shed in
reverse sequential order so that the highest numbered channel
that is active will be shed first. When a phase is shed, that phase
will not go into a tri-state mode until that phase would have had
its PWM go HIGH.
When leaving Power Savings Mode, through the reception of a
Serial VID command with Bit 7 set to 1, the ISL6328 will
sequentially turn on phases starting with lowest numbered
inactive channel. When a phase is being reactivated, it will not
leave a tri-state until the PWM of that phase goes HIGH.
If, while in Power Savings Mode, a Serial VID command is
received that forces a VID level change while maintaining Bit 7 at
0, the ISL6328 will first exit the Power Savings Mode state as
FN7621.1
June 7, 2011
ISL6328
described previously. The output voltage will then be stepped up
or down to the appropriate VID level. Finally, the ISL6328 will
then re-enter Power Savings Mode.
While in Power Savings Mode, the ISL6328 implements two
features that effectively enhance the efficiency of the regulator
even more. These features are Diode Emulation and Gate Voltage
Optimization.
upper specification limit, a larger negative spike can be sustained
without crossing the lower limit. By adding a well controlled output
impedance, the output voltage under load can effectively be level
shifted down so that a larger positive spike can be sustained without
crossing the upper specification limit.
EXTERNAL CIRCUIT
DROOP
CONTROL
DIODE EMULATION
While in Power Savings Mode, the active phases will behave as if they
are in a standard buck configuration. To accomplish this, the lower
MOSFET is turned on only while there is current flowing to the load. This
behavior emulates the diode in a standard buck. The conduction loss
across the RDS(on) of the MOSFET, however, is much less than a diode,
resulting in a measurable power savings.
COMP
CC
IAVG
RC
While in Power Savings Mode, the gate drive voltage for the lower
MOSFETs of the active phases is reduced from the nominal 12V
that is utilized in Normal mode to 5.75V. Lowering the gate drive
voltage can have an appreciable effect on the efficiency of the
converter.
In order to utilize 5V gate drive at all times, 5V should be tied to
the PVCC pin and the GVOT pin should be shorted to the PVCC
pin. This configuration will allow for 5V gate drive in all modes of
operation.
Voltage Regulation
The integrating compensation network shown in Figure 8 insures
that the steady-state error in the output voltage is limited only to
the error in the reference voltage and offset errors in the OFS
current source, remote-sense and error amplifiers. Intersil
specifies the guaranteed tolerance of the ISL6328 to include the
combined tolerances of each of these elements.
The output of the error amplifier, VCOMP, is used by the
modulator to generate the PWM signals. The PWM signals
control the timing of the Internal MOSFET drivers and regulate
the converter output so that the voltage at FB is equal to the
voltage at REF. This will regulate the output voltage to be equal to
Equation 15. The internal and external circuitry that controls
voltage regulation is illustrated in Figure 8.
(EQ. 15)
The ISL6328 incorporates differential remote-sense amplification in
the feedback path. The differential sensing removes the voltage
error encountered when measuring the output voltage relative to the
controller ground reference point resulting in a more accurate
means of sensing output voltage.
Load-Line (Droop) Regulation
By adding a well controlled output impedance, the output voltage
can effectively be level shifted in a direction which works to
achieve a cost-effective solution can help to reduce the outputvoltage spike that results from fast load-current demand
changes.
The magnitude of the spike is dictated by the ESR and ESL of the output
capacitors selected. By positioning the no-load voltage level near the
18
IOFS
FB
GATE VOLTAGE OPTIMIZATION
V OUT = V REF – V OFS – V DROOP
ISL6328 INTERNAL CIRCUIT
+
RFB
+
-
+
VOUT
-
ERROR
AMPLIFIER
2k
(VDROOP + VOFS)
VSEN
VCOMP
∑
VID
DAC
RGND
FIGURE 8. OUTPUT VOLTAGE AND LOAD-LINE REGULATION WITH
OFFSET ADJUSTMENT
As shown in Figure 8, with droop enabled, the average current of
all active channels, IAVG, flows from FB through a load-line
regulation resistor RFB. The resulting voltage drop across RFB is
proportional to the output current, effectively creating an output
voltage droop with a steady-state value defined as:
V DROOP = I AVG ⋅ R FB
(EQ. 16)
The regulated output voltage is reduced by the droop voltage
VDROOP. The output voltage as a function of load current is
shown in Equation 17.
⎛ I OUT DCR
⎞
V OUT = V REF – V OFS – ⎜ ----------- ⋅ --------------- ⋅ K ⋅ R FB⎟
⎝ N R ISEN
⎠
(EQ. 17)
In Equation 17, VREF is the reference voltage, VOFS is the
programmed offset voltage, IOUT is the total output current of the
converter, K is the DC gain of the RC filter across the inductor (K
is defined in Equation 8), N is the number of active channels, and
DCR is the distributed inductor impedance value.
Droop Control
The DRPCTRL (Droop Control) pin is used to enable and/or
disable load line regulation on both the Core and Northbridge
regulators. The pin is also used to set the number of phases in
Power Savings Mode (PSI) mode. A single resistor tied from the
DRPCTRL pin to either GND or VCC will program the ISL6328 to
either enable or disable droop on both Core and Northbridge
simultaneously.
FN7621.1
June 7, 2011
ISL6328
TABLE 4. DRPCTRL FUNCTIONALITY
VOUT
RESISTOR VALUE
CORE DROOP
NB DROOP
100kΩ
Disabled
Disabled
50kΩ
Disabled
Enabled
20kΩ
Enabled
Enabled
0Ω
Enabled
Disabled
+
VOFS
-
RFB
VREF
E/A
FB
IOFS
If the DRPCTRL resistor is tied to ground, then the number of
phases in PSI mode is 1. If the DRPCTRL resistor is tied to VCC,
then the minimum number of phases in PSI mode is 2.
Output-Voltage Offset Programming
The ISL6328 allows the designer to accurately adjust the offset
voltage by connecting a resistor, ROFS, from the OFS pin to VCC or
GND. When ROFS is connected between OFS and VCC, the voltage
across it is regulated to 1.6V. This causes a proportional current
(IOFS) to flow into the FB pin and out of the OFS pin. If ROFS is
connected to ground, the voltage across it is regulated to 0.3V,
and IOFS flows into the OFS pin and out of the FB pin. The offset
current flowing through the resistor between VDIFF and FB will
generate the desired offset voltage which is equal to the product
(IOFS x RFB). These functions are shown in Figures 9 and 10.
Once the desired output offset voltage has been determined, use
the following formulas to set ROFS:
For Positive Offset (connect ROFS to GND):
0.3 ⋅ R FB
R OFS = ----------------------V OFFSET
(EQ. 18)
For Negative Offset (connect ROFS to VCC):
1.6 ⋅ R FB
R OFS = ----------------------V OFFSET
(EQ. 19)
-
OFS
ROFS
ISL6328
GND
-
1.6V
+
+
0.3V
GND
VCC
FIGURE 10. POSITIVE OFFSET OUTPUT VOLTAGE
PROGRAMMING
Dynamic VID
The AMD processor does not step the output voltage commands up or
down to the target voltage, but instead passes only the target voltage to
the ISL6328 through the SVI interface. The ISL6328 manages the
resulting VID-on-the-Fly transition in a controlled manner, supervising a
safe output voltage transition without discontinuity or disruption. The
ISL6328 begins slewing the DAC at 3.0mV/µs until the DAC and target
voltage are equal. Thus, the total time required for a dynamic VID
transition is dependent only on the size of the DAC change.
Advanced Adaptive Zero Shoot-Through
Deadtime Control (Patent Pending)
VOFS
+
RFB
The integrated drivers incorporate a unique adaptive deadtime
control technique to minimize deadtime, resulting in high efficiency
from the reduced freewheeling time of the lower MOSFET
body-diode conduction, and to prevent the upper and lower
MOSFETs from conducting simultaneously. This is accomplished by
ensuring either rising gate turns on its MOSFET with minimum and
sufficient delay after the other has turned off.
VREF
E/A
FB
IOFS
VCC
-
ROFS
OFS
-
ISL6328
0.3V
GND
FIGURE 9. NEGATIVE OFFSET OUTPUT VOLTAGE
PROGRAMMING
19
1.6V
+
+
VCC
During turn-off of the lower MOSFET, the PHASE voltage is
monitored until it reaches a -0.3V/+0.8V (forward/reverse inductor
current). At this time the UGATE is released to rise. An auto-zero
comparator is used to correct the rDS(ON) drop in the phase voltage
preventing false detection of the -0.3V phase level during rDS(ON)
conduction period. In the case of zero current, the UGATE is released
after 35ns delay of the LGATE dropping below 0.5V. When LGATE
first begins to transition low, this quick transition can disturb the
PHASE node and cause a false trip, so there is 20ns of blanking
time once LGATE falls until PHASE is monitored.
Once the PHASE is high, the advanced adaptive shoot-through
circuitry monitors the PHASE and UGATE voltages during a PWM
falling edge and the subsequent UGATE turn-off. If either the
UGATE falls to less than 1.75V above the PHASE or the PHASE
falls to less than +0.8V, the LGATE is released to turn-on.
FN7621.1
June 7, 2011
ISL6328
Initialization
Prior to initialization, proper conditions must exist on the EN, VCC,
GVOT, PVCC, ISEN2-, ISEN3- and ISEN4- pins. When the conditions
are met, the controller begins soft-start. Once the output voltage is
within the proper window of operation, the controller asserts
VDDPWRGD.
Power-On Reset
The ISL6328 requires VCC, PVCC and GVOT inputs to exceed their
rising POR thresholds before the ISL6328 has sufficient bias to
guarantee proper operation.
The bias voltage applied to VCC must reach the internal power-on reset
(POR) rising threshold. Once this threshold is reached, the ISL6328 has
enough bias to begin checking the driver POR inputs, EN, and channel
detect portions of the initialization cycle. Hysteresis between the rising
and falling thresholds assure the ISL6328 will not advertently turn off
unless the bias voltage drops substantially (see “Electrical
Specifications” on page 9).
The bias voltage applied to the PVCC pin powers the internal
MOSFET drivers of each output channel. In order for the ISL6328
to begin operation, the PVCC input must exceed its POR rising
threshold to guarantee proper operation of the internal drivers.
Hysteresis between the rising and falling thresholds assure that
once enabled, the ISL6328 will not inadvertently turn off unless
the PVCC bias voltage drops substantially (see “Electrical
Specifications” on page 9). Depending on the number of active
CORE channels determined by the Phase Detect block, the
external driver POR checking is supported by the Enable
Comparator.
Enable Comparator
The ISL6328 features a dual function enable input (EN) for
enabling the controller and power sequencing between the
controller and external drivers or another voltage rail. The enable
comparator holds the ISL6328 in shutdown until the voltage at
EN rises above 0.86V. The enable comparator has about 110mV
of hysteresis to prevent bounce. It is important that the driver ICs
reach their rising POR level before the ISL6328 becomes
enabled. The schematic in Figure 12 demonstrates sequencing
the ISL6328 with the ISL66xx family of Intersil MOSFET drivers,
which require 12V bias.
When selecting the value of the resistor divider the driver
maximum rising POR threshold should be used for calculating
the proper resistor values. This will prevent improper sequencing
events from creating false trips during soft-start.
If the controller is configured for 1- or 2-phase CORE operation,
then the resistor divider can be used for sequencing the
controller with another voltage rail. The resistor divider to EN
should be selected using a similar approach as the previous
driver discussion.
Phase Detection
The ISEN2-, ISEN3- and ISEN4- pins are monitored prior to softstart to determine the number of active CORE channel phases.
If ISEN4- is tied to VCC, the controller will set the channel firing
order and timing for 3-phase operation. If ISEN4- and ISEN3- are
20
tied to VCC, the controller will set the channel firing order and
timing for a 2-phase operation. The controller will configure itself
as a single phase regulator if ISEN4-, ISEN3- and ISEN2- are tied
to VCC (see “PWM Operation” on page 13 for details).
Soft-Start Output Voltage Targets
Once the POR and Phase Detect blocks and enable comparator
are satisfied, the controller will begin the soft-start sequence and
will ramp the CORE and NB output voltages up to the SVI
interface designated target level. Prior to soft-starting both CORE
and NB outputs, the ISL6328 must check the state of the SVI
interface inputs to determine the correct target voltages for both
outputs. When the controller is enabled, the state of the SVD and
SVC inputs are checked and the target output voltages set for
both CORE and NB outputs are set by the DAC (see “Serial VID
Interface (SVI)” on page 15). These targets will only change if the
EN signal is pulled low or after a POR reset of VCC.
Soft-Start
The soft-start sequence is composed of three periods, as shown
in Figure 11. At the beginning of soft-start, the DAC immediately
obtains the output voltage target. A 100µs fixed delay time, TDA,
proceeds the output voltage rise. After this delay period the
ISL6328 will begin ramping both CORE and NB output voltages
to the programmed DAC level at a fixed rate of 3.25mV/µs. The
amount of time required to ramp the output voltage to the final
DAC voltage is referred to as TDB, and can be calculated as
shown in Equation 20.
V DAC
TDB = -------------------------–3
3.0 ⋅ 10
(EQ. 20)
After the DAC voltage reaches the final VID setting, PGOOD will
be set to high.
VNB
400mV/DIV
TDA
VCORE
400mV/DIV
TDB
EN
5V/DIV
VDDPWRGD
5V/DIV
100µs/DIV
FIGURE 11. SOFT-START WAVEFORMS
Pre-Biased Soft-Start
The ISL6328 also has the ability to start up into a pre-charged output,
without causing any unnecessary disturbance. The FB pin is monitored
during soft-start, and should it be higher than the equivalent internal
ramping reference voltage, the output drives hold both MOSFETs off.
Once the internal ramping reference exceeds the FB pin
potential, the output drives are enabled, allowing the output to
FN7621.1
June 7, 2011
ISL6328
ramp from the pre-charged level to the final level dictated by the
DAC setting. Should the output be pre-charged to a level
exceeding the DAC setting, the output drives are enabled at the
end of the soft-start period, leading to an abrupt correction in the
output voltage down to the DAC-set level.
OUTPUT PRECHARGED
ABOVE DAC LEVEL
OUTPUT PRECHARGED
BELOW DAC LEVEL
At the inception of an overvoltage event, both on-board lower gate pins
are commanded low as are the active PWM outputs to the external
drivers, the PGOOD signal is driven low, and the ISL6328 latches off
normal PWM action. This turns on the all of the lower MOSFETs and
pulls the output voltage below a level that might cause damage to the
load. The lower MOSFETs remain driven ON until VSEN falls below
400mV. The ISL6328 will continue to protect the load in this fashion as
long as the overvoltage condition recurs. Once an overvoltage condition
ends the ISL6328 latches off, and must be reset by toggling POR,
before a soft-start can be re-initiated.
Pre-POR Overvoltage Protection
VCORE
400mV/DIV
EN
5V/DIV
100µs/DIV
Prior to PVCC, VCC and GVOT exceeding their POR levels, the
ISL6328 is designed to protect either load from any overvoltage
events that may occur. This is accomplished by means of an
internal 10kΩ resistor tied from PHASE to LGATE, which turns on
the lower MOSFET to control the output voltage until the
overvoltage event ceases or the input power supply cuts off. For
complete protection, the low side MOSFET should have a gate
threshold well below the maximum voltage rating of the
load/microprocessor.
FIGURE 12. SOFT-START WAVEFORMS FOR ISL6328-BASED
MULTIPHASE CONVERTER
-
Both CORE and NB output support start up into a pre-charged
output.
Fault Monitoring and Protection
The ISL6328 actively monitors both CORE and NB output
voltages and currents to detect fault conditions. Fault monitors
trigger protective measures to prevent damage to either load.
One common power-good indicator is provided for linking to
external system monitors. The schematic in Figure 13 outlines
the interaction between the fault monitors and the power-good
signal.
Power-Good Signal
The power-good pin (VDDPWRGD) is an open-drain logic output
that signals whether or not the ISL6328 is regulating both NB
and CORE output voltages within the proper levels, and whether
any fault conditions exist. This pin should be tied to a +5V source
through a resistor.
During shutdown and soft-start, VDDPWRGD pulls low and
releases high after a successful soft-start and both output
voltages are operating between the undervoltage and
overvoltage limits. PGOOD transitions low when an undervoltage,
overvoltage, or overcurrent condition is detected on either output
or when the controller is disabled by a POR reset or EN. In the
event of an overvoltage or overcurrent condition, the controller
latches off and PGOOD will not return high. Pending a POR reset
of the ISL6328 and successful soft-start, the PGOOD will return
high.
Overvoltage Protection
The ISL6328 constantly monitors the sensed output voltage on the
VSEN pin to detect if an overvoltage event occurs. When the output
voltage rises above the OVP trip level and exceeds the PGOOD OV
limit actions are taken by the ISL6328 to protect the
microprocessor load.
21
170µA
OCL
+
100µA
REPEAT FOR EACH
CORE CHANNEL
-
OCP
INB
I1
+
-
100µA
+
IAVG
OCP
CORE ONLY
NB ONLY
SOFT-START, FAULT
AND CONTROL LOGIC
NB ONLY
1.8V
+
OVP
-
VSEN_NB+
DAC - 300mV
CORE ONLY
1.8V
UV
+
+
OVP
DAC + 250mV
-
VDDPWRGD
OV
+
VSEN
DAC - 300mV
UV
+
ISL6328 INTERNAL CIRCUITRY
FIGURE 13. POWER GOOD AND PROTECTION CIRCUITRY
FN7621.1
June 7, 2011
ISL6328
In the event that during normal operation if the PVCC, VCC or
GVOT voltage falls back below the POR threshold, the pre-POR
overvoltage protection circuitry reactivates to protect from any
more pre-POR overvoltage events.
Undervoltage Detection
The undervoltage threshold is set at VDAC - 300mV typical. When
the output voltage (VSEN-RGND) is below the undervoltage
threshold, PGOOD gets pulled low. No other action is taken by the
controller. PGOOD will return high if the output voltage rises
above VDAC - 250mV typical.
Open Sense Line Protection
In the case that either of the remote sense lines, VSEN or GND,
become open, the ISL6328 is designed to detect this and shut
down the controller. This event is detected by monitoring small
currents that are fed out the VSEN and RGND pins. In the event of
an open sense line fault, the controller will continue to remain off
until the fault goes away, at which point the controller will reinitiate a soft-start sequence.
start, as shown in Figure 14. If the fault remains, the trip-retry
cycles will continue until either the fault is cleared or for a total of
seven attempts. If the fault is not cleared on the final attempt,
the controller disables UGATE and LGATE signals for both Core
and Northbridge and latches off requiring a POR of VCC to reset
the ISL6328.
It is important to note that during soft-start, the overcurrent trip
point is increased by a factor of 1.35. If the fault draws enough
current to trip overcurrent during normal run mode, it may not
draw enough current during the soft-start ramp period to trip
overcurrent while the output is ramping up. If a fault of this type
is affecting the output, then the regulator will complete soft-start
and the trip-retry counter will be reset to zero. Once the regulator
has completed soft-start, the overcurrent trip point will return to
it’s nominal setting and an overcurrent shutdown will be initiated.
This will result in a continuous hiccup mode.
OUTPUT CURRENT, 50A/DIV
Overcurrent Protection
The ISL6328 takes advantage of the proportionality between the
load current and the average current, IAVG, to detect an
overcurrent condition. See “Continuous Current Sampling” on
page 14 and “Channel-Current Balance” on page 15 for more
detail on how the average current is measured. Once the average
current exceeds 100µA, a comparator triggers the converter to
begin overcurrent protection procedures.
The overcurrent trip threshold is dictated by the DCR of the
inductors, the number of active channels, the DC gain of the
inductor RC filter and the RISEN resistor. The overcurrent trip
threshold is shown in Equation 21.
V IN – N ⋅ V OUT V OUT
N
1
I OCP = 100mA ⋅ ------------ ⋅ --- ⋅ R ISEN – ------------------------------------ ⋅ ------------V IN
2 ⋅ L ⋅ fS
DCR K
(EQ. 21)
Where:
R2
K = -------------------R1 + R2
See “Continuous Current Sampling” on
page 14.
fSW = Switching Frequency
Equation 21 is valid for both the Core regulator and the
Northbridge regulator. This equation includes the DC load current
as well as the total ripple current contributed by all the phases.
For the Northbridge regulator, N is 1.
During soft-start, the overcurrent trip point is boosted by a factor
of 1.35. Instead of comparing the average measured current to
100µA, the average current is compared to 135µA. Immediately
after soft-start is over, the comparison level changes to 100µA.
This is done to allow for start-up into an active load while still
supplying output capacitor in-rush current.
OVERCURRENT PROTECTION SHUTDOWN
At the beginning of overcurrent shutdown, the controller sets all
of the UGATE and LGATE signals low, puts PWM3 and PWM4 (if
active) in a high-impedance state, and forces VDDPWRGD low.
This turns off all of the upper and lower MOSFETs. The system
remains in this state for fixed period of 12ms. If the controller is
still enabled at the end of this wait period, it will attempt a soft22
0A
OUTPUT VOLTAGE,
500mV/DIV
0V
3ms/DIV
FIGURE 14. OVERCURRENT BEHAVIOR IN HICCUP MODE
Note that the energy delivered during trip-retry cycling is much
less than during full-load operation, so there is no thermal
hazard.
CORE REGULATOR OVERCURRENT
The ISL6328 features a Dual Overcurrent Protection (OCP)
feature on the Core that allows AMD processors to “throttle” the
load current beyond the normal OCP threshold for brief periods of
time. These current spikes occur when idle processor cores are
enabled. Dual OCP is not enabled until soft-start is complete.
When the Core average sensed current, IAVG, exceeds 100µA,
the Core regulator does not immediately initiate overcurrent
protection shutdown. The ISL6328 will, instead, source an
amount of current out of the OCP pin that is equivalent to the
amount of sensed current that exceeds 100µA. A capacitor tied
between the OCP pin and ground will immediately begin
charging. If the voltage across the capacitor, and thus the voltage
on the OCP pin, exceeds 2V then the ISL6328 will immediately
initiate overcurrent protection shutdown. If IAVG decreases to a
level below 100µA prior to the voltage on the OCP pin exceeding
2V, then the OCP pin is internally pulled to ground and the
voltage on the OCP capacitor is discharged. If, at any time, the
average sensed current exceeds 145µA, the ISL6328 will
immediately initiate overcurrent protection shutdown.
FN7621.1
June 7, 2011
ISL6328
It is recommended that the maximum current spike correspond
to an average sensed current level of 120µA. It is also
recommended that if the maximum current spike load was
applied to the Core regulator that overcurrent protection
shutdown initiate after 1ms. This would require a 0.01µF
capacitor be tied to the OCP pin. To calculate the OCP capacitor,
use Equation 22.
20μA ⋅ t DELAY
C OCP = -------------------------------------2V
(EQ. 22)
NORTHBRIDGE REGULATOR OVERCURRENT
The Northbridge regulator does not incorporate dual OCP. When
the sensed current of the Northbridge exceeds 100µA,
Overcurrent Protection Shutdown is initiated. The overcurrent
shutdown for the Northbridge regulator will only disable the
MOSFET drivers for the Northbridge. Once 7 retry attempts have
been executed unsuccessfully, the controller will disable UGATE
and LGATE signals for both Core and Northbridge and will latch
off requiring a POR of VCC to reset the ISL6328.
OVERCURRENT PROTECTION IN POWER SAVINGS
MODE
While in Power Savings Mode, the OCP trip point will be lower
than when running in Normal Mode and there is no
accommodation for current throttling. Equation 21, with N = 1,
will yield the OCP trip point for the Core regulator while in Power
Savings mode.
If an overcurrent event should occur while the system is in Power
Savings Mode, the ISL6328 will restart in the Normal state with
the PSI_L bit set to 1.
Individual Channel Overcurrent Limiting
The ISL6328 has the ability to limit the current in each individual
channel of the Core regulator without shutting down the entire
regulator. This is accomplished by continuously comparing the
sensed currents of each channel with a constant 170µA OCL
reference current. If a channel’s individual sensed current
exceeds this OCL limit, the UGATE signal of that channel is
immediately forced low, and the LGATE signal is forced high. This
turns off the upper MOSFET(s), turns on the lower MOSFET(s),
and stops the rise of current in that channel, forcing the current
in the channel to decrease. That channel’s UGATE signal will not
be able to return high until the sensed channel current falls back
below the 170µA reference.
General Design Guide
This design guide is intended to provide a high-level explanation of
the steps necessary to create a multiphase power converter. It is
assumed that the reader is familiar with many of the basic skills and
techniques referenced in the following. In addition to this guide,
Intersil provides complete reference designs that include
schematics, bills of materials, and example board layouts for all
common microprocessor applications.
Power Stages
The first step in designing a multiphase converter is to determine
the number of phases. This determination depends heavily on
the cost analysis which in turn depends on system constraints
23
that differ from one design to the next. Principally, the designer
will be concerned with whether components can be mounted on
both sides of the circuit board, whether through-hole components
are permitted, the total board space available for power-supply
circuitry, and the maximum amount of load current. Generally
speaking, the most economical solutions are those in which each
phase handles between 25A and 30A. All surface-mount designs
will tend toward the lower end of this current range. If
through-hole MOSFETs and inductors can be used, higher
per-phase currents are possible. In cases where board space is
the limiting constraint, current can be pushed as high as 40A per
phase, but these designs require heat sinks and forced air to cool
the MOSFETs, inductors and heat-dissipating surfaces.
MOSFETS
The choice of MOSFETs depends on the current each MOSFET will
be required to conduct, the switching frequency, the capability of
the MOSFETs to dissipate heat, and the availability and nature of
heat sinking and air flow.
LOWER MOSFET POWER CALCULATION
The calculation for power loss in the lower MOSFET is simple,
since virtually all of the loss in the lower MOSFET is due to current
conducted through the channel resistance (rDS(ON)). In Equation
23, IM is the maximum continuous output current, IPP is the
peak-to-peak inductor current (see Equation 2), and d is the duty
cycle (VOUT/VIN).
2 ⋅ (1 – d)
I L, PP
⎛ I M⎞ 2
P LOW, 1 = r DS ( ON ) ⋅ ⎜ -----⎟ ⋅ ( 1 – d ) + ----------------------------------12
⎝ N⎠
(EQ. 23)
An additional term can be added to the lower-MOSFET loss
equation to account for additional loss accrued during the dead
time when inductor current is flowing through the lower-MOSFET
body diode. This term is dependent on the diode forward voltage
at IM, VD(ON), the switching frequency, fS, and the length of dead
times, td1 and td2, at the beginning and the end of the lowerMOSFET conduction interval respectively.
⎛ I M I PP⎞
⎛ IM I ⎞
PP-⎟ ⋅ t
P LOW, 2 = V D ( ON ) ⋅ f S ⋅ ⎜ ----- + --------⎟ ⋅ t d1 + ⎜ ----- – ------2 ⎠
2 ⎠ d2
⎝N
⎝N
(EQ. 24)
The total maximum power dissipated in each lower MOSFET is
approximated by the summation of PLOW,1 and PLOW,2.
UPPER MOSFET POWER CALCULATION
In addition to rDS(ON) losses, a large portion of the upper-MOSFET
losses are due to currents conducted across the input voltage
(VIN) during switching. Since a substantially higher portion of the
upper-MOSFET losses are dependent on switching frequency, the
power calculation is more complex. Upper MOSFET losses can be
divided into separate components involving the upper-MOSFET
switching times, the lower-MOSFET body-diode reverse-recovery
charge, Qrr, and the upper MOSFET rDS(ON) conduction loss.
When the upper MOSFET turns off, the lower MOSFET does not
conduct any portion of the inductor current until the voltage at
the phase node falls below ground. Once the lower MOSFET
begins conducting, the current in the upper MOSFET falls to zero
as the current in the lower MOSFET ramps up to assume the full
inductor current. In Equation 25, the required time for this
FN7621.1
June 7, 2011
ISL6328
commutation is t1 and the approximated associated power loss
is PUP,1.
1.6
⎛ I M I PP⎞ ⎛ t 1 ⎞
P UP,1 ≈ V IN ⋅ ⎜ ----- + --------⎟ ⋅ ⎜ ----- ⎟ ⋅ f S
2 ⎠ ⎝ 2⎠
⎝N
1.4
(EQ. 25)
At turn on, the upper MOSFET begins to conduct and this
transition occurs over a time t2. In Equation 26, the approximate
power loss is PUP,2.
⎛ I M I PP⎞
P UP, 2 ≈ V IN ⋅ ⎜ ----- – --------⎟
2 ⎠
⎝N
⎛t ⎞
⋅ ⎜ ----2- ⎟ ⋅ f S
⎝ 2⎠
(EQ. 26)
A third component involves the lower MOSFET reverse-recovery
charge, Qrr. Since the inductor current has fully commutated to
the upper MOSFET before the lower-MOSFET body diode can
recover all of Qrr, it is conducted through the upper MOSFET
across VIN. The power dissipated as a result is PUP,3.
(EQ. 27)
P UP,3 = V IN ⋅ Q rr ⋅ f S
CBOOT_CAP (µF)
1.2
1.0
0.8
0.6
QGATE = 100nC
0.4
50nC
0.2
0.0
20nC
0.0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1.0
ΔVBOOT_CAP (V)
FIGURE 15. BOOTSTRAP CAPACITANCE vs BOOT RIPPLE VOLTAGE
Finally, the resistive part of the upper MOSFET is given in
Equation 28 as PUP,4.
2
I PP2
⎛ I M⎞
P UP,4 ≈ r DS ( ON ) ⋅ ⎜ -----⎟ ⋅ d + ---------12
⎝ N⎠
Gate Drive Voltage Versatility
(EQ. 28)
The total power dissipated by the upper MOSFET at full load can
now be approximated as the summation of the results from
Equations 25, 26, 27 and 28. Since the power equations depend
on MOSFET parameters, choosing the correct MOSFETs can be an
iterative process involving repetitive solutions to the loss
equations for different MOSFETs and different switching
frequencies.
Internal Bootstrap Device
All three integrated drivers feature an internal bootstrap schottky
diode. Simply adding an external capacitor across the BOOT and
PHASE pins completes the bootstrap circuit. The bootstrap
function is also designed to prevent the bootstrap capacitor from
overcharging due to the large negative swing at the PHASE node.
This reduces voltage stress on the boot to phase pins.
The bootstrap capacitor must have a maximum voltage rating
above PVCC + 4V and its capacitance value can be chosen from
Equation 29:
Q GATE
C BOOT_CAP ≥ --------------------------------ΔV BOOT_CAP
(EQ. 29)
⎛ Q G1 ⋅ PVCC
⎞
Q GATE = ⎜ ------------------------------ ⋅ N Q1⎟
V
⎝
⎠
GS1
where QG1 is the amount of gate charge per upper MOSFET at
VGS1 gate-source voltage and NQ1 is the number of control
MOSFETs. The ΔVBOOT_CAP term is defined as the allowable
droop in the rail of the upper gate drive.
The ISL6328 provides the user flexibility in choosing the gate
drive voltage for efficiency optimization. The controller ties the
upper and lower drive rails together. Simply applying a voltage
from 5V up to 12V on PVCC sets both gate drive rail voltages
simultaneously.
Package Power Dissipation
When choosing MOSFETs it is important to consider the amount
of power being dissipated in the integrated drivers located in the
controller. Since there are a total of three drivers in the controller
package, the total power dissipated by all three drivers must be
less than the maximum allowable power dissipation for the QFN
package.
Calculating the power dissipation in the drivers for a desired
application is critical to ensure safe operation. Exceeding the
maximum allowable power dissipation level will push the IC
beyond the maximum recommended operating junction
temperature of +125°C. The maximum allowable IC power
dissipation for the 6x6 QFN package is approximately 3.5W at
room temperature. See “Layout Considerations” on page 29 for
thermal transfer improvement suggestions.
When designing the ISL6328 into an application, it is
recommended that the following calculation is used to ensure
safe operation at the desired frequency for the selected
MOSFETs. The total gate drive power losses, PQg_TOT, due to the
gate charge of MOSFETs and the integrated driver’s internal
circuitry and their corresponding average driver current can be
estimated with Equations 30 and 31, respectively.
P Qg_TOT = P Qg_Q1 + P Qg_Q2 + I Q ⋅ VCC
3
P Qg_Q1 = --- ⋅ Q G1 ⋅ PVCC ⋅ F SW ⋅ N Q1 ⋅ N PHASE
2
P Qg_Q2 = Q G2 ⋅ PVCC ⋅ F SW ⋅ N Q2 ⋅ N PHASE
3
I DR = ⎛ --- ⋅ Q G1 ⋅ N + Q G2 ⋅ N Q2⎞ ⋅ N PHASE ⋅ F SW + I Q
⎝2
⎠
Q1
24
(EQ. 30)
(EQ. 31)
FN7621.1
June 7, 2011
ISL6328
Inductor DCR Current Sensing Component
Fine Tuning
I
VIN
L
L
MOSFET
LGATE(n)
INDUCTOR
VL(s)
+
+
DRIVER
DCR
BOOT
R1
D
RLO1
RG1
CDS
RGI1
ISENn+
In
CGS
-
Q1
ISENn+
VC(s)
RISEN
S
ISEN
PHASE
FIGURE 16. TYPICAL UPPER-GATE DRIVE TURN-ON PATH
FIGURE 18. DCR SENSING CONFIGURATION
PVCC
D
CGD
RHI2
R2
+
UGATE
VC(s)
C
ISL6328 INTERNAL CIRCUIT
G
COUT
-
CGD
RHI1
VOUT
-
PVCC
n
UGATE(n)
-
In Equations 30 and 31, PQg_Q1 is the total upper gate drive
power loss and PQg_Q2 is the total lower gate drive power loss;
the gate charge (QG1 and QG2) is defined at the particular gate to
source drive voltage PVCC in the corresponding MOSFET data
sheet; IQ is the driver total quiescent current with no load at both
drive outputs; NQ1 and NQ2 are the number of upper and lower
MOSFETs per phase, respectively; NPHASE is the number of active
phases. The IQ*VCC product is the quiescent power of the
controller without capacitive load and is typically 75mW at
300kHz.
G
LGATE
RLO2
RG2
CDS
RGI2
CGS
1. If the regulator is not utilizing droop, modify the circuit by
placing the frequency set resistor between FS and Ground for
the duration of this procedure.
Q2
S
2. Capture a transient event with the oscilloscope set to about
L/DCR/2 (sec/div). For example, with L = 1µH and
DCR = 1mΩ, set the oscilloscope to 500µs/div.
FIGURE 17. TYPICAL LOWER-GATE DRIVE TURN-ON PATH
The total gate drive power losses are dissipated among the resistive
components along the transition path and in the bootstrap diode. The
portion of the total power dissipated in the controller itself is the power
dissipated in the upper drive path resistance (PDR_UP), the lower drive
path resistance (PDR_UP), and in the boot strap diode (PBOOT). The rest
of the power will be dissipated by the external gate resistors (RG1 and
RG2) and the internal gate resistors (RGI1 and RGI2) of the MOSFETs.
Figures 16 and 17 show the typical upper and lower gate drives turn-on
transition path. The total power dissipation in the controller itself, PDR,
can be roughly estimated as:
P DR = P DR_UP + P DR_LOW + P BOOT + ( I Q ⋅ VCC )
P Qg_Q1
P BOOT = ------------------3
R HI1
R LO1
⎛
⎞ P Qg_Q1
P DR_UP = ⎜ ----------------------------------- + -------------------------------------⎟ ⋅ ------------------R
+
R
R
+
R
3
⎝ HI1
EXT1
LO1
EXT1⎠
R HI2
R LO2
⎛
⎞ P Qg_Q2
P DR_LOW = ⎜ ----------------------------------- + -------------------------------------⎟ ⋅ ------------------R
+
R
R
+
R
2
⎝ HI2
EXT2
LO2
EXT2⎠
R GI1
R GI2
R EXT2 = R G2 + -----------R EXT1 = R G1 + -----------N Q2
N Q1
25
Due to errors in the inductance and/or DCR it may be necessary
to adjust the value of R1 and R2 to match the time constants
correctly. The effects of time constant mismatch can be seen in
the form of droop overshoot or undershoot during the initial load
transient spike, as shown in Figure 19. Follow the steps below to
ensure the R-C and inductor L/DCR time constants are matched
accurately.
3. Record ΔV1 and ΔV2 as shown in Figure 19.
ΔV2
ΔV1
VOUT
ITRAN
ΔI
(EQ. 32)
FIGURE 19. TIME CONSTANT MISMATCH BEHAVIOR
FN7621.1
June 7, 2011
ISL6328
4. Select new values, R1,NEW and R2,NEW, for the time constant
resistors based on the original values, R1,OLD and R2,OLD,
using Equations 33 and 34.
ΔV 1
R 1, NEW = R 1, OLD ⋅ ---------ΔV
(EQ. 33)
ΔV 1
R 2, NEW = R 2, OLD ⋅ ---------ΔV 2
(EQ. 34)
2
5. Replace R1 and R2 with the new values and check to see that
the error is corrected. Repeat the procedure if necessary.
Loadline Regulation Resistor
The loadline regulation resistor, labeled RFB in Figure 8, sets
the desired loadline required for the application. Equation 35
can be used to calculate RFB.
V DROOP
MAX
R FB = ----------------------------------------------------I
⎛ OUT MAX
⎞
⎜ ----------------------- ⋅ DCR⎟
N
⎝
⎠
--------------------------------------------- ⋅ K
R ISEN
(EQ. 35)
To choose the value for RFB in this situation, please refer to
“Compensation Without Loadline Regulation” on page 26.
Compensation With Loadline Regulation
The load-line regulated converter behaves in a similar manner to
a peak current mode controller because the two poles at the
output filter L-C resonant frequency split with the introduction of
current information into the control loop. The final location of
these poles is determined by the system function, the gain of the
current signal, and the value of the compensation components,
RC and CC.
components depend on the relationships of f0 to the L-C pole
frequency and the ESR zero frequency. For each of the following
three, there is a separate set of equations for the compensation
components.
In Equation 36, L is the per-channel filter inductance divided by
the number of active channels; C is the sum total of all output
capacitors; ESR is the equivalent series resistance of the bulk
output filter capacitance; and VP-P is the peak-to-peak sawtooth
signal amplitude as described in the “Electrical Specifications”
on page 9.
Once selected, the compensation values in Equation 36 assure a
stable converter with reasonable transient performance. In most
cases, transient performance can be improved by making
adjustments to RC. Slowly increase the value of RC while
observing the transient performance on an oscilloscope until no
further improvement is noted. Normally, CC will not need
adjustment. Keep the value of CC from Equation 36 unless some
performance issue is noted
The optional capacitor C2, is sometimes needed to bypass noise
away from the PWM comparator (see Figure 20). Keep a position
available for C2, and be prepared to install a high-frequency
capacitor of between 22pF and 150pF in case any leading edge
jitter problem is noted.
Case 1:
2 ⋅ π ⋅ f 0 ⋅ V pp ⋅ L ⋅ C
R C = R FB ⋅ -----------------------------------------------------0.66 ⋅ V
IN
0.66 ⋅ V IN
C C = -------------------------------------------------2 ⋅ π ⋅ V PP ⋅ R FB ⋅ f 0
C2 (OPTIONAL)
Case 2:
RC
CC
1
------------------------------- > f 0
2⋅π⋅ L⋅C
1
1
------------------------------ ≤ f 0 < ---------------------------------2 ⋅ π ⋅ C ⋅ ESR
2⋅π⋅ L⋅C
V PP ⋅ ( 2 ⋅ π ) 2 ⋅ f 02 ⋅ L ⋅ C
R C = R FB ⋅ --------------------------------------------------------------0.66 ⋅ V
COMP
IN
0.66 ⋅ V IN
C C = ---------------------------------------------------------------------------------2
( 2 ⋅ π ) ⋅ f 02 ⋅ V PP ⋅ R FB ⋅ L ⋅ C
FB
ISL6328
RFB
Case 3:
VSEN
FIGURE 20. COMPENSATION CONFIGURATION FOR
LOAD-LINE REGULATED ISL6328 CIRCUIT
Since the system poles and zero are affected by the values of the
components that are meant to compensate them, the solution to
the system equation becomes fairly complicated. Fortunately,
there is a simple approximation that comes very close to an
optimal solution. Treating the system as though it were a voltagemode regulator, by compensating the L-C poles and the ESR zero
of the voltage mode approximation, yields a solution that is
always stable with very close to ideal transient performance.
(EQ. 36)
1
f 0 > ----------------------------------2 ⋅ π ⋅ C ⋅ ESR
2 ⋅ π ⋅ f 0 ⋅ V pp ⋅ L
R C = R FB ⋅ ------------------------------------------0.66 ⋅ V IN ⋅ ESR
0.66 ⋅ V IN ⋅ ESR ⋅ C
C C = -------------------------------------------------------------2 ⋅ π ⋅ V PP ⋅ R FB ⋅ f 0 ⋅ L
Compensation Without Loadline Regulation
The non load-line regulated converter is accurately modeled as a
voltage-mode regulator with two poles at the L-C resonant
frequency and a zero at the ESR frequency. A type III controller,
as shown in Figure 21, provides the necessary compensation.
Select a target bandwidth for the compensated system, f0. The
target bandwidth must be large enough to assure adequate
transient performance, but smaller than 1/3 of the per-channel
switching frequency. The values of the compensation
26
FN7621.1
June 7, 2011
ISL6328
1
------------------------------- > f 0
2⋅π⋅ L⋅C
C2
RC
CC
2 ⋅ π ⋅ f 0 ⋅ V pp ⋅ L ⋅ C
R C = R FB ⋅ -----------------------------------------------------0.66 ⋅ V
COMP
Case 1:
FB
C1
IN
0.66 ⋅ V IN
C C = -------------------------------------------------2 ⋅ π ⋅ V PP ⋅ R FB ⋅ f 0
ISL6328
RFB
R1
1
1
------------------------------- ≤ f 0 < ----------------------------------2 ⋅ π ⋅ C ⋅ ESR
2⋅π⋅ L⋅C
VSEN
FIGURE 21. COMPENSATION CIRCUIT WITHOUT LOAD-LINE
REGULATION
The first step is to choose the desired bandwidth, f0, of the
compensated system. Choose a frequency high enough to assure
adequate transient performance but not higher than 1/3 of the
switching frequency. The type-III compensator has an extra
high-frequency pole, fHF. This pole can be used for added noise
rejection or to assure adequate attenuation at the error-amplifier
high order pole and zero frequencies. A good general rule is to
choose fHF = 10f0, but it can be higher if desired. Choosing fHF to
be lower than 10f0 can cause problems with too much phase shift
below the system bandwidth.
C ⋅ ESR
R 1 = R FB ⋅ ---------------------------------------L ⋅ C – C ⋅ ESR
Case 2:
V PP ⋅ ( 2 ⋅ π ) 2 ⋅ f 02 ⋅ L ⋅ C
R C = R FB ⋅ --------------------------------------------------------------0.66 ⋅ V
IN
(EQ. 38)
0.66 ⋅ V IN
C C = ---------------------------------------------------------------------------------2
( 2 ⋅ π ) ⋅ f 02 ⋅ V PP ⋅ R FB ⋅ L ⋅ C
1
f 0 > ----------------------------------2 ⋅ π ⋅ C ⋅ ESR
Case 3:
2 ⋅ π ⋅ f 0 ⋅ V pp ⋅ L
R C = R FB ⋅ ------------------------------------------0.66 ⋅ V IN ⋅ ESR
0.66 ⋅ V IN ⋅ ESR ⋅ C
C C = -------------------------------------------------------------2 ⋅ π ⋅ V PP ⋅ R FB ⋅ f 0 ⋅ L
Output Filter Design
L ⋅ C – C ⋅ ESR
C 1 = ---------------------------------------R FB
0.75 ⋅ V IN
C 2 = ----------------------------------------------------------------------------------------------------2
( 2 ⋅ π ) ⋅ f 0 ⋅ f HF ⋅ ( L ⋅ C ) ⋅ R FB ⋅ V P – P
(EQ. 37)
2
V PP ⋅ ⎛ 2π⎞ ⋅ f 0 ⋅ f HF ⋅ L ⋅ C ⋅ R FB
⎝ ⎠
R C = ------------------------------------------------------------------------------------0.75 ⋅ V ⋅ ( 2 ⋅ π ⋅ f HF ⋅ L ⋅ C – 1 )
IN
0.75 ⋅ V IN ⋅ ( 2 ⋅ π ⋅ f HF ⋅ L ⋅ C – 1 )
C C = ----------------------------------------------------------------------------------------------------( 2 ⋅ π ) 2 ⋅ f 0 ⋅ f HF ⋅ ( L ⋅ C ) ⋅ R FB ⋅ V P – P
In the solutions to the compensation equations, there is a single
degree of freedom. For the solutions presented in Equation 38,
RFB is selected arbitrarily. The remaining compensation
components are then selected according to Equation 38.
The output inductors and the output capacitor bank together to
form a low-pass filter responsible for smoothing the pulsating
voltage at the phase nodes. The output filter also must provide
the transient energy until the regulator can respond. Because it
has a low bandwidth compared to the switching frequency, the
output filter limits the system transient response. The output
capacitors must supply or sink load current while the current in
the output inductors increases or decreases to meet the
demand.
In high-speed converters, the output capacitor bank is usually the
most costly (and often the largest) part of the circuit. Output filter
design begins with minimizing the cost of this part of the circuit.
The critical load parameters in choosing the output capacitors are
the maximum size of the load step, ΔI, the load-current slew rate,
di/dt, and the maximum allowable output-voltage deviation under
transient loading, ΔVMAX. Capacitors are characterized according
to their capacitance, ESR, and ESL (equivalent series inductance).
In Equation 38, L is the per-channel filter inductance divided by
the number of active channels; C is the sum total of all output
capacitors; ESR is the equivalent-series resistance of the bulk
output-filter capacitance; and VPP is the peak-to-peak sawtooth
signal amplitude as described in “Electrical Specifications” on
page 9.
27
FN7621.1
June 7, 2011
ISL6328
di
ΔV ≈ ESL ⋅ ----- + ESR ⋅ ΔI
dt
(EQ. 39)
The filter capacitor must have sufficiently low ESL and ESR so
that ΔV < ΔVMAX.
Most capacitor solutions rely on a mixture of high frequency
capacitors with relatively low capacitance in combination with bulk
capacitors having high capacitance but limited high-frequency
performance. Minimizing the ESL of the high-frequency capacitors
allows them to support the output voltage as the current increases.
Minimizing the ESR of the bulk capacitors allows them to supply the
increased current with less output voltage deviation.
The ESR of the bulk capacitors also creates the majority of the
output-voltage ripple. As the bulk capacitors sink and source the
inductor AC ripple current (see “Interleaving” on page 12 and
Equation 3), a voltage develops across the bulk capacitor ESR
equal to IC,PP (ESR). Thus, once the output capacitors are
selected, the maximum allowable ripple voltage, VPP(MAX),
determines the lower limit on the inductance.
⎛V – N ⋅ V
⎞
OUT⎠ ⋅ V OUT
⎝ IN
L ≥ ESR ⋅ -----------------------------------------------------------f S ⋅ V IN ⋅ V PP( MAX )
(EQ. 40)
Since the capacitors are supplying a decreasing portion of the
load current while the regulator recovers from the transient, the
capacitor voltage becomes slightly depleted. The output
inductors must be capable of assuming the entire load current
before the output voltage decreases more than ΔVMAX. This
places an upper limit on inductance.
Equation 41 gives the upper limit on L for the cases when the
trailing edge of the current transient causes a greater
output-voltage deviation than the leading edge. Equation 42
addresses the leading edge. Normally, the trailing edge dictates
the selection of L because duty cycles are usually less than 50%.
Nevertheless, both inequalities should be evaluated, and L
should be selected based on the lower of the two results. In each
equation, L is the per-channel inductance, C is the total output
capacitance, and N is the number of active channels.
28
2 ⋅ N ⋅ C ⋅ VO
L ≤ ------------------------------- ⋅ ΔV MAX – ( ΔI ⋅ ESR )
( ΔI ) 2
(EQ. 41)
⋅N⋅C
---------------------------- ⋅ ΔV MAX – ( ΔI ⋅ ESR ) ⋅ ⎛ V IN – V O⎞
L ≤ 1.25
⎝
⎠
( ΔI ) 2
(EQ. 42)
Switching Frequency
There are a number of variables to consider when choosing the
switching frequency, as there are considerable effects on the
upper MOSFET loss calculation. These effects are outlined in
“MOSFETs” on page 23, and they establish the upper limit for the
switching frequency. The lower limit is established by the
requirement for fast transient response and small output-voltage
ripple as outlined in “Output Filter Design” on page 27. Choose
the lowest switching frequency that allows the regulator to meet
the transient-response requirements.
Switching frequency is determined by the selection of the
frequency-setting resistor, RT. Figure 22 and Equation 43 are provided
to assist in selecting the correct value for RT.
R T = 10
[10.61 – ( 1.035 ⋅ log ( f S ) ) ]
(EQ. 43)
1k
RT (kΩ)
At the beginning of the load transient, the output capacitors
supply all of the transient current. The output voltage will initially
deviate by an amount approximated by the voltage drop across
the ESL. As the load current increases, the voltage drop across
the ESR increases linearly until the load current reaches its final
value. The capacitors selected must have sufficiently low ESL and
ESR so that the total output-voltage deviation is less than the
allowable maximum. Neglecting the contribution of inductor
current and regulator response, the output voltage initially
deviates by an amount:
100
10
10k
100k
1M
10M
SWITCHING FREQUENCY (Hz)
FIGURE 22. RT vs SWITCHING FREQUENCY
Input Capacitor Selection
The input capacitors are responsible for sourcing the AC
component of the input current flowing into the upper MOSFETs.
Their RMS current capacity must be sufficient to handle the AC
component of the current drawn by the upper MOSFETs which is
related to duty cycle and the number of active phases.
FN7621.1
June 7, 2011
ISL6328
IL(P-P) = 0
IL(P-P) = 0.25 IO
0.3
IL(P-P) = 0.5 IO
IL(P-P) = 0.75 IO
INPUT-CAPACITOR CURRENT (IRMS/IO)
INPUT-CAPACITOR CURRENT (IRMS/IO)
0.3
0.2
0.1
0
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (VO/VIN)
The voltage rating of the capacitors should also be at least 1.25x
greater than the maximum input voltage. Figures 24 and 25 provide
the same input RMS current information for three-phase and
2-phase designs respectively. Use the same approach for selecting
the bulk capacitor type and number.
INPUT-CAPACITOR CURRENT (IRMS/IO)
IL(P-P) = 0
IL(P-P) = 0.5 IO
IL(P-P) = 0.75 IO
IL(P-P) = 0.5 IO
IL(P-P) = 0.75 IO
0
0
0.2
0.4
0.6
0.8
1.0
Layout Considerations
MOSFETs switch very fast and efficiently. The speed with which
the current transitions from one device to another causes voltage
spikes across the interconnecting impedances and parasitic
circuit elements. These voltage spikes can degrade efficiency,
radiate noise into the circuit and lead to device overvoltage
stress. Careful component selection, layout, and placement
minimizes these voltage spikes. Consider, as an example, the
turnoff transition of the upper PWM MOSFET. Prior to turnoff, the
upper MOSFET was carrying channel current. During the turnoff,
current stops flowing in the upper MOSFET and is picked up by
the lower MOSFET. Any inductance in the switched current path
generates a large voltage spike during the switching interval.
Careful component selection, tight layout of the critical
components, and short, wide circuit traces minimize the
magnitude of voltage spikes.
There are two sets of critical components in a DC/DC converter
using a ISL6328 controller. The power components are the most
critical because they switch large amounts of energy. Next are
small signal components that connect to sensitive nodes or
supply critical bypassing current and signal coupling.
0.2
0.1
0
IL(P-P) = 0
FIGURE 25. NORMALIZED INPUT-CAPACITOR RMS
CURRENT FOR 2-PHASE CONVERTER
For a four-phase design, use Figure 23 to determine the
input-capacitor RMS current requirement set by the duty cycle,
maximum sustained output current (IO), and the ratio of the
peak-to-peak inductor current (IL(P-P)) to IO. Select a bulk
capacitor with a ripple current rating which will minimize the
total number of input capacitors required to support the RMS
current calculated.
IL(P-P) = 0.25 IO
0.1
DUTY CYCLE (VIN/VO)
FIGURE 23. NORMALIZED INPUT-CAPACITOR RMS CURRENT vs
DUTY CYCLE FOR 4-PHASE CONVERTER
0.3
0.2
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (VIN/VO)
FIGURE 24. NORMALIZED INPUT-CAPACITOR RMS CURRENT
FOR 3-PHASE CONVERTER
Low capacitance, high-frequency ceramic capacitors are needed in
addition to the input bulk capacitors to suppress leading and
falling edge voltage spikes. The spikes result from the high current
slew rate produced by the upper MOSFET turn on and off. Select
low ESL ceramic capacitors and place one as close as possible to
each upper MOSFET drain to minimize board parasitics and
maximize suppression.
29
The power components should be placed first, which include the
MOSFETs, input and output capacitors, and the inductors. It is
important to have a symmetrical layout for each power train,
preferably with the controller located equidistant from each.
Symmetrical layout allows heat to be dissipated equally across all
power trains. Equidistant placement of the controller to the CORE
and NB power trains it controls through the integrated drivers
helps keep the gate drive traces equally short, resulting in equal
trace impedances and similar drive capability of all sets of
MOSFETs.
When placing the MOSFETs try to keep the source of the upper
FETs and the drain of the lower FETs as close as thermally possible.
Input high-frequency capacitors, CHF, should be placed close to the
drain of the upper FETs and the source of the lower FETs. Input
bulk capacitors, CBULK, case size typically limits following the
same rule as the high-frequency input capacitors. Place the input
FN7621.1
June 7, 2011
ISL6328
bulk capacitors as close to the drain of the upper FETs as possible
and minimize the distance to the source of the lower FETs.
Locate the output inductors and output capacitors between the
MOSFETs and the load. The high-frequency output decoupling
capacitors (ceramic) should be placed as close as practicable to the
decoupling target, making use of the shortest connection paths to
any internal planes, such as vias to GND next or on the capacitor
solder pad.
The critical small components include the bypass capacitors
(CFILTER) for VCC and PVCC, and many of the components
surrounding the controller including the feedback network and
current sense components. Locate the VCC/PVCC bypass
capacitors as close to the ISL6328 as possible. It is especially
important to locate the components associated with the
feedback circuit close to their respective controller pins, since
they belong to a high-impedance circuit loop, sensitive to EMI
pick-up.
A multi-layer printed circuit board is recommended. Figure 26 shows
the connections of the critical components for the converter. Note
that capacitors CIN and COUT could each represent numerous
physical capacitors. Dedicate one solid layer, usually the one
underneath the component side of the board, for a ground plane
and make all critical component ground connections with vias to
this layer. Dedicate another solid layer as a power plane and break
this plane into smaller islands of common voltage levels. Keep the
metal runs from the PHASE terminal to output inductors short. The
power plane should support the input power and output power
nodes. Use copper filled polygons on the top and bottom circuit
layers for the phase nodes. Use the remaining printed circuit layers
for small signal wiring.
Routing UGATE, LGATE, and PHASE Traces
Great attention should be paid to routing the UGATE, LGATE, and PHASE
traces since they drive the power train MOSFETs using short, high
current pulses. It is important to size them as large and as short as
possible to reduce their overall impedance and inductance. They should
be sized to carry at least one ampere of current (0.02” to 0.05”). Going
between layers with vias should also be avoided, but if so, use two vias
for interconnection when possible.
Extra care should be given to the LGATE traces in particular since
keeping their impedance and inductance low helps to significantly
reduce the possibility of shoot-through. It is also important to route
each channels UGATE and PHASE traces in as close proximity as
possible to reduce their inductances.
Current Sense Component
Placement and Trace Routing
One of the most critical aspects of the ISL6328 regulator layout
is the placement of the inductor DCR current sense components
and traces. The R-C current sense components must be placed
as close to their respective ISEN+ and ISEN- pins on the ISL6328
as possible.
The sense traces that connect the R-C sense components to each side
of the output inductors should be routed on the bottom of the board,
away from the noisy switching components located on the top of the
board. These traces should be routed side by side, and they should be
very thin traces. It’s important to route these traces as far away from
any other noisy traces or planes as possible. These traces should pick
up as little noise as possible. These traces should also originate from
the geometric center of the inductor pin pads and that location should
be the single point of contact the trace makes with its respective net.
Thermal Management
For maximum thermal performance in high current, high
switching frequency applications, connecting the thermal GND
pad of the ISL6328 to the ground plane with multiple vias is
recommended. This heat spreading allows the part to achieve
its full thermal potential. It is also recommended that the
controller be placed in a direct path of airflow if possible to help
thermally manage the part.
30
FN7621.1
June 7, 2011
ISL6328
VCC
+5V
VCC RSVD
ISEN1TCOMP1
CS1CS1+
ISEN1+
CS2-
ISEN2-
CS2+
ISEN2+
TCOMP2
PWM3
PWM4
CS3-
ISEN3-
PVCC
CS3+
ISEN3+
GVOT
CS4-
ISEN4-
PWM3
PWM4
+12V
ISL6328
CS4+
+12V
BOOT1
ISEN4+
RNTC*
CS_NB-
ISEN_NB-
CS_NB+
ISEN_NB+
UGATE1
PHASE1
UGATE1
PHASE1
LGATE1
DRPCTRL
CS1CS1+
BOOT2
CS3CS3+
+12V
FS
LGATE1
PWM3
PWM4
+12V
PVCC
VCC
BOOT2
+12V
GND
OFS
UGATE2
PHASE2
OCP
+12V
ISL6614
BOOT1PWM1
PWM2
+12V
SVC
SVD
PWROK
VDDPWRGD
LGATE2
VSEN
RGND
APA
UGATE2
PHASE2
CS2CS2+
Core
+12V
UGATE_NB
PHASE_NB
EN
LGATE_NB
PGND
LGATE2
KELVIN
SENSE
LINES
CORE_FB
BOOT_NB
ENABLE
CS4CS4+
CPU
North
Bridge
CS_NBCS_NB+
KELVIN
SENSE
LINE
VSEN_NB
CORE_FB
FB_PSI
COMP
RED COMPONENTS:
LOCATE CLOSE TO IC TO
MINIMIZE CONNECTION PATH
FB_NB
FB
COMP_NB
BLUE COMPONENTS:
LOCATE NEAR LOAD
(MINIMIZE CONNECTION PATH)
GND
MAGENTA COMPONENTS:
LOCATE CLOSE TO SWITCHING TRANSISTORS
(MINIMIZE CONNECTION PATH)
KEY
HEAVY TRACE ON CIRCUIT PLANE LAYER
ISLAND ON POWER PLANE LAYER
ISLAND ON CIRCUIT PLANE LAYER
KELVIN TYPE TRACE/SENSE LINE
(Keep these traces away from any switching nodes)
VIA CONNECTION TO GROUND PLANE
*LOCATE NTC RESISTOR CLOSE TO PHASE1 INDUCTOR
FIGURE 26. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS
31
FN7621.1
June 7, 2011
ISL6328
Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make
sure you have the latest Rev.
DATE
REVISION
6/7/11
FN7621.1
CHANGE
Initial Release to web.
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32
FN7621.1
June 7, 2011
ISL6328
Package Outline Drawing
L48.6x6B
48 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 0, 9/09
4X 4.4
6.00
44X 0.40
A
B
6
PIN 1
INDEX AREA
6
PIN #1 INDEX AREA
48
37
1
6.00
36
4 .40 ± 0.15
25
12
0.15
(4X)
13
24
0.10 M C A B
0.05 M C
TOP VIEW
48X 0.45 ± 0.10
4 48X 0.20
BOTTOM VIEW
SEE DETAIL "X"
(
SEATING PLANE
0.08 C
( 44 X 0 . 40 )
( 5. 75 TYP )
C
0.10 C
BASE PLANE
MAX 1.00
SIDE VIEW
4. 40 )
C
0 . 2 REF
5
( 48X 0 . 20 )
( 48X 0 . 65 )
0 . 00 MIN.
0 . 05 MAX.
DETAIL "X"
TYPICAL RECOMMENDED LAND PATTERN
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3. Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
5. Tiebar shown (if present) is a non-functional feature.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 indentifier may be
either a mold or mark feature.
33
FN7621.1
June 7, 2011
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