ON LM2575D2T-015G 1.0 a, adjustable output voltage, step−down switching regulator Datasheet

LM2575
1.0 A, Adjustable Output
Voltage, Step−Down
Switching Regulator
The LM2575 series of regulators are monolithic integrated circuits
ideally suited for easy and convenient design of a step−down
switching regulator (buck converter). All circuits of this series are
capable of driving a 1.0 A load with excellent line and load regulation.
These devices are available in fixed output voltages of 3.3 V, 5.0 V,
12 V, 15 V, and an adjustable output version.
These regulators were designed to minimize the number of external
components to simplify the power supply design. Standard series of
inductors optimized for use with the LM2575 are offered by several
different inductor manufacturers.
Since the LM2575 converter is a switch−mode power supply, its
efficiency is significantly higher in comparison with popular
three−terminal linear regulators, especially with higher input voltages.
In many cases, the power dissipated by the LM2575 regulator is so
low, that no heatsink is required or its size could be reduced
dramatically.
The LM2575 features include a guaranteed ±4% tolerance on output
voltage within specified input voltages and output load conditions, and
±10% on the oscillator frequency (±2% over 0°C to 125°C). External
shutdown is included, featuring 80 mA typical standby current. The
output switch includes cycle−by−cycle current limiting, as well as
thermal shutdown for full protection under fault conditions.
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TO−220
TV SUFFIX
CASE 314B
1
5
Heatsink surface connected to Pin 3
TO−220
T SUFFIX
CASE 314D
1
5
Pin
Features
• 3.3 V, 5.0 V, 12 V, 15 V, and Adjustable Output Versions
• Adjustable Version Output Voltage Range of 1.23 V to 37 V ±4%
•
•
•
•
•
•
•
•
•
•
Maximum Over Line and Load Conditions
Guaranteed 1.0 A Output Current
Wide Input Voltage Range: 4.75 V to 40 V
Requires Only 4 External Components
52 kHz Fixed Frequency Internal Oscillator
TTL Shutdown Capability, Low Power Standby Mode
High Efficiency
Uses Readily Available Standard Inductors
Thermal Shutdown and Current Limit Protection
Moisture Sensitivity Level (MSL) Equals 1
Pb−Free Packages are Available*
Vin
Output
Ground
Feedback
ON/OFF
D2PAK
D2T SUFFIX
CASE 936A
1
5
Heatsink surface (shown as terminal 6 in
case outline drawing) is connected to Pin 3
ORDERING INFORMATION
See detailed ordering and shipping information in the package
dimensions section on page 24 of this data sheet.
Applications
•
•
•
•
•
•
1.
2.
3.
4.
5.
Simple and High−Efficiency Step−Down (Buck) Regulators
Efficient Pre−Regulator for Linear Regulators
On−Card Switching Regulators
Positive to Negative Converters (Buck−Boost)
Negative Step−Up Converters
Power Supply for Battery Chargers
DEVICE MARKING INFORMATION
See general marking information in the device marking
section on page 25 of this data sheet.
*For additional information on our Pb−Free strategy and soldering details, please
download the ON Semiconductor Soldering and Mounting Techniques
Reference Manual, SOLDERRM/D.
© Semiconductor Components Industries, LLC, 2005
November, 2005 − Rev. 8
1
Publication Order Number:
LM2575/D
LM2575
Typical Application (Fixed Output Voltage Versions)
Feedback
7.0 V − 40 V
Unregulated
DC Input
+Vin
Cin
100 mF
4
LM2575
Output
1
3 GND 5
2
ON/OFF
L1
330 mH
D1
1N5819
Cout
330 mF
5.0 V Regulated
Output 1.0 A Load
Representative Block Diagram and Typical Application
Unregulated
DC Input
+Vin
3.1 V Internal
Regulator
1
ON/OFF
ON/OFF
5
Cin
4
Feedback
R2
Fixed Gain
Error Amplifier Comparator
R1
1.0 k
Current
Limit
Output
Voltage Versions
R2
(W)
3.3 V
5.0 V
12 V
15 V
1.7 k
3.1 k
8.84 k
11.3 k
For adjustable version
R1 = open, R2 = 0 W
Driver
Latch
Freq
Shift
18 kHz
1.235 V
Band−Gap
Reference
L1
Output
1.0 Amp
Switch
52 kHz
Oscillator
Reset
2
GND
Thermal
Shutdown
Regulated
Output
Vout
Cout
D1
3
Load
This device contains 162 active transistors.
Figure 1. Block Diagram and Typical Application
ABSOLUTE MAXIMUM RATINGS (Absolute Maximum Ratings indicate limits beyond which damage to the device may occur.)
Rating
Symbol
Value
Unit
Maximum Supply Voltage
Vin
45
V
ON/OFF Pin Input Voltage
−
−0.3 V ≤ V ≤ +Vin
V
Output Voltage to Ground (Steady−State)
−
−1.0
V
Power Dissipation
Case 314B and 314D (TO−220, 5−Lead)
Thermal Resistance, Junction−to−Ambient
Thermal Resistance, Junction−to−Case
Case 936A (D2PAK)
Thermal Resistance, Junction−to−Ambient (Figure 34)
Thermal Resistance, Junction−to−Case
PD
RqJA
RqJC
PD
RqJA
RqJC
Internally Limited
65
5.0
Internally Limited
70
5.0
W
°C/W
°C/W
W
°C/W
°C/W
Storage Temperature Range
Tstg
−65 to +150
°C
Minimum ESD Rating (Human Body Model: C = 100 pF, R = 1.5 kW)
−
3.0
kV
Lead Temperature (Soldering, 10 s)
−
260
°C
Maximum Junction Temperature
TJ
150
°C
Maximum ratings are those values beyond which device damage can occur. Maximum ratings applied to the device are individual stress limit
values (not normal operating conditions) and are not valid simultaneously. If these limits are exceeded, device functional operation is not implied,
damage may occur and reliability may be affected.
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LM2575
OPERATING RATINGS (Operating Ratings indicate conditions for which the device is intended to be functional, but do not guarantee
specific performance limits. For guaranteed specifications and test conditions, see the Electrical Characteristics.)
Rating
Symbol
Value
Unit
Operating Junction Temperature Range
TJ
−40 to +125
°C
Supply Voltage
Vin
40
V
SYSTEM PARAMETERS ([Note 1] Test Circuit Figure 14)
ELECTRICAL CHARACTERISTICS (Unless otherwise specified, Vin = 12 V for the 3.3 V, 5.0 V, and Adjustable version, Vin = 25 V
for the 12 V version, and Vin = 30 V for the 15 V version. ILoad = 200 mA. For typical values TJ = 25°C, for min/max values TJ is the
operating junction temperature range that applies [Note 2], unless otherwise noted.)
Characteristics
Symbol
Min
Typ
Max
Unit
Output Voltage (Vin = 12 V, ILoad = 0.2 A, TJ = 25°C)
Vout
3.234
3.3
3.366
V
Output Voltage (4.75 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A)
TJ = 25°C
TJ = −40 to +125°C
Vout
3.168
3.135
3.3
−
3.432
3.465
η
−
75
−
%
Output Voltage (Vin = 12 V, ILoad = 0.2 A, TJ = 25°C)
Vout
4.9
5.0
5.1
V
Output Voltage (8.0 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A)
TJ = 25°C
TJ = −40 to +125°C
Vout
4.8
4.75
5.0
−
5.2
5.25
η
−
77
−
%
Output Voltage (Vin = 25 V, ILoad = 0.2 A, TJ = 25°C)
Vout
11.76
12
12.24
V
Output Voltage (15 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A)
TJ = 25°C
TJ = −40 to +125°C
Vout
11.52
11.4
12
−
12.48
12.6
η
−
88
−
%
Output Voltage (Vin = 30 V, ILoad = 0.2 A, TJ = 25°C)
Vout
14.7
15
15.3
V
Output Voltage (18 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A)
TJ = 25°C
TJ = −40 to +125°C
Vout
14.4
14.25
15
−
15.6
15.75
η
−
88
−
%
Feedback Voltage (Vin = 12 V, ILoad = 0.2 A, Vout = 5.0 V, TJ = 25°C)
VFB
1.217
1.23
1.243
V
Feedback Voltage (8.0 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A, Vout = 5.0 V)
TJ = 25°C
TJ = −40 to +125°C
VFB
1.193
1.18
1.23
−
1.267
1.28
−
77
−
LM2575−3.3 (Note 1 Test Circuit Figure 14)
Efficiency (Vin = 12 V, ILoad = 1.0 A)
V
LM2575−5 ([Note 1] Test Circuit Figure 14)
Efficiency (Vin = 12 V, ILoad = 1.0 A)
V
LM2575−12 (Note 1 Test Circuit Figure 14)
Efficiency (Vin = 15V, ILoad = 1.0 A)
V
LM2575−15 (Note 1 Test Circuit Figure 14)
Efficiency (Vin = 18 V, ILoad = 1.0 A)
V
LM2575 ADJUSTABLE VERSION (Note 1 Test Circuit Figure 14)
η
Efficiency (Vin = 12 V, ILoad = 1.0 A, Vout = 5.0 V)
V
%
1. External components such as the catch diode, inductor, input and output capacitors can affect switching regulator system performance.
When the LM2575 is used as shown in the Figure 14 test circuit, system performance will be as shown in system parameters section.
2. Tested junction temperature range for the LM2575:
Tlow = −40°C
Thigh = +125°C
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LM2575
DEVICE PARAMETERS
ELECTRICAL CHARACTERISTICS (Unless otherwise specified, Vin = 12 V for the 3.3 V, 5.0 V, and Adjustable version, Vin = 25 V
for the 12 V version, and Vin = 30 V for the 15 V version. ILoad = 200 mA. For typical values TJ = 25°C, for min/max values TJ is the
operating junction temperature range that applies [Note 2], unless otherwise noted.)
Symbol
Characteristics
Min
Typ
Max
−
−
25
−
100
200
−
47
42
52
−
−
−
58
63
−
−
1.0
−
1.2
1.3
94
98
−
1.7
1.4
2.3
−
3.0
3.2
−
−
0.8
6.0
2.0
20
−
−
5.0
−
9.0
11
15
−
80
−
200
400
Unit
ALL OUTPUT VOLTAGE VERSIONS
Feedback Bias Current (Vout = 5.0 V Adjustable Version Only)
TJ = 25°C
TJ = −40 to +125°C
Ib
Oscillator Frequency Note 3
TJ = 25°C
TJ = 0 to +125°C
TJ = −40 to +125°C
fosc
Saturation Voltage (Iout = 1.0 A Note 4)
TJ = 25°C
TJ = −40 to +125°C
Vsat
Max Duty Cycle (“on”) Note 5
DC
Current Limit (Peak Current Notes 4 and 3)
TJ = 25°C
TJ = −40 to +125°C
ICL
Output Leakage Current Notes 6 and 7, TJ = 25°C
Output = 0 V
Output = −1.0 V
IL
Quiescent Current Note 6
TJ = 25°C
TJ = −40 to +125°C
IQ
Standby Quiescent Current (ON/OFF Pin = 5.0 V (“off”))
TJ = 25°C
TJ = −40 to +125°C
nA
kHz
V
A
mA
mA
mA
Istby
ON/OFF Pin Logic Input Level (Test Circuit Figure 14)
Vout = 0 V
TJ = 25°C
TJ = −40 to +125°C
Vout = Nominal Output Voltage
TJ = 25°C
TJ = −40 to +125°C
%
V
VIH
2.2
2.4
1.4
−
−
−
−
−
1.2
−
1.0
0.8
−
−
15
0
30
5.0
VIL
mA
ON/OFF Pin Input Current (Test Circuit Figure 14)
ON/OFF Pin = 5.0 V (“off”), TJ = 25°C
ON/OFF Pin = 0 V (“on”), TJ = 25°C
IIH
IIL
3. The oscillator frequency reduces to approximately 18 kHz in the event of an output short or an overload which causes the regulated output
voltage to drop approximately 40% from the nominal output voltage. This self protection feature lowers the average dissipation of the IC by
lowering the minimum duty cycle from 5% down to approximately 2%.
4. Output (Pin 2) sourcing current. No diode, inductor or capacitor connected to output pin.
5. Feedback (Pin 4) removed from output and connected to 0 V.
6. Feedback (Pin 4) removed from output and connected to +12 V for the Adjustable, 3.3 V, and 5.0 V versions, and +25 V for the 12 V and
15 V versions, to force the output transistor “off”.
7. Vin = 40 V.
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4
LM2575
0.6
Vin = 20 V
ILoad = 200 mA
Normalized at
TJ = 25°C
0.4
0.2
0
−0.2
−0.4
−0.6
−50
−25
0
1.0
Vout , OUTPUT VOLTAGE CHANGE (%)
Vout , OUTPUT VOLTAGE CHANGE (%)
TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 14)
25
50
75
100
0.6
0.2
5.0
10
15
20
25
30
Vin, INPUT VOLTAGE (V)
Figure 2. Normalized Output Voltage
Figure 3. Line Regulation
35
40
3.0
1.1
IO , OUTPUT CURRENT (A)
Vsat , SATURATION VOLTAGE (V)
0
TJ, JUNCTION TEMPERATURE (°C)
1.0
0.9
−40°C
0.8
25°C
0.7
0.6
125°C
0.5
2.5
2.0
1.5
1.0
0.5
Vin = 25 V
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
0
−50
1.0
50
75
100
Figure 5. Current Limit
125
20
ILoad = 1.0 A
IQ , QUIESCENT CURRENT (mA)
DVout = 5%
Rind = 0.2 W
1.2
ILoad = 200 mA
0.8
0.6
0.4
−50
25
Figure 4. Switch Saturation Voltage
1.4
1.0
0
TJ, JUNCTION TEMPERATURE (°C)
1.8
1.6
−25
SWITCH CURRENT (A)
2.0
INPUT−OUTPUT DIFFERENTIAL (V)
12 V and 15 V
0
1.2
0.4
3.3 V, 5.0 V and Adj
0.4
−0.2
125
ILoad = 200 mA
TJ = 25°C
0.8
Vout = 5.0 V
Measured at
Ground Pin
TJ = 25°C
18
16
14
ILoad = 1.0 A
12
10
ILoad = 200 mA
8.0
6.0
−25
0
25
50
75
100
4.0
125
0
5.0
10
15
20
25
30
TJ, JUNCTION TEMPERATURE (°C)
Vin, INPUT VOLTAGE (V)
Figure 6. Dropout Voltage
Figure 7. Quiescent Current
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35
40
120
TJ = 25°C
100
80
60
40
20
0
0
5.0
10
15
20
25
30
35
120
Istby , STANDBY QUIESCENT CURRENT (μ A)
Istby , STANDBY QUIESCENT CURRENT (μ A)
LM2575
40
100
80
60
40
20
0
−50
50
75
100
Figure 9. Standby Quiescent Current
IFB , FEEDBACK PIN CURRENT (nA)
−2.0
−4.0
−6.0
−8.0
−25
0
25
50
75
100
20
0
−20
−40
−50
125
−25
0
25
50
75
100
TJ, JUNCTION TEMPERATURE (°C)
Figure 11. Feedback Pin Current
I Load, LOAD CURRENT (A) Vout , OUTPUT VOLTAGE
CHANGE (mV)
Figure 10. Oscillator Frequency
10 V
0
OUTPUT 1.0 A
CURRENT
(PIN 2)
0
100
0
−100
INDUCTOR 1.0 A
CURRENT
0.5 A
OUTPUT 20 mV
RIPPLE
/DIV
VOLTAGE
5.0 ms/DIV
Figure 12. Switching Waveforms
1.0
0.5
0
100 ms/DIV
Figure 13. Load Transient Response
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125
Adjustable
Version Only
TJ, JUNCTION TEMPERATURE (°C)
OUTPUT
VOLTAGE
(PIN 2)
25
Figure 8. Standby Quiescent Current
40
−10
−50
0
TJ, JUNCTION TEMPERATURE (°C)
Vin = 12 V
Normalized at 25°C
0
−25
Vin, INPUT VOLTAGE (V)
2.0
NORMALIZED FREQUENCY (%)
Vin = 12 V
VON/OFF = 5.0 V
125
LM2575
5.0 Output Voltage Versions
Feedback
4
Vin
+
Output
3
Vin
Unregulated
DC Input
8.0 V − 40 V
2
ON/OFF
5
GND
Vout
Regulated
Output
L1
330 mH
LM2575−5
1
Cin
100 mF/50 V
D1
1N5819
Cout
330 mF
/16 V
Load
−
Adjustable Output Voltage Versions
Feedback
4
Vin
+
LM2575
Adjustable
1
3
Unregulated
DC Input
8.0 V − 40 V
GND
5
Vout
Regulated
Output
L1
330 mH
Output
2
ON/OFF
Cin
100 mF/50 V
D1
1N5819
Cout
330 mF
/16 V
R2
Load
R1
−
V out + V
ref
ǒ
R2 + R1
ǒ1 ) R2
Ǔ
R1
V out
V ref
Ǔ
–1
Where Vref = 1.23 V, R1
between 1.0 kW and 5.0 kW
Figure 14. Typical Test Circuit
PCB LAYOUT GUIDELINES
As in any switching regulator, the layout of the printed
circuit board is very important. Rapidly switching currents
associated with wiring inductance, stray capacitance and
parasitic inductance of the printed circuit board traces can
generate voltage transients which can generate
electromagnetic interferences (EMI) and affect the desired
operation. As indicated in the Figure 14, to minimize
inductance and ground loops, the length of the leads
indicated by heavy lines should be kept as short as possible.
For best results, single−point grounding (as indicated) or
ground plane construction should be used.
On the other hand, the PCB area connected to the Pin 2
(emitter of the internal switch) of the LM2575 should be
kept to a minimum in order to minimize coupling to sensitive
circuitry.
Another sensitive part of the circuit is the feedback. It is
important to keep the sensitive feedback wiring short. To
assure this, physically locate the programming resistors near
to the regulator, when using the adjustable version of the
LM2575 regulator.
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LM2575
PIN FUNCTION DESCRIPTION
Pin
Symbol
Description (Refer to Figure 1)
1
Vin
This pin is the positive input supply for the LM2575 step−down switching regulator. In order to minimize
voltage transients and to supply the switching currents needed by the regulator, a suitable input bypass
capacitor must be present (Cin in Figure 1).
2
Output
3
GND
4
Feedback
This pin senses regulated output voltage to complete the feedback loop. The signal is divided by the
internal resistor divider network R2, R1 and applied to the non−inverting input of the internal error amplifier.
In the Adjustable version of the LM2575 switching regulator this pin is the direct input of the error amplifier
and the resistor network R2, R1 is connected externally to allow programming of the output voltage.
5
ON/OFF
It allows the switching regulator circuit to be shut down using logic level signals, thus dropping the total
input supply current to approximately 80 mA. The input threshold voltage is typically 1.4 V. Applying a
voltage above this value (up to +Vin) shuts the regulator off. If the voltage applied to this pin is lower than
1.4 V or if this pin is connected to ground, the regulator will be in the “on” condition.
This is the emitter of the internal switch. The saturation voltage Vsat of this output switch is typically 1.0 V.
It should be kept in mind that the PCB area connected to this pin should be kept to a minimum in order to
minimize coupling to sensitive circuitry.
Circuit ground pin. See the information about the printed circuit board layout.
DESIGN PROCEDURE
Buck Converter Basics
The inductor current during this time is:
The LM2575 is a “Buck” or Step−Down Converter which
is the most elementary forward−mode converter. Its basic
schematic can be seen in Figure 15.
The operation of this regulator topology has two distinct
time periods. The first one occurs when the series switch is
on, the input voltage is connected to the input of the inductor.
The output of the inductor is the output voltage, and the
rectifier (or catch diode) is reverse biased. During this
period, since there is a constant voltage source connected
across the inductor, the inductor current begins to linearly
ramp upwards, as described by the following equation:
I
L(on)
+
I
For the buck converter with ideal components, the duty
cycle can also be described as:
V
d + out
V
in
During this “on” period, energy is stored within the core
material in the form of magnetic flux. If the inductor is
properly designed, there is sufficient energy stored to carry
the requirements of the load during the “off” period.
D1
Figure 16 shows the buck converter idealized waveforms
of the catch diode voltage and the inductor current.
Von(SW)
Vout
Cout
Diode Voltage
Vin
L
t
d + on , where T is the period of switching.
T
L
L
ǒVout – VDǓ toff
This period ends when the power switch is once again
turned on. Regulation of the converter is accomplished by
varying the duty cycle of the power switch. It is possible to
describe the duty cycle as follows:
ǒVin – VoutǓ ton
Power
Switch
L(off)
+
RLoad
Power
Switch
Off
Power
Switch
On
Power
Switch
Off
Power
Switch
On
Time
VD(FWD)
Inductor Current
Figure 15. Basic Buck Converter
The next period is the “off” period of the power switch.
When the power switch turns off, the voltage across the
inductor reverses its polarity and is clamped at one diode
voltage drop below ground by catch dioded. Current now
flows through the catch diode thus maintaining the load
current loop. This removes the stored energy from the
inductor.
Ipk
ILoad(AV)
Imin
Diode
Power
Switch
Diode
Power
Switch
Time
Figure 16. Buck Converter Idealized Waveforms
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LM2575
Procedure (Fixed Output Voltage Version) In order to simplify the switching regulator design, a step−by−step design
procedure and example is provided.
Procedure
Example
Given Parameters:
Vout = Regulated Output Voltage (3.3 V, 5.0 V, 12 V or 15 V)
Vin(max) = Maximum DC Input Voltage
ILoad(max) = Maximum Load Current
Given Parameters:
Vout = 5.0 V
Vin(max) = 20 V
ILoad(max) = 0.8 A
1. Controller IC Selection
According to the required input voltage, output voltage and
current, select the appropriate type of the controller IC output
voltage version.
1. Controller IC Selection
According to the required input voltage, output voltage,
current polarity and current value, use the LM2575−5
controller IC
2. Input Capacitor Selection (Cin)
To prevent large voltage transients from appearing at the input
and for stable operation of the converter, an aluminium or
tantalum electrolytic bypass capacitor is needed between the
input pin +Vin and ground pin GND. This capacitor should be
located close to the IC using short leads. This capacitor should
have a low ESR (Equivalent Series Resistance) value.
2. Input Capacitor Selection (Cin)
A 47 mF, 25 V aluminium electrolytic capacitor located near
to the input and ground pins provides sufficient bypassing.
3. Catch Diode Selection (D1)
A. For this example the current rating of the diode is 1.0 A.
3. Catch Diode Selection (D1)
A. Since the diode maximum peak current exceeds the
regulator maximum load current the catch diode current
rating must be at least 1.2 times greater than the maximum
load current. For a robust design the diode should have a
current rating equal to the maximum current limit of the
LM2575 to be able to withstand a continuous output short
B. The reverse voltage rating of the diode should be at least
1.25 times the maximum input voltage.
B. Use a 30 V 1N5818 Schottky diode, or any of the
suggested fast recovery diodes shown in the Table 4.
4. Inductor Selection (L1)
A. According to the required working conditions, select the
correct inductor value using the selection guide from
Figures 17 to 21.
B. From the appropriate inductor selection guide, identify the
inductance region intersected by the Maximum Input
Voltage line and the Maximum Load Current line. Each
region is identified by an inductance value and an inductor
code.
C. Select an appropriate inductor from the several different
manufacturers part numbers listed in Table 1 or Table 2.
When using Table 2 for selecting the right inductor the
designer must realize that the inductor current rating must
be higher than the maximum peak current flowing through
the inductor. This maximum peak current can be calculated
as follows:
ǒVin–VoutǓ ton
I
+I
)
p(max) Load(max)
2L
4. Inductor Selection (L1)
A. Use the inductor selection guide shown in Figures 17
to 21.
B. From the selection guide, the inductance area intersected
by the 20 V line and 0.8 A line is L330.
C. Inductor value required is 330 mH. From the Table 1 or
Table 2, choose an inductor from any of the listed
manufacturers.
where ton is the “on” time of the power switch and
V
ton + out x 1
V
fosc
in
For additional information about the inductor, see the
inductor section in the “External Components” section of
this data sheet.
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LM2575
Procedure (Fixed Output Voltage Version) (continued)In order to simplify the switching regulator design, a step−by−step design
procedure and example is provided.
Procedure
Example
5. Output Capacitor Selection (Cout)
A. Since the LM2575 is a forward−mode switching regulator
with voltage mode control, its open loop 2−pole−2−zero
frequency characteristic has the dominant pole−pair
determined by the output capacitor and inductor values. For
stable operation and an acceptable ripple voltage,
(approximately 1% of the output voltage) a value between
100 mF and 470 mF is recommended.
B. Due to the fact that the higher voltage electrolytic capacitors
generally have lower ESR (Equivalent Series Resistance)
numbers, the output capacitor’s voltage rating should be at
least 1.5 times greater than the output voltage. For a 5.0 V
regulator, a rating at least 8V is appropriate, and a 10 V or
16 V rating is recommended.
5. Output Capacitor Selection (Cout)
A. Cout = 100 mF to 470 mF standard aluminium electrolytic.
B. Capacitor voltage rating = 16 V.
Procedure (Adjustable Output Version: LM2575−Adj)
Procedure
Example
Given Parameters:
Vout = Regulated Output Voltage
Vin(max) = Maximum DC Input Voltage
ILoad(max) = Maximum Load Current
Given Parameters:
Vout = 8.0 V
Vin(max) = 12 V
ILoad(max) = 1.0 A
1. Programming Output Voltage
To select the right programming resistor R1 and R2 value (see
Figure 14) use the following formula:
1. Programming Output Voltage (selecting R1 and R2)
Select R1 and R2:
ǒ
ǒ
V out + 1.23 1 )
Ǔ
R2
V out + V
1)
ref
R1 where Vref = 1.23 V
ǒ
R2 + R1
Resistor R1 can be between 1.0 k and 5.0 kW. (For best
temperature coefficient and stability with time, use 1% metal
film resistors).
V out
R2 + R1
–1
V
ref
ǒ
Ǔ
V out
V
ref
Ǔ
R2
R1 Select R1 = 1.8 kW
Ǔ
*1
+ 1.8 k
8.0 V
ǒ1.23
* 1Ǔ
V
R2 = 9.91 kW, choose a 9.88 k metal film resistor.
2. Input Capacitor Selection (Cin)
To prevent large voltage transients from appearing at the input
and for stable operation of the converter, an aluminium or
tantalum electrolytic bypass capacitor is needed between the
input pin +Vin and ground pin GND This capacitor should be
located close to the IC using short leads. This capacitor should
have a low ESR (Equivalent Series Resistance) value.
2. Input Capacitor Selection (Cin)
A 100 mF aluminium electrolytic capacitor located near the
input and ground pin provides sufficient bypassing.
For additional information see input capacitor section in the
“External Components” section of this data sheet.
3. Catch Diode Selection (D1)
A. Since the diode maximum peak current exceeds the
regulator maximum load current the catch diode current
rating must be at least 1.2 times greater than the maximum
load current. For a robust design, the diode should have a
current rating equal to the maximum current limit of the
LM2575 to be able to withstand a continuous output short.
B. The reverse voltage rating of the diode should be at least
1.25 times the maximum input voltage.
3. Catch Diode Selection (D1)
A. For this example, a 3.0 A current rating is adequate.
B. Use a 20 V 1N5820 or MBR320 Schottky diode or any
suggested fast recovery diode in the Table 4.
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LM2575
Procedure (Adjustable Output Version: LM2575−Adj) (continued)
Procedure
Example
4. Inductor Selection (L1)
A. Use the following formula to calculate the inductor Volt x
microsecond [V x ms] constant:
V out
6
E x T + V – V out
x 10 [V x ms]
in
V on
F[Hz]
ǒ
4. Inductor Selection (L1)
A. Calculate E x T [V x ms] constant:
Ǔ
E x T + (12 – 8.0) x 8.0 x 1000 + 51 [V x ms]
12
52
B. Match the calculated E x T value with the corresponding
number on the vertical axis of the Inductor Value Selection
Guide shown in Figure 21. This E x T constant is a measure
of the energy handling capability of an inductor and is
dependent upon the type of core, the core area, the number
of turns, and the duty cycle.
C. Next step is to identify the inductance region intersected by
the E x T value and the maximum load current value on the
horizontal axis shown in Figure 21.
D. From the inductor code, identify the inductor value. Then
select an appropriate inductor from the Table 1 or Table 2.
The inductor chosen must be rated for a switching
frequency of 52 kHz and for a current rating of 1.15 x IIoad.
The inductor current rating can also be determined by
calculating the inductor peak current:
I
p(max)
+I
Load(max)
)
B. E x T = 51 [V x ms]
C. ILoad(max) = 1.0 A
Inductance Region = L220
D. Proper inductor value = 220 mH
Choose the inductor from the Table 1 or Table 2.
ǒVin – VoutǓ ton
2L
where ton is the “on” time of the power switch and
V out
x 1
f osc
in
For additional information about the inductor, see the
inductor section in the “External Components” section of
this data sheet.
t on +
V
5. Output Capacitor Selection (Cout)
A. Since the LM2575 is a forward−mode switching regulator
with voltage mode control, its open loop 2−pole−2−zero
frequency characteristic has the dominant pole−pair
determined by the output capacitor and inductor values.
5. Output Capacitor Selection (Cout)
A.
Cout w 7.785 12 + 53 μF
8.220
To achieve an acceptable ripple voltage, select
Cout = 100 mF electrolytic capacitor.
For stable operation, the capacitor must satisfy the
following requirement:
V in(max)
[μF]
Cout w 7.785
V out x L [μH]
B. Capacitor values between 10 mF and 2000 mF will satisfy
the loop requirements for stable operation. To achieve an
acceptable output ripple voltage and transient response, the
output capacitor may need to be several times larger than
the above formula yields.
C. Due to the fact that the higher voltage electrolytic capacitors
generally have lower ESR (Equivalent Series Resistance)
numbers, the output capacitor’s voltage rating should be at
least 1.5 times greater than the output voltage. For a 5.0 V
regulator, a rating of at least 8V is appropriate, and a 10 V
or 16 V rating is recommended.
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LM2575
60
20
15
10
H1000
Vin , MAXIMUM INPUT VOLTAGE (V)
Vin , MAXIMUM INPUT VOLTAGE (V)
INDUCTOR VALUE SELECTION GUIDE
L680
L470
8.0
L330
7.0
L220
L150
6.0
L100
5.0
0.2
0.3
0.4
0.5
0.6
0.8
L680
12
L470
10
L330
9.0
L220
8.0
L150
0.3
0.4
0.5
0.6
0.7 0.8 0.9 1.0
IL, MAXIMUM LOAD CURRENT (A)
Figure 17. LM2575−3.3
Figure 18. LM2575−5.0
60
H2200
Vin , MAXIMUM INPUT VOLTAGE (V)
Vin , MAXIMUM INPUT VOLTAGE (V)
H1000
15
IL, MAXIMUM LOAD CURRENT (A)
H1500
H1000
H680
H470
20
18
17
L680
16
L470
L330
15
14
0.2
H1500
7.0
0.2
1.0
60
40
30
25
60
40
25
20
L220
0.3
0.4
0.5
0.6
H2200
40
35
30
H1500
H1000
25
H680
20
19
L680
L470
L330
18
L220
17
0.2
0.7 0.8 0.9 1.0
0.3
0.4
0.5
0.6
IL, MAXIMUM LOAD CURRENT (A)
IL, MAXIMUM LOAD CURRENT (A)
Figure 19. LM2575−12
Figure 20. LM2575−15
ET, VOLTAGE TIME (Vμ s)
200
150
125
H2200
H1500
H1000
H680
100
80
70
60
50
L470
L330
L220
30
20
0.2
H470
L680
40
L150
L100
0.3
0.4
0.5
0.6
0.7 0.8 0.9 1.0
IL, MAXIMUM LOAD CURRENT (A)
Figure 21. LM2575−Adj
NOTE: This Inductor Value Selection Guide is applicable for continuous mode only.
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H470
22
0.7 0.8 0.9 1.0
LM2575
Table 1. Inductor Selection Guide
Inductor
Code
Inductor
Value
Pulse Eng
Renco
AIE
Tech 39
L100
100 mH
PE−92108
RL2444
415−0930
77 308 BV
L150
150 mH
PE−53113
RL1954
415−0953
77 358 BV
L220
220 mH
PE−52626
RL1953
415−0922
77 408 BV
L330
330 mH
PE−52627
RL1952
415−0926
77 458 BV
L470
470 mH
PE−53114
RL1951
415−0927
−
L680
680 mH
PE−52629
RL1950
415−0928
77 508 BV
H150
150 mH
PE−53115
RL2445
415−0936
77 368 BV
H220
220 mH
PE−53116
RL2446
430−0636
77 410 BV
H330
330 mH
PE−53117
RL2447
430−0635
77 460 BV
H470
470 mH
PE−53118
RL1961
430−0634
−
H680
680 mH
PE−53119
RL1960
415−0935
77 510 BV
H1000
1000 mH
PE−53120
RL1959
415−0934
77 558 BV
H1500
1500 mH
PE−53121
RL1958
415−0933
−
H2200
2200 mH
PE−53122
RL2448
415−0945
77 610 BV
Table 2. Inductor Selection Guide
Inductance
Current
(mH)
(A)
THT
SMT
THT
SMT
THT
SMT
SMT
0.32
67143940
67144310
RL−1284−68−43
RL1500−68
PE−53804
PE−53804−S
DO1608−68
0.58
67143990
67144360
RL−5470−6
RL1500−68
PE−53812
PE−53812−S
DO3308−683
0.99
67144070
67144450
RL−5471−5
RL1500−68
PE−53821
PE−53821−S
DO3316−683
1.78
67144140
67144520
RL−5471−5
−
PE−53830
PE−53830−S
DO5022P−683
0.48
67143980
67144350
RL−5470−5
RL1500−100
PE−53811
PE−53811−S
DO3308−104
0.82
67144060
67144440
RL−5471−4
RL1500−100
PE−53820
PE−53820−S
DO3316−104
1.47
67144130
67144510
RL−5471−4
−
PE−53829
PE−53829−S
DO5022P−104
0.39
−
67144340
RL−5470−4
RL1500−150
PE−53810
PE−53810−S
DO3308−154
0.66
67144050
67144430
RL−5471−3
RL1500−150
PE−53819
PE−53819−S
DO3316−154
1.20
67144120
67144500
RL−5471−3
−
PE−53828
PE−53828−S
DO5022P−154
0.32
67143960
67144330
RL−5470−3
RL1500−220
PE−53809
PE−53809−S
DO3308−224
0.55
67144040
67144420
RL−5471−2
RL1500−220
PE−53818
PE−53818−S
DO3316−224
1.00
67144110
67144490
RL−5471−2
−
PE−53827
PE−53827−S
DO5022P−224
0.42
67144030
67144410
RL−5471−1
RL1500−330
PE−53817
PE−53817−S
DO3316−334
0.80
67144100
67144480
RL−5471−1
−
PE−53826
PE−53826−S
DO5022P−334
68
100
150
220
330
Schott
Renco
Pulse Engineering
Coilcraft
NOTE: Table 1 and Table 2 of this Indicator Selection Guide shows some examples of different manufacturer products suitable for design
with the LM2575.
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LM2575
Table 3. Example of Several Inductor Manufacturers Phone/Fax Numbers
Pulse Engineering Inc.
Phone
Fax
+ 1−619−674−8100
+ 1−619−674−8262
Pulse Engineering Inc. Europe
Phone
Fax
+ 353 93 24 107
+ 353 93 24 459
Renco Electronics Inc.
Phone
Fax
+ 1−516−645−5828
+ 1−516−586−5562
AIE Magnetics
Phone
Fax
+ 1−813−347−2181
Coilcraft Inc.
Phone
Fax
+ 1−708−322−2645
+ 1−708−639−1469
Coilcraft Inc., Europe
Phone
Fax
+ 44 1236 730 595
+ 44 1236 730 627
Tech 39
Phone
Fax
+ 33 8425 2626
+ 33 8425 2610
Schott Corp.
Phone
Fax
+ 1−612−475−1173
+ 1−612−475−1786
Table 4. Diode Selection Guide gives an overview about both surface−mount and through−hole diodes for an
effective design. Device listed in bold are available from ON Semiconductor.
Schottky
Ultra−Fast Recovery
1.0 A
3.0 A
1.0 A
VR
SMT
THT
SMT
THT
20 V
SK12
1N5817
SR102
SK32
MBRD320
1N5820
MBR320
SR302
30 V
MBRS130LT3
SK13
1N5818
SR103
11DQ03
SK33
MBRD330
1N5821
MBR330
SR303
31DQ03
40 V
MBRS140T3
SK14
10BQ040
10MQ040
1N5819
SR104
11DQ04
MBRS340T3
MBRD340
30WQ04
SK34
1N5822
MBR340
SR304
31DQ04
50 V
MBRS150
10BQ050
MBR150
SR105
11DQ05
MBRD350
SK35
30WQ05
MBR350
SR305
11DQ05
SMT
THT
MURS120T3
MUR120
11DF1
HER102
SMT
THT
MURS320T3
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3.0 A
10BF10
MURD320
31DF1
HER302
MUR320
30WF10
MUR420
LM2575
EXTERNAL COMPONENTS
(below 0.05 W), there is a possibility of an unstable feedback
loop, resulting in oscillation at the output. This situation can
occur when a tantalum capacitor, that can have a very low
ESR, is used as the only output capacitor.
Input Capacitor (Cin)
The Input Capacitor Should Have a Low ESR
For stable operation of the switch mode converter a low
ESR (Equivalent Series Resistance) aluminium or solid
tantalum bypass capacitor is needed between the input pin
and the ground pin to prevent large voltage transients from
appearing at the input. It must be located near the regulator
and use short leads. With most electrolytic capacitors, the
capacitance value decreases and the ESR increases with
lower temperatures. For reliable operation in temperatures
below −25°C larger values of the input capacitor may be
needed. Also paralleling a ceramic or solid tantalum
capacitor will increase the regulator stability at cold
temperatures.
At Low Temperatures, Put in Parallel Aluminium
Electrolytic Capacitors with Tantalum Capacitors
Electrolytic capacitors are not recommended for
temperatures below −25°C. The ESR rises dramatically at
cold temperatures and typically rises 3 times at −25°C and
as much as 10 times at −40°C. Solid tantalum capacitors
have much better ESR spec at cold temperatures and are
recommended for temperatures below −25°C. They can be
also used in parallel with aluminium electrolytics. The value
of the tantalum capacitor should be about 10% or 20% of the
total capacitance. The output capacitor should have at least
50% higher RMS ripple current rating at 52 kHz than the
peak−to−peak inductor ripple current.
RMS Current Rating of Cin
The important parameter of the input capacitor is the RMS
current rating. Capacitors that are physically large and have
large surface area will typically have higher RMS current
ratings. For a given capacitor value, a higher voltage
electrolytic capacitor will be physically larger than a lower
voltage capacitor, and thus be able to dissipate more heat to
the surrounding air, and therefore will have a higher RMS
current rating. The consequence of operating an electrolytic
capacitor above the RMS current rating is a shortened
operating life. In order to assure maximum capacitor
operating lifetime, the capacitor’s RMS ripple current rating
should be:
Catch Diode
Locate the Catch Diode Close to the LM2575
The LM2575 is a step−down buck converter; it requires a
fast diode to provide a return path for the inductor current
when the switch turns off. This diode must be located close
to the LM2575 using short leads and short printed circuit
traces to avoid EMI problems.
Use a Schottky or a Soft Switching
Ultra−Fast Recovery Diode
Irms > 1.2 x d x ILoad
Since the rectifier diodes are very significant source of
losses within switching power supplies, choosing the
rectifier that best fits into the converter design is an
important process. Schottky diodes provide the best
performance because of their fast switching speed and low
forward voltage drop.
They provide the best efficiency especially in low output
voltage applications (5.0 V and lower). Another choice
could be Fast−Recovery, or Ultra−Fast Recovery diodes. It
has to be noted, that some types of these diodes with an
abrupt turnoff characteristic may cause instability or EMI
troubles.
A fast−recovery diode with soft recovery characteristics
can better fulfill a quality, low noise design requirements.
Table 4 provides a list of suitable diodes for the LM2575
regulator. Standard 50/60 Hz rectifier diodes such as the
1N4001 series or 1N5400 series are NOT suitable.
where d is the duty cycle, for a buck regulator
V
t
d + on + out
T
V
in
|V out|
t on
and d +
+
for a buck*boost regulator.
T
|V out| ) V
in
Output Capacitor (Cout)
For low output ripple voltage and good stability, low ESR
output capacitors are recommended. An output capacitor
has two main functions: it filters the output and provides
regulator loop stability. The ESR of the output capacitor and
the peak−to−peak value of the inductor ripple current are the
main factors contributing to the output ripple voltage value.
Standard aluminium electrolytics could be adequate for
some applications but for quality design low ESR types are
recommended.
An aluminium electrolytic capacitor’s ESR value is
related to many factors such as the capacitance value, the
voltage rating, the physical size and the type of construction.
In most cases, the higher voltage electrolytic capacitors have
lower ESR value. Often capacitors with much higher
voltage ratings may be needed to provide low ESR values
that are required for low output ripple voltage.
Inductor
The magnetic components are the cornerstone of all
switching power supply designs. The style of the core and
the winding technique used in the magnetic component’s
design has a great influence on the reliability of the overall
power supply.
Using an improper or poorly designed inductor can cause
high voltage spikes generated by the rate of transitions in
current within the switching power supply, and the
possibility of core saturation can arise during an abnormal
operational mode. Voltage spikes can cause the
semiconductors to enter avalanche breakdown and the part
can instantly fail if enough energy is applied. It can also
The Output Capacitor Requires an ESR Value
That Has an Upper and Lower Limit
As mentioned above, a low ESR value is needed for low
output ripple voltage, typically 1% to 2% of the output
voltage. But if the selected capacitor’s ESR is extremely low
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LM2575
cause significant RFI (Radio Frequency Interference) and
EMI (Electro−Magnetic Interference) problems.
the physical volume the inductor must fit within, and the
amount of EMI (Electro−Magnetic Interference) shielding
that the core must provide. The inductor selection guide
covers different styles of inductors, such as pot core, E−core,
toroid and bobbin core, as well as different core materials
such as ferrites and powdered iron from different
manufacturers.
For high quality design regulators the toroid core seems to
be the best choice. Since the magnetic flux is completely
contained within the core, it generates less EMI, reducing
noise problems in sensitive circuits. The least expensive is
the bobbin core type, which consists of wire wound on a
ferrite rod core. This type of inductor generates more EMI
due to the fact that its core is open, and the magnetic flux is
not completely contained within the core.
When multiple switching regulators are located on the
same printed circuit board, open core magnetics can cause
interference between two or more of the regulator circuits,
especially at high currents due to mutual coupling. A toroid,
pot core or E−core (closed magnetic structure) should be
used in such applications.
Continuous and Discontinuous Mode of Operation
Do Not Operate an Inductor Beyond its
Maximum Rated Current
Exceeding an inductor’s maximum current rating may
cause the inductor to overheat because of the copper wire
losses, or the core may saturate. Core saturation occurs when
the flux density is too high and consequently the cross
sectional area of the core can no longer support additional
lines of magnetic flux.
This causes the permeability of the core to drop, the
inductance value decreases rapidly and the inductor begins
to look mainly resistive. It has only the dc resistance of the
winding. This can cause the switch current to rise very
rapidly and force the LM2575 internal switch into
cycle−by−cycle current limit, thus reducing the dc output
load current. This can also result in overheating of the
inductor and/or the LM2575. Different inductor types have
different saturation characteristics, and this should be kept
in mind when selecting an inductor.
POWER SWITCH
CURRENT (A)
1.0
0
1.0
0.1
0
0
INDUCTOR
CURRENT (A)
INDUCTOR
CURRENT (A)
POWER SWITCH
CURRENT (A)
The LM2575 step−down converter can operate in both the
continuous and the discontinuous modes of operation. The
regulator works in the continuous mode when loads are
relatively heavy, the current flows through the inductor
continuously and never falls to zero. Under light load
conditions, the circuit will be forced to the discontinuous
mode when inductor current falls to zero for certain period
of time (see Figure 22 and Figure 23). Each mode has
distinctively different operating characteristics, which can
affect the regulator performance and requirements. In many
cases the preferred mode of operation is the continuous
mode. It offers greater output power, lower peak currents in
the switch, inductor and diode, and can have a lower output
ripple voltage. On the other hand it does require larger
inductor values to keep the inductor current flowing
continuously, especially at low output load currents and/or
high input voltages.
To simplify the inductor selection process, an inductor
selection guide for the LM2575 regulator was added to this
data sheet (Figures 17 through 21). This guide assumes that
the regulator is operating in the continuous mode, and
selects an inductor that will allow a peak−to−peak inductor
ripple current to be a certain percentage of the maximum
design load current. This percentage is allowed to change as
different design load currents are selected. For light loads
(less than approximately 200 mA) it may be desirable to
operate the regulator in the discontinuous mode, because the
inductor value and size can be kept relatively low.
Consequently, the percentage of inductor peak−to−peak
current increases. This discontinuous mode of operation is
perfectly acceptable for this type of switching converter.
Any buck regulator will be forced to enter discontinuous
mode if the load current is light enough.
HORIZONTAL TIME BASE: 5.0 ms/DIV
Figure 22. Continuous Mode Switching
Current Waveforms
0.1
0
HORIZONTAL TIME BASE: 5.0 ms/DIV
Figure 23. Discontinuous Mode Switching
Current Waveforms
Selecting the Right Inductor Style
Some important considerations when selecting a core type
are core material, cost, the output power of the power supply,
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LM2575
GENERAL RECOMMENDATIONS
Output Voltage Ripple and Transients
Source of the Output Ripple
Heatsinking and Thermal Considerations
The Through−Hole Package TO−220
Since the LM2575 is a switch mode power supply
regulator, its output voltage, if left unfiltered, will contain a
sawtooth ripple voltage at the switching frequency. The
output ripple voltage value ranges from 0.5% to 3% of the
output voltage. It is caused mainly by the inductor sawtooth
ripple current multiplied by the ESR of the output capacitor.
The LM2575 is available in two packages, a 5−pin
TO−220(T, TV) and a 5−pin surface mount D2PAK(D2T).
There are many applications that require no heatsink to keep
the LM2575 junction temperature within the allowed
operating range. The TO−220 package can be used without
a heatsink for ambient temperatures up to approximately
50°C (depending on the output voltage and load current).
Higher ambient temperatures require some heatsinking,
either to the printed circuit (PC) board or an external
heatsink.
Short Voltage Spikes and How to Reduce Them
The regulator output voltage may also contain short
voltage spikes at the peaks of the sawtooth waveform (see
Figure 24). These voltage spikes are present because of the
fast switching action of the output switch, and the parasitic
inductance of the output filter capacitor. There are some
other important factors such as wiring inductance, stray
capacitance, as well as the scope probe used to evaluate these
transients, all these contribute to the amplitude of these
spikes. To minimize these voltage spikes, low inductance
capacitors should be used, and their lead lengths must be
kept short. The importance of quality printed circuit board
layout design should also be highlighted.
The Surface Mount Package D 2PAK and its
Heatsinking
The other type of package, the surface mount D2PAK, is
designed to be soldered to the copper on the PC board. The
copper and the board are the heatsink for this package and
the other heat producing components, such as the catch
diode and inductor. The PC board copper area that the
package is soldered to should be at least 0.4 in2 (or 100 mm2)
and ideally should have 2 or more square inches (1300 mm2)
of 0.0028 inch copper. Additional increasing of copper area
beyond approximately 3.0 in2 (2000 mm2) will not improve
heat dissipation significantly. If further thermal
improvements are needed, double sided or multilayer PC
boards with large copper areas should be considered.
Voltage spikes caused by switching action of the output
switch and the parasitic inductance of the output capacitor
UNFILTERED
OUTPUT
VOLTAGE
Thermal Analysis and Design
The following procedure must be performed to determine
whether or not a heatsink will be required. First determine:
1. PD(max) maximum regulator power dissipation in
the application.
2. TA(max) maximum ambient temperature in the
application.
3. TJ(max)
maximum allowed junction temperature
(125°C for the LM2575). For a conservative
design, the maximum junction temperature
should not exceed 110°C to assure safe
operation. For every additional 10°C
temperature rise that the junction must
withstand, the estimated operating lifetime
of the component is halved.
4. RqJC
package thermal resistance junction−case.
5. RqJA
package thermal resistance junction−ambient.
(Refer to Absolute Maximum Ratings in this data sheet or
RqJC and RqJA values).
VERTICAL
RESOLUTION:
20 mV/DIV
FILTERED
OUTPUT
VOLTAGE
HORIZONTAL TIME BASE: 10 ms/DIV
Figure 24. Output Ripple Voltage Waveforms
Minimizing the Output Ripple
In order to minimize the output ripple voltage it is possible
to enlarge the inductance value of the inductor L1 and/or to
use a larger value output capacitor. There is also another way
to smooth the output by means of an additional LC filter
(20 mH, 100 mF), that can be added to the output (see
Figure 33) to further reduce the amount of output ripple and
transients. With such a filter it is possible to reduce the
output ripple voltage transients 10 times or more. Figure 24
shows the difference between filtered and unfiltered output
waveforms of the regulator shown in Figure 33.
The upper waveform is from the normal unfiltered output
of the converter, while the lower waveform shows the output
ripple voltage filtered by an additional LC filter.
The following formula is to calculate the total power
dissipated by the LM2575:
PD = (Vin x IQ) + d x ILoad x Vsat
where d is the duty cycle and for buck converter
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17
LM2575
Unregulated
DC Input
12 V to 25 V
V
t
d + on + O ,
T
V
in
IQ
Vin
VO
ILoad
(quiescent current) and Vsat can be found in the
LM2575 data sheet,
is minimum input voltage applied,
is the regulator output voltage,
is the load current.
Cin
100 mF
/50 V
Feedback
4
+Vin
LM2575−12
1
3
GND
5
Output
2
ON/OFF
D1
1N5819
Cout
1800 mF
/16 V
Regulated
Output
−12 V @ 0.35 A
The dynamic switching losses during turn−on and
turn−off can be neglected if proper type catch diode is used.
Figure 25. Inverting Buck−Boost Regulator Using the
LM2575−12 Develops −12 V @ 0.35 A
Packages Not on a Heatsink (Free−Standing)
For a free−standing application when no heatsink is used,
the junction temperature can be determined by the following
expression:
ADDITIONAL APPLICATIONS
Inverting Regulator
TJ = (RqJA) (PD) + TA
An inverting buck−boost regulator using the LM2575−12
is shown in Figure 25. This circuit converts a positive input
voltage to a negative output voltage with a common ground
by bootstrapping the regulators ground to the negative
output voltage. By grounding the feedback pin, the regulator
senses the inverted output voltage and regulates it.
In this example the LM2575−12 is used to generate a
−12 V output. The maximum input voltage in this case
cannot exceed +28 V because the maximum voltage
appearing across the regulator is the absolute sum of the
input and output voltages and this must be limited to a
maximum of 40 V.
This circuit configuration is able to deliver approximately
0.35 A to the output when the input voltage is 12 V or higher.
At lighter loads the minimum input voltage required drops
to approximately 4.7 V, because the buck−boost regulator
topology can produce an output voltage that, in its absolute
value, is either greater or less than the input voltage.
Since the switch currents in this buck−boost configuration
are higher than in the standard buck converter topology, the
available output current is lower.
This type of buck−boost inverting regulator can also
require a larger amount of startup input current, even for
light loads. This may overload an input power source with
a current limit less than 1.5 A.
Such an amount of input startup current is needed for at
least 2.0 ms or more. The actual time depends on the output
voltage and size of the output capacitor.
Because of the relatively high startup currents required by
this inverting regulator topology, the use of a delayed startup
or an undervoltage lockout circuit is recommended.
where (RqJA)(PD) represents the junction temperature rise
caused by the dissipated power and TA is the maximum
ambient temperature.
Packages on a Heatsink
If the actual operating junction temperature is greater than
the selected safe operating junction temperature determined
in step 3, than a heatsink is required. The junction
temperature will be calculated as follows:
TJ = PD (RqJA + RqCS + RqSA) + TA
where
L1
100 mH
RqJC is the thermal resistance junction−case,
RqCS is the thermal resistance case−heatsink,
RqSA is the thermal resistance heatsink−ambient.
If the actual operating temperature is greater than the
selected safe operating junction temperature, then a larger
heatsink is required.
Some Aspects That can Influence Thermal Design
It should be noted that the package thermal resistance and
the junction temperature rise numbers are all approximate,
and there are many factors that will affect these numbers,
such as PC board size, shape, thickness, physical position,
location, board temperature, as well as whether the
surrounding air is moving or still.
Other factors are trace width, total printed circuit copper
area, copper thickness, single− or double−sided, multilayer
board, the amount of solder on the board or even color of the
traces.
The size, quantity and spacing of other components on
the board can also influence its effectiveness to dissipate
the heat.
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18
LM2575
Using a delayed startup arrangement, the input capacitor
can charge up to a higher voltage before the switch−mode
regulator begins to operate.
The high input current needed for startup is now partially
supplied by the input capacitor Cin.
Shutdown
Input
5.0 V
The inverting regulator operates in a different manner
than the buck converter and so a different design procedure
has to be used to select the inductor L1 or the output
capacitor Cout.
The output capacitor values must be larger than is
normally required for buck converter designs. Low input
voltages or high output currents require a large value output
capacitor (in the range of thousands of mF).
The recommended range of inductor values for the
inverting converter design is between 68 mH and 220 mH. To
select an inductor with an appropriate current rating, the
inductor peak current has to be calculated.
The following formula is used to obtain the peak inductor
current:
0
5
ON/OFF 3
GND
Off
R3
470
On
R2
47 k
MOC8101
NOTE: This picture does not show the complete circuit.
Figure 27. Inverting Buck−Boost Regulator Shut Down
Circuit Using an Optocoupler
With the inverting configuration, the use of the ON/OFF
pin requires some level shifting techniques. This is caused
by the fact, that the ground pin of the converter IC is no
longer at ground. Now, the ON/OFF pin threshold voltage
(1.4 V approximately) has to be related to the negative
output voltage level. There are many different possible shut
down methods, two of them are shown in Figures 27 and 28.
I
(V ) |V |)
O ) V in x t on
[ Load in
2L 1
V
in
|V |
O
where t on +
x 1 , and f osc + 52 kHz.
V ) |V | fosc
in
O
Under normal continuous inductor current operating
conditions, the worst case occurs when Vin is minimal.
Note that the voltage appearing across the regulator is the
absolute sum of the input and output voltage, and must not
exceed 40 V.
Cin
C1
100 mF
/50 V 0.1 mF
Cin
R1
100 mF 47 k
−Vout
peak
Unregulated
DC Input
12 V to 25 V
LM2575−XX
1
Design Recommendations:
I
+Vin
+Vin
+V
0
Off
Shutdown
Input
On
R2
5.6 k
+Vin
+Vin
1
Feedback
+Vin
1
LM2575−12
5
R1
47 k
ON/OFF 3
4
Output
LM2575−XX
Cin
100 mF
L1
100 mH
Q1
2N3906
2
GND
Cout
1800 mF
/16 V
D1
1N5819
R2
47 k
5
ON/OFF 3
R1
12 k
GND
−Vout
NOTE: This picture does not show the complete circuit.
Regulated
Output
−12 V @ 0.35 A
Figure 28. Inverting Buck−Boost Regulator Shut Down
Circuit Using a PNP Transistor
Figure 26. Inverting Buck−Boost
Regulator with Delayed Startup
Negative Boost Regulator
This example is a variation of the buck−boost topology
and is called a negative boost regulator. This regulator
experiences relatively high switch current, especially at low
input voltages. The internal switch current limiting results in
lower output load current capability.
The circuit in Figure 29 shows the negative boost
configuration. The input voltage in this application ranges
from −5.0 V to −12 V and provides a regulated −12 V output.
It has been already mentioned above, that in some
situations, the delayed startup or the undervoltage lockout
features could be very useful. A delayed startup circuit
applied to a buck−boost converter is shown in Figure 26.
Figure 32 in the “Undervoltage Lockout” section describes
an undervoltage lockout feature for the same converter
topology.
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19
LM2575
cause some problems by coupling the ripple into the
ON/OFF pin, the regulator could be switched periodically
on and off with the line (or double) frequency.
If the input voltage is greater than −12 V, the output will rise
above −12 V accordingly, but will not damage the regulator.
Cout
1000 mF
/16 V
4
+Vin
LM2575−12
1
Cin
100 mF
/50 V
Feedback
Output
3
GND
5
L1
Unregulated
DC Input
−Vin = −5.0 V to −12 V
150 mH
2
ON/OFF
+Vin
+Vin
D1
1N5817
LM2575−XX
1
C1
0.1 mF
Regulated
Output
Vout = −12 V
Cin
100 mF
5
ON/OFF 3
R1
47 k
GND
R2
47 k
Load Current from
200 mA for Vin = −5.2 V
to 500 mA for Vin = −7.0 V
NOTE: This picture does not show the complete circuit.
Figure 30. Delayed Startup Circuitry
Figure 29. Negative Boost Regulator
Undervoltage Lockout
Design Recommendations:
Some applications require the regulator to remain off until
the input voltage reaches a certain threshold level. Figure 31
shows an undervoltage lockout circuit applied to a buck
regulator. A version of this circuit for buck−boost converter
is shown in Figure 32. Resistor R3 pulls the ON/OFF pin
high and keeps the regulator off until the input voltage
reaches a predetermined threshold level, which is
determined by the following expression:
The same design rules as for the previous inverting
buck−boost converter can be applied. The output capacitor
Cout must be chosen larger than would be required for a
standard buck converter. Low input voltages or high output
currents require a large value output capacitor (in the range
of thousands of mF). The recommended range of inductor
values for the negative boost regulator is the same as for
inverting converter design.
Another important point is that these negative boost
converters cannot provide current limiting load protection in
the event of a short in the output so some other means, such
as a fuse, may be necessary to provide the load protection.
V
th
[V
Z1
ǒ
Ǔ
(Q1)
) 1 ) R2 V
R1 BE
+Vin
+Vin
Delayed Startup
1
There are some applications, like the inverting regulator
already mentioned above, which require a higher amount of
startup current. In such cases, if the input power source is
limited, this delayed startup feature becomes very useful.
To provide a time delay between the time the input voltage
is applied and the time when the output voltage comes up,
the circuit in Figure 30 can be used. As the input voltage is
applied, the capacitor C1 charges up, and the voltage across
the resistor R2 falls down. When the voltage on the ON/OFF
pin falls below the threshold value 1.4 V, the regulator starts
up. Resistor R1 is included to limit the maximum voltage
applied to the ON/OFF pin, reduces the power supply noise
sensitivity, and also limits the capacitor C1 discharge
current, but its use is not mandatory.
When a high 50 Hz or 60 Hz (100 Hz or 120 Hz
respectively) ripple voltage exists, a long delay time can
R2
10 k
Cin
100 mF 5
R3
47 k
LM2575−5.0
ON/OFF 3
GND
Z1
1N5242B
Q1
2N3904
R1
10 k
Vth ≈ 13 V
NOTE: This picture does not show the complete circuit.
Figure 31. Undervoltage Lockout Circuit for
Buck Converter
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20
LM2575
Adjustable Output, Low−Ripple Power Supply
+Vin
+Vin
1
R2
15 k
Cin
100 mF 5
R3
68 k
A 1.0 A output current capability power supply that
features an adjustable output voltage is shown in Figure 33.
This regulator delivers 1.0 A into 1.2 V to 35 V output.
The input voltage ranges from roughly 8.0 V to 40 V. In order
to achieve a 10 or more times reduction of output ripple, an
additional L−C filter is included in this circuit.
LM2575−5.0
ON/OFF 3
GND
Z1
1N5242B
Vth ≈ 13 V
Q1
2N3904
R1
15 k
Vout = −5.0 V
NOTE: This picture does not show the complete circuit.
Figure 32. Undervoltage Lockout Circuit for
Buck−Boost Converter
Feedback
4
+Vin
L1
150 mH
LM2575−Adj
1
Output
3
GND
5
L2
20 mH
2
ON/OFF
Cin
100 mF
/50 V
C1
100 mF
Cout
2200 mF
D1
1N5819
R1
1.1 k
Optional Output
Ripple Filter
Figure 33. Adjustable Power Supply with Low Ripple Voltage
JUNCTION-TO-AIR (° C/W)
R θ JA, THERMAL RESISTANCE
80
3.5
PD(max) for TA = 50°C
70
3.0
Free Air
Mounted
Vertically
60
ÎÎÎÎ
ÎÎÎÎ
ÎÎÎÎ
ÎÎÎÎ
2.0 oz. Copper
L
Minimum
Size Pad
50
2.5
2.0
L
40
1.5
RqJA
30
1.0
0
5.0
10
15
20
25
30
L, LENGTH OF COPPER (mm)
Figure 34. D2PAK Thermal Resistance and Maximum
Power Dissipation versus P.C.B. Copper Length
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21
Regulated
Output Voltage
1.2 V to 35 V @1.0 A
R2
50 k
PD, MAXIMUM POWER DISSIPATION (W)
Unregulated
DC Input
+
LM2575
THE LM2575−5.0 STEP−DOWN VOLTAGE REGULATOR WITH 5.0 V @ 1.0 A OUTPUT POWER
CAPABILITY. TYPICAL APPLICATION WITH THROUGH−HOLE PC BOARD LAYOUT
Feedback
4
+Vin
Unregulated
DC Input
+Vin = +7.0 V to +40 V
LM2575−5.0
1
Output
3
C1
100 mF
/50 V
GND
L1
330 mH
Regulated Output
+Vout1 = 5.0 V @ 1.0 A
2
ON/OFF
5
J1
D1
1N5819
GNDin
Cout
330 mF
/16 V
GNDout
C1
C2
D1
L1
−
−
−
−
100 mF, 50 V, Aluminium Electrolytic
330 mF, 16 V, Aluminium Electrolytic
1.0 A, 40 V, Schottky Rectifier, 1N5819
330 mH, Tech 39: 77 458 BV, Toroid Core, Through−Hole, Pin 3 = Start, Pin 7 = Finish
Figure 35. Schematic Diagram of the LM2575−5.0 Step−Down Converter
GNDin
C1
GNDout
U1 LM2575
J1
L1
D1
C2
DC−DC Converter
+Vin
+Vout1
NOTE: Not to scale.
NOTE: Not to scale.
Figure 36. Printed Circuit Board
Component Side
Figure 37. Printed Circuit Board
Copper Side
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22
LM2575
THE LM2575−ADJ STEP−DOWN VOLTAGE REGULATOR WITH 8.0 V @ 1.0 A OUTPUT POWER
CAPABILITY. TYPICAL APPLICATION WITH THROUGH−HOLE PC BOARD LAYOUT
Regulated
Output Unfiltered
Vout1 = 8.0 V @1.0 A
4
Unregulated
DC Input
Feedback
+Vin
LM2575−Adj
1
+Vin = +10 V to + 40 V
Output
3
GND
5
L1
330 mH
L2
25 mH
Vout2 = 8.0 V @1.0 A
2
ON/OFF
C1
100 mF
/50 V
R2
10 k
D1
1N5819
C2
330 mF
/16 V
−
−
−
−
−
−
−
−
C3
100 mF
/16 V
R1
1.8 k
V
C1
C2
C3
D1
L1
L2
R1
R2
Regulated
Output Filtered
ǒ
R2
out + V ref ) 1 ) R1
Ǔ
Vref = 1.23 V
100 mF, 50 V, Aluminium Electrolytic
R1 is between 1.0 k and 5.0 k
330 mF, 16 V, Aluminium Electrolytic
100 mF, 16 V, Aluminium Electrolytic
1.0 A, 40 V, Schottky Rectifier, 1N5819
330 mH, Tech 39: 77 458 BV, Toroid Core, Through−Hole, Pin 3 = Start, Pin 7 = Finish
25 mH, TDK: SFT52501, Toroid Core, Through−Hole
1.8 k
10 k
Figure 38. Schematic Diagram of the 8.0 V @ 1.0 V Step−Down Converter Using the LM2575−Adj
(An additional LC filter is included to achieve low output ripple voltage)
GNDin
GNDout
U1 LM2575
C3
C1
C2
L1
D1
J1
L2
+Vin
+Vout2
+Vout1
R2 R1
NOTE: Not to scale.
NOTE: Not to scale.
Figure 39. PC Board Component Side
Figure 40. PC Board Copper Side
References
•
•
•
•
National Semiconductor LM2575 Data Sheet and Application Note
National Semiconductor LM2595 Data Sheet and Application Note
Marty Brown “Practical Switching Power Supply Design”, Academic Press, Inc., San Diego 1990
Ray Ridley “High Frequency Magnetics Design”, Ridley Engineering, Inc. 1995
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23
LM2575
ORDERING INFORMATION
Device
Nominal
Output Voltage
Operating
Temperature Range
Package
LM2575TV−ADJ
TO−220 (Vertical Mount)
LM2575TV−ADJG
TO−220 (Vertical Mount)
(Pb−Free)
LM2575T−ADJ
TO−220 (Straight Lead)
LM2575T−ADJG
TO−220 (Straight Lead)
(Pb−Free)
LM2575D2T−ADJ
1.23 V to 37 V
TJ = −40° to +125°C
D2PAK (Surface Mount)
(Pb−Free)
LM2575D2T−ADJR4
D2PAK (Surface Mount)
LM2575D2T−ADJR4G
D2PAK (Surface Mount)
(Pb−Free)
LM2575TV−3.3
TO−220 (Vertical Mount)
LM2575TV−3.3G
TO−220 (Vertical Mount)
(Pb−Free)
LM2575T−3.3
TO−220 (Straight Lead)
LM2575T−3.3G
TO−220 (Straight Lead)
(Pb−Free)
3.3 V
TJ = −40° to +125°C
D2PAK (Surface Mount)
(Pb−Free)
LM2575D2T−3.3R4
D2PAK (Surface Mount)
LM2575D2T−3.3R4G
D2PAK (Surface Mount)
(Pb−Free)
LM2575TV−005
TO−220 (Vertical Mount)
LM2575TV−005G
TO−220 (Vertical Mount)
(Pb−Free)
LM2575T−005
TO−220 (Straight Lead)
LM2575T−005G
TO−220 (Straight Lead)
(Pb−Free)
5.0 V
TJ = −40° to +125°C
D2PAK (Surface Mount)
(Pb−Free)
LM2575D2T−5R4
D2PAK (Surface Mount)
LM2575D2T−5R4G
D2PAK (Surface Mount)
(Pb−Free)
LM2575TV−012
TO−220 (Vertical Mount)
LM2575TV−012G
TO−220 (Vertical Mount)
(Pb−Free)
LM2575T−012
TO−220 (Straight Lead)
LM2575T−012G
TO−220 (Straight Lead)
(Pb−Free)
12 V
TJ = −40° to +125°C
50 Units/Rail
800 Tape & Reel
50 Units/Rail
D2PAK (Surface Mount)
LM2575D2T−005G
LM2575D2T−012
800 Tape & Reel
D2PAK (Surface Mount)
LM2575D2T−3.3G
LM2575D2T−005
50 Units/Rail
D2PAK (Surface Mount)
LM2575D2T−ADJG
LM2575D2T−3.3
Shipping †
800 Tape & Reel
50 Units/Rail
D2PAK (Surface Mount)
LM2575D2T−012G
D2PAK (Surface Mount)
(Pb−Free)
LM2575D2T−12R4
D2PAK (Surface Mount)
LM2575D2T−12R4G
D2PAK (Surface Mount)
(Pb−Free)
800 Tape & Reel
†For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging
Specifications Brochure, BRD8011/D.
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24
LM2575
ORDERING INFORMATION
Nominal
Output Voltage
Device
Operating
Temperature Range
Shipping †
Package
LM2575TV−015
TO−220 (Vertical Mount)
LM2575TV−015G
TO−220 (Vertical Mount)
(Pb−Free)
LM2575T−015
TO−220 (Straight Lead)
LM2575T−015G
TO−220 (Straight Lead)
(Pb−Free)
15 V
LM2575D2T−015
TJ = −40° to +125°C
50 Units/Rail
D2PAK (Surface Mount)
LM2575D2T−015G
D2PAK (Surface Mount)
(Pb−Free)
LM2575D2T−15R4
D2PAK (Surface Mount)
LM2575D2T−15R4G
D2PAK (Surface Mount)
(Pb−Free)
800 Tape & Reel
†For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging
Specifications Brochure, BRD8011/D.
MARKING DIAGRAMS
TO−220
TV SUFFIX
CASE 314B
LM
2575T−xxx
AWLYWWG
TO−220
T SUFFIX
CASE 314D
LM
2575T−xxx
AWLYWWG
D2PAK
D2T SUFFIX
CASE 936A
LM
2575−xxx
AWLYWWG
1
1
5
5
1
5
xxx
A
WL
Y
WW
G
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25
= 3.3, 5.0, 12, 15, or ADJ
= Assembly Location
= Wafer Lot
= Year
= Work Week
= Pb−Free Package
LM2575
PACKAGE DIMENSIONS
TO−220
TV SUFFIX
CASE 314B−05
ISSUE L
Q
OPTIONAL
CHAMFER
E
A
U
K
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: INCH.
3. DIMENSION D DOES NOT INCLUDE
INTERCONNECT BAR (DAMBAR) PROTRUSION.
DIMENSION D INCLUDING PROTRUSION SHALL
NOT EXCEED 0.043 (1.092) MAXIMUM.
C
B
−P−
L
S
W
DIM
A
B
C
D
E
F
G
H
J
K
L
N
Q
S
U
V
W
V
F
J
5X
G
5X
0.24 (0.610)
M
T
H
D
0.10 (0.254)
M
T P
N
M
−T−
SEATING
PLANE
INCHES
MIN
MAX
0.572
0.613
0.390
0.415
0.170
0.180
0.025
0.038
0.048
0.055
0.850
0.935
0.067 BSC
0.166 BSC
0.015
0.025
0.900
1.100
0.320
0.365
0.320 BSC
0.140
0.153
−−−
0.620
0.468
0.505
−−−
0.735
0.090
0.110
MILLIMETERS
MIN
MAX
14.529 15.570
9.906 10.541
4.318
4.572
0.635
0.965
1.219
1.397
21.590 23.749
1.702 BSC
4.216 BSC
0.381
0.635
22.860 27.940
8.128
9.271
8.128 BSC
3.556
3.886
−−− 15.748
11.888 12.827
−−− 18.669
2.286
2.794
TO−220
T SUFFIX
CASE 314D−04
ISSUE F
−T−
B
−Q−
DETAIL A−A
B1
SEATING
PLANE
C
E
A
U
L
J
H
G
D
DIM
A
B
B1
C
D
E
G
H
J
K
L
Q
U
1234 5
K
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: INCH.
3. DIMENSION D DOES NOT INCLUDE
INTERCONNECT BAR (DAMBAR) PROTRUSION.
DIMENSION D INCLUDING PROTRUSION SHALL
NOT EXCEED 10.92 (0.043) MAXIMUM.
5 PL
0.356 (0.014)
M
T Q
M
B
B1
DETAIL A−A
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26
INCHES
MIN
MAX
0.572
0.613
0.390
0.415
0.375
0.415
0.170
0.180
0.025
0.038
0.048
0.055
0.067 BSC
0.087
0.112
0.015
0.025
0.977
1.045
0.320
0.365
0.140
0.153
0.105
0.117
MILLIMETERS
MIN
MAX
14.529 15.570
9.906 10.541
9.525 10.541
4.318
4.572
0.635
0.965
1.219
1.397
1.702 BSC
2.210
2.845
0.381
0.635
24.810 26.543
8.128
9.271
3.556
3.886
2.667
2.972
LM2575
PACKAGE DIMENSIONS
D2PAK
D2T SUFFIX
CASE 936A−02
ISSUE C
−T−
OPTIONAL
CHAMFER
A
E
B
V
H
1 2 3 4 5
M
D
0.010 (0.254)
U
S
K
M
T
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: INCH.
3. TAB CONTOUR OPTIONAL WITHIN DIMENSIONS A
AND K.
4. DIMENSIONS U AND V ESTABLISH A MINIMUM
MOUNTING SURFACE FOR TERMINAL 6.
5. DIMENSIONS A AND B DO NOT INCLUDE MOLD
FLASH OR GATE PROTRUSIONS. MOLD FLASH
AND GATE PROTRUSIONS NOT TO EXCEED 0.025
(0.635) MAXIMUM.
TERMINAL 6
L
G
DIM
A
B
C
D
E
G
H
K
L
M
N
P
R
S
U
V
P
N
R
C
SOLDERING FOOTPRINT*
8.38
0.33
1.702
0.067
10.66
0.42
3.05
0.12
16.02
0.63
SCALE 3:1
1.016
0.04
mm Ǔ
ǒinches
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
http://onsemi.com
27
INCHES
MIN
MAX
0.386
0.403
0.356
0.368
0.170
0.180
0.026
0.036
0.045
0.055
0.067 BSC
0.539
0.579
0.050 REF
0.000
0.010
0.088
0.102
0.018
0.026
0.058
0.078
5 _ REF
0.116 REF
0.200 MIN
0.250 MIN
MILLIMETERS
MIN
MAX
9.804
10.236
9.042
9.347
4.318
4.572
0.660
0.914
1.143
1.397
1.702 BSC
13.691
14.707
1.270 REF
0.000
0.254
2.235
2.591
0.457
0.660
1.473
1.981
5 _ REF
2.946 REF
5.080 MIN
6.350 MIN
LM2575
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LM2575/D