HP HCPL-7510-060E Isolated linear sensing ic Datasheet

Agilent HCPL-7510
Isolated Linear Sensing IC
Data Sheet
Description
The HCPL-7510 isolated linear
current sensing IC family is
designed for current sensing in
low-power electronic motor
drives. In a typical
implementation, motor current
flows through an external
resistor and the resulting
analog voltage drop is sensed
by the HCPL-7510. An output
voltage is created on the other
side of the HCPL-7510 optical
isolation barrier. This singleended output voltage is
proportional to the motor
current. Since common-mode
voltage swings of several
hundred volts in tens of
nanoseconds are common in
modern switching inverter
motor drives, the HCPL-7510
was designed to ignore very
high common-mode transient
slew rates (of at least 10 kV/
µs).
The high CMR capability of the
HCPL-7510 isolation amplifier
provides the precision and
stability needed to accurately
monitor motor current in high
noise motor control environments, providing for smoother
control (less “torque ripple”)
in various types of motor
control applications.
Functional Diagram
IDD1
IDD2
VDD1
1
VIN+
2
+
VIN–
3
–
GND1
4
SHIELD
8
VDD2
+
7
VOUT
–
6
VREF
5
GND2
The product can also be used
for general analog signal
isolation applications. For
general applications, we
recommend the HCPL-7510
(gain tolerance of ±5%). The
HCPL-7510 utilizes sigma delta
(S-D) analog-to-digital
converter technology to
delivery offset and gain
accuracy and stability over
time and temperature. This
performance is delivered in a
compact, auto-insert, 8-pin
DIP package that meets worldwide regulatory safety
standards. (A gull-wing
surface mount option #300 is
also available).
Features
• 15 kV/µs common-mode rejection
at Vcm = 1000 V
• Compact, auto-insertable 8-pin
DIP package
• 60 ppm/°C gain drift vs.
temperature
• –0.6 mV input offset voltage
• 8 µV/°C input offset voltage vs.
temperature
• 100 kHz bandwidth
• 0.06% nonlinearity, single-ended
amplifier output for low power
application.
• Worldwide safety approval:
UL 1577 (3750 Vrms/1 min.), CSA
and IEC/EN/DIN EN 60747-5-2
(Option 060 only)
Σ-∆
∆)
• Advanced sigma-delta (Σ
A/D converter technology
Applications
• Low-power inverter current
sensing
• Motor phase and rail current
sensing
• Switched mode power supply
signal isolation
• General purpose low-power
current sensing and monitoring
• General purpose analog signal
isolation
CAUTION: It is advised that normal static precautions be taken in handling and assembly of
this component to prevent damage and /or degradation which may be induced by ESD.
Ordering Information
Specify part number followed by option number (if desired).
Example:
HCPL-7510-XXXX
No option = Standard DIP package, 50 per tube.
300 = Gull Wing Surface Mount Option, 50 per tube.
500 = Tape and Reel Packaging Option.
060 = IEC/EN/DIN EN 60747-5-2 Option.
XXXE = Lead Free Option
Package Outline Drawings
HCPL-7510 Standard DIP Package
9.80 ± 0.25
(0.386 ± 0.010)
8
7
6
5
DATE CODE
A 7510
YYWW
1
1.19 (0.047) MAX.
2
3
4
7.62 ± 0.25
(0.300 ± 0.010)
1.78 (0.070) MAX.
6.35 ± 0.25
(0.250 ± 0.010)
3.56 ± 0.13
(0.140 ± 0.005)
4.70 (0.185) MAX.
0.51 (0.020) MIN.
2.92 (0.115) MIN.
1.080 ± 0.320
(0.043 ± 0.013)
0.65 (0.025) MAX.
2.54 ± 0.25
(0.100 ± 0.010)
DIMENSIONS IN MILLIMETERS AND (INCHES).
NOTE: FLOATING LEAD PROTUSION IS 0.5 mm (20 mils) MAX.
2
5 TYP.
0.20 (0.008)
0.33 (0.013)
HCPL-7510 Gull Wing Surface Mount Option 300 Outline Drawing
Land Pattern Recommendation
9.80 ± 0.25
(0.386 ± 0.010)
8
6
7
1.016 (0.040)
5
A 7510
6.350 ± 0.25
(0.250 ± 0.010)
YYWW
1
2
3
10.9 (0.430)
4
2.0 (0.080)
1.27 (0.050)
9.65 ± 0.25
(0.380 ± 0.010)
1.780
(0.070)
MAX.
1.19
(0.047)
MAX.
7.62 ± 0.25
(0.300 ± 0.010)
0.20 (0.008)
0.33 (0.013)
3.56 ± 0.13
(0.140 ± 0.005)
0.635 ± 0.25
(0.025 ± 0.010)
1.080 ± 0.320
(0.043 ± 0.013)
2.54
(0.100)
BSC
0.635 ± 0.130
(0.025 ± 0.005)
DIMENSIONS IN MILLIMETERS (INCHES).
TOLERANCES (UNLESS OTHERWISE SPECIFIED):
NOTE: FLOATING LEAD PROTUSION IS 0.5 mm (20 mils) MAX.
3
xx.xx = 0.01
xx.xxx = 0.005
12 NOM.
LEAD COPLANARITY
MAXIMUM: 0.102 (0.004)
Solder Reflow Temperature Profile
300
PREHEATING RATE 3˚C + 1˚C/–0.5˚C/SEC.
REFLOW HEATING RATE 2.5˚C ± 0.5˚C/SEC.
PEAK
TEMP.
240˚C
PEAK
TEMP.
230˚C
200
TEMPERATURE (˚C)
PEAK
TEMP.
245˚C
2.5˚C ± 0.5˚C/SEC.
30
SEC.
160˚C
150˚C
140˚C
SOLDERING
TIME
200˚C
30
SEC.
3˚C + 1˚C/–0.5˚C
100
PREHEATING TIME
150˚C, 90 + 30 SEC.
50 SEC.
TIGHT
TYPICAL
LOOSE
0
50
0
ROOM TEMPERATURE
100
150
TIME (SECONDS)
Recommended Pb-Free IR Profile
TEMPERATURE (˚C)
tp
Tp
217 ˚C
TL
Tsmax
Tsmin
TIME WITHIN 5 ˚C of ACTUAL
PEAK TEMPERATURE
20-40 SEC.
260 +0/-5 ˚C
RAMP-UP
3 ˚C/SEC. MAX.
150 - 200 ˚C
ts
PREHEAT
60 to 180 SEC.
RAMP-DOWN
6 ˚C/SEC. MAX.
tL
60 to 150 SEC.
25
t 25 ˚C to PEAK
TIME (SECONDS)
NOTES:
THE TIME FROM 25 ˚C to PEAK TEMPERATURE = 8 MINUTES MAX.
Tsmax = 200 ˚C, Tsmin = 150 ˚C
4
200
250
Regulatory Information
The HCPL-7510 has been approved by the following organizations:
IEC/EN/DIN EN 60747-5-2
Approved under:
IEC 60747-5-2:1997 + A1:2002
EN 60747-5-2:2001 + A1:2002
DIN EN 60747-5-2 (VDE 0884 Teil 2):2003-01.
UL
Approved under UL 1577, component recognition
program up to VISO = 3750 VRMS. File E55361.
CSA
Approved under CSA Component Acceptance
Notice #5, File CA 88324.
IEC/EN/DIN EN 60747-5-2 Insulation Characteristics[1]
Description
Symbol
Characteristic Unit
Installation classification per DIN EN 0110-1/1997-04, Table 1
for rated mains voltage - 150 Vrms
for rated mains voltage - 300 Vrms
for rated mains voltage - 600 Vrms
I – IV
I – III
I – II
Climatic Classification
55/100/21
Pollution Degree (DIN EN 0110-1/1997-04)
2
Maximum Working Insulation Voltage
VIORM
891
Vpeak
Input to Output Test Voltage, Method b[2]
VIORM x 1.875 = VPR, 100% production test with tm = 1 sec, partial discharge <5 pC
VPR
1670
Vpeak
Input to Output Test Voltage, Method a[2]
VIORM x 1.5 = VPR, type and sample test, tm = 60 sec, partial discharge <5 pC
VPR
1336
Vpeak
Highest Allowable Overvoltage (transient overvoltage tini = 10 sec)
VIOTM
6000
Vpeak
Safety-limiting values – maximum values allowed in the event of a failure.
Case Temperature
Input Current[3]
Output Power[3]
TS
175
IS, INPUT
400
PS, OUTPUT 600
Insulation Resistance at TS, VIO = 500 V
RS
Ω
>109
800
OUTPUT POWER – PS, INPUT CURRENT – IS
Notes:
1. Insulation characteristics are guaranteed only within the safety maximum ratings which must be
ensured by protective circuits within the application. Surface Mount Classifications is Class A in
accordance with CECC00802.
2. Refer to the optocoupler section of the Isolation and Control Components Designer’s Catalog,
under Product Safety Regulations section,
(IEC/EN/DIN EN 60747-5-2) for a detailed description of Method a and Method b partial
discharge test profiles.
3. Refer to the following figure for dependence of PS and IS on ambient temperature.
°C
mA
mW
PS (mW)
IS (mA)
700
600
500
400
300
200
100
0
0
25
50
75
100 125 150 175 200
TS – CASE TEMPERATURE – C
5
Insulation and Safety Related Specifications
Parameter
Symbol
Value
Unit
Conditions
Minimum External Air Gap
(clearance)
L(101)
7.4
mm
Measured from input terminals to output terminals,
shortest distance through air.
Minimum External Tracking
(creepage)
L(102)
8.0
mm
Measured from input terminals to output terminals,
shortest distance path along body.
0.5
mm
Through insulation distance conductor to conductor,
usually the straight line distance thickness between the
emitter and detector.
>175
V
DIN IEC 112 Part 1
Minimum Internal Plastic Gap
(internal clearance)
Tracking Resistance
(comparative tracking index)
CTI
Isolation Group
IIIa
Material Group (DIN EN 0110-1/1997-04)
Absolute Maximum Ratings
Parameter
Symbol
Min.
Max.
Units
Storage Temperature
TS
–55
125
°C
Operating Temperature
TA
–40
100
°C
Supply Voltage
VDD1_max, VDD1_max
0
6
V
Steady-State Input Voltage
VIN+, VIN-
–2.0
VDD1 + 0.5-
V
Two Second Transient Input Voltage
VIN+, VIN-
–6.0
VDD1 + 0.5-
V
Output Voltage
VOUT
–0.5
VDD2 + 0.5-
V
Reference Input Voltage
VREF
0.0
VDD2 + 0.5-
V
Reference Input Current
IREF
20-
mA
Lead Solder Temperature
260°C for 10 sec., 1.6 mm below seating plane
Solder Reflow Temperature Profile
See Package Outline Drawings section
Note
Recommended Operating Conditions
Parameter
Symbol
Min.
Max.
Units
Operating Temperature
TA
–40
85
°C
Supply Voltage
VDD1, VDD2
4.5
5.5
V
Input Voltage (accurate and linear)
VIN+, VIN-
–200
200
mV
Input Voltage (functional)
VIN+, VIN-
–2.0
2.0
V
Reference Input Voltage
VREF
4.0
VDD2
V
6
Note
Electrical Specifications (DC)
Unless otherwise noted, all typicals and figures are at the nominal operation conditions of VIN+ = 0 V, VIN- = 0 V,
VREF = 4.0 V, VDD1 = VDD2 = 5.0 V and TA = 25°C; all Minimum/Maximum specifications are within the Recommended
Operating Conditions.
Parameter
Symbol
Min.
Typ.
Max.
Units
Test
Conditions
Fig.
Note
Input Offset Voltage
VOS
–6
–1
6
mV
VIN+ = 0 V
6
1
Magnitude of Input Offset
Change vs. Temperature
∆Vos/∆T
8
20
µV/°C
Gain
G
VREF/0.512
+ 3%
V/V
-0.2 V < VIN+ 8
< 0.2 V
TA = 25°C
Magnitude of Gain Change
vs. Temperature
∆G/∆T
60
300
ppm/°C
-0.2 V < VIN+ 9
< 0.2 V
VOUT 200 mV Nonlinearity
NL200
0.06
0.55
%
-0.2 V < VIN+ 10
< 0.2 V
Magnitude of VOUT 200 mV
Nonlinearity Change
vs. Temperature
|dNL200/dT|
0.0004
%/°C
-0.2 V < VIN+ 11
< 0.2 V
VOUT 100 mV Nonlinearity
NL100
0.04
0.4
%
-0.1 V < VIN+
< 0.1 V
Input Supply Current
IDD1
11.7
16
mA
1,2,3
Output Supply Current
IDD2
9.9
16
mA
1,2,3
Reference Voltage Input
Current
IREF
0.26
1
mA
Input Current
IIN+
–0.6
5
µA
Magnitude of Input Bias
Current vs. Termperature
Coefficient
|dIIN/dT|
0.45
nA/°C
Maximum Input Voltage
before VOUT Clipping
|VIN+|MAX
256
mV
Equivalent Input Impedance
RIN
700
kΩ
VOUT Output Impedance
ROUT
15
Ω
Input DC Common-Mode
Rejection Ratio
CMRRIN
63
dB
7
VREF/0.512
– 3%
7
VIN+ = 0 V
2
3,4
3,5
4
5
7
Switching Specifications (AC)
Over recommended operating conditions unless otherwise specified.
Parameter
Symbol Min.
Typ.
Max.
Units
Test Conditions
Fig. Note
VIN to VOUT Signal Delay (50 – 10%)
tPD10
2.2
4
µs
VIN+ = 0 mV to 200 mV step
13
VIN to VOUT Signal Delay (50 – 50%)
tPD50
3.4
5
µs
VIN to VOUT Signal Delay (50 – 90%)
tPD90
5.2
9.9
µs
VOUT Rise Time (10 – 90%)
tR
3.0
7
µs
VOUT Fall Time (10 – 90%)
tF
3.2
7
µs
VOUT Bandwidth (-3 dB)
BW
VIN+ = 200 mVpk-pk
14
VOUT Noise
NOUT
Common Mode Transient
Immunity
CMTI
50
10
100
kHz
31.5
mVrms VIN+ = 0 V
15
kV/µs
TA = 25°C, VCM = 1000 V
15
Package Characteristics
Parameter
Symbol
Min.
Input-Output Momentary
Withstand Voltage
VISO
3750
Input-Output Resistance
RI-O
Input-Output Capacitance
CI-O
Typ.
Max.
Units
Test Conditions
Vrms
TA = 25°C, RH < 50%
>109
Ω
VI-O = 500 V
1.4
pF
Freq = 1 MHz
Fig.
Note
6
Notes:
General Note: Typical values were taken from a sample of nominal units operating at nominal conditions (VDD1 = VDD2 = 5 V, VREF = 4.0 V, Temperature =
25°C) unless otherwise stated. Nominal plots shown from Figure 1 to 11 represented the drift of these nominal units from their nominal operating
conditions.
1. Input Offset Voltage is defined as the DC Input Voltage required to obtain an output voltage of VREF/2.
2. Gain is defined as the slope of the best-fit line of the output voltage vs. the differential input voltage (VIN+ - VIN-) over the specified input range. Gain
is derived from VREF/512 mV; e.g. VREF = 5.0, gain will be 9.77 V/V.
3. Nonlinearity is defined as half of the peak-to-peak output deviation from the best-fit gain line, expressed as a percentage of the full-scale output
voltage range.
4. NL200 is the nonlinearity specified over an input voltage range of ±200 mV.
5. NL100 is the nonlinearity specified over an input voltage range of ±100 mV.
6. In accordance with UL1577, each optocoupler is proof tested by applying an insulation test voltage •4500 Vrms for 1 second (leakage detection current
limit, I I-O < 5 µA). This test is performed before the 100% production test for the partial discharge (method b) shown in
IEC/EN/DIN EN 60747-5-2 Insulation Characteristic Table, if applicable.
7. CMRR is defined as the ratio of the differential signal gain (signal applied differentially between pins 2 and 3) to the common-mode gain (input pins
tied together and the signal applied to both inputs at the same time), expressed in dB.
8
11
10
IDD1
9
IDD2
8
4.5
4.7
5.1
4.9
5.3
11.0
12.0
10.5
11.0
10.0
9.5
9.0
8.5
IDD1
8.0
IDD2
7.5
5.5
0
-20
40
20
60
80
8.0
7.0
IDD2
0
3.5
2.0
-0.4
-0.6
-0.8
-1.0
-1.2
DVOS – INPUT OFFSET CHANGE – µV
2.5
-0.2
3.0
2.5
2.0
1.5
1.0
0.5
-0.1
0
0.1
0.2
0
-0.3
0.3
VIN – INPUT VOLTAGE – V
0
-0.1
0.1
0.2
0.5
0
-0.5
-1.0
-1.5
VDD1
VDD2
1.0
0
-0.5
-1.0
-1.5
4.7
4.9
5.3
5.1
5.5
Figure 6. Input offset change vs. supply
voltage.
0.7
0.015
VDD1
0.6
VDD2
0.5
∆GAIN – GAIN CHANGE – %
∆GAIN – GAIN CHANGE – %
1.0
0.3
VDD – SUPPLY VOLTAGE – V
0.020
TYPICAL
MAXIMUM
1.5
0.2
0.5
-2.0
4.5
0.3
Figure 5. Output voltage vs. input voltage.
2.0
0.1
0
1.5
VIN – INPUT VOLTAGE – V
Figure 4. Input current vs. input voltage.
-2.0
-40
-0.2
-0.1
Figure 3. Supply current vs. input voltage.
4.0
-0.2
-0.2
VIN – INPUT VOLTAGE – V
0.2
-1.4
-0.3
IDD1
6.0
4.0
-0.3
100
Figure 2. Supply current vs. temperature.
VO – OUTPUT VOLTAGE – V
IIN – INPUT CURRENT – µA
9.0
TA – TEMPERATURE – C
Figure 1. Supply current vs. supply voltage.
∆VOS – INPUT OFFSET CHANGE – mV
10.0
5.0
7.0
-40
VDD – SUPPLY VOLTAGE – V
0.010
0.005
0
-0.005
0.4
0.3
0.2
0.1
0
-0.1
-0.2
-20
0
20
40
60
80
100
TA – TEMPERATURE – C
Figure 7. Input offset change vs. temperature.
9
IDD – SUPPLY CURRENT – mA
12
IDD – SUPPLY CURRENT – mA
IDD – SUPPLY CURRENT – mA
13
-0.010
4.5
4.7
4.9
5.1
5.3
VDD – SUPPLY VOLTAGE – V
Figure 8. Gain change vs. supply voltage.
5.5
-0.3
-40
-20
0
20
40
60
80
TA – TEMPERATURE – C
Figure 9. Gain change vs. temperature.
100
0.09
0.048
NL – NONLINEARITY – %
NL – NONLINEARITY – %
0.050
0.046
0.044
VDD1
0.042
0.08
0.07
0.06
VDD2
4.7
4.9
5.1
5.3
0.05
-40
5.5
-20
Figure 10. Nonlinearity vs. supply voltage.
60
40
VDD2
1
80
100
6
8
0.1 µF
0.1 µF
2
7
VOUT
HCPL-7510
6
3
0.1 µF
20
Figure 11. Nonlinearity vs. temperature.
VDD1
VIN
0
TA – TEMPERATURE – C
VDD – SUPPLY VOLTAGE – V
4
5
VREF
TPD – PROPAGATION DELAY – µs
0.040
4.5
5
4
3
2
Tp5010
Tp5050
Tp5090
Trise
1
0
-40
-20
0
20
40
60
80
100
TA – TEMPERATURE – C
Figure 12. Propagation delay test circuit.
Figure 13. Propagation delay vs. temperature.
VDD2
78L05
IN OUT
0.1
µF
1
0.1
µF
0
1
8
0.1 µF
7
2
GAIN – dB
-1
HCPL-7510
9V
-2
3
6
4
5
-3
-4
-5
-6
0.1
PULSE GEN.
1.0
10.0
100.0
1000.0
–
+
FREQUENCY – kHz
VCM
Figure 14. Bandwidth.
10
Figure 15. CMTI test circuit.
VOUT
VREF
Application Information
Power Supplies and Bypassing
The recommended supply
connections are shown in
Figure 16. A floating power
supply (which in many
applications could be the same
supply that is used to drive
the high-side power transistor)
is regulated to 5 V using a
simple zener diode (D1); the
value of resistor R4 should be
chosen to supply sufficient
current from the existing
floating supply. The voltage
from the current sensing
resistor (Rsense) is applied to
the input of the HCPL-7510
through an RC anti-aliasing
filter (R2 and C2). Although
the application circuit is
relatively simple, a few
recommendations should be
followed to ensure optimal
performance.
The power supply for the
HCPL-7510 is most often
obtained from the same supply
used to power the power
transistor gate drive circuit. If
a dedicated supply is required,
in many cases it is possible to
add an additional winding on
an existing transformer.
HV+
Otherwise, some sort of simple
isolated supply can be used,
such as a line powered
transformer or a highfrequency DC-DC converter.
An inexpensive 78L05 threeterminal regulator can also be
used to reduce the floating
supply voltage to 5 V. To help
attenuate high- frequency
power supply noise or ripple,
a resistor or inductor can be
used in series with the input
of the regulator to form a
low-pass filter with the
regulator’s input bypass
capacitor.
+
FLOATING
POSITIVE
SUPPLY
GATE DRIVE
CIRCUIT
-
R4
R2
MOTOR
D1
5.1 V
39 Ω
+ R1 -
1 VDD1
2 VIN+
C2
0.01 µF
3 VIN4 GND1
RSENSE
HVFigure 16. Recommended supply and sense resistor connections.
11
C1
0.1 µF
HCPL-7510
As shown in Figure 17, 0.1 µF
bypass capacitors (C1, C2)
should be located as close as
possible to the pins of the
HCPL-7510. The bypass
capacitors are required
because of the high-speed
digital nature of the signals
inside the HCPL-7510. A 0.01
µF bypass capacitor (C2) is
also recommended at the
input due to the switchedcapacitor nature of the input
circuit. The input bypass
capacitor also forms part of
the anti-aliasing filter, which
is recommended to prevent
high frequency noise from
aliasing down to lower
frequencies and interfering
with the input signal. The
input filter also performs an
important reliability function—it
reduces transient spikes from
ESD events flowing through the
current sensing resistor.
PC Board Layout
The design of the printed
circuit board (PCB) should
follow good layout practices,
such as keeping bypass
capacitors close to the supply
pins, keeping output signals
away from input signals, the
use of ground and power
planes, etc. In addition, the
layout of the PCB can also
affect the isolation transient
immunity (CMTI) of the
HCPL-7510, due primarily to
stray capacitive coupling
between the input and the
output circuits. To obtain
optimal CMTI performance, the
layout of the PC board should
minimize any stray coupling
by maintaining the maximum
possible distance between the
input and output sides of the
circuit and ensuring that any
ground or power plane on the
PC board does not pass
directly below or extend much
wider than the body of the
HCPL-7510.
FLOATING
POSITIVE
SUPPLY
HV+
GATE DRIVE
CIRCUIT
U1
78L05
IN
C1
0.1 µF
OUT
C2
0.1 µF
R5
68 Ω
MOTOR
+ R1 -
C3
0.01 µF
µC
+5 V
1 VDD1
VDD2 8
2 VIN+
VOUT 7
3 VIN-
VREF 6
4 GND1
A/D
C4
C5
GND2 5
HCPL-7510
C6 = 150 pF
C4 = C5 = 0.1 µF
Figure 17. Recommended HCPL-7510 application circuit.
12
VREF
GND
RSENSE
HV-
C6
Current Sensing Resistors
The current sensing resistor
should have low resistance (to
minimize power dissipation),
low inductance (to minimize
di/dt induced voltage spikes
which could adversely affect
operation), and reasonable
tolerance (to maintain overall
circuit accuracy). Choosing a
particular value for the
resistor is usually a
compromise between
minimizing power dissipation
and maximizing accuracy.
Smaller sense resistance
decreases power dissipation,
while larger sense resistance
can improve circuit accuracy
by utilizing the full input
range of the HCPL-7510.
The first step in selecting a
sense resistor is determining
how much current the resistor
will be sensing. The graph in
Figure 18 shows the RMS
current in each phase of a
three-phase induction motor
as a function of average motor
output power (in horsepower,
hp) and motor drive supply
voltage. The maximum value
of the sense resistor is
determined by the current
being measured and the
maximum recommended input
voltage of the isolation
amplifier. The maximum sense
resistance can be calculated by
taking the maximum
recommended input voltage
and dividing by the peak
current that the sense resistor
should see during normal
operation. For example, if a
motor will have a maximum
RMS current of 10 A and can
experience up to 50%
overloads during normal
operation, then the peak
current is 21.1 A (=10 x 1.414
x 1.5). Assuming a maximum
input voltage of 200 mV, the
maximum value of sense
13
resistance in this case would
be about 10 mΩ. The
maximum average power
dissipation in the sense
resistor can also be easily
calculated by multiplying the
sense resistance times the
square of the maximum RMS
current, which is about 1 W in
the previous example. If the
power dissipation in the sense
resistor is too high, the
resistance can be decreased
below the maximum value to
decrease power dissipation.
The minimum value of the
sense resistor is limited by
precision and accuracy
requirements of the design. As
the resistance value is
reduced, the output voltage
across the resistor is also
reduced, which means that the
offset and noise, which are
fixed, become a larger
percentage of the signal
amplitude. The selected value
of the sense resistor will fall
somewhere between the
minimum and maximum
values, depending on the
particular requirements of a
specific design.
When sensing currents large
enough to cause significant
heating of the sense resistor,
the temperature coefficient
(tempco) of the resistor can
introduce nonlinearity due to
the signal dependent
temperature rise of the
resistor. The effect increases
as the resistor-to-ambient
thermal resistance increases.
This effect can be minimized
by reducing the thermal
resistance of the current
sensing resistor or by using a
resistor with a lower tempco.
Lowering the thermal
resistance can be accomplished
by repositioning the current
sensing resistor on the PC
board, by using larger PC
board traces to carry away
more heat, or by using a heat
sink. For a two-terminal
current sensing resistor, as the
value of resistance decreases,
the resistance of the leads
become a significant
percentage of the total
resistance. This has two
primary effects on resistor
accuracy. First, the effective
resistance of the sense resistor
can become dependent on
factors such as how long the
leads are, how they are bent,
how far they are inserted into
the board, and how far solder
wicks up the leads during
assembly (these issues will be
discussed in more detail
shortly). Second, the leads are
typically made from a
material, such as copper,
which has a much higher
tempco than the material from
which the resistive element
itself is made, resulting in a
higher tempco overall. Both of
these effects are eliminated
when a four-terminal current
sensing resistor is used. A
four-terminal resistor has two
additional terminals that are
Kelvin-connected directly
across the resistive element
itself; these two terminals are
used to monitor the voltage
across the resistive element
while the other two terminals
are used to carry the load
current. Because of the Kelvin
connection, any voltage drops
across the leads carrying the
load current should have no
impact on the measured
voltage.
MOTOR OUTPUT POWER – HORSEPOWER
40
440
380
220
120
35
30
25
20
15
10
5
0
0
5
10
15
20
25
30
35
MOTOR PHASE CURRENT – A (rms)
Figure 18. Motor output horsepower vs. motor
phase current and supply voltage.
When laying out a PC board
for the current sensing
resistors, a couple of points
should be kept in mind. The
Kelvin connections to the
resistor should be brought
together under the body of the
resistor and then run very
close to each other to the
input of the HCPL-7510; this
minimizes the loop area of the
connection and reduces the
possibility of stray magnetic
fields from interfering with the
measured signal. If the sense
resistor is not located on the
same PC board as the HCPL7510 circuit, a tightly twisted
pair of wires can accomplish
the same thing. Also, multiple
layers of the PC board can be
14
used to increase current
carrying capacity. Numerous
plated-through vias should
surround each non-Kelvin
terminal of the sense resistor
to help distribute the current
between the layers of the PC
board. The PC board should
use 2 or 4 oz. copper for the
layers, resulting in a current
carrying capacity in excess of
20 A. Making the current
carrying traces on the PC
board fairly large can also
improve the sense resistor’s
power dissipation capability by
acting as a heat sink. Liberal
use of vias where the load
current enters and exits the
PC board is also
recommended.
Sense Resistor Connections
The recommended method for
connecting the HCPL-7510 to
the current sensing resistor is
shown in Figure 17. VIN+ (pin
2 of the HPCL-7510) is
connected to the positive
terminal of the sense resistor,
while VIN- (pin 3) is shorted
to GND1 (pin 4), with the
powersupply return path
functioning as the sense line
to the negative terminal of the
current sense resistor. This
allows a single pair of wires
or PC board traces to connect
the HCPL-7510 circuit to the
sense resistor. By referencing
the input circuit to the
negative side of the sense
resistor, any load current
induced noise transients on
the resistor are seen as a
common- mode signal and will
not interfere with the currentsense signal. This is important
because the large load
currents flowing through the
motor drive, along with the
parasitic inductances inherent
in the wiring of the circuit,
can generate both noise spikes
and offsets that are relatively
large compared to the small
voltages that are being
measured across the current
sensing resistor. If the same
power supply is used both for
the gate drive circuit and for
the current sensing circuit, it
is very important that the
connection from GND1 of the
HCPL-7510 to the sense
resistor be the only return
path for supply current to the
gate drive power supply in
order to eliminate potential
ground loop problems. The
only direct connection between
the HCPL-7510 circuit and the
gate drive circuit should be
the positive power supply line.
FREQUENTLY ASKED QUESTIONS ABOUT THE HCPL-7510
1. THE BASICS
1.1: Why should I use the HCPL-7510 for sensing current when Hall-effect sensors are available which don’t need an
isolated supply voltage?
Available in an auto-insertable, 8-pin DIP package, the HCPL-7510 is smaller than and has better
linearity, offset vs. temperature and Common Mode Rejection (CMR) performance than most Halleffect sensors. Additionally, often the required input-side power supply can be derived from the
same supply that powers the gate-drive optocoupler.
2. SENSE RESISTOR AND INPUT FILTER
Ω resistors? I have never seen one that low.
2.1: Where do I get 10 mΩ
Although less common than values above 10 Ω, there are quite a few manufacturers of resistors
suitable for measuring currents up to 50 A when combined with the HCPL-7510. Example product
information may be found at Dale’s web site (http://www.vishay.com/vishay/dale) and Isotek’s web
site (http://www.isotekcorp.com) and Iwaki Musen Kenkyusho’s website (http://
www.iwakimusen.co.jp) and Micron Electric’s website (http://www.micron-e.co.jp).
2.2: Should I connect both inputs across the sense resistor instead of grounding VIN- directly to pin 4?
This is not necessary, but it will work. If you do, be sure to use an RC filter on both pin 2 (VIN+)
and pin 3 (VIN-) to limit the input voltage at both pads.
2.3: Do I really need an RC filter on the input? What is it for? Are other values of R and C okay?
The input anti-aliasing filter (R=39 Ω, C=0.01 µF) shown in the typical application circuit is
recommended for filtering fast switching voltage transients from the input signal.
(This helps to attenuate higher signal frequencies which could otherwise alias with the input
sampling rate and cause higher input offset voltage.)
Some issues to keep in mind using different filter resistors or capacitors are:
1. (Filter resistor:) The equivalent input resistance for HCPL-7510 is around 700 kΩ. It is
therefore best to ensure that the filter resistance is not a significant percentage of this value;
otherwise the offset voltage will be increased through the resistor divider effect. [As an
example, if Rfilt = 5.5 kΩ, then VOS = (Vin * 1%) = 2 mV for a maximum 200 mV input and
VOS will vary with respect to Vin.]
2. The input bandwidth is changed as a result of this different R-C filter configuration. In fact this
is one of the main reasons for changing the input-filter R-C time constant.
3. (Filter capacitance:) The input capacitance of the HCPL-7510 is approximately 1.5 pF. For
proper
operation the switching input-side sampling capacitors must be charged from a
relatively fixed (low impedance) voltage source. Therefore, if a filter capacitor is used it is best
for this capacitor to be a few orders of magnitude greater than the CINPUT (A value of at least
100 pF works well.)
2.4: How do I ensure that the HCPL-7510 is not destroyed as a result of short circuit conditions which
voltage drops across the sense resistor that exceed the ratings of the HCPL-7510’s inputs?
cause
Select the sense resistor so that it will have less than 5 V drop when short circuits occur. The
only other requirement is to shut down the drive before the sense resistor is damaged or its
solder joints melt. This ensures that the input of the HCPL-7510 can not be damaged by sense
resistors going open-circuit.
3. ISOLATION AND INSULATION
3.1: How many volts will the HCPL-7510 withstand?
The momentary (1 minute) withstand voltage is 3750 V rms per UL 1577 and CSA Component
Acceptance Notice #5.
15
4. ACCURACY
4.1: Does the gain change if the internal LED light output degrades with time?
No. The LED is used only to transmit a digital pattern. Agilent has accounted for LED degradation
in the design of the product to ensure long life.
5. MISCELLANEOUS
5.1: How does the HCPL-7510 measure negative signals with only a +5 V supply?
The inputs have a series resistor for protection against large negative inputs. Normal signals are
no more than 200 mV in amplitude. Such signals do not forward bias any junctions sufficiently to
interfere with accurate operation of the switched capacitor input circuit.
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Copyright © 2005 Agilent Technologies, Inc.
February 2, 2005
obsoletes 5989-0317EN
5989-2162EN