TI1 DAC5688 Dual-channel, 16-bit, 800 msps, 2xâ 8x interpolating digital-to-analog converter (dac) Datasheet

DAC5688
www.ti.com
SLLS880C – DECEMBER 2007 – REVISED AUGUST 2010
DUAL-CHANNEL, 16-BIT, 800 MSPS, 2x–8x INTERPOLATING
DIGITAL-TO-ANALOG CONVERTER (DAC)
Check for Samples: DAC5688
FEATURES
DESCRIPTION
•
•
The DAC5688 is a dual-channel 16-bit 800 MSPS
digital-to-analog converter (DAC) with dual CMOS
digital data bus, integrated 2x-8x interpolation filters,
a fine frequency mixer with 32-bit complex
numerically controlled oscillator (NCO), on-board
clock multiplier, IQ compensation, and internal
voltage reference. Different modes of operation
enable or bypass various signal processing blocks.
The DAC5688 offers superior linearity, noise,
crosstalk and PLL phase noise performance.
1
•
•
•
•
•
•
•
•
•
•
Dual, 16-Bit, 800 MSPS DACs
Dual, 16-Bit, 250 MSPS CMOS Input Data
– 16 Sample Input FIFO
– Flexible input data bus options
High Performance
– 81 dBc ACLR WCDMA TM1 at 70 MHz
2x-32x Clock Multiplying PLL/VCO
Selectable 2x–8x Interpolation Filters
– Stop-band Attenuation > 80 dB
Complex Mixer with 32-Bit NCO
Digital Quadrature Modulator Correction
– Gain, Phase and Offset Correction
Digital Inverse SINC Filter
3- or 4-Wire Serial Control Interface
On Chip 1.2-V Reference
Differential Scalable Output: 2 to 20 mA
Package: 64-pin 9×9mm QFN
The DAC5688 dual CMOS data bus provides 250
MSPS input data transfer per DAC channel. Several
input data options are available: dual-bus data,
single-bus interleaved data, even and odd
multiplexing at half-rate, and an input FIFO with either
external or internal clock to ease interface timing.
Input data can interpolated 2x, 4x or 8x by on-board
digital interpolating FIR filters with over 80 dB of
stop-band attenuation.
The DAC5688 allows both complex or real output. An
optional 32-bit NCO/mixer in complex mode provides
frequency upconversion and the dual DAC output
produces a complex Hilbert Transform pair. A digital
Inverse SINC filter compensates for natural DAC
sin(x)/x frequency roll-off. The digital Quadrature
Modulator Correction (QMC) feature allows IQ
compensation of phase, gain and offset to maximize
sideband rejection and minimize LO feed-through of
an external quadrature modulator performing the final
single sideband RF up-conversion.
APPLICATIONS
•
•
•
•
•
Cellular Base Stations
Broadband Wireless Access (BWA)
WiMAX 802.16
Fixed Wireless Backhaul
Cable Modem Termination System (CMTS)
The DAC5688 is pin compatible with the DAC5689
which does not include a clock-multiplying PLL. The
DAC5688 is characterized for operation over the
industrial temperature range of –40°C to 85°C and is
available in a 64-pin 9x9mm QFN package.
ORDERING INFORMATION (1)
(1)
(2)
(3)
Order Code
TA = –40°C to 85°C
Package Qty
Tape and Reel Format
DAC5688IRGCT
250
DAC5688IRGCR
2000
Package
Drawing/Type (2)
(3)
RGC / 64QFN Quad Flatpack No-Lead
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
website at www.ti.com.
Thermal Pad Size: 7,4 mm × 7,4 mm
MSL Peak Temperature: Level-3-260C-168 HR
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2007–2010, Texas Instruments Incorporated
DAC5688
SLLS880C – DECEMBER 2007 – REVISED AUGUST 2010
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
FUNCTIONAL BLOCK DIAGRAM
CLKVDD
VFUSE
LPF
DVDD
CLK2
CLK2C
CLKOUT
CLKO_CLK1
1.2 V
Reference
Internal Clock Generation and
2x - 32x PLL Clock Multiplier
2-8x Fdata
BIASJ
LOCK_CLK1C
LOCK
A
QMC
gain
A-Offset
SYNC
TXENABLE
FIR2
FIR4
DA[15:0]
Input FIFO /
Demux
x2
x2
x2
2x – 8x Interpolation
67 taps
DB[15:0]
x2
19 taps
11 taps
x2
x2
Quadrature Modulator
Correction (QMC):
Phase & Gain
FIR3
Full Mixer (FMIX)
FIR1
x
sin(x)
IOUTA1
IOUTA2
9 taps
x
sin(x)
16-b DAC
IOUTB1
IOUTB2
QMC
B-Offset
32-bit NCO
SIF Control
16-b DAC
sin
cos
RESETB
EXTIO
EXTLO
B
gain
AVDD
Updated: 2-Oct-07
IOVDD
SDIO SDO SDENB SCLK
GND
2
DVDD
RESETB
50
49
51
IOUTA2
IOUTA1
AVDD
AVDD
53
AVDD
55
54
52
BIASJ
EXTIO
57
56
AVDD
EXTLO
59
58
IOUTB1
IOUTB2
60
AVDD
62
61
LPF
DVDD
64
63
PINOUT
CLKVDD
1
48
SDENB
CLK2
2
47
SCLK
CLK2C
3
46
SDIO
GND
4
45
SDO
SYNC
5
44
VFUSE
TXENABLE
DA15
6
43
DB15
42
DB14
DA14
8
41
DB13
IOVDD
9
40
DB12
DVDD
10
39
DVDD
DA13
11
38
DB11
DA12
12
37
DB10
DA11
13
36
DB9
DA10
14
35
DB8
DA9
15
34
DB7
DA8
16
33
DB6
DAC5688
7
31
32
DB5
30
DB3
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DB4
28
29
DB2
DB0
DB1
26
27
LOCK_CLK1C
24
25
DA0
CLKO_CLK1
22
23
DA1
21
DA3
DA2
19
20
DA5
DA6
DA4
17
18
DA7
RGC Package
64QFN, 9x9mm
(Top View)
Copyright © 2007–2010, Texas Instruments Incorporated
Product Folder Link(s): DAC5688
DAC5688
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SLLS880C – DECEMBER 2007 – REVISED AUGUST 2010
PIN FUNCTIONS
PIN
NAME
NO.
I/O
DESCRIPTION
AVDD
51, 54,
55, 59,
62
I
Analog supply voltage. (3.3V)
BIASJ
57
O
Full-scale output current bias. For 20mA full-scale output current, connect a 960 Ω resistor to GND.
CLK2
2
I
With the clock multiplier PLL enabled, CLK2 provides lower frequency reference clock. If the PLL is disabled, CLK2
directly provides clock for DAC up to 800 MHz.
CLK2C
3
I
Complementary CLK2 input.
CLKO_CLK1
25
I/O
In Dual Clock Modes, provides lower frequency input clock (CLK1). Optionally provides clock (CLKO) output for data
bus. Internal pull-down.
LOCK_ CLK1C
26
I/O
Complementary CLK1 signal if configured as a differential input. In PLL mode, optionally outputs PLL lock status. Internal
pull-down.
CLKVDD
1
I
Internal clock buffer supply voltage. (1.8V)
It is recommended to isolate this supply from DVDD.
DA[15..0]
7, 8,
11–24
I
A-Channel Data Bits 0 through 15.
DA15 is most significant data bit (MSB) – pin 7
DA0 is least significant data bit (LSB) – pin 24
Internal pull-down. The order of bus can be reversed via CONFIG4 reva bit.
DB[15..0]
40–43,
27–38
I
B-Channel Data Bits 0 through 15.
DB15 is most significant data bit (MSB) – pin 43
DB0 is least significant data bit (LSB) – pin 27
Internal pull-down. The order of bus can be reversed via CONFIG4 revb bit.
DVDD
10, 39,
50, 63
I
Digital supply voltage. (1.8V)
For best performance it is recommended to isolate pins 10 and 39 from all other 1.8V supplies.
EXTIO
56
I/O
Used as external reference input when internal reference is disabled (i.e., EXTLO connected to AVDD). Used as internal
reference output when EXTLO = GND, requires a 0.1mF decoupling capacitor to GND when used as reference output
EXTLO
58
O
Connect to GND for internal reference, or AVDD for external reference.
4,
Thermal
Pad
I
Pin 4 and the Thermal Pad located on the bottom of the QFN package is ground for AVDD, DVDD and IOVDD supplies.
IOUTA1
52
O
A-Channel DAC current output. An offset binary data pattern of 0x0000 at the DAC input results in a full scale current
sink and the least positive voltage on the IOUTA1 pin. Similarly, a 0xFFFF data input results in a 0 mA current sink and
the most positive voltage on the IOUTA1 pin. In single DAC mode, outputs appear on the IOUTA1/A2 pair only.
IOUTA2
53
O
A-Channel DAC complementary current output. The IOUTA2 has the opposite behavior of the IOUTA1 described above.
An input data value of 0x0000 results in a 0mA sink and the most positive voltage on the IOUTA2 pin.
IOUTB1
61
O
B-Channel DAC current output. Refer to IOUTA1 description above.
IOUTB2
60
O
B-Channel DAC complementary current output. Refer to IOUTA2 description above.
IOVDD
9
I
3.3V supply voltage for all digital I/O. Note: This supply input should remain at 3.3V regardless of the 1.8V or 3.3V
selectable digital input switching thresholds via CONFIG26 io_1p8_3p3.
LPF
64
I
PLL loop filter connection. If not using the clock multiplying PLL, leave the LPF pin open. Set PLL_sleep and clear
PLL_ena control bits for reduced power dissipation.
SYNC
5
I
Optional SYNC input for internal clock dividers, FIFO, NCO and QMC blocks. Internal pull-down.
RESETB
49
I
Resets the chip when low. Internal pull-up.
SCLK
47
I
Serial interface clock. Internal pull-down.
SDENB
48
I
Active low serial data enable, always an input to the DAC5688. Internal pull-up.
SDIO
46
I/O
Bi-directional serial data in 3-pin mode (default). In 4-pin interface mode (CONFIG5 sif4), the SDIO pin is an input only.
Internal pull-down.
SDO
45
O
Uni-directional serial interface data in 4-pin mode (CONFIG5 sif4). The SDO pin is tri-stated in 3-pin interface mode
(default). Internal pull-down.
TXENABLE
6
I
Transmit enable input. Internal pull-down. TXENABLE has two purposes. In all modes, TXENABLE must be high for the
DATA to the DAC to be enabled. When TXENABLE is low, the digital logic section is forced to all 0, and any input data is
ignored. In interleaved data mode, TXENABLE can be used to synchronize the data to channels A and B. The first
A-channels sample should be aligned with the rising edge of TXENABLE.
VFUSE
44
I
Digital supply voltage. (1.8V) This supply pin is also used for factory fuse programming. Connect to DVDD pins for
normal operation.
GND
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Copyright © 2007–2010, Texas Instruments Incorporated
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DAC5688
SLLS880C – DECEMBER 2007 – REVISED AUGUST 2010
www.ti.com
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted) (1)
VALUE
UNIT
–0.5 to 2.3
V
–0.5 to 2.3
V
–0.5 to 2.3
V
AVDD (2)
–0.5 to 4
V
IOVDD (2)
–0.5 to 4
V
DVDD (2)
Supply Voltage Range
VFUSE
(2)
CLKVDD (2)
Supply Voltage Range
AVDD to DVDD
–2 to 2.6
V
CLKVDD to DVDD
–0.5 to 0.5
V
IOVDD to AVDD
–0.5 to 0.5
V
–0.5 to CLKVDD + 0.5
V
–0.5 to IOVDD + 0.5
V
–0.5 to IOVDD + 0.5
V
–0.5 to IOVDD + 0.5
V
–0.5 to AVDD + 0.5
V
CLK2, CLK2C
(2)
CLKO_CLK1, LOCK_CLK1C, SLEEP, TXENABLE (2)
DA[15..0] ,DB[15..0]
(2)
SDO, SDIO, SCLK, SDENB, RESETB
IOUTA1/B1, IOUTA2/B2
(2)
(2)
LPF, EXTIO, EXTLO, BIASJ
(2)
Peak input current (any input)
Peak total input current (all inputs)
–0.5 to AVDD + 0.5
V
20 mA
mA
–30 mA
mA
Operating free-air temperature range, TA: DAC5688I
–40 to 85
°C
Storage temperature range
–65 to 150
°C
(1)
(2)
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of these or any other conditions beyond those indicated under recommended operating conditions is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
Measured with respect to GND.
THERMAL INFORMATION
THERMAL CONDUCTIVITY
(1) (2)
DAC5688
64ld QFN
TJ
Maximum junction temperature
qJA
Theta junction-to-ambient thermal resistance (still air)
22
Theta junction-to-ambient thermal resistance (200 lfm)
15
yJT
Psi junction-to-top of package characterization parameter
0.2
qJB
Theta junction-to-board characterization parameter
3.5
(1)
(2)
4
UNITS
125
°C
°C/W
Air flow or heat sinking reduces qJA and may be required for sustained operation at 85°C under maximum operating conditions.
It is strongly recommended to solder the device thermal pad to the board ground plane.
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Copyright © 2007–2010, Texas Instruments Incorporated
Product Folder Link(s): DAC5688
DAC5688
www.ti.com
SLLS880C – DECEMBER 2007 – REVISED AUGUST 2010
ELECTRICAL CHARACTERISTICS (DC SPECIFICATIONS)
over recommended operating free-air temperature range, AVDD, IOVDD = 3.3 V, DVDD, CLKVDD = 1.8 V, IOUTFS = 20 mA
PARAMETER
TEST CONDITIONS
RESOLUTION
MIN
TYP
MAX
UNIT
16
Bits
DC ACCURACY
INL
Integral nonlinearity
DNL
Differential nonlinearity
1 LSB = IOUTFS/216
±4
LSB
±2
LSB
ANALOG OUTPUT
Coarse gain linearity
± 0.04
Offset error mid code offset
Gain error
%FSR
With external reference
1
%FSR
With internal reference
0.7
Gain mismatch
With internal reference, dual DAC mode
Minimum full scale output current
Nominal full-scale current, IOUTFS = 16 × IBIAS current.
–2
%FSR
2
%FSR
2
Maximum full scale output current
Output compliance range (1)
LSB
0.01
mA
20
IOUTFS = 20 mA
AVDD
– 0.5V
Output resistance
Output capacitance
AVDD
+ 0.5V
V
300
kΩ
5
pF
REFERENCE OUTPUT
VREF
Reference output voltage
Internal Reference Mode
1.14
Reference output current (2)
1.2
1.26
V
100
nA
REFERENCE INPUT
VEXTIO
Input voltage range
External Reference Mode
0.1
Input resistance
Small signal bandwidth
1.25
V
1
CONFIG26: isbiaslpf_a and isbiaslpf_b = 0
95
CONFIG26: isbiaslpf_a and isbiaslpf_b = 1
472
Input capacitance
MΩ
kHz
100
pF
TEMPERATURE COEFFICIENTS
Offset drift
Gain drift
±1
With external reference
±15
With internal reference
±30
Reference voltage drift
ppm of
FSR/°C
±8
ppm/°C
POWER SUPPLY
PSRR
AVDD, IOVDD
3.0
3.3
3.6
DVDD, CLKVDD
1.7
1.8
1.9
Power supply rejection ratio
AVDD + IOVDD current, 3.3V
DVDD + CLKVDD current, 1.8V
Mode 1: ×8 Interp, PLL on, QMC = off, ISINC = off,
DAC A+B on, FIN = 5 MHz Tone, NCO = 145 MHz,
FOUT = 150 MHz, FDAC = 500 MHz
Power Dissipation
AVDD + IOVDD current, 3.3V
DVDD + CLKVDD current, 1.8V
P
Mode 2: ×8 Interp, PLL off, QMC = on, ISINC = on,
DAC A+B on, FIN = 5 MHz Tone, NCO = 91 MHz
FOUT = 96 MHz, FDAC = 614.4 MHz
Power Dissipation
AVDD + IOVDD current, 3.3V
DVDD + CLKVDD current, 1.8V
Mode 3 (Max): ×4 Interp, PLL on, QMC = on, ISINC = on,
DAC A+B on, FIN = 5 MHz Tone, NCO = 135 MHz,
FOUT = 140 MHz, FDAC = 800 MHz
Power Dissipation
AVDD + IOVDD current, 3.3V
DVDD + CLKVDD current, 1.8V
Power Dissipation
(1)
(2)
V
±0.2
%FSR/V
150
mA
450
mA
1300
mW
140
mA
520
mA
1400
mW
150
mA
700
1750
Mode 4 (Sleep): ×8 Interp, PLL off, QMC = off, ISINC = off,
DAC A+B off, FIN = 5 MHz Tone, NCO = off,
FOUT = off, FDAC = 800 MHz,
V
mA
1950
12
mA
15
65
mW
mA
100
mW
The upper limit of the output compliance is determined by the CMOS process. Exceeding this limit may result in transistor breakdown,
resulting in reduced reliability of the DAC5688 device. The lower limit of the output compliance is determined by the load resistors and
full-scale output current. Exceeding the upper limit adversely affects distortion performance and integral nonlinearity.
Use an external buffer amplifier with high impedance input to drive any external load.
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Copyright © 2007–2010, Texas Instruments Incorporated
Product Folder Link(s): DAC5688
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DAC5688
SLLS880C – DECEMBER 2007 – REVISED AUGUST 2010
www.ti.com
ELECTRICAL CHARACTERISTICS (AC SPECIFICATIONS)
Over recommended operating free-air temperature range, AVDD, IOVDD = 3.3 V, DVDD, CLKVDD = 1.8 V, IOUTFS = 20 mA
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
ANALOG OUTPUT (1)
fDAC
Maximum output update rate
ts(DAC)
Output settling time to 0.1%
Transition: Code 0x0000 to 0xFFFF
tpd
Output propagation delay
DAC outputs are updated on falling edge of DAC clock.
Does not include Digital Latency (see below).
tr(IOUT)
Output rise time
tf(IOUT)
Output fall time
Digital Latency
AC PERFORMANCE
SFDR
SNR
IMD
ns
2
ns
10% to 90%
220
ps
90% to 10%
220
ps
No Interp, NCO off, QMC off, ISINC = off
109
x2 Interpolation, NCO off, QMC off, ISINC = off
172
x4 Interpolation, NCO off, QMC off, ISINC = off
276
x8 Interpolation, NCO off, QMC off, ISINC = off
488
x8 Interpolation, NCO on, QMC off, ISINC = off
512
x8 Interpolation, NCO on, QMC on, ISINC = off
528
x8 Interpolation, NCO on, QMC on, ISINC = on
548
Spurious free dynamic
range
×4 Interp, PLL off, CLK2 = 800 MHz,
DAC A+B on,
0 dBFS Single tone, FOUT = FIN
First Nyquist Zone < fDATA/2
FOUT= 10.1 MHz
83
FOUT= 20.1 MHz
79
×4 Interp, PLL off, CLK2 = 800 MHz,
DAC A+B on,
0 dBFS Single tone, FIN = 10.1 MHz,
FOUT = FIN + NCO
NCO= 10 MHz, FOUT= 20.1 MHz
72
NCO= 60 MHz, FOUT= 70.1 MHz
68
NCO= 140 MHz, FOUT= 150.1 MHz
64
NCO= 290 MHz, FOUT= 300.1 MHz
57
NCO= 40 MHz, FOUT= 51±0.5 MHz
85
NCO= 60 MHz, FOUT= 71±0.5 MHz
83
NCO= 130 MHz, FOUT= 141±0.5 MHz
74
Third-order
Two-Tone intermodulation
(Each tone at –6 dBFS)
×4 Interp, PLL off, CLK2 = 800 MHz,
DAC A+B on,
FIN = 10.5 and 11. 5 MHz,
FOUT = FIN + NCO
Four-tone Intermodulation
to Nyquist
(Each tone at –12 dBFS)
×4 Interp, PLL off, CLK2 = 800 MHz, DAC A+B on,
FIN = 9.8, 10.4, 11.6 and 12.2 MHz (600kHz spacing), NCO = 129 MHz,
FOUT = FIN + NCO = 140±1.2 MHz
×8 Interp, PLL off, CLK2 = 737.28 MHz,
DAC A+B on, FIN = 23 .04 MHz, NCO = off
ACLR
(3)
Adjacent Channel
Leakage Ratio
Noise Floor,
Noise Spectral Density
(NSD)
(3)
(1)
(2)
(3)
6
MSPS
10.4
DAC
clock
cycles
(2)
Signal-to-Noise Ratio
IMD3
800
×8 Interp, PLL off, CLK2 = 737.28 MHz,
DAC A+B on, FIN = Baseband I/Q,
FOUT = NCO
×8 Interp, PLL off, CLK2 = 737.28 MHz,
DAC A+B on,
FIN = FOUT = Baseband I/Q,
50 MHz offset, 1 MHz BW
73
Single Carrier, FOUT = 23.04 MHz
81
Single Carrier, FOUT = 70MHz
81
Single Carrier, FOUT = 140MHz
78
dBc
dBc
dBc
dBc
dBc
Four Carrier, FOUT = 140MHz
70
Single Carrier Noise Floor
101
dBm
Single Carrier NSD in 1 MHz BW
161
dBm/Hz
Four Carrier Noise Floor
101
dBm
Four Carrier NSD in 1 MHz BW
161
dBm/Hz
Measured differential across IOUTA1 and IOUTA2 or IOUTB1 and IOUTB2 with 25 Ω each to AVDD.
4:1 transformer output termination, 50Ω doubly terminated load
W-CDMA with 3.84 MHz BW, 5-MHz spacing, centered at IF. TESTMODEL 1, 10 ms
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DAC5688
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SLLS880C – DECEMBER 2007 – REVISED AUGUST 2010
ELECTRICAL CHARACTERISTICS (DIGITAL SPECIFICATIONS)
Over recommended operating free-air temperature range, AVDD, IOVDD = 3.3V, DVDD, CLKVDD = 1.8V.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
CMOS INTERFACE: SDO, SDIO, SCLK, SDENB, RESETB, DA[15:0], DB[15:0], SYNC, TXENABLE, CLKO_CLK1, LOCK_CLK1C
CONFIG26 io_1p8_3p3 = 0 (3.3V levels)
2.30
CONFIG26 io_1p8_3p3 = 1 (1.8V levels)
1.25
VIH
High-level input voltage
VIL
Low-level input voltage
IIH
High-level input current
±20
mA
IIL
Low-level input current
±20
mA
CI
CMOS Input capacitance
2
pF
VOH
VOL
V
CONFIG26 io_1p8_3p3 = 0 (3.3V levels)
1.00
CONFIG26 io_1p8_3p3 = 1 (1.8V levels)
0.54
SDO, SDIO
ILOAD = –100 mA
IOVDD
– 0.2
SDO, SDIO
ILOAD = –2 mA
0.8 ×
IOVDD
SDO, SDIO
ILOAD = 100 mA
0.2
SDO, SDIO
ILOAD = 2 mA
0.5
Input data rate
V
V
0
250
V
MSPS
ts(SDENB)
Setup time, SDENB to rising edge of SCLK
20
ns
ts(SDIO)
Setup time, SDIO valid to rising edge of SCLK
10
ns
th(SDIO)
Hold time, SDIO valid to rising edge of SCLK
5
ns
tSCLK
Period of SCLK
100
ns
tSCLKH
High time of SCLK
40
ns
tSCLK
Low time of SCLK
40
td(Data)
Data output delay after falling edge of SCLK
10
ns
tRESET
Minimum RESETB pulse width
25
ns
ns
TIMING PARALLEL DATA INPUT: (DUAL CLOCK and DUAL SYNCHRONOUS CLOCK MODES: Figure 32)
ts
Setup time
th
Hold time
t_align
Max timing offset between CLK1 and CLK2
rising edges
CLK1/C = input
DUAL SYNCHRONOUS BUS MODE only
(Typical characteristic)
1
1
ns
1
ns
- 0.55
ns
2fCLK 2
TIMING PARALLEL DATA INPUT (EXTERNAL CLOCK MODE: Figure 33 and PLL CLOCK MODE: Figure 34)
ts
Setup time
th
Hold time
td(CLKO)
Delay time
1
CLKO_CLK1 = input or output. Note: Delay
time increases with higher capacitive
loads.
ns
1
ns
4.5
ns
CLOCK INPUT (CLK2/CLK2C)
CLK2/C Duty cycle
40%
CLK2/C Differential voltage (1)
60%
0.4
CLK2/C Input common mode
1
V
2/3 ×
CLKVDD
V
CLK2C Input Frequency
800
MHz
CLOCK INPUT (CLK1/CLK1C)
CLK1/C Duty cycle
40%
CLK1/C Differential voltage
0.4
CLK1/C Input common mode
60%
1.0
V
IOVDD
/2
CLK1/C Input Frequency
V
250
MHz
160
MHz
CLOCK OUTPUT (CLKO)
CLKO Output Frequency (2)
(1)
(2)
with 3pF load
Driving the clock input with a differential voltage lower than 1V will result in degraded performance.
Specified by design and simulation. Not production tested. It is recommended to buffer CLKO.
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ELECTRICAL CHARACTERISTICS (DIGITAL SPECIFICATIONS)
Over recommended operating free-air temperature range, AVDD, IOVDD = 3.3V, DVDD, CLKVDD = 1.8V.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
PHASE LOCKED LOOP
Phase noise at 600 kHz offset
Phase noise at 6 MHz offset
100 MHz, 0-dBFS tone,
fDATA = 200 MSPS, CLK2/C = 200 MHz,
PLL_m = '00111', PLL_n = '001', (M/N=4)
PLL_gain = '11', PLL_range = '1000' (8)
x4 Interpolation
PLL_gain = '00', PLL_range = '0000' (0)
PLL_gain = '01', PLL_range = '0001' (1)
PLL_gain = '01', PLL_range = '0010' (2)
PLL_gain = '01', PLL_range = '0011' (3)
PLL_gain = '10', PLL_range = '0100' (4)
PLL_gain = '10', PLL_range = '0101' (5)
PLL/VCO Operating Frequency,
Typical VCO Gain
PLL_gain = '10', PLL_range = '0110' (6)
PLL_gain = '10', PLL_range = '0111' (7)
PLL_gain = '11', PLL_range = '1000' (8)
PLL_gain = '11', PLL_range = '1001' (9)
PLL_gain = '11', PLL_range = '1010' (A)
PLL_gain = '11', PLL_range = '1011' (B)
PFD Maximum Frequency
8
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–125
dBc/ Hz
–146
140
270
215
270
440
290
370
490
530
650
680
720
750
830
860
MHz
MHz/V
890
245
880
MHz
MHz/V
260
840
MHz
MHz/V
275
800
MHz
MHz/V
230
750
MHz
MHz/V
245
710
MHz
MHz/V
260
660
MHz
MHz/V
285
600
MHz
MHz/V
230
530
MHz
MHz/V
255
450
MHz
MHz/V
MHz
MHz/V
910
MHz
235
MHz/V
160
MHz
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TYPICAL CHARACTERISTICS
Figure 1. Integral Nonlinearity
Figure 2. Differential Nonlinearity
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Figure 3. Single Tone Spectral Plot
Figure 4. Single Tone Spectral Plot
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TYPICAL CHARACTERISTICS (continued)
85
Fdata = 200 MSPS
FIN = 10.1 MHz, Sweep FMIX
x4 Interpolation
PLL off
80
SFDR - dBc
75
70
65
60
55
50
Figure 5. In-Band SFDR vs. Intermediate Frequency
100
150
200
FOUT (MHz)
250
300
350
0
Fdata = 200 MSPS
IF = FNCO
x4 Interpolation, PLL off
105
Fdata = 200 MSPS, IQ
FIN = 20 MHz ±0.5 MHz
IF = 20 MHz
x4 Interpolation
PLL off
-10
100
-20
95
-30
-6 dBFS
Power - dBm
90
85
80
75
-40
-50
-60
-70
70
-80
65
0 dBFS
60
-90
55
0
50
100
150
200
250
fi - Input Frequency - MHz
300
350
-100
18
Figure 7. Two Tone IMD vs Intermediate Frequency
10
50
Figure 6. Out-Of-Band SFDR vs Intermediate Frequency
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110
IMD3 - dBc
0
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19
20
IF - MHz
21
22
Figure 8. Two Tone IMD Spectral Plot
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SLLS880C – DECEMBER 2007 – REVISED AUGUST 2010
TYPICAL CHARACTERISTICS (continued)
85
0
Fdata = 200 MSPS, IQ
FIN = 0 MHz
IF = 140 MHz
x4 Interpolation, FMIX
PLL off
-10
-20
80
-30
75
-40
ACLR - dBc
Power - dBm
Fdata = 92.16 MSPS
IF = FNCO
x8 Interpolation
Fdac = 737.28 MSPS
-50
-60
PLL OFF
70
65
-70
PLL ON
-80
60
-90
-100
138
139
140
IF - MHz
141
142
55
50
0
100
150
200
IF (MHz)
250
300
350
Figure 9. Two Tone IMD Spectral Plot
Figure 10. WCDMA ACLR vs Intermediate Frequency
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Figure 11. WCDMA TM1:Single Carrier, PLL Off
Figure 12. WCDMA TM1:Single Carrier, PLL On
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TYPICAL CHARACTERISTICS (continued)
12
Figure 13. WCDMA TM1:Single Carrier, PLL Off
Figure 14. WCDMA TM1:Single Carrier, PLL On
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Figure 15. WCDMA TM1:Two Carriers, PLL Off
Figure 16. WCDMA TM1:Two Carriers, PLL On
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TYPICAL CHARACTERISTICS (continued)
Figure 17. WCDMA TM1:Four Carriers, PLL Off
Figure 18. WCDMA TM1:Four Carriers, PLL On
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TEST METHODOLOGY
Typical AC specifications were characterized with the DAC5688EVM. A sinusoidal master clock frequency is
generated by an HP8665B signal generator which drives an Agilent 8133A pulse generator to generate a square
wave output clock for the TSW3100 Pattern Generator and EVM input clock. On the EVM, the input clock is
driven by an CDCM7005 clock distribution chip that is configured to simply buffer the external clock or divide it
down for necessary test configurations.
The DAC5688 output is characterized with a Rohde and Schwarz FSU spectrum analyzer. For WCDMA signal
characterization, it is important to use a spectrum analyzer with high IP3 and noise subtraction capability so that
the spectrum analyzer does not limit the ACPR measurement.
DEFINITION OF SPECIFICATIONS
Adjacent Carrier Leakage Ratio (ACLR): Defined for a 3.84Mcps 3GPP W-CDMA input signal measured in a
3.84MHz bandwidth at a 5MHz offset from the carrier with a 12dB peak-to-average ratio.
Analog and Digital Power Supply Rejection Ratio (APSRR, DPSRR): Defined as the percentage error in the
ratio of the delta IOUT and delta supply voltage normalized with respect to the ideal IOUT current.
Differential Nonlinearity (DNL): Defined as the variation in analog output associated with an ideal 1 LSB
change in the digital input code.
Gain Drift: Defined as the maximum change in gain, in terms of ppm of full-scale range (FSR) per °C, from the
value at ambient (25°C) to values over the full operating temperature range.
Gain Error: Defined as the percentage error (in FSR%) for the ratio between the measured full-scale output
current and the ideal full-scale output current.
Integral Nonlinearity (INL): Defined as the maximum deviation of the actual analog output from the ideal output,
determined by a straight line drawn from zero scale to full scale.
Intermodulation Distortion (IMD3, IMD): The two-tone IMD3 or four-tone IMD is defined as the ratio (in dBc) of
the worst 3rd-order (or higher) intermodulation distortion product to either fundamental output tone.
Offset Drift: Defined as the maximum change in DC offset, in terms of ppm of full-scale range (FSR) per °C,
from the value at ambient (25°C) to values over the full operating temperature range.
Offset Error: Defined as the percentage error (in FSR%) for the ratio between the measured mid-scale output
current and the ideal mid-scale output current.
Output Compliance Range: Defined as the minimum and maximum allowable voltage at the output of the
current-output DAC. Exceeding this limit may result reduced reliability of the device or adversely affecting
distortion performance.
Reference Voltage Drift: Defined as the maximum change of the reference voltage in ppm per degree Celsius
from value at ambient (25°C) to values over the full operating temperature range.
Spurious Free Dynamic Range (SFDR): Defined as the difference (in dBc) between the peak amplitude of the
output signal and the peak spurious signal.
Signal to Noise Ratio (SNR): Defined as the ratio of the RMS value of the fundamental output signal to the
RMS sum of all other spectral components below the Nyquist frequency, including noise, but excluding the first
six harmonics and dc.
14
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REGISTER DESCRIPTIONS
Table 1. Register Map
Name
Address
Default
(MSB)
Bit 7
Bit 6
Bit 5
STATUS0
CONFIG1
0x00
0x01
PLL_lock
unused
unused
0x01
0x0B
CONFIG2
0x02
0xE1
CONFIG3
0x03
0x00
CONFIG4
0x04
0x00
ser_dac_data_ena
CONFIG5
0x05
0x22
sif4
CONFIG6
0x06
0x00
phaseoffset(7:0)
CONFIG7
0x07
0x00
phaseoffset(15:8)
CONFIG8
0x08
0x00
phaseadd(7:0)
CONFIG9
0x09
0x00
phaseadd(15:8)
CONFIG10
0x0A
0x00
phaseadd(23:16)
CONFIG11
0x0B
0x00
phaseadd(31:24)
CONFIG12
0x0C
0x00
qmc_gaina(7:0)
CONFIG13
0x0D
0x00
qmc_gainb(7:0)
CONFIG14
0x0E
0x00
qmc_phase(7:0)
CONFIG15
0x0F
0x24
CONFIG16
0x10
0x00
CONFIG17
0x11
0x00
CONFIG18
0x12
0x00
qmc_offseta(12:8)
unused
unused
unused
CONFIG19
0x13
0x00
qmc_offsetb(12:8)
unused
unused
unused
CONFIG20
0x14
0x00
CONFIG21
0x15
0x00
CONFIG22
0x16
0x15
CONFIG23
0x17
0x15
CONFIG24
0x18
0x80
CONFIG25
0x19
0x00
unused
unused
unused
CONFIG26
0x1A
0x0D
io_1p8_3p3
unused
sleepb
CONFIG27
0x1B
0xFF
CONFIG28
0x1C
0x00
CONFIG29
0x1D
0x00
CONFIG30
0x1E
0x00
insel_mode(1:0)
diffclk_ena
clk1_in_ena
Bit 4
Bit 3
Bit 2
device_ID(2:0)
synchr_clkin
twos
inv_inclk
clk1c_in_ena
clko_SE_hold
fir4_ena
qmc_offset_ena
clko_dly(1:0)
output_delay(1:0)
clkdiv_sync_ena
qmc_phase(9:8)
(LSB)
Bit 0
version(1:0)
unused
diffclk_dly(1:0)
sif_sync_sig
Bit 1
interp_value(1:0)
qmc_corr_ena
mixer_ena
reserved
B_equals_A
A_equals_B
unused
reva
revb
clkdiv_sync_sel
reserved
clkdiv_shift
mixer_gain
unused
qmc_gaina(10:8)
qmc_gainb(10:8)
qmc_offseta(7:0)
qmc_offsetb(7:0)
ser_dac_data(7:0)
ser_dac_data(15:8)
nco_sel(1:0)
nco_reg_sel(1:0)
unused
unused
qmcorr_reg_sel(1:0)
fifo_sel(2:0)
qmoffset_reg_sel(1:0)
aflag_sel
unused
unused
unused
unused
unused
unused
unused
unused
unused
unused
unused
sleepa
isbiaslpf_a
isbiaslpf_b
PLL_sleep
PLL_ena
fifo_sync_strt(3:0)
coarse_daca(3:0)
coarse_dacb(3:0)
reserved
PLL_m(4:0)
PLL_LPF_reset
VCO_div2
PLL_gain(1:0)
PLL_n(2:0)
PLL_range(3:0)
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Register name: STATUS0 - Address: 0x00, Default 0x01
Bit 7
Bit 6
Bit 5
PLL_lock
0
unused
0
unused
0
PLL_lock
device_ID(2:0)
version(1:0)
:
:
:
Bit 4
Bit 3
0
device_ID (2:0)
0
Bit 2
Bit 1
Bit 0
version(1:0)
0
0
1
Bit 1
Bit 0
Asserted when the internal PLL is locked. (Read Only)
Returns ‘000’ for DAC5688. (Read Only)
A hardwired register that contains the version of the chip. (Read Only)
Register name: CONFIG1 Address: 0x01, Default 0x0B
Bit 7
Bit 6
insel_mode (1:0)
0
0
insel_mode(1:0)
:
Bit 5
Bit 4
Bit 3
Bit 2
unused
0
synchr_clkin
0
twos
1
inv_inclk
0
Controls the expected format of the input data. For the interleaved modes, TXENABLE or the MSB of the port
that does not have data can be used to tell the chip which sample is the A sample. For TXENABLE the sample
aligned with the rising edge is A. For the MSB, it is presumed that this signal will toggle with A and B. The MSB
should be ‘1’ for A and ‘0’ for B. (*** See CONFIG23 ***)
insel_mode
00
01
10
11
synchr_clkin
:
twos
:
inv_inclk
:
interp_value(1:0)
:
Function
Normal input on A and B.
Interleaved input on A, which is de-interleaved and placed on
both A and B data paths. (*** See CONFIG23 ***)
Interleaved input on B, which is de-interleaved and placed on
both A and B data paths. (*** See CONFIG23 ***)
Half rate data on A and B inputs. This data is merge together
to form a single stream of data on the A data path.
This turns on the synchronous mode of the dual-clock in mode. In this mode, the CLK2/C and CLK1/C must be
synchronous in phase since the slower clock is used to synchronize dividers in the clock distribution circuit.
When set (default), the input data format is expected to be 2’s complement. When cleared, the input is
expected to be offset-binary.
This allows the input clock, the clock driving the input side of the FIFO to be inverted. This allows easier
registering of the data (more setup/hold time) in the single-clock mode of the device
These bits define the interpolation factor:
interp_value
00
01
10
11
16
interp_valule(1:0)
1
1
Interpolation Factor
1X
2X
4X
8X
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Register name: CONFIG2 Address: 0x02, Default 0xE1
Bit 7
Bit 6
Bit 5
Bit 4
diffclk_ena
1
clk1_in_ena
1
clk1c_ in_ena
1
clko_SE_hold
0
diffclk_ena
:
clk1_in_ena
:
clk1c_in_ena
:
clko_SE_hold
:
fir4_ena
qmc_offset_ena
qmc_corr_ena
mixer_ena
:
:
:
:
Bit 3
Bit 2
fir4_ ena
0
Bit 1
qmc_ offset_ena
0
qmc_ corr_ena
0
Bit 0
mixer_ena
1
When set (default), CLK1 and CLK1C pins are used as a differential clock input. Otherwise, CLK1 pin is used as
a single ended input.
When set (default), the CLKO_CLK1 pin is configured as the CLK1 input. If cleared, the pin is configured to
output an internally generated CLKO as a clock signal for the input data.
When set (default), the LOCK_CLK1C pin is configured as the CLK1C input. If cleared, the pin is configured to
output the PLL_LOCK status.
When set, the single ended (SE) clock is held to a value of ‘1’ so that the signal doesn’t toggle when using the
differential clock input.
When set, the FIR4 Inverse SINC filter is enabled. Otherwise it is bypassed
When set, the digital Quadrature Modulator Correction (QMC) offset correction circuitry is enabled.
When set, the QMC phase and gain correction circuitry is enabled.
When set, the Full Mixer (FMIX) is enabled. Otherwise it is bypassed.
Register name: CONFIG3 Address: 0x03, Default 0x00
Bit 7
Bit 6
Bit 5
diffclk_dly(1:0)
0
:
0
Bit 2
0
0
0
Bit 1
Bit 0
Reserved(3:0)
0
0
To allow for a wider range of interfacing, the differential input clock (CLK1/CLK1C) has programmable delay
added to its tree.
diffclk_dly
00
01
10
11
clko_dly(1:0)
Bit 3
clko_dly(1:0)
0
diffclk_dly(1:0)
Bit 4
:
Approximate additional delay
0
1.0 ns
2.0 ns
3.0 ns
Same as above except these bits effect the single ended or internally generated clock
Register name: CONFIG4 Address: 0x04, Default 0x00
Bit 7
Bit 6
ser_dac_ data_ena
0
ser_dac_data_ena
output_delay(1:0)
B_equals_A
A_equals_B
:
:
:
:
:
:
Bit 4
B_equals_A
0
Bit 3
A_equals_B
0
Bit 2
Bit 1
Bit 0
unused
0
reva
0
revb
0
Muxes the ser_dac_data(15:0) to both DACs when asserted.
Delays the output to both DACs from 0 to 3 DAC clock cycles
When set, the DACA data is driving the DACB output.
When set, the DACB data is driving the DACA output.
Bit 4
B_equals_A
0
0
1
1
reva
revb
Bit 5
output_delay(1:0)
0
0
Bit 3
A_equals_B
0
1
0
1
DACB
Output
B data
B data
A data
A data
DACA
Output
A data
B data
A data
B data
Description
Normal Output
Both DACs driven by B data
Both DACs driven by A data
Swapped Output
Reverse the input bits of the A input port. MSB becomes LSB.
Reverse the input bits of the B input port. MSB becomes LSB
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Register name: CONFIG5 Address: 0x05, Default 0x22
Bit 7
sif4
0
Bit 6
sif_ sync_sig
0
sif4
sif_sync_sig
clkdiv_sync_ena
clkdiv_sync_sel
clkdiv_shift
:
:
:
:
:
mixer_gain
:
Bit 5
Bit 4
clkdiv_sync_ena
1
clkdiv_sync_sel
0
Bit 3
Reserved
0
Bit 2
clkdiv_shift
0
Bit 1
mixer_gain
1
Bit 0
unused
0
When set, the serial interface (SIF) is a 4 bit interface, otherwise it is a 3 bit interface.
SIF created sync signal. Set to ‘1’ to cause a sync and then clear to ‘0’ to remove it.
Enables syncing of the clock divider using the sync or TXENABLE pins when the bit is asserted.
Selects the input pin to sync the clock dividers. (0 = SYNC, 1 = TXENABLE)
When set, a rising edge on the selected sync (see clkdiv_sync_sel) for the clock dividers will cause a slip in the
synchronous counter by 1T and is useful for multi-DAC time alignment.
When set, adds 6dB to the mixer gain output.
Register name: CONFIG6 Address: 0x06, Default 0x00 (Synced)
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
0
0
0
Bit 2
Bit 1
Bit 0
0
0
0
phaseoffset(7:0)
0
phaseoffset(7:0)
0
:
0
0
0
See CONFIG7 below.
Register name: CONFIG7 Address: 0x07, Default 0x00 (Synced)
Bit 7
Bit 6
0
phaseoffset(15:8)
0
:
Bit 5
0
Bit 4
Bit 3
phaseoffset(15:8)
0
0
This is the phase offset added to the NCO accumulator just before generation of the SIN and COS values. The
phase offset is added to the upper 16bits of the NCO accumulator results and these 16 bits are used in the
sin/cosine lookup tables.
Register name: CONFIG8 Address: 0x08, Default 0x00 (Synced)
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
0
0
0
Bit 2
Bit 1
Bit 0
0
0
0
Bit 2
Bit 1
Bit 0
0
0
0
phaseadd(7:0)
0
phaseadd(7:0)
0
:
0
0
0
See CONFIG11 below.
Register name: CONFIG9 Address: 0x09, Default 0x00 (Synced)
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
phaseadd(15:8)
0
phaseadd(15:8)
0
:
0
0
0
See CONFIG11 below.
Register name: CONFIG10 Address: 0x0A, Default 0x00 (Synced)
Bit 7
Bit 6
0
phaseadd(23:16)
18
0
:
Bit 5
0
Bit 4
Bit 3
phaseadd(23:16)
0
0
See CONFIG11 below.
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Register name: CONFIG11 Address: 0x0B, Default 0x00 (Synced)
Bit 7
Bit 6
0
phaseadd(31:24)
0
:
Bit 5
0
Bit 4
Bit 3
phaseadd(31:24)
0
0
Bit 2
Bit 1
Bit 0
0
0
0
The phaseadd(31:24) value is used to determine the frequency of the NCO. The two’s complement formatted value
can be positive or negative and the LSB is equal to Fs/(2^32).
Register name: CONFIG12 Address: 0x0C, Default 0x00 (Synced)
Bit 7
Bit 6
Bit 5
Bit 4
0
0
0
0
Bit 3
Bit 2
Bit 1
Bit 0
0
0
0
qmc_gaina(7:0)
qmc_gaina(7:0)
:
0
Lower 8 bits of the 11-bit Quadrature Modulator Correction (QMC) gain word for DACA. The upper 3 bits are in
the CONFIG15 register. The full 11-bit qmc_gaina(10:0) word is formatted as UNSIGNED with a range of 0 to
1.9990 and the default gain is 1.0000. The implied decimal point for the multiplication is between bit 9 and bit 10.
Refer to formatting reference below.
qmc_gaina(10:0)
[Binary]
00000000000
00000000001
…..
01111111111
10000000000
10000000001
…..
11111111111
qmc_gaina(10:0)
[Decimal]
Format
Gain Value
0
0 + 0/1024 =
1
0 + 1/1024 =
…..
1023 0 + 1023/1024 =
[Default] 1024
1 + 0/1024 =
1025
1 + 1/1024 =
…..
2047 1 + 1023/1024 =
0.0000000
0.0009766
….
0.9990234
1.0000000
1.0009766
….
1.9990234
Register name: CONFIG13 Address: 0x0D, Default 0x00 (Synced)
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
0
0
0
qmc_gainb(7:0)
0
qmc_gainb(7:0)
0
0
0
0
: Lower 8 bits of the 11-bit QMC gain word for DACB. The upper 3 bits are in CONFIG15 register. Refer to CONFIG12
above for formatting.
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Register name: CONFIG14 Address: 0x0E, Default 0x00 (Synced)
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
0
0
0
qmc_phase(7:0)
0
0
qmc_phase(7:0)
:
0
0
0
Lower 8 bits of the 10-bit Quadrature Modulator Correction (QMC) phase word. The upper 2 bits are in the
CONFIG15 register. The full 11-bit qmc_phase(9:0) correction word is formatted as two’s complement and
scaled to occupy a range of –0.125 to 0.12475 and a default phase correction 0.00. To accomplish QMC phase
correction, this value is multiplied by the current ‘Q’ sample, then summed into the ‘I’ sample. Refer to formatting
reference below.
qmc_phase(9:0)
[Binary]
10000000000
10000000001
…..
11111111111
00000000000
00000000001
…..
01111111111
qmc_phase(9:0)
[Decimal]
–512
–511
…..
–1
[Default] 0
1
…..
511
Format
(–1 + 0/512) / 8 =
(–1 + 1/512) / 8 =
(–1 + 511/512) / 8 =
(+0 + 0/512) / 8 =
(+0 + 1/512) / 8 =
(+0 + 511/512) / 8 =
Phase
Correction
–0.1250000
–0.1234559
….
–0.0002441
+0.0000000
+0.0002441
….
+0.1247559
Register name: CONFIG15 Address: 0x0F, Default 0x24 (Synced)
Bit 7
Bit 6
Bit 5
Bit 4
1
qmc_gaina(10:8)
0
qmc_phase(9:8)
0
qmc_phase(9:8)
qmc_gaina(10:8)
qmc_gainb(10:8)
0
:
:
:
Bit 3
0
Bit 2
Bit 1
Bit 0
1
qmc_gainb(10:8)
0
0
Upper 2 bits of qmc_phase term. Defaults to zero.
Upper 3 bits of qmc_gaina term. Defaults to unity gain.
Upper 3 bits of the qmc_gainb term. Defaults to unity gain.
Register name: CONFIG16 Address: 0x10, Default 0x00 (Synced)
Bit 7
0
Bit 6
Bit 5
0
0
qmc_offseta(7:0)
:
Bit 4
Bit 3
qmc_offseta(7:0)
0
0
Bit 2
Bit 1
Bit 0
0
0
0
Lower 8 bits of the DACA offset correction. The upper 5 bits are in CONFIG18 register. The offset is
measured in DAC LSBs.
Register name: CONFIG17 Address: 0x11, Default 0x00 (Synced)
Bit 7
0
Bit 6
0
Bit 5
0
Bit 4
Bit 3
qmc_offsetb(7:0)
0
0
Bit 2
Bit 1
Bit 0
0
0
0
qmc_offsetb(7:0) : Lower 8 bits of the DACB offset correction. The upper 5 bits are in
CONFIG19 register. The offset is measured in DAC LSBs.
20
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Register name: CONFIG18 Address: 0x12, Default 0x00 (Synced)
Bit 7
Bit 6
Bit 5
0
qmc_offseta(12:8)
0
0
qmc_offseta(12:8)
Bit 4
0
Bit 3
Bit 2
Bit 1
Bit 0
0
unused
0
unused
0
unused
0
: Upper 5 bits of the DACA offset correction.
Register name: CONFIG19 Address: 0x13, Default 0x00 (Synced)
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
0
0
qmc_offsetb(12:8)
0
0
0
unused
0
unused
0
unused
0
Bit 2
Bit 1
Bit 0
0
0
0
qmc_offsetb(12:8)
: Upper 5 bits of the DACB offset correction.
Register name: CONFIG20 Address: 0x14, Default 0x00
Bit 7
Bit 6
0
Bit 5
0
Bit 4
0
ser_dac_data(7:0)
:
Bit 3
ser_dac_data(7:0)
0
0
Lower 8 bits of the serial interface controlled DAC value. This data is routed to both DACs when enabled
via ser_dac_data_ena in CONFIG4. Value is expected in 2s complement format.
Register name: CONFIG21 Address: 0x15, Default 0x00
Bit 7
Bit 6
Bit 5
0
0
0
ser_dac_data(15:8)
:
Bit 4
Bit 3
ser_dac_data(15:8)
0
0
Bit 2
Bit 1
Bit 0
0
0
0
Upper 8 bits of the serial interface controlled DAC value. This data is routed to both DACs when enabled via
ser_dac_data_ena in CONFIG4. Value is expected in 2's complement format.
Register name: CONFIG22 Address: 0x16, Default 0x15
Bit 7
Bit 6
Bit 5
nco_sel(1:0)
0
nco_sel(1:0)
nco_reg_sel(1:0)
qmcorr_reg_sel(1:0)
qmoffsest_reg_sel(1:0)
Bit 4
Bit 3
nco_reg_sel(1:0)
0
1
0
:
:
:
:
Selects the
Selects the
Selects the
Selects the
signal
signal
signal
signal
*_sel (1:0)
00
01
10
11
to use
to use
to use
to use
as
as
as
as
Bit 2
qmcorr_reg_sel(1:0)
0
1
the
the
the
the
sync
sync
sync
sync
for
for
for
for
Bit 1
Bit 0
qmoffset_reg_sel(1:0)
0
1
the NCO accumulator.
loading the NCO registers.
loading the QM correction registers.
loading the QM offset correction registers.
Sync selected
TXENABLE from FIFO output
SYNC from FIFO output
sync_SIF_sig (via CONFIG5)
Always zero
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Register name: CONFIG23 Address: 0x17, Default 0x15
Bit 7
Bit 6
unused
0
unused
0
fifo_sel(2:0)
:
Bit 5
Bit 4
0
fifo_sel(2:0)
1
:
Bit 2
Bit 1
Bit 0
0
aflag_ sel
1
unused
0
unused
1
Selects the sync source for the FIFO from the table below. For the case where the sync is dependent on the first
transition of the input data MSB: Once the transition occurs, the only way to get another sync it to reset the device
or to program fifo_sel to another value
fifo_sel (2:0)
000
001
010
011
100
101
110
111
aflag_sel
Bit 3
Sync selected
TXENABLE from pin
SYNC from pin
sync_SIF_sig (via CONFIG5)
Always zero
1st transition on DA MSB
1st transition on DB MSB
Always zero
Always one
When set, the MSB of the input opposite of incoming data is used to determine the A sample. When cleared,
rising edge of TXENABLE is used. Refer to Figure 37.
Register name: CONFIG24 Address: 0x18, Default 0x80
Bit 7
Bit 6
Bit 5
fifo_sync_strt(3:0)
0
0
1
fifo_sync_strt(3:0)
:
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
0
Unused
0
Unused
0
Unused
0
Unused
0
When the sync to the FIFO occurs, this is the value loaded into the FIFO output position counter. With this
value the initial difference between input and output pointers can be controlled. This may be helpful in
syncing multiple chips or controlling the delay through the device.
Register name: CONFIG25 Address: 0x19, Default 0x00
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
Unused
0
Unused
0
Unused
0
Unused
0
Unused
0
Unused
0
Unused
0
Unused
0
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
io_1p8_3p3
0
Unused
0
sleepb
0
sleepa
0
isbiaslpfb_a
1
isbiaslpf_b
1
PLL_ sleep
0
PLL_ena
1
Register name: CONFIG26 Address: 0x1A, Default 0x0D
io_1p8_3p3
:
sleepb
:
sleepa
:
isbiaslpfb_a
:
isbiaslpfb_b
:
PLL_sleep
PLL_ena
:
:
22
Used to program the digital input voltage threshold levels. ‘0’=3.3V tolerate pads and ‘1’=1.8V tolerate pads.
Applies to following digital pins: CLKO_CLK1, LOCK_CLK1C, DA[15:0], DB[15:0], SYNC, RESETB, SCLK,
SDENB, SDIO (input only) and TXENABLE.
When set, DACB is put into sleep mode. Putting the DAC into single DAC mode does not automatically assert this
signal, so for minimum power in single DAC mode, also program this register bit.
When set, DACA is put into sleep mode. Note: If DACA channel is in sleep mode (sleepa = '1') the DACB channel
is also forced in to sleep mode.
Turns on the low pass filter for the current source bias in the DACA when cleared. The low pass filter will set a
corner at ~472kHz when low and ~95 kHz when high.
Turns on the low pass filter for the current source bias in the DACB when cleared. The low pass filter will set a
corner at ~472kHz when low and ~95 kHz when high.
When set, the PLL is put into sleep mode. Bypassing the PLL does not automatically but it into sleep mode.
When set, the PLL is on and its output clock is being used as the DAC clock.
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Register name: CONFIG27 Address: 0x1B, Default 0xFF
Bit 7
Bit 6
Bit 5
Bit 4
coarse_daca(3:0)
1
1
1
coarse_daca(3:0)
:
:
1
1
Bit 2
Bit 1
coarse_dacb(3:0)
1
1
Bit 0
1
Scales the output current is 16 equal steps.
V EXTIO
Rbias
coarse_dacb(3:0)
Bit 3
(DACA_gain ) 1)
Same as above except for DACB.
Register name: CONFIG28 Address: 0x1C, Default 0x00
Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
Reserved
0
Reserved
0
Reserved
0
Reserved
0
Reserved
0
Reserved
0
Reserved
0
Reserved
0
Bit 7
Bit 6
Bit 5
Bit 2
Bit 1
Bit 0
0
PLL_m(4:0)
0
0
PLL_n(2:0)
0
0
Register name: CONFIG29 Address: 0x1D, Default 0x00
0
Bit 4
0
Bit 3
0
PLL_m
:
M portion of the M/N divider of the PLL thermometer encoded:
PLL_m(4:0)
M value
00000
1
00001
2
00011
4
00111
8
01111
16
11111
32
All other values
Invalid
PLL_n
:
N portion of the M/N divider of the PLL thermometer encoded. If supplying a high rate CLK2/C frequency, the PLL_n value
should be used to divide down the input CLK2/C to maintain a maximum PFD operating of 160 MHz.
PLL_n(2:0)
n value
000
1
001
2
011
4
111
8
All other values
Invalid
PLL Function:
ƒ vco +
ƫ
ƪ(M)
(N)
ƒ ref
where ƒref is the frequency of the external DAC clock input on the CLK2/C pins
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Register name: CONFIG30 Address: 0x1E, Default 0x00
Bit 7
Bit 6
PLL_LPF_ reset
0
VCO_div2
0
PLL_LPF_reset
VCO_div2
:
:
PLL_gain(1:0)
:
PLL_range(3:0)
:
24
Bit 5
Bit 4
Bit 3
Bit 2
PLL_gain(1:0)
0
Bit 1
Bit 0
PLL_range(3:0)
0
0
0
0
0
When set, can be used to hold the PLL loop filter at 0 volts.
When set, the PLL CLOCK output is 1/2 the PLL VCO frequency. Used to run the VCO at 2X the desired clock
frequency to reduce phase noise for lower DAC clock rates.
Used to adjust the PLL’s Voltage Controlled Oscillator (VCO) gain, KVCO. Refer to the Electrical Characteristics
table. By increasing the PLL_gain, the VCO can cover a broader range of frequencies; however, the higher gain
also increases the phase noise of the PLL. In general, lower PLL_gain settings result in lower phase noise. The
KVCO of the VCO can also affect the PLL stability and is used to determine the loop filter components. See section
on determining the PLL filter components for more detail.
Used to adjust the bias current of the VCO. By increasing the bias current, the oscillator can reach higher
frequencies. Refer to the Electrical Characteristics table.
'0000' – minimum bias current and lowest VCO frequency range
'1111' – maximum bias current and highest VCO frequency range
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SLLS880C – DECEMBER 2007 – REVISED AUGUST 2010
DETAILED DESCRIPTION
EXAMPLE SYSTEM DIAGRAM
DAC5688 DAC
DAC
I-Signal
Term
InvSINC
QMC
(Gain/Phase)
NCO/Mixer
FIFO &
Demux
DB[15:0]
16
I/Q FIR1-3
(8x)
DAC
Antenna
LPF
5V
PA
Q-Signal
Term
LPF
To TX
Feedback
TXENABLE
Clock, Sync & Control
CLK1
90
opt.
PLL
CLK2
CLK2C
Digital
Up
Converter
(DUC)
TRF3703 AQM
5V
DA[15:0]
16
76.8 MHz
Loop
Filter
To RX
Path
0
~ 2.1 GHz
TRF3761- X PLL /VCO
Div
1/2/4
100
CK
76.8 MHz
614.4 MHz
10 MHz
REF
OSC
Term
CDCM 7005
÷8
÷8
Status &
Control
REF_IN
÷1
Clock Divider /
Distribution
Duplexer
GC5016
PLL
Synth
Loop
Filter
VCO
NDivider
VCTRL_IN
Loop
Filter
RDiv
PFD
Charge
Pump
CPOUT
Status& Control
VCXO
614.4 MHz
Figure 19. Example System Diagram: Direct Conversion with 8x interpolation
SERIAL INTERFACE
The serial port of the DAC5688 is a flexible serial interface which communicates with industry standard
microprocessors and microcontrollers. The interface provides read/write access to all registers used to define the
operating modes of DAC5688. It is compatible with most synchronous transfer formats and can be configured as
a 3 or 4 pin interface by SIF4 in register CONFIG5. In both configurations, SCLK is the serial interface input
clock and SDENB is serial interface enable. For 3 pin configuration, SDIO is a bidirectional pin for both data in
and data out. For 4 pin configuration, SDIO is data in only and SDO is data out only. Data is input into the device
with the rising edge of SCLK. Data is output from the device on the falling edge of SCLK.
Each read/write operation is framed by signal SDENB (Serial Data Enable Bar) asserted low for 2 to 5 bytes,
depending on the data length to be transferred (1–4 bytes). The first frame byte is the instruction cycle which
identifies the following data transfer cycle as read or write, how many bytes to transfer, and what address to
transfer the data. Table 2 indicates the function of each bit in the instruction cycle and is followed by a detailed
description of each bit. Frame bytes 2 to 5 comprise the data transfer cycle.
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Table 2. Instruction Byte of the Serial Interface
Bit
Description
R/W
[N1 : N0]
7
6
5
4
3
2
1
0
R/W
N1
N0
A4
A3
A2
A1
A0
Identifies the following data transfer cycle as a read or write operation. A high indicates a read operation from DAC5688 and
a low indicates a write operation to DAC5688.
Identifies the number of data bytes to be transferred per Table 3. Data is transferred MSB first.
Table 3. Number of Transferred Bytes Within One
Communication Frame
[A4 : A0]
N1
N0
Description
0
0
Transfer 1 Byte
0
1
Transfer 2 Bytes
1
0
Transfer 3 Bytes
1
1
Transfer 4 Bytes
Identifies the address of the register to be accessed during the read or write operation. For multi-byte transfers, this address
is the starting address. Note that the address is written to the DAC5688 MSB first and counts down for each byte
Figure 20 shows the serial interface timing diagram for a DAC5688 write operation. SCLK is the serial interface
clock input to DAC5688. Serial data enable SDENB is an active low input to DAC5688. SDIO is serial data in.
Input data to DAC5688 is clocked on the rising edges of SCLK.
Data Transfer Cycle(s)
Instruction Cycle
SDENB
SCLK
SDIO
r/w N1
N0 A4
A3 A2
A1
A 0 D7 D6
t s (SDENB)
D5 D4
D3 D2 D1
D0
t SCLK
SDENB
SCLK
SDIO
t h ( SDIO)
t s (SDIO)
t SCLKH t SCLKL
Figure 20. Serial Interface Write Timing Diagram
Figure 21 shows the serial interface timing diagram for a DAC5688 read operation. SCLK is the serial interface
clock input to DAC5688. Serial data enable SDENB is an active low input to DAC5688. SDIO is serial data in
during the instruction cycle. In 3 pin configuration, SDIO is data out from DAC5688 during the data transfer
cycle(s), while SDO is in a high-impedance state. In 4 pin configuration, SDO is data out from DAC5688 during
the data transfer cycle(s). The SDIO/SDO data is output on the falling edge of SCLK. At the end of the data
transfer, SDO will output low on the final falling edge of SCLK until the rising edge of SDENB when it will 3-state.
26
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Instruction Cycle
SDENB
Data Transfer Cycle(s)
SCLK
SDIO
r/w N1 N0
A3 A2 A1 A0 D7 D6 D5 D4 D3 D2 D1 D0 0
-
D7 D6 D5 D4 D3 D2 D1 D0 0
SDO
4 pin configuration 3 pin configuration
output
output
SDENB
SCLK
SDIO
SDO
Data n
Data n-1
t d (Data)
Figure 21. Serial Interface Read Timing Diagram
FIR FILTERS
Figure 22 shows the magnitude spectrum response for FIR1, a 67-tap interpolating half-band filter. The transition
band is from 0.4 to 0.6 × fIN (the input data rate for the FIR filter) with <0.002-dB of pass-band ripple and > 80-dB
stop-band attenuation. Figure 23 shows the transition band region from 0.37 to 0.47 × fIN. Up to 0.458 × fIN there
is less than 0.5 dB of attenuation.
Figure 24 shows the magnitude spectrum response for the 19-tap FIR2 filter. The transition band is from 0.25 to
0.75 × fIN (the input data rate for the FIR filter). For 4x interpolation modes, the composite filter response is
shown in Figure 25.
Figure 26 shows the magnitude spectrum response for the 11-tap FIR3 filter. For 8x interpolation modes, the
composite filter response is shown in Figure 27.
The DAC5688 also has a 9-tap non-interpolating inverse sinc filter (FIR4) running at the DAC update rate (fDAC)
that can be used to flatten the frequency response of the sample and hold output. The DAC sample and hold
output set the output current and holds it constant for one DAC clock cycle until the next sample, resulting in the
well known sin(x)/x or sinc(x) frequency response shown in Figure 28 (red dash-dotted line). The inverse sinc
filter response (Figure 28, blue dashed line) has the opposite frequency response between 0 to 0.4 × fDAC,
resulting in the combined response (Figure 28, green solid line). Between 0 to 0.4 × fDAC, the inverse sinc filter
compensates the sample and hold rolloff with less than 0.03-dB error.
The inverse sinc filter has a gain > 1 at all frequencies. Therefore, the signal input to FIR4 must be reduced from
full scale to prevent saturation in the filter. The amount of backoff required depends on the signal frequency, and
is set such that at the signal frequencies the combination of the input signal and filter response is less than 1 (0
dB). For example, if the signal input to FIR4 is at 0.25 × fDAC, the response of FIR4 is 0.9 dB, and the signal must
be backed off from full scale by 0.9 dB. The gain function in the QMC block can be used to set reduce amplitude
of the input signal. The advantage of FIR4 having a positive gain at all frequencies is that the user is then able to
optimized backoff of the signal based on the signal frequency.
The filter taps for all digital filters are listed in Table 4. Note that the loss of signal amplitude may result in lower
SNR due to decrease in signal amplitude.
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Magnitude Spectrum for FIR1
Magnitude Spectrum for FIR1
0.1
20
0
0
-20
-0.1
-40
dB
dB
-60
-80
-0.2
-0.3
-100
-0.4
-120
-0.5
-140
-160
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
0.37
1.0
0.38
0.39
0.4
f/Fin
Figure 22. Magnitude Spectrum for FIR1
0.41
0.42
f/Fin
0.43
0.44
0.45
0.46
0.47
Figure 23. FIR1 Transition Band
vertical spacer
vertical spacer
vertical spacer
4x Interpolation Composite Filtering Response
Magnitude Spectrum for FIR 2
20
0
0
-20
-20
-40
dB
-40
dB
-60
-60
-80
-80
-100
-100
-120
-120
-140
-140
-160
0
-160
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
0.5
1
f/Fin
Figure 24. Magnitude Spectrum for FIR2
28
1
f/Fin
2
Figure 25. 4x Interpolation Composite Response
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8x Interpolation Composite Filtering Response
Magnitude Spectrum for FIR 3
20
0
0
-20
-20
-40
-40
-60
dB
dB
-60
-80
-80
-100
-100
-120
-120
-140
-140
-160
0
-160
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
0.5
1
2
f/Fin
1
f/Fin
Figure 26. Magnitude Spectrum for FIR3
vertical spacer
vertical spacer
3
4
Figure 27. 8x Interpolation Composite Response
FIR 4 Inverse Corrected Spectrum
5
4
3
2
dB
1
0
-1
-2
-3
-4
-5
0
0.05
0.1
0.15
0.2
0.25 0.3
f/fDAC
0.35
0.4
0.45
0.5
Figure 28. Magnitude Spectrum for FIR4
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Table 4. FIR Filter Coefficients
2X Interpolating Half-band Filters
2
30
FIR1
FIR2
FIR3
FIR4
67 Taps
19 Taps
11 Taps
9 Taps
2
9
9
31
31
1
1
–4
0
0
0
0
0
0
-4
–5
–5
–58
–58
–219
–219
13
13
0
0
0
0
0
0
–50
–50
11
11
214
214
1212
1212
592 (1)
0
0
0
0
–21
–21
–638
–638
0
0
0
0
37
37
0
0
–61
–61
0
0
97
97
0
0
–148
–148
0
0
218
218
0
0
–314
–314
0
0
444
444
0
0
–624
–624
0
0
877
877
0
0
–1260
–1260
0
0
1916
1916
0
0
–3372
–3372
0
0
10395
10395
16384
(1)
Non-Interpolating
Inverse-SINC Filter
2521
4096
2048
(1)
2521
(1)
(1)
Center Taps are highlighted in BOLD.
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Full Complex Mixer (FMIX)
The full complex Mixer (FMIX) block uses a Numerically Controlled Oscillator (NCO) with a 32-bit frequency
register phaseadd(31:0) and a 16-bit phase register phaseoffset(15:0) to provide sin and cos for mixing. The
NCO tuning frequency is programmed in CONFIG8 through CONFIG11 registers. Phase offset is programmed in
CONFIG6 and CONFIG7 registers. A block-diagram of the NCO is shown below in Figure 29.
32
16
Frequency
Register
32
32
Accumulator
CLK
32
16
16
sin
Look Up
Table
RESET
16
cos
16
FDAC
NCO SYNC
via
nco_sel(1:0)
Phase
Register
Figure 29. Block-Diagram of the NCO
Synchronization of the NCO occurs by resetting the NCO accumulator to zero. The synchronization source is
selected by CONFIG22 nco_sel(1:0). Frequency word fref in the phaseadd register is added to the accumulator
every clock cycle, fDAC. The output frequency of the NCO is
ƒref ƒNCO_CLK
ƒ NCO +
2 32
(1)
Treating channels A and B as a complex vector I + I×Q where I(t) = A(t) and Q(t) = B(t), the output of FMIX
IOUT(t) and QOUT(t) is
I OUT(t) + ǒI IN(t) cosǒ2pf NCOt ) dǓ * Q IN(t) sinǒ2pf NCOt ) dǓǓ
2 (mixer_gain*1)
+
(2)
(3)
Where t is the time since the last resetting of the NCO accumulator, d is the phase offset value and mixer_gain
is either 0 or 1. d is given by:
d + 2p phase(15 : 0)ń216
(4)
The maximum output amplitude of FMIX occurs if IIN(t) and QIN(t) are simultaneously full scale amplitude and the
sine and cosine arguments 2pfNCOt + d (2N-1)×p/4 (N = 1, 2, ...).
With CONFIG5 mixer_gain = 0, the gain through FMIX is sqrt(2)/2 or –3 dB. This loss in signal power is in most
cases undesirable, and it is recommended that the gain function of the QMC block be used to increase the signal
by 3 dB to compensate. With mixer_gain = 1, the gain through FMIX is sqrt(2) or + 3 dB, which can cause
clipping of the signal if IIN(t) and QIN(t) are simultaneously near full scale amplitude and should therefore be used
with caution.
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Quadrature Modulator Correction (QMC)
The Quadrature Modulator Correction (QMC) block provides a means for adjusting the gain and phase of the
complex signal. At a quadrature modulator output, gain and phase imbalances result in an undesired sideband
signal.
The block diagram for the QMC is shown in Figure 30. The QMC block contains 3 programmable parameters:
qmc_gaina(10:0), qmc_gainb(10:0) and qmc_phase(9:0).
Registers qmc_gaina(10:0) and qmc_gainb(10:0) control the I and Q path gains and are 11 bit values with a
range of 0 to approximately 2. This value is used to scale the signal range. Register qmc_phase(9:0) controls the
phase imbalance between I and Q and is a 10-bit value that ranges from –1/8 to approximately +1/8. This value
is multiplied by each Q sample then summed into the I sample path. This operation is a simplified approximation
of a true phase rotation and covers the range from –7.5 to +7.5 degrees in 1024 steps.
qmc_gaina(10:0)
11
I(t)
X
S
10
X
Q(t)
qmc _phase (9:0)
X
11
qmc_gain b (10:0)
Figure 30. QMC Block Diagram
DAC Offset Control
The qmc_offseta(12:0) and qmc_offsetb(12:0) values can be used to independently adjust the I and Q path DC
offsets. Both offset values are in represented in 2s-complement format with a range from –4096 to 4095.
The offset value adds a digital offset to the digital data before digital-to-analog conversion. Since the offset is
added directly to the data it may be necessary to back off the signal to prevent saturation. Both data and offset
values are LSB aligned.
qma_offset
{-4096, - 4095, … , 4095 }
13
I
S
Q
S
13
qmb_offset
{-4096, - 4095, … , 4095 }
Figure 31. DAC Offset Block
32
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CLOCK MODES
The DAC5688 supports several different clocking modes for generating the internal clocks for the logic and DAC.
The clocking modes are selected by programming the register bits below and summarized in Table 5.
Register
Control Bits
CONFIG1
synchr_clkin
CONFIG2
clk1_in_ena, clk1c_in_ena, diffclk_ena
CONFIG26
PLL_ena
Table 5. Summary of Clock Modes and Options
CLKO_
CLK1
I/O
synchr_clkin
clk1_in_en
Programming Bits
clk1c_in_ena
diffclk_ena
PLL_ena
1
1
1
0
1
X
0
0
0
1
1
1
0
Input
0
1
X
0
0
Output
0
0
X
0
0
Clocking Mode
Option
Dual Synchronous Clock Mode
Diff. CLK1
Input
1
S/E CLK1
Input
1
Diff. CLK1
Input
S/E CLK1
Dual Clock Mode
External Clock Mode
CLKO
PLL Clock Mode
Diff. CLK1
Input
0
1
1
1
1
S/E CLK1
Input
0
1
X
0
1
Output
0
0
X
0
1
CLKO
DUAL SYNCHRONOUS CLOCK MODE
In DUAL SYNCHRONOUS CLOCK MODE, the user provides the CLK2/C clock signal at the DAC sample rate
and also provides a divided down CLK1 at the input data rate. The CLK1 signal can be differential or
single-ended. Refer to Figure 16 for the timing diagram. In this mode the relationship between CLK2 and CLK1
(t_align) is critical and used as a synchronizing mechanism for the internal logic. This facilitates multi-DAC
synchronization by using dual external clock inputs CLK1 and CLK2 while FIFO data is always written and read
from location zero. It is highly recommended that a clock synchronizer device such as the CDCM7005 provide
both CLK2/C and CLK1/C inputs. Although CLK1 could be single-ended it is recommended to use a differential
clock to ensure proper skews between the two clock inputs.
DUAL CLOCK MODE
In DUAL CLOCK MODE, the user provides the CLK2/C clock signal at the DAC sample rate and also provides a
divided down CLK1 at the input data rate. The CLK1 signal can be differential or single-ended. Refer to Figure 32
for the timing diagram. Unlike the DUAL SYNCHRONOUS CLOCK MODE, the t_align parameter is not critical
because these clocks are not used as a synchronizing mechanism for the internal logic and the FIFO is used as
an elastic buffer for the data. Synchronizing in this mode is provided by separate control inputs.
CLK 2
D < t _align (only in dual synchronous clock mode)
CLK 1
DA [0 : 15 ]
DB [0 : 15 ]
ts
th
Figure 32. DUAL (SYNCHRONOUS) CLOCK MODE Timing Diagram
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EXTERNAL CLOCK MODE
In EXTERNAL CLOCK MODE, the user provides a clock signal at the DAC output sample rate through CLK2/C.
The CLKO_CLK1 pin is configured as an output in this mode and will toggle at a required frequency for the
configured interpolation rate and data mode. The CLKO_CLK1 clock can be used to drive the input data source
(such as digital upconverter) that sends the data to the DAC. Note that the CKO_CLK1 delay relative to the input
CLK2 rising edge (td(CLKO) in Figure 33) will increase with increasing loads.
CLK 2
t d(CLKO)
CLKO _ CLK 1
(output )
DA [0 : 15 ]
DB [0 : 15 ]
ts
th
Figure 33. EXTERNAL CLOCK MODE Timing Diagram
PLL CLOCK MODE
In PLL CLOCK MODE, the user provides an external reference clock to the CLK2/C input pins. Refer to
Figure 34. An internal clock multiplying PLL uses the lower-rate reference clock to generate a high-rate clock for
the DAC. This function is very useful when a high-rate clock is not already available at the system level;
however, the internal VCO phase noise in PLL Clock Mode may degrade the quality of the DAC output signal
when compared to an external low jitter clock source.
CLK0_CLK1
(input or output)
DA [0 :15 ]
DB [0 :15 ]
ts
th
Figure 34. PLL CLOCK MODE Timing Diagram
The internal PLL has a type four phase-frequency detector (PFD) comparing the CLK2/C reference clock with a
feedback clock to drive a charge pump controlling the VCO operating voltage and maintaining synchronization
between the two clocks. An external low-pass filter is required to control the loop response of the PLL. See the
Low-Pass Filter section for the filter setting calculations. This is the only mode where the LPF filter applies.
The input reference clock N-Divider is selected by CONFIG29 PLL_n(2:0) for values of ÷1, ÷2, ÷4 or ÷8. The
VCO feedback clock M-Divider is selected by CONFIG29 PLL_m(4:0) for values of ÷1, ÷2, ÷4, ÷8, ÷16 or ÷32.
The combination of M-Divider and N-Divider form the clock multiplying ratio of M/N. If the reference clock
frequency is greater than 160MHz, use a N-Divider of ÷2, ÷4 or ÷8 to avoid exceeding the maximum PFD
operating frequency.
For DAC sample rates less than the maximum VCO operating frequency of 910/2 or 455 MHz. The phase noise
of PLL may improved by using the output divider via CONFIG30 VCO_div2. If not using the PLL, clear
CONFIG26 PLL_ena and set CONFIG26 PLL_sleep to reduce power consumption. In some cases, it may be
useful to reset the VCO control voltage by toggling CONFIG30 PLL_LPF_reset.
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(Pin 64)
External
Loop
Filter
LPF
(3.3V, Pin 9)
IOVDD
(1.8V, Pin 1)
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PLL Bypass
Clock Multiplying PLL
CLK2
FREF
CLK2C
N–Divider FREF /N
(1, 2, 4, 8)
PFD
FVCO /M
FVCO
VCO
Charge
Pump
To internal
DAC clock
distribution
FPLL
M- Divider
( 1,2,4,8,16,32)
FVCO
FVCO /2
÷2
PLL Sleep
PLL_sleep
(CONFIG26)
PLL_gain (1:0),
PLL_range(3:0)
(CONFIG30)
PLL_m (4:0)
(CONFIG29)
PLL_n (2:0)
(CONFIG29)
VCO_div2
(CONFIG11)
PLL_ena
(CONFIG26)
PLL_LPF_reset
(CONFIG30)
Figure 35. Functional Block Diagram for PLL
DATA BUS MODES
The DAC5688 supports three DATA BUS MODES:
1. DUAL BUS MODE
2. INTERLEAVED BUS MODE
3. HALF RATE BUS MODE
DUAL BUS MODE
In DUAL BUS MODE, the user inputs data on both DA[15:0] and DB[15:0] ports. This mode is selected by setting
CONFIG1 insel_mode(1:0) = ‘00’. Refer to Figure 36.
CLK1
DA[15:0]
A0
A1
A2
A3
AN
AN+1
DB[15:0]
B0
B1
B2
B3
BN
BN+1
Figure 36. DUAL BUS MODE (Dual Clock Mode)
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INTERLEAVED BUS MODE
In INTERLEAVED BUS MODE, the user inputs dual-channel data as an interleaved single data stream to either
DA[15:] or DB[15:0] ports. The DAC5688 de-interleaves the input data stream and routes to both A and B data
paths. For input data on DA[15:0], set CONFIG1 insel_mode[15:0] = ‘01’. For input data on DB[15:0], set
CONFIG1 insel_mode[15:0] = ‘10’. In this bus mode, a separate input flag is required to distinguish an A sample
from a B sample in the interleaved data stream. This flag can either be the single event rising edge of
TXENABLE or the continuous toggling MSB of the port inactive data port. For the TXENABLE flag option, set the
CONFIG23 aflag_sel bit and the A sample will be expected to be aligned with the rising edge of TXENABLE. For
the toggling MSB option, clear the CONFIG23 aflag_sel bit and the A sample will be expected for each ‘1’ of the
MSB with the B sample is flagged for each ‘0’ of the MSB. Refer to Figure 37.
CLK1
Single event rising edge
flags “A” sample if
aflag_sel = ‘1’
TXENABLE
Toggling MSB
flags “A” sample if
aflag_sel = ‘0’
DB15
DA [15:0]
A0
B0
A1
B1
AN
BN
Figure 37. INTERLEAVED BUS MODE on DA[15:0] port (Dual Clock Mode)
HALF RATE BUS MODE
In HALF RATE BUS MODE, the user inputs data on both DA[15:0] and DB[15:0] ports at half rate and input logic
merges both data streams into one DAC channel (A). This mode is selected by setting CONFIG1
insel_mode[15:0] = ‘11’. Refer to Figure 38.
CLK 1
DA[15:0]
A0
A2
A4
A6
AN
DB[15:0]
A1
A3
A5
A7
A N+1
Figure 38. HALF RATE BUS MODE (Dual Clock Mode)
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CLK2 and CLK2C Inputs
Figure 39 shows an equivalent circuit for the DAC input clock (CLK2/C).
CLKVDD
333 W
CLK2
2 KW
Note: Input common mode level is
approximately 2/3 * CLKVDD or 1.2 V.
2 KW
CLK2C
666 W
GND
Figure 39. CLK2/C Equivalent Input Circuit
Figure 40 shows the preferred configuration for driving the CLK2/CLK2C input clock with a differential ECL/PECL
source.
0 .1 mF
+
CLK 2
Differential
ECL
or (LV)PECL
Source
C AC
-
100 W
CLK2C
0 .1 mF
150 W
RT
150 W
Figure 40. Preferred Clock Input Configuration With a Differential ECL/PECL Clock Source
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CLKO_CLK1 and LOCK_CLK1C Pins
Figure 41 shows the functionality of the CLKO_CLK1 and LOCK_CLK1C pins. Refer to Table 5. The controls for
these pins are found in the CONFIG2 register and are used in selection of device clocking mode. In single-ended
mode (CONFIG2 diffclk_ena = ‘0’) refer to Figure 43, both CLKO_CLK1 and LOCK_CLK1C pins have an
internal pull-down resistor approximately equivalent to 100kΩ.
clk1_in_ena
clko_SE_hold
EN
Internal CLKO
0
Internal CLK 1
CLKO _ CLK 1
EN
1
LOCK _ CLK 1 C
Internal LOCK
EN
clk1c_in_ena
diffclk_ena
Figure 41. CLKO_CLK1 and LOCK_CLK1C pins bi-directional control
In differential mode (CONFIG2 diffclk_ena = ‘1’) the CLKO_CLK1 and LOCK_CLK1C input pins are configured
as a differential CLK1/C clock input. Refer Figure 39 for the equivalent circuit.
IOVDD
IOVDD
10 KW
CLKO _ CLK 1
IOVDD
GND
10 KW
IOVDD
GND 10 KW
Note: Input common mode level is
approximately 0.5* IOVDD or 1.65 V.
LOCK _ CLK 1 C
10 KW
GND
GND
Figure 42. CLKO_CLK1 and LOCK_CLK1C Differential Input Mode Equivalent Circuit
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CMOS DIGITAL INPUTS
Figure 43 shows a schematic of the equivalent CMOS digital inputs of the DAC5688. SDIO, SCLK, SYNC,
TXENABLE, DA[15:0] and DB[15:0] have pull-down resistors while RESETB and SDENB have pull-up resistors
internal the DAC5688. See specification table for logic thresholds. The pull-up and pull-down circuitry is
approximately equivalent to 100kΩ.
The input switches levels for all CMOS digital inputs can be changed from 3.3V input levels to 1.8V input levers
by programming the CONFIG26 io_1p8_3p3 register bit. If io_1p8_3p3 is cleared, the input thresholds are set
for 3.3V CMOS levels. If io_1p8_3p3 is set, the input thresholds are set for 1.8V levels.
IOVDD
400 W
Internal
digital in
RESETB
SDENB
400 W
Internal
digital in
100 kW
SCLK
SYNC
TXENABLE
DA[15:0]
DB[15:0]
CLKO _CLK 1**
100 kW
IOVDD
** As an input
GND
GND
Figure 43. CMOS/TTL Digital Equivalent Input
REFERENCE OPERATION
The DAC5688 uses a bandgap reference and control amplifier for biasing the full-scale output current. The
full-scale output current is set by applying an external resistor RBIAS to pin BIASJ. The bias current IBIAS through
resistor RBIAS is defined by the on-chip bandgap reference voltage and control amplifier. The default full-scale
output current equals 16 times this bias current and can thus be expressed as:
IOUTFS = 16 × IBIAS = 16 × VEXTIO / RBIAS
Each DAC has a 4-bit independent coarse gain control via coarse_daca(3:0) and coarse_dacb (3:0) in the
CONFIG27 register. Using gain control, the IOUTFS can be expressed as:
IOUTAFS = (DACA_gain + 1) × IBIAS = (DACA_gain + 1) × VEXTIO / RBIAS
IOUTBFS = (DACB_gain + 1) × IBIAS = (DACB_gain + 1) × VEXTIO / RBIAS
where VEXTIO is the voltage at terminal EXTIO. The bandgap reference voltage delivers an accurate voltage of
1.2 V. This reference is active when terminal EXTLO is connected to AGND. An external decoupling capacitor
CEXT of 0.1 mF should be connected externally to terminal EXTIO for compensation. The bandgap reference can
additionally be used for external reference operation. In that case, an external buffer with high impedance input
should be applied in order to limit the bandgap load current to a maximum of 100 nA. The internal reference can
be disabled and overridden by an external reference by connecting EXTLO to AVDD. Capacitor CEXT may hence
be omitted. Terminal EXTIO thus serves as either input or output node.
The full-scale output current can be adjusted from 20 mA down to 2 mA by varying resistor RBIAS or changing the
externally applied reference voltage. The internal control amplifier has a wide input range, supporting the
full-scale output current range of 20 dB.
DAC TRANSFER FUNCTION
The CMOS DAC’s consist of a segmented array of NMOS current sinks, capable of sinking a full-scale output
current up to 20 mA. Differential current switches direct the current to either one of the complementary output
nodes IOUT1 or IOUT2. (DACA = IOUTA1 or IOUTA2 and DACB = IOUTB1 or IOUTB2.) Complementary output
currents enable differential operation, thus canceling out common mode noise sources (digital feed-through,
on-chip and PCB noise), dc offsets, even order distortion components, and increasing signal output power by a
factor of two.
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The full-scale output current is set using external resistor RBIAS in combination with an on-chip bandgap voltage
reference source (+1.2 V) and control amplifier. Current IBIAS through resistor RBIAS is mirrored internally to
provide a maximum full-scale output current equal to 16 times IBIAS.
The relation between IOUT1 and IOUT2 can be expressed as:
IOUT1 = – IOUTFS – IOUT2
We will denote current flowing into a node as – current and current flowing out of a node as + current. Since the
output stage is a current sink the current can only flow from AVDD into the IOUT1 and IOUT2 pins. The output
current flow in each pin driving a resistive load can be expressed as:
IOUT1 = IOUTFS × (65536 – CODE) / 65536
IOUT2 = IOUTFS × CODE / 65536
where CODE is the decimal representation of the DAC data input word.
For the case where IOUT1 and IOUT2 drive resistor loads RL directly, this translates into single ended voltages
at IOUT1 and IOUT2:
VOUT1 = AVDD – | IOUT1 | × RL
VOUT2 = AVDD – | IOUT2 | × RL
Assuming that the data is full scale (65536 in offset binary notation) and the RL is 25 Ω, the differential voltage
between pins IOUT1 and IOUT2 can be expressed as:
VOUT1 = AVDD – | –0mA | × 25 Ω = 3.3 V
VOUT2 = AVDD – | –20mA | × 25 Ω = 2.8 V
VDIFF = VOUT1 – VOUT2 = 0.5V
Note that care should be taken not to exceed the compliance voltages at node IOUT1 and IOUT2, which would
lead to increased signal distortion.
DAC OUTPUT SINC RESPONSE
Due to sampled nature of a high-speed DAC’s, the well known sin(x)/x (or SINC) response can significantly
attenuate higher frequency output signals. Refer to Figure 44 which shows the unitized SINC attenuation roll-off
with respect to the final DAC sample rate in 4 Nyquist zones. For example, if the final DAC sample rate FS = 1.0
GSPS, then a tone at 440MHz will be attenuated by 3.0dB. Although the SINC response can create challenges
in frequency planning, one side benefit is the natural attenuation of Nyquist images. The increased over-sampling
ratio of the input data provided by the DAC5688’s 2x, 4x and 8x digital interpolation modes improve the SINC
roll-off (droop) within the original signal’s band of interest.
Figure 44. Unitized DAC sin(x)/x (SINC) Response
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ANALOG CURRENT OUTPUTS
Figure 45 shows a simplified schematic of the current source array output with corresponding switches.
Differential switches direct the current of each individual NMOS current source to either the positive output node
IOUT1 or its complementary negative output node IOUT2. The output impedance is determined by the stack of
the current sources and differential switches, and is typically >300 kΩ in parallel with an output capacitance of 5
pF.
The external output resistors are referred to an external ground. The minimum output compliance at nodes
IOUT1 and IOUT2 is limited to AVDD – 0.5 V, determined by the CMOS process. Beyond this value, transistor
breakdown may occur resulting in reduced reliability of the DAC5688 device. The maximum output compliance
voltage at nodes IOUT1 and IOUT2 equals AVDD + 0.5 V. Exceeding the minimum output compliance voltage
adversely affects distortion performance and integral non-linearity. The optimum distortion performance for a
single-ended or differential output is achieved when the maximum full-scale signal at IOUT1 and IOUT2 does not
exceed 0.5 V.
AVDD
RLOAD
IOUT 1
RLOAD
IOUT 2
S(1)
S(2)
S(1)C
S(N)
S(2)C
S(N)C
...
Figure 45. Equivalent Analog Current Output
The DAC5688 can be easily configured to drive a doubly terminated 50Ω cable using a properly selected RF
transformer. Figure 46 and Figure 47 show the 50Ω doubly terminated transformer configuration with 1:1 and 4:1
impedance ratio, respectively. Note that the center tap of the primary input of the transformer has to be
connected to AVDD to enable a cd current flow. Applying a 20mA full-scale output current would lead to a 0.5
VPP for a 1:1 transformer and a 1 VPP output for a 4:1 transformer. The low dc-impedance between IOUT1 or
IOUT2 and the transformer center tap sets the center of the ac-signal at AVDD, so the 1 VPP output for the 4:1
transformer results in an output between AVDD + 0.5 V and AVDD – 0.5 V.
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AVDD
(3 .3 V )
50 W
1:1
IOUT 1
RLOAD
100 W
50 W
IOUT 2
50 W
AVDD (3.3 V)
Figure 46. Driving a Doubly Terminated 50Ω Cable Using a 1:1 Impedance Ratio Transformer
AVDD (3 .3 V )
100 W
4:1
IOUT 1
RLOAD
50 W
IOUT 2
100 W
AVDD (3.3 V)
Figure 47. Driving a Doubly Terminated 50Ω Cable Using a 4:1 Impedance Ratio Transformer
PASSIVE INTERFACE TO ANALOG QUADRATURE MODULATORS
A common application in communication systems is to interface the DAC to an IQ modulator like the TRF3703
family of modulators from Texas Instruments. The input of the modulator is generally of high impedance and
requires a specific common-mode voltage. A simple resistive network can be used to maintain 50Ω load
impedance for the DAC5688 and also provide the necessary common-mode voltages for both the DAC and the
modulator.
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Vin ~ Varies
Vout ~ 2.8 to 3.8 V
I1
Signal Conditioning
IOUTA1
IOUTA2
IOUTB1
IOUTB2
I2
S
Q1
RF
Q2
Quadrature modulator
Figure 48. DAC to Analog Quadrature Modulator Interface
The DAC5688 has a maximum 20mA full-scale output and a voltage compliance range of AVDD ± 0.5 V. The
TRF3703 IQ modulator family can be operated at three common-mode voltages: 1.5V, 1.7V, and 3.3V.
Figure 49 shows the recommended passive network to interface the DAC5688 to the TRF3703-17 which has a
common mode voltage of 1.7V. The network generates the 3.3V common mode required by the DAC output and
1.7V at the modulator input, while still maintaining 50Ω load for the DAC.
V1
R1
I
R2
DAC5688
I
R3
TRF3703-17
V2
R3
R2
/I
/I
R1
V1
Figure 49. DAC5688 to TRF3703-17 Interface
If V1 is set to 5V and V2 is set to -5V, the corresponding resistor values are R1 = 57Ω, R2 = 80Ω, and R3 =
336Ω. The loss developed through R2 is about -1.86 dB. In the case where there is no –5V supply available and
V2 is set to 0V, the resistor values are R1 = 66Ω, R2 = 101Ω, and R3 = 107Ω. The loss with these values is
–5.76dB.
Figure 50 shows the recommended network for interfacing with the TRF3703-33 which requires a common mode
of 3.3V. This is the simplest interface as there is no voltage shift. Because there is no voltage shift there is any
loss in the network. With V1 = 5V and V2 = 0V, the resistor values are R1 = 66Ω and R3 = 208Ω.
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DAC5688
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V1
R1
I
I
R3
DAC5688
TRF3703-33
V2
R3
/I
/I
R1
V1
Figure 50. DAC5688 to TRF3703-33 Interface
In most applications a baseband filter is required between the DAC and the modulator to eliminate the DAC
images. This filter can be placed after the common-mode biasing network. For the DAC to modulator network
shown in Figure 51, R2 and the filter load R4 need to be considered into the DAC impedance. The filter has to be
designed for the source impedance created by the resistor combination of R3 // (R2+R1). The effective
impedance seen by the DAC is affected by the filter termination resistor resulting in R1 // (R2+R3 // (R4/2)).
V1
R1
R2
I
R3
V2
DAC5688
Filter
R4
TRF3703
R3
R2
/I
R1
V1
Figure 51. DAC5688 to Modulator Interface with Filter
Factoring in R4 into the DAC load, a typical interface to the TRF3703-17 with V1 = 5V and V2 = 0V results in the
following values: R1 = 72Ω, R2 = 116Ω, R3 = 124Ω and R4 = 150Ω. This implies that the filter needs to be
designed for 75Ω input and output impedance (single-ended impedance). The common mode levels for the DAC
and modulator are maintained at 3.3V and 1.7V and the DAC load is 50Ω. The added load of the filter
termination causes the signal to be attenuated by –10.8 dB.
A filter can be implemented in a similar manner to interface with the TRF3703-33. In this case it is much simpler
to balance the loads and common mode voltages due to the absence of R2. An added benefit is that there is no
loss in this network. With V1 = 5V and V2 = 0V the network can be designed such that R1 = 115Ω, R3 = 681Ω,
and R4 = 200Ω. This results in a filter impedance of R1 // R2=100Ω, and a DAC load of R1 // R3 // (R4/2) which
is equal to 50Ω. R4 is a differential resistor and does not affect the common mode level created by R1 and R3.
The common-mode voltage is set at 3.3 V for a full-scale current of 20mA.
For more information on how to interface the DAC5688 to an analog quadrature modulator please refer to the
application reports Passive Terminations for Current Output DACs (SLAA399) and Design of Differential Filters
for High-Speed Signal Chains (SLWA053).
44
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DAC5688
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SLLS880C – DECEMBER 2007 – REVISED AUGUST 2010
RECOMMENDED STARTUP SEQUENCE
The following startup sequence is recommend to initialization the DAC5688:
1. Supply all 1.8V (CLKVDD, DVDD, VFUSE) and 3.3V (AVDD and IOVDD) voltages.
2. Toggle RESETB pin for a minimum 25 nSec active low pulse width.
3. Provide a stable CLK2/C input clock.
4. Program all desired SIF registers.
5. Provide a sync signal to all digital blocks. The sync input source may be either TXENABLE pin, SYNC pin or
a software sync via CONFIG5 sif_sync_sig bit; however, only the TXENABLE or SYNC pins are
recommended for multi-DAC synchronization. Refer to CONFIG5, CONFIG22 and CONFIG23 registers for
sync source selection. Note: Registers CONFIG6 through CONFIG13 all require a sync input to transfer the
contents of the control register inputs to the active digital blocks.
6. Provide data flow.
MULTI-DAC SYNCHRONIZATION
If the system has two or more DACs requiring synchronization, the sync signal in Step 5 of the RECOMMENDED
STARTUP SEQUENCE must be provided to all the DACs simultaneously. The sync input source must be either
the TXENABLE pin or the SYNC pin (the software sync is not recommended).
In some applications such as beamforming it is required that the multiple DACs in the system have constant
latency thus resulting in phase aligned outputs. As a result of the clock domain transfer on the DAC5688 FIFO,
the outputs of all DACs can only be synchronized to within ±1 DAC clock cycle in the External and Dual Clock
modes. In order to guarantee exact phase alignment between all devices it is required to set up the device in
Dual Synchronous Clock mode.
DESIGNING THE PLL LOOP FILTER
To minimize phase noise given for a given fDAC and M/N, the values of PLL_gain and PLL_range are selected
so that GVCO is minimized and within the MIN and MAX frequency for a given setting.
The external loop filter components C1, C2, and R1 are set by the GVCO, M/N, the loop phase margin fd and the
loop bandwidth wd. Except for applications where abrupt clock frequency changes require a fast PLL lock time, it
is suggested that fd be set to at least 80 degrees for stable locking and suppression of the phase noise side
lobes. Phase margins of 60 degrees or less can be sensitive to board layout and decoupling details.
See Figure 52 for the recommend external loop filter topology. C1, C2, and R1 are calculated by the following
equations
t3 2
C1 + t1 1 * t2
R1 +
C2 + t1 * t2
t3
t3
t1(t3 * t2)
(5)
ǒ
Ǔ
where
t1 +
K dKvco
ǒtan f d ) sec f dǓ
w 2d
t2 +
1
ǒ
w d tan f d ) sec f dǓ
t3 +
tan f d ) sec f d
wd
(6)
charge pump current: Iqp = 1 mA
vco gain: KVCO = 2p × GVCO rad/V
PFD Frequency: wd ≤ 160 MHz
phase detector gain: Kd = Iqp ÷ (2 × p × M) A/rad
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DAC5688
SLLS880C – DECEMBER 2007 – REVISED AUGUST 2010
www.ti.com
An Excel spreadsheet is provided by Texas Instruments for automatically calculating the values for C1, R1 and
C2.
DAC5688 PLL
PLL
LPF
Pin 64
R1
C2
C1
External
Loop
Filter
Figure 52. Recommended External Loop Filter Topology
46
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DAC5688
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SLLS880C – DECEMBER 2007 – REVISED AUGUST 2010
REVISION HISTORY
NOTE: Page numbers of previous versions may differ from current version.
Changes from Revision A (March 2008) to Revision B
Page
•
Changed Dual-Channel to first of title ................................................................................................................................... 1
•
Changed sin(x)/x from upper case to lower case ................................................................................................................. 1
•
Added sentence to DESCRIPTION section.."The DAC5688....multiplying PLL." ................................................................. 1
•
Changed to join last column 2 bottom rows as one .............................................................................................................. 1
•
Deleted "and External" from description of pin 25 ................................................................................................................ 3
•
Added sentence to description of pin 1 ................................................................................................................................ 3
•
Added sentence to description of pin 10,39,50,63 ............................................................................................................... 3
•
Added text to description of TXENABLE, pin 6 .................................................................................................................... 3
•
Deleted part of condition - Measured differential....to AVDD ................................................................................................ 5
•
Changed min value from 1.71 to 1.7, max value from 2.15 to 1.9 ....................................................................................... 5
•
Deleted min value -0.2 and max value 0.2, and added typ value of +/-0.2 .......................................................................... 5
•
Deleted "PLL = off" from 7 rows of Digital Latency description ............................................................................................ 6
•
Changed test conditions "NCO off" to "NCO on"; last row of Digital Latency, 2 places ....................................................... 6
•
Changed test conditions "NCO off" to "NCO" on next to last row of Digital Latency; and "QMC off" to "QMC on" ............. 6
•
Deleted min value -40 and max value 40 and added +/-20 to typ value in IIH row ............................................................. 7
•
Deleted min value -40 and max value 40 and added +/-20 to typ value in IIL row .............................................................. 7
•
Deleted 0.22xIOVDD from max value and added 0.5 in row of VOL ................................................................................... 7
•
Added 2 notes to EC digital specifications table .................................................................................................................. 7
•
Changed sentence in Offset Error: under TEST METHODOLOGY ................................................................................... 14
•
Added new register map table under REGISTER DESCRIPTIONS section ...................................................................... 15
•
Changed text in Register STATUS0 ................................................................................................................................... 16
•
Changed Bit 0 of Register CONFIG2 from 0 to 1 ............................................................................................................... 17
•
Changed text of Register CONFIG3 description ................................................................................................................ 17
•
Changed text of Register CONFIG4 description ................................................................................................................ 17
•
Deleted "Reserved" explanation from Register CONFIG5 description. .............................................................................. 18
•
Changed phaseoffset(15:0) to phaseoffset(15:8) in Register CONFIG7 description ......................................................... 18
•
Changed "Phaseadd(31:0)" to "phaseadd(31:24)" in Register CONFIG11 description ..................................................... 19
•
Deleted explanatory "Note" from Register CONFG14 description ..................................................................................... 20
•
Added text to Register CONFIG20 description. .................................................................................................................. 21
•
Added text to Register CONFIG21 description. .................................................................................................................. 21
•
Changed "Default 0x00" to "Default 0x15" for Register CONFIG23 Address .................................................................... 22
•
Changed text in Register CONFIG23 description ............................................................................................................... 22
•
Changed Register CONFIG28 description from "Reserved(7:0)" to "cleared" ................................................................... 23
•
Deleted "cleared" in description for Register CONFG28 .................................................................................................... 23
•
Deleted explanatory NOTE from Register CONFIG30 description. .................................................................................... 24
•
Added sentence to 1st paragraph of section "SERIAL INTERFACE" description. ............................................................. 25
•
Changed graphic entity for Figure 22, Magnitude Spectrum for FIR1 ................................................................................ 28
•
Changed text in section "Full Complex Mixer (FMIX)." ....................................................................................................... 31
•
Changed QOUT equation in Full Complex Mixer (FMIX) section. ........................................................................................ 31
•
Changed description for section "Quadrature Modulator Correction (QMC)." .................................................................... 32
•
Changed description for section "DAC Offset Control." ...................................................................................................... 32
•
Changed text in section "DUAL SYNCHRONOUS CLOCK MODE." ................................................................................. 33
•
Added text in section "DUAL CLOCK MODE." ................................................................................................................... 33
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DAC5688
SLLS880C – DECEMBER 2007 – REVISED AUGUST 2010
www.ti.com
•
Changed Figure 36 , Figure 37, Figure 38, caption from "......(PLL Clock Mode)" to "......(Dual Clock Mode)" ................. 35
•
Changed graphic entity for Figure 40 ................................................................................................................................. 37
•
Added section "PASSIVE INTERFACE TO ANALOG QUADRATURE MODULATORS." ................................................. 42
•
Changed text in section "RECOMMENDED STARTUP SEQUENCE " ............................................................................. 45
Changes from Revision B (August 2010) to Revision C
•
48
Page
Changed cos in equation 2 to sin ....................................................................................................................................... 31
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PACKAGE OPTION ADDENDUM
www.ti.com
18-Feb-2014
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
DAC5688IRGC25
ACTIVE
VQFN
RGC
64
25
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 85
DAC5688I
DAC5688IRGCR
ACTIVE
VQFN
RGC
64
2000
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 85
DAC5688I
DAC5688IRGCRG4
ACTIVE
VQFN
RGC
64
2000
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 85
DAC5688I
DAC5688IRGCT
ACTIVE
VQFN
RGC
64
250
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 85
DAC5688I
DAC5688IRGCTG4
ACTIVE
VQFN
RGC
64
250
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 85
DAC5688I
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
18-Feb-2014
(6)
Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
16-Feb-2012
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
DAC5688IRGCR
VQFN
RGC
64
2000
330.0
16.4
9.3
9.3
1.5
12.0
16.0
Q2
DAC5688IRGCT
VQFN
RGC
64
250
330.0
16.4
9.3
9.3
1.5
12.0
16.0
Q2
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
16-Feb-2012
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
DAC5688IRGCR
VQFN
RGC
64
2000
336.6
336.6
28.6
DAC5688IRGCT
VQFN
RGC
64
250
336.6
336.6
28.6
Pack Materials-Page 2
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