AN880 APPLICATION NOTE ® THE L6569: A NEW HIGH VOLTAGE IC DRIVER FOR ELECTRONIC LAMP BALLAST by G. Calabrese and T. Castagnet INTRODUCTION Electronic lamp ballasts are now popular in both consumer and industrial lighting. They offer power saving, flicker free operation and reduced sizes. Improvements to the light control and cost reduction of the ballast will broaden their market acceptance. Today designers focus on reducing the cost of the ballast, but also work to add features to the ballast like saving energy by dimming the light, or increasing the life time with better preheat and protections. Such requirements have contributed to the development of dedicated high voltage controllers like the L6569, which are able to drive the floating transistor of a symmetric half bridge inverter. This device is a simple, monolithic oscillator-half bridge driver that allows quick design of the ballast. HIGH VOLTAGE IC DRIVERS IN BALLAST APPLICATIONS The voltage fed half bridge Voltage fed series resonant half bridge inverters are currently used for Compact Fluorescent Lamp ballasts (CFL), for Halogen Lamp transformers, and for many European Tube Lamp (TL) ballasts. This simple converter is preferred for new designs, because it minimizes the off state voltage of the power transistors to the peak line voltage, and requires only one resonant choke. In addition this choke protects the half bridge against short circuits across lamp terminals. However overheating and overcurrent occur during open load operation. The inverter robustness must be improved, or some protections are required. The half bridge inverter operates in Zero Voltage Switching (ZVS) resonant mode [1], to reduce the transistor switching losses and the electromagnetic interference generated by the output wiring and the lamp. Fully integrated ballast controllers By varying the switching frequency, the half bridge inverter is able to modulate the lamp power. However most current designs use a sinFebruary 2003 Figure 1: CFL series resonant half bridge inverter. Figure 2: Current and voltage of the STD3NA50 MOSFETs when driven in ZVS with the L6569. ID VDS GND LVG GND RF GND 2 µs/dv ; 50 V/dv ; 0.1 A/dv gle frequency with a saturable pulse transformer (see fig. 1) to drive the transistors. This type of design has a higher component count, a higher tolerance on the switching frequency, and it cannot adjust the lamp power. The only way to design a cost effective, compact and smart control of the lamp is to use a dedicated I.C. that is able to drive the upper transistor of an symmetric half bridge inverter. Such controllers require a high voltage capability for the floating transistor driver [2]. MOSFETs are preferred over Bipolar transistors as power switches because their gate driver requires a lower supply current and a smaller silicon size [3]. 1/14 AN880 APPLICATION NOTE THE L6569 AND ITS APPLICATIONS The L6569 The L6569 is able to directly control a symmetric half bridge inverter of a fluorescent lamp ballast, or a low voltage halogen lamp transformer.Two 270mA buffers drive the inverter MOSFETs in complementary fashion with a 1.25µs built-in dead time to prevent cross conduction. The buffer for the upper Mosfet is driven through a 600V level shifter realized in BCD off line technology. The oscillator, similar to a CMOS 555 timer, operates from 25 to 150 kHz with a +/-5% maximum tolerance. The internal 15V shunt regulator has a 9V Under Voltage Lock Out with an 1V hysteresis, and the circuit requires only 150 µA at power up. The L6569 integrates a high voltage Lateral DMOS transistor in place of the usual external diode [2] to charge the bootstrap capacitor for the upper buffer. Figure 5 shows DMOS operating as a synchronous rectifier. The applications The primary application for the L6569 is the Compact Fluorescent Lamp. With the oscillator, the supply and the Mosfet drivers it is the core of the application, and designers can customize the circuit to their requirements. Figure 3: Block diagram of the L6569. VS BOOT CHARGE PUMP UVLO LEVEL SHIFTER RF HIGH SIDE DRIVER HVG OUT CF LOGIC CONTROL with DEAD TIME LVG LOW SIDE DRIVER GND Figure 4: Basic application diagram using the L6569 and two STD4NK50Z MOSFETs. 100nF 180KΩ 22Ω 10µF 10KΩ AC LINE STD4NK50Z L6569 LAMP 10µF 1nF 22Ω D02IN1385 2/14 AN880 APPLICATION NOTE Figure 5: Bootstrap capacitor charge. ON 15.6 V 600V 120Ω CHARGE PUMP CIRCUIT ON LOGIC L6569 Figure 6: Basic diagram for 2x105 W lamp ballast in full bridge configuration. HV VS EXTERNAL OSCILLATOR BOOT VS RF CF GND HVG L6569 100nF 100nF 47 47 OUT LVG BOOT HVG OUT 47 47 VS LVG RF L6569 CF GND STB9NK50Z D02IN1386 Typical industrial TL ballasts requires complex control with dimming or automation interface. Here the L6569 is a driver between the power and control blocks. To use it with an external oscillator, pin CF is used as an 0-12V logic input, and the L6569 becomes a high voltage buffer. Applications with power above 150W require a full bridge inverter. Figure 6 shows how two L6569 drive such a MOSFET bridge. If no external control is required, the first L6569 master can control the switching with its oscillator, and synchronizes the other driver as (slave). The L6569 start up Two versions of the L6569 are available with different start up characteristics. The L6569 drives the lower MOSFET ON at power-up until the supply voltage reaches the Under Voltage Lock Out. The bootstrap capacitor is precharged to 4.6V and both the lower and the upper MOSFETs will switch immediately with the oscillator. This is intended for inverters which use only one DC blocking capacitor connected to the power ground, as shown on figure 4 for CFL ballast. 3/14 AN880 APPLICATION NOTE The L6569A holds both MOSFETs OFF until the Under Voltage Lock Out is reached. This is intended for inverters using 2 decoupling capacitors in half bridge as shown on figure 12. The inverter is totally off, so that the voltage at the capacitors center node is not unbalanced by the leakage path during power on. CONSIDERATIONS ON THE L6569 ENVIRONMENT To illustrate the benefits of the L6569 in the CFL applications, a demonstration board was developed to supply Sylvania 18W DULUX lamp (ref: CF18DT/E). The following chapters summarize the application considerations applied in this design. The schematic, lay out and components list are shown in appendix A. Symmetric half bridge operation To supply a fluorescent lamp, the ballast has to achieve 3 functions: pre heat, ignition, and normal lamp operation. The serial resonance occurs between the choke and the capacitor in parallel with the lamp. The choice of these components determines the lamp ignition voltage and the nominal lamp current. Since the inverter using the L6569 and MOSFETs can operate at a higher frequency than conventional solutions, the size of the passive components will be reduced. Such inverter can operate up to 150 kHz in ZVS mode, and the switching losses of the power transistors only limits the frequency. In new design this frequency should be set between 50 and 100 kHz. For instance with an 18W lamp, a frequency increase from 33 to 50 kHz will lead to a 40% reduction of the choke size. To operate in Zero Voltage Switching (ZVS), the switching frequency is higher than the resonant frequency. All operation phases of the ballast are secure in this mode. When the bootstrap transistor is conducting, no pulse current will flow from pin BOOT to pin VS, as it might happen in Zero Current Switching. The bootstrap transistor remains in its Safe Operating Area, and its dissipation is negligible. The MOSFET drive The ZVS drive technique requires only a fast turn off capability as shown on figure 2, and the transistor buffers are designed with a stronger sink current. The two MOSFET buffers of the L6569 can sink a 400 mA peak current on capacitive load. Typically these buffers can drive any MOSFETs in TO220 package. Figure 7 shows an example with the STP8NA50 that has an 0.85 Ω resistance RDS-ON. 4/14 Figure 7: Current and voltage of the STP8NA50 MOSFET at turn off with the L6569. TGD = 245 ns ,Tc = 95 ns, E = 93 µJ @ Tj = 50°C, RG = 22 Ω. TGD Tc ID VGS GND GND VD GND 50 ns/dv ; 1 A/dv ; 5 V/dv ; 50V/dv The built-in dead time circuit acts when a MOSFET turns off, delaying the turn on of the opposite transistor for 1.25 µs. The voltage VOUT between the 2 MOSFETs must switch within the minimum dead time (0.85 µs), as shown on figure 8, to avoid bridge cross conductions and transistors overheat. Figure 8: STD3NA50 MOSFET turn off when driven by the L6569. TC + TGD < TD TD VDS ID TC LVG TGD RF GND GND GND 200 ns/dv ; 50 V/dv ; 0.1 A/dv The MOSFET voltage selection Since the ballast is connected to the ac mains, it must handle any spurious voltage spikes. When the front end RFI filter and the clamping device, such as a varistor, absorbes totally the spike energy, MOSFETs can have the same 600V minimum breakdown voltage BVDSS as the L6569. Otherwise when the upper MOSFET is on, the residual default may be applied to the L6569. Although the pin OUT breakdown voltage is higher than 600V, it has a poor avalanche robustness. Therefore the lower MOSFET protects the driver by having a lower BVDSS. A MOSFET with a minimum BVDSS up to 500V will achieve safely this task. AN880 APPLICATION NOTE Figure 9. L6569 driver protection against voltage spikes. BV OUT > 600V H.V.+ ON 15V V OUT I L6569 OFF mA, a secondary winding on the resonant choke is an easy supply alternative. The auxiliary supply of the converter The circuit consumption is defined by the MOSFETs gate charge, the I.C. consumption, the oscillator, and the shunt regulator. Several circuits are possible. In many applications a snubber is used to reduce the dissipation in the MOSFETs. When this snubber is used in conjunction with a start up resistor (RS in Figure 10), a non dissipative supply is achieved almost for free. At start up the I.C. is consuming 150 µA, and therefore only a small supply resistor is required. During operation the capacitor provides the supply current. To avoid cross conduction, the capacitance is limited by the driver dead time TD . Hence the capacitive supply current IC is also limited.For a CFL ballast this circuit easily supplies the required operating current. Using a CF18DT lamp ( IL > 230 mA) the required capacitance is 470 pF on 230 Vac line. At 50 kHz the average capacitive current is 6 mA, as described in appendix B. When the required driver current is higher than 10 The ballast shutdown The L6569 allows several ways (see figg. 11, 12 and 13) to shutdown the ballast [4]: by acting on the CF input oscillator pin to turn off the upper MOSFET or by acting on the VS supply pin with the Under Voltage Lock Out. Acting on CF (Fig. 11) a limiting resistor RL has to be used, and it has to be: RL ⋅ CF > 1µs. When the shutdown is realized acting on Vs pin, (see fig. 12) a limiting resistor Rs must be used to slow down the discharge of the supply filter Cs. The constant time of the discharge must be greater than 10 periods of the switching frequency: 10 RS ≥ Cs ⋅ fsw Connecting the CF pin to ground GND stops the oscillator, and the lower MOSFET will remain ON. Therefore the bootstrap capacitor remains Figure 10: Non dissipative auxiliary supply using the transistor snubber. 1mA WHEN STARTING 220kΩ Rs 6 mA WHEN 50 kHz SWITCHING C 470 pF 310 V Cs bootstrap circuit L6569 5/14 AN880 APPLICATION NOTE charged and the circuit can restart immediately. higher than the SCR holding current (see figure This method is suitable when the inverter uses 12), the SCR will remain on and the two MOSonly one DC blocking capacitor connected to the FETs off. Removing power or commutating the power ground, as used on figure 11 for Compact SCR allows a new start up [4]. Fluorescent Lamp. Pulling the VS voltage below Otherwise a disable circuitry that turns off the two the UVLO turns off the oscillator and gives the MOSFETs (see figure13), can achieve the shutsame bridge configuration. down function. Compared to the SCR solution, For the L6569A, discharging the VS supply below the shutdown is immediate and the inverter can the UVLO turns off both MOSFETs. An SCR like restart on the disable order. the X0202MA may be used for the reset function. If the current flowing through the supply resistor is Figure 11: L6569 shutdown through the CF oscillator pin. L 6569 RF RL CF ON Figure 12: Shutdown with a thyristor & a serial resistor to slow down the supply voltage decay. L 6569A C R Rs OFF R OFF shutdown Figure 13: L6569 disable circuitry with both MOSFETs off. H.V. 100nF Vs VS BOOT RF HVG CF OUT GND LVG 5.6k 200 200 22 L6569 V IN 4.7 k W HCF4011 BC327 DISABLE 6/14 AN880 APPLICATION NOTE THE LAMP SEEN BY THE ELECTRONIC DESIGNER The lamp equivalent impedance The compact fluorescent lamps are specified at 25 kHz (IEC 929). The MOSFETs and the L6569 allow to increase the switching frequency, but the sensitivity of the lamp to the frequency needs to be analyzed. A few samples of the CF18DT/E lamp were tested by varying the frequency and the current of the lamp. The figure 14 shows the lamp impedance versus its current as it varies from 0.1A to 0.23A with 5 frequencies from 25 to 150 kHz (TAMB = 25°C). Figure 14: Variation of the lamp impedance versus its current for several switching frequencies. R lamp (Ohms) 1200 25 kHz 1000 50 kHz 800 100 kHz 600 The preheat Preheat techniques are used in CFL ballasts to reduce the ignition lamp voltage. During this phase the lamp is characterized by a high impedance that forces the electrical conduction through the preheat filaments. These filaments initially have a low resistance that will increase by 5 times during the preheat. The preheat typically lasts from 400 ms to 1 s, and is achieved by controlling either the current or the voltage of the filaments. For a current control the filaments are in series with the resonant network as shown on figure 16a. When the inverter frequency is constant, a positive temperature coefficient thermistor (PTC) in parallel with the lamp achieves the task by adjusting both the filament current and the preheat duration. The board uses a 150Ω PTC with two 8.2 nF capacitors. The preheat lasts 0.8s and the filament current is 0.45 Arms. The PTC is a cheap device, but it is dissipative and works only once at power-up. Figure 16: Basic preheat current control diagram (a); preheat filament energy curve (b) 150 kHz 400 200 0 0.05 0.1 0.15 0.2 I lamp (A) 0.25 0.3 From the tests the impedance appears insensitive to the frequency for such lamps. The specified impedance might be valid for higher frequency operation. The relative lamp light output was measured as proposed in reference [5]. The light flux increases slightly in that frequency range, but can be considered constant. Obviously the impedance is sensitive to the current with a negative coefficient, and the ballast operates with a non linear impedance [6]. When current is half the nominal one, the impedance is 2.6 times higher, and the voltage is only 25% higher (see figure 15). Figure 15: Variation of the average impedance and voltage of the lamp R (Ohms) 1200 U (V) 200 Rlamp 900 150 600 100 300 50 0 0.05 0.1 0.15 I (A) 0.2 0 0.25 I CTL (A) LAMP E If E=R.I² (B) I CTL t Vlamp The preheat can be achieved with a filament voltage control. The filaments are supplied by two auxiliary windings of the resonant choke as shown figure 17a. During the preheat the L6569 frequency is increased, and the choke operates 7/14 AN880 APPLICATION NOTE as a transformer supplying the voltage to the filaments. Only few components are added around the L6569 (see figure 18), and the control of the preheat energy is less sensitive to the preheat duration and the inverter frequency (see figure 17b). The start up initialization The initial conditions of the power switching start up requires care; especially if the resonant and switching frequencies are close to each other. The resonant network is not loaded and the full Figure 17: Basic preheat voltage control diagram (a); preheat filament energy curve (b) (A) VCTL LAMP E Vf VCTL (B) E=V²/R t Figure 18: Double frequency control for the L6569 with programmed frequency and duration. 1_Vs 2_RF RF 3_CF R CF L6569 C CF_ST 8/14 AN880 APPLICATION NOTE line voltage VDC is applied when the oscillator starts. The ballast has to start directly with its nominal conditions to remove any transient oscillation. Hence the operation runs in ZVS mode with no spurious lamp ignition. This situation does not occur with the saturable transformer drive, because the saturation limits naturally the current by increasing the frequency. In the example the resonant capacitors are preset to be compatible with the choke current rise (see figure 19). The blocking capacitor is precharged to approximately half VDC by 2 biasing resistors, and the lower Mosfet also discharges the resonant capacitor to ground (see figure 20). Therefore the blocking capacitor never goes above 2/3 of the line voltage VDC (250V rating), the operation is safe in ZVS mode. The L6569 is here preferred to the L6569A, because the lower The lamp removal protection Used in TL ballast, the lamp removal protection is frequently also requested in the "plug-in" CFL ballast . Depending of the topology and the preheat mode, the lamp removal behaves as: - a noload resonant mode when the choke and the capacitor are still connected to the inverter ; a required overcurrent protection increases the frequency to reduce the current; - an open circuit mode when the lamp filaments are inserted in the resonant circuit. When the circuit is open, the choke is not supplied. The MOSFETs turn off slowly generating bridge cross conduction, and undesirable dissipation losses (see figure 21). The detection stops the switching to eliminate the cross conduction. Figure 19: Waveforms of the choke current and the capacitor voltages in steady state preheat. Figure 21: Drain current and voltage STP8NA50 MOSFET operating with noload. ID = 2 A peak IDEAL INITIAL TIME VD II VGS GND ID GND GND GND VB VBI GND 5 µs/dv ; 50 V/dv ; 0.5 A/dv 100 ns/dv ; 50 V/dv ; 5V/dv ; 1 A/dv Mosfet is on at power-up. Figure 20: Configuration of the resonant network during the initialization of the driver. VS<UVLO II 2.4 mH L6569 ON VBI VB 2 x 180 kΩ 4nF 100 nF 9/14 AN880 APPLICATION NOTE Figure 22: Open load detection example. L6569 LAMP 3_CF 4_GND 10.R 18V 100.R 11.R Several ways can achieve the protection task. First it can be done by sensing the resonant current through a MOSFET source resistor or a secondary winding on the choke. The switching is stopped when a large current reduction is detected by analog means. A logic circuit can also detect the presence of the lamp filaments. One end of a filament is always connected to a fixed voltage. If the other end of the filament is connected through a high impedance resistor to another voltage, the absence of the filament can be easily detected by monitoring the resistor voltage change as shown on figure 22. CONCLUSION The foregoing note shows how high voltage drivers, like the L6569, simplify the design of the lamp ballast. These devices includes all the circuitry to drive MOSFETs in half bridge inverter. Since the optimized switching frequency increases above 50 kHz with a low tolerance, the size of the passive resonant components is re- 10/14 10.R duced, and the ballast becomes cheaper.With its supply and its oscillator the L6569 is versatile, and its flexibility permits to design any improved power control. BIBLIOGRAPHY [1]: AN 527 "Electronic fluorescent lamp ballast" A.Vitanza, R.Scollo, SGS-THOMSON [2]: Smart Power ICs, Chapter 8, High voltage integrated circuits for off-line power applications. C.Diazzi, SGS-THOMSON [3]: AN 512 "Characteristics of power semiconductors" JM Peter, SGS-THOMSON [4] : "The L6569 half bridge driver: the shutdown function" [5] : "Compatibility test of dimming electronic ballasts used in daylighting and environment controls", A.Buddenberg, Rensselaer Polytechnic Institute [6]: "PSpice High Frequency Dynamic Fluorescent Lamp Model", Bryce Hesterman, APEC’96, p.641 AN880 APPLICATION NOTE APPENDIX A: CFL DEMONSTRATION BOARD WITH THE L6569 A demonstration board was developed as an example for Compact Fluorescent Lamp ballast. It is optimised for a CF18DT/E/830 18W lamp from Osram-Sylvania. Using the L6569 the circuit achieves preheat, ignition and normal lamp operation. The power transistors are two STD3NA50 500V-3Ω MOSFETs in I-PACK package. The resonant ballast The value of the choke (L1) and the two capacitors (C7 & C8) in parallel with the lamp determine the lamp ignition voltage and the nominal lamp current. During the ignition the lamp impedance is essentially infinite, and the filaments resistance is only the serial load. To generate the ignition voltage, the switching frequency is set close to the resonance frequency. In normal operation the choke resonates with the capacitors C7 & C8 (parallel loading), but also with the decoupling capacitor C6 (serial loading). The current mode preheat uses a 150Ω Positive Temperature Coefficient thermistor. Inserted in the capacitive series (C7 & C8), the PTC produces a 0.45 Arms filament current during the initial 0.8 s (reference: 307C1253BHEAB from CERA-MITE). Board description The three sections of the board are an AC input rectifier, the half bridge inverter, and the resonant ballast. By changing the connection on the input mains, the ballast can operate either on 120 Vac mains with a voltage doubler rectification, or on 230 Vac mains with a full wave rectification. The input resistor R1 limits the initial inrush current charging the bulk capacitors. The L6569 operates with a single 50 kHz switching frequency programmed by R4 and C1. Two fast diodes D2 & D3 synchronize the oscillator to keep the switching in ZVS mode. The control circuit requires 4.5 mA to supply the I.C., the MOS gate drives, and the oscillator. Its supply delivers at least 6.5 mA as described in appendix B. The start up resistor also balances the voltage across the two bulk capacitors. Figure 23. CFL ballast diagram for a 18W CFD18T/E lamp with 120/230 Vac inputs. Basic ballast electrical characteristics Input voltage: 120 or 230 Vac by input connections change Switching frequency: 50kHz Average dc line voltage range: Vdc from 260 to 355 Nominal supply current: 0.17A rms @ 310 Vdc Nominal output power: 17W Minimum ignition voltage: 700V peak @ 260 Vdc Nominal preheat current: 0.45 Arms during 0.8s @ 310 Vdc Figure 23. R8 120K 1/2W C2 47µF 250V R4 27K 1/4W 4 x 1N4006 D7 D4 D6 D5 R5 100K 220V N R1 15 1W 110V 1/2W C5 47µF 250V C3 4.7µF 25V D8 1N4148 VS BOOT RF HVG L6569 CF C1 560pF 50V OUT LVG R9 180K 1/4W R10 10K R2 22 1/4W R3 22 1/4W GND C9 470pF 630V R6 47 1/4W Q1 STD4NK50 C4100nF 50V 1/4W L1=2.4mH Q2 STD4NK50 R7 180K 1/4W (*) BYW100-100 D1 ZPD 18V C7 8.2nF 630V (*) C6 100nF 250V C8 8.2nF 630V (*) RV1 PTC 150 350V D2 L1=2.4mH core TH LCC E2006-B4 Ref also VOGH 575 0409200 2.4mH C7-C8=PS8n2J H3 630-2A TH D96IN419C D3 BYW100-100 CFL LAMP SYLVANIA DELUX T/E 18W (*) Polypropylene, Capacitors 630V rated VL 11/14 AN880 APPLICATION NOTE Figure 24: PCB Layout of the board. Comp. Side Copper Side Figure 25: PCB component placement diagram. The resonant choke The inductance of the choke is 2.4 mH with a minimum saturation current of 0.65 A. In the practical example it has been done with: Core: Thomson LCC E2006 material B4; Air gap: 2 spacers of 0.4 mm each (total 0.8 mm); AL = 75 nH; Bobbin: HC2006BA-06; Number of turns: 175; Measured saturation current: 1 A peak @ 25 °C; 12/14 Customization of the board Some flexibility is added to the board to extend its evaluation. The MOSFETs have two foot prints to mount either I-PAK or TO220 packages. And two choke footprints are also avalaible for E1905A and E2006A magnetic cores. AN880 APPLICATION NOTE APPENDIX B: Rating of the capacitive supply with the L6569 driver The supply is made with the snubber and a start up resistor RS. A snubber circuit is used to minimize the MOSFETs dissipation. It also achieves a non dissipative supply as shown on figure 10. The MOSFETs gate charge, the driver consumption, the oscillator, and the shunt regulator, define the circuit consumption. We can estimate this current is IS AV: ISAV > 2 ⋅ IG + IQS + IOSC + IREG = = 2 ⋅ QG ⋅ fsw + IQS + VS + IREG RF ⋅ 2 Where QG the MOSFET gate charge IQS the driver supply current VS the supply voltage RF the oscillator resistor and VS the driver supply voltage IREG the shunt regulator current. When VS is lower than the UVLO threshold UUVLO, the driver is only consuming. Its current must be minimal to reduce the dissipation of the resistor RS. The L6569 has a 150 µA start up current, and the maximum resistance is 2MΩ for a 230Vac line application. We can also reduce the resistor value to get a faster start up time TS. plies the lamp current during the lower MOS turn off. To avoid any cross conduction its capacitance is limited by the driver dead time TD (see figure 26). Hence the capacitive supply current IC is also limited. TD ⋅ IL C< VDC ICAV = C ⋅ VDC ⋅ FSW < IL ⋅ TD ⋅ FSW Where IL is the peak lamp current, and FSW the switching frequency. For a ballast such as a CFL one this circuit supplies easily the required current. For instance with a CF18DT lamp ( IL > 230 mA) the capacitor is 1nF on 120Vac line, 470 pF on 230 Vac line. At 50 kHz the average capacitive current is 6 mA in both cases. Figure 26: Cross conduction of the snubber capacitor with the upper MOSFET: capacitor current and voltage waveforms. TD VHVG + VOUT IC GND GND RS ⋅ CS ⋅ UUVLO TS = VDC RF GND 200 ns/dv ; 50 V/dv ; 0.1 A/dv Where CS is the supply capacitor, and VDC the line voltage. When the timer oscillates, the capacitor C sup- 13/14 AN880 APPLICATION NOTE Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of STMicroelectronics. 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