AD AD823AR Dual, 16 mhz, rail-to-rail fet input amplifier Datasheet

Dual, 16 MHz, Rail-to-Rail
FET Input Amplifier
AD823
CONNECTION DIAGRAM
OUT1 1
8
+VS
–IN1 2
7
OUT2
+IN1 3
6
–IN2
–VS 4
5
+IN2
AD823
Figure 1. 8-Lead PDIP and SOIC
RL = 100kΩ
CL = 50pF
+VS = +3V
G = +1
3V
GND
500mV
APPLICATIONS
00901-002
Single-supply operation
Output swings rail-to-rail
Input voltage range extends below ground
Single-supply capability from 3 V to 36 V
High load drive
Capacitive load drive of 500 pF, G = +1
Output current of 15 mA, 0.5 V from supplies
Excellent ac performance on 2.6 mA/amplifier
−3 dB bandwidth of 16 MHz, G = +1
350 ns settling time to 0.01% (2 V step)
Slew rate of 22 V/μs
Good dc performance
800 μV maximum input offset voltage
2 μV/°C offset voltage drift
25 pA maximum input bias current
Low distortion: −108 dBc worst harmonic @ 20 kHz
Low noise: 16 nV/√Hz @ 10 kHz
No phase inversion with inputs to the supply rails
00901-001
FEATURES
200µs
Figure 2. Output Swing, +VS = +3 V, G = +1
Battery-powered precision instrumentation
Photodiode preamps
Active filters
12-bit to 16-bit data acquisition systems
Medical instrumentation
2
1
+VS = +5V
G = +1
0
The AD823 is a dual precision, 16 MHz, JFET input op amp
that can operate from a single supply of 3.0 V to 36 V or from
dual supplies of ±1.5 V to ±18 V. It has true single-supply
capability with an input voltage range extending below ground
in single-supply mode. Output voltage swing extends to within
50 mV of each rail for IOUT ≤ 100 μA, providing outstanding
output dynamic range.
An offset voltage of 800 μV maximum, an offset voltage drift of
2 μV/°C, input bias currents below 25 pA, and low input voltage
noise provide dc precision with source impedances up to a
Gigaohm. It provides 16 MHz, −3 dB bandwidth, −108 dB THD
@ 20 kHz, and a 22 V/μs slew rate with a low supply current of
2.6 mA per amplifier. The AD823 drives up to 500 pF of direct
capacitive load as a follower and provides an output current of
15 mA, 0.5 V from the supply rails. This allows the amplifier to
handle a wide range of load conditions.
–2
–3
–4
–5
–6
–7
–8
1k
10k
100k
1M
FREQUENCY (Hz)
10M
00901-003
GENERAL DESCRIPTION
OUTPUT (dB)
–1
Figure 3. Small Signal Bandwidth, G = +1
This combination of ac and dc performance, plus the outstanding
load drive capability, results in an exceptionally versatile amplifier for applications such as A/D drivers, high speed active
filters, and other low voltage, high dynamic range systems.
The AD823 is available over the industrial temperature range of
−40°C to +85°C and is offered in both 8-lead PDIP and 8-lead
SOIC packages.
Rev. B
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113 ©1995–2007 Analog Devices, Inc. All rights reserved.
AD823
TABLE OF CONTENTS
Features .............................................................................................. 1
Typical Performance Characteristics ..............................................7
Applications....................................................................................... 1
Theory of Operation ...................................................................... 13
General Description ......................................................................... 1
Output Impedance ..................................................................... 14
Connection Diagram ....................................................................... 1
Application Notes ........................................................................... 15
Revision History ............................................................................... 2
Input Characteristics.................................................................. 15
Specifications..................................................................................... 3
Output Characteristics............................................................... 15
Absolute Maximum Ratings............................................................ 6
Outline Dimensions ....................................................................... 18
Thermal Resistance ...................................................................... 6
Ordering Guide .......................................................................... 19
ESD Caution.................................................................................. 6
REVISION HISTORY
2/07—Rev. A to Rev. B
Updated Format..................................................................Universal
Changes to DC Performance .......................................................... 5
Updated Outline Dimensions ....................................................... 18
Changes to Ordering Guide ......................................................... 19
5/04—Rev. 0 to Rev. A
Changes to Specifications ................................................................ 2
Changes to Ordering Guide ......................................................... 17
Updated Outline Dimensions ....................................................... 17
5/95—Revision 0: Initial Version
Rev. B | Page 2 of 20
AD823
SPECIFICATIONS
At TA = 25°C, +VS = +5 V, RL = 2 kΩ to 2.5 V, unless otherwise noted.
Table 1.
Parameter
DYNAMIC PERFORMANCE
−3 dB Bandwidth, VO ≤ 0.2 V p-p
Full Power Response
Slew Rate
Settling Time
to 0.1%
to 0.01%
NOISE/DISTORTION PERFORMANCE
Input Voltage Noise
Input Current Noise
Harmonic Distortion
Crosstalk
f = 1 kHz
f = 1 MHz
DC PERFORMANCE
Initial Offset
Maximum Offset Over temperature
Offset Drift
Input Bias Current
at TMAX
Input Offset Current
at TMAX
Open-Loop Gain
TMIN to TMAX
INPUT CHARACTERISTICS
Input Common-Mode Voltage Range
Input Resistance
Input Capacitance
Common-Mode Rejection Ratio
OUTPUT CHARACTERISTICS
Output Voltage Swing
IL = ±100 μA
IL = ±2 mA
IL = ±10 mA
Output Current
Short-Circuit Current
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current
Power Supply Rejection Ratio
Conditions
Min
Typ
G = +1
VO = 2 V p-p
G = −1, VO = 4 V Step
12
16
3.5
22
MHz
MHz
V/μs
G = −1, VO = 2 V Step
G = −1, VO = 2 V Step
320
350
ns
ns
f = 10 kHz
f = 1 kHz
RL = 600 Ω to 2.5 V, VO = 2 V p-p, f = 20 kHz
16
1
−108
nV/√Hz
fA/√Hz
dBc
RL = 5 kΩ
RL = 5 kΩ
−105
−63
dB
dB
VCM = 0 V to 4 V
VCM = 0 V to 4 V
0.2
0.3
2
3
0.5
2
0.5
45
VO = 0.2 V to 4 V, RL = 2 kΩ
14
20
20
−0.2 to +3
VCM = 0 V to 3 V
60
VOUT = 0.5 V to 4.5 V
Sourcing to 2.5 V
Sinking to 2.5 V
G = +1
Rev. B | Page 3 of 20
70
0.8
2.0
25
5
20
Unit
mV
mV
μV/°C
pA
nA
pA
nA
V/mV
V/mV
−0.2 to +3.8
1013
1.8
76
V
Ω
pF
dB
0.025 to 4.975
0.08 to 4.92
0.25 to 4.75
16
40
30
500
V
V
V
mA
mA
mA
pF
3
TMIN to TMAX, total
VS = 5 V to 15 V, TMIN to TMAX
Max
5.2
80
36
5.6
V
mA
dB
AD823
At TA = 25°C, +VS = +3.3 V, RL = 2 kΩ to 1.65 V, unless otherwise noted.
Table 2.
Parameter
DYNAMIC PERFORMANCE
−3 dB Bandwidth, VO ≤ 0.2 V p-p
Full Power Response
Slew Rate
Settling Time
to 0.1%
to 0.01%
NOISE/DISTORTION PERFORMANCE
Input Voltage Noise
Input Current Noise
Harmonic Distortion
Crosstalk
f = 1 kHz
f = 1 MHz
DC PERFORMANCE
Initial Offset
Maximum Offset Over temperature
Offset Drift
Input Bias Current
at TMAX
Input Offset Current
at TMAX
Open-Loop Gain
TMIN to TMAX
INPUT CHARACTERISTICS
Input Common-Mode Voltage Range
Input Resistance
Input Capacitance
Common-Mode Rejection Ratio
OUTPUT CHARACTERISTICS
Output Voltage Swing
IL = ±100 μA
IL = ±2 mA
IL = ±10 mA
Output Current
Short-Circuit Current
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current
Power Supply Rejection Ratio
Conditions
Min
Typ
G = +1
VO = 2 V p-p
G = −1, VO = 2 V Step
12
15
3.2
20
MHz
MHz
V/μs
G = −1, VO = 2 V Step
G = −1, VO = 2 V Step
250
300
ns
ns
f = 10 kHz
f = 1 kHz
RL = 100 Ω, VO = 2 V p-p, f = 20 kHz
16
1
−93
nV/√Hz
fA/√Hz
dBc
RL = 5 kΩ
RL = 5 kΩ
−105
−63
dB
dB
VCM = 0 V to 2 V
VCM = 0 V to 2 V
0.2
0.5
2
3
0.5
2
0.5
30
VO = 0.2 V to 2 V, RL = 2 kΩ
13
15
12
−0.2 to +1
VCM = 0 V to 1 V
54
VOUT = 0.5 V to 2.5 V
Sourcing to 1.5 V
Sinking to 1.5 V
G = +1
Rev. B | Page 4 of 20
70
1.5
2.5
25
5
20
Unit
mV
mV
μV/°C
pA
nA
pA
nA
V/mV
V/mV
−0.2 to +1.8
1013
1.8
70
V
Ω
pF
dB
0.025 to 3.275
0.08 to 3.22
0.25 to 3.05
15
40
30
500
V
V
V
mA
mA
mA
pF
3
TMIN to TMAX, total
VS = 3.3 V to 15 V, TMIN to TMAX
Max
5.0
80
36
5.7
V
mA
dB
AD823
At TA = 25°C, VS = ±15 V, RL = 2 kΩ to 0 V, unless otherwise noted.
Table 3.
Parameter
DYNAMIC PERFORMANCE
−3 dB Bandwidth, VO ≤ 0.2 V p-p
Full Power Response
Slew Rate
Settling Time
to 0.1%
to 0.01%
NOISE/DISTORTION PERFORMANCE
Input Voltage Noise
Input Current Noise
Harmonic Distortion
Crosstalk
f = 1 kHz
f = 1 MHz
DC PERFORMANCE
Initial Offset
Maximum Offset Over temperature
Offset Drift
Input Bias Current
at TMAX
Input Offset Current
at TMAX
Open-Loop Gain
TMIN to TMAX
INPUT CHARACTERISTICS
Input Common-Mode Voltage Range
Input Resistance
Input Capacitance
Common-Mode Rejection Ratio
OUTPUT CHARACTERISTICS
Output Voltage Swing
IL = ±100 μA
IL = ±2 mA
IL = ±10 mA
Output Current
Short-Circuit Current
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current
Power Supply Rejection Ratio
Conditions
Min
Typ
G = +1
VO = 2 V p-p
G = −1, VO = 10 V Step
12
16
4
25
MHz
MHz
V/μs
G = −1, VO = 10 V Step
G = −1, VO = 10 V Step
550
650
ns
ns
f = 10 kHz
f = 1 kHz
RL = 600 Ω, VO = 10 V p-p, f = 20 kHz
16
1
−90
nV/√Hz
fA/√Hz
dBc
RL= 5 kΩ
RL= 5 kΩ
−105
−63
dB
dB
VCM = 0 V
VCM = −10 V
VCM = 0 V
0.7
1.0
2
5
60
0.5
2
0.5
60
VO = +10 V to −10 V, RL = 2 kΩ
17
30
30
−15.2 to +13
VCM = −15 V to +13 V
66
VOUT = −14.5 V to +14.5 V
Sourcing to 0 V
Sinking to 0 V
G = +1
Rev. B | Page 5 of 20
70
3.5
7
30
5
20
Unit
mV
mV
μV/°C
pA
pA
nA
pA
nA
V/mV
V/mV
−15.2 to +13.8
1013
1.8
82
V
Ω
pF
dB
−14.95 to +14.95
−14.92 to +14.92
−14.75 to +14.75
17
80
60
500
V
V
V
mA
mA
mA
pF
3
TMIN to TMAX, total
VS = 5 V to 15 V, TMIN to TMAX
Max
7.0
80
36
8.4
V
mA
dB
AD823
ABSOLUTE MAXIMUM RATINGS
Table 4
2.0
TJ = 150°C
8-LEAD PDIP
1.3 W
0.9 W
±VS
±1.2 V
See Figure 4
−65°C to +125°C
−40°C to +85°C
300°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
1.5
1.0
8-LEAD SOIC
0.5
0
–50 –40 –30 –20 –10 0 10 20 30 40 50 60
AMBIENT TEMPERATURE (°C)
70
80 90
00901-004
Rating
36 V
MAXIMUM POWER DISSIPATION (W)
Parameter
Supply Voltage
Internal Power Dissipation
PDIP (N)
SOIC (R)
Input Voltage (Common Mode)
Differential Input Voltage
Output Short-Circuit Duration
Storage Temperature Range N, R
Operating Temperature Range
Lead Temperature Range
(Soldering, 10 sec)
Figure 4. Maximum Power Dissipation vs. Temperature
THERMAL RESISTANCE
θJA is specified for the worst-case conditions, that is, a device
soldered in a circuit board for surface-mount packages.
Specification is for device in free air.
Table 5. Thermal Resistance
Package Type
8-Lead PDIP
8-Lead SOIC
ESD CAUTION
Rev. B | Page 6 of 20
θJA
90
160
Unit
°C/W
°C/W
AD823
TYPICAL PERFORMANCE CHARACTERISTICS
80
100
+VS = +5V
314 UNITS
σ = 40µV
70
+VS = +5V
317 UNITS
σ = 0.4pA
90
80
60
70
60
UNITS
UNITS
50
40
50
40
30
30
20
20
10
–100
–50
0
50
100
INPUT OFFSET VOLTAGE (µV)
150
200
0
00901-005
–150
0
Figure 5. Typical Distribution of Input Offset Voltage
22
18
10000
3
4
5
6
7
INPUT BIAS CURRENT (pA)
8
9
10
+VS = +5V
VCM = 0V
INPUT BIAS CURRENT (pA)
1000
16
14
UNITS
2
Figure 8. Typical Distribution of Input Bias Current
+VS = +5V
–55°C TO +125°C
103 UNITS
20
1
00901-008
10
0
–200
12
10
8
6
4
100
10
1
–5
–4
–3 –2 –1 0
3
4
5
1
2
INPUT OFFSET VOLTAGE DRIFT (µV/°C)
6
7
0.1
0
00901-006
0
–6
Figure 6. Typical Distribution of Input Offset Voltage Drift
3
25
50
75
TEMPERATURE (°C)
100
125
00901-009
2
Figure 9. Input Bias Current vs. Temperature
1000
+VS = +5V
VS = ±15V
INPUT BIAS CURRENT (pA)
1
0
–1
–2
100
10
1
–4
–5
–4
–3
–2
–1
0
1
2
3
COMMON-MODE VOLTAGE (V)
4
5
0.1
–16
Figure 7. Input Bias Current vs. Common-Mode Voltage
–12
–8
–4
0
4
8
COMMON-MODE VOLTAGE (V)
12
16
Figure 10. Input Bias Current vs. Common-Mode Voltage
Rev. B | Page 7 of 20
00901-010
–3
00901-007
INPUT BIAS CURRENT (pA)
2
AD823
110
95
VS = ±2.5V
RL = 2kΩ
+VS = +5V
94
93
OPEN-LOOP GAIN (dB)
OPEN-LOOP GAIN (dB)
100
90
80
92
91
90
89
88
70
500k
86
–55
Figure 11. Open-Loop Gain vs. Load Resistance
100
95
125
RL = 2kΩ
CL = 20pF
100
80
80
OPEN-LOOP GAIN (dB)
100
RL = 1kΩ
10
RL = 100Ω
0.1
0.5
1.0
–2.5 –2.0 –1.5 –1.0 –0.5
0
OUTPUT VOLTAGE (V)
1.5
2.0
2.5
00901-012
1
PHASE
60
40
40
GAIN
20
20
0
0
–20
100
1k
10k
100k
1M
FREQUENCY (Hz)
100
–40
ALL
OTHERS
VS = ±2.5V
VOUT = 2V p-p
VS = ±15V
RL = 1kΩ
VOUT = 10V p-p
RL = 600Ω
–80
–90
+VS = +5V
VOUT = 2V p-p
RL = 5kΩ
–100
–110
100
+VS = +3V
VOUT = 2V p-p
RL = 5kΩ
1k
10k
100k
FREQUENCY (Hz)
1M
Figure 13. Total Harmonic Distortion vs. Frequency
+VS = +5V
30
10
3
00901-013
–70
INPUT VOLTAGE NOISE (nV/√Hz)
–50
+VS = +3V
VOUT = 2V p-p
RL = 100Ω
–20
100M
10M
Figure 15. Open-Loop Gain and Phase Margin vs. Frequency
Figure 12. Open-Loop Gain vs. Output Voltage, VS = ±2.5 V
–60
60
PHASE MARGIN (Degrees)
RL = 10kΩ
OPEN-LOOP GAIN (k V )
V
5
35
65
TEMPERATURE (°C)
Figure 14. Open-Loop Gain vs. Temperature
1000
THD (dB)
–25
00901-015
100k
10k
LOAD RESISTANCE (Ω)
10
100
1k
10k
FREQUENCY (Hz)
100k
Figure 16. Input Voltage Noise vs. Frequency
Rev. B | Page 8 of 20
1M
00901-016
1k
00901-011
60
100
00901-014
87
AD823
5
90
CL = 20pF
RL = 2kΩ
G = +1
4
VS = ±15V
80
+VS = +5V
CLOSED-LOOP GAIN (dB)
3
70
2
CMRR (dB)
1
0
+27°C
–1
–55°C
60
50
+125°C
–2
40
–3
20
10
00901-017
–5
0.30 3.27 6.24 9.21 12.18 15.15 18.12 21.09 24.06 27.03 30.00
FREQUENCY (MHz)
10
0.1
10k
100k
FREQUENCY (Hz)
1M
10M
VS – VOH
25°C
0.1
0.01
0.1
1
10
LOAD CURRENT (mA)
100
10
1%
0.1%
6
0.01%
QUIESCENT CURRENT (mA)
VS = ±15V
CL = 20pF
4
2
0
–2
–4
0.1%
1%
–6
VOL
25°C
Figure 21. Output Saturation Voltage vs. Load Current
0.01%
+125°C
8
+25°C
6
–55°C
4
2
–10
100
200
300
400
500
600
700
SETTLING TIME (ns)
0
0
5
10
15
SUPPLY VOLTAGE (±V)
Figure 22. Quiescent Current vs. Supply Voltage
Figure 19. Output Step Size vs. Settling Time (Inverter)
Rev. B | Page 9 of 20
20
00901-022
–8
00901-019
OUTPUT STEP SIZE FROM 0V TO VSHOWN (V)
8
10M
1
Figure 18. Output Resistance vs. Frequency, +VS = +5 V, Gain = +1
10
1M
00901-021
OUTPUT SATURATION VOLTAGE (V)
1
00901-018
OUTPUT RESISTANCE (Ω)
10
1k
10k
100k
FREQUENCY (Hz)
+VS = +5V
+VS = +5V
GAIN = +1
0.01
100
1k
Figure 20. Common-Mode Rejection Ratio vs. Frequency
Figure 17. Closed-Loop Gain vs. Frequency
100
100
00901-020
30
–4
AD823
100
90
+VS = +5V
80
70
+PSRR
60
50
–PSRR
40
30
CL
15
12
9
ФM = 45°
6
ФM = 20°
20
10k
100k
FREQUENCY (Hz)
1M
10M
0
00901-023
1k
1
0
8
–30
RL = 2kΩ
G = +1
–40
9
10
+VS = +5V
–50
CROSSTALK (dB)
20
VS = ±15V
10
+VS = +5V
–70
–80
–90
–100
–110
10M
–130
1k
00901-024
100k
1M
FREQUENCY (Hz)
10k
Figure 24. Large Signal Frequency Response
100k
FREQUENCY (Hz)
Figure 27. Crosstalk vs. Frequency
VIN = 20V p-p
VS = ±15V
G = +1
VIN = 2.9V p-p
+VS = +3V
G = –1
500mV
5V
10µs
100kΩ
20µs
+15V
3V
VIN = 2.9V p-p
20kHz, 20V p-p
VOUT
50Ω
100kΩ
50pF
–15V
604Ω
50pF
Figure 25. Output Swing, +VS = +3 V, G = −1
Figure 28. Output Swing, VS = ±15 V, G = +1
Rev. B | Page 10 of 20
00901-028
100kΩ
10M
1M
00901-027
–120
+VS = +3V
0
10k
–60
00901-025
OUTPUT VOLTAGE (V p-p)
3
4
5
6
7
CAPACITOR (pF × 1000)
Figure 26. Series Resistance vs. Capacitive Load
Figure 23. Power Supply Rejection vs. Frequency
30
2
00901-026
3
10
0
100
RS
VIN
18
SERIES RESISTANCE (Ω)
POWER SUPPLY REJECTION (dB)
21
+VS = +5V
AD823
5V
RL = 300Ω
CL = 50pF
RF = RG = 2kΩ
+VS = +5V
G = –1
RL = 100kΩ
CL = 50pF
+VS = +3V
G = +1
3V
200µs
500mV
Figure 29. Output Swing, +VS = +5 V, G = −1
00901-032
500mV
GND
00901-029
GND
200µs
Figure 32. Output Swing, +VS = +3 V, G = +1
5V
VIN = 100mV STEP
+VS = +3V
G = +1
RL = 2kΩ
CL = 50pF
+VS = +5V
G = +1
1.55V
Figure 30. Pulse Response, +VS = +3 V, G = +1
RL = 2kΩ
CL = 50pF
+VS = +5V
G = +2
500mV
100ns
GND
Figure 33. Pulse Response, +VS = +5 V, G = +1
RL = 2kΩ
CL = 470pF
+VS = +5V
G = +1
100ns
GND
00901-031
5V
500mV
00901-033
50ns
Figure 31. Pulse Response, +VS = +5 V, G = +2
500mV
200ns
Figure 34. Pulse Response, +VS = +5 V, G = +1, CL = 470 pF
Rev. B | Page 11 of 20
00901-034
25mV
00901-030
1.45V
AD823
RL = 100kΩ
CL = 50pF
VS = ±15V
G = +1
+10V
5V
500ns
00901-035
–10V
Figure 35. Pulse Response, VS = ±15 V, G = +1
Rev. B | Page 12 of 20
AD823
THEORY OF OPERATION
The AD823 is fabricated on the Analog Devices, Inc. proprietary
complementary bipolar (CB) process that enables the construction
of PNP and NPN transistors with similar fT’s in the 600 MHz to
800 MHz region. In addition, the process also features N-Channel
JFETs that are used in the input stage of the AD823. These
process features allow the construction of high frequency, low
distortion op amps with picoamp input currents. This design
uses a differential output input stage to maximize bandwidth
and headroom (see Figure 36). The smaller signal swings
required on the S1P/S1N outputs reduce the effect of the
nonlinear currents due to junction capacitances and improve
the distortion performance. With this design, harmonic
distortion of better than −91 dB @ 20 kHz into 600 Ω with
VOUT = 4 V p-p on a single 5 V supply is achieved. The
complementary common emitter design of the output stage
provides excellent load drive without the need for emitter
followers, thereby improving the output range of the device
considerably with respect to conventional op amps. The
AD823 can drive 20 mA with the outputs within 0.6 V of the
supply rails. The AD823 also offers outstanding precision for a
high speed op amp. Input offset voltages of 1 mV maximum
and offset drift of 2 μV/°C are achieved through the use of the
Analog Devices advanced thin film trimming techniques.
A nested integrator topology is used in the AD823 (see Figure 37).
The output stage can be modeled as an ideal op amp with a
single-pole response and a unity-gain frequency set by
transconductance gm2 and Capacitor C2. R1 is the output
impedance of the input stage; gm is the input transconductance.
C1 and C5 provide Miller compensation for the overall op amp.
The unity-gain frequency occurs at gm/C5. Solving the node
equations for this circuit yields
V OUT
Vi
=
A0
(sR1[C1( A2 + 1)] + 1) × ⎛⎜⎜ s ⎡⎢
g m2 ⎤ ⎞
⎥ + 1⎟⎟
⎣
⎝ C2 ⎦ ⎠
where:
A0 = gmgm2 R2R1 (open-loop gain of op amp)
A2 = gm2 R2 (open-loop gain of output stage).
The first pole in the denominator is the dominant pole of the
amplifier and occurs at ~18 Hz. This equals the input stage
output impedance R1 multiplied by the Miller-multiplied value
of C1. The second pole occurs at the unity-gain bandwidth of
the output stage, which is 23 MHz. This type of architecture
allows more open-loop gain and output drive to be obtained
than a standard 2-stage architecture would allow.
VCC
R42
R37
VBE + 0.3V V1
Q43
I5
Q55
Q44
A=1
I6
Q57
A = 19
Q61
Q72
J1
Q49
Q18
Q46
J6
R44
S1P
S1N
VOUT
Q54
Q21
VINN
Q62
Q60
VCC
C1
Q48
Q53
I1
C6
R33
C2
R28
VB
Q35
I2
Q17
A = 19
R43
I3
Q56
VEE
Figure 36. Simplified Schematic
Rev. B | Page 13 of 20
Q52
I4
Q59
A=1
00901-036
VINP
Q58
AD823
OUTPUT IMPEDANCE
Rev. B | Page 14 of 20
S1N
gmVI
C1
R1
VOUT
S1P
C2
gmVI
R1
C5
gm2
R2
00901-037
The low frequency open-loop output impedance of the commonemitter output stage used in this design is approximately 30 kΩ.
Although this is significantly higher than a typical emitter
follower output stage, when it is connected with feedback, the
output impedance is reduced by the open-loop gain of the op
amp. With 109 dB of open-loop gain, the output impedance is
reduced to <0.2 Ω. At higher frequencies, the output impedance
rises as the open-loop gain of the op amp drops; however, the
output also becomes capacitive due to the integrator capacitors
C1 and C2. This prevents the output impedance from ever
becoming excessively high (see Figure 18), which can cause
stability problems when driving capacitive loads. In fact, the AD823
has excellent cap-load drive capability for a high frequency op
amp. Figure 34 shows the AD823 connected as a follower while
driving 470 pF direct capacitive load. Under these conditions,
the phase margin is approximately 20°. If greater phase margin
is desired, a small resistor can be used in series with the output
to decouple the effect of the load capacitance from the op amp
(see Figure 26). In addition, running the part at higher gains
also improves the capacitive load drive capability of the op amp.
Figure 37. Small Signal Schematic
AD823
APPLICATION NOTES
INPUT CHARACTERISTICS
In the AD823, N-Channel JFETs are used to provide a low offset,
low noise, high impedance input stage. Minimum input commonmode voltage extends from 0.2 V below −VS to 1 V < +VS. Driving
the input voltage closer to the positive rail causes a loss of amplifier
bandwidth and increased common-mode voltage error.
The AD823 does not exhibit phase reversal for input voltages up
to and including +VS. Figure 38 shows the response of an
AD823 voltage follower to a 0 V to 5 V (+VS) square wave input.
The input and output are superimposed. The output polarity
tracks the input polarity up to +VS, with no phase reversal. The
reduced bandwidth above a 4 V input causes the rounding of
the output wave form. For input voltages greater than +VS, a
resistor in series with the AD823’s noninverting input prevents
phase reversal, at the expense of greater input voltage noise.
This is illustrated in Figure 39.
1V
2µs
100
90
10
00901-038
GND 0%
1V
Figure 38. AD823 Input Response: RP = 0, VIN = 0 to +VS
1V
A current limiting resistor should be used in series with the
input of the AD823 if there is a possibility of the input voltage
exceeding the positive supply by more than 300 mV, or if an
input voltage is applied to the AD823 when ±VS = 0. The
amplifier becomes damaged if left in that condition for more
than 10 seconds. A 1 kΩ resistor allows the amplifier to
withstand up to 10 V of continuous overvoltage and increases
the input voltage noise by a negligible amount.
Input voltages less than −VS are a completely different story.
The amplifier can safely withstand input voltages 20 V below
−VS as long as the total voltage from the positive supply to the
input terminal is less than 36 V. In addition, the input stage
typically maintains picoamp level input currents across that
input voltage range.
The AD823 is designed for 16 nV/√Hz wideband input voltage
noise and maintains low noise performance to low frequencies
(see Figure 16). This noise performance, along with the AD823’s
low input current and current noise, means that the AD823
contributes negligible noise for applications with source
resistances greater than 10 kΩ and signal bandwidths greater
than 1 kHz.
OUTPUT CHARACTERISTICS
The AD823’s unique bipolar rail-to-rail output stage swings
within 25 mV of the supplies with no external resistive load.
The AD823’s approximate output saturation resistance is 25 Ω
sourcing and sinking. This can be used to estimate the output
saturation voltage when driving heavier current loads. For
instance, when driving 5 mA, the saturation voltage to the rails
is approximately 125 mV.
10 µs
100
+VS
Because the input stage uses N-Channel JFETs, input current
during normal operation is negative; the current flows out from
the input terminals. If the input voltage is driven more positive
than +VS − 0.4 V, the input current reverses direction as internal
device junctions become forward biased. This is illustrated in
Figure 7.
90
If the AD823’s output is driven hard against the output
saturation voltage, it recovers within 250 ns of the input
returning to the amplifier’s linear operating region.
A/D Driver
10
0%
1V
RP
VIN
5V
AD823
VOUT
00901-039
GND
The rail-to-rail output of the AD823 makes it useful as an A/D
driver in a single-supply system. Because it is a dual op amp, it
can be used to drive both the analog input of the A/D as well as
its reference input. The high impedance FET input of the
AD823 is well suited for minimal loading of high output
impedance devices.
Figure 39. AD823 Input Response:
VIN = 0 to +VS + 200 mV, VOUT = 0 to +VS, RP = 49.9 kΩ
Rev. B | Page 15 of 20
AD823
Figure 40 shows a schematic of an AD823 being used to drive
both the input and reference input of an AD1672, a 12-bit,
3-MSPS, single-supply ADC. One amplifier is configured as a
unity-gain follower to drive the analog input of the AD1672,
which is configured to accept an input voltage that ranges from
0 V to 2.5 V.
The distortion analysis is important for systems requiring good
frequency domain performance. Other systems may require
good time domain performance. The noise and settling time
performance of the AD823 provides the necessary information
for its applicability for these systems.
1
VIN = 2.15V p-p
G = +1
FI = 490kHz
9
6
5
7
3
8
+5VD
0.1µF
0.1µF
+5VA
10µF
00901-041
10µF
2
4
1
3
VIN
49.9Ω
20
REFOUT
21
AIN1
22
AIN2
AD823
5
VREF
(1.25V)
AD1672
7
6
4
1kΩ
1kΩ
23
REFIN
24
REFCOM
25
NCOMP2
26
NCOMP1
27
16
15
13
14
12
11
10
9
8
7
6
5
4
3
2
1
3 V, Single-Supply Stereo Headphone Driver
OTR
The AD823 exhibits good current drive and total harmonic
distortion plus noise (THD+N) performance, even at 3 V
single supplies. At 20 kHz, THD+N equals −62 dB (0.079%) for
a 300 mV p-p output signal. This is comparable to other singlesupply op amps that consume more power and cannot run on
3 V power supplies.
BIT1 (MSB)
BIT2
BIT3
BIT4
BIT5
BIT6
BIT7
BIT8
BIT9
BIT10
BIT11
BIT12 (LSB)
REF
COM
CLOCK
ACOM
Figure 41. FFT of AD1672 Output Driven by AD823
0.1µF
19
18
00901-040
8
+5VD
DCOM
2
10µF
+VDD
0.1µF
+VCC
28 19
Figure 40. AD823 Driving Input and Reference of the
AD1672, a 12-Bit, 3-MSPS ADC
The circuit was tested with a 500 kHz sine wave input that was
heavily low-pass filtered (60 dB) to minimize the harmonic content
at the input to the AD823. The digital output of the AD1672 was
analyzed by performing a fast Fourier transform (FFT).
In Figure 42, each channel’s input signal is coupled via a 1 μF
Mylar capacitor. Resistor dividers set the dc voltage at the
noninverting inputs so that the output voltage is midway
between the power supplies (+1.5 V). The gain is 1.5. Each half
of the AD823 can then be used to drive a headphone channel. A
5 Hz high-pass filter is realized by the 500 μF capacitors and the
headphones that can be modeled as 32 Ω load resistors to
ground. This ensures that all signals in the audio frequency
range (20 Hz to 20 kHz) are delivered to the headphones.
During the testing, it was observed that at 500 kHz, the output
of the AD823 cannot go below ~350 mV (operating with
negative supply at ground) without seriously degrading the
second harmonic distortion. Another test was performed with a
200 Ω pull-down resistor to ground that allowed the output to
go as low as 200 mV without seriously affecting the second
harmonic distortion. There was, however, a slight increase in
the third harmonic term with the resistor added, but it was still
less than the second harmonic.
Figure 41 is an FFT plot of the results of driving the AD1672
with the AD823 with no pull-down resistor. The input
amplitude was 2.15 V p-p and the lower voltage excursion was
350 mV. The input frequency was 490 kHz, which was chosen
to spread the location of the harmonics.
Rev. B | Page 16 of 20
3V
95.3kΩ
95.3kΩ
1µF
MYLAR
47.5kΩ
0.1µF
3
CHANNEL 1
1/2
2
95.3kΩ
8
AD823
10kΩ
1
+
0.1µF
+
500µF
L
4.99kΩ
HEADPHONES
32Ω IMPEDANCE
10kΩ
R
4.99kΩ
6
1µF
47.5kΩ
5
CHANNEL 2
1/2
AD823
4
500µF
7
+
MYLAR
Figure 42. 3 V Single-Supply Stereo Headphone Driver
00901-042
+5VA
15dB/DIV
The other amplifier is configured as a gain of 2 to drive the
reference input from a 1.25 V reference. Although the AD1672
has its own internal reference, there are systems that require
greater accuracy than the internal reference provides. On the other
hand, if the AD1672 internal reference is used, the second AD823
amplifier can be used to buffer the reference voltage for driving
other circuitry while minimally loading the reference source.
AD823
Second-Order Low-Pass Filter
Single-Supply Half-Wave and Full-Wave Rectifiers
Figure 43 depicts the AD823 configured as a second-order
Butterworth low-pass filter. With the values as shown, the
corner frequency equals 200 kHz. Component selection is
shown in the following equations:
An AD823 configured as a unity-gain follower and operated
with a single supply can be used as a simple half-wave rectifier.
The AD823 inputs maintain picoamp level input currents even
when driven well below the minus supply. The rectifier puts
that behavior to good use, maintaining an input impedance of
over 1011 Ω for input voltages from within 1 V of the positive
supply to 20 V below the negative supply.
R1 = R2 = User Selected (Typical Values: 10 kΩ to 100 kΩ)
C1( farads ) =
C2 =
1.414
2πf cutoff × R1
The full-wave and half-wave rectifier shown in Figure 45
operates as follows: when VIN is above ground, R1 is bootstrapped through the unity-gain follower A1 and the loop of
Amplifier A2. This forces the inputs of A2 to be equal, thus no
current flows through R1 or R2, and the circuit output tracks
the input. When VIN is below ground, the output of A1 is forced
to ground. The noninverting input of Amplifier A2 sees the
ground level output of A1; therefore, A2 operates as a unitygain inverter. The output at Node C is then a full-wave rectified
version of the input. Node B is a buffered half-wave rectified
version of the input. Input voltage supply to ±18 V can be
rectified, depending on the voltage supply used.
0.707
2πf cutoff × R1
C2
56pF
R1
20kΩ
+5V
C3
0.1µF
R2
20kΩ
VIN
C1
28pF
1/2
AD823
VOUT
50pF
R1
100kΩ
00901-043
C4
0.1µF
–5V
+VS
Figure 43. Second-Order Low-Pass Filter
A
A plot of the filter is shown in Figure 44; better than 50 dB of
high frequency rejection is provided.
3
VIN
2
6
1
A1
4
1/2
AD823
5
A2
A2
7
1/2
AD823
C
FULL-WAVE
RECTIFIED OUTPUT
HALF-WAVE
RECTIFIED OUTPUT
VDB – VOUT
–20
Figure 45. Full-Wave and Half-Wave Rectifier
–30
–40
2V
–50
200µs
100
A 90
10k
100k
1M
FREQUENCY (Hz)
10M
100M
Figure 44. Frequency Response of Filter
B
10
C 0%
2V
Figure 46. Single-Supply Half-Wave and Full-Wave Rectifier
Rev. B | Page 17 of 20
00901-046
–60
1k
00901-044
B
–10
00901-044
HIGH FREQUENCY REJECTION (dB)
0.01µF
8
0
R2
100kΩ
AD823
OUTLINE DIMENSIONS
0.375 (9.53)
0.365 (9.27)
0.355 (9.02)
8
5
1
4
0.295 (7.49)
0.285 (7.24)
0.275 (6.98)
0.325 (8.26)
0.310 (7.87)
0.300 (7.62)
0.100 (2.54)
BSC
0.015
(0.38)
MIN
0.180
(4.57)
MAX
0.150 (3.81)
0.130 (3.30)
0.110 (2.79)
0.022 (0.56)
0.018 (0.46)
0.014 (0.36)
0.150 (3.81)
0.135 (3.43)
0.120 (3.05)
0.015 (0.38)
0.010 (0.25)
0.008 (0.20)
SEATING
PLANE
0.060 (1.52)
0.050 (1.27)
0.045 (1.14)
COMPLIANT TO JEDEC STANDARDS MO-095AA
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
Figure 47. 8-Lead Plastic Dual In-Line Package [PDIP]
Narrow Body
(N-8)
Dimensions shown in inches and (millimeters)
5.00 (0.1968)
4.80 (0.1890)
8
4.00 (0.1574)
3.80 (0.1497) 1
5
1.27 (0.0500)
BSC
0.25 (0.0098)
0.10 (0.0040)
6.20 (0.2440)
4 5.80 (0.2284)
1.75 (0.0688)
1.35 (0.0532)
0.51 (0.0201)
COPLANARITY
SEATING 0.31 (0.0122)
0.10
PLANE
0.50 (0.0196)
× 45°
0.25 (0.0099)
8°
0.25 (0.0098) 0° 1.27 (0.0500)
0.40 (0.0157)
0.17 (0.0067)
COMPLIANT TO JEDEC STANDARDS MS-012AA
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
Figure 48. 8-Lead Standard Small Outline Package [SOIC_N]
Narrow Body
(R-8)
Dimensions shown in millimeters and (inches)
Rev. B | Page 18 of 20
AD823
ORDERING GUIDE
Models
AD823AN
AD823ANZ 1
AD823AR
AD823AR-REEL
AD823AR-REEL7
AD823ARZ1
AD823ARZ-RL1
AD823ARZ-R71
1
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
Package Description
8-Lead PDIP
8-Lead PDIP
8-Lead SOIC_N
8-Lead SOIC_N, 13” Reel
8-Lead SOIC_N, 7” Reel
8-Lead SOIC_N
8-Lead SOIC_N, 13” Reel
8-Lead SOIC_N, 7” Reel
Z = RoHS Compliant Part.
Rev. B | Page 19 of 20
Package Option
N-8
N-8
R-8
R-8
R-8
R-8
R-8
R-8
AD823
NOTES
©1995–2007 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
C00901-0-2/07(B)
Rev. B | Page 20 of 20