Zero-Drift, Single-Supply, Rail-to-Rail Input/Output Operational Amplifiers AD8571/AD8572/AD8574 +IN A 3 V– TOP VIEW 6 (Not to Scale) 4 5 AD8571 8 NC NC 1 7 V+ –IN A 2 OUT A +IN A 3 NC NC = NO CONNECT 8 NC 7 V+ 6 OUT A TOP VIEW V– 4 (Not to Scale) 5 NC NC = NO CONNECT Figure 1. 8-Lead MSOP (RM Suffix) Figure 2. 8-Lead SOIC (R Suffix) OUT A 1 8 V+ 8 V+ –IN A 2 AD8572 7 OUT B –IN A 2 AD8572 7 OUT B +IN A 3 6 –IN B +IN A 3 6 –IN B 4 5 +IN B V– TOP VIEW (Not to Scale) V– TOP VIEW (Not to Scale) 4 5 +IN B OUT A 1 Figure 3. 8-Lead TSSOP (RU Suffix) OUT A 1 13 –IN D GENERAL DESCRIPTION +IN A 3 12 +IN D This family of amplifiers has ultralow offset, drift, and bias current. The AD8571, AD8572, and AD8574 are single, dual, and quad amplifiers, respectively, featuring rail-to-rail input and output swings. All are guaranteed to operate from 2.7 V to 5 V single supply. +IN B 5 V+ 4 Figure 4. 8-Lead SOIC (R Suffix) 14 OUT D –IN A 2 The AD857x family provides benefits previously found only in expensive auto-zeroing or chopper-stabilized amplifiers. Using Analog Devices, Inc. topology, these zero-drift amplifiers combine low cost with high accuracy. (No external capacitors are required.) Using a patented spread-spectrum auto-zero technique, the AD857x family eliminates the intermodulation effects from interaction of the chopping function with the signal frequency in ac applications. AD8571 AD8574 9 –IN C OUT B 7 8 OUT C Figure 5. 14-Lead TSSOP (RU Suffix) OUT A 1 14 OUT D –IN A 2 13 –IN D 12 +IN D +IN A 3 V+ 4 11 V– TOP VIEW (Not to Scale) 10 +IN C –IN B 6 01104-004 2 01104-005 1 +IN B 5 AD8574 11 V– TOP VIEW (Not to Scale) 10 +IN C –IN B 6 9 –IN C OUT B 7 8 OUT C 01104-006 Temperature sensors Pressure sensors Precision current sensing Strain gage amplifiers Medical instrumentation Thermocouple amplifiers NC –IN A 01104-002 APPLICATIONS PIN CONFIGURATIONS 01104-003 Low offset voltage: 1 μV Input offset drift: 0.005 μV/°C Rail-to-rail input and output swing 5 V/2.7 V single-supply operation High gain, CMRR, PSRR: 130 dB Ultralow input bias current: 20 pA Low supply current: 750 μA/op amp Overload recovery time: 50 μs No external capacitors required 01104-001 FEATURES Figure 6. 14-Lead SOIC (R Suffix) The AD857x family is specified for the extended industrial/ automotive (−40°C to +125°C) temperature range. The AD8571 single amplifier is available in 8-lead MSOP and narrow 8-lead SOIC packages. The AD8572 dual amplifier is available in 8-lead narrow SOIC and 8-lead TSSOP surface mount packages. The AD8574 quad amplifier is available in narrow 14-lead SOIC and 14-lead TSSOP packages. With an offset voltage of only 1 μV and drift of 0.005 μV/°C, the AD857x family is perfectly suited for applications where error sources cannot be tolerated. Position and pressure sensors, medical equipment, and strain gage amplifiers benefit greatly from nearly zero drift over their operating temperature range. Many more systems require the rail-to-rail input and output swings provided by the AD857x family. Rev. B Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 ©2006 Analog Devices, Inc. All rights reserved. AD8571/AD8572/AD8574 TABLE OF CONTENTS Features .............................................................................................. 1 1/f Noise Characteristics ........................................................... 17 Applications....................................................................................... 1 Random Auto-Zero Correction Eliminates Intermodulation Distortion .................................................................................... 17 General Description ......................................................................... 1 Pin Configurations ........................................................................... 1 Revision History ............................................................................... 2 Specifications..................................................................................... 3 5 V Electrical Characteristics...................................................... 3 2.7 V Electrical Characteristics................................................... 4 Absolute Maximum Ratings............................................................ 5 Thermal Characteristics .............................................................. 5 ESD Caution.................................................................................. 5 Typical Performance Characteristics ............................................. 6 Functional Description .................................................................. 14 Amplifier Architecture .............................................................. 14 Basic Auto-Zero Amplifier Theory.......................................... 14 Auto-Zero Phase......................................................................... 14 Amplification Phase ................................................................... 15 Broadband and External Resistor Noise Considerations.......... 18 Output Overdrive Recovery...................................................... 18 Input Overvoltage Protection ................................................... 18 Output Phase Reversal............................................................... 18 Capacitive Load Drive ............................................................... 19 Power-Up Behavior .................................................................... 19 Applications..................................................................................... 20 5 V Precision Strain Gage Circuit ............................................ 20 3 V Instrumentation Amplifier ................................................ 20 High Accuracy Thermocouple Amplifier ............................... 20 Precision Current Meter............................................................ 21 Precision Voltage Comparator.................................................. 21 Outline Dimensions ....................................................................... 22 Ordering Guide .......................................................................... 23 High Gain, CMRR, PSRR.......................................................... 16 Maximizing Performance Through Proper Layout ................ 16 REVISION HISTORY 09/06—Rev. A to Rev. B 07/03—Rev. 0 to Rev. A Updated Format..................................................................Universal Renumbered Figures ..........................................................Universal Changes to Figure 50...................................................................... 14 Changes to Figure 51...................................................................... 15 Changes to Figure 66...................................................................... 21 Updated Outline Dimensions ....................................................... 22 Changes to Ordering Guide .......................................................... 23 Renumbered Figures ..........................................................Universal Changes to Ordering Guide .............................................................4 Change to Figure 15. ...................................................................... 16 Updated Outline Dimensions....................................................... 19 10/99—Revision 0: Initial Version Rev. B | Page 2 of 24 AD8571/AD8572/AD8574 SPECIFICATIONS 5 V ELECTRICAL CHARACTERISTICS VS = 5 V, VCM = 2.5 V, VO = 2.5 V, TA = 25°C, unless otherwise noted. Table 1. Parameter INPUT CHARACTERISTICS Offset Voltage Symbol Conditions Min VOS Typ Max Unit 1 5 10 50 1.5 70 200 5 μV μV pA nA pA pA V dB dB dB dB μV/°C −40°C ≤ TA ≤ +125°C Input Bias Current IB 10 1.0 20 150 −40°C ≤ TA ≤ +125°C Input Offset Current IOS −40°C ≤ TA ≤ +125°C Input Voltage Range Common-Mode Rejection Ratio CMRR Large Signal Voltage Gain 1 AVO Offset Voltage Drift OUTPUT CHARACTERISTICS Output Voltage High ∆VOS/∆T VOH Output Voltage Low VOL Short-Circuit Limit ISC VCM = 0 V to 5 V −40°C ≤ TA ≤ +125°C RL = 10 kΩ, VO = 0.3 V to 4.7 V −40°C ≤ TA ≤ +125°C −40°C ≤ TA ≤ +125°C RL = 100 kΩ to GND −40°C to +125°C RL = 10 kΩ to GND −40°C to +125°C RL = 100 kΩ to V+ −40°C to +125°C RL = 10 kΩ to V+ −40°C to +125°C 0 120 115 125 120 4.99 4.99 4.95 4.95 ±25 −40°C to +125°C Output Current IO −40°C to +125°C POWER SUPPLY Power Supply Rejection Ratio Supply Current/Amplifier DYNAMIC PERFORMANCE Slew Rate Overload Recovery Time Gain Bandwidth Product NOISE PERFORMANCE Voltage Noise Voltage Noise Density Current Noise Density 1 PSRR ISY SR VS = 2.7 V to 5.5 V −40°C ≤ TA ≤ +125°C VO = 0 V −40°C ≤ TA ≤ +125°C RL = 10 kΩ GBP en p-p en p-p en in 0 Hz to 10 Hz 0 Hz to 1 Hz f = 1 kHz f = 10 Hz Gain testing is dependent upon test bandwidth. Rev. B | Page 3 of 24 120 115 140 130 145 135 0.005 4.998 4.997 4.98 4.975 1 2 10 15 ±50 ±40 ±30 ±15 130 130 850 1000 0.4 0.05 1.5 1.3 0.41 51 2 0.04 10 10 30 30 975 1075 0.3 V V V V mV mV mV mV mA mA mA mA dB dB μA μA V/μs ms MHz μV p-p μV p-p nV/√Hz fA/√Hz AD8571/AD8572/AD8574 2.7 V ELECTRICAL CHARACTERISTICS VS = 2.7 V, VCM = 1.35 V, VO = 1.35 V, TA = 25°C, unless otherwise noted. Table 2. Parameter INPUT CHARACTERISTICS Offset Voltage Symbol Conditions Min VOS Typ Max Unit 1 5 10 50 1.5 50 200 2.7 μV μV pA nA pA pA V dB dB dB dB μV/°C −40°C ≤ TA ≤ +125°C Input Bias Current IB 10 1.0 10 150 −40°C ≤ TA ≤ +125°C Input Offset Current IOS −40°C ≤ TA ≤ +125°C Input Voltage Range Common-Mode Rejection Ratio CMRR Large Signal Voltage Gain 1 AVO Offset Voltage Drift OUTPUT CHARACTERISTICS Output Voltage High ∆VOS/∆T VOH Output Voltage Low VOL Short-Circuit Limit ISC VCM = 0 V to 2.7 V −40°C ≤ TA ≤ +125°C RL = 10 kΩ, VO = 0.3 V to 2.4 V −40°C ≤ TA ≤ +125°C −40°C ≤ TA ≤ +125°C RL = 100 kΩ to GND −40°C to +125°C RL = 10 kΩ to GND −40°C to +125°C RL = 100 kΩ to V+ −40°C to +125°C RL = 10 kΩ to V+ −40°C to +125°C 0 115 110 110 105 2.685 2.685 2.67 2.67 ±10 −40°C to +125°C Output Current IO −40°C to +125°C POWER SUPPLY Power Supply Rejection Ratio Supply Current/Amplifier DYNAMIC PERFORMANCE Slew Rate Overload Recovery Time Gain Bandwidth Product NOISE PERFORMANCE Voltage Noise Voltage Noise Density Current Noise Density 1 PSRR ISY SR VS = 2.7 V to 5.5 V −40°C ≤ TA ≤ +125°C VO = 0 V −40°C ≤ TA ≤ +125°C 2.697 2.696 2.68 2.675 1 2 10 15 ±15 ±10 ±10 ±5 130 130 750 950 0.04 10 10 20 20 900 1000 V V V V mV mV mV mV mA mA mA mA dB dB μA μA RL = 10 kΩ 0.5 0.05 1 V/μs ms MHz 0 Hz to 10 Hz f = 1 kHz f = 10 Hz 2.0 94 2 μV p-p nV/√Hz fA/√Hz GBP en p-p en in 120 115 130 130 140 130 0.005 Gain testing is dependent upon test bandwidth. Rev. B | Page 4 of 24 AD8571/AD8572/AD8574 ABSOLUTE MAXIMUM RATINGS THERMAL CHARACTERISTICS Table 3. Parameter Supply Voltage Input Voltage Differential Input Voltage1 ESD (Human Body Model) Output Short-Circuit Duration to GND Storage Temperature Range RM, RU, and R Packages Operating Temperature Range AD8571A/AD8572A/AD8574A Junction Temperature Range RM, RU, and R Packages Lead Temperature Range (Soldering, 60 sec) 1 Rating 6V GND to VS + 0.3 V ±5.0 V 2000 V Indefinite −65°C to +150°C −40°C to +125°C θJA is specified for the worst-case conditions, that is, θJA is specified for a device soldered in a circuit board for SOIC and TSSOP packages. Table 4. Thermal Resistance Package Type 8-Lead MSOP (RM) 8-Lead TSSOP (RU) 8-Lead SOIC (R) 14-Lead TSSOP (RU) 14-Lead SOIC (R) −65°C to +150°C 300°C ESD CAUTION Differential input voltage is limited to ±5.0 V or the supply voltage, whichever is less. Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. Rev. B | Page 5 of 24 θJA 190 240 158 180 120 θJC 44 43 43 36 36 Unit °C/W °C/W °C/W °C/W °C/W AD8571/AD8572/AD8574 TYPICAL PERFORMANCE CHARACTERISTICS 180 180 VS = 2.7V VCM = 1.35V TA = 25°C 140 120 100 80 60 40 140 120 100 80 60 40 –1.5 –0.5 0.5 1.5 0 –2.5 01104-007 0 –2.5 2.5 OFFSET VOLTAGE (µV) –0.5 0.5 1.5 2.5 OFFSET VOLTAGE (µV) Figure 10. Input Offset Voltage Distribution at 5 V Figure 7. Input Offset Voltage Distribution at 2.7 V 50 12 VS = 5V TA = –40°C, +25°C, +85°C NUMBER OF AMPLIFIERS 30 +85°C 20 10 +25°C 0 –10 –40°C 8 6 4 2 1 2 3 4 INPUT COMMON-MODE VOLTAGE (V) 5 01104-008 –20 0 VS = 5V VCM = 2.5V TA = –40°C TO +125°C 10 0 0 1 2 3 4 5 6 INPUT OFFSET DRIFT (nV/°C) Figure 8. Input Bias Current vs. Common-Mode Voltage 01104-011 40 INPUT BIAS CURRENT (pA) –1.5 01104-010 20 20 –30 VS = 5V VCM = 2.5V TA = 25°C 160 NUMBER OF AMPLIFIERS NUMBER OF AMPLIFIERS 160 Figure 11. Input Offset Voltage Drift Distribution at 5 V 10k 1500 VS = 5V TA = 125°C VS = 5V TA = 25°C 1000 OUTPUT VOLTAGE (mV) 500 0 –500 –1000 100 SOURCE 10 SINK 1 –2000 0 1 2 3 4 COMMON-MODE VOLTAGE (V) 5 0.1 0.0001 0.001 0.01 0.1 1 10 100 LOAD CURRENT (mA) Figure 12. Output Voltage to Supply Rail vs. Output Current at 5 V Figure 9. Input Bias Current vs. Common-Mode Voltage Rev. B | Page 6 of 24 01104-012 –1500 01104-009 INPUT BIAS CURRENT (pA) 1k AD8571/AD8572/AD8574 800 VS = 2.7V TA = 25°C TA = 25°C SUPPLY CURRENT PER AMPLIFIER (µA) 100 SINK SOURCE 10 1 0.001 0.01 0.1 1 10 500 400 300 200 100 0 01104-013 0.1 0.0001 600 100 LOAD CURRENT (mA) 1 2 3 4 5 6 SUPPLY VOLTAGE (V) Figure 13. Output Voltage to Supply Rail vs. Output Current at 2.7 V Figure 16. Supply Current vs. Supply Voltage 1000 60 VCM = 2.5V VS = 5V 50 750 OPEN-LOOP GAIN (dB) INPUT BIAS CURRENT (pA) 0 01104-016 OUTPUT VOLTAGE (mV) 1k 700 500 250 VS = 2.7V CL = 0pF RL = ∞ 40 0 30 45 20 90 10 135 0 180 –10 225 –20 270 PHASE SHIFT (Degrees) 10k –25 0 25 50 75 100 125 150 TEMPERATURE (°C) –40 10k 01104-014 –50 1M 10M 100M FREQUENCY (Hz) Figure 17. Open-Loop Gain and Phase Shift vs. Frequency at 2.7 V Figure 14. Bias Current vs. Temperature 60 50 5V OPEN-LOOP GAIN (dB) 0.8 2.7V 0.6 0.4 0.2 VS = 5V CL = 0pF RL = ∞ 40 0 30 45 20 90 10 135 0 180 –10 225 –20 270 PHASE SHIFT (Degrees) 1.0 0 –75 –50 –25 0 25 50 75 100 TEMPERATURE (°C) 125 150 Figure 15. Supply Current vs. Temperature –40 10k 100k 1M 10M 100M FREQUENCY (Hz) Figure 18. Open-Loop Gain and Phase Shift vs. Frequency at 5 V Rev. B | Page 7 of 24 01104-018 –30 01104-015 SUPPLY CURRENT (mA) 100k 01104-017 –30 0 –75 AD8571/AD8572/AD8574 60 300 VS = 2.7V CL = 0pF RL = 2kΩ 240 20 OUTPUT IMPEDANCE (Ω) AV = –100 30 AV = –10 10 0 AV = +1 –10 210 180 150 90 60 –30 30 1k 10k 100k 1M 10M FREQUENCY (Hz) Figure 19. Closed-Loop Gain vs. Frequency at 2.7 V AV = 100 120 –20 –40 100 VS = 5V 270 01104-019 CLOSED-LOOP GAIN (dB) 40 AV = 10 AV = 1 0 100 1k 10k 100k 1M 10M FREQUENCY (Hz) Figure 22. Output Impedance vs. Frequency at 5 V 60 40 CLOSED-LOOP GAIN (dB) VS = 2.7V CL = 300pF RL = 2kΩ AV = 1 VS = 5V CL = 0pF RL = 2kΩ 50 AV = –100 30 20 AV = –10 10 0 AV = +1 –10 –30 1k 10k 100k 1M 10M FREQUENCY (Hz) 500mV 01104-020 2µs –40 100 01104-023 –20 Figure 23. Large Signal Transient Response at 2.7 V Figure 20. Closed-Loop Gain vs. Frequency at 5 V 300 VS = 5V CL = 300pF RL = 2kΩ AV = 1 VS = 2.7V 270 210 180 AV = 100 150 120 AV = 10 90 30 AV = 1 0 100 1k 10k 100k 1M FREQUENCY (Hz) 5µs 10M 1V Figure 24. Large Signal Transient Response at 5 V Figure 21. Output Impedance vs. Frequency at 2.7 V Rev. B | Page 8 of 24 01104-024 60 01104-021 OUTPUT IMPEDANCE (Ω) 240 01104-022 50 AD8571/AD8572/AD8574 45 VS = ±1.35V CL = 50pF RL = ∞ AV = 1 50mV +OS 30 25 20 –OS 15 10 5 0 10 100 1k 10k CAPACITANCE (pF) Figure 25. Small Signal Transient Response at 2.7 V Figure 28. Small Signal Overshoot vs. Load Capacitance at 5 V VS = ±2.5V CL = 50pF RL = ∞ AV = 1 0V VS = ±2.5V VIN = –200mV p-p (RET TO GND) CL = 0pF RL = 10kΩ AV = –100 VIN VOUT 0V 20µs 1V 01104-029 50mV 01104-026 5µs BOTTOM SCALE: 1V/DIV TOP SCALE: 200mV/DIV Figure 26. Small Signal Transient Response at 5 V Figure 29. Positive Overvoltage Recovery 50 40 VS = ±1.35V RL = 2kΩ TA = 25°C VIN 0V 35 30 +OS 25 0V –OS 20 VS = ±2.5V VIN = 200mV p-p (RET TO GND) CL = 0pF RL = 10kΩ AV = –100 15 VOUT 5 0 10 20µs 100 1k 10k CAPACITANCE (pF) 1V BOTTOM SCALE: 1V/DIV TOP SCALE: 200mV/DIV Figure 30. Negative Overvoltage Recovery Figure 27. Small Signal Overshoot vs. Load Capacitance at 2.7 V Rev. B | Page 9 of 24 01104-030 10 01104-027 SMALL SIGNAL OVERSHOOT (%) 45 01104-028 5µs VS = ±2.5V RL = 2kΩ TA = 25°C 35 01104-025 SMALL SIGNAL OVERSHOOT (%) 40 AD8571/AD8572/AD8574 140 VS = ±2.5V RL = 2kΩ AV = –100 VIN = 60mV p-p VS = ±1.35V 120 PSRR (dB) 100 80 60 40 0 100 1k 10k 100k 1M 10M FREQUENCY (Hz) Figure 31. No Phase Reversal Figure 34. PSRR vs. Frequency at ±1.35 V 140 140 VS = ±2.5V 120 100 100 60 80 60 40 40 20 20 1k 10k 100k 1M 10M FREQUENCY (Hz) 0 100 –PSRR 1k 10k 100k 1M 10M 01104-035 80 +PSRR 1M 01104-036 PSRR (dB) 120 01104-032 CMRR (dB) VS = 2.7V 0 100 01104-034 1V +PSRR 20 01104-031 200µs –PSRR FREQUENCY (Hz) Figure 32. CMRR vs. Frequency at 2.7 V Figure 35. PSRR vs. Frequency at ±2.5 V 140 3.0 VS = 5V 120 2.5 OUTPUT SWING (V p-p) 80 60 40 0 100 2.0 VS = ±1.35V RL = 2kΩ AV = 1 THD + N < 1% TA = 25°C 1.5 1.0 0.5 20 1k 10k 100k 1M FREQUENCY (Hz) 10M 01104-033 CMRR (dB) 100 0 100 1k 10k 100k FREQUENCY (Hz) Figure 36. Maximum Output Swing vs. Frequency at 2.7 V Figure 33. CMRR vs. Frequency at 5 V Rev. B | Page 10 of 24 AD8571/AD8572/AD8574 5.5 VS = ±2.5V RL = 2kΩ AV = 1 THD + N < 1% TA = 25°C 5.0 4.0 312 en (nV/ Hz) 3.5 3.0 2.5 2.0 1.5 260 208 156 104 1.0 1k 10k 100k 1M FREQUENCY (Hz) 0 01104-037 0 100 0.5 1.0 1.5 2.0 2.5 FREQUENCY (kHz) Figure 37. Maximum Output Swing vs. Frequency at 5 V 01104-040 52 0.5 Figure 40. Voltage Noise Density at 2.7 V from 0 Hz to 2.5 kHz VS = ±1.35V AV = 120,000 VS = 2.7V RS = 0Ω 112 en (nV/ Hz) 96 0V 80 64 48 32 50mV 0 5 10 15 20 25 FREQUENCY (kHz) Figure 38. 0.1 Hz to 10 Hz Noise at 2.7 V 01104-041 16 01104-038 1s Figure 41. Voltage Noise Density at 2.7 V from 0 Hz to 25 kHz VS = ±2.5V AV = 120,000 VS = 5V RS = 0Ω 182 en (nV/ Hz) 156 130 104 78 52 50mV 26 0 0.5 1.0 1.5 2.0 2.5 FREQUENCY (kHz) Figure 42. Voltage Noise Density at 5 V from 0 Hz to 2.5 kHz Figure 39. 0.1 Hz to 10 Hz Noise at 5 V Rev. B | Page 11 of 24 01104-042 1s 01104-039 OUTPUT SWING (V p-p) 4.5 VS = 2.7V RS = 0Ω 364 AD8571/AD8572/AD8574 150 VS = 5V RS = 0Ω VS = 2.7V TO 5.5V 80 64 48 32 0 5 10 15 20 25 FREQUENCY (kHz) 135 130 125 –75 01104-043 16 140 –50 –25 0 25 50 75 100 125 150 01104-045 en (nV/ Hz) 96 145 150 01104-046 POWER SUPPLY REJECTION (dB) 112 TEMPERATURE (°C) Figure 43. Voltage Noise Density at 5 V from 0 Hz to 25 kHz Figure 45. Power Supply Rejection vs. Temperature 50 VS = 5V RS = 0Ω 40 SHORT-CIRCUIT CURRENT (mA) 210 150 120 90 60 ISC– 20 10 0 –10 ISC+ –20 –30 30 –40 0 5 FREQUENCY (kHz) 10 01104-044 en (nV/ Hz) 180 VS = 2.7V 30 –50 –75 –50 –25 0 25 50 75 100 125 TEMPERATURE (°C) Figure 46. Output Short-Circuit Current vs. Temperature Figure 44. Voltage Noise Density at 5 V from 0 Hz to 10 Hz Rev. B | Page 12 of 24 AD8571/AD8572/AD8574 100 225 ISC– 40 20 0 –20 –40 ISC+ –60 –80 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 250 175 150 RL = 1kΩ 125 100 75 50 RL = 10kΩ RL = 100kΩ 25 –50 –25 0 25 50 75 100 125 TEMPERATURE (°C) 150 01104-048 OUTPUT VOLTAGE SWING (mV) VS = 5V 200 0 –75 RL = 1kΩ 150 125 100 75 50 RL = 10kΩ RL = 100kΩ 0 –75 –50 –25 0 25 50 75 100 125 TEMPERATURE (°C) Figure 49. Output Voltage to Supply Rail vs. Temperature Figure 47. Output Short-Circuit Current vs. Temperature 225 175 25 01104-047 –100 –75 VS = 5V 200 Figure 48. Output Voltage to Supply Rail vs. Temperature Rev. B | Page 13 of 24 150 01104-049 VS = 5V 60 OUTPUT VOLTAGE SWING (mV) SHORT-CIRCUIT CURRENT (mA) 80 250 AD8571/AD8572/AD8574 FUNCTIONAL DESCRIPTION The AD8571/AD8572/AD8574 are CMOS amplifiers that achieve their high degree of precision through random frequency auto-zero stabilization. The autocorrection topology allows the AD857x to maintain its low offset voltage over a wide temperature range, and the randomized auto-zero clock eliminates any intermodulation distortion (IMD) errors at the amplifier output. The AD857x can be run from a single-supply voltage as low as 2.7 V. The extremely low offset voltage of 1 μV and no IMD products allows the amplifier to be easily configured for high gains without risk of excessive output voltage errors. This makes the AD857x an ideal amplifier for applications requiring both dc precision and low distortion for ac signals. The extremely small temperature drift of 5 nV/°C ensures a minimum of offset voltage error over its entire temperature range of −40°C to +125°C. These combined features make the AD857x an excellent choice for a variety of sensitive measurement and automotive applications. AMPLIFIER ARCHITECTURE Each AD857x op amp consists of two amplifiers: a main amplifier and a secondary amplifier that is used to correct the offset voltage of the main amplifier. Both consist of a rail-to-rail input stage, allowing the input common-mode voltage range to reach both supply rails. The input stage consists of an NMOS differential pair operating concurrently with a parallel PMOS differential pair. The outputs from the differential input stages are combined in another gain stage whose output is used to drive a rail-to-rail output stage. The wide voltage swing of the amplifier is achieved by using two output transistors in a common-source configuration. The output voltage range is limited by the drain-to-source resistance of these transistors. As the amplifier is required to source or sink more output current, the voltage drop across these transistors increases due to their on resistance (rds). Simply put, the output voltage does not swing as close to the rail under heavy output current conditions as it does with light output current. This is a characteristic of all rail-to-rail output amplifiers. Figure 12 and Figure 13 show how close the output voltage can get to the rails with a given output current. The output of the AD857x is shortcircuit protected to approximately 50 mA of current. The AD857x amplifiers have exceptional gain, yielding greater than 120 dB of open-loop gain with a load of 2 kΩ. Because the output transistors are configured in a common-source configuration, the gain of the output stage, and thus the open-loop gain of the amplifier, is dependent on the load resistance. Open-loop gain decreases with smaller load resistances. This is another characteristic of rail-to-rail output amplifiers. BASIC AUTO-ZERO AMPLIFIER THEORY Autocorrection amplifiers are not a new technology. Various IC implementations have been available for more than 15 years and some improvements have been made over time. The AD857x design offers a number of significant performance improvements over older versions while attaining a very substantial reduction in device cost. This section offers a simplified explanation of how the AD857x is able to offer extremely low offset voltages and high open-loop gains. As noted in the Amplifier Architecture section, each AD857x op amp contains two internal amplifiers. One is used as the primary amplifier, the other as an autocorrection, or nulling, amplifier. Each amplifier has an associated input offset voltage that can be modeled as a dc voltage source in series with the noninverting input. In Figure 50 and Figure 51, these are labeled as VOSX, where X denotes the amplifier associated with the offset: A for the nulling amplifier, B for the primary amplifier. The open-loop gain for the +IN and −IN inputs of each amplifier is given as AX. Both amplifiers also have a third voltage input with an associated open-loop gain of BX. There are two modes of operation determined by the action of two sets of switches in the amplifier: an auto-zero phase and an amplification phase. AUTO-ZERO PHASE In this phase, all φA switches are closed and all φB switches are opened. Here, the nulling amplifier is taken out of the gain loop by shorting its two inputs together. Of course, there is a degree of offset voltage, shown as VOSA, inherent in the nulling amplifier that maintains a potential difference between the +IN and −IN inputs. The nulling amplifier feedback loop is closed through φA2 and VOSA appears at the output of the nulling amp and on CM1, an internal capacitor in the AD857x. Mathematically, we can express this in the time domain as VOA [t ] = A A VOSA [t ] − B A VOA [t ] (1) this also can be expressed as VOA [t ] = A AVOSA [t ] 1 + BA (2) This shows that the offset voltage of the nulling amplifier times a gain factor appears at the output of the nulling amplifier and thus on the CM1 capacitor. Rev. B | Page 14 of 24 AD8571/AD8572/AD8574 VOSB + VIN+ AB VIN– VOUT BB ΦB VOA VOSA + ΦA long-term wear time, both of which are much slower than the auto-zero clock frequency of the AD857x. This effectively makes the VOS time invariant, and Equation 5 can be rewritten as ΦB AA CM2 VOA [t ] = AAVIN [t ] + ⎛ VOSA ⎞⎟ VOA [t ] = A A ⎜ VIN [t ] + ⎜ 1 + B A ⎟⎠ ⎝ Figure 50. Auto-Zero Phase of the Amplifier AMPLIFICATION PHASE When the φB switches close and the φA switches open for the amplification phase, this offset voltage remains on CM1 and essentially corrects any error from the nulling amplifier. The voltage across CM1 is designated as VNA. The potential difference between the two inputs to the primary amplifier is designated as VIN, or VIN = (VIN+ − VIN–). The output of the nulling amplifier can then be expressed as VOA [t ] = A A (V IN [t ] − VOSA [t ] − B A V NA [t ] (3) VOSB + AB VIN– VOUT BB VOA VOSA + ΦB AA CM2 ΦA Here, the auto-zeroing becomes apparent. Note that the VOS term is reduced by a 1 + BA factor. This shows how the nulling amplifier has greatly reduced its own offset voltage error even before correcting the primary amplifier. Thus, the primary amplifier output voltage is the voltage at the output of the AD857x amplifier. It is equal to VOUT [t ] = A B (V IN [t ] + VOSB ) + B B V NB ⎡ ⎛ ⎞⎤ V VOUT [t ] = ABVIN [t ] + ABVOSB + B B ⎢ A A ⎜ VIN [t ] + OSA ⎟⎥ 1 + B A ⎟⎠⎥⎦ ⎢⎣ ⎜⎝ (9) CM1 VNA Figure 51. Output Phase of the Amplifier Because φA is now open and there is no place for CM1 to discharge, the voltage (VNA) at the present time (t) is equal to the voltage at the output of the nulling amp (VOA) at the time when φA was closed. If the period of the autocorrection switching frequency is designated as TS, then the amplifier switches between phases every 0.5 × TS. Therefore, in the amplification phase 1 ⎤ ⎡ VNA [t ] = VNA ⎢t − TS ⎥ 2 ⎦ ⎣ (8) In the amplification phase, VOA = VNB, so this can be rewritten as VOUT [t ] = V IN [t ](A B + A A B B ) + VNB –BA (7) combining terms yields 01104-051 ΦB (6) or CM1 01104-050 ΦA VNA ΦA 1 + BA VNB –BA VIN+ AA (1 + BA )VOSA − AA BAVOSA A A B BVOSA 1 + BA (10) The AD857x architecture is optimized in such a way that AA = AB and BA = BB and BA >> 1. In addition, the gain product to AABB is much greater than AB. Thus, Equation 10 can be simplified to VOUT [t ] = V IN [t ]A A B A + A A (VOSA + VOSB ) (4) 1 ⎤ ⎡ A A B AVOSA ⎢t − TS ⎥ 2 ⎦ (5) ⎣ VOA [t ] = A AVIN [t ] + A AVOSA [t ] − 1 + BA (11) Most obvious is the gain product of both the primary and nulling amplifiers. This AABA term is what gives the AD857x its extremely high open-loop gain. To understand how VOSA and VOSB relate to the overall effective input offset voltage of the complete amplifier, set up the generic amplifier equation of VOUT = k × (VIN + VOS , EFF ) and substituting Equation 4 and Equation 2 into Equation 3 yields + A BVOSB (12) where k is the open-loop gain of an amplifier and VOS, EFF is its effective offset voltage. Putting Equation 12 into the form of Equation 11 gives VOUT [t ] = V IN [t ]A A B A + VOS, EFF A A B A (13) Therefore For the sake of simplification, it can be assumed that the autocorrection frequency is much faster than any potential change in VOSA or VOSB. This is a good assumption since changes in offset voltage are a function of temperature variation or Rev. B | Page 15 of 24 VOS , EFF ≈ VOSA + VOSB BA (14) AD8571/AD8572/AD8574 V+ Thus, the offset voltages of both the primary and nulling amplifiers are reduced by the Gain Factor BA. This takes a typical input offset voltage from several millivolts down to an effective input offset voltage of submicrovolts. This autocorrection scheme makes the AD857x family of amplifiers extremely precise. R1 R2 AD8572 GUARD RING GUARD RING VREF To achieve the maximum performance of the extremely high input impedance and low offset voltage of the AD857x, care should be taken in the circuit board layout. The PC board surface must remain clean and free of moisture to avoid leakage currents between adjacent traces. Surface coating of the circuit board reduces surface moisture and provides a humidity barrier, reducing parasitic resistance on the board. The use of guard rings around the amplifier inputs further reduces leakage currents. Figure 52 shows how the guard ring should be configured and Figure 53 shows the top view of how a surface mount layout can be arranged. The guard ring does not need to be a specific width, but it should form a continuous loop around both inputs. By setting the guard ring voltage equal to the voltage at the noninverting input, parasitic capacitance is minimized as well. For further reduction of leakage currents, components can be mounted to the PC board using Teflon® standoff insulators. V Figure 53. Top View of AD8572 SOIC Layout with Guard Rings Other potential sources of offset error are thermoelectric voltages on the circuit board. This voltage, also called Seebeck voltage, occurs at the junction of two dissimilar metals and is proportional to the temperature of the junction. The most common metallic junctions on a circuit board are solder-toboard trace and solder-to-component lead. Figure 54 shows a cross-section view of the thermal voltage error sources. When the temperature of the PC board at one end of the component (TA1) differs from the temperature at the other end (TA2), the Seebeck voltages are not equal, resulting in a thermal voltage error. This thermocouple error can be reduced by using dummy components to match the thermoelectric error source. Placing the dummy component as close as possible to its partner ensures both Seebeck voltages are equal, thus canceling the thermocouple error. Maintaining a constant ambient temperature on the circuit board further reduces this error. The use of a ground plane helps distribute heat throughout the board and also reduces EMI noise pickup. COMPONENT LEAD VSC1 VTS1 – – + + SURFACE MOUNT COMPONENT SOLDER VSC2 – + + VTS2 – PC BOARD TA1 TA2 COPPER TRACE IF TA1 ≠ TA2, THEN VTS1 + VSC1 ≠ VTS2 + VSC2 Figure 54. Mismatch in Seebeck Voltages Causes a Thermoelectric Voltage Error VOUT IN RF AD8572 R1 RS = R1 01104-052 AD8572 VOUT VIN VOUT Figure 52. Guard Ring Layout and Connections to Reduce PC Board Leakage Currents Rev. B | Page 16 of 24 AD8571/AD8572/ AD8574 AV = 1 + (RF /R1) Figure 55. Using Dummy Components to Cancel Thermoelectric Voltage Errors 01104-055 VIN 01104-054 MAXIMIZING PERFORMANCE THROUGH PROPER LAYOUT VOUT 01104-053 VREF V– Common-mode and power supply rejection are indications of the amount of offset voltage an amplifier has as a result of a change in its input common-mode or power supply voltages. As shown in the Amplification Phase section, the autocorrection architecture of the AD857x allows it to effectively minimize offset voltages. The technique also corrects for offset errors caused by common-mode voltage swings and power supply variations. This results in superb CMRR and PSRR figures in excess of 130 dB. Because the autocorrection occurs continuously, these figures can be maintained across the entire temperature range of the device, from −40°C to +125°C. AD8572 R1 VIN2 HIGH GAIN, CMRR, PSRR VIN R2 VIN1 AD8571/AD8572/AD8574 0 1/f NOISE CHARACTERISTICS 6 7 8 9 OUTPUT SIGNAL 01104-057 6 7 8 9 10 VS = 5V AV = 60dB –60 –80 –100 –120 0 1 2 3 4 5 6 7 8 9 10 FREQUENCY (kHz) Figure 58. Spectral Analysis of AD857x in High Gain with an Input Signal 01104-056 5 5 –40 –120 4 4 –20 –100 3 3 0 –80 2 2 Figure 58 shows the spectral output of an AD8572 configured in a high gain (60 dB) with a 1 mV input signal applied. Note the absence of any IMD products in the spectrum. The signal-tonoise (SNR) ratio of the output signal is better than 60 dB, or 0.1%. –60 1 1 Figure 57. Spectral Analysis of AD857x Output with 60 dB Gain –40 –160 0 FREQUENCY (kHz) VS = 5V AV = 0dB –140 –80 –120 0 –20 –60 –100 RANDOM AUTO-ZERO CORRECTION ELIMINATES INTERMODULATION DISTORTION The AD857x can be used as a conventional op amp for gains up to 1 MHz. The auto-zero correction frequency of the device continuously varies, based on a pseudorandom generator with a uniform distribution from 2 kHz to 4 kHz. The randomization of the autocorrection clock creates a continuous randomization of intermodulation distortion (IMD) products that show up as simple broadband noise at the output of the amplifier. This noise naturally combines with the amplifier voltage noise in a root-squared-sum fashion, resulting in an output free of IMD. Figure 56 shows the spectral output of an AD8572 with the amplifier configured for unity gain and the input grounded. Figure 57 shows the spectral output with the amplifier configured for a gain of 60 dB. –40 01104-058 Because the AD857x amplifiers are self-correcting op amps, they do not have increasing flicker noise at lower frequencies. In essence, low frequency noise is treated as a slowly varying offset error and is greatly reduced as a result of autocorrection. The correction becomes more effective as the noise frequency approaches dc, offsetting the tendency of the noise to increase exponentially as frequency decreases. This allows the AD857x to have lower noise near dc than standard low noise amplifiers that are susceptible to 1/f noise. VS = 5V AV = 60dB –20 OUTPUT SIGNAL Another advantage of auto-zero amplifiers is their ability to cancel flicker noise. Flicker noise, also known as 1/f noise, is noise inherent in the physics of semiconductor devices and increases 3 dB for every octave decrease in frequency. The 1/f corner frequency of an amplifier is the frequency at which the flicker noise is equal to the broadband noise of the amplifier. At lower frequencies, flicker noise dominates, causing higher degrees of error for sub-Hertz frequencies or dc precision applications. 10 FREQUENCY (kHz) Figure 56. Spectral Analysis of AD857x Output in Unity Gain Configuration Rev. B | Page 17 of 24 AD8571/AD8572/AD8574 BROADBAND AND EXTERNAL RESISTOR NOISE CONSIDERATIONS The total broadband noise output from any amplifier is primarily a function of three types of noise: input voltage noise from the amplifier, input current noise from the amplifier, and Johnson noise from the external resistors used around the amplifier. Input voltage noise, or en, is strictly a function of the amplifier used. The Johnson noise from a resistor is a function of the resistance and the temperature. Input current noise, or in, creates an equivalent voltage noise proportional to the resistors used around the amplifier. These noise sources are not correlated with each other and their combined noise sums in a rootsquared-sum fashion. The full equation is given as e n,TOTAL = [e n 2 + 4kTrs + (i n rs ) 2 ]1 / 2 (15) where: en = input voltage noise of the amplifier. in = input current noise of the amplifier. rs = source resistance connected to the noninverting terminal. k = Boltzmann’s constant (1.38 × 10−23 J/K). T = ambient temperature in Kelvin (K = 273.15 + °C). The input voltage noise density, en, of the AD857x is 51 nV/√Hz, and the input noise, in, is 2 fA/√Hz. The en, TOTAL is dominated by input voltage noise provided the source resistance is less than 172 kΩ. With source resistance greater than 172 kΩ, the overall noise of the system is dominated by the Johnson noise of the resistor itself. Because the input current noise of the AD857x is very small, in does not become a dominant term unless rS is greater than 4 GΩ, which is an impractical value of source resistance. The total noise, en, TOTAL, is expressed in volts-per-square-root Hertz, and the equivalent rms noise over a certain bandwidth can be found as e n = e n,TOTAL × BW (16) where BW is the bandwidth of interest in Hertz. OUTPUT OVERDRIVE RECOVERY The AD857x amplifiers have an excellent overdrive recovery of only 200 μs from either supply rail. This characteristic is particularly difficult for autocorrection amplifiers, because the nulling amplifier requires a substantial amount of time to error correct the main amplifier back to a valid output. Figure 29 and Figure 30 show the positive and negative overdrive recovery time for the AD857x. The output overdrive recovery for an autocorrection amplifier is defined as the time it takes for the output to correct to its final voltage from an overload state. It is measured by placing the amplifier in a high gain configuration with an input signal that forces the output voltage to the supply rail. The input voltage is then stepped down to the linear region of the amplifier, usually to halfway between the supplies. The time from the input signal step-down to the output settling to within 100 μV of its final value is the overdrive recovery time. Many competitors’ autocorrection amplifiers require a number of auto-zero clock cycles to recover from output overdrive and some can take several milliseconds for the output to settle properly. INPUT OVERVOLTAGE PROTECTION Although the AD857x is a rail-to-rail input amplifier, care should be taken to ensure that the potential difference between the inputs does not exceed 5 V. Under normal operating conditions, the amplifier corrects its output to ensure the two inputs are at the same voltage. However, if the device is configured as a comparator, or is under some unusual operating condition, the input voltages may be forced to different potentials. This could cause excessive current to flow through internal diodes in the AD857x used to protect the input stage against overvoltage. If either input exceeds either supply rail by more than 0.3 V, large amounts of current begin to flow through the ESD protection diodes in the amplifier. These diodes are connected between the inputs and each supply rail to protect the input transistors against an electrostatic discharge event and are normally reverse-biased. However, if the input voltage exceeds the supply voltage, these ESD diodes become forward-biased. Without current-limiting, excessive amounts of current can flow through these diodes causing permanent damage to the device. If inputs are subject to overvoltage, appropriate series resistors should be inserted to limit the diode current to less than 2 mA. OUTPUT PHASE REVERSAL Output phase reversal occurs in some amplifiers when the input common-mode voltage range is exceeded. As common-mode voltage is moved outside of the common-mode range, the outputs of these amplifiers suddenly jump in the opposite direction to the supply rail. This is the result of the differential input pair shutting down, causing a radical shifting of internal voltages that results in the erratic output behavior. The AD857x amplifier has been carefully designed to prevent any output phase reversal, provided both inputs are maintained within the supply voltages. If one or both inputs could exceed either supply voltage, a resistor should be placed in series with the input to limit the current to less than 2 mA to ensure the output does not reverse its phase. Rev. B | Page 18 of 24 AD8571/AD8572/AD8574 CAPACITIVE LOAD DRIVE Table 5. Snubber Network Values for Driving Capacitive Loads The AD857x has excellent capacitive load driving capabilities and can safely drive up to 10 nF from a single 5 V supply. Although the device is stable, capacitive loading limits the bandwidth of the amplifier. Capacitive loads also increase the amount of overshoot and ringing at the output. An RC snubber network, shown in Figure 59, can be used to compensate the amplifier against capacitive load ringing and overshoot. AD8571/ AD8572/ AD8574 – VIN + 200mV p-p Rx 60Ω Cx 0.47µF VOUT CL 4.7nF 01104-059 5V CLOAD 1 nF 4.7 nF 10 nF Rx 200 Ω 60 Ω 20 Ω Cx 1 nF 0.47 μF 10 μF POWER-UP BEHAVIOR On power-up, the AD857x settles to a valid output within 5 μs. Figure 61 shows an oscilloscope photo of the output of the amplifier along with the power supply voltage, and Figure 62 shows the test circuit. With the amplifier configured for unity gain, the device takes approximately 5 μs to settle to its final output voltage, hundreds of microseconds faster than many other autocorrection amplifiers. Figure 59. Snubber Network Configuration for Driving Capacitive Loads Although the snubber does not recover the loss of amplifier bandwidth from the load capacitance, it does allow the amplifier to drive larger values of capacitance while maintaining a minimum of overshoot and ringing. Figure 60 shows the output of an AD857x driving a 1 nF capacitor with and without a snubber network. VOUT 0V V+ 10μs 0V WITH SNUBBER 5µs 01104-061 1V BOTTOM TRACE = 2V/DIV TOP TRACE = 1V/DIV Figure 61. AD857x Output Behavior on Power-Up WITHOUT SNUBBER 100mV 100kΩ Figure 60. Overshoot and Ringing are Substantially Reduced Using a Snubber Network The optimum value for the resistor and capacitor is a function of the load capacitance and is best determined empirically since actual CLOAD includes stray capacitances and can differ substantially from the nominal capacitive load. Table 5 shows some snubber network values that can be used as starting points. Rev. B | Page 19 of 24 VSY = 0V TO 5V AD8571/ AD8572/ AD8574 VOUT Figure 62. AD857x Test Circuit for Turn-On Time 01104-062 VS = 5V CLOAD = 4.7nF 01104-060 100kΩ AD8571/AD8572/AD8574 APPLICATIONS 5 V PRECISION STRAIN GAGE CIRCUIT The extremely low offset voltage of the AD8572 makes it an ideal amplifier for any application requiring accuracy with high gains, such as a weigh scale or strain gage. Figure 63 shows a configuration for a single supply, precision strain gage measurement system. In an ideal difference amplifier, the ratio of the resistors is set exactly equal to AV = R2 R1 = R4 (19) R3 setting the output voltage of the system to 2 × (R1 + R 2 ) (17) RB where RB is the resistance of the load cell. Using the values given in Figure 63, the output voltage linearly varies from 0 V with no strain to 4 V under full strain. 2 5V 1kΩ Q1 2N2222 OR EQUIVALENT 6 2.5V 3 A2 4 AD8572-B 12kΩ 4.0V REF192 VOUT = AV (V 1 − V 2) Due to finite component tolerance, the ratio between the four resistors is not exactly equal, and any mismatch results in a reduction of common-mode rejection from the system. Referring to Figure 64, the exact common-mode rejection ratio can be expressed as CMRR = R1R4 + 2R2R4 + R2R3 In the three-op amp instrumentation amplifier configuration shown in Figure 65, the output difference amplifier is set to unity gain with all four resistors equal in value. If the tolerance of the resistors used in the circuit is given as δ, the worst-case CMRR of the instrumentation amplifier is CMRRMIN = 40mV FULL-SCALE R2 100Ω A1 AD8572-A R3 17.4kΩ NOTE: USE 0.1% TOLERANCE RESISTORS. VOUT 0V TO 4V R4 100Ω 1 (22) 2δ AD8574-A V2 01104-063 350Ω LOAD CELL RG R R R R R VOUT R Figure 63. 5 V Precision Strain Gage Amplifier 3 V INSTRUMENTATION AMPLIFIER V1 The high common-mode rejection, high open-loop gain, and operation down to 3 V of supply voltage make the AD857x an excellent choice of op amp for discrete single-supply instrumentation amplifiers. The common-mode rejection ratio of the AD857x is greater than 120 dB, but the CMRR of the system is also a function of the external resistor tolerances. The gain of the difference amplifier shown in Figure 64 is given as ⎛ R4 ⎞ ⎛ ⎛ R2 ⎞ R1 ⎞ VOUT = V 1 ⎜ ⎟ ⎜1 + ⎟ − V 2⎜ ⎟ R 3 R 4 R 2 + ⎠ ⎝ R1 ⎠ ⎝ ⎠⎝ (18) R2 (21) 2R1R4 − 2R2R3 20kΩ R1 17.4kΩ (20) AD8574-B VOUT = 1 + AD8574-C RTRIM 2R (V1 – V2) RG 01104-065 A REF192 provides a 2.5 V precision reference voltage for A2. The A2 amplifier boosts this voltage to provide a 4.0 V reference for the top of the strain gage resistor bridge. Q1 provides the current drive for the 350 Ω bridge network. A1 is used to amplify the output of the bridge with the full-scale output voltage equal to Figure 65. Discrete Instrumentation Amplifier Configuration Thus, using 1% tolerance resistors results in a worst-case system CMRR of 0.02, or 34 dB. Therefore, either high precision resistors or an additional trimming resistor, as shown in Figure 65, should be used to achieve high common-mode rejection. The value of this trimming resistor should be equal to the value of R multiplied by its tolerance. For example, using 10 kΩ resistors with 1% tolerance would require a series trimming resistor equal to 100 Ω. R1 R3 V1 R4 IF AD8571/ AD8572/ AD8574 R2 R2 R4 = , THEN VOUT = R1 R1 R3 (V1 – V2) HIGH ACCURACY THERMOCOUPLE AMPLIFIER VOUT 01104-064 V2 Figure 64. Using the AD857x as a Difference Amplifier Figure 66 shows a K-type thermocouple amplifier configuration with cold junction compensation. Even from a 5 V supply, the AD8571 can provide enough accuracy to achieve a resolution of better than 0.02°C from 0°C to 500°C. D1 is used as a temperature measuring device to correct the cold-junction error from Rev. B | Page 20 of 24 AD8571/AD8572/AD8574 the thermocouple and should be placed as close as possible to the two terminating junctions. With the thermocouple measuring tip immersed in a 0°C ice bath, R6 should be adjusted until the output is at 0 V. ⎛ R2 ⎞ VOUT = V + − ⎜ × R SENSE × I L ⎟ ⎝ R1 ⎠ Using the values shown in Figure 66, the output voltage tracks temperature at 10 mV/°C. For a wider range of temperature measurement, R9 can be decreased to 62 kΩ. This creates a 5 mV/°C change at the output, allowing measurements of up to 1000°C. REF02EZ 2 12V 6 0.1µF RSENSE 0.1Ω 3V IL V+ 3V 4 R5 40.2kΩ R9 124kΩ 5V + R2 2.74kΩ R8 453Ω 8 1/2 AD8572 8 1 R6 200Ω R4 5.62kΩ 3 R3 53.6kΩ 4 1 4 S M1 Si9433 0.1µF 2 0.1µF D MONITOR OUTPUT AD8572 0V TO 5V (0°C TO 500°C) G R2 2.49kΩ 01104-067 + 2 01104-066 – 3 10µF D1 – R1 100Ω + 1N4148 (24) For the component values shown in Figure 68, the output transfer function decreases from V at –2.5 V/A. 5V R1 10.7kΩ K-TYPE THERMOCOUPLE 40.7µV/°C Figure 68 shows the low-side monitor equivalent. In this circuit, the input common-mode voltage to the AD8572 is at or near ground. Again, a 0.1 Ω resistor provides a voltage drop proportional to the return current. The output voltage is given as Figure 67. High-Side Load Current Monitor Figure 66. Precision K-Type Thermocouple Amplifier with Cold-Junction Compensation V+ PRECISION CURRENT METER R2 2.49kΩ Because of its low input bias current and superb offset voltage at single-supply voltages, the AD857x is an excellent amplifier for precision current monitoring. Its rail-to-rail input allows the amplifier to be used as either a high-side or a low-side current monitor. Using both amplifiers in the AD8572 provides a simple method to monitor both current supply and return paths for load or fault detection. The 0.1 Ω resistor creates a voltage drop to the noninverting input of the AD857x. The output of the amplifier is corrected until this voltage appears at the inverting input. This creates a current through R1 that in turn flows through R2. The monitor output is given by ⎛R ⎞ Monitor Output = R 2 × ⎜ SENSE ⎟ × I L ⎝ R1 ⎠ (23) Q1 V+ R1 100Ω 1/2 AD8572 RSENSE 0.1Ω RETURN TO GROUND 01104-068 Figure 67 shows a high-side current monitor configuration. Here, the input common-mode voltage of the amplifier is at or near the positive supply voltage. The rail-to-rail input of the amplifier provides a precise measurement, even with the input common-mode voltage at the supply voltage. The CMOS input structure does not draw any input bias current, ensuring a minimum of measurement error. VOUT Figure 68. Low-Side Load Current Monitor PRECISION VOLTAGE COMPARATOR The AD857x can be operated open-loop and used as a precision comparator. The AD857x has less than 50 μV of offset voltage when run in this configuration. The slight increase of offset voltage stems from the fact that the autocorrection architecture operates with lowest offset in a closed-loop configuration, that is, one with negative feedback. With 50 mV of overdrive, the device has a propagation delay of 15 μs on the rising edge and 8 μs on the falling edge. Care should be taken to ensure the maximum differential voltage of the device is not exceeded. For more information, refer to the Input Overvoltage Protection section. Using the components shown in Figure 67, the monitor output transfer function is 2.5 V/A. Rev. B | Page 21 of 24 AD8571/AD8572/AD8574 OUTLINE DIMENSIONS 3.20 3.00 2.80 8 3.20 3.00 2.80 3.10 3.00 2.90 1 8 5.15 4.90 4.65 5 4.50 4.40 4.30 4 1 PIN 1 0.65 BSC 4 0.65 BSC 1.10 MAX 0.38 0.22 COPLANARITY 0.10 6.40 BSC PIN 1 0.95 0.85 0.75 0.15 0.00 5 0.80 0.60 0.40 8° 0° 0.23 0.08 0.15 0.05 1.20 MAX COPLANARITY 0.10 SEATING PLANE 0.30 0.19 SEATING 0.20 PLANE 0.09 8° 0° 0.75 0.60 0.45 COMPLIANT TO JEDEC STANDARDS MO-153-AA COMPLIANT TO JEDEC STANDARDS MO-187-AA Figure 71. 8-Lead Thin Shrink Small Outline Package [TSSOP] (RU-8) Dimensions shown in millimeters Figure 69. 8-Lead Mini Small Outline Package [MSOP] (RM-8) Dimensions shown in millimeters 5.00 (0.1968) 4.80 (0.1890) 8 1 5 4 1.27 (0.0500) BSC 0.25 (0.0098) 0.10 (0.0040) COPLANARITY 0.10 SEATING PLANE 6.20 (0.2440) 5.80 (0.2284) 1.75 (0.0688) 1.35 (0.0532) 0.51 (0.0201) 0.31 (0.0122) 14 0.50 (0.0196) 0.25 (0.0099) 45° 6.40 BSC 1 8° 0° 0.25 (0.0098) 0.17 (0.0067) 8 4.50 4.40 4.30 7 PIN 1 1.05 1.00 0.80 1.27 (0.0500) 0.40 (0.0157) COMPLIANT TO JEDEC STANDARDS MS-012-A A CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. 0.65 BSC 1.20 MAX 0.15 0.05 060506-A 4.00 (0.1574) 3.80 (0.1497) 5.10 5.00 4.90 0.30 0.19 0.20 0.09 SEATING COPLANARITY PLANE 0.10 8° 0° COMPLIANT TO JEDEC STANDARDS MO-153-AB-1 Figure 72. 14-Lead Thin Shrink Small Outline Package [TSSOP] (RU-14) Dimensions shown in millimeters Figure 70. 8-Lead Standard Small Outline Package [SOIC_N] Narrow Body (R-8) Dimensions shown in millimeters and inches Rev. B | Page 22 of 24 0.75 0.60 0.45 AD8571/AD8572/AD8574 8.75 (0.3445) 8.55 (0.3366) 8 14 1 7 6.20 (0.2441) 5.80 (0.2283) 1.27 (0.0500) BSC 0.25 (0.0098) 0.10 (0.0039) COPLANARITY 0.10 0.50 (0.0197) 0.25 (0.0098) 1.75 (0.0689) 1.35 (0.0531) SEATING PLANE 0.51 (0.0201) 0.31 (0.0122) 45° 8° 0° 0.25 (0.0098) 0.17 (0.0067) 1.27 (0.0500) 0.40 (0.0157) COMPLIANT TO JEDEC STANDARDS MS-012-AB CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. 060606-A 4.00 (0.1575) 3.80 (0.1496) Figure 73. 14-Lead Standard Small Outline Package [SOIC_N] Narrow Body (R-14) Dimensions shown in millimeters and (inches) ORDERING GUIDE Model AD8571AR AD8571AR-REEL AD8571AR-REEL7 AD8571ARZ 1 AD8571ARZ-REEL1 AD8571ARZ-REEL71 AD8571ARM-R2 AD8571ARM-REEL AD8571ARMZ-R21 AD8571ARMZ-REEL1 AD8572AR AD8572AR-REEL AD8572AR-REEL7 AD8572ARZ1 AD8572ARZ-REEL1 AD8572ARZ-REEL71 AD8572ARU AD8572ARU-REEL AD8572ARUZ1 AD8572ARUZ-REEL1 AD8574AR AD8574AR-REEL AD8574AR-REEL7 AD8574ARZ1 AD8574ARZ-REEL1 AD8574ARZ-REEL71 AD8574ARU AD8574ARU-REEL AD8574ARUZ1 AD8574ARUZ-REEL1 1 Temperature Range −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C Package Description 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead MSOP 8-Lead MSOP 8-Lead MSOP 8-Lead MSOP 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead TSSOP 8-Lead TSSOP 8-Lead TSSOP 8-Lead TSSOP 14-Lead SOIC_N 14-Lead SOIC_N 14-Lead SOIC_N 14-Lead SOIC_N 14-Lead SOIC_N 14-Lead SOIC_N 14-Lead TSSOP 14-Lead TSSOP 14-Lead TSSOP 14-Lead TSSOP Z = Pb-free part, # denote lead-free product may be top or bottom marked. Rev. B | Page 23 of 24 Package Option R-8 R-8 R-8 R-8 R-8 R-8 RM-8 RM-8 RM-8 RM-8 R-8 R-8 R-8 R-8 R-8 R-8 RU-8 RU-8 RU-8 RU-8 R-14 R-14 R-14 R-14 R-14 R-14 RU-14 RU-14 RU-14 RU-14 Branding AJA AJA AJA# AJA # AD8571/AD8572/AD8574 NOTES ©2006 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. C01104-0-9/06(B) Rev. B | Page 24 of 24