MPS MP2354 2a, 23v, 380khz step-down converter Datasheet

TM
MP2354
2A, 23V, 380KHz
Step-Down Converter
The Future of Analog IC Technology
TM
DESCRIPTION
FEATURES
The MP2354 is a monolithic step down switch
mode converter with a built in internal power
MOSFET. It achieves 2A continuous output
current over a wide input supply range with
excellent load and line regulation.
•
•
Current mode operation provides fast transient
response and eases loop stabilization.
Fault condition protection includes cycle-by-cycle
current limiting and thermal shutdown. In
shutdown mode the regulator draws 20µA of
supply current.
EVALUATION BOARD REFERENCE
Board Number
Dimensions
EV2354DS-00A
2.3”X x 1.4”Y x 0.5”Z
•
•
•
•
•
•
•
•
•
•
0.18Ω Internal Power MOSFET Switch
Stable with Low ESR Output Ceramic
Capacitors
Up to 95% Efficiency
2A Output Current
Wide 4.75V to 23V Operating Input Range
Fixed 380KHz Frequency
Thermal Shutdown
Cycle-by-Cycle Over Current Protection
Programmable Under Voltage Lockout
Frequency Synchronization Input
Operating Temperature: –40°C to +85°C
Available in an 8-Pin SO Package
APPLICATIONS
•
•
•
Distributed Power Systems
Battery Chargers
Pre-Regulator for Linear Regulators
“MPS” and “The Future of Analog IC Technology” are Trademarks of Monolithic
Power Systems, Inc.
TYPICAL APPLICATION
Efficiency vs
Output Current
INPUT
4.75V to 23V
OPEN
IF NOT USED
8
1
3
2
VIN
RUN
BST
LX
MP2354
SYNC
GND
5
FB
90
4
6
B230A
OUTPUT
3.3V / 2A
COMP
7
3.3nF
EFFICIENCY (%)
OPEN
AUTOMATIC
STARTUP
95
10nF
5.0V
3.3V
85
2.5V
80
75
70
65
60
MP2354_TAC_S01
0
0.5
1.0
1.5
2.0
2.5
OUTPUT CURRENT (A)
MP2354_TAC_EC01
MP2354 Rev. 1.4
1/6/2006
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TM
MP2354 – 2A, 23V, 380KHz STEP-DOWN CONVERTER
ABSOLUTE MAXIMUM RATINGS (1)
PACKAGE REFERENCE
TOP VIEW
SYNC
1
8
RUN
BST
2
7
COMP
VIN
3
6
FB
LX
4
5
GND
Supply Voltage (VIN) .................................... 25V
Switch Voltage (VLX) ....................... –1V to +26V
Bootstrap Voltage (VBST)....................... VLX + 6V
Feedback Voltage (VFB) ................... –0.3 to +6V
Enable/UVLO Voltage (VRUN)........... –0.3 to +6V
Comp Voltage (VCOMP) ..................... –0.3 to +6V
Sync Voltage (VSYNC) ....................... –0.3 to +6V
Junction Temperature...............................150°C
Lead Temperature ....................................260°C
Storage Temperature ..............–65°C to +150°C
Recommended Operating Conditions
(2)
Input Voltage (VIN) ......................... 4.75V to 23V
Operating Temperature .............–40°C to +85°C
MP2354_PD01-SOIC8
Thermal Resistance
(3)
θJA
θJC
SOIC8.................................... 105 ..... 50... °C/W
Part Number*
Package
Temperature
MP2354DS
SOIC8
–40°C to +85°C
*
For Tape & Reel, add suffix –Z (eg. MP2354DS–Z)
For Lead Free, add suffix –LF (eg. MP2354DS–LF–Z)
Notes:
1) Exceeding these ratings may damage the device.
2) The device is not guaranteed to function outside of its
operating conditions.
3) Measured on approximately 1” square of 1 oz copper.
ELECTRICAL CHARACTERISTICS
VIN = 12V, TA = +25°C, unless otherwise noted.
Parameter
Feedback Voltage
Symbol Condition
VFB
Upper Switch On Resistance RDS(ON)1
Lower Switch On Resistance RDS(ON)2
Upper Switch Leakage
Current Limit (4)
Current Sense
Transconductance
GCS
Output Current to Comp Pin
Voltage
Error Amplifier Voltage Gain
AVEA
Error Amplifier
GEA
Transconductance
Oscillator Frequency
fS
Short Circuit Frequency
Sync Frequency
Maximum Duty Cycle
DMAX
Minimum Duty Cycle
DMIN
MP2354 Rev. 1.4
1/6/2006
4.75V ≤ VIN ≤ 23V
Min
Typ
Max
Units
1.198
1.222
1.246
V
2.7
0.18
10
0
3.4
VRUN = 0V, VLX = 0V
∆IC = ±10µA
VFB = 0V
Sync Drive 0V to 2.7V
VFB = 1.0V
VFB = 1.5V
10
Ω
Ω
µA
A
1.95
A/V
400
V/V
500
700
1000
µA/V
342
380
35
418
KHz
KHz
KHz
%
%
445
600
90
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2
TM
MP2354 – 2A, 23V, 380KHz STEP-DOWN CONVERTER
ELECTRICAL CHARACTERISTICS (continued)
VIN = 12V, TA = +25°C, unless otherwise noted.
Parameter
RUN Shutdown Threshold
RUN Pull Up Current
EN UVLO Threshold Rising
EN UVLO Threshold
Hysteresis
Symbol Condition
ICC > 100µA
VRUN = 0V
VEN Rising
Min
0.7
1.0
2.37
Typ
1.0
1.3
2.5
Max
1.3
Supply Current (Shutdown)
VRUN ≤ 0.4V
20
35
µA
Supply Current (Quiescent)
VRUN ≥ 2.8V, VFB = 1.5V
1.0
1.2
mA
2.62
210
Thermal Shutdown
155
Units
V
µA
V
mV
°C
Note:
4) Equivalent output current = 1.5A ≥ 50% Duty Cycle
2.0A ≤ 50% Duty Cycle
Assumes ripple current = 30% of load current.
Slope compensation changes current limit above 40% duty cycle.
PIN FUNCTIONS
Pin #
Name
1
SYNC
2
BST
3
VIN
4
LX
5
GND
6
FB
7
COMP
8
RUN
MP2354 Rev. 1.4
1/6/2006
Description
Synchronization Input. This pin is used to synchronize the internal oscillator frequency to an
external source. There is an internal 11kΩ pull down resistor to GND, therefore leave SYNC
unconnected if unused.
Bootstrap (C5). This capacitor is needed to drive the power switch’s gate above the supply
voltage. It is connected between LX and BST pins to form a floating supply across the power
switch driver. The voltage across C5 is about 5V and is supplied by the internal +5V supply
when the LX pin voltage is low.
Supply Voltage. The MP2354 operates from a +4.75V to +23V unregulated input. C1 is needed
to prevent large voltage spikes from appearing at the input.
Switch. This connects the inductor to either VIN through M1 or to GND through M2.
Ground. This pin is the voltage reference for the regulated output voltage. For this reason care
must be taken in its layout. This node should be placed outside of the D1 to C1 ground path to
prevent switching current spikes from inducing voltage noise into the part.
Feedback. An external resistor divider from the output to GND, tapped to the FB pin sets the
output voltage. To prevent current limit run away during a short circuit fault condition the
frequency foldback comparator lowers the oscillator frequency when the FB voltage is below
700mV.
Compensation. This node is the output of the transconductance error amplifier and the input to
the current comparator. Frequency compensation is done at this node by connecting a series
R-C to ground. See the Compensation section for exact details.
Enable/UVLO. A voltage greater than 2.62V enables operation. Leave RUN unconnected for
automatic startup. An Under Voltage Lockout (UVLO) function can be implemented by the
addition of a resistor divider from VIN to GND. For complete low current shutdown it’s the RUN
pin voltage needs to be less than 700mV.
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TM
MP2354 – 2A, 23V, 380KHz STEP-DOWN CONVERTER
TYPICAL PERFORMANCE CHARACTERISTICS
Circuit of Figure 2, VIN = 12V, VO = 3.3V, L1 = 15µH, C1 = 10µF, C2 = 22µF, TA = +25°C, unless
otherwise noted.
Heavy Load Operation
Light Load Operation
2A Load
No Load
VO, AC
50mV/div.
VO, AC
20mV/div.
VIN, AC
200mV/div.
VIN, AC
20mV/div.
IL
1A/div.
IL
1A/div.
VLX
10V/div.
VLX
10V/div.
MP2354-TPC01
MP2354-TPC02
Startup from Shutdown
Load Transient
2A Resistive Load
VRUN
2V/div.
VO, AC
200mV/div.
VOUT
1V/div.
IL
1A/div.
ILOAD
1A/div.
IL
1A/div.
MP2354-TPC03
MP2354-TPC04
Short Circuit Protection
Short Circuit Recovery
VOUT
2V/div.
VOUT
2V/div.
IL
1A/div.
IL
1A/div.
MP2354-TPC05
MP2354 Rev. 1.4
1/6/2006
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MP2354-TPC06
4
TM
MP2354 – 2A, 23V, 380KHz STEP-DOWN CONVERTER
OPERATION
MP2354 reverts to its initial M1 off, M2 on state.
If the Current Sense Amplifier plus Slope
Compensation signal does not exceed the
COMP voltage, then the falling edge of the CLK
resets the Flip-Flop.
The MP2354 is a current mode regulator. The
COMP pin voltage is proportional to the peak
inductor current. At the beginning of a cycle: the
upper transistor M1 is off; the lower transistor
M2 is on (refer to Figure 1), the COMP pin
voltage is higher than the current sense
amplifier output; and the current comparator’s
output is low. The rising edge of the 380KHz
CLK signal sets the RS Flip-Flop. Its output
turns off M2 and turns on M1 thus connecting
the SW pin and inductor to the input supply.
The increasing inductor current is sensed and
amplified by the Current Sense Amplifier. Ramp
compensation is summed to Current Sense
Amplifier output and compared to the Error
Amplifier output by the Current Comparator.
When the Current Sense Amplifier plus Slope
Compensation signal exceeds the COMP pin
voltage, the RS Flip-Flop is reset and the
The output of the Error Amplifier integrates the
voltage difference between the feedback and
the 1.23V bandgap reference. The polarity is
such that an FB pin voltage lower than 1.222V
increases the COMP pin voltage. Since the
COMP pin voltage is proportional to the peak
inductor current an increase in its voltage
increases current delivered to the output. The
lower 10Ω switch ensures that the bootstrap
capacitor voltage is charged during light load
conditions. External Schottky Diode D1 carries
the inductor current when M1 is off.
VIN 3
CURRENT
SENSE
AMPLIFIER
INTERNAL
REGULATORS
OSCILLATOR
SYNC 1
35/380kHz
+
0.7V
--
RUN 8
-2.50V/
2.29V
+
FREQUENCY
FOLDBACK
COMPARATOR
+
SLOPE
COMP
5V
--
CLK
+
SHUTDOWN
COMPARATOR
--
S
Q
R
Q
CURRENT
COMPARATOR
2
BST
4
LX
5
GND
LOCKOUT
COMPARATOR
1.8V
--
+
--
0.7V
1.22V
6
FB
+
ERROR
AMPLIFIER
7
COMP
MP2354_BD01
Figure 1—Functional Block Diagram
MP2354 Rev. 1.4
1/6/2006
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TM
MP2354 – 2A, 23V, 380KHz STEP-DOWN CONVERTER
APPLICATION INFORMATION
C5
10nF
INPUT
4.75V to 23V
OPEN
AUTOMATIC
STARTUP
OPEN
IF NOT USED
8
1
3
2
VIN
BST
4
LX
RUN
MP2354
SYNC
FB
GND
5
D1
B230A
COMP
7
C6
OPEN
6
OUTPUT
2.5V / 2A
C3
3.3nF
MP2354_TAC_F02
Figure 2—Typical Application Circuit
Sync Pin Operation
Inductor
The SYNC pin driving waveform should be a
The inductor is required to supply constant
square wave with a rise time less than 20ns.
current to the output load while being driven by
Minimum High voltage level is 2.7V. Low level
the switched input voltage. A larger value
is less than 0.8V. The frequency of the external
inductor will result in less ripple current that will
sync signal needs to be greater than 445KHz.
result in lower output ripple voltage. However,
the larger value inductor will have a larger
A rising edge on the SYNC pin forces a reset of
physical size, higher series resistance, and/or
the oscillator. The upper transistor M1 is
lower saturation current. A good rule for
switched off immediately if it is not already off.
determining the inductance to use is to allow
250ns later M1 turns on connecting LX to VIN.
the peak-to-peak ripple current in the inductor
Setting the Output Voltage
to be approximately 30% of the maximum
The output voltage is set using a resistive
switch current limit. Also, make sure that the
voltage divider from the output to FB (see
peak inductor current is below the maximum
Figure 2). The voltage divider divides the output
switch current limit. The inductance value can
voltage down by the ratio:
be calculated by:
VFB =
VOUT × R2
R1 + R2
Where VFB is the feedback voltage and VOUT is
the output voltage.
Thus the output voltage is:
VOUT =
1.23 × (R1 + R2)
R2
R2 can be as high as 100kΩ, but a typical value
is 10kΩ. Using that value, R1 is determined by:
R1 = 8.18 × (VOUT − 1.23 )(kΩ )
For example, for a 3.3V output voltage, R2 is
10kΩ, and R1 is 17kΩ.
MP2354 Rev. 1.4
1/6/2006
L=
⎛
⎞
VOUT
V
× ⎜1 − OUT ⎟⎟
fS × ∆IL ⎜⎝
VIN ⎠
Where VIN is the input voltage, fS is the 380KHz
switching frequency, and ∆IL is the peak-topeak inductor ripple current.
Choose an inductor that will not saturate under
the maximum inductor peak current. The peak
inductor current can be calculated by:
ILP = ILOAD +
⎛
VOUT
V
× ⎜⎜1 − OUT
2 × fS × L ⎝
VIN
⎞
⎟⎟
⎠
Where ILOAD is the load current.
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TM
MP2354 – 2A, 23V, 380KHz STEP-DOWN CONVERTER
Table 1 lists a number of suitable inductors
from various manufacturers. The choice of
which style inductor to use mainly depends on
the price vs. size requirements and any EMI
requirement.
Table 1—Inductor Selection Guide
Vendor/
Model
Package
Dimensions
(mm)
W
L
H
Core
Type
Core
Material
Open
Open
Shielded
Shielded
Shielded
Shielded
Ferrite
Ferrite
Ferrite
Ferrite
Ferrite
Ferrite
7.0
7.3
5.5
5.5
6.7
10.1
7.8
8.0
5.7
5.7
6.7
10.0
5.5
5.2
5.5
5.5
3.0
3.0
Shielded
Ferrite
5.0
5.0
3.0
Shielded
Shielded
Open
Ferrite
Ferrite
Ferrite
7.6
10.0
9.7
7.6
10.0
1.5
5.1
4.3
4.0
Open
Open
Ferrite
Ferrite
9.4
9.4
13.0
13.0
3.0
5.1
Sumida
CR75
CDH74
CDRH5D28
CDRH5D28
CDRH6D28
CDRH104R
Toko
D53LC
Type A
D75C
D104C
D10FL
For simplification, choose the input capacitor
whose RMS current rating greater than half of
the maximum load current.
The input capacitor can be electrolytic, tantalum
or ceramic. When using electrolytic or tantalum
capacitors, a small, high quality ceramic
capacitor, i.e. 0.1µF, should be placed as close
to the IC as possible. When using ceramic
capacitors, make sure that they have enough
capacitance to provide sufficient charge to
prevent excessive voltage ripple at input. The
input voltage ripple caused by capacitance can
be estimated by:
∆VIN =
Input Capacitor
The input current to the step-down converter is
discontinuous, therefore a capacitor is required
to supply the AC current to the step-down
converter while maintaining the DC input
voltage. Use low ESR capacitors for the best
performance. Ceramic capacitors are preferred,
but tantalum or low-ESR electrolytic capacitors
may also suffice. Choose X5R or X7R
dielectrics when using ceramic capacitors.
Since the input capacitor (C1) absorbs the input
switching current it requires an adequate ripple
current rating. The RMS current in the input
capacitor can be estimated by:
I C1 = ILOAD
⎛
⎞
V
V
× OUT ×⎜⎜1− OUT ⎟⎟
VIN ⎝
VIN ⎠
The worst-case condition occurs where:
I C1
MP2354 Rev. 1.4
1/6/2006
I
= LOAD
2
⎞
⎟⎟
⎠
Output Capacitor
The output capacitor is required to maintain the
DC output voltage. Ceramic, tantalum, or low
ESR electrolytic capacitors are recommended.
Low ESR capacitors are preferred to keep the
output voltage ripple low. The output voltage
ripple can be estimated by:
Coilcraft
DO3308
DO3316
⎛
ILOAD
V
V
× OUT × ⎜1 − OUT
fS × C1 VIN ⎜⎝
VIN
∆VOUT =
VOUT ⎛
V
× ⎜⎜1 − OUT
fS × L ⎝
VIN
⎞
⎞ ⎛
1
⎟
⎟⎟ × ⎜ R ESR +
⎜
8 × f S × C2 ⎟⎠
⎠ ⎝
Where L is the inductor value, C2 is the output
capacitance value and RESR is the equivalent
series resistance (ESR) value of the output
capacitor.
In the case of ceramic capacitors, the
impedance at the switching frequency is
dominated by the capacitance. The output
voltage ripple is mainly caused by the
capacitance. For simplification, the output
voltage ripple can be estimated by:
∆VOUT =
⎛
V
× ⎜⎜1 − OUT
VIN
× L × C2 ⎝
VOUT
8 × fS
2
⎞
⎟⎟
⎠
In the case of tantalum or electrolytic
capacitors, the ESR dominates the impedance
at the switching frequency. For simplification,
the output ripple can be approximated to:
∆VOUT =
VOUT ⎛
V
⎞
× ⎜1 − OUT ⎟⎟ × R ESR
f S × L ⎜⎝
VIN ⎠
The characteristics of the output capacitor also
affect the stability of the regulation system. The
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MP2354 can be optimized for a wide range of
capacitance and ESR values.
The DC gain of the voltage feedback loop is
given by:
A VDC = R LOAD × G CS × A VEA ×
VFB
VOUT
Where AVEA is the error amplifier voltage gain,
400V/V;
GCS
is
the
current
sense
transconductance, 1.95A/V; RLOAD is the load
resistor value.
Choose a diode whose maximum reverse
voltage rating is greater than the maximum
input voltage, and whose current rating is
greater than the maximum load current.
The system has two poles of importance. One
is due to the compensation capacitor (C3) and
the output resistor of error amplifier, and the
other is due to the output capacitor and the load
resistor. These poles are located at:
M
IN PS
D TE CO
O
R
N NA NF
O
I
L
D
T
D US EN
IS
T
E
I
T
R ON AL
IB
L
U
Y
T
E
Output Rectifier Diode
The output rectifier diode supplies the current to
the inductor when the upper transistor M1 is off.
Use a Schottky diode to reduce losses due to
the diode forward voltage and recovery times.
Table 2 provides the Schottky diode part
numbers based on the maximum input voltage
and current rating.
Table 2—Schottky Rectifier Selection Guide
VIN (Max)
15V
20V
26V
2A Load Current
Part Number
Vendor (5)
30BQ015
4
B220
1
SK23
6
SR22
6
20BQ030
4
B230
1
SK23
6
SR23
3, 6
SS23
2, 3
Note:
5) Refer to Table 3 for Rectifier Manufacturers
Table 3—Schottky Diode Manufacturers
#
1
2
3
4
5
6
Vendor
Diodes, Inc.
Fairchild Semiconductor
General Semiconductor
International Rectifier
On Semiconductor
Pan Jit International
Web Site
www.diodes.com
www.fairchildsemi.com
www.gensemi.com
www.irf.com
www.onsemi.com
www.panjit.com.tw
Compensation
MP2354 employs current mode control for easy
compensation and fast transient response. The
system stability and transient response are
controlled through the COMP pin. COMP pin is
the output of the internal transconductance
error amplifier. A series capacitor-resistor
combination sets a pole-zero combination to
control the characteristics of the control system.
fP1 =
GEA
2π × C3 × A VEA
fP2 =
1
2π × C2 × R LOAD
Where
GEA
is
the
transconductance, 770µA/V.
error
amplifier
The system has one zero of importance, due to
the compensation capacitor (C3) and the
compensation resistor (R3). This zero is located
at:
f Z1 =
1
2π × C3 × R3
Smaller fZ1 provides more phase margin, but
longer transient settling time. A trade-off has to
be made between the stability and the transient
response. A typical value is less than one-fourth
of the crossover frequency.
The system may have another zero of
importance, if the output capacitor has a large
capacitance and/or a high ESR value. The zero,
due to the ESR and capacitance of the output
capacitor, is located at:
fESR =
1
2π × C2 × R ESR
TM
MP2354 – 2A, 23V, 380KHz STEP-DOWN CONVERTER
In this case, a third pole set by
compensation capacitor (C6) and
compensation resistor (R3) is used
compensate the effect of the ESR zero on
loop gain. This pole is located at:
f P3 =
the
the
to
the
1
2π × C6 × R3
The goal of compensation design is to shape
the converter transfer function to get a desired
loop gain. The system crossover frequency
where the feedback loop has the unity gain is
important. Lower crossover frequencies result
in slower line and load transient responses,
while higher crossover frequencies could cause
system unstable. A good rule of thumb is to set
the crossover frequency to approximately onetenth of the switching frequency. Switching
frequency for the MP2354 is 380KHz, so the
desired crossover frequency is around 38KHz.
Table 4 lists the typical values of compensation
components for some standard output voltages
with various output capacitors and inductors.
The values of the compensation components
have been optimized for fast transient
responses and good stability at given
conditions.
Table 4—Compensation Values for Typical
Output Voltage/Capacitor Combinations
VOUT
L1
C2
R3
C3
C6
2.5V
10µH
min.
22µF
Ceramic
5.6kΩ
4.7nF
None
3.3V
15µH
min.
22µF
Ceramic
7.5kΩ
3.3nF
None
5V
15µH
min.
22µF
Ceramic
11kΩ
2.2nF
None
12V
22µH
min.
22µF
Ceramic
27kΩ
1nF
None
2.5V
10µH
min.
560µF Al.
30mΩ ESR
140kΩ
1nF
120pF
3.3V
15µH
min.
560µF Al
30mΩ ESR
187kΩ
1nF
82pF
5V
15µH
min.
470µF Al.
30mΩ ESR
237kΩ
1nF
56pF
12V
22µH
min.
220µF Al.
30mΩ ESR
267kΩ
1nF
22pF
MP2354 Rev. 1.4
1/6/2006
To optimize the compensation components for
conditions not listed in Table 4, the following
procedure can be used.
1) Choose the compensation resistor (R3) to
set the desired crossover frequency. Determine
the R3 value by the following equation:
R3 =
2π × C2 × f C VOUT
×
G EA × G CS
VFB
2) Choose the compensation capacitor (C3) to
achieve the desired phase margin. For
applications with typical inductor values, setting
the compensation zero, fZ1, to less than one
forth of the crossover frequency provides
sufficient phase margin. Determine the C3
value by the following equation:
C3 >
4
2π × R3 × f C
Where R3 is the compensation resistor value.
3) Determine if the second compensation
capacitor (C6) is required. It is required if the
ESR zero of the output capacitor is located at
less than half of the 380KHz switching
frequency, or the following relationship is valid:
f
1
< S
π
×
×
2 C2 R ESR
2
If this is the case, then add the second
compensation capacitor (C6) to set the pole fP3
at the location of the ESR zero.
Determine the C6 value by the equation:
C6 =
C2 × RESR
R3
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9
TM
MP2354 – 2A, 23V, 380KHz STEP-DOWN CONVERTER
5V
External Bootstrap Diode
It is recommended that an external bootstrap
diode be added when the system has a 5V
fixed input or the power supply generates a 5V
output. This helps improve the efficiency of the
regulator. The bootstrap diode can be a low
cost one such as IN4148 or BAT54.
BS
10nF
MP2354
SW
MP2354_F03
Figure 3—External Bootstrap Diode
This diode is also recommended for high duty
cycle operation (when
VOUT
>65%) and high
VIN
output voltage (VOUT>12V) applications.
PACKAGE INFORMATION
SOIC8
PIN 1 IDENT.
0.229(5.820)
0.244(6.200)
0.0075(0.191)
0.0098(0.249)
0.150(3.810)
0.157(4.000)
SEE DETAIL "A"
0.011(0.280) x 45o
0.020(0.508)
0.013(0.330)
0.020(0.508)
0.050(1.270)BSC
0.189(4.800)
0.197(5.004)
0.053(1.350)
0.068(1.730)
0o-8o
0.049(1.250)
0.060(1.524)
0.016(0.410)
0.050(1.270)
DETAIL "A"
SEATING PLANE
0.001(0.030)
0.004(0.101)
NOTE:
1) Control dimension is in inches. Dimension in bracket is millimeters.
NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications.
Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS
products into any application. MPS will not assume any legal responsibility for any said applications.
MP2354 Rev. 1.4
1/6/2006
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© 2006 MPS. All Rights Reserved.
10