TI1 LM5642 High voltage, dual synchronous buck converter with oscillator synchronization Datasheet

LM5642, LM5642X
www.ti.com
SNVS219J – MAY 2004 – REVISED MAY 2011
LM5642/LM5642X High Voltage, Dual Synchronous Buck Converter with Oscillator
Synchronization
Check for Samples: LM5642, LM5642X
FEATURES
DESCRIPTION
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•
The LM5642 series consists of two current mode
synchronous buck regulator controllers operating
180° out of phase with each other at a normal
switching frequency of 200kHz for the LM5642 and at
375kHz for the LM5642X.
1
2
•
•
•
•
•
•
•
•
•
•
•
•
•
•
Two Synchronous Buck Regulators
180° Out of Phase Operation
200 kHz Fixed Nominal Frequency: LM5642
375 kHz Fixed Nominal Frequency: LM5642X
Synchronizable Switching Frequency from 150
kHz to 250 kHz for the LM5642 and 200 kHz to
500 kHz for the LM5642X
4.5V to 36V Input Range
50 µA Shutdown Current
Adjustable Output from 1.3V to 90% of Vin
0.04% (Typical) Line and Load Regulation
Accuracy
Current Mode Control with or without a Sense
Resistor
Independent Enable/Soft-start Pins Allow
Simple Sequential Startup Configuration.
Configurable for Single Output Parallel
Operation. (See Figure 4)
Adjustable Cycle-by-cycle Current Limit
Input Under-voltage Lockout
Output Over-voltage Latch Protection
Output Under-voltage Protection with Delay
Thermal Shutdown
Self Discharge of Output Capacitors when the
Regulator is OFF
TSSOP and HTSSOP (Exposed PAD) Packages
APPLICATIONS
•
•
•
•
•
•
Embedded Computer Systems
Navigation Systems
Telecom Systems
Set-Top Boxes
WebPAD
Point Of Load Power Architectures
Out of phase operation reduces the input RMS ripple
current, thereby significantly reducing the required
input capacitance. The switching frequency can be
synchronized to an external clock between 150 kHz
and 250 kHz for the LM5642 and between 200 kHz
and 500 kHz for the LM5642X. The two switching
regulator outputs can also be paralleled to operate as
a dual-phase, single output regulator.
The output of each channel can be independently
adjusted from 1.3V to 90% of Vin. An internal 5V rail
is also available externally for driving bootstrap
circuitry.
Current-mode feedback control assures excellent line
and load regulation and wide loop bandwidth for
excellent response to fast load transients. Current is
sensed across either the Vds of the top FET or
across an external current-sense resistor connected
in series with the drain of the top FET.
The LM5642 features analog soft-start circuitry that is
independent of the output load and output
capacitance making the soft-start behavior more
predictable and controllable than traditional soft-start
circuits.
Over-voltage protection is available for both outputs.
A UV-Delay pin is also available to allow delayed shut
off time for the IC during an output under-voltage
event.
Typical Application Circuit
VIN
4.5V - 36V
UV_Delay
Vout1
1.3V-0.9VIN
SYNC
LM5642/LM5642X
SS/ON1
SS/ON2
Vout2
1.3V-0.9VIN
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2004–2011, Texas Instruments Incorporated
LM5642, LM5642X
SNVS219J – MAY 2004 – REVISED MAY 2011
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Connection Diagram
KS1
1
28
RSNS1
ILIM1
2
27
SW1
COMP1
3
26
HDRV1
FB1
4
25
CBOOT1
SYNC
5
24
VDD1
UVDELAY
6
23
LDRV1
VLIN5
7
22
VIN
SGND
8
21
PGND
ON/SS1
9
20
ON/SS2
10
KS1
1
28
RSNS1
ILIM1
2
27
SW1
COMP1
3
26
HDRV1
FB1
4
25
CBOOT1
SYNC
5
24
VDD1
UVDELAY
6
23
LDRV1
VLIN5
7
22
VIN
DAP
SGND
8
21
PGND
LDRV2
ON/SS1
9
20
LDRV2
19
VDD2
ON/SS2
10
19
VDD2
FB2
11
18
CBOOT2
COMP2
12
17
HDRV2
ILIM2
13
16
SW2
KS2
14
15
RSNS2
FB2
11
18
CBOOT2
COMP2
12
17
HDRV2
ILIM2
13
16
SW2
KS2
14
15
RSNS2
Figure 1. Top View
Figure 2. Top View
PIN DESCRIPTIONS
KS1 (Pin 1)
The positive (+) Kelvin sense for the internal current sense amplifier of Channel 1. Use a separate trace to
connect this pin to the current-sense point. It should be connected to VIN as close as possible to the currentsense resistor. When no current-sense resistor is used, connect as close as possible to the drain node of the
upper MOSFET.
ILIM1 (Pin 2)
Current limit threshold setting for Channel 1. It sinks a constant current of 9.9 µA, which is converted to a voltage
across a resistor connected from this pin to VIN. The voltage across the resistor is compared with either the VDS
of the top MOSFET or the voltage across the external current sense resistor to determine if an over-current
condition has occurred in Channel 1.
COMP1 (Pin 3)
Compensation pin for Channel 1. This is the output of the internal transconductance error amplifier. The loop
compensation network should be connected between this pin and the signal ground, SGND (Pin 8).
FB1 (Pin 4)
Feedback input for channel 1. Connect to VOUT through a voltage divider to set the Channel 1 output voltage.
SYNC (Pin 5)
The switching frequency of the LM5642 can be synchronized to an external clock.
SYNC = LOW: Free running at 200 kHz for LM5642, and at 375kHz for LM5642X. Channels are 180° out of
phase.
SYNC = HIGH: Waiting for external clock
SYNC = Falling Edge: Channel 1 HDRV pin goes high. Channel 2 HDRV pin goes high after 2.5 µs delay. The
maximum SYNC pulse width must be greater than 100 ns.
For SYNC = Low operation, connect this pin to signal ground through a 220 kΩ resistor.
UV_DELAY (Pin 6)
A capacitor from this pin to ground sets the delay time for UVP. The capacitor is charged from a 5 µA current
source. When UV_DELAY charges to 2.3V (typical), the system immediately latches off. Connecting this pin to
ground will disable the output under-voltage protection.
VLIN5 (Pin 7)
The output of an internal 5V LDO regulator derived from VIN. It supplies the internal bias for the chip and powers
the bootstrap circuitry for gate drive. Bypass this pin to signal ground with a minimum of 4.7 µF ceramic capacitor.
SGND (Pin 8)
The ground connection for the signal-level circuitry. It should be connected to the ground rail of the system.
ON/SS1 (Pin 9)
Channel 1 enable pin. This pin is internally pulled up to one diode drop above VLIN5. Pulling this pin below 1.2V
(open-collector type) turns off Channel 1. If both ON/SS1 and ON/SS2 pins are pulled below 1.2V, the whole chip
goes into shut down mode. Adding a capacitor to this pin provides a soft-start feature that minimizes inrush
current and output voltage overshoot.
ON/SS2 (Pin 10)
Channel 2 enable pin. See the description for Pin 9, ON/SS1. May be connected to ON/SS1 for simultaneous
startup or for parallel operation.
FB2 (Pin 11)
Feedback input for channel 2. Connect to VOUT through a voltage divider to set the Channel 2 output voltage.
2
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SNVS219J – MAY 2004 – REVISED MAY 2011
PIN DESCRIPTIONS (continued)
COMP2 (Pin 12)
Compensation pin for Channel 2. This is the output of the internal transconductance error amplifier. The loop
compensation network should be connected between this pin and the signal ground SGND (Pin 8).
ILIM2 (Pin 13)
Current limit threshold setting for Channel 2. See ILIM1 (Pin 2).
KS2 (Pin 14)
The positive (+) Kelvin sense for the internal current sense amplifier of Channel 2. See KS1 (Pin 1).
RSNS2 (Pin 15)
The negative (-) Kelvin sense for the internal current sense amplifier of Channel 2. Connect this pin to the low side
of the current sense resistor that is placed between VIN and the drain of the top MOSFET. When the Rds of the
top MOSFET is used for current sensing, connect this pin to the source of the top MOSFET. Always use a
separate trace to form a Kelvin connection to this pin.
SW2 (Pin 16)
Switch-node connection for Channel 2, which is connected to the source of the top MOSFET of Channel 2. It
serves as the negative supply rail for the top-side gate driver, HDRV2.
HDRV2 (Pin 17)
Top-side gate-drive output for Channel 2. HDRV is a floating drive output that rides on the corresponding
switching-node voltage.
CBOOT2 (Pin 18)
Bootstrap capacitor connection. It serves as the positive supply rail for the Channel 2 top-side gate drive. Connect
this pin to VDD2 (Pin 19) through a diode, and connect the low side of the bootstrap capacitor to SW2 (Pin16).
VDD2 (Pin 19)
The supply rail for the Channel 2 low-side gate drive. Connected to VLIN5 (Pin 7) through a 4.7Ω resistor and
bypassed to power ground with a ceramic capacitor of at least 1µF. Tie this pin to VDD1 (Pin 24).
LDRV2 (Pin 20)
Low-side gate-drive output for Channel 2.
PGND (Pin 21)
The power ground connection for both channels. Connect to the ground rail of the system.
VIN (Pin 22)
The power input pin for the chip. Connect to the positive (+) input rail of the system. This pin must be connected
to the same voltage rail as the top FET drain (or the current sense resistor when used).
LDRV1 (Pin 23)
Low-side gate-drive output for Channel 1.
VDD1 (Pin 24)
The supply rail for Channel 1 low-side gate drive. Tie this pin to VDD2 (Pin 19).
CBOOT1 (Pin 25)
Bootstrap capacitor connection. This pin serves as the positive supply rail for the Channel 1 top-side gate drive.
See CBOOT2 (Pin 18).
HDRV1 (Pin 26)
Top-side gate-drive output for Channel 1. See HDRV2 (Pin 17).
SW1 (Pin 27)
Switch-node connection for Channel 1. See SW2 (Pin16).
RSNS1 (Pin 28)
The negative (-) Kelvin sense for the internal current sense amplifier of Channel 1. See RSNS2 (Pin 15).
PGND (DAP)
The power ground connection for both channels. Connect to the ground rail of the system. Use of multiple vias to
internal ground plane or GND layer helps to dissipate heat generated by output power.
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ABSOLUTE MAXIMUM RATINGS (1) (2)
Voltages from the indicated pins to SGND/PGND:
VIN, ILIM1, ILIM2, KS1, KS2
−0.3V to 38V
SW1, SW2, RSNS1, RSNS2
−0.3 to (VIN + 0.3)V
−0.3V to 6V
FB1, FB2, VDD1, VDD2
−0.3V to (VLIN5 +0.3)V
SYNC, COMP1, COMP2, UV Delay
ON/SS1, ON/SS2
(3)
−0.3V to (VLIN5 +0.6)V
CBOOT1, CBOOT2
43V
−0.3V to 7V
CBOOT1 to SW1, CBOOT2 to SW2
−0.3V to (VDD+0.3)V
LDRV1, LDRV2
HDRV1 to SW1, HDRV2 to SW2
−0.3V
HDRV1 to CBOOT1, HDRV2 to CBOOT2
+0.3V
Power Dissipation (TA = 25°C) (4)
TSSOP
1.1W
HTSSOP
3.4W
−65°C to +150°C
Ambient Storage Temp. Range
Soldering Dwell Time, Temp. (5)
ESD Rating
(1)
(2)
(3)
(4)
(5)
(6)
Wave
4 sec, 260°C
Infrared
10sec, 240°C
Vapor Phase
75sec, 219°C
(6)
2kV
Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating Range indicates conditions for
which the device is intended to be functional, but does not ensure specific performance limits. For ensured specifications and test
conditions, see the Electrical Characteristics. The ensured specifications apply only for the test conditions. Some performance
characteristics may degrade when the device is not operated under the listed test conditions.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
ON/SS1 and ON/SS2 are internally pulled up to one diode drop above VLIN5. Do not apply an external pull-up voltage to these pins. It
may cause damage to the IC.
The maximum allowable power dissipation is calculated by using PDMAX = (TJMAX - TA)/θJA, where TJMAX is the maximum junction
temperature, TA is the ambient temperature and θJA is the junction-to-ambient thermal resistance of the specified package. The power
dissipation ratings results from using 125°C, 25°C, and 90.6°C/W for TJMAX, TA, and θJA respectively. A θJA of 90.6°C/W represents the
worst-case condition of no heat sinking of the 28-pin TSSOP. The HTSSOP package has a θJA of 29°C/W. The HTSSOP package
thermal ratings results from the IC being mounted on a 4 layer JEDEC standard board using the same temperature conditions as the
TSSOP package above. A thermal shutdown will occur if the temperature exceeds the maximum junction temperature of the device.
See http://www.ti.com for other methods of soldering plastic small-outline packages.
For testing purposes, ESD was applied using the human-body model, a 100pF capacitor discharged through a 1.5 kΩ resistor.
OPERATING RATINGS
(1)
VIN (VLIN5 tied to VIN)
4.5V to 5.5V
VIN (VIN and VLIN5 separate)
5.5V to 36V
−40°C to +125°C
Junction Temperature
(1)
4
Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating Range indicates conditions for
which the device is intended to be functional, but does not ensure specific performance limits. For ensured specifications and test
conditions, see the Electrical Characteristics. The ensured specifications apply only for the test conditions. Some performance
characteristics may degrade when the device is not operated under the listed test conditions.
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LM5642, LM5642X
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SNVS219J – MAY 2004 – REVISED MAY 2011
ELECTRICAL CHARACTERISTICS
Unless otherwise specified, VIN = 28V, GND = PGND = 0V, VLIN5 = VDD1 = VDD2. Limits appearing in boldface type apply
over the specified operating junction temperature range, (-40°C to +125°C, if not otherwise specified). Specifications
appearing in plain type are measured using low duty cycle pulse testing with TA = 25°C (1), (2). Min/Max limits are specified by
design, test, or statistical analysis.
Symbol
Parameter
Conditions
Min
Typ
Max
Units
System
ΔVOUT/VOUT
VFB1_FB2
IVIN
Load Regulation
VIN = 28V, Vcompx = 0.5V to 1.5V
0.04
%
Line Regulation
5.5V ≤ VIN ≤ 36V, Vcompx =1.25V
0.04
%
Feedback Voltage
5.5V ≤ VIN ≤ 36V
1.2154
-20°C to 85°C
1.2179
Input Supply Current
1.2364
1.2364
1.2574
1.2549
V
VON_SSx > 2V
5.5V ≤ VIN ≤ 36V
1.1
2.0
mA
Shutdown (3)
VON_SS1 = VON_SS2= 0V
50
110
µA
5
5.30
V
±2
±7.0
mV
8.4
9.9
11.4
µA
0.5
2.4
5.0
µA
2
5.5
10
µA
0.7
1.12
1.4
V
VLIN5
VLIN5 Output Voltage
IVLIN5 = 0 to 25mA,
5.5V ≤ VIN ≤ 36V
VCLos
Current Limit Comparator
Offset (VILIMX −VRSNSX)
VIN = 6V
ICL
Current Limit Sink Current
Iss_SC1,
Iss_SC2
Soft-Start Source Current
VON_ss1 = VON_ss2 = 1.5V (on)
Iss_SK1,
Iss_SK2
Soft-Start Sink Current
VON_ss1 = VON_ss2 = 1.5V
VON_SS1,
VON_SS2
Soft-Start On Threshold
VSSTO
Soft-Start Timeout
Threshold
Isc_uvdelay
UV_DELAY Source Current
UV-DELAY = 2V
Isk_uvdelay
UV_DELAY Sink Current
UV-DELAY = 0.4V
VUVDelay
UV_DELAY Threshold
Voltage
VUVP
FB1, FB2, Under Voltage
Protection Latch Threshold
4.70
(4)
3.4
V
2
5
9
µA
0.2
0.48
1.2
mA
2.3
As a percentage of nominal output voltage
(falling edge)
75
Hysteresis
80.7
V
86
3.7
%
%
VOVP
VOUT Overvoltage
Shutdown Latch Threshold
As a percentage measured at VFB1, VFB2
107
114
122
%
Swx_R
SW1, SW2 ON-Resistance
VSW1 = VSW2 = 0.4V
420
487
560
Ω
(1)
(2)
(3)
(4)
A typical is the center of characterization data measured with low duty cycle pulse tsting at TA = 25°C. Typicals are not ensured.
All limits are specified. All electrical characteristics having room-temperature limits are tested during production with TA = TJ = 25°C. All
hot and cold limits are specified by correlating the electrical characteristics to process and temperature variations and applying statistical
process control.
Both switching controllers are off. The linear regulator VLIN5 remains on.
When SS1 and SS2 pins are charged above this voltage and either of the output voltages at Vout1 or Vout2 is still below the regulation
limit, the under voltage protection feature is initialized.
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ELECTRICAL CHARACTERISTICS (continued)
Unless otherwise specified, VIN = 28V, GND = PGND = 0V, VLIN5 = VDD1 = VDD2. Limits appearing in boldface type apply
over the specified operating junction temperature range, (-40°C to +125°C, if not otherwise specified). Specifications
appearing in plain type are measured using low duty cycle pulse testing with TA = 25°C (1), (2). Min/Max limits are specified by
design, test, or statistical analysis.
Symbol
Parameter
Conditions
Min
Typ
Max
Units
Gate Drive
ICBOOT
CBOOTx Leakage Current
VCBOOT1 = VCBOOT2 = 7V
10
nA
ISC_DRV
HDRVx and LDRVx Source
Current
VCBOOT1 = VCBOOT2 = 5V, VSWx=0V,
HDRVx=LDRVx=2.5V
0.5
A
Isk_HDRV
HDRVx Sink Current
VCBOOTx = VDDx = 5V, VSWx = 0V, HDRVX
= 2.5V
0.8
A
Isk_LDRV
LDRVx Sink Current
VCBOOTx = VDDx = 5V, VSWx = 0V, LDRVX
= 2.5V
1.1
A
RHDRV
HDRV1 & 2 Source OnResistance
VCBOOT1 = VCBOOT2 = 5V,
VSW1 = VSW2 = 0V
3.1
Ω
1.5
Ω
3.1
Ω
1.1
Ω
HDRV1 & 2 Sink OnResistance
RLDRV
LDRV1 & 2 Source OnResistance
LDRV1 & 2 Sink OnResistance
VCBOOT1 = VCBOOT2 = 5V,
VSW1 = VSW2 = 0V
VDD1 = VDD1 = 5V
Oscillator and Sync Controls
Fosc
Oscillator Frequency
Don_max
Maximum On-Duty Cycle
Ton_min
Minimum On-Time
SSOT_delta
HDRV1 and HDRV2 Delta
On Time
VHS
SYNC Pin Min High Input
VLS
SYNC Pin Max Low Input
5.5 ≤ VIN ≤ 36V, LM5642
166
200
226
5.5 ≤ VIN ≤ 36V, LM5642X
311
375
424
VFB1 = VFB2 = 1V, Measured at pins
HDRV1 and HDRV2
96
98.9
%
166
ns
ON/SS1 = ON/SS2 = 2V
20
2
250
1.52
kHz
ns
V
1.44
0.8
V
80
±200
nA
Error Amplifier
IFB1, IFB2
Feedback Input Bias
Current
VFB1_FIX = 1.5V, VFB2_FIX = 1.5V
Icomp1_SC,
Icomp2_SC
COMP Output Source
Current
VFB1_FIX = VFB2_FIX = 1V,
VCOMP1 = VCOMP2 = 1V
6
-20°C to 85°C
18
VFB1_FIX = VFB2_FIX = 1.5V and
VCOMP1 = VCOMP2 = 0.5V
6
-20°C to 85°C
18
Icomp1_SK,
Icomp2_SK
COMP Output Sink Current
gm1, gm2
Transconductance
GISNS1,
GISNS2
Current Sense Amplifier
(1&2) Gain
127
µA
118
µA
720
VCOMPx = 1.25V
µmho
4.2
5.2
7.5
3.6
4.0
4.4
Voltage References and Linear Voltage Regulators
UVLO
6
VLIN5 Under-voltage
Lockout
Threshold Rising
ON/SS1, ON/SS2 transition
from low to high
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C3 100 pF
C2 10 nF
Vin = 24V+10%
22
C1
ILIM1
VIN
IC1
LM5642
1 PF
KS1
RSNS1
6
C34
SYNC
UV_DELAY
100 nF
5
HDRV1
CBOOT1
SW1
220 k:
S1
C11
ON/SS1
LDRV1
PGND
10 nF
FB1
10
S2
C12
VDD
19
VDD1
VDD2
R27
7
4.7 :
C27
C26
3
C19
1 PF 4.7 PF
R23
8.45
k:
8.2 nF
12
1
12 k:
28
25
27
R2 100:
C6
R7
10 m:
C4 100 pF
R6
26
C7
Q1
100:
D3A BAS40-06
VDD
L1
4.2 PH
7 m:
ILIM2
KS2
RSNS2
Q2
C13 10 nF C14 100 pF
COMP2
HDRV2
CBOOT2
C20
15 nF
SW2
R24
13.7
k:
8
LDRV2
SGND
FB2
R15
10 m:
C16 100 pF
17
18
16 C25
11
C9
+ 330
PF
6.3V
10 m:
VIN
6.8 k:
14
20
R10
2.26
k:
R14 100:
R13
13
15
Vo1 = 1.8V, 7A
4.99 k:
4
VLIN5
COMP1
R11
Si4840DY
21
10 PF
50V
2.8Arms
Si4850EY
100 nF
23
ON/SS2
10 nF
24
R1
SYNC
R28
9
2
VIN
R16
Q4
100:
D3B BAS40-06
VDD
Si4850EY
L2
100 nF
Q5
C16
10 PF
50V
2.8Arms
10 P+
12 m:
Si4840DY
R20
Vo2 = 3.3V, 4A
R19
8.25
k:
+
C23
330 PF
6.3V
10 m:
4.99 k:
Figure 3. Typical 2 Channel Application Circuit
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C3 100 pF
C2 10 nF
Vin = 30Vr 10%
22
VIN
C1
1 PF
ILIM1
IC1
LM5642
1
KS1
28
R2 100:
R1
2
16.9 k:
C34
26
HDRV1
100
nF
CBOOT1
5
SW1
SYNC
R28
220 k:
9
S1
UV_DELAY
C11
R6
ON/SS1
22 nF
LDRV1
FB1
24
VDD
19
ON/SS2
VDD1
7
C27
C26
1 PF
4.7 PF
3
C19
FB2
ILIM2
VDD2
KS2
R27
4.7:
VLIN5
RSNS2
Q2
23
27 nF
HDRV2
CBOOT2
R23
SW2
8 SGND
Si4850EY
L1
Vo = 1.8V, 20A
2.7 PH
4.5 m:
Q3
R10
2.26
k:
R11
21
Si4470DY x 2
4
11
13
14
15
R13
20
1 PF
C16
R15
10 m:
R16
Q4
100:
D3B BAS40-06
VDD
16 C25
C10
R14 100:
C16 100 pF
18
C9
1000 PF
16V
22 m:
VIN
16.9 k:
17
+
4.99 k:
C13 10 nF C14 100 pF
COMP1
12 COMP2
11.5 k:
27
Q1
100:
D3A BAS40-06
VDD
C7
100 nF
SYNC
PGND
10
25
C6
10 PF
50V
2.8Arms
R7
10 m:
C4 100 pF
RSNS1
6
VIN
100 nF
Q5
Q6
10 PF
50V
2.8Arms
Si4850EY
L2
2.7 PH
4.5 m:
LDRV2
Si4470DY x 2
C23
+ 1000 PF
16V
22 m:
C24
1 PF
Figure 4. Typical Single Channel Application Circuit
8
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BLOCK DIAGRAM
VIN
Voltage
and
Current
generator
BG
SD Disable
BG
reference
Bias
Generator
Vref
Current
bias
IREF
Input Power
Supply
+
+
-
5V LDO
(Allways ON)
VLIN5
From another Ch.
10 PA
COMPx
ILIM
Comp
Ch1 and Ch2 are identical
ILIMx
+
KSx
+
-
CHx
output
ISENSE
amp
error amp
FBx
-
PWM comp
Normal:
ON
+
BG
2 PA
PWM logic
control
R Q
HDRVx
SS:
ON
S Q
SWx
Corrective
ramp
ON/OFF
and
S/S
control
ON/SSx
CBOOTx
Shifter
and latch
+
-
RSNSx
0.50V
S/S level
+
Cycle
Skip
comp
+
-
Shoot through
protection
sequencer
CHx
Output
+
VDDx
LDRVx
7 PA
PGNDx
fault
5 PA
UV_DELAY
FAULT
TSD
UVLO
Active
discharge
Rdson =
500:
UVP
R
Q
S
Q
R
Q
S
Reset by
POR or SD
Q
UV
UVP
OVP
UVPG1
comparator
OVP
To Ch2
From
another
CH.
0
2.5 Ps
delay
OSC
200 kHz LM5642
or
375 kHz LM5642X
SYNC
SGND
Figure 5. Block Diagram
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TYPICAL PERFORMANCE CHARACTERISTICS
Softstart Waveforms (No-Load Both Channels)
UVP Startup Waveform (VIN = 24V)
ON/SS1, 2V/div
Vo2, 2V/div
Vo1,
1V/div
Vo1, 2V/div
ON/SS1 and 2,
5V/div, VIN = 36V
Vo2, 2V/div
Io1, 5A/div
Vo1, 2V/div
ON/SS1 and 2,
5V/div, VIN = 24V
UV DELAY, 2V/div
20ms/DIV
4 ms/DIV
Figure 6.
Figure 7.
Over-Current and UVP Shutdown (VIN = 24V, Io2 = 0A)
Shutdown Waveforms (VIN = 24V, No-Load)
Io1, 5A/div
Vo2, 1V/div
Vo1, 1V/div
Vo2, 1V/div
Vo1, 1V/div
ON/SS1 and 2, 5V/div
UV DELAY, 1V/div
100ms/DIV
20ms/DIV
Figure 8.
Figure 9.
Ch.1 Load Transient Response (VIN = 24V, Vo1 = 1.8V)
Ch.2 Load Transient Response (VIN = 24V, Vo2 = 3.3V)
Io2, 2A/DIV
Io1, 2A/DIV
Vo2, 100mV/DIV
Vo1, 100mV/DIV
100Ps/DIV
100Ps/DIV
Figure 10.
10
Figure 11.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Ch. 2 Load Transient Response (VIN = 36V, Vo2 = 3.3V)
Ch.1 Load Transient Response (VIN = 36V, Vo1 = 1.8V)
Io2, 2A/DIV
Io1, 2A/DIV
Vo2, 100mV/DIV
Vo1, 100mV/DIV
100Ps/DIV
100Ps/DIV
Figure 12.
Figure 13.
Input Supply Current vs Temperature
(Shutdown Mode VIN = 28V)
Input Supply Current vs VIN
Shutdown Mode (25°C)
55
53.5
53
50
52
45
IQ (PA)
IQ (PA)
52.5
40
51.5
51
50.5
35
50
30
-40
-20
0
25
50
75
100
49.5
5.5
125
8
12
16
20
24
28
32
36
32
36
TEMPERATURE (oC)
VIN
Figure 14.
Figure 15.
VLIN5 vs Temperature
VLIN5 vs VIN (25°C)
5.095
5.1
5.08
5.09
VIN = 36V
5.06
5.04
VLIN5 (V)
VLIN5 (V)
5.085
VIN = 5.5V
5.02
5.08
5.075
5
5.07
4.98
4.96
-40
5.065
-20
0
25
50
75
100
125
5.5
TEMPERATURE (oC)
8
12
16
20
24
28
VIN (V)
Figure 16.
Figure 17.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Operating Frequency vs Temperature
(VIN = 28V)
FB Reference Voltage vs Temperature
204
1.2365
202
1.236
200
FREQUENCY (kHz)
1.237
1.2355
VREF (V)
1.235
1.2345
1.234
1.2335
198
196
194
192
190
1.233
188
1.2325
186
1.232
-40
-20
0
25
50
75
100
184
-40
125
-20
o
0
25
50
75
100
125
o
TEMPERATURE ( C)
TEMPERATURE ( C)
Figure 18.
Figure 19.
Error Amplifier Tranconductance Gain
vs
Temperature
Efficiency vs Load Current Using Resistor Sense
Ch.1 = 1.8V, Ch.2 = Off
750
100
700
90
650
80
EFFICIENCY (%)
EA gm (Pmho)
VIN = 24V
600
550
70
60
500
50
450
40
400
VIN = 36V
30
-40
-20
0
25
50
75
100
125
0
o
TEMPERATURE ( C)
2
3
4
5
6
7
LOAD CURRENT (A)
Figure 20.
12
1
Figure 21.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Efficiency vs Load Current
Ch.2 = 3.3V, Ch.1 = Off
Efficiency vs Load Current Using Vds Sense
Ch.2 = 1.8V, Ch.2 = Off
100
100
VIN = 24V
VIN = 24V
90
VIN = 36V
90
EFFICIENCY (%)
EFFICIENCY (%)
80
80
70
VIN = 36V
70
60
50
60
40
50
30
0
1
2
3
4
5
0
1
2
3
4
5
6
7
LOAD CURRENT (A)
LOAD CURRENT (A)
Figure 22.
Figure 23.
Efficiency vs Load Current Using Vds Sense
Ch.2 = 3.3V, Ch.1 = Off
100
VIN = 24V
EFFICIENCY (%)
90
VIN = 36V
80
70
60
50
0
1
2
3
4
5
LOAD CURRENT (A)
Figure 24.
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OPERATING DESCRIPTIONS
SOFT START
The ON/SS1 pin has dual functionality as both channel enable and soft start control. Referring to the soft start
block diagram is shown in Figure 25, the LM5642 will remain in shutdown mode while both soft start pins are
grounded.
In a normal application (with a soft start capacitor connected between the ON/SS1 pin and SGND) soft start
functions as follows: As the input voltage rises (note, Iss starts to flow when VIN ≥ 2.2V), the internal 5V LDO
starts up, and an internal 2.4 µA current charges the soft start capacitor. During soft start, the error amplifier
output voltage at the COMPx pin is clamped at 0.55V and the duty cycle is controlled only by the soft start
voltage. As the SSx pin voltage ramps up, the duty cycle increases proportional to the soft start ramp, causing
the output voltage to ramp up. The rate at which the duty cycle increases depends on the capacitance of the soft
start capacitor. The higher the capacitance, the slower the output voltage ramps up. When the corresponding
output voltage exceeds 98% (typical) of the set target voltage, the regulator switches from soft start to normal
operating mode. At this time, the 0.55V clamp at the output of the error amplifier releases and peak current
feedback control takes over. Once in peak current feedback control mode, the output voltage of the error
amplifier will travel within a 0.5V and 2V window to achieve PWM control. See Figure 26.
The amount of capacitance needed for a desired soft-start time can be approximated in the following equation:
Iss x tss
Css =
Vss
where
•
•
Iss = 2.4 µA for one channel and 4.8µA if the channels are paralleled
tss is the desired soft-start time
(1)
Finally,
Vss = 1.5 §
·
+1
© Vin ¹
Vo
(2)
During soft start, over-voltage protection and current limit remain in effect. The under voltage protection feature is
activated when the ON/SS pin exceeds the timeout threshold (3.4V typical). If the ON/SSx capacitor is too small,
the duty cycle may increase too rapidly, causing the device to latch off due to output voltage overshoot above the
OVP threshold. This becomes more likely in applications with low output voltage, high input voltage and light
load. A capacitance of 10 nF is recommended at each soft start pin to provide a smooth monotonic output ramp.
+
2PA
disable
R Q
S>R
S Q
fault
ONx
+
-
ON/SSx
ON: 2.4PA source
Fault: 5.5PA sink
7PA
1.2V/
1.05V
ON/OFF
comparator
+
-
S/S level
S/S buffer
Figure 25. Soft-Start and ON/OFF
14
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low clamp
+
-
0.45V
COMPx
+
high clamp
SS:0.55V
OP:2V
Figure 26. Voltage Clamp at COMPx Pin
FBx
from other CH.
OVP
1.13BG
OVPx
+
shutdown
latch OVP
HDRV: off
LDRV:on
S Q
OVP 1/2
R Q
UVP
in: 0.84BG
out:0.80BG
+
5u
A
UVPx
UV_DELAY
ONx
SS Timeout
from other CH.
S
SD
power on
reset
Q
R Q
shutdown
latch UVP
HDRV: off
LDRV:off TSD
UVLO
fault
Figure 27. OVP and UVP
OVER VOLTAGE PROTECTION (OVP)
If the output voltage on either channel rises above 113% of nominal, over voltage protection activates. Both
channels will latch off. When the OVP latch is set, the high side FET driver, HDRVx, is immediately turned off
and the low side FET driver, LDRVx, is turned on to discharge the output capacitor through the inductor. To reset
the OVP latch, either the input voltage must be cycled, or both channels must be switched off (both ON/SS pins
pulled low).
UNDER VOLTAGE PROTECTION (UVP) AND UV DELAY
If the output voltage on either channel falls below 80% of nominal, under voltage protection activates. As shown
in Figure 27, an under-voltage event will shut off the UV_DELAY MOSFET, which will allow the UV_DELAY
capacitor to charge with 5µA (typical). If the UV_DELAY pin voltage reaches the 2.3V threshold both channels
will latch off. UV_DELAY will then be disabled and the UV_DELAY pin will return to 0V. During UVP, both the
high side and low side FET drivers will be turned off. If no capacitor is connected to the UV_DELAY pin, the UVP
latch will be activated immediately. To reset the UVP latch, either the input voltage must be cycled, or both
ON/SS pins must be pulled low. The UVP function can be disabled by connecting the UV_DELAY pin to ground.
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THERMAL SHUTDOWN
The LM5642 IC will enter thermal shutdown if the die temperature exceeds 160°C. The top and bottom FETs of
both channels will be turned off immediately. In addition, both soft start capacitors will begin to discharge through
separate 5.5 µA current sinks. The voltage on both capacitors will settle to approximately 1.1V, where it will
remain until the thermal shutdown condition has cleared. The IC will return to normal operating mode when the
die temperature has fallen to below 146°C. At this point the two soft start capacitors will begin to charge with
their normal 2.4 µA current sources. This allows a controlled return to normal operation, similar to the soft start
during turn-on. If the thermal shutdown condition clears before the voltage on the soft start capacitors has fallen
to 1.1V, the capacitors will first be discharged to 1.1V, and then immediately begin charging back up.
OUTPUT CAPACITOR DISCHARGE
Each channel has an embedded 480Ω MOSFET with the drain connected to the SWx pin. This MOSFET will
discharge the output capacitor of its channel if its channel is off, or the IC enters a fault state caused by one of
the following conditions:
1. UVP
2. UVLO
If an output over voltage event occurs, the HDRVx will be turned off and LDRVx will be turned on immediately to
discharge the output capacitors of both channels through the inductors.
BOOTSTRAP DIODE SELECTION
The bootstrap diode and capacitor form a supply that floats above the switch node voltage. VLIN5 powers this
supply, creating approximately 5V (minus the diode drop) which is used to power the high side FET drivers and
driver logic. When selecting a bootstrap diode, Schottky diodes are preferred due to their low forward voltage
drop, but care must be taken for circuits that operate at high ambient temperature. The reverse leakage of some
Schottky diodes can increase by more than 1000x at high temperature, and this leakage path can deplete the
charge on the bootstrap capacitor, starving the driver and logic. Standard PN junction diodes and fast rectifier
diodes can also be used, and these types maintain tighter control over reverse leakage current across
temperature.
SWITCHING NOISE REDUCTION
Power MOSFETs are very fast switching devices. In synchronous rectifier converters, the rapid increase of drain
current in the top FET coupled with parasitic inductance will generate unwanted Ldi/dt noise spikes at the source
node of the FET (SWx node) and also at the VIN node. The magnitude of this noise will increase as the output
current increases. This parasitic spike noise may produce excessive electromagnetic interference (EMI), and can
also cause problems in device performance. Therefore, it must be suppressed using one of the following
methods.
When using resistor based current sensing, it is strongly recommended to add R-C filters to the current sense
amplifier inputs as shown in Figure 29. This will reduce the susceptibility to switching noise, especially during
heavy load transients and short on time conditions. The filter components should be connected as close as
possible to the IC.
As shown in Figure 28, adding a resistor in series with the HDRVx pin will slow down the gate drive, thus slowing
the rise and fall time of the top FET, yielding a longer drain current transition time.
Usually a 3.3Ω to 4.7Ω resistor is sufficient to suppress the noise. Top FET switching losses will increase with
higher resistance values.
Small resistors (1-5 ohms) can also be placed in series with the CBOOTx pin to effectively reduce switch node
ringing. A CBOOT resistor will slow the rise time of the FET, whereas a resistor at HDRV will increase both rise
and fall times.
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CBOOTx
HDRVx
SWx
Rsw
4R7
0.1 PF
Figure 28. HDRV Series Resistor
CURRENT SENSING AND LIMITING
As shown in Figure 29, the KSx and RSNSx pins are the inputs of the current sense amplifier. Current sensing is
accomplished either by sensing the Vds of the top FET or by sensing the voltage across a current sense resistor
connected from VIN to the drain of the top FET. The advantages of sensing current across the top FET are
reduced parts count, cost and power loss.
The RDS-ON of the top FET is not as stable over temperature and voltage as a sense resistor, hence great care
must be used in layout for VDS sensing circuits. At input voltages above 30V, the maximum recommended output
current is 5A per channel.
Keeping the differential current-sense voltage below 200mV ensures linear operation of the current sense
amplifier. Therefore, the RDS-ON of the top FET or the current sense resistor must be small enough so that the
current sense voltage does not exceed 200 mV when the top FET is on. There is a leading edge blanking circuit
that forces the top FET on for at least 166ns. Beyond this minimum on time, the output of the PWM comparator
is used to turn off the top FET. Additionally, a minimum voltage of at least 50 mV across Rsns is recommended
to ensure a high SNR at the current sense amplifier.
Assuming a maximum of 200 mV across Rsns, the current sense resistor can be calculated as follows:
where
•
•
Imax is the maximum expected load current, including overload multiplier (ie: 120%)
Irip is the inductor ripple current (see Equation 17)
(3)
The above equation gives the maximum allowable value for Rsns. Conduction losses will increase with larger
Rsns, thus lowering efficiency.
The peak current limit is set by an external resistor connected between the ILIMx pin and the KSx pin. An
internal 10 µA current sink on the ILIMx pin produces a voltage across the resistor to set the current limit
threshold which is then compared to the current sense voltage. A 10 nF capacitor across this resistor is required
to filter unwanted noise that could improperly trip the current limit comparator.
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10 PA
LIMx
comp
LIMx
13k
+
-
POWER
SUPPLY
KSx
10 nF
100
+
ISENSE
amp
20m
RSNSx 100
100 pF
100 pF
Figure 29. Current Sense and Current Limit
Current limit is activated when the inductor current is high enough to cause the voltage at the RSNSx pin to be
lower than that of the ILIMx pin. This toggles the Ilim comparator, thus turning off the top FET immediately. The
comparator is disabled when the top FET is turned off and during the leading edge blanking time. The equation
for current limit resistor, Rlim, is as follows:
where
•
Ilim is the load current at which the current limit comparator will be tripped
(4)
When sensing current across the top FET, replace Rsns with the RDS-ON of the FET. This calculated Rlim value
specifies that the minimum current limit will not be less than Imax. It is recommended that a 1% tolerance resistor
be used.
When sensing across the top FET (VDS sensing), RDS-ON will show more variation than a current-sense resistor,
largely due to temperature variation. RDS-ON will increase proportional to temperature according to a specific
temperature coefficient. Refer to the FET manufacturer's datasheet to determine the range of RDS-ON values over
operating temperature or see the Component Selection section (Equation 27) for a calculation of maximum RDSON. This will prevent RDS-ON variations from prematurely tripping the current limit comparator as the operating
temperature increases.
To ensure accurate current sensing using VDS sensing, special attention in board layout is required. The KSx and
RSNSx pins require separate traces to form a Kelvin connection at the corresponding current sense nodes. In
addition, the filter components R14, R16, C14, C15 should be removed.
INPUT UNDER VOLTAGE LOCKOUT (UVLO)
The input under-voltage lock out threshold, which is sensed via the VLIN5 internal LDO output, is 4.0V (typical).
Below this threshold, both HDRVx and LDRVx will be turned off and the internal 480Ω MOSFETs will be turned
on to discharge the output capacitors through the SWx pins. When the input voltage is below the UVLO
threshold, the ON/SS pins will sink 5mA to discharge the soft start capacitors and turn off both channels. As the
input voltage increases again above 4.0V, UVLO will be de-activated, and the device will restart through a normal
soft start phase. If the voltage at VLIN5 remains below 4.5V, but above the 4.0V UVLO threshold, the device
cannot be ensured to operate within specification.
If the input voltage is between 4.0V and 5.2V, the VLIN5 pin will not regulate, but will follow approximately 200
mV below the input voltage.
18
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DUAL-PHASE PARALLEL OPERATION
In applications with high output current demand, the two switching channels can be configured to operate as a
two phase converter to provide a single output voltage with current sharing between the two switching channels.
This approach greatly reduces the stress and heat on the output stage components while lowering input ripple
current. The inductor ripple currents also cancel to a varying degree which results in lowered output ripple
voltage. Figure 4 shows an example of a typical two-phase circuit. Because precision current sense is the
primary design criteria to ensure accurate current sharing between the two channels, both channels must use
external sense resistors for current sensing. To minimize the error between the error amplifiers of the two
channels, tie the feedback pins FB1 and FB2 together and connect to a single voltage divider for output voltage
sensing. Also, tie the COMP1 and COMP2 together and connect to the compensation network. ON/SS1 and
ON/SS2 must be tied together to enable and disable both channels simultaneously.
EXTERNAL FREQUENCY SYNC
The LM5642 series has the ability to synchronize to external sources in order to set the switching frequency. This
allows the LM5642 to use frequencies from 150 kHz to 250 kHz and the LM5642X to use frequencies from 200
kHz to 500 kHz. Lowering the switching frequency allows a smaller minimum duty cycle, DMIN, and hence a
greater range between input and output voltage. Increasing switching frequency allows the use of smaller output
inductors and output capacitors (see Component Selection). In general, synchronizing all the switching
frequencies in multi-converter systems makes filtering of the switching noise easier.
The sync input can be from a system clock, from another switching converter in the system, or from any other
periodic signal with a logic low-level less than 1.4V and a logic high level greater than 2V. Both CMOS and TTL
level inputs are acceptable.
The LM5642 series uses a fixed delay between Channel 1 and Channel 2. The nominal switching frequency of
200kHz for the LM5642 corresponds to a switching period of 5µs. Channel 2 always turns its high-side switch on
2.5µs after Channel 1 Figure 30 (a). When the converter is synchronized to a frequency other than 200kHz, the
switching period is reduced or increased, while the fixed delay between Channel 1 and Channel 2 remains
constant. The phase difference between channels is therefore no longer 180°. At the extremes of the sync range,
the phase difference drops to 135° Figure 30 (b) and Figure 30 (c). The result of this lower phase difference is a
reduction in the maximum duty cycle of one channel that will not overlap the duty cycle of the other. As shown in
Input Capacitor Selection section, when the duty cycle D1 for Channel 1 overlaps the duty cycle D2 for Channel
2, the input rms current increases, requiring more input capacitors or input capacitors with higher ripple current
ratings. The new, reduced maximum duty cycle can be calculated by multiplying the sync frequency (in Hz) by
2.5x10-6 (the fixed delay in seconds). The same logic applies to the LM5642X. However the LM5642X has a
nominal switching frequency of 375kHz which corresponds to a period of 2.67µs. Therefore channel 2 of the
LM5642X always begins it's period after 1.33µs.
DMAX = FSYNC*2.5x10-6
(5)
At a sync frequency of 150 kHz, for example, the maximum duty cycle for Channel 1 that will not overlap
Channel 2 would be 37.5%. At 250 kHz, it is the duty cycle for Channel 2 that is reduced to a DMAX of 37.5%.
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FSW = 200 kHz
D1
5 Ps
5 Ps
D2
2.5 Ps
(a)
FSW = 150 kHz
D1
6.67 Ps
6.67 Ps
D2
2.5 Ps
(b)
FSW = 250 kHz
D1
4 Ps
4 Ps
D2
2.5 Ps
(c)
Figure 30. Period Fixed Delay Example
Component Selection
OUTPUT VOLTAGE SETTING
The output voltage for each channel is set by the ratio of a voltage divider as shown in Figure 31. The resistor
values can be determined by the following equation:
where
•
Vfb = 1.238V
(6)
Although increasing the value of R1 and R2 will increase efficiency, this will also decrease accuracy. Therefore, a
maximum value is recommended for R2 in order to keep the output within .3% of Vnom. This maximum R2 value
should be calculated first with the following equation:
where
•
20
200nA is the maximum current drawn by FBx pin
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Vout
R2
FBx
R1
GND
Figure 31. Output Voltage Setting
Example: Vnom = 5V, Vfb = 1.2364V, Ifbmax = 200nA.
(8)
Choose 60K
(9)
The Cycle Skip and Dropout modes of the LM5642 series regulate the minimum and maximum output
voltage/duty cycle that the converter can deliver. Both modes check the voltage at the COMP pin. Minimum
output voltage is determined by the Cycle Skip Comparator. This circuitry skips the high side FET ON pulse
when the COMP pin voltage is below 0.5V at the beginning of a cycle. The converter will continue to skip every
other pulse until the duty cycle (and COMP pin voltage) rise above 0.5V, effectively halving the switching
frequency.
Maximum output voltage is determined by the Dropout circuitry, which skips the low side FET ON pulse
whenever the COMP pin voltage exceeds the ramp voltage derived from the current sense. Up to three low side
pulses may be skipped in a row before a minimum on-time pulse must be applied to the low side FET.
Figure 32 shows the range of ouput voltage (for Io = 3A) with respect to input voltage that will keep the converter
from entering either Skip Cycle or Dropout mode.
For input voltages below 5.5V, VLIN5 must be connected to Vin through a small resistor (approximately 4.7
ohm). This will ensure that VLIN5 does not fall below the UVLO threshold.
35
30
VOUT
25
20
15
Operating Region
10
5
0
4
8
12
16
20
24
28
32
36
VIN
Figure 32. Output Voltage Range
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Output Capacitor Selection
In applications that exhibit large, fast load current swings, the slew rate of such a load current transient will likely
be beyond the response speed of the regulator. Therefore, to meet voltage transient requirements during worstcase load transients, special consideration should be given to output capacitor selection. The total combined
ESR of the output capacitors must be lower than a certain value, while the total capacitance must be greater
than a certain value. Also, in applications where the specification of output voltage regulation is tight and ripple
voltage must be low, starting from the required output voltage ripple will often result in fewer design iterations.
ALLOWED TRANSIENT VOLTAGE EXCURSION
The allowed output voltage excursion during a load transient (ΔVc_s) is:
where
•
•
±δ% is the output voltage regulation window
±ε% is the output voltage initial accuracy
(10)
Example: Vnom = 5V, δ% = 7%, ε% = 3.4%, Vrip = 40mV peak to peak.
(11)
MAXIMUM ESR CALCULATION
Unless the rise and fall times of a load transient are slower than the response speed of the control loop, if the
total combined ESR (Re) is too high, the load transient requirement will not be met, no matter how large the
capacitance.
The maximum allowed total combined ESR is:
(12)
Since the ripple voltage is included in the calculation of ΔVc_s, the inductor ripple current should not be included
in the worst-case load current excursion. Simply use the worst-case load current excursion for ΔIc_s.
Example: ΔVc_s = 160 mV, ΔIc_s = 3A. Then Re_max = 53.3 mΩ.
Maximum ESR criterion can be used when the associated capacitance is high enough, otherwise more
capacitors than the number determined by this criterion should be used in parallel.
MINIMUM CAPACITANCE CALCULATION
In a switch mode power supply, the minimum output capacitance is typically dictated by the load transient
requirement. If there is not enough capacitance, the output voltage excursion will exceed the maximum allowed
value even if the maximum ESR requirement is met. The worst-case load transient is an unloading transient that
happens when the input voltage is the highest and when the current switching cycle has just finished. The
corresponding minimum capacitance is calculated as follows:
(13)
Notice it is already assumed the total ESR, Re, is no greater than Re_max, otherwise the term under the square
root will be a negative value. Also, it is assumed that L has already been selected, therefore the minimum L
value should be calculated before Cmin and after Re (see Inductor Selection below). Example: Re = 20 mΩ,
Vnom = 5V, ΔVc_s = 160 mV, ΔIc_s = 3A, L = 8 µH
(14)
Generally speaking, Cmin decreases with decreasing Re, ΔIc_s, and L, but with increasing Vnom and ΔVc_s.
22
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Inductor Selection
The size of the output inductor can be determined from the desired output ripple voltage, Vrip, and the
impedance of the output capacitors at the switching frequency. The equation to determine the minimum
inductance value is as follows:
(15)
In the above equation, Re is used in place of the impedance of the output capacitors. This is because in most
cases, the impedance of the output capacitors at the switching frequency is very close to Re. In the case of
ceramic capacitors, replace Re with the true impedance at the switching frequency.
Example: Vin = 36V, Vo = 3.3V, VRIP = 60 mV, Re = 20 mΩ, F = 200 kHz.
Lmin =
3.3 x 0.02
36 - 3.3
x
= 5PH
200kHz x 36
.060
(16)
The actual selection process usually involves several iterations of all of the above steps, from ripple voltage
selection, to capacitor selection, to inductance calculations. Both the highest and the lowest input and output
voltages and load transient requirements should be considered. If an inductance value larger than Lmin is
selected, make sure that the Cmin requirement is not violated.
Priority should be given to parameters that are not flexible or more costly. For example, if there are very few
types of capacitors to choose from, it may be a good idea to adjust the inductance value so that a requirement of
3.2 capacitors can be reduced to 3 capacitors.
Since inductor ripple current is often the criterion for selecting an output inductor, it is a good idea to doublecheck this value. The equation is:
(17)
Also important is the ripple content, which is defined by Irip /Inom. Generally speaking, a ripple content of less
than 50% is ok. Larger ripple content will cause too much power loss in the inductor.
Example: Vin = 36V, Vo = 3.3V, F = 200 kHz, L = 5 µH, 3A max IOUT
Irip =
36 - 3.3
3.3
x
= 3A
36
200kHz x 5x10-6
(18)
3A is 100% ripple which is too high.
In this case, the inductor should be reselected on the basis of ripple current.
Example: 40% ripple, 40% • 3A = 1.2A
1.2A =
L=
36 - 3.3
3.3
x
L x 200kHz 36
(19)
36 - 3.3
3.3
x
= 12.5PH
200kHz x 1.2A 36
(20)
When choosing the inductor, the saturation current should be higher than the maximum peak inductor current
and the RMS current rating should be higher than the maximum load current.
Input Capacitor Selection
The fact that the two switching channels of the LM5642 are 180° out of phase will reduce the RMS value of the
ripple current seen by the input capacitors. This will help extend input capacitor life span and result in a more
efficient system. Input capacitors must be selected that can handle both the maximum ripple RMS current at
highest ambient temperature as well as the maximum input voltage. In applications in which output voltages are
less than half of the input voltage, the corresponding duty cycles will be less than 50%. This means there will be
no overlap between the two channels' input current pulses.
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The equation for calculating the maximum total input ripple RMS current for duty cycles under 50% is:
where
•
•
•
•
I1 is maximum load current of Channel 1
I2 is the maximum load current of Channel 2
D1 is the duty cycle of Channel 1
D2 is the duty cycle of Channel 2
(21)
Example: Imax_1 = 3.6A, Imax_2 = 3.6A, D1 = 0.42, and D2 = 0.275
(22)
Choose input capacitors that can handle 1.66A ripple RMS current at highest ambient temperature. In
applications where output voltages are greater than half the input voltage, the corresponding duty cycles will be
greater than 50%, and there will be overlapping input current pulses. Input ripple current will be highest under
these circumstances. The input RMS current in this case is given by:
(23)
Where, again, I1 and I2 are the maximum load currents of channel 1 and 2, and D1 and D2 are the duty cycles.
This equation should be used when both duty cycles are expected to be higher than 50%.
If the LM5642 is being used with an external clock frequency other than 200kHz, or 375 kHz for the LM5642X,
the preceding equations for input rms current can still be used. The selection of the first equation or the second
changes because overlap can now occur at duty cycles that are less than 50%. From the EXTERNAL
FREQUENCY SYNC section, the maximum duty cycle that ensures no overlap between duty cycles (and hence
input current pulses) is:
DMAX = FSYNC* 2.5 x 10-6
(24)
There are now three distinct possibilities which must be considered when selecting the equation for input rms
current. The following applies for the LM5642, and also the LM5642X by replacing 200 kHz with 375 kHz:
1. Both duty cycles D1 and D2 are less than DMAX. In this case, the first, simple equation can always be used.
2. One duty cycle is greater than DMAX and the other duty cycle is less than DMAX. In this case, the system
designer can take advantage of the fact that the sync feature reduces DMAX for one channel, but lengthens it
for the other channel. For FSYNC < 200kHz, D1 is reduced to DMAX while D2 actually increases to (1-DMAX).
For FSYNC > 200kHz, D2 is reduced to DMAX while D1 increases to (1-DMAX). By using the channel reduced to
DMAX for the lower duty cycle, and the channel that has been increased for the higher duty cycle, the first,
simple rms input current equation can be used.
3. Both duty cycles are greater than DMAX. This case is identical to a system at 200 kHz where either duty cycle
is 50% or greater. Some overlap of duty cycles is specified, and hence the second, more complicated rms
input current equation must be used.
Input capacitors must meet the minimum requirements of voltage and ripple current capacity. The size of the
capacitor should then be selected based on hold up time requirements. Bench testing for individual applications
is still the best way to determine a reliable input capacitor value. Input capacitors should always be placed as
close as possible to the current sense resistor or the drain of the top FET. When high ESR capacitors such as
tantalum are used, a 1µF ceramic capacitor should be added as closely as possible to the high-side FET drain
and low-side FET source.
24
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MOSFET Selection
BOTTOM FET SELECTION
During normal operation, the bottom FET is switching on and off at almost zero voltage. Therefore, only
conduction losses are present in the bottom FET. The most important parameter when selecting the bottom FET
is the on-resistance (RDS-ON). The lower the on-resistance, the lower the power loss. The bottom FET power loss
peaks at maximum input voltage and load current. The equation for the maximum allowed on-resistance at room
temperature for a given FET package, is:
where
•
•
•
•
Tj_max is the maximum allowed junction temperature in the FET
Ta_max is the maximum ambient temperature
Rθja is the junction-to-ambient thermal resistance of the FET
TC is the temperature coefficient of the on-resistance which is typically in the range of 4000ppm/°C
(25)
If the calculated RDS-ON (MAX) is smaller than the lowest value available, multiple FETs can be used in parallel.
This effectively reduces the Imax term in the above equation, thus reducing RDS-ON. When using two FETs in
parallel, multiply the calculated RDS-ON (MAX) by 4 to obtain the RDS-ON (MAX) for each FET. In the case of three
FETs, multiply by 9.
(26)
If the selected FET has an Rds value higher than 35.3Ω, then two FETs with an RDS-ON less than 141 mΩ (4 x
35.3 mΩ) can be used in parallel. In this case, the temperature rise on each FET will not go to Tj_max because
each FET is now dissipating only half of the total power.
TOP FET SELECTION
The top FET has two types of losses: switching loss and conduction loss. The switching losses mainly consist of
crossover loss and losses related to the low-side FET body diode reverse recovery. Since it is rather difficult to
estimate the switching loss, a general starting point is to allot 60% of the top FET thermal capacity to switching
losses. The best way to precisely determine switching losses is through bench testing. The equation for
calculating the on resistance of the top FET is thus:
(27)
Example: Tj_max = 100°C, Ta_max = 60°C, Rqja = 60°C/W, Vin_min = 5.5V, Vnom = 5V, and Iload_max = 3.6A.
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(28)
When using FETs in parallel, the same guidelines apply to the top FET as apply to the bottom FET.
Loop Compensation
The general purpose of loop compensation is to meet static and dynamic performance requirements while
maintaining stability. Loop gain is what is usually checked to determine small-signal performance. Loop gain is
equal to the product of control-output transfer function and the feedback transfer function (the compensation
network transfer function). Generally speaking it is desirable to have a loop gain slope that is roughly -20dB
/decade from a very low frequency to well beyond the crossover frequency. The crossover frequency should not
exceed one-fifth of the switching frequency. The higher the bandwidth, the faster the load transient response
speed will be. However, if the duty cycle saturates during a load transient, further increasing the small signal
bandwidth will not help. Since the control-output transfer function usually has very limited low frequency gain, it is
a good idea to place a pole in the compensation at zero frequency, so that the low frequency gain will be
relatively large. A large DC gain means high DC regulation accuracy (i.e. DC voltage changes little with load or
line variations). The rest of the compensation scheme depends highly on the shape of the control-output plot.
20
0
Asymptoti
c
GAIN
(dB)
PHASE (°)
-45
0
-90
-20
Phas
e
-135
-40
Gain
-60
10
100
-180
1M
1
10
100
k
k
k
FREQUENCY
(Hz)
Figure 33. Control-Output Transfer Function
GAIN (dB)
As shown in Figure 33, the control-output transfer function consists of one pole (fp), one zero (fz), and a double
pole at fn (half the switching frequency). The following can be done to create a -20dB /decade roll-off of the loop
gain: Place the first pole at 0Hz, the first zero at fp, the second pole at fz, and the second zero at fn. The
resulting feedback transfer function is shown in Figure 34.
-20
dB/
dec
(fp1 is at zero frequency)
-20
dB/
dec
B
fz1
fp2
fz2
FREQUENCY
Figure 34. Feedback Transfer Function
26
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The control-output corner frequencies, and thus the desired compensation corner frequencies, can be
determined approximately by the following equations:
(29)
fP =
1
1 - D - .5
+
2SRO CO 2SfLCO
(30)
Since fp is determined by the output network, it will shift with loading (Ro). It is best to use a minimum Iout value
of approximately 100mA when determining the maximum Ro value.
Example: Re = 20 mΩ, Co = 100 uF, Romax = 5V/100 mA = 50Ω:
(31)
(32)
First determine the minimum frequency (fpmin) of the pole across the expected load range, then place the first
compensation zero at or below that value. Once fpmin is determined, Rc1 should be calculated using:
where
•
•
•
B is the desired gain in V/V at fp (fz1)
gm is the transconductance of the error amplifier
R1 and R2 are the feedback resistors
(33)
A gain value around 10dB (3.3v/v) is generally a good starting point.
Example: B = 3.3v/v, gm = 650m, R1 = 20 kKΩ, R2 = 60.4 kΩ:
(34)
Bandwidth will vary proportional to the value of Rc1. Next, Cc1 can be determined with the following equation:
(35)
Example: fpmin = 995 Hz, Rc1 = 20 kΩ:
(36)
The compensation network (Figure 35) will also introduce a low frequency pole which will be close to 0 Hz.
A second pole should also be placed at fz. This pole can be created with a single capacitor Cc2 and a shorted
Rc2 (see Figure 35). The minimum value for this capacitor can be calculated by:
(37)
Cc2 may not be necessary, however it does create a more stable control loop. This is especially important with
high load currents and in current sharing mode.
Example: fz = 80 kHz, Rc1 = 20 kΩ:
(38)
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A second zero can also be added with a resistor in series with Cc2. If used, this zero should be placed at fn,
where the control to output gain rolls off at -40dB/dec. Generally, fn will be well below the 0dB level and thus will
have little effect on stability. Rc2 can be calculated with the following equation:
(39)
Vo
Vc
CC1
RC1
gm
R2
CC2
RC2
compensation
network
R1
Figure 35. Compensation Network
PCB Layout Considerations
To produce an optimal power solution with the LM5642 series, good layout and design of the PCB are as
important as the component selection. The following are several guidelines to aid in creating a good layout.
KELVIN TRACES FOR SENSE LINES
When using the current sense resistor to sense the load current connect the KS pin using a separate trace to
VIN, as close as possible to the current-sense resistor. The RSNS pin should be connected using a separate
trace to the low-side of the current sense resistor. The traces should be run parallel to each other to give
common mode rejection. Although it can be difficult in a compact design, these traces should stay away from the
output inductor and switch node if possible, to avoid coupling stray flux fields. When a current-sense resistor is
not used the KS pin should be connected as close as possible to the drain node of the upper MOSFET and the
RSNS pin should be connected as close as possible to the source of the upper MOSFET using Kelvin traces. To
further help minimize noise pickup on the sense lines is to use RC filtering on the KS and RSNS pins.
SEPARATE PGND AND SGND
Good layout techniques include a dedicated ground plane, usually on an internal layer. Signal level components
like the compensation and feedback resistors should be connected to a section of this internal SGND plane. The
SGND section of the plane should be connected to the power ground at only one point. The best place to
connect the SGND and PGND is right at the PGND pin..
MINIMIZE THE SWITCH NODE
The plane that connects the power FETs and output inductor together radiates more EMI as it gets larger. Use
just enough copper to give low impedance to the switching currents, preferably in the form of a wide, but short,
trace run.
LOW IMPEDANCE POWER PATH
The power path includes the input capacitors, power FETs, output inductor, and output capacitors. Keep these
components on the same side of the PCB and connect them with thick traces or copper planes (shapes) on the
same layer. Vias add resistance and inductance to the power path, and have relatively high impedance
connections to the internal planes. If high switching currents must be routed through vias and/or internal planes,
use multiple vias in parallel to reduce their resistance and inductance. The power components should be kept
close together. The longer the paths that connect them, the more they act as antennas, radiating unwanted EMI.
Please see AN-1229 (literature number SNVA054) for further PCB layout considerations.
28
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Table 1. Bill Of Materials for Figure 3 24V to 1.8, 3.3V LM5642
ID
Part Number
Type
Size
U1
LM5642
Dual
Synchronous
Controller
TSSOP-28
Q1, Q4
Si4850EY
N-MOSFET
SO-8
Q2, Q5
Si4840DY
N-MOSFET
SO-8
D3
BAS40-06
Schottky Diode
L1
RLF12560T-4R2N100
Inductor
L2
RLF12545T-100M5R1
Inductor
12.5x12.5x 4.5mm
Parameters
Qty
Vendor
1
TI
60V
2
Vishay
40V
2
Vishay
SOT-23
40V
1
Vishay
12.5x12.5x 6mm
4.2µH, 7mΩ 10A
1
TDK
10µH, 12mΩ 5.1A
1
TDK
C1
C3216X7R1H105K
Capacitor
1206
1µF, 50V
1
TDK
C3, C4, C14,
C15
VJ1206Y101KXXAT
Capacitor
1206
100pF, 25V
3
Vishay
C27
C2012X5R1C105K
Capacitor
0805
1µF, 16V
1
TDK
C6, C16
C5750X5R1H106M
Capacitor
2220
10µF 50V, 2.8A
2
TDK
C9, C23
6TPD330M
Capacitor
7.3x4.3x 3.8mm
330µF, 6.3V, 10mΩ
2
Sanyo
C2, C11, C12,
C13
VJ1206Y103KXXAT
Capacitor
1206
10nF, 25V
4
Vishay
C7, C25, C34
VJ1206Y104KXXAT
Capacitor
1206
100nF, 25V
3
Vishay
C19
VJ1206Y822KXXAT
Capacitor
1206
8.2nF 10%
1
Vishay
C20
VJ1206Y153KXXAT
Capacitor
1206
15nF 10%
1
Vishay
C26
C3216X7R1C475K
Capacitor
1206
4.7µF 25V
1
TDK
R1
CRCW1206123J
Resistor
1206
12kΩ 5%
1
Vishay
R2, R6, R14,
R16
CRCW1206100J
Resistor
1206
100Ω 5%
1
Vishay
R13
CRCW1206682J
Resistor
1206
6.8kΩ 12%
1
Vishay
R7, R15
WSL-2512 .010 1%
Resistor
2512
10mΩ 1W
2
Vishay
R8, R9, R12,
R17, R18, R21,
R31, R32
CRCW1206000Z
Resistor
1206
0Ω
8
Vishay
R10
CRCW12062261F
Resistor
1206
2.26kΩ 1%
1
Vishay
R23
CRCW12068451F
Resistor
1206
8.45kΩ 1%
1
Vishay
R24
CRCW12061372F
Resistor
1206
13.7kΩ 1%
1
Vishay
R11, R20
CRCW12064991F
Resistor
1206
4.99kΩ 1%
2
Vishay
R19
CRCW12068251F
Resistor
1206
8.25kΩ 1%
1
Vishay
R27
CRCW12064R7J
Resistor
1206
4.7Ω 5%
1
Vishay
R28
CRCW1206224J
Resistor
1206
220kΩ 5%
1
Vishay
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Table 2. Bill of Materials for Figure 4 30V to 1.8V, 20A LM5642
ID
Part Number
Type
Size
U1
LM5642
Dual
Synchronou
s Controller
TSSOP-28
Parameters
Qty
Vendor
1
TI
Q1, Q4
Si4850EY
N-MOSFET
SO-8
60V
2
Vishay
Q2, Q3, Q5, Q6
Si4470DY
N-MOSFET
SO-8
60V
4
Vishay
D3
BAS40-06
Schottky
Diode
SOT-23
40V
1
Vishay
L1,L2
RLF12560T-2R7N110
Inductor
12.5x12.5x 6mm
2.7µH,4.5mΩ 11.5A
2
TDK
C1
C3216X7R1H105K
Capacitor
1206
1µF, 50V
1
TDK
C10, C24, C27
C2012X5R1C105K
Capacitor
0805
1µF, 16V
3
TDK
C6, C16, C28,
C30
C5750X5R1H106M
Capacitor
2220
10µF 50V, 2.8A
4
TDK
C9, C23
16MV1000WX
Capacitor
10mm D20mm H
1000µF, 16V, 22mΩ
2
Sanyo
C2, C13
VJ1206Y103KXXAT
Capacitor
1206
10nF, 25V
2
Vishay
C11
VJ1206Y223KXXAT
Capacitor
1206
22nF, 25V
1
Vishay
C7,C25, C34
VJ1206Y104KXXAT
Capacitor
1206
100nF, 25V
3
Vishay
C19
VJ1206Y273KXXAT
Capacitor
1206
27nF 10%
1
Vishay
C26
C3216X7R1C475K
Capacitor
1206
4.7µF 25V
1
TDK
R1, R13
CRCW1206123J
Resistor
1206
16.9kΩ 1%
1
Vishay
R2, R6, R14,
R16
CRCW1206100J
Resistor
1206
100Ω 5%
1
Vishay
R7, R15
WSL-2512 .010 1%
Resistor
2512
10mΩ 1W
2
Vishay
R8, R9, R12,
R17, R18, R21,
R31, R32
CRCW1206000Z
Resistor
1206
0Ω
8
Vishay
R10
CRCW12062261F
Resistor
1206
2.26kΩ 1%
1
Vishay
R11
CRCW12064991F
Resistor
1206
4.99kΩ 1%
1
Vishay
R23
CRCW12061152F
Resistor
1206
11.5kΩ 1%
1
Vishay
R27
CRCW12064R7J
Resistor
1206
4.7Ω 5%
1
Vishay
R28
CRCW1206224J
Resistor
1206
220kΩ 5%
1
Vishay
30
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Table 3. Bill Of Materials Based on Figure 3 Vin= 9-16V, VO1,2=1.5V,1.8V, 5A LM5642X
ID
Part Number
Type
Size
U1
LM5642X
Dual
Synchronous
Controller
TSSOP-28
Q1, Q4
Si4850EY
N-MOSFET
SO-8
Q2, Q5
Si4840DY
N-MOSFET
SO-8
Parameters
Qty
Vendor
1
TI
60V
2
Vishay
40V
2
Vishay
D3
BA54A
Schottky Diode
SOT-23
30V
1
Vishay
L1, L2
RLF12545T-4R2N100
Inductor
12.5x12.5x 4.5mm
4.2µH, 7mΩ 6.5A
2
TDK
C1
C3216X7R1H105K
Capacitor
1206
1µF, 50V
1
TDK
C3, C4, C14,
C15
VJ1206Y101KXXAT
Capacitor
1206
100pF, 25V
4
Vishay
C27
C2012X5R1C105K
Capacitor
0805
1µF, 16V
1
TDK
C6, C28
C5750X7R1H106M
Capacitor
2220
10µF 50V, 2.8A
2
TDK
C9, C23
C4532X7R0J107M
Capacitor
1812
100µF, 6.3V, 1mΩ
2
TDK
C2, C11, C12,
C13
VJ1206Y103KXXAT
Capacitor
1206
10nF, 25V
4
Vishay
C7, C25, C34
VJ1206Y104KXXAT
Capacitor
1206
100nF, 25V
3
Vishay
C18, C20
VJ1206Y473KXXAT
Capacitor
1206
47nF 10%
2
Vishay
C26
C3216X7R1C475K
Capacitor
1206
4.7µF 25V
1
TDK
R1, R13
CRCW12061912F
Resistor
1206
19.1kΩ 1%
2
Vishay
R2, R6, R14,
R16
CRCW1206100J
Resistor
1206
100Ω 5%
1
Vishay
R7, R15
WSL-1206 .020 1%
Resistor
1206
20mΩ 1W
2
Vishay
R8, R9, R12,
R17, R18, R21,
R31, R32
CRCW1206000Z
Resistor
1206
0Ω
8
Vishay
R10, R19
CRCW12061001F
Resistor
1206
1kΩ 1%
2
Vishay
R11
CRCW12062611F
Resistor
1206
2.61kΩ 1%
1
Vishay
R20
CRCW12062321F
Resistor
1206
2.32kΩ 1%
1
Vishay
R22, R24
CRCW12063011F
Resistor
1206
3.01kΩ 1%
2
Vishay
R27
CRCW12064R7J
Resistor
1206
4.7Ω 5%
1
Vishay
R28
CRCW1206224J
Resistor
1206
220kΩ 5%
1
Vishay
Submit Documentation Feedback
Copyright © 2004–2011, Texas Instruments Incorporated
Product Folder Links: LM5642 LM5642X
31
LM5642, LM5642X
SNVS219J – MAY 2004 – REVISED MAY 2011
www.ti.com
Table 4. Bill Of Materials Based on Figure 3 Vin= 9-16V, VO1,2=3.3V,5V, 5A LM5642X
ID
Part Number
Type
Size
U1
LM5642X
Dual
Synchronous
Controller
TSSOP-28
Q1, Q4
Si4850EY
N-MOSFET
SO-8
Q2, Q5
Si4840DY
N-MOSFET
SO-8
Parameters
Qty
Vendor
1
TI
60V
2
Vishay
40V
2
Vishay
D3
BA54A
Schottky Diode
SOT-23
30V
1
Vishay
L1, L2
RLF12545T-5R6N6R1
Inductor
12.5x12.5x 4.5mm
5.6µH, 9mΩ 6.1A
2
TDK
C1
C3216X7R1H105K
Capacitor
1206
1µF, 50V
1
TDK
C3, C4, C14,
C15
VJ1206Y101KXXAT
Capacitor
1206
100pF, 25V
4
Vishay
C27
C2012X5R1C105K
Capacitor
0805
1µF, 16V
1
TDK
C6, C28
C5750X7R1H106M
Capacitor
2220
10µF 50V, 2.8A
2
TDK
C9, C23
C4532X7R0J107M
Capacitor
1812
100µF, 6.3V, 1mΩ
2
TDK
C2, C11, C12,
C13
VJ1206Y103KXXAT
Capacitor
1206
10nF, 25V
4
Vishay
C7, C25, C34
VJ1206Y104KXXAT
Capacitor
1206
100nF, 25V
3
Vishay
C18, C20
VJ1206Y393KXXAT
Capacitor
1206
39nF 10%
2
Vishay
C26
C3216X7R1C475K
Capacitor
1206
4.7µF 25V
1
TDK
R1, R13
CRCW12061912F
Resistor
1206
19.1kΩ 1%
2
Vishay
R2, R6, R14,
R16
CRCW1206100J
Resistor
1206
100Ω 5%
1
Vishay
R7, R15
WSL-1206 .020 1%
Resistor
1206
20mΩ 1W
2
Vishay
R8, R9, R12,
R17, R18, R21,
R31, R32
CRCW1206000Z
Resistor
1206
0Ω
8
Vishay
R10, R19
CRCW12061002F
Resistor
1206
10kΩ 1%
2
Vishay
R11
CRCW12066191F
Resistor
1206
6.19kΩ 1%
1
Vishay
R20
CRCW12063321F
Resistor
1206
3.32kΩ 1%
1
Vishay
R22, R24
CRCW12063831F
Resistor
1206
3.83kΩ 1%
2
Vishay
R27
CRCW12064R7J
Resistor
1206
4.7Ω 5%
1
Vishay
R28
CRCW1206224J
Resistor
1206
220kΩ 5%
1
Vishay
32
Submit Documentation Feedback
Copyright © 2004–2011, Texas Instruments Incorporated
Product Folder Links: LM5642 LM5642X
PACKAGE OPTION ADDENDUM
www.ti.com
9-Mar-2013
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package Qty
Drawing
Eco Plan
Lead/Ball Finish
(2)
MSL Peak Temp
Op Temp (°C)
Top-Side Markings
(3)
(4)
LM5642MH/NOPB
ACTIVE
HTSSOP
PWP
28
48
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
LM5642
MH
LM5642MHX/NOPB
ACTIVE
HTSSOP
PWP
28
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
LM5642
MH
LM5642MTC
ACTIVE
TSSOP
PW
28
48
TBD
Call TI
Call TI
-40 to 125
LM5642
MTC
LM5642MTC/NOPB
ACTIVE
TSSOP
PW
28
48
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
LM5642
MTC
LM5642MTCX
ACTIVE
TSSOP
PW
28
2500
TBD
Call TI
Call TI
-40 to 125
LM5642
MTC
LM5642MTCX/NOPB
ACTIVE
TSSOP
PW
28
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
LM5642
MTC
LM5642XMH
ACTIVE
HTSSOP
PWP
28
48
TBD
Call TI
Call TI
-40 to 125
LM5642
XMH
LM5642XMH/NOPB
ACTIVE
HTSSOP
PWP
28
48
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
LM5642
XMH
LM5642XMHX
ACTIVE
HTSSOP
PWP
28
2500
TBD
Call TI
Call TI
-40 to 125
LM5642
XMH
LM5642XMHX/NOPB
ACTIVE
HTSSOP
PWP
28
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
LM5642
XMH
LM5642XMT
ACTIVE
TSSOP
PW
28
48
TBD
Call TI
Call TI
-40 to 125
LM5642
XMT
LM5642XMT/NOPB
ACTIVE
TSSOP
PW
28
48
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
LM5642
XMT
LM5642XMTX
ACTIVE
TSSOP
PW
28
2500
TBD
Call TI
Call TI
-40 to 125
LM5642
XMT
LM5642XMTX/NOPB
ACTIVE
TSSOP
PW
28
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
LM5642
XMT
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
9-Mar-2013
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
Only one of markings shown within the brackets will appear on the physical device.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
17-Nov-2012
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
LM5642MHX/NOPB
HTSSOP
PWP
28
2500
330.0
16.4
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
6.8
10.2
1.6
8.0
16.0
Q1
LM5642MTCX
TSSOP
PW
28
2500
330.0
16.4
6.8
10.2
1.6
8.0
16.0
Q1
LM5642MTCX/NOPB
TSSOP
PW
28
2500
330.0
16.4
6.8
10.2
1.6
8.0
16.0
Q1
LM5642XMHX
HTSSOP
PWP
28
2500
330.0
16.4
6.8
10.2
1.6
8.0
16.0
Q1
LM5642XMHX/NOPB
HTSSOP
PWP
28
2500
330.0
16.4
6.8
10.2
1.6
8.0
16.0
Q1
LM5642XMTX
TSSOP
PW
28
2500
330.0
16.4
6.8
10.2
1.6
8.0
16.0
Q1
LM5642XMTX/NOPB
TSSOP
PW
28
2500
330.0
16.4
6.8
10.2
1.6
8.0
16.0
Q1
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
17-Nov-2012
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
LM5642MHX/NOPB
HTSSOP
PWP
28
2500
349.0
337.0
45.0
LM5642MTCX
TSSOP
PW
28
2500
358.0
343.0
63.0
LM5642MTCX/NOPB
TSSOP
PW
28
2500
358.0
343.0
63.0
LM5642XMHX
HTSSOP
PWP
28
2500
349.0
337.0
45.0
LM5642XMHX/NOPB
HTSSOP
PWP
28
2500
349.0
337.0
45.0
LM5642XMTX
TSSOP
PW
28
2500
358.0
343.0
63.0
LM5642XMTX/NOPB
TSSOP
PW
28
2500
358.0
343.0
63.0
Pack Materials-Page 2
MECHANICAL DATA
PWP0028A
MXA28A (Rev D)
www.ti.com
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