AD AD548KN Precision, low power bifet op amp Datasheet

a
FEATURES
Enhanced Replacement for LF441 and TL061
DC Performance:
200 mA max Quiescent Current
10 pA max Bias Current, Warmed Up (AD548C)
250 mV max Offset Voltage (AD548C)
2 mV/8C max Drift (AD548C)
2 mV p-p Noise, 0.1 Hz to 10 Hz
AC Performance:
1.8 V/ms Slew Rate
1 MHz Unity Gain Bandwidth
Available in Plastic, Hermetic Cerdip and Hermetic
Metal Can Packages and in Chip Form
Available in Tape and Reel in Accordance with
EIA-481A Standard
MIL-STD-883B Parts Available
Dual Version Available: AD648
Surface Mount (SOIC) Package Available
PRODUCT DESCRIPTION
The AD548 is a low power, precision monolithic operational
amplifier. It offers both low bias current (10 pA max, warmed
up) and low quiescent current (200 µA max) and is fabricated
with ion-implanted FET and laser wafer trimming technologies.
Input bias current is guaranteed over the AD548’s entire
common-mode voltage range.
The economical J grade has a maximum guaranteed input offset
voltage of less than 2 mV and an input offset voltage drift of less
than 20 µV/°C. The C grade reduces input offset voltage to less
than 0.25 mV and offset voltage drift to less than 2 µV/°C. This
level of dc precision is achieved utilizing Analog’s laser wafer
drift trimming process. The combination of low quiescent current and low offset voltage drift minimizes changes in input offset voltage due to self-heating effects. Four additional grades are
offered over the commercial, industrial and military temperature
ranges.
The AD548 is recommended for any dual supply op amp application requiring low power and excellent dc and ac performance. In applications such as battery-powered, precision
instrument front ends and CMOS DAC buffers, the AD548’s
excellent combination of low input offset voltage and drift, low
bias current and low 1/f noise reduces output errors. High common-mode rejection (86 dB, min on the “C” grade) and high
open-loop gain ensures better than 12-bit linearity in high impedance, buffer applications.
The AD548 is pinned out in a standard op amp configuration
and is available in six performance grades. The AD548J and
AD548K are rated over the commercial temperature range of
0°C to +70°C. The AD548A, AD548B and AD548C are rated
Precision, Low Power
BiFET Op Amp
AD548
CONNECTION DIAGRAMS
Plastic Mini-DIP (N) Package,
Cerdip (Q) Package and
SOIC (R)Package
OFFSET NULL 1
8 NC
INVERTING
2
INPUT
NONINVERTING 3
INPUT
V– 4
7 V+
6 OUTPUT
5 OFFSET
NULL
AD548
TOP VIEW
10kΩ
TO-99 (H) Package
4
NC
OFFSET NULL
AD548
V+
6
3
TOP VIEW
7
INVERTING 2
INPUT
NONINVERTING
INPUT
–15V
VOS TRIM
8
1
5
1
5
4
OUTPUT
OFFSET
NULL
V–
NOTE : PIN 4 CONNECTED TO CASE
NC = NO CONNECT
over the industrial temperature range of –40°C to +85°C. The
AD548S is rated over the military temperature range of –55°C
to +125°C and is available processed to MIL-STD-883B, Rev. C.
The AD548 is available in an 8-pin plastic mini-DIP, cerdip,
TO-99 metal can, surface mount (SOIC), or in chip form.
PRODUCT HIGHLIGHTS
1. A combination of low supply current, excellent dc and ac
performance and low drift makes the AD548 the ideal op
amp for high performance, low power applications.
2. The AD548 is pin compatible with industry standard op
amps such as the LF441, TL061, and AD542, enabling designers to improve performance while achieving a reduction
in power dissipation of up to 85%.
3. Guaranteed low input offset voltage (2 mV max) and drift
(20 µV/°C max) for the AD548J are achieved utilizing
Analog Devices’ laser drift trimming technology, eliminating
the need for external trimming.
4. Analog Devices specifies each device in the warmed-up condition, insuring that the device will meet its published specifications in actual use.
5. A dual version, the AD648 is also available.
6. Enhanced replacement for LF441 and TL061.
REV. B
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700
Fax: 617/326-8703
AD548–SPECIFICATIONS (@ +258C and V = 615 V dc unless otherwise noted)
S
Model
Min
INPUT OFFSET VOLTAGE 1
Initial Offset
TMIN to TMAX
vs. Temperature
vs. Supply
vs. Supply, TMIN to TMAX
Long-Term Offset Stability
AD548J/A/S
Typ
Max
0.75
2.0
3.0/3.0/3.0
20
80
76/76/76
Max
5
Min
0.5
0.7/0.8
5
86
80
5
INPUT IMPEDANCE
Differential
Common Mode
AD548K/B
Typ
0.3
15
INPUT BIAS CURRENT
Either Input 2, VCM = 0
Either Input 2 at TMAX, VCM = 0
Max Input Bias Current Over
Common-Mode Voltage Range
Offset Current, VCM = 0
Offset Current at T MAX
INPUT VOLTAGE RANGE
Differential3
Common Mode
Common-Mode Rejection
VCM = ± 10 V
TMIN to TMAX
VCM = ± 11 V
TMIN to TMAX
Min
0.10
3
30
10
0.25/0.65/10
2
1 × 1012i3
3 × 1012i3
Max
Units
0.25
0.4
2.0
mV
mV
µV/°C
dB
dB
µV/Month
10
0.65
pA
nA
15
5
0.35
pA
pA
nA
86
80
15
20
0.45/1.3/20
AD548C
Typ
15
10
0.25/0.65
3
15
5
0.15/0.35
2
1 × 1012i3
3 × 1012i3
1 × 1012i3
3 × 1012i3
ΩipF
ΩipF
± 11
± 20
± 12
± 11
± 20
± 12
± 11
± 20
± 12
V
V
76
76/76/76
70
70/70/70
90
90
84
84
82
82
76
76
92
92
86
86
86
86
76
76
98
98
90
90
dB
dB
dB
dB
INPUT VOLTAGE NOISE
Voltage 0.1 Hz to 10 Hz
f = 10 Hz
f = 100 Hz
f = 1 kHz
f = 10 kHz
2
80
40
30
30
2
80
40
30
30
2
80
40
30
30
INPUT CURRENT NOISE
f = 1 kHz
1.8
1.8
1.8
fA/√Hz
1.0
30
1.8
8
MHz
kHz
V/µs
µs
FREQUENCY RESPONSE
Unity Gain, Small Signal
Full Power Response
Slew Rate, Unity Gain
Settling Time to ± 0.01%
1.0
30
1.8
8
0.8
300
300/300/300
150
150/150/150
1000
700
500
300
OUTPUT CHARACTERISTICS
Voltage @ RL ≥ 10 kΩ,
TMIN to TMAX
Voltage @ RL ≥ 5 kΩ,
TMIN to TMAX
Short Circuit Current
± 12
± 12/± 12/± 12
± 11
± 11/± 11/± 11
± 13
POWER SUPPLY
Rated Performance
Operating Range
Quiescent Current
± 4.5
OPEN LOOP GAIN
VO = ± 10 V, RL ≥ 10 kΩ
TMIN to TMAX, RL ≥ 10 kΩ
VO = ± 10 V, RL ≥ 5 kΩ
TMIN to TMAX, RL ≥ 5 kΩ
TEMPERATURE RANGE
Operating, Rated Performance
Commercial (0°C to +70°C)
Industrial (–40°C to +85°C)
Military (–55°C to +125°C)
PACKAGE OPTIONS
SOIC (R-8)
Plastic (N-8)
Cerdip (Q-8)
Metal Can (H-08A)
Tape and Reel
Chips Available
0.8
1.0
0.8
300
300
150
150
1000
700
500
300
300
300
150
150
1000
700
500
300
V/mV
V/mV
V/mV
V/mV
± 12
± 12
± 11
± 11
± 13
± 12
± 12
± 11
± 11
± 13
V
± 12.3
V
15
mA
15
± 15
170
± 4.5
± 12.3
± 15
170
AD548J
AD548A
AD548S
AD548JR
AD548JN
AD548AQ
AD548AH
AD548JR-REEL
AD548JCHIPS
1.0
15
± 18
200
µV p-p
nV/√Hz
nV/√Hz
nV/√Hz
nV/√Hz
1.0
30
1.8
8
1.0
± 12.3
4.0
± 18
200
AD548K
AD548B
± 4.5
± 15
170
± 18
200
V
V
µA
AD548C
AD548KR, AD548BR
AD548KN
AD548CQ
AD548BH
AD548KR-REEL, AD548BR-REEL
NOTES
1
Input Offset Voltage specifications are guaranteed after 5 minutes of operation at TA = +25°C.
2
Bias Current specifications are guaranteed maximum at either input after 5 minutes of operation at TA = +25°C. For higher temperature, the current doubles every 10°C.
3
Defined as voltages between inputs, such that neither exceeds ± 10 V from ground.
Specifications subject to change without notice.
–2–
REV. C
AD548
ABSOLUTE MAXIMUM RATINGS l
METALIZATION PHOTOGRAPH
Dimensions shown in inches and (mm).
Contact factory for latest dimensions
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .± 18 V
Internal Power Dissipation2 . . . . . . . . . . . . . . . . . . . . 500 mW
Input Voltage3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .± 18 V
Output Short Circuit Duration . . . . . . . . . . . . . . . . . Indefinite
Differential Input Voltage . . . . . . . . . . . . . . . . . . +VS and –VS
Storage Temperature Range (Q, H) . . . . . . . . –65°C to +150°C
(N, R) . . . . . . . . –65°C to +125°C
Operating Temperature Range
AD548J/K . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to +70°C
AD548A/B/C . . . . . . . . . . . . . . . . . . . . . . . . –40°C to +85°C
AD548S . . . . . . . . . . . . . . . . . . . . . . . . . . . –55°C to +125°C
Lead Temperature Range (Soldering 60 sec) . . . . . . . . +300°C
NOTES
1
Stresses above those listed under “Absolute Maximum Ratings” may cause
permanent damage to the device. This is a stress rating only and functional
operation of the device at these or any other conditions above those indicated in the
operational sections of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
2
Thermal Characteristics: 8-Pin SOIC Package: θJA = 160°C/W, θJC = 42°C/W;
8-Pin Plastic Package: θJA = 90°C/W; 8-Pin Cerdip Package: θJC = 22°C/W, θJA =
110°C/W; 8-Pin Metal Can Package: θJC = 65°C/W, θJA = 150°C/W.
3
For supply voltages less than ± 18 V, the absolute maximum input voltage is equal
to the supply voltage.
Typical Characteristics
20
20
30
15
+VIN
10
–VIN
5
0
5
10
15
SUPPLY VOLTAGE – ±V
–VOUT
10
25°C
RL = 10k
5
0
20
5
10
15
SUPPLY VOLTAGE – ±V
20
INPUT BIAS CURRENT – pA
180
160
140
20
15
10
5
10
100
1k
LOAD RESISTANCE – Ω
10k
Figure 3. Output Voltage Swing
vs. Load Resistance
10
200
25
0
0
Figure 2. Output Voltage Swing
vs. Supply Voltage
Figure 1. Input Voltage Range
vs. Supply Voltage
QUIESCENT CURRENT – µA
15
100nA
10nA
8
INPUT BIAS CURRENT
0
OUTPUT VOLTAGE SWING – Volts p-p
OUTPUT VOLTAGE SWING – ±V
INPUT VOLTAGE – ±V
+VOUT
6
4
1nA
100pA
10pA
1pA
2
100fA
120
0
0
5
10
15
SUPPLY VOLTAGE – ±V
20
Figure 4. Quiescent Current vs.
Supply Voltage
0
4
8
12
16
SUPPLY VOLTAGE – ±V
20
Figure 5. Input Bias Current
vs. Supply Voltage
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD548 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
REV. C
–3–
10fA
–55
–25
5
35
65
TEMPERATURE – °C
95
125
Figure 6. Input Bias Current vs.
Temperature
WARNING!
ESD SENSITIVE DEVICE
AD548–Typical Characteristics
30
1500
25
1250
OPEN LOOP GAIN – V/mV
10
INPUT BIAS CURRENT – pA
8
I∆VOSI – µV
20
6
4
15
10
2
750
500
250
0
–6
–2
2
6
10
0
10
100
60
GAIN
40
20
20
0
0
–20
–20
10k
100k
1M
PHASE IN DEGREES
80
60
1k
50
60
0
–55
70
–25
5
–40
10M
100
90
80
70
125
Figure 9. Open Loop Gain vs.
Temperature
2
4
6
8 10 12 14
SUPPLY VOLTAGE – ±V
16
+SUPPLY
100
80
60
40
–SUPPLY
20
0
100
18
1k
10k
100k
FREQUENCY – Hz
1M
Figure 12. PSRR vs. Frequency
Figure 11. Open Loop Voltage Gain
vs. Supply Voltage
22
90
95
–20
0
FREQUENCY – Hz
Figure 10. Open Loop Frequency
Response
65
120
110
60
35
TEMPERATURE – °C
POWER SUPPLY REJECTION – dB
PHASE
–40
40
120
OPEN LOOP VOLTAGE GAIN – dB
100
40
30
Figure 8. Change in Offset Voltage
vs. Warm-Up Time
Figure 7. Input Bias Current vs.
Common-Mode Voltage
80
20
WARM-UP TIME – Seconds
COMMON-MODE VOLTAGE – V
OPEN LOOP GAIN – dB
1000
5
0
–10
RL = 10k
10
10mV
OUTPUT VOLTAGE – V p-p
80
CMRR – dB
70
60
50
40
OUTPUT VOLTAGE SWING – V
20
18
16
14
12
10
8
6
4
1mV
5
0
–5
1mV
30
2
0
20
1k
10k
100k
FREQUENCY – Hz
–10
10
1M
100
1k
10k
100k
0
1M
0.1
0.01
UNITY GAIN
FOLLOWER
0.001
1k
10k
FREQUENCY – Hz
Figure 16. Total Harmonic
Distortion vs. Frequency
100k
140
INPUT NOISE VOLTAGE – µV p-p
INPUT NOISE VOLTAGE – nV/√Hz
FOLLOWER
WITH GAIN = 10
120
100
80
60
40
20
0
10
100
1k
10k
100k
FREQUENCY – Hz
Figure 17. Input Noise Voltage
Spectral Density
–4–
6
8
Figure 15. Output Swing and Error
Voltage vs. Output Settling Time
160
1
4
SETTLING TIME – µs
Figure 14. Large Signal Frequency
Response
4
100
2
FREQUENCY – Hz
Figure 13. CMRR vs. Frequency
TOTAL HARMONIC DISTORTION – %
10mV
WHENEVER JOHNSON NOISE IS GREATER THAN
AMPLIFIER NOISE, AMPLIFIER NOISE CAN BE
CONSIDERED NEGLIGIBLE FOR APPLICATION
10,000
1,000
1kHz BANDWIDTH
RESISTOR JOHNSON
NOISE
100
10
10Hz
BANDWIDTH
1
AMPLIFIER GENERATED NOISE
0
100k
1M
10M
100M
1G
10G
100G
SOURCE IMPEDANCE – Ω
Figure 18. Total Noise vs. Source
Impedance
REV. C
Typical Characteristics–AD548
Figure 19a. Unity Gain Follower
Figure 20a. Utility Gain Inverter
Figure 19b. Unity Gain Follower
Pulse Response (Large Signal)
Figure 20b. Utility Gain Inverter
Pulse Response (Large Signal)
APPLICATION NOTES
The AD548 is a JFET-input op amp with a guaranteed maximum IB of less than 10 pA, and offset and drift laser-trimmed to
0.25 mV and 2 µV/°C respectively (AD548C). AC specs include 1 MHz bandwidth, 1.8 V/µs typical slew rate and 8 µs settling time for a 20 V step to ± 0.01%—all at a supply current less
than 200 µA. To capitalize on the device’s performance, a number of error sources should be considered.
The minimal power drain and low offset drift of the AD548
reduce self-heating or “warm-up” effects on input offset voltage,
making the AD548 ideal for on/off battery powered applications. The power dissipation due to the AD548’s 200 µA supply
current has a negligible effect on input current, but heavy output loading will raise the chip temperature. Since a JFET’s input current doubles for every 10°C rise in chip temperature, this
can be a noticeable effect.
Figure 19c. Unity Gain Follower
Pulse Response (Small Signal)
Figure 20c. Unity Gain Inverter
Pulse Response (Small Signal)
Applying the AD548
Figure 21. Offset Null Configuration
LAYOUT
OFFSET NULLING
To take full advantage of the AD548’s 10 pA max input current,
parasitic leakages must be kept below an acceptable level. The
practical limit of the resistance of epoxy or phenolic circuit
board material is between 1 × 1012 Ω and 3 × 1012 Ω. This can
result in an additional leakage of 5 pA between an input of 0 V
and a –15 V supply line. Teflon or a similar low leakage material
(with a resistance exceeding 1017 Ω) should be used to isolate
high impedance input lines from adjacent lines carrying high
voltages. The insulator should be kept clean, since contaminants
will degrade the surface resistance.
Unlike bipolar input amplifiers, zeroing the input offset voltage
of a BiFET op amp will not minimize offset drift. Using balance
Pins 1 and 5 to adjust the input offset voltage as shown in Figure 21 will induce an added drift of 0.24 µV/°C per 100 µV of
nulled offset. The low initial offset (0.25 mV) of the AD548C
results in only 0.6 µV/°C of additional drift.
A metal guard completely surrounding the high impedance
nodes and driven by a voltage near the common-mode input potential can also be used to reduce some parasitic leakages. The
guarding pattern in Figure 22 will reduce parasitic leakage due
to finite board surface resistance; but it will not compensate for
a low volume resistivity board.
The amplifier is designed to be functional with power supply
voltages as low as ± 4.5 V. It will exhibit a higher input offset
voltage than at the rated supply voltage of ± 15 V, due to power
supply rejection effects. The common-mode range of the
AD548 extends from 3 V more positive than the negative supply
to 1 V more negative than the positive supply. Designed to
cleanly drive up to 10 kΩ and 100 pF loads, the AD548 will
drive a 2 kΩ load with reduced open loop gain.
REV. C
–5–
AD548
Figure 22. Board Layout for Guarding Inputs
Figure 24. AD548 Used as DAC Output Amplifier
That is:
INPUT PROTECTION
The AD548 is guaranteed to withstand input voltages equal to
the power supply potential. Exceeding the negative supply voltage on either input will forward bias the substrate junction of
the chip. The induced current may destroy the amplifier due to
excess heat.
 R 
VOS Output =VOS Input 1+ FB 
RO 

RFB is the feedback resistor for the op amp, which is internal to
the DAC. RO is the DAC’s R-2R ladder output resistance. The
value of RO is code dependent. This has the effect of changing
the offset error voltage at the amplifier’s output. An output amplifier with a sub millivolt input offset voltage is needed to
preserve the linearity of the DAC’s transfer function.
Input protection is required in applications such as a flame
detector in a gas chromatograph, where a very high potential
may be applied to the input terminals during a sensor fault condition. Figure 23 shows a simple current limiting scheme that
can be used. RPROTECT should be chosen such that the maximum overload current is 1.0 mA (l00 kΩ for a 100 V overload,
for example).
The AD548 in this configuration provides a 700 kHz small signal bandwidth and 1.8 V/µs typical slew rate. The 33 pF capacitor across the feedback resistor optimizes the circuit’s response.
The oscilloscope photos in Figures 25 and 26 show small and
large signal outputs of the circuit in Figure 24. Upper traces
show the input signal VIN. Lower traces are the resulting output
voltage with the DAC’s digital input set to all 1s. The AD548
settles to ± 0.01% for a 20 V input step in 14 µs.
Exceeding the negative common-mode range on either input
terminal causes a phase reversal at the output, forcing the
amplifier output to the corresponding high or low state. Exceeding the negative common-mode on both inputs simultaneously
forces the output high. Exceeding the positive common-mode
range on a single input doesn’t cause a phase reversal, but if
both inputs exceed the limit the output will be forced high. In
all cases, normal amplifier operation is resumed when input
voltages are brought back within the common-mode range.
5V
20V
5µS
100
90
10
0%
Figure 25. Response to ± 20 V p-p Reference Square Wave
50mV
Figure 23. Input Protection of IV Converter
200mV
2µS
100
90
D/A CONVERTER OUTPUT BUFFER
The circuit in Figure 24 shows the AD548 and AD7545 12-bit
CMOS D/A converter in a unipolar binary configuration. VOUT
will be equal to VREF attenuated by a factor depending on the
digital word. VREF sets the full scale. Overall gain is trimmed by
adjusting RIN. The AD548’s low input offset voltage, low drift
and clean dynamics make it an attractive low power output
buffer.
10
0%
Figure 26. Response to ± 100 mV p-p Reference Square
Wave
The input offset voltage of the AD548 output amplifier results
in an output error voltage. This error voltage equals the input
offset voltage of the op amp times the noise gain of the
amplifier.
–6–
REV. C
Application Hints–AD548
PHOTODIODE PREAMP
The performance of the photodiode preamp shown in Figure 27
is enhanced by the AD548’s low input current, input voltage
offset and offset voltage drift. The photodiode sources a current
proportional to the incident light power on its surface. RF converts
the photodiode current to an output voltage equal to RF × IS.
Figure 27.
An error budget illustrating the importance of low amplifier
input current, voltage offset and offset voltage drift to minimize
output voltage errors can be developed by considering the equivalent circuit for the small (0.2 mm2 area) photodiode shown in
Figure 27. The input current results in an error proportional to
the feedback resistance used. The amplifier’s offset will produce
an error proportional to the preamp’s noise gain (I + RF/RSH),
where RSH is the photodiode shunt resistance. The amplifier’s
input current will double with every 10°C rise in temperature,
and the photodiode’s shunt resistance halves with every 10°C
rise. The error budget in Figure 28 assumes a room temperature
photodiode RSH of 500 MΩ, and the maximum input current
and input offset voltage specs of an AD548C.
TEMP
8C
RSH (MV)
VOS (mV) (1+ RF/RSH) VOS
–25
0
+25
+50
+75
+85
15,970
2,830
500
88.5
15.6
7.8
150
200
250
300
350
370
151 µV
207 µV
300 µV
640 µV
2.6 mV
5.1 mV
IB (pA)
IBRF
TOTAL
0.30
2.26
10.00
56.6
320
640
30 µV
262 µV
1.0 mV
5.6 mV
32 mV
64 mV
181 µV
469 µV
1.30 mV
6.24 mV
34.6 mV
69.1 mV
Figure 28. Photo Diode Pre-Amp Errors Over Temperature
The capacitance at the amplifier’s negative input (the sum of the
photodiode’s shunt capacitance, the op amp’s differential input
capacitance, stray capacitance due to wiring, etc.) will cause a
rise in the preamp’s noise gain over frequency. This can result in
excess noise over the bandwidth of interest. CF reduces the
noise gain “peaking” at the expense of bandwidth.
INSTRUMENTATION AMPLIFIER
The AD548C’s maximum input current of 10 pA makes it an
excellent building block for the high input impedance instrumentation amplifier shown in Figure 29. Total current drain for
this circuit is under 600 µA. This configuration is optimal for
conditioning differential voltages from high impedance sources.
The overall gain of the circuit is controlled by RG, resulting in
the following transfer function:
VOUT
VIN
REV. C
=1+
(R1 + R2 )
RG
Figure 29. Low Power Instrumentation Amplifier
Gains of 1 to 100 can be accommodated with gain nonlinearities
of less than 0.01%. Referred to input errors, which contribute
an output error proportional to in amp gain, include a maximum untrimmed input offset voltage of 0.5 mV and an input
offset voltage drift over temperature of 4 µV/°C. Output errors,
which are independent of gain, will contribute an additional
0.5 mV offset and 4 µV/°C drift. The maximum input current is
15 pA over the common-mode range, with a common-mode
impedance of over 1 × 1012 Ω. Resistor pairs R3/R5 and R4/R6
should be ratio matched to 0.01% to take full advantage of the
AD548’s high common-mode rejection. Capacitors C1 and C1′
compensate for peaking in the gain over frequency caused by
input capacitance when gains of 1 to 3 are used.
The –3 dB small signal bandwidth for this low power instrumentation amplifier is 700 kHz for a gain of 1 and 10 kHz for a
gain of 100. The typical output slew rate is 1.8 V/µs.
LOG RATIO AMPLIFIER
Log ratio amplifiers are useful for a variety of signal conditioning applications, such as linearizing exponential transducer outputs and compressing analog signals having a wide dynamic
range. The AD548’s picoamp level input current and low input
offset voltage make it a good choice for the front-end amplifier
of the log ratio circuit shown in Figure 30. This circuit produces
an output voltage equal to the log base 10 of the ratio of the input currents I1 and I2. Resistive inputs R1 and R2 are provided
for voltage inputs.
Input currents I1 and I2 set the collector currents of Q1 and Q2,
a matched pair of logging transistors. Voltages at points A and
B are developed according to the following familiar diode
equation:
V BE = (kT/q) ln (I C /I ES )
In this equation, k is Boltzmann’s constant, T is absolute temperature, q is an electron charge, and IES is the reverse saturation
current of the logging transistors. The difference of these two
voltages is taken by the subtractor section and scaled by a factor
of approximately 16 by resistors R9, R10, and R8. Temperature
–7–
AD548
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
C999a–19–12/86
TO-99 (H) Package
Plastic Mini-DIP (N) Package
Figure 30. Log Ratio Amplifier
compensation is provided by resistors R8 and R15, which have a
positive 3500 ppm/°C temperature coefficient. The transfer
function for the output voltage is:
Cerdip (Q) Package
VOUT = 1V log 10 (I 2 / I 1 )
Frequency compensation is provided by R11, R12, C1, and C2.
Small signal bandwidth is approximately 300 kHz at input currents above 100 µA and will proportionally decrease with lower
signal levels. D1, D2, R13, and R14 compensate for the effects
of the two logging transistors’ ohmic emitter resistance.
This circuit ensures a 1% log conformance error over an input
current range of 300 pA to 1 mA, with low level accuracy
limited by the AD548’s input current. The low level input voltage accuracy of this circuit is limited by the input offset voltage
and drift of the AD548.
PRINTED IN U.S.A.
To trim this circuit, set the two input currents to 10 µA and adjust VOUT to zero by adjusting the potentiometer on A3. Then
set I2 to 1 µA and adjust the scale factor such that the output
voltage is 1 V by trimming potentiometer R10. Offset adjustment for A1 and A2 is provided to increase the accuracy of the
voltage inputs.
SOIC (R) Package
–8–
REV. C