MOTOROLA MC145482SD 5v 13-bit linear pcm codec-filter Datasheet

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SEMICONDUCTOR TECHNICAL DATA
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The MC145482 is a 13–bit linear PCM Codec–Filter with 2s complement data
format, and is offered in 20–pin SOG and SSOP packages. This device
performs the voice digitization and reconstruction as well as the band limiting
and smoothing required for the voice coding in digital communication systems.
This device is designed to operate in both synchronous and asynchronous
applications and contains an on–chip precision reference voltage.
This device has an input operational amplifier whose output is the input to the
encoder section. The encoder section immediately low–pass filters the analog
signal with an active R–C filter to eliminate very high frequency noise from being
modulated down to the passband by the switched capacitor filter. From the
active R–C filter, the analog signal is converted to a differential signal. From this
point, all analog signal processing is done differentially. This allows processing
of an analog signal that is twice the amplitude allowed by a single–ended
design, which reduces the significance of noise to both the inverted and
non–inverted signal paths. Another advantage of this differential design is that
noise injected via the power supplies is a common–mode signal that is
cancelled when the inverted and non–inverted signals are recombined. This
dramatically improves the power supply rejection ratio.
After the differential converter, a differential switched capacitor filter band–
passes the analog signal from 200 Hz to 3400 Hz before the signal is digitized
by the differential 13–bit linear A/D converter. The digital output is 2s
complement format.
The decoder digital input accepts 2s complement data and reconstructs it
using a differential 13–bit linear D/A converter. The output of the D/A is
low–pass filtered at 3400 Hz and sinX/X compensated by a differential switched
capacitor filter. The signal is then filtered by an active R–C filter to eliminate the
out–of–band energy of the switched capacitor filter.
The MC145482 PCM Codec–Filter has a high impedance VAG reference pin
which allows for decoupling of the internal circuitry that generates the
mid–supply VAG reference voltage to the VSS power supply ground. This
reduces clock noise on the analog circuitry when external analog signals are
referenced to the power supply ground.
The MC145482 13–bit linear PCM Codec–Filter accepts both Short Frame
Sync and Long Frame Sync clock formats, and utilizes CMOS due to its reliable
low–power performance and proven capability for complex analog/digital VLSI
functions.
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DW SUFFIX
SOG PACKAGE
CASE 751D
20
1
SD SUFFIX
SSOP
CASE 940C
20
1
ORDERING INFORMATION
MC145482DW
MC145482SD
SOG Package
SSOP
PIN ASSIGNMENT
VAG Ref
1
20
VAG
RO–
2
19
TI+
PI
3
18
TI–
PO–
4
17
TG
PO+
5
16
HB
VDD
6
15
VSS
FSR
7
14
FST
DR
8
13
DT
BCLKR
9
12
BCLKT
10
11
MCLK
PDI
Single 5 V Power Supply
13–Bit Linear ADC/DAC Conversions with 2s Complement Data Format
Typical Power Dissipation of 25 mW, Power–Down of 0.01 mW
Fully–Differential Analog Circuit Design for Lowest Noise
Transmit Band–Pass and Receive Low–Pass Filters On–Chip
Transmit High–Pass Filter May be Bypassed by Pin Selection
Active R–C Pre–Filtering and Post–Filtering
On–Chip Precision Reference Voltage of 1.575 V for a 0 dBm TLP
@ 600 Ω
Full–Duplex Sample Rates from 7 k to 16 k Samples/s
3–Terminal Input Op Amp Can be Used, or a 2–Channel Input Multiplexer
Receive Gain Control from 0 dB to – 21 dB in 3 dB Steps in Synchronous
Operation
Push–Pull 300 Ω Power Drivers with External Gain Adjust
This document contains information on a product under development. Motorola reserves the right to change or discontinue this product without notice.
REV 0
3/97 TN97032700

Motorola, Inc. 1997
MOTOROLA
MC145482
1
RECEIVE
SHIFT
REGISTER
RO –
DR
DAC
FREQ
PI
–
+
PO –
FSR
BCLKR
SHARED
DAC
SEQUENCE
AND
CONTROL
–1
PO +
VDD
VDD
VSS
R*
FST
R*
1
VAG
MCLK
BCLKT
1.575 V
REF
VAG Ref
PDI
VSS
TG
TI –
TI +
–
+
ADC
FREQ
HB
TRANSMIT
SHIFT
REGISTER
DT
Figure 1. MC145482 13–Bit Linear PCM Codec–Filter Block Diagram
DEVICE DESCRIPTION
A PCM Codec–Filter is used for digitizing and reconstructing the human voice. These devices are used primarily for
the telephone network to facilitate voice switching and transmission. Once the voice is digitized, it may be switched by
digital switching methods or transmitted long distance (T1,
microwave, satellites, etc.) without degradation. The name
codec is an acronym from ‘‘COder’’ for the analog–to–digital
converter (ADC) used to digitize voice, and ‘‘DECoder’’ for
the digital–to–analog converter (DAC) used for reconstructing voice. A codec is a single device that does both the ADC
and DAC conversions.
To digitize intelligible voice requires a signal–to–distortion
ratio of about 30 dB over a dynamic range of about 40 dB.
This may be accomplished with a linear 13–bit ADC and
DAC. The MC145482 satisfies these requirements and may
be used as the analog front–end for voice coders using DSP
technology to further compress the digital data stream.
In a sampling environment, Nyquist theory says that to
properly sample a continuous signal, it must be sampled at a
frequency higher than twice the signal’s highest frequency
component. Voice contains spectral energy above 3 kHz, but
MC145482
2
its absence is not detrimental to intelligibility. To reduce the
digital data rate, which is proportional to the sampling rate, a
sample rate of 8 kHz was adopted, consistent with a bandwidth of 3 kHz. This sampling requires a low–pass filter to
limit the high frequency energy above 3 kHz from distorting
the in–band signal. The telephone line is also subject to
50/60 Hz power line coupling, which must be attenuated
from the signal by a high–pass filter before the analog–to–
digital converter. The MC145482 includes a high–pass filter
for compatibility with existing telephone applications, but it
may be removed from the analog input signal path by the
high–pass bypass pin.
The digital–to–analog conversion process reconstructs a
staircase version of the desired in–band signal, which has
spectral images of the in–band signal modulated about the
sample frequency and its harmonics. These spectral images
are called aliasing components, which need to be attenuated
to obtain the desired signal. The low–pass filter used to attenuate these aliasing components is typically called a reconstruction or smoothing filter.
The MC145482 PCM Codec–Filter has the codec, both
presampling and reconstruction filters, and a precision voltage reference on–chip.
MOTOROLA
PIN DESCRIPTIONS
POWER SUPPLY
VDD
Positive Power Supply (Pin 6)
This is the most positive power supply and is typically connected to + 5 V. This pin should be decoupled to VSS with a
0.1 µF ceramic capacitor.
when a logic 1 is applied to this pin. The device goes through
a power–up sequence when this pin is taken to a logic 1
state, which prevents the DT PCM output from going low impedance for at least two FST cycles. The VAG and VAG Ref
circuits and the signal processing filters must settle out before the DT PCM output or the RO– receive analog output
will represent a valid analog signal.
ANALOG INTERFACE
VSS
Negative Power Supply (Pin 15)
TI+
Transmit Analog Input (Non–Inverting) (Pin 19)
This is the most negative power supply and is typically
connected to 0 V.
This is the non–inverting input of the transmit input gain
setting operational amplifier. This pin accommodates a differential to single–ended circuit for the input gain setting op
amp. This allows input signals that are referenced to the V SS
pin to be level shifted to the VAG pin with minimum noise.
This pin may be connected to the VAG pin for an inverting
amplifier configuration if the input signal is already referenced to the VAG pin. The common mode range of the TI+
and TI– pins is from 1.2 V, to V DD minus 1.2 V. This is an FET
gate input.
The TI+ pin also serves as a digital input control for the
transmit input multiplexer. Connecting the TI+ pin to V DD will
place this amplifier’s output (TG) into a high–impedance
state, and selects the TG pin to serve as a high–impedance
input to the transmit filter. Connecting the TI+ pin to VSS will
also place this amplifier’s output (TG) into a high–impedance
state, and selects the TI– pin to serve as a high–impedance
input to the transmit filter.
VAG
Analog Ground Output (Pin 20)
This output pin provides a mid–supply analog ground. This
pin should be decoupled to VSS with a 0.01 µF ceramic capacitor. All analog signal processing within this device is referenced to this pin. If the audio signals to be processed are
referenced to V SS, then special precautions must be utilized
to avoid noise between V SS and the VAG pin. Refer to the applications information in this document for more information.
The VAG pin becomes high impedance when this device is in
the powered–down mode.
VAG Ref
Analog Ground Reference Bypass (Pin 1)
This pin is used to capacitively bypass the on–chip circuitry that generates the mid–supply voltage for the VAG output
pin. This pin should be bypassed to VSS with a 0.1 µF ceramic capacitor using short, low inductance traces. The VAG Ref
pin is only used for generating the reference voltage for the
VAG pin. Nothing is to be connected to this pin in addition to
the bypass capacitor. All analog signal processing within this
device is referenced to the VAG pin. If the audio signals to be
processed are referenced to VSS, then special precautions
must be utilized to avoid noise between VSS and the VAG pin.
Refer to the applications information in this document for
more information. When this device is in the powered–down
mode, the VAG Ref pin is pulled to the VDD power supply with
a non–linear, high–impedance circuit.
CONTROL
HB
Transmit High–Pass Filter Bypass (Pin 16)
TI–
Transmit Analog Input (Inverting) (Pin 18)
This is the inverting input of the transmit gain setting operational amplifier. Gain setting resistors are usually connected from this pin to TG and from this pin to the analog
signal source. The common mode range of the TI+ and TI–
pins is from 1.2 V to VDD – 1.2 V. This is an FET gate input.
The TI– pin also serves as one of the transmit input mulitplexer pins when the TI+ pin is connected to VSS. When TI+
is connected to VDD, this pin is ignored. See the pin descriptions for the TI+ and the TG pins for more information.
TG
Transmit Gain (Pin 17)
This pin selects whether the transmit high–pass filter will
be used or bypassed, which allows frequencies below
200 Hz to appear at the input of the ADC to be digitized. This
high–pass filter is a third order filter for attenuating power line
frequencies, typically 50/60 Hz. A logic low selects this filter.
A logic high deselects or bypasses this filter. When the filter
is bypassed, the transmit frequency response extends down
to dc.
This is the output of the transmit gain setting operational
amplifier and the input to the transmit band–pass filter. This
op amp is capable of driving a 2 kΩ load. Connecting the TI+
pin to VDD will place the TG pin into a high–impedance state,
and selects the TG pin to serve as a high–impedance input to
the transmit filter. All signals at this pin are referenced to the
VAG pin. When TI+ is connected to VSS, this pin is ignored.
See the pin descriptions for TI+ and TI– pins for more information. This pin is high impedance when the device is in
the powered–down mode.
PDI
Power–Down Input (Pin 10)
RO–
Receive Analog Output (Inverting) (Pin 2)
This pin puts the device into a low power dissipation mode
when a logic 0 is applied. When this device is powered down,
all of the clocks are gated off and all bias currents are turned
off, which causes RO–, PO–, PO+, TG, VAG, and DT to become high impedance. The device will operate normally
This is the inverting output of the receive smoothing filter
from the digital–to–analog converter. This output is capable
of driving a 2 kΩ load to 1.575 V peak referenced to the VAG
pin. If the device is operated half–channel with the FST pin
clocking and FSR pin held low, the receive filter input will be
MOTOROLA
MC145482
3
connected to the VAG voltage. This minimizes transients at
the RO– pin when full–channel operation is resumed by
clocking the FSR pin. This pin is high impedance when the
device is in the powered–down mode.
PI
Power Amplifier Input (Pin 3)
This is the inverting input to the PO– amplifier. The non–
inverting input to the PO– amplifier is internally tied to the
VAG pin. The PI and PO– pins are used with external resistors in an inverting op amp gain circuit to set the gain of the
PO+ and PO– push–pull power amplifier outputs. Connecting PI to VDD will power down the power driver amplifiers and
the PO+ and PO– outputs will be high impedance.
PO–
Power Amplifier Output (Inverting) (Pin 4)
This is the inverting power amplifier output, which is used
to provide a feedback signal to the PI pin to set the gain of
the push–pull power amplifier outputs. This pin is capable of
driving a 300 Ω load to PO+. The PO+ and PO– outputs are
differential (push–pull) and capable of driving a 300 Ω load to
3.15 V peak, which is 6.3 V peak–to–peak. The bias voltage
and signal reference of this output is the VAG pin. The VAG
pin cannot source or sink as much current as this pin, and
therefore low impedance loads must be between PO+ and
PO–. The PO+ and PO– differential drivers are also capable
of driving a 100 Ω resistive load or a 100 nF Piezoelectric
transducer in series with a 20 Ω resister with a smalll increase in distortion. These drivers may be used to drive resistive loads of ≥ 32 Ω when the gain of PO– is set to 1/4 or
less. Connecting PI to VDD will power down the power driver
amplifiers, and the PO+ and PO– outputs will be high impedance. This pin is also high impedance when the device is
powered down by the PDI pin.
PO+
Power Amplifier Output (Non–Inverting) (Pin 5)
This is the non–inverting power amplifier output, which is
an inverted version of the signal at PO–. This pin is capable
of driving a 300 Ω load to PO–. Connecting PI to VDD will
power down the power driver amplifiers and the PO+ and
PO– outputs will be high impedance. This pin is also high impedance when the device is powered down by the PDI pin.
See PI and PO– for more information.
DIGITAL INTERFACE
MCLK
Master Clock (Pin 11)
This is the master clock input pin. The clock signal applied
to this pin is used to generate the internal 256 kHz clock and
sequencing signals for the switched–capacitor filters, ADC,
and DAC. The internal prescaler logic compares the clock on
this pin to the clock at FST (8 kHz) and will automatically
accept 256, 512, 1536, 1544, 2048, 2560, or 4096 kHz. For
MCLK frequencies of 256 and 512 kHz, MCLK must be syn-
MC145482
4
chronous and approximately rising edge aligned to FST. For
optimum performance at frequencies of 1.536 MHz and
higher, MCLK should be synchronous and approximately rising edge aligned to the rising edge of FST. In many applications, MCLK may be tied to the BCLKT pin.
FST
Frame Sync, Transmit (Pin 14)
This pin accepts an 8 kHz clock that synchronizes the output of the serial PCM data at the DT pin. This input is compatible with both Long Frame Sync and Short Frame Sync. If
both FST and FSR are held low for several 8 kHz frames, the
device will power down. FST must be clocking for the device
to power up affter being powered down by the frame syncs.
BCLKT
Bit Clock, Transmit (Pin 12)
This pin controls the transfer rate of transmit PCM data. In
the synchronous modes of sign–bit extended and receive
gain adjust, the BCLKT also controls the transfer rate of the
receive PCM data. This pin can accept any bit clock frequency from 256 to 4096 kHz for Long Frame Sync and Short
Frame Sync timing.
DT
Data, Transmit (Pin 13)
This pin is controlled by FST and BCLKT and is high impedance except when outputting PCM data. This pin is high
impedance when the device is in the powered–down mode.
FSR
Frame Sync, Receive (Pin 7)
This pin accepts an 8 kHz clock, which synchronizes the
input of the serial PCM data at the DR pin. FSR can be
asynchronous to FST in the Long Frame Sync or Short
Frame Sync modes.
BCLKR
Bit Clock, Receive (Pin 9)
This pin accepts any bit clock frequency from 256 to 4096
kHz. The BCLKR pin is also used as a mode select pin when
not being clocked for several 8 kHz frames. The BCKLT pin
is used to clock the receive PCM data transfers when the
BCLKR pin is not being clocked. When the BCLKR pinis a
logic 0, the sign–bit extended synchronous mode is selected,
which uses 16–bit transfers with the first four bits being the
sign bit. When the BCLKR pin is a logic 1, the receive gain
adjust synchronous mode is selected, which uses a 13–bit
transfer for the transmit PCM data, but uses a 16–bit transfer
for the receive side, with the 13–bit voice data being first, followed by three bits which control the attenuation of the receive analog output.
DR
Data, Receive (Pin 8)
This pin is the PCM data input. See the pin descriptions for
FSR, BCLKR, and BCKLT for more information.
MOTOROLA
FUNCTIONAL DESCRIPTION
ANALOG INTERFACE AND SIGNAL PATH
The transmit portion of this device includes a low–noise,
three–terminal op amp capable of driving a 2 kΩ load. This
op amp has inputs of TI+ (Pin 19) and TI– (Pin 18) and its
output is TG (Pin 17). This op amp is intended to be configured in an inverting gain circuit. The analog signal may be
applied directly to the TG pin if this transmit op amp is independently powered down by connecting the TI+ input to the
VDD power supply. The TG pin becomes high impedance
when the transmit op amp is powered down. The TG pin is
internally connected to a 3–pole anti–aliasing pre–filter. This
pre–filter incorporates a 2–pole Butterworth active low–pass
filter, followed by a single passive pole. This pre–filter is followed by a single–ended to differential converter that is
clocked at 512 kHz. All subsequent analog processing utilizes fully–differential circuitry. The next section is a fully–differential, 5–pole switched–capacitor low–pass filter with a
3.4 kHz frequency cutoff. After this filter is a 3–pole
switched–capacitor high–pass filter having a cutoff frequency of about 200 Hz. This high–pass stage has a transmission zero at dc that eliminates any dc coming from the
analog input or from accumulated op amp offsets in the preceding filter stages. The high–pass filter may be bypassed or
removed from the signal path by the HB pin. When the high–
pass filter is bypassed, the frequency response extends
down to include dc. The last stage of the high–pass filter is
an autozeroed sample and hold amplifier.
One bandgap voltage reference generator and digital–to–
analog converter (DAC) are shared by the transmit and receive sections. The autozeroed, switched–capacitor
bandgap reference generates precise positive and negative
reference voltages that are virtually independent of temperature and power supply voltage. A capacitor array (CDAC) is
combined with a resistor string (RDAC) to implement the
13–bit linear DAC structure. The encode process uses the
DAC, the voltage reference, and a frame–by–frame autozeroed comparator to implement a successive approximation conversion algorithm. All of the analog circuitry involved
in the data conversion (the voltage reference, RDAC, CDAC,
and comparator) are implemented with a differential architecture.
The receive section includes the DAC described above, a
sample and hold amplifier, a 5–pole, 3400 Hz switched capacitor low–pass filter with sinX/X correction, and a 2–pole
active smoothing filter to reduce the spectral components of
the switched capacitor filter. The output of the smoothing filter is buffered by an amplifier, which is output at the RO– pin.
This output is capable of driving a 2 kΩ load to the VAG pin.
The MC145482 also has a pair of power amplifiers that are
connected in a push–pull configuration. The PI pin is the inverting input to the PO– power amplifier. The non–inverting
input is internally tied to the VAG pin. This allows this amplifier
to be used in an inverting gain circuit with two external resistors. The PO+ amplifier has a gain of minus one, and is internally connected to the PO– output. This complete power
amplifier circuit is a differential (push–pull) amplifier with adjustable gain. The power amplifier may be powered down independently of the rest of the chip by connecting the PI pin to
VDD.
The calibration level for both ADC and DAC of this 13–bit
linear PCM Codec–Filter is referenced to Mu–Law with the
MOTOROLA
same bit voltage weighting about the zero crossing. This results in the 0 dBm0 calibration level being 3.20 dB below the
peak sinusoidal level before clipping. Based on the reference
voltage of 1.575 V, the calibration level is 0.775 Vrms or
0 dBm at 600 Ω.
The MC145482 has the ability to attenuate the receive
analog output when used in the receive gain adjust mode.
This mode is accessed by applying a logic high to the
BCLKR pin while the rest of the clock pins are clocked normally. This allows three additional bits that will be used to
control the gain of the analog output to be clocked into the
DR pin following the 13 bits of voice data. Table 1 shows the
attenuation values and the corresponding digital codes.
Table 1. Receive Gain Adjust Mode
Coefficients and Attenuation Weightings
Coefficient
Attenuation in dB
000
0
001
–3
010
–6
011
–9
100
– 12
101
– 15
110
– 18
111
– 21
POWER–DOWN
There are two methods of putting this device into a low
power consumption mode, which makes the device nonfunctional and consumes virtually no power. PDI is the power–
down input pin which, when taken low, powers down the
device. Another way to power the device down is to hold both
the FST and FSR pins low while the BCLKT and MCLK pins
are clocked. When the chip is powered down, the VAG, TG,
RO–, PO+, PO–, and DT outputs are high impedance and
the VAG Ref pin is pulled to the VDD power supply with a non–
linear, high–impedance circuit. To return the chip to the power–up state, PDI must be high and the FST frame sync pulse
must be present while the BCLKT and MCLK pins are
clocked. The DT output will remain in a high–impedance
state for at least two 8 kHz FST pulses after power–up.
MASTER CLOCK
Since this codec–filter design has a single DAC architecture, the MCLK pin is used as the master clock for all analog
signal processing including analog–to–digital conversion,
digital–to–analog conversion, and for transmit and receive filtering functions of this device. The clock frequency applied to
the MCLK pin may be 256 kHz, 512 kHz, 1.536 MHz,
1.544 MHz, 2.048 MHz, 2.56 MHz, or 4.096 MHz. This device has a prescaler that automatically determines the proper
divide ratio to use for the MCLK input, which achieves the required 256 kHz internal sequencing clock. The clocking requirements of the MCLK input are independent of the PCM
data transfer mode (i.e., Long Frame Sync, Short Frame
Sync, whether the device is used in the synchronous modes
or not).
MC145482
5
DIGITAL I/O
The MC145482 is a 13–bit linear device using 2s complement data format. Table 2 shows the 13–bit data word format
for the maximum positive code and negative zero and full–
scale.
Table 3 shows the series of eight 13–bit PCM words that
correspond to a digital milliwatt. The digital milliwatt is the
1 kHz calibration signal reconstructed by the DAC that defines the absolute gain or 0 dBm0 transmission level point
(TLP) of the DAC. The calibration level for this 13–bit linear
ADC and DAC is referenced to Mu–Law with the same bit
voltage weighting about the zero crossing. This results in the
0 dBm0 calibration level being 3.20 dB below the peak sinusoidal level before clipping. Refer to Figures 2a–2d for a
summary and comparison of the four PCM data interface
modes of this device.
Table 2. PCM Codes for Zero and Full–Scale
Level
Sign Bit
Magnitude Bits
+ Full Scale
0
1111 1111 1111
+ One Step
0
0000 0000 0001
Zero
0
0000 0000 0000
– One Step
1
1111 1111 1111
– Full Scale
1
0000 0000 0000
Table 3. PCM Codes for 1 kHz Digital Milliwatt
Level
Sign Bit
Magnitude Bits
π/8
3π/8
5π/8
7π/8
9π/8
11π/8
13π/8
15π/8
MC145482
6
MOTOROLA
FST (FSR)
BCLKT (BCLKR)
DT
DR
DON’T CARE
1
2
3
4
5
6
7
8
9
10
11
12 13
1
2
3
4
5
6
7
8
9
10
11
12
13
DON’T CARE
Figure 2a. Long Frame Sync (Transmit and Receive Have Individual Clocking)
FST (FSR)
BCLKT (BCLKR)
DT
DR
DON’T CARE
1
2
3
4
5
6
7
8
9
10
11
12 13
1
2
3
4
5
6
7
8
9
10
11
12
13
DON’T CARE
Figure 2b. Short Frame Sync (Transmit and Receive Have Individual Clocking)
FST (FSR)
SHORT OR
LONG FRAME
SYNC
BCLKT
DT
DR
DON’T CARE
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15 16
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
DON’T CARE
Figure 2c. Sign–Extended (BCLKR = 0)
Transmit and receive both use BCLKT, and the first four data bits are the sign bit.
FST may occur at a different time than FSR.
FST (FSR)
SHORT OR
LONG FRAME
SYNC
BCLKT
DT
DR
DON’T CARE
1
2
3
4
5
6
7
8
9
10
11
12 13
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
DON’T CARE
Figure 2d. Receive Gain Adjust (BCLKR = 1)
Transmit and receive both use BCLKT. FST may occur at a different time than FSR.
Bits 14, 15, and 16, clocked into DR, are used for attenuation control for the receive analog output.
Figure 2. Digital Timing Modes for the PCM Data Interface
MOTOROLA
MC145482
7
PRINTED CIRCUIT BOARD LAYOUT
CONSIDERATIONS
The MC145482 is manufactured using high–speed CMOS
VLSI technology to implement the complex analog signal
processing functions of a PCM Codec–Filter. The fully–differential analog circuit design techniques used for this device
result in superior performance for the switched capacitor filters, the analog–to–digital converter (ADC) and the digital–
to–analog converter (DAC). Special attention was given to
the design of this device to reduce the sensitivities of noise,
including power supply rejection and susceptibility to radio
frequency noise. This special attention to design includes a
fifth order low–pass filter, followed by a third order high–pass
filter whose output is converted to a digital signal with greater
than 75 dB of dynamic range, all operating on a single 5 V
power supply. This results in an LSB size for small audio signals of about 386 µV. The typical idle channel noise level of
this device is less than one LSB. In addition to the dynamic
range of the codec–filter function of this device, the input
gain–setting op amp has the capability of greater than 35 dB
of gain intended for an electret microphone interface.
This device was designed for ease of implementation, but
due to the large dynamic range and the noisy nature of the
environment for this device (digital switches, radio telephones, DSP front–end, etc.) special care must be taken to
assure optimum analog transmission performance.
PC BOARD MOUNTING
It is recommended that the device be soldered to the PC
board for optimum noise performance. If the device is to be
used in a socket, it should be placed in a low parasitic pin
inductance (generally, low–profile) socket.
POWER SUPPLY, GROUND, AND NOISE
CONSIDERATIONS
This device is intended to be used in switching applications which often require plugging the PC board into a rack
with power applied. This is known as ‘‘hot–rack insertion.’’ In
these applications care should be taken to limit the voltage
on any pin from going positive of the VDD pins, or negative of
the VSS pins. One method is to extend the ground and power
contacts of the PCB connector. The device has input protection on all pins and may source or sink a limited amount of
current without damage. Current limiting may be accomplished by series resistors between the signal pins and the
connector contacts.
The most important considerations for PCB layout deal
with noise. This includes noise on the power supply, noise
generated by the digital circuitry on the device, and cross
coupling digital or radio frequency signals into the audio signals of this device. The best way to prevent noise is to:
1. Keep digital signals as far away from audio signals as
possible.
2. Keep radio frequency signals as far away from the audio
signals as possible.
3. Use short, low inductance traces for the audio circuitry
to reduce inductive, capacitive, and radio frequency
noise sensitivities.
4. Use short, low inductance traces for digital and RF
circuitry to reduce inductive, capacitive, and radio
frequency radiated noise.
MC145482
8
5. Bypass capacitors should be connected from the VDD,
VAG Ref, and VAG pins to VSS with minimal trace length.
Ceramic monolithic capacitors of about 0.1 µF are
acceptable for the VDD and VAG Ref pins to decouple the
device from its own noise. The VDD capacitor helps
supply the instantaneous currents of the digital circuitry
in addition to decoupling the noise which may be
generated by other sections of the device or other
circuitry on the power supply. The VAG Ref decoupling
capacitor is effecting a low–pass filter to isolate the
mid–supply voltage from the power supply noise generated on–chip, as well as external to the device. The VAG
decoupling capacitor should be about 0.01 µF. This
helps to reduce the impedance of the VAG pin to VSS at
frequencies above the bandwidth of the VAG generator,
which reduces the susceptiblility to RF noise.
6. Use a short, wide, low inductance trace to connect the
VSS ground pin to the power supply ground. The VSS pin
is the digital ground and the most negative power supply
pin for the analog circuitry. All analog signal processing
is referenced to the VAG pin, but because digital and RF
circuitry will probably be powered by this same ground,
care must be taken to minimize high frequency noise in
the VSS trace. Depending on the application, a double–
sided PCB with a VSS ground plane connecting all of the
digital and analog VSS pins together would be a good
grounding method. A multilayer PC board with a ground
plane connecting all of the digital and analog VSS pins
together would be the optimal ground configuration.
These methods will result in the lowest resistance and
the lowest inductance in the ground circuit. This is
important to reduce voltage spikes in the ground circuit
resulting from the high speed digital current spikes. The
magnitude of digitally induced voltage spikes may be
hundreds of times larger than the analog signal the
device is required to digitize.
7. Use a short, wide, low inductance trace to connect the
V DD power supply pin to the 5 V power supply.
Depending on the application, a double–sided PCB with
VDD bypass capacitors to the VSS ground plane, as
described above, may complete the low impedance
coupling for the power supply. For a multilayer PC board
with a power plane, connecting all of the V DD pins to the
power plane would be the optimal power distribution
method. The integrated circuit layout and packaging
considerations for the 5 V V DD power circuit are
essentially the same as for the VSS ground circuit.
8. The VAG pin is the reference for all analog signal
processing. In some applications the audio signal to be
digitized may be referenced to the VSS ground. To
reduce the susceptibility to noise at the input of the ADC
section, the three–terminal op amp may be used in a
differential to single–ended circuit to provide level
conversion from the VSS ground to the VAG ground with
noise cancellation. The op amp may be used for more
than 35 dB of gain in microphone interface circuits, which
will require a compact layout with minimum trace lengths
as well as isolation from noise sources. It is recommended that the layout be as symmetrical as possible to
avoid any imbalances which would reduce the noise
cancelling benefits of this differential op amp circuit.
Refer to the application schematics for examples of this
circuitry.
MOTOROLA
If possible, reference audio signals to the VAG pin
instead of to the VSS pin. Handset receivers and telephone line interface circuits using transformers may be
audio signal referenced completely to the VAG pin. Re-
MOTOROLA
fer to the application schematics for examples of this
circuitry. The VAG pin cannot be used for ESD or line
protection.
MC145482
9
MAXIMUM RATINGS (Voltages Referenced to VSS Pin)
Rating
Symbol
Value
Unit
VDD
– 0.5 to 6
V
Voltage on Any Analog Input or Output Pin
VSS – 0.3 to VDD + 0.3
V
Voltage on Any Digital Input or Output Pin
VSS – 0.3 to VDD + 0.3
V
TA
– 40 to + 85
°C
Tstg
– 85 to +150
°C
DC Supply Voltage
Operating Temperature Range
Storage Temperature Range
POWER SUPPLY (TA = – 40 to + 85°C)
Min
Typ
Max
Unit
4.75
5.0
5.25
V
(No Load, PI ≥ VDD – 0.5 V)
(No Load, PI ≤ VDD – 1.5 V)
—
—
5.0
5.2
—
—
mA
Power–Down Current (VIH for Logic Levels
PDI = VSS
Must be ≥ VDD – 0.5 V)
FST and FSR = VSS, PDI = VDD
—
—
0.001
0.01
—
—
mA
Symbol
Min
Max
Unit
Characteristics
DC Supply Voltage
Active Current Dissipation (VDD = 5 V)
DIGITAL LEVELS (VDD = 4.75 to 5.25 V, VSS = 0 V, TA = – 40 to + 85°C)
Characteristics
Input Low Voltage
VIL
—
0.6
V
Input High Voltage
VIH
2.4
—
V
Output Low Voltage (DT Pin, IOL= 2.5 mA)
VOL
—
0.4
V
Output High Voltage (DT Pin, IOH = – 2.5 mA)
VOH
VDD – 0.5
—
V
Input Low Current (VSS ≤ Vin ≤ VDD)
IIL
– 10
+ 10
µA
Input High Current (VSS ≤ Vin ≤ VDD)
IIH
– 10
+ 10
µA
Output Current in High Impedance State (VSS ≤ DT ≤ VDD)
IOZ
– 10
+ 10
µA
Cin
—
10
pF
Cout
—
15
pF
Input Capacitance of Digital Pins (Except DT)
Input Capacitance of DT Pin when High–Z
MC145482
10
MOTOROLA
ANALOG ELECTRICAL CHARACTERISTICS (VDD = 4.75 to 5.25 V, VSS = 0 V, TA = – 40 to + 85°C)
Min
Typ
Max
Unit
Input Current
TI+, TI–
—
± 0.1
± 1.0
µA
Input Resistance to VAG (VAG – 0.5 V ≤ Vin ≤ VAG + 0.5 V)
TI+, TI–
10
—
—
MΩ
Input Capacitance
TI+, TI–
—
—
10
pF
Input Offset Voltage of TG Op Amp
TI+, TI–
—
—
±5
mV
Input Common Mode Voltage Range
TI+, TI–
1.2
VDD – 1.2
V
Input Common Mode Rejection Ratio
TI+, TI–
—
TBD
—
dB
Gain Bandwidth Product (10 kHz) of TG Op Amp (RL ≥ 10 kΩ)
—
3000
—
kHz
DC Open Loop Gain of TG Op Amp (RL ≥ 10 kΩ)
—
95
—
dB
Equivalent Input Noise (C–Message) Between TI+ and TI– at TG
—
– 30
—
dBrnC
Output Load Capacitance for TG Op Amp
0
—
100
pF
0.5
—
VDD – 0.5
V
± 1.0
—
—
mA
Characteristics
Output Voltage Range for TG
(RL = 2 kΩ to VAG)
Output Current (0.5 V ≤ Vout ≤ VDD – 0.5 V)
TG, RO–
Output Load Resistance to VAG
TG, RO–
2
—
—
kΩ
Output Impedance
RO–
—
1
—
Ω
Output Load Capacitance
RO–
0
—
500
pF
—
—
± 25
mV
VDD/2 – 0.1
VDD/2
VDD/2 + 0.1
V
± 2.0
± 10
—
mA
TBD
TBD
TBD
TBD
—
—
dBC
± 1.0
µA
DC Output Offset Voltage of RO– Referenced to VAG
VAG Output Voltage Referenced to VSS (No Load)
VAG Output Current with ± 25 mV Change in Output Voltage
Power Supply Rejection Ratio
(0 to 100 kHz @100 mVrms Applied to VDD,
C–Message Weighting, All Analog Signals
Referenced to VAG Pin)
Transmit
Receive
Power Drivers PI, PO+, PO–
Input Current (VAG – 0.5 V ≤ PI ≤ VAG + 0.5 V)
PI
—
± 0.05
Input Resistance (VAG – 0.5 V ≤ PI ≤ VAG + 0.5 V)
PI
10
—
—
MΩ
Input Offset Voltage
PI
—
—
± 20
mV
—
—
± 50
mV
± 10
—
—
mA
PO+ or PO– Output Resistance (Inverted Unity Gain for PO–)
—
1
—
Ω
Gain Bandwidth Product (10 kHz, Open Loop for PO–)
—
1000
—
kHz
Output Offset Voltage of PO+ Relative to PO– (Inverted Unity Gain for PO–)
Output Current (VSS + 0.7 V ≤ PO+ or PO– ≤ VDD – 0.7 V)
Load Capacitance (PO+ or PO– to VAG, or PO+ to PO–)
Gain of PO+ Relative to PO– (RL = 300 Ω, + 3 dBm0, 1 kHz)
Total Signal to Distortion at PO+ and PO– with a Differential Load of:
300 Ω
100 nF in series with ≥ 20 Ω
≥ 100 Ω
Power Supply Rejection Ratio
(0 to 25 kHz @ 100 mVrms Applied to VDD.
PO– Connected to PI. Differential or Measured
Referenced to VAG Pin.)
MOTOROLA
0 to 4 kHz
4 to 25 kHz
0
—
1000
pF
– 0.2
0
+ 0.2
dB
45
—
—
60
40
40
—
—
—
dBC
TBD
—
TBD
TBD
—
—
dB
MC145482
11
ANALOG TRANSMISSION PERFORMANCE
(VDD = 4.75 to 5.25 V, VSS = 0 V, All Analog Signals Referenced to VAG, 0 dBm0 = 0.775 Vrms = 0 dBm @ 600 Ω, FST = FSR = 8 kHz,
BCLKT = MCLK = 2.048 MHz Synchronous Operation, TA = – 40 to + 85°C, Unless Otherwise Noted)
A/D
Characteristics
Ch
i i
Peak Single Frequency Tone Amplitude without Clipping
Tmax
Min
Typ
D/A
Max
Min
Typ
Max
Units
U i
—
1.575
—
—
1.575
—
Vpk
– 0.25
—
+ 0.25
– 0.25
—
+ 0.25
dB
—
—
TBD
TBD
—
—
—
—
TBD
TBD
—
—
dB
—
TBD
—
—
TBD
—
dB
+ 3 dBm0
0 dBm0
– 10 dBm0
– 20 dBm0
– 30 dBm0
–40 dBm0
–50 dBm0
–60 dBm0
—
—
—
—
—
—
—
—
55
58
58
53
44
34
24
14
—
—
—
—
—
—
—
—
—
—
—
—
—
—
—
—
60
60
60
55
46
36
26
16
—
—
—
—
—
—
—
—
dBC
Idle Channel Noise (For End–to–End and A/D, See Note 1)
(C–Message Weighted)
(Psophometric Weighted)
—
—
—
—
17
– 69
—
—
—
—
11
– 79
dBr nc0
dBm0p
—
—
—
—
– 1.0
– 0.20
– 0.35
– 0.9
—
—
—
—
—
—
–3
—
—
—
—
–3
—
—
– 40
– 30
– 26
—
– 0.4
+ 0.20
+ 0.20
0
—
– 14
– 32
– 0.5
– 0.5
– 0.5
– 0.5
– 0.5
– 0.20
– 0.35
– 0.9
—
—
—
—
—
—
—
—
—
—
—
–3
—
—
0
0
0
0
0
+ 0.20
+ 0.20
0
—
– 14
– 30
dB
Out–of–Band Spurious at VAG Ref (300 to 3400 Hz @ 0 dBm0 in)
4600 to 7600 Hz
7600 to 8400 Hz
8400 to 100,000 Hz
—
—
—
—
—
—
—
—
—
—
—
—
—
—
—
– 30
– 40
– 30
dB
Idle Channel Noise Selective (8 kHz, Input = VAG, 30 Hz Bandwidth)
—
—
—
—
—
– 70
dBm0
Absolute Delay (1600 Hz) (HB = 0)
—
—
315
—
—
205
µs
—
—
—
—
—
—
—
—
—
—
—
—
—
—
210
130
70
35
70
95
145
– 40
– 40
– 40
– 30
—
—
—
—
—
—
—
—
—
—
—
—
—
—
85
110
175
µs
—
—
– 75
—
—
– 75
dB
Absolute Gain (0 dBm0 @ 1.02 kHz, TA = 25°C, VDD = 5.0 V)
Absolute Gain Variation with Temperature
0 to + 70°C
– 40 to + 85°C
Absolute Gain Variation with Power Supply (TA = 25°C)
Total Distortion, 1.02 kHz Tone (C–Message Weighting)
Frequency Response
(Relative to 1.02 kHz @ 0 dBm0) (HB = 0)
15 Hz
50 Hz
60 Hz
165 Hz
200 Hz
300 to 3000 Hz
3300 Hz
3400 Hz
3600 Hz
4000 Hz
4600 Hz to 100 kHz
Group Delay Referenced to 1600 Hz (HB = 0)
Crosstalk of 1020 Hz @ 0 dBm0 from A/D or D/A (Note 2)
500 to 600 Hz
600 to 800 Hz
800 to 1000 Hz
1000 to 1600 Hz
1600 to 2600 Hz
2600 to 2800 Hz
2800 to 3000 Hz
NOTES:
1. Extrapolated from a 1020 Hz @ – 50 dBm0 distortion measurement to correct for encoder enhancement.
2. Selectively measured while stimulated with 2667 Hz @ – 50 dBm0.
MC145482
12
MOTOROLA
DIGITAL SWITCHING CHARACTERISTICS, LONG FRAME SYNC AND SHORT FRAME SYNC
(VDD = 4.75 to 5.25 V, VSS = 0 V, All Digital Signals Referenced to VSS, TA = – 40 to + 85°C, CL = 150 pF, FST = FSR = 8 kHz,
Unless Otherwise Noted)
Ref.
No.
Characteristics
Min
Typ
Max
Unit
kHz
1
Master Clock Frequency for MCLK
—
—
—
—
—
—
—
256
512
1536
1544
2048
2560
4096
—
—
—
—
—
—
—
1
MCLK Duty Cycle for 256 kHz Operation
45
—
55
%
2
Minimum Pulse Width High for MCLK (Frequencies of 512 kHz or Greater)
50
—
—
ns
3
Minimum Pulse Width Low for MCLK (Frequencies of 512 kHz or Greater)
50
—
—
ns
4
Rise Time for All Digital Signals
—
—
50
ns
5
Fall Time for All Digital Signals
—
—
50
ns
6
Setup Time from MCLK Low to FST High
50
—
—
ns
7
Setup Time from FST High to MCLK Low
50
—
—
ns
8
Bit Clock Data Rate for BCLKT or BCLKR
256
—
4096
kHz
9
Minimum Pulse Width High for BCLKT or BCLKR
50
—
—
ns
10
Minimum Pulse Width Low for BCLKT or BCLKR
50
—
—
ns
11
Hold Time from BCLKT (BCLKR) Low to FST (FSR) High
20
—
—
ns
12
Setup Time for FST (FSR) High to BCLKT (BCLKR) Low
80
—
—
ns
13
Setup Time from DR Valid to BCLKR Low
0
—
—
ns
14
Hold Time from BCLKR Low to DR Invalid
50
—
—
ns
LONG FRAME SPECIFIC TIMING
15
Hold Time from 2nd Period of BCLKT (BCLKR) Low to FST (FSR) Low
50
—
—
ns
16
Delay Time from FST or BCLKT, Whichever is Later, to DT for Valid MSB Data
—
—
60
ns
17
Delay Time from BCLKT High to DT for Valid Data
—
—
60
ns
18
Delay Time from the Later of the 13th (16th for Sign–Extended Mode) BCLKT
Falling Edge, or the Falling Edge of FST to DT Output High Impedance
10
—
60
ns
19
Minimum Pulse Width Low for FST or FSR
50
—
—
ns
SHORT FRAME SPECIFIC TIMING
20
Hold Time from BCLKT (BCLKR) Low to FST (FSR) Low
50
—
—
ns
21
Setup Time from FST (FSR) Low to MSB Period of BCLKT (BCLKR) Low
50
—
—
ns
22
Delay Time from BCLKT High to DT Data Valid
10
—
60
ns
23
Delay Time from the 13th (16th for Sign–Extended Mode) BCLKT Low to DT
Output High Impedance
10
—
60
ns
MOTOROLA
MC145482
13
1
7
4
3
6
5
2
MCLK
8
1
BCLKT
2
3
4
5
12
6
7
13
14
9
11
15
10
FST
16
18
17
18
16
1
DT
2
3
4
5
6
13
8
BCLKR
(BCLKT)
1
2
3
11
4
5
15
6
7
13
14
9
12
10
FSR
14
13
DR
1
2
3
4
5
6
13
Figure 3. Long Frame Sync Timing
MC145482
14
MOTOROLA
1
7
4
3
6
5
2
MCLK
12
8
1
BCLKT
2
3
4
5
6
7
13
14
9
20
21
11
10
FST
23
22
22
1
DT
2
3
4
5
6
13
8
1
BCLKR
2
3
4
5
20
6
7
13
14
9
21
11
10
12
FSR
14
13
DR
1
2
3
4
5
6
13
Figure 4. Short Frame Sync Timing
MOTOROLA
MC145482
15
1
2
0.1 µF
2X20 k
AUDIO OUT
3
4
–
+
5
6
+5V
7
0.1 µF
8
9
10
VAG Ref
VAG
RO–
TI+
PI
TI–
17
PO+
HB 16
FSR
FST
DT
DR
BCLKT
BCLKR
MCLK
PDI
10 kΩ
1.0 µF
ANALOG IN
Y
18
TG
VSS
10 kΩ
19
PO–
VDD
0.01 µF
20
10 kΩ
10 kΩ
1.0 µF
15
14
8 kHz
13
PCM OUT
12
2.048 MHz
11
PCM IN
Figure 5. MC145482 Test Circuit — Signals Referenced to VAG Pin
0.1 µF
1
AUDIO OUT
2X20 k
RL ≥ 2 kΩ
2
3
68 µF
AUDIO OUT
RL ≥ 150 Ω
+
4
5
10 kΩ
+5V
6
7
0.1 µF
8
9
10
VAG Ref
VAG
TI+
PI
TI–
PO–
TG
PO+
HB 16
VSS
FSR
FST
DR
DT
BCLKR
PDI
BCLKT
MCLK
10 kΩ
10 kΩ
19
RO–
VDD
0.01 µF
20
1.0 µF
18
17
10 kΩ
10 kΩ
ANALOG IN
Y
1.0 µF
15
14
13
12
8 kHz
PCM OUT
2.048 MHz
11
PCM IN
Figure 6. MC145482 Test Circuit — Signals Referenced to VSS
MC145482
16
MOTOROLA
+3 V
1 kΩ
SIDETONE
0.01 µF
0.1 µF
68 µF
1 kΩ
420 pF
1
VAG Ref
2
3
4
REC
5
6
+5V
7
0.1 µF
8
9
10
VAG
20
RO–
TI+
PI
TI–
PO–
TG
PO+
HB
VDD
VSS
FSR
FST
DR
DT
13
BCLKT
12
BCLKR
PDI
MCLK
75 kΩ
1 kΩ 0.1 µF
19
17
MIC
1 kΩ
18
0.1 µF
75 kΩ
16
15
420 pF
14
8 kHz
PCM OUT
2.048 MHz
11
PCM IN
Figure 7. MC145482 Handset Interface
1.0 µF
R0 = 600 Ω
10 kΩ
0.1 µF
TIP
1
2X20 k
N = 0.5
1/4 R0
N = 0.5
2
3
4
5
– 48 V
N = 0.5
+5V
6
7
RING
0.1 µF
8
9
10
VAG Ref
VAG
RO–
TI+
PI
TI–
PO–
TG
PO+
HB
VDD
VSS
FSR
FST
DR
DT
BCLKR
PDI
BCLKT
MCLK
20
19
18
17
0.1 µF
20 kΩ
16
15
14
13
12
8 kHz
PCM OUT
2.048 MHz
11
PCM IN
Figure 8. MC145482 Step–Up Transformer Line Interface
MOTOROLA
MC145482
17
PACKAGE DIMENSIONS
DW SUFFIX
SOG PACKAGE
CASE 751D–04
NOTES:
1. DIMENSIONING AND TOLERANCING PER
ANSI Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSIONS A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.150
(0.006) PER SIDE.
5. DIMENSION D DOES NOT INCLUDE
DAMBAR PROTRUSION. ALLOWABLE
DAMBAR PROTRUSION SHALL BE 0.13
(0.005) TOTAL IN EXCESS OF D DIMENSION
AT MAXIMUM MATERIAL CONDITION.
–A–
20
11
–B–
10X
P
0.010 (0.25)
1
M
B
M
10
20X
D
0.010 (0.25)
M
T A
B
S
DIM
A
B
C
D
F
G
J
K
M
P
R
J
S
F
R
X 45 _
C
–T–
18X
G
SEATING
PLANE
MILLIMETERS
MIN
MAX
12.65
12.95
7.40
7.60
2.35
2.65
0.35
0.49
0.50
0.90
1.27 BSC
0.25
0.32
0.10
0.25
0_
7_
10.05
10.55
0.25
0.75
INCHES
MIN
MAX
0.499
0.510
0.292
0.299
0.093
0.104
0.014
0.019
0.020
0.035
0.050 BSC
0.010
0.012
0.004
0.009
0_
7_
0.395
0.415
0.010
0.029
M
K
SD SUFFIX
SSOP
CASE 940C–02
20
NOTES:
1. CONTROLLING DIMENSION: MILLIMETER.
2. DIMENSIONS AND TOLERANCES PER ANSI
Y14.5M, 1982.
3. DIMENSIONS A AND B DO NOT INCLUDE
MOLD FLASH OR PROTRUSIONS AND ARE
MEASURED AT THE PARTING LINE. MOLD
FLASH OR PROTRUSIONS SHALL NOT
EXCEED 0.15MM PER SIDE.
4. DIMENSION IS THE LENGTH OF TERMINAL
FOR SOLDERING TO A SUBSTRATE.
5. TERMINAL POSITIONS ARE SHOWN FOR
REFERENCE ONLY.
6. THE LEAD WIDTH DIMENSION DOES NOT
INCLUDE DAMBAR PROTRUSION.
ALLOWABLE DAMBAR PROTRUSION SHALL
BE 0.08MM TOTAL IN EXCESS OF THE LEAD
WIDTH DIMENSION.
11
B
–R–
1
C
10
0.076 (0.003)
A
–P–
N
0.25 (0.010)
M
R
M
L
J
M
G
H
MC145482
18
D
0.120 (0.005)
F
NOTE 4
M
T P
DIM
A
B
C
D
F
G
H
J
L
M
N
MILLIMETERS
MIN
MAX
7.10
7.30
5.20
5.38
1.75
1.99
0.25
0.38
0.65
1.00
0.65 BSC
0.59
0.75
0.10
0.20
7.65
7.90
0_
8_
0.05
0.21
INCHES
MIN
MAX
0.280
0.287
0.205
0.212
0.069
0.078
0.010
0.015
0.026
0.039
0.026 BSC
0.023
0.030
0.004
0.008
0.301
0.311
0_
8_
0.002
0.008
S
MOTOROLA
Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty, representation or guarantee regarding
the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and
specifically disclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters which may be provided in Motorola
data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals”
must be validated for each customer application by customer’s technical experts. Motorola does not convey any license under its patent rights nor the rights of
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Mfax is a trademark of Motorola, Inc.
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MOTOROLA
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MC145482/D
MC145482
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