LINER LT5557 400mhz to 3.8ghz 3.3v high signal level downconverting mixer Datasheet

LT5557
400MHz to 3.8GHz
3.3V High Signal Level
Downconverting Mixer
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FEATURES
DESCRIPTIO
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The LT®5557 active mixer is optimized for high linearity,
wide dynamic range downconverter applications. The IC
includes a high speed differential LO buffer amplifier
driving a double-balanced mixer. Broadband, integrated
transformers on the RF and LO inputs provide singleended 50Ω interfaces. The differential IF output allows
convenient interfacing to differential IF filters and amplifiers, or is easily matched to drive a single-ended 50Ω load,
with or without an external transformer.
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■
■
■
■
■
■
■
■
3.3V Operation for Reduced Power
50Ω Single-Ended RF and LO Ports
Wide RF Frequency Range: 400MHz to 3.8GHz*
High Input IP3: 25.6dBm at 900MHz
24.7dBm at 1950MHz
23.7dBm at 2.6GHz
Conversion Gain: 3.3dB at 900MHz
2.9dB at 1950MHz
–3dBm LO Drive Level
Low LO Leakage
Low Noise Figure: 10.6dB at 900MHz
11.7dB at 1950MHz
Very Few External Components
16-Lead (4mm × 4mm) QFN Package
The RF input is internally matched to 50Ω from 1.6GHz to
2.3GHz, and the LO input is internally matched to 50Ω
from 1GHz to 5GHz. The frequency range of both ports is
easily extended with simple external matching. The IF
output is partially matched and usable for IF frequencies
up to 600MHz.
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APPLICATIO S
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The LT5557’s high level of integration minimizes the total
solution cost, board space and system-level variation.
Cellular, CDMA, WCDMA, TD-SCDMA and UMTS
Infrastructure
WiMAX
Wireless Infrastructure Receiver
Wireless Infrastructure PA Linearization
900MHz/2.4GHz/3.5GHz WLAN
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
*Operation over a wider frequency range is possible with reduced performance. Consult factory for
information and assistance.
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TYPICAL APPLICATIO
High Signal Level Downmixer for Multi-Carrier Wireless Infrastructure
Conversion Gain, IIP3, SSB NF and
LO Leakage vs RF Frequency
LO INPUT
–3dBm (TYP)
26
24
22
100nH
IF+
1nF
150nH
RF
INPUT
RF
IF –
BIAS
GND
EN
4.7pF
VCC2 VCC1
18
16
14
12
1μF
5557 TA01a
10
LOW-SIDE LO
IF = 240MHz
PLO = –3dBm
TA = 25°C
VCC = 3.3V
20
20
30
SSB NF
40
10
LO-IF
8
6
4
100nH
3.3V
1nF
IF
OUTPUT
240MHz
0
IIP3
2
1700
LO-RF
50
GC
1800
LO LEAKAGE (dBm)
4.7pF
GC (dB), NF (dB), IIP3 (dBm)
LT5557
60
2100
2000
1900
RF FREQUENCY (MHz)
2200
5557 TA01b
5557fa
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LT5557
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AXI U
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PACKAGE/ORDER I FOR ATIO
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ABSOLUTE
RATI GS
(Note 1)
CAUTION: This part is sensitive to electrostatic discharge
(ESD). It is very important that proper ESD precautions be
observed when handling the LT5557.
ORDER PART
NUMBER
NC
NC
LO
NC
TOP VIEW
16 15 14 13
NC 1
12 GND
NC 2
RF 3
LT5557EUF#PBF
11 IF+
17
10
IF–
9 GND
6
7
8
VCC2
NC
5
VCC1
NC 4
EN
Supply Voltage (VCC1, VCC2, IF+, IF–) ......................... 4V
Enable Voltage ............................... –0.3V to VCC + 0.3V
LO Input Power (380MHz to 4.2GHz) ............... +10dBm
LO Input DC Voltage ............................ –1V to VCC + 1V
RF Input Power (400MHz to 3.8GHz) ............... +12dBm
RF Input DC Voltage ............................................ ±0.1V
Operating Temperature Range ............... – 40°C to 85°C
Storage Temperature Range ................ – 65°C to 125°C
Junction Temperature (TJ)................................... 125°C
UF PART MARKING
UF PACKAGE
16-LEAD (4mm × 4mm) PLASTIC QFN
5557
TJMAX = 125°C, θJA = 37°C/W
EXPOSED PAD (PIN 17) IS GND
MUST BE SOLDERED TO PCB
Order Options Tape and Reel: Add #TR
Lead Free Part Marking: http://www.linear.com/leadfree/
Consult LTC Marketing for parts specified with wider operating temperature ranges.
DC ELECTRICAL CHARACTERISTICS
VCC = 3.3V, EN = High, TA = 25°C, unless otherwise specified. Test circuit shown in Figure 1. (Note 3)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
2.9
3.3
3.9
V
60
92
mA
mA
mA
mA
100
μA
Power Supply Requirements (VCC)
Supply Voltage
Supply Current
VCC1 (Pin 7)
VCC2 (Pin 6)
IF+ + IF– (Pin 11 + Pin 10)
Total Supply Current
25.1
3.3
53.2
81.6
Enable (EN) Low = Off, High = On
Shutdown Current
EN = Low
Input High Voltage (On)
2.7
V
Input Low Voltage (Off)
EN Pin Input Current
EN = 3.3V DC
53
0.3
V
90
μA
Turn-ON Time
2.8
μs
Turn-OFF Time
2.9
μs
AC ELECTRICAL CHARACTERISTICS
Test circuit shown in Figure 1. (Notes 2, 3)
PARAMETER
CONDITIONS
RF Input Frequency Range
No External Matching (Midband)
With External Matching (Low Band or High Band)
400
No External Matching
With External Matching
380
LO Input Frequency Range
IF Output Frequency Range
Requires Appropriate IF Matching
RF Input Return Loss
ZO = 50Ω, 1600MHz to 2300MHz (No External Matching)
LO Input Return Loss
ZO = 50Ω, 1000MHz to 5000MHz (No External Matching)
IF Output Impedance
Differential at 240MHz
LO Input Power
1200MHz to 4200MHz
380MHz to 1200MHz
MIN
TYP
MAX
UNITS
3800
MHz
MHz
1600 to 2300
1000 to 4200
MHz
MHz
0.1 to 600
MHz
>12
–8
–5
dB
>10
dB
529Ω||2.6pF
R||C
–3
0
2
5
dBm
dBm
5557fa
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LT5557
AC ELECTRICAL CHARACTERISTICS
Standard Downmixer Application: VCC = 3.3V, EN = High, TA = 25°C,
PRF = – 6dBm (–6dBm/tone for 2-tone IIP3 tests, Δf = 1MHz), fLO = fRF – fIF, PLO = –3dBm (0dBm for 450MHz and 900MHz tests),
IF output measured at 240MHz, unless otherwise noted. Test circuit shown in Figure 1. (Notes 2, 3, 4)
PARAMETER
CONDITIONS
Conversion Gain
RF = 450MHz, IF = 70MHz, High Side LO
RF = 900MHz, IF = 140MHz
RF = 1750MHz
RF = 1950MHz
RF = 2150MHz
RF = 2600MHz, IF = 360MHz
RF = 3600MHz, IF = 450MHz
MIN
Conversion Gain vs Temperature
TA = – 40°C to 85°C, RF = 1950MHz
Input 3rd Order Intercept
TYP
2.9
3.3
3.0
2.9
2.9
2.5
1.7
MAX
UNITS
dB
dB
dB
dB
dB
dB
dB
–0.0217
dB/°C
RF = 450MHz, IF = 70MHz, High Side LO
RF = 900MHz, IF = 140MHz
RF = 1750MHz
RF = 1950MHz
RF = 2150MHz
RF = 2600MHz, IF = 360MHz
RF = 3600MHz, IF = 450MHz
24.1
25.6
25.5
24.7
24.3
23.7
23.5
dBm
dBm
dBm
dBm
dBm
dBm
dBm
Single-Sideband Noise Figure
RF = 450MHz, IF = 70MHz, High Side LO
RF = 900MHz, IF = 140MHz
RF = 1750MHz
RF = 1950MHz
RF = 2150MHz
RF = 2600MHz, IF = 360MHz
RF = 3600MHz, IF = 450MHz
12.7
10.6
11.3
11.7
12.8
13.2
15.4
dB
dB
dB
dB
dB
dB
dB
LO to RF Leakage
fLO = 380MHz to 1600MHz
fLO = 1600MHz to 4000MHz
<–50
<–45
dBm
dBm
LO to IF Leakage
fLO = 380MHz to 2200MHz
fLO = 2200MHz to 4000MHz
<–42
<–38
dBm
dBm
RF to LO Isolation
fRF = 400MHz to 1700MHz
fRF = 1700MHz to 3800MHz
>50
>42
dB
dB
RF to IF Isolation
fRF = 400MHz to 2300MHz
fRF = 2300MHz to 3800MHz
>41
>37
dB
dB
2RF-2LO Output Spurious Product
(fRF = fLO + fIF/2)
900MHz: fRF = 830MHz at –6dBm, fIF = 140MHz
1950MHz: fRF = 1830MHz at –6dBm, fIF = 240MHz
–61
–53
dBc
dBc
3RF-3LO Output Spurious Product
(fRF = fLO + fIF/3)
900MHz: fRF = 806.67MHz at –6dBm, fIF = 140MHz
1950MHz: fRF = 1790MHz at –6dBm, fIF = 240MHz
–83
–70
dBc
dBc
Input 1dB Compression
RF = 450MHz, IF = 70MHz, High Side LO
RF = 900MHz, IF = 140MHz
RF = 1950MHz
RF = 2600MHz, IF = 360MHz
RF = 3600MHz, IF = 450MHz
10.0
8.8
8.8
8.6
9.1
dBm
dBm
dBm
dBm
dBm
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: 450MHz and 900MHz performance measured with external LO and
RF matching. 2600MHz and 3600MHz performance measured with
external RF matching. See Figure 1 and Applications Information.
Note 3: Specifications over the –40°C to 85°C temperature range are
assured by design, characterization and correlation with statistical process
controls.
Note 4: SSB Noise Figure measurements performed with a small-signal
noise source and bandpass filter on RF input, and no other RF signal
applied.
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LT5557
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TYPICAL PERFOR A CE CHARACTERISTICS
VCC = 3.3V, Test circuit shown in Figure 1.
Midband (No external RF/LO matching) 240MHz IF output, PRF = –6dBm (–6dBm/tone for 2-tone IIP3 tests, Δf = 1MHz), PLO = –3dBm,
unless otherwise noted.
Conversion Gain, IIP3 and NF
vs RF Frequency
LO Leakage and RF Isolation vs
LO and RF Frequency
26
24
86
16
14
12
SSB NF
8
35
–40
LO-IF
25
–50
4
GC
TA = 25°C
IF = 240MHz
2
1.6
1.7
1.8 1.9 2.0 2.1
RF FREQUENCY (GHz)
2.2
LO-RF
–60
1.2
2.3
GC (dB), IIP3 (dBm)
GC (dB), IIP3 (dBm)
IIP3
GC
–25
25
50
0
TEMPERATURE (°C)
75
100
27
25
23
21
19
17
15
13
11
9
7
5
3
1
–50
–9
–7
–3
–1
1
–5
LO INPUT POWER (dBm)
3
5557 G07
GC (dB), NF (dB), IIP3 (dBm)
IIP3
GC
–40°C
3.1
3.5
3.7
3.3
SUPPLY VOLTAGE (V)
3.9
1950MHz Conversion Gain, IIP3
and NF vs Supply Voltage
1750MHz
1950MHz
2150MHz
IF = 240MHz
GC
–25
25
50
0
TEMPERATURE (°C)
75
100
26
24
22
20
18
16
14
12
10
8
6
4
2
0
2.9
IIP3
–40°C
25°C
85°C
SSB NF
LOW-SIDE LO
IF = 240MHz
GC
3.1
3.5
3.7
3.3
SUPPLY VOLTAGE (V)
26
24
22
20
18
16
14
12
10
8
6
4
2
0
2150MHz Conversion Gain, IIP3
and NF vs LO Power
IIP3
–40°C
25°C
85°C
SSB NF
LOW-SIDE LO
IF = 240MHz
GC
–9
–7
–3
–1
1
–5
LO INPUT POWER (dBm)
3.9
5557 G06
1950MHz Conversion Gain, IIP3
and NF vs LO Power
LOW-SIDE LO
IF = 240MHz
–10°C
80
5557 G05
1750MHz Conversion Gain, IIP3
and NF vs LO Power
SSB NF
81
5557 G03
IIP3
5557 G04
–40°C
25°C
85°C
25°C
77
2.9
Conversion Gain and IIP3
vs Temperature (High-Side LO)
IF = 240MHz
60°C
82
5557 G02
Conversion Gain and IIP3
vs Temperature (Low-Side LO)
1750MHz
1950MHz
2150MHz
83
78
15
2.7
1.5
2.1
2.4
1.8
LO/RF FREQUENCY (GHz)
5557 G01
27
25
23
21
19
17
15
13
11
9
7
5
3
1
–50
85°C
84
79
TA = 25°C
PLO = –3dBm
GC (dB), NF (dB), IIP3 (dBm)
10
85
45
RF-LO
RF-IF
3
5557 G08
GC (dB), NF (dB), IIP3 (dBm)
18
–30
SUPPLY CURRENT (mA)
LOW-SIDE LO
HIGH-SIDE LO
LO LEAKAGE (dBm)
20
6
GC (dB), NF (dB), IIP3 (dBm)
87
RF ISOLATION (dB)
GC (dB), NF (dB), IIP3 (dBm)
55
IIP3
22
27
25
23
21
19
17
15
13
11
9
7
5
3
1
Supply Current vs Supply Voltage
–20
26
24
22
20
18
16
14
12
10
8
6
4
2
0
IIP3
–40°C
25°C
85°C
SSB NF
LOW-SIDE LO
IF = 240MHz
GC
–9
–7
–3
–1
1
–5
LO INPUT POWER (dBm)
3
5557 G09
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LT5557
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TYPICAL PERFOR A CE CHARACTERISTICS
VCC = 3.3V, Test circuit shown in Figure 1.
Midband (No external RF/LO matching) 240MHz IF output, PRF = –6dBm (–6dBm/tone for 2-tone IIP3 tests, Δf = 1MHz), PLO = –3dBm,
unless otherwise noted.
IFOUT, 2 × 2 and 3 × 3 Spurs
vs RF Input Power (Single Tone)
IF Output Power, IM3 and IM5 vs
RF Input Power (2 Input Tones)
10
15
–10
–20
–30
TA = 25°C
RF1 = 1949.5MHz
RF2 = 1950.5MHz
LO = 1710MHz
–50
–60
–70
TA = 25°C
LO = 1710MHz
IF = 240MHz
–15
–25
–35
–45
2RF-2LO
(RF = 1830MHz)
–55
–65
–75
–80
IM3
–85
IM5
–95
–15 –12 –9 –6 –3 0
3
6
RF INPUT POWER (dBm)
0
–15
–12
–9
–6
–3
RF INPUT POWER (dBm/TONE)
35
25
20
15
–70
3RF-3LO
(RF = 1790MHz)
–9
–7
–5
–1
1
–3
LO INPUT POWER (dBm)
30
85°C
25°C
–40°C
25
20
TA = 25°C
LOW-SIDE LO
IF = 240MHz
24
15
10
21
18
15
12
9
6
5
5
SSB Noise Figure Distribution at
1950MHz
27
LOW-SIDE LO
IF = 240MHz
3
5557 G12
10
0
–65
12
IIP3 Distribution at 1950MHz
30
DISTRIBUTION (%)
30
–60
5557 G11
TA = 25°C
LOW-SIDE LO
IF = 240MHz
35
2RF-2LO
(RF = 1830MHz)
–55
–80
9
5557 G10
Conversion Gain Distribution at
1950MHz
40
–50
–75
3RF-3LO
(RF = 1790MHz)
DISTRIBUTION (%)
–90
–100
–18
TA = 25°C
LO = 1710MHz
IF = 240MHz
PRF = –6dBm
–45
RELATIVE SPUR LEVEL (dBc)
IFOUT
–40
–40
IFOUT
5 (RF = 1950MHz)
–5
OUTPUT POWER (dBm)
OUTPUT POWER/TONE (dBm)
0
DISTRIBUTION (%)
2 × 2 and 3 × 3 Spurs
vs LO Power (Single Tone)
3
0
2.6
2.7
2.9
2.8
3.0
3.1
CONVERSION GAIN (dB)
23
3.2
24
26
25
IIP3 (dBm)
0
28
27
11.0
11.2
11.4 11.6 11.8 12.0
SSB NOISE FIGURE (dB)
5557 G26
5557 G25
12.2
5557 G27
450MHz application (with external RF/LO matching) 70MHz IF output, PRF = –6dBm (–6dBm/tone for 2-tone IIP3 tests,
Δf = 1MHz), high-side LO at 0dBm, unless otherwise noted.
Conversion Gain, IIP3 and NF
vs RF Frequency
450MHz Conversion Gain,
IIP3 and NF vs LO Power
25
18
16
SSB NF
12
10
8
17
15
SSB NF
13
11
HIGH-SIDE LO
IF = 70MHz
9
7
GC
500
5557 G13
900MHz
APPLICATION
–50
450MHz
APPLICATION
1
475
450
425
RF FREQUENCY (MHz)
–45
–55
3
GC
LO-RF
LO-IF
TA = 25°C
PLO = 0dBm
–40
–40°C
25°C
85°C
19
5
6
2
400
GC (dB), NF (dB), IIP3 (dBm)
GC (dB), NF (dB), IIP3 (dBm)
HIGH-SIDE LO
TA = 25°C
IF = 70MHz
20
4
IIP3
21
22
14
–35
23
IIP3
LO LEAKAGE (dBm)
26
24
LO Leakage vs LO Frequency
450MHz and 900MHz Applications
–6
–4
2
0
–2
LO INPUT POWER (dBm)
4
6
5557 G14
–60
400
800
600
1000
LO FREQUENCY (MHz)
1200
5557 G15
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LT5557
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TYPICAL PERFOR A CE CHARACTERISTICS
15
27
25
23
21
19
17
15
13
11
9
7
5
3
1
5
IIP3
–5
OUTPUT POWER (dBm)
28
26
IIP3
24
22
LOW-SIDE LO
20
TA = 25°C
18
IF = 140MHz
16
14
12
SSB NF
10
8
6
GC
4
2
800
900
750
950 1000 1050
850
RF FREQUENCY (MHz)
GC (dB), NF (dB), IIP3 (dBm)
GC (dB), NF (dB), IIP3 (dBm)
VCC = 3.3V, Test circuit shown in Figure 1.
900MHz application (with external RF/LO matching), 140MHz IF output, PRF = –6dBm (–6dBm/tone for 2-tone IIP3 tests, Δf = 1MHz),
low-side LO at 0dBm, unless otherwise noted.
900MHz Conversion Gain, IIP3 and
IFOUT, 2 × 2 and 3 × 3 Spurs
Conversion Gain, IIP3 and NF vs
NF vs LO Power
vs RF Input Power (Single-Tone)
RF Frequency
–40°C
25°C
85°C
SSB NF
LOW-SIDE LO
IF = 140MHz
GC
IFOUT
(RF = 900MHz)
–15
–25
TA = 25°C
LO = 760MHz
IF = 140MHz
–35
–45
2RF-2LO
(RF = 830MHz)
–55
–65
–75
3RF-3LO
(RF = 806.67MHz)
–95
–15 –12 –9 –6 –3 0
3
6
RF INPUT POWER (dBm)
–85
LO INPUT POWER (dBm)
5557 G16
9
12
5557 G18
5557 G17
26
24
22
20
18
16
14
12
10
8
6
4
2
0
–20
45
RF-LO
RF-IF
IIP3
–30
–40°C
25°C
85°C
SSB NF
LOW-SIDE LO
LO LEAKAGE (dBm)
GC (dB), NF (dB), IIP3 (dBm)
26
24
IIP3
22
LOW-SIDE LO
20
HIGH-SIDE LO
18
16
14
12 SSB NF
10
8
TA = 25°C
6
GC
4
2
0
2.4
2.6
2.3
2.7
2.5
RF FREQUENCY (GHz)
35
–40
25
LO-RF
–50
15
LO-IF
GC
–60
–9
–7
RF ISOLATION (dB)
GC (dB), NF (dB), IIP3 (dBm)
2.3-2.7GHz application (with external RF matching) 360MHz IF output, PRF = –6dBm (–6dBm/tone for 2-tone IIP3 tests, Δf = 1MHz),
PLO = –3dBm, unless otherwise noted.
LO Leakage and RF Isolation vs
Conversion Gain, IIP3 and SSB
2.6GHz Conversion Gain, IIP3 and
LO and RF Frequency
NF vs RF Frequency
NF vs LO Power
–3
–1
1
–5
LO INPUT POWER (dBm)
5
1.9
3
2.1
2.3
2.5
2.7
2.9
LO/RF FREQUENCY (GHz)
5557 G21
5557 G20
5557 G19
3.1
3.3-3.8GHz application (with external RF matching) 450MHz IF output, PRF = –6dBm (–6dBm/tone for 2-tone IIP3 tests, Δf = 1MHz),
low-side LO at –3dBm, unless otherwise noted.
LO Leakage and RF Isolation vs LO
Conversion Gain, IIP3 and SSB NF
3.6GHz Conversion Gain, IIP3 and
vs RF Frequency
SSB NF vs LO Power
and RF Frequency
24
IIP3
22
10
8
6
TA = 25°C
4
GC
2
GC (dB), NF (dB), IIP3 (dBm)
14
12
RF-LO
–40
18
SSB NF
16
14
12
–40°C
25°C
85°C
10
8
6
35
LO-IF
25
LO-RF
GC
2
0
0
3.3
–50
–60
4
45
RF-IF
RF ISOLATION (dB)
18
16 SSB NF
55
IIP3
20
20
GC (dB), NF (dB), IIP3 (dBm)
–30
22
LO LEAKAGE (dBm)
24
3.4
3.7
3.6
3.5
RF FREQUENCY (GHz)
3.8
5557 G22
–9
–7
–1
–3
–5
LO INPUT POWER (dBm)
1
3
5557 G23
–70
2.8
3.0
3.4
3.6
3.2
LO/RF FREQUENCY (GHz)
3.8
15
5557 G24
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LT5557
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PI FU CTIO S
NC (Pins 1, 2, 4, 8, 13, 14, 16): Not Connected Internally.
These pins should be grounded on the circuit board for the
best LO-to-RF and LO-to-IF isolation.
be externally connected to the VCC2 pin and decoupled
with 1000pF and 1μF capacitors.
GND (Pins 9, 12): Ground. These pins are internally
connected to the backside ground for improved isolation.
They should be connected to the RF ground on the circuit
board, although they are not intended to replace the
primary grounding through the backside contact of the
package.
RF (Pin 3): Single-Ended Input for the RF Signal. This pin
is internally connected to the primary side of the RF input
transformer, which has low DC resistance to ground. If the
RF source is not DC blocked, then a series blocking
capacitor must be used. The RF input is internally matched
from 1.6GHz to 2.3GHz. Operation down to 400MHz or up
to 3.8GHz is possible with simple external matching.
IF–, IF + (Pins 10, 11): Differential Outputs for the IF
Signal. An impedance transformation may be required to
match the outputs. These pins must be connected to VCC
through impedance matching inductors, RF chokes or a
transformer center tap. Typical current consumption is
26.6mA each (53.2mA total).
EN (Pin 5): Enable Pin. When the input enable voltage is
higher than 2.7V, the mixer circuits supplied through Pins
6, 7, 10 and 11 are enabled. When the input voltage is less
than 0.3V, all circuits are disabled. Typical input current is
53μA for EN = 3.3V and 0μA when EN = 0V. The EN pin
should not be left floating. Under no conditions should the
EN pin voltage exceed VCC + 0.3V, even at start-up.
LO (Pin 15): Single-Ended Input for the Local Oscillator
Signal. This pin is internally connected to the primary side
of the LO transformer, which is internally DC blocked. An
external blocking capacitor is not required. The LO input is
internally matched from 1GHz to 5GHz. Operation down to
380MHz is possible with simple external matching.
VCC2 (Pin 6): Power Supply Pin for the Bias Circuits.
Typical current consumption is 3.3mA. This pin should be
externally connected to the VCC1 pin and decoupled with
1000pF and 1μF capacitors.
Exposed Pad (Pin 17): Circuit Ground Return for the
Entire IC. This must be soldered to the printed circuit board
ground plane.
VCC1 (Pin 7): Power Supply Pin for the LO Buffer Circuits.
Typical current consumption is 25.1mA. This pin should
W
BLOCK DIAGRA
15
LO
REGULATOR
LIMITING
AMPLIFIERS
VREF
GND 12
VCC1
3
EXPOSED
17
PAD
IF+
IF–
RF
DOUBLE-BALANCED
MIXER
11
10
GND 9
BIAS
EN
5
VCC1
VCC2
6
7
5557 BD
5557fa
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LT5557
TEST CIRCUITS
LOIN
L4
EXTERNAL MATCHING
FOR LO BELOW 1GHz
2
ZO
50Ω
14
NC
BIAS
13
0.015"
NC
NC
GND
NC
IF
T1
3
+ 11
C3
IF –
RF
4
GND
NC
EN
4
2
L1
10
1
9
•
•
IFOUT
240MHz
6
VCC2 VCC1 NC
5
EN
C5
3.9pF
GND
12
LT5557
3
L (mm)
C5
RFIN
15
LO
NC
1
LOW-PASS MATCH
FOR 450MHz, 900MHz
AND 3.6GHz RF
DC1131A
BOARD
STACK-UP
(NELCO N4000-13)
0.062"
16
RFIN
RF
GND
εR = 3.7
0.015"
C4
6
7
8
VCC (2.9V to 3.9V)
C1
C2
L5
3.6nH
5557 F01
*HIGH-PASS MATCH
FOR 2.6GHz RF
APPLICATION
RF
LO
RF MATCH
IF
LO MATCH
IF MATCH
L
C5
L4
C4
L1
C3
450MHz High Side 70MHz
6.5mm
12pF
10nH
8.2pF
270nH
15pF
900MHz Low Side 140MHz
1.7mm
3.9pF
2.7nH
3.9pF
180nH
3.9pF
–
–
47nH
1.2pF
–
–
39nH
–
2.6GHz
360MHz
3.6GHz
450MHz
HIGH-PASS*
2.9mm
1pF
REF DES
VALUE
SIZE
PART NUMBER
REF DES
VALUE
C1
1000pF
0402
AVX 04025C102JAT
L4, C4, C5
C2
1μF
0603
AVX 0603ZD105KAT
L1
82nH
C3
2.2pF
0402
AVX 04025A2R2BAT
T1
8:1
SIZE
PART NUMBER
0402
See Applications Information
0603
Toko LLQ1608-F82NG
Mini-Circuits TC8-1+
Figure 1. Standard Downmixer Test Schematic—Transformer-Based Bandpass IF Matching (240MHz IF)
LOIN
L4
0.018"
DISCRETE
IF BALUN
C4
16
EXTERNAL MATCHING
FOR LO BELOW 1GHz
1
2
RFIN
ZO
50Ω
L (mm)
C5
NC
14
NC
13
NC
NC
GND
NC
IF
12
3
L1
L3
4
IF –
RF
GND
NC
EN
5
7
BIAS
0.018"
GND
DC910A
BOARD
STACK-UP
(FR-4)
C3
IFOUT
240MHz
C7
10
9
L2
VCC2 VCC1 NC
6
RF
GND
C6
+ 11
LT5557
EN
LOW-PASS MATCH
FOR 450MHz, 900MHz
AND 3.6GHz RF
15
LO
εR = 4.4
0.062"
8
VCC (2.9V to 3.9V)
C2
C1
5557 F02
REF DES
VALUE
SIZE
PART NUMBER
REF DES
C1, C3
1000pF
0402
AVX 04025C102JAT
L4, C4, C5
1μF
0603
AVX 0603ZD105KAT
L1, L2
100nH
0603
Toko LL1608-FSLR10J
4.7pF
0402
AVX 04025A4R7CAT
L3
150nH
0603
Toko LL1608-FSLR15J
C2
C6, C7
VALUE
SIZE
PART NUMBER
0402
See Applications Information
Figure 2. Downmixer Test Schematic—Discrete IF Balun Matching (240MHz IF)
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Introduction
The LT5557 consists of a high linearity double-balanced
mixer, RF buffer amplifier, high speed limiting LO buffer
amplifier and bias/enable circuits. The RF and LO inputs
are both single ended. The IF output is differential. Low
side or high side LO injection can be used.
Two evaluation circuits are available. The standard evaluation circuit, shown in Figure 1, incorporates transformerbased IF matching and is intended for applications that
require the highest dynamic range and the widest IF
bandwidth. The second evaluation circuit, shown in Figure 2, replaces the IF transformer with a discrete IF balun
for reduced solution cost and size. The discrete IF balun
delivers higher conversion gain, but slightly degraded IIP3
and noise figure, and reduced IF bandwidth.
band edge can be optimized with a series 3.9pF capacitor
at Pin 3, which improves the 1.6GHz return loss to greater
than 25dB. Likewise, the 2.3GHz match can be improved
to greater than 25dB with a series 1.5nH inductor. A series
2.7nH/2.2pF network will simultaneously optimize the lower
and upper band edges and expand the RF input bandwidth
to 1.2GHz-2.5GHz. Measured RF input return losses for
these three cases are also plotted in Figure 4a.
Alternatively, the input match can be shifted as low as
400MHz or up to 3800MHz by adding a shunt capacitor (C5)
to the RF input. A 450MHz input match is realized with C5
= 12pF, located 6.5mm away from Pin 3 on the evaluation
board’s 50Ω input transmission line. A 900MHz input match
requires C5 = 3.9pF, located at 1.7mm. A 3.6GHz input
match is realized with C5 = 1pF, located at 2.9mm. This
0
RF Input Port
The RF input is internally matched from 1.6GHz to 2.3GHz,
requiring no external components over this frequency
range. The input return loss, shown in Figure 4a, is typically 12dB at the band edges. The input match at the lower
LOW-PASS MATCH
FOR 450MHz, 900MHz
and 3.6GHz RF
RFIN
TO
MIXER
ZO = 50Ω
L = L (mm)
3
RF
C5
–10
–15
–20
SERIES 2.7nH
AND 2.2pF
–25
SERIES 3.9pF
–30
0.2
0.7
L5
HIGH-PASS MATCH
FOR 2.6GHz RF
AND WIDEBAND RF
3.7
4.2
0
–5
–10
–15
450MHz
L = 6.5mm
C5 = 12pF
900MHz
L = 1.7mm
C5 = 3.9pF
–20
–25
C5
1.7 2.2 2.7 3.2
FREQUENCY (GHz)
5557 F04a
5557 F03
RFIN
1.2
SERIES 1.5nH
(4a) Series Reactance Matching
RF PORT RETURN LOSS (dB)
The mixer’s RF input, shown in Figure 3, consists of an
integrated transformer and a high linearity differential
amplifier. The primary terminals of the transformer are
connected to the RF input (Pin 3) and ground. The secondary side of the transformer is internally connected to the
amplifier’s differential inputs. The DC resistance of the
primary is 4.2Ω. If the RF source has DC voltage present,
then a coupling capacitor must be used in series with the
RF input pin.
RF PORT RETURN LOSS (dB)
NO EXT MATCH
–5
–30
0.2
0.7
1.2
3.6GHz
L = 2.9mm
C5 = 1pF
NO EXT
MATCH
1.7 2.2 2.7 3.2
FREQUENCY (GHz)
3.7
2.6GHz
SERIES 3.9pF
SHUNT 3.6nH
4.2
5557 F04b
(4b) Series Shunt Matching
Figure 3. RF Input Schematic
Figure 4. RF Input Return Loss With
and Without External Matching
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series transmission line/shunt capacitor matching topology allows the LT5557 to be used for multiple frequency
standards without circuit board layout modifications. The
series transmission line can also be replaced with a series
chip inductor for a more compact layout.
Input return losses for the 450MHz, 900MHz, 2.6GHz and
3.6GHz applications are plotted in Figure 4b. The input
return loss with no external matching is repeated in Figure
4b for comparison. The 2.6GHz RF input match uses the
high-pass matching network shown in Figures 1 and 3
with C5 = 3.9pF and L5 = 3.6nH. The high-pass input
matching network is also used to create a wideband or
dual-band input match. For example, with C5 = 3.3pF and
L5 = 10nH, the RF input is matched from 800MHz to
2.2GHz, with optimum matching in the 800MHz to 1.1GHz
and 1.6GHz to 2.2GHz bands, simultaneously.
RF input impedance and S11 versus frequency (with no
external matching) are listed in Table 1 and referenced to
Pin 3. The S11 data can be used with a microwave circuit
simulator to design custom matching networks and simulate board-level interfacing to the RF input filter.
LO Input Port
The mixer’s LO input, shown in Figure 5, consists of an
integrated transformer and high speed limiting differential
amplifiers. The amplifiers are designed to precisely drive
the mixer for the highest linearity and the lowest noise
figure. An internal DC blocking capacitor in series with the
transformer’s primary eliminates the need for an external
blocking capacitor.
The LO input is internally matched from 1 to 5GHz. The
input match can be shifted down, as low as 750MHz, with
a single shunt capacitor (C4) on Pin 15. One example is
plotted in Figure 6 where C4 = 2.7pF produces a 750MHz
to 1GHz match.
LO input matching below 750MHz requires the series
inductor (L4)/shunt capacitor (C4) network shown in
Figure 5. Two examples are plotted in Figure 6 where L4 =
2.7nH/C4 = 3.9pF produces a 650MHz to 830MHz match
and L4 = 10nH/C4 = 8.2pF produces a 460MHz to 560MHz
match. The evaluation boards do not include pads for L4,
so the circuit trace needs to be cut near Pin 15 to insert L4.
A low cost multilayer chip inductor is adequate for L4.
Table 1. RF Input Impedance vs Frequency
INPUT
IMPEDANCE
MAG
S11
ANGLE
50
4.6 + j2.3
0.832
174.7
300
9.1 + j11.2
0.706
153.8
450
12.0 + j14.5
0.639
145.8
600
14.7 + j17.4
0.588
138.7
900
20.5 + j23.3
0.506
123.4
1300
34.4 + j30.3
0.380
97.5
1700
59.6 + j23.8
0.229
55.8
1950
69.2 + j2.8
0.163
6.9
2200
59.2 – j18.1
0.184
–53.5
2450
41.5 – j24.5
0.274
–94.2
2700
28.3 – j21.3
0.374
–120.3
3000
19.0 – j13.5
0.481
–145.5
3300
13.9 – j5.1
0.568
–167.3
3600
10.8 + j3.4
0.645
171.9
3900
9.4 + j12.3
0.700
151.4
EXTERNAL
MATCHING
FOR LO < 1GHz
LOIN
TO
MIXER
L4
15
LO
C4
LIMITER
VREF
REGULATOR
VCC2
5557 F05
Figure 5. LO Input Schematic
0
LO PORT RETURN LOSS (dB)
FREQUENCY
(MHz)
–10
NO EXT
MATCH
–20
L4 = 10nH
C4 = 8.2pF
L4 = 2.7nH
C4 = 3.9pF
–30
0.3
L4 = 0
C4 = 2.7pF
1
5
LO FREQUENCY (GHz)
5557 G06
Figure 6. LO Input Return Loss
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The optimum LO drive is –3dBm for LO frequencies above
1.2GHz, although the amplifiers are designed to accommodate several dB of LO input power variation without
significant mixer performance variation. Below 1.2GHz,
0dBm LO drive is recommended for optimum noise figure,
although –3dBm will still deliver good conversion gain
and linearity.
Custom matching networks can be designed using the
port impedance data listed in Table 2. This data is referenced to the LO pin with no external matching.
Table 2. LO Input Impedance vs Frequency
S11
The IF output impedance can be modeled as 560Ω in
parallel with 2.6pF at low frequencies. An equivalent
small-signal model (including bondwire inductance) is
shown in Figure 8. Frequency-dependent differential IF
output impedance is listed in Table 3. This data is referenced to the package pins (with no external components)
and includes the effects of IC and package parasitics. The
IF output can be matched for IF frequencies as low as
several kHz or as high as 600MHz.
Table 3. IF Output Impedance vs Frequency
FREQUENCY (MHz)
DIFFERENTIAL OUTPUT
IMPEDANCE (RIF || XIF)
FREQUENCY
(MHz)
INPUT
IMPEDANCE
MAG
ANGLE
1
560 || – j63.7k (2.6pF)
50
10.0 – j326
0.991
–17.4
70
556 || – j870 (2.6pF)
300
8.5 – j41.9
0.820
–99.2
140
551 || – j440 (2.6pF)
523 || – j320 (2.6pF)
500
11.8 – j10.1
0.632
–155.9
190
700
18.8 + j10.9
0.474
151.8
240
529 || – j254 (2.6pF)
900
35.0 + j27.4
0.350
100.8
300
509 || – j200 (2.66pF)
1200
72.9 + j19.3
0.241
31.3
360
483 || – j163 (2.7pF)
1500
70.0 – j12.6
0.196
–26.1
450
448 || – j125 (2.83pF)
1800
55.0 – j17.0
0.167
–64.3
600
396 || – j92 (2.88pF)
2200
47.8 – j9.7
0.102
–97.2
2600
53.6 – j1.9
0.039
–26.8
3000
66.7 + j0.7
0.143
2.1
3500
82.1 – j13.9
0.263
–17.4
4000
69.0 – j30.1
0.290
–43.5
4500
43.7 – j13.2
0.154
–107.5
5000
36.4 + j19.8
0.271
111.6
Two methods of differential to single-ended IF matching
are described:
• Transformer - Based Bandpass
• Discrete IF balun
IF+
8:1
11
IF Output Port
C3
The IF outputs, IF+ and IF–, are internally connected to the
collectors of the mixer switching transistors (see Figure 7). Both pins must be biased at the supply voltage,
which can be applied through the center tap of a transformer or through matching inductors. Each IF pin draws
26.6mA of supply current (53.2mA total). For optimum
single-ended performance, these differential outputs
should be combined externally through an IF transformer
or a discrete IF balun circuit. The standard evaluation
board (see Figure 1) includes an IF transformer for
impedance transformation and differential to single-ended
transformation. A second evaluation board (see Figure 2)
realizes the same functionality with a discrete IF balun
circuit.
IFOUT
50Ω
L1 VCC
–
IF
10
VCC
5557 F07
Figure 7. IF Output with External Matching
0.7nH
RS
IF+
11
CS
RIF || XIF
IF
–
10
0.7nH
5557 F08
Figure 8. IF Output Small-Signal Model
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0
Transformer-Based Bandpass IF Matching
IF PORT RETURN LOSS (dB0
The standard evaluation board (shown in Figure 1) uses an
L-C bandpass IF matching network, with an 8:1 transformer
connected across the IF pins. The L-C network maximizes
mixer performance at the desired IF frequency. The
transformer performs impedance transformation and
provides a single-ended 50Ω output.
–10
–20
B
A
C
The value of L1 is calculated as:
–30
50
L1 = 1/[(2πfIF)2 • CIF]
where REFF, the effective IF resistance when loaded with
the transformer and inductor loss, is approximately 200Ω.
Below 40MHz, the magnitude of the internal IF reactance
is relatively high compared to the internal resistance. In
this case, L1 (and C3) can be eliminated, and the 8:1
transformer alone is adequate for IF matching.
The LT5557 was characterized with IF frequencies of
70MHz, 140MHz, 240MHz, 360MHz and 450MHz. The
values of L1 and C3 used for these frequencies are
tabulated in Figure 1 and repeated in Figure 9. In all cases,
L1 is a high-Q 0603 wire-wound chip inductor, for highest
conversion gain. Low-cost multi-layer chip inductors can
be substituted, with a slight reduction in conversion gain.
The measured IF output return losses are plotted in
Figure 9.
D
E
250
350
450
IF FREQUENCY (MHz)
550
5557 G09
A:
B:
C:
D:
E:
where CIF is the sum of C3 and the internal IF capacitance
(listed in Table 3). The value of C3 is selected such that L1
falls on a standard value, while satisfying the desired IF
bandwidth. The IF bandwidth can be estimated as:
BWIF = 1/(2πREFFCIF)
150
70MHz, L1 = 270nH, C3 = 15pF
140MHz, L1 = 180nH, C3 = 3.9pF
240MHz, L1 = 82nH, C3 = 2.2pF
360MHz, L1 = 47nH, C3 = 1.2pF
450MHz, L1 = 39nH, C3 = 0pF
Figure 9. IF Output Return Loss with
Transformer-Based Bandpass Matching
Discrete IF Balun Matching
For many applications, it is possible to replace the IF
transformer with the discrete IF balun shown in Figure 2.
The values of L1, L2, C6 and C7 are calculated to realize a
180 degree phase shift at the desired IF frequency and
provide a 50Ω single-ended output, using the equations
listed below. Inductor L3 is calculated to cancel the
internal 2.6pF capacitance. L3 also supplies bias voltage to
the IF+ pin. Low cost multilayer chip inductors are adequate for L1, L2 and L3. C3 is a DC blocking capacitor.
L1, L2 =
C6,C7 =
L3 =
RIF • ROUT
ωIF
1
ωIF • RIF • ROUT
XIF
ωIF
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Compared to the transformer-based IF matching technique, this network delivers approximately 1dB higher
conversion gain (since the IF transformer loss is eliminated), though noise figure and IIP3 are degraded slightly.
The most significant performance difference, as shown in
Figure 12, is the limited IF bandwidth available from the
discrete approach. For low IF frequencies, the absolute
bandwidth is small, whereas higher IF frequencies offer
wider bandwidth.
Table 5. Discrete IF Balun Element Values (ROUT = 50Ω)
IF FREQUENCY
(MHz)
L1, L2
C6, C7
L3
190
120nH
6.0pF
270nH || 3.3kΩ
240
100nH
4.7pF
150nH
360
56nH
3.0pF
82nH
450
47nH
2.2pF
47nH
IF PORT RETURN LOSS (dB)
–10
360 MHz
–20
240 MHz
190 MHz
–30
50
150
450 MHz
250
350
450
IF FREQUENCY (MHz)
550
5557 F10
Figure 10. IF Output Return Losses with Discrete Balun Matching
–10
26
24
IIP3
22
190IF
240IF
360IF
450IF
GC (dB), IIP3 (dBm)
20
18
16
–20
LOW-SIDE LO (–3dBm)
–30
TA = 25°C
–40
14
12
–50
10
LO-IF
8
6
–60
GC
4
2
1700
LO-IF LEAKAGE (dBm)
Discrete IF balun element values for four common IF
frequencies (190MHz, 240MHz, 360MHz and 450MHz)
are listed in Table 4. The 190MHz application circuit uses
a 3.3kΩ resistor in parallel with L3 as described above.
The corresponding measured IF output return losses are
shown in Figure 10. Typical conversion gain, IIP3 and LOIF leakage, versus RF input frequency, for all four examples is shown in Figure 11. Typical conversion gain,
IIP3 and noise figure versus IF output frequency is shown
in Figure 12.
0
2100
2000
1900
1800
RF INPUT FREQUENCY (MHz)
–70
2200
5557 F11
Figure 11. Conversion Gain, IIP3 and LO-IF Leakage
vs RF Input Frequency and IF Output Frequency
(in MHz) Using Discrete IF Balun Matching
26
24
22
GC (dB), NF (dB), IIP3 (dBm)
These equations give a good starting point, but it is usually
necessary to adjust the component values after building
and testing the circuit. The final solution can be achieved
with less iteration by considering the parasitics of L3 in the
above calculations. Specifically, the effective parallel resistance of L3 (calculated from the manufacturer’s Q data)
will reduce the value of RIF, which in turn influences the
calculated values of L1 (=L2) and C6 (=C7). Also, the
effective parallel capacitance of L3 (taken from the manufacturers SRF data) must be considered, since it is in
parallel with XIF (from table 3). Frequently, the calculated
value for L1 does not fall on a standard value for the
desired IF. In this case, a simple solution is to load the IF
output with a high-value external chip resistor in parallel
with L3, which reduces the value of RIF, until L1 is a
standard value.
IIP3
20
190IF
240IF
360IF
450IF
18
16
14
12
10
SSB NF
8
6
4
RF = 1950MHz
LOW-SIDE LO (–3dBm)
TA = 25°C
GC
2
150
200 250 300 350 400 450 500
IF OUTPUT FREQUENCY (MHz) 5557 F12
Figure 12. Conversion Gain, IIP3 and SSB NF vs IF Output
Frequency Using Discrete IF Balun Matching
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Differential IF Output Matching
For fully differential IF architectures, the mixer’s IF outputs can be matched directly into a SAW filter or IF
amplifier, thus eliminating the IF transformer. One example is shown in Figure 13, where the mixer’s 500Ω
differential output resistance is matched into a 100Ω
differential SAW filter using the tapped-capacitor technique. Inductors L1 and L2 form the inductive portion of
the matching network, cancel the internal 2.6pF capacitance, and supply DC bias current to the mixer core.
Capacitors C6 through C9 are the capacitive portion of the
matching, and perform the impedance step-down.
The calculations for tapped-capacitor matching are covered in the literature, and are not repeated here. Other
differential matching options include low-pass, highpass and band-pass. The choice depends on the system
performance goals, IF frequency, IF bandwidth and filter
(or amplifier) input impedance. Contact the factory for
applications assistance.
Enable Interface
Figure 14 shows a simplified schematic of the EN pin
interface. The voltage necessary to turn on the LT5557 is
2.7V. To disable the chip, the enable voltage must be less
than 0.3V. If the EN pin is allowed to float, the chip will tend
to remain in its last operating state. Thus it is not recommended that the enable function be used in this manner.
If the shutdown function is not required, then the EN pin
should be connected directly to VCC.
The voltage at the EN pin should never exceed the power
supply voltage (VCC) by more than 0.3V. If this should
occur, the supply current could be sourced through the
EN pin ESD diode, potentially damaging the IC.
C6
C8
L1
IF +
SAW
FILTER
VCC2
IF
AMP
LT5557
EN
5
22k
IF –
SUPPLY
DECOUPLING
L2
C1
5557 F13
C7
VCC
C2
C9
Figure 13. Differential IF Matching Using
the Tapped-Capacitor Technique
Figure 14. Enable Input Circuit
Standard Evaluation Board Layout (DC1131A)
Discrete IF Evaluation Board Layout (DC910A)
5557 F14
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PACKAGE DESCRIPTIO
UF Package
16-Lead Plastic QFN (4mm × 4mm)
(Reference LTC DWG # 05-08-1692)
0.72 ±0.05
4.35 ± 0.05
2.15 ± 0.05
2.90 ± 0.05 (4 SIDES)
PACKAGE OUTLINE
0.30 ±0.05
0.65 BSC
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
BOTTOM VIEW—EXPOSED PAD
4.00 ± 0.10
(4 SIDES)
0.75 ± 0.05
R = 0.115
TYP
15
PIN 1 NOTCH R = 0.20 TYP
OR 0.35 × 45° CHAMFER
16
0.55 ± 0.20
PIN 1
TOP MARK
(NOTE 6)
1
2.15 ± 0.10
(4-SIDES)
2
(UF16) QFN 10-04
0.200 REF
0.00 – 0.05
0.30 ± 0.05
0.65 BSC
NOTE:
1. DRAWING CONFORMS TO JEDEC PACKAGE OUTLINE MO-220 VARIATION (WGGC)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LT5557
RELATED PARTS
PART NUMBER DESCRIPTION
COMMENTS
Infrastructure
LT5511
High Linearity Upconverting Mixer
RF Output to 3GHz, 17dBm IIP3, Integrated LO Buffer
LT5512
1KHz-3GHz High Signal Level Active Mixer
20dBm IIP3 from 30MHz to 900MHz, Integrated LO Buffer, HF/VHF/UHF Optimized
LT5514
Ultralow Distortion, IF Amplifier/ADC Driver
with Digitally Controlled Gain
850MHz Bandwidth, 47dBm OIP3 at 100MHz, 10.5dB to 33dB Gain Control Range
LT5515
1.5GHz to 2.5GHz Direct Conversion Quadrature
Demodulator
20dBm IIP3, Integrated LO Quadrature Generator
LT5516
0.8GHz to 1.5GHz Direct Conversion Quadrature
Demodulator
21.5dBm IIP3, Integrated LO Quadrature Generator
LT5517
40MHz to 900MHz Quadrature Demodulator
21dBm IIP3, Integrated LO Quadrature Generator
LT5519
0.7GHz to 1.4GHz High Linearity Upconverting Mixer 17.1dBm IIP3 at 1GHz, Integrated RF Output Transformer with 50Ω Matching,
Single-Ended LO and RF Ports Operation
LT5520
1.3GHz to 2.3GHz High Linearity Upconverting Mixer 15.9dBm IIP3 at 1.9GHz, Integrated RF Output Transformer with 50Ω Matching,
Single-Ended LO and RF Ports Operation
LT5521
10MHz to 3700MHz High Linearity
Upconverting Mixer
24.2dBm IIP3 at 1.95GHz, NF = 12.5dB, 3.15V to 5.25V Supply, Single-Ended
LO Port Operation
LT5522
400MHz to 2.7GHz High Signal Level
Downconverting Mixer
4.5V to 5.25V Supply, 25dBm IIP3 at 900MHz, NF = 12.5dB, 50Ω Single-Ended RF
and LO Ports
LT5525
High Linearity, Low Power Downconverting Mixer
Single-Ended 50Ω RF and LO Ports, 17.6dBm IIP3 at 1900MHz, ICC = 28mA
LT5526
High Linearity, Low Power Downconverting Mixer
3V to 5.3V Supply, 16.5dBm IIP3, 100kHz to 2GHz RF, NF = 11dB, ICC = 28mA,
–65dBm LO-RF Leakage
LT5527
400MHz to 3.7GHz, 5V High Signal Level
Downconverting Mixer
23.5dBm IIP3 at 1.9GHz, NF = 12.5dB, Single-Ended RF and LO Ports
LT5528
1.5GHz to 2.4GHz High Linearity Direct I/Q
Modulator
21.8dBm OIP3 at 2GHz, –159dBm/Hz Noise Floor, 50Ω Interface at all Ports
LT5568
600MHz to 1.2GHz High Linearity Direct I/Q
Modulator
22.9dBm OIP3, –160.3dBm/Hz Noise Floor, –46dBc Image Rejection,
–43dBm Carrier Leakage
RF Power Detectors
LTC®5505
RF Peak Detectors with >40dB Dynamic Range
300MHz to 3GHz, Temperature Compensated, –32dBm to 12dBm
LTC5507
100kHz to 1000MHz RF Peak Power Detector
100kHz to 1GHz, Temperature Compensated, –34dBm to 14dBm
LTC5508
300MHz to 7GHz RF Peak Power Detector
44dB Dynamic Range, Temperature Compensated, SC70 Package,
–32dBm to 12dBm
LTC5509
300MHz to 3GHz RF Peak Power Detector
36dB Dynamic Range, Low Power Consumption, SC70 Package, –30dBm to 6dBm
LTC5530
300MHz to 7GHz Precision RF Peak Power Detector Precision VOUT Offset Control, Shutdown, Adjustable Gain, –32dBm to 10dBm
300MHz to 7GHz Precision RF Peak Power Detector Precision VOUT Offset Control, Shutdown, Adjustable Offset, –32dBm to 10dBm
LTC5531
LTC5532
300MHz to 7GHz Precision RF Peak Power Detector Precision VOUT Offset Control, Adjustable Gain and Offset,
±35mV Offset Voltage Tolerence
LTC5533
300MHz to 11GHz Dual Precision RF Peak Detector
–32dBm to 12dBm, Adjustable Offset, 45dB Ch-Ch Isolation
LT5534
50MHz to 3GHz RF Log Detector with 60dB
Dynamic Range
±1dB Output Variation over Temperature, 38ns Response Time
LTC5536
Precision 600MHz to 7GHz RF Peak Detector
with Fast Comparator Output
25ns Response Time, Comparator Reference Input, Latch Enable Input,
–26dBm to +12dBm Input Range
LT5537
90dB Dynamic Range RF Log Detector
LF to 1GHz, –79dBm to 12dBm, Very Low Tempco
Low Voltage RF Building Block
LT5546
500MHz Quadrature Demodulator with VGA and
17MHz Baseband Bandwidth
17MHz Baseband Bandwidth, 40MHz to 500MHz IF, 1.8V to 5.25V Supply, –7dB to
56dB Linear Power Gain
Linear Technology Corporation
LT/CGRAFX 0407 REV A • PRINTED IN THE USA
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16
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2006