LINER LT3570 1.5a buck converter, 1.5a boost converter and ldo controller Datasheet

LT3570
1.5A Buck Converter,
1.5A Boost Converter and
LDO Controller
DESCRIPTION
FEATURES
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The LT®3570 is a buck and boost converter with internal
power switches and LDO controller. Each converter is
designed with a 1.5A current limit and an input range from
2.5V to 36V, making the LT3570 ideal for a wide variety
of applications. Switching frequencies up to 2.1MHz are
programmed with an external timing resistor and the
oscillator can be synchronized to an external clock up to
2.75MHz.
2.5V to 36V Input Voltage Range
Programmable Switching Frequency
from 500kHz to 2.1MHz
Synchronizable Up to 2.75MHz
VOUT(MIN): 0.8V
Independent Soft-Start for Each Converter
Separate VIN Supplies for Each Converter
Duty Cycle Range: 0% to 90% at 1MHz
Available in 24-Lead (4mm × 4mm) QFN and
20-Lead TSSOP Packages
The LT3570 features a programmable soft-start function
that limits the feedback voltage during start-up helping
prevent overshoot and limiting inrush current. The LDO
controller is capable of delivering up to 10mA of base
current to an external NPN transistor.
APPLICATIONS
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Cable and Satellite Set-Top Boxes
Automotive Systems
Telecom Systems
“Dying Gasp” Systems
TFT LCD Displays
L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
TYPICAL APPLICATION
VIN
5V
VIN1 VIN2 VIN3 BIAS
SHDN1
SHDN2
SHDN3
D1
VOUT1
12V
275mA
SHDN1
SHDN2
SHDN3
D3
100nF
SW2
D2
6.8μH
SW1
143k
10μF
FB1
SS1
VC1
10.0k
Efﬁciency
BOOST
LT3570
100
VOUT2
3.3V
1A
3.3μH
95
32.4k
FB2
SS2
VC2
EFFICIENCY (%)
10μF
22μF
22k
10nF
10.2k
90
85
fSW = 1.2MHz
VIN = 5V
VOUT1 = 12V
VOUT2 = 3.3V
VOUT3 = 2.5V
IOUT1 = 275mA
IOUT3 = 100mA
80
1nF
75
22k
NPN_DRV
Q1
1nF
10nF
22.1k
RT
SYNC
FB3
GND
VOUT3
2.5V
100mA
2.2μF
70
0
0.2
0.4
0.6
IOUT2 (A)
0.8
1.0
3570 TA01b
10.2k
15.8k
3570 TA01a
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LT3570
ABSOLUTE MAXIMUM RATINGS
(Note 1)
VIN1, VIN2, VIN3, VBIAS Voltage ..................................40V
BOOST Voltage .........................................................60V
BOOST Pin Above SW2 .............................................25V
NPN_DRV Voltage .......................................................8V
SW1 Voltage .............................................................40V
SHDN1, SHDN2, SHDN3 Voltage ..............................40V
SYNC, RT Voltage ........................................................3V
SS1, SS2 Voltage ........................................................3V
FB1, FB2, FB3 Voltage ...............................................10V
VC1, VC2 Voltage..........................................................3V
Maximum Junction Temperature........................... 125°C
Operating Temperature Range (Note 2).. –40°C to 125°C
Storage Temperature Range
TSSOP ............................................... –65°C to 150°C
QFN.................................................... –65°C to 125°C
Lead Temperature (Soldering, 10 sec)
TSSOP Only ...................................................... 300°C
PIN CONFIGURATION
FB2
TOP VIEW
FB3
NPN_DRV
VIN3
BIAS
BOOST
TOP VIEW
24 23 22 21 20 19
VIN2 1
18 VC2
VIN2 2
17 SS2
SW2 3
16 GND
25
SW1 4
15 RT
FB1
1
20 VC1
SHDN1
2
19 SS1
SHDN2
3
18 VIN1
SHDN3
4
17 GND
SYNC
5
RT
6
21
16 SW1
15 SW2
GND 5
14 SYNC
SS2
7
14 VIN2
GND 6
13 SHDN3
VC2
8
13 BOOST
FB2
9
12 VIN3
FB3 10
SHDN2
SHDN1
FB1
9 10 11 12
VC1
SS1
8
VIN1
7
11 NPN_DRV
FE PACKAGE
20-LEAD PLASTIC TSSOP
UF PACKAGE
24-LEAD (4mm s 4mm) PLASTIC QFN
TJMAX = 125°C, θJA = 37°C/W
EXPOSED PAD (PIN 25) IS GND, MUST BE SOLDERED TO PCB
TJMAX = 125°C, θJA = 38°C/W
EXPOSED PAD (PIN 21) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3570EUF#PBF
LT3570EUF#TRPBF
3570
24-Lead (4mm × 4mm) Plastic QFN
–40°C to 125°C
LT3570IUF#PBF
LT3570IUF#TRPBF
3570
24-Lead (4mm × 4mm) Plastic QFN
–40°C to 125°C
LT3570EFE#PBF
LT3570EFE#TRPBF
LT3570FE
20-Lead Plastic TSSOP
–40°C to 125°C
LT3570IFE#PBF
LT3570IFE#TRPBF
LT3570FE
20-Lead Plastic TSSOP
–40°C to 125°C
Consult LTC Marketing for parts speciﬁed with wider operating temperature ranges. *The temperature grade is identiﬁed by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based ﬁnish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel speciﬁcations, go to: http://www.linear.com/tapeandreel/
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LT3570
ELECTRICAL CHARACTERISTICS
The l denotes the speciﬁcations which apply over the full operating
temperature range, otherwise speciﬁcations are at TA = 25°C. VIN1,2,3 = 12V, VSHDN1,2,3 = 12V unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
2.1
2.5
V
2.1
2.5
V
0
1.5
μA
Minimum Operating Voltage (VIN1)
(Note 3)
l
Minimum Operating Voltage (VIN2)
(Note 3)
l
Shutdown Current (Note 4)
VSHDN1,2,3 = 0V
VIN1 Quiescent Current
VSHDN1 = 12V, VSHDN2,3 = 0V, VC1 = 0.4V (Not Switching)
VSHDN1 = 0V, VSHDN2,3 = 12V
3.2
65
4.5
150
mA
μA
VIN2 Quiescent Current
VSHDN1,3 = 0V, VSHDN2 = 12V, VC2 = 0.4V (Not Switching)
VSHDN1,3 = 12V, VSHDN2 = 0V
3.5
3.5
4.5
4.5
mA
mA
VIN3 Quiescent Current
VSHDN1,2 = 0V, VSHDN3 = 12V
VSHDN1,2 = 12V, VSHDN3 = 0V
700
0
950
1.5
μA
μA
Bias Quiescent Current
VBIAS = 2.5V
2.3
3.1
mA
VSHDN1,2,3 Pin Threshold
IVIN2 > 100μA
1.4
V
1.25
1.4
V
30
0.1
50
1.5
μA
μA
500
2100
550
2300
kHz
kHz
0.3
l
VSHDN1,2,3 Pin UVLO
1.1
VSHDNX Pin Current
VSHDNX = 12V, VSHDNY,Z = 0V (Note 5)
VSHDN1,2,3 = 0V
Switching Frequency
RT = 44.2k
RT = 7.87k
450
1900
Maximum Duty Cycle
RT = 44.2k
RT = 7.87k
95
80
%
%
Synchronous Frequency Threshold
0.3
1.5
Synchronous Frequency Range
650
2750
Synchronous Frequency Minimum On/Off
Time
50
l
FB1,2,3 Pin Voltage
772
V
kHz
ns
788
804
0.01
mV
FB1,2,3 Pin Voltage Line Regulation
VVIN1,2,3 = 2.5V to 36V, VC1,2 = 1V
FB1,2 Pin Bias Current
VFB1,2 = 800mV, VC1,2 = 1V (Note 6)
30
200
%/V
nA
FB3 Pin Bias Current
VFB3 = 800mV (Note 6)
30
200
nA
SS1,2 Pin Source Current
VSS1,2 = 500mV
4.5
μA
VC1,2 Pin Source Current
VFB1,2 = 600mV
12
μA
VC1,2 Pin Sink Current
VFB1,2 = 1V
12
μA
Error Ampliﬁer 1 Transconductance
190
μMho
Error Ampliﬁer 1 Voltage Gain
100
V/V
VC1 Pin Switching Threshold
750
mV
SW1
VC1 to SW1 Current Gain
5.9
SW1 Current Limit
(Note 7)
1.5
2.4
SW1 VCESAT
ISW1 = 1A (Note 7)
240
SW1 Leakage Current
SW1 = 40V, VSHDN1 = 0V
0.2
A/V
3.1
A
mV
5
μA
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LT3570
ELECTRICAL CHARACTERISTICS
The l denotes the speciﬁcations which apply over the full operating
temperature range, otherwise speciﬁcations are at TA = 25°C. VIN1,2,3 = 12V, VSHDN1,2,3 = 12V unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
SW2
Error Ampliﬁer 2 Transconductance
195
μMho
Error Ampliﬁer 2 Voltage Gain
100
V/V
VC2 Pin Switching Threshold
700
mV
VC2 to SW2 Current Gain
5.4
SW2 Current Limit
(Note 7)
1.5
2.4
SW2 VCESAT
ISW2 = 1A (Note 7)
240
SW2 Leakage Current
SW2 = 0V, VIN2 = 40V, VSHDN2 = 0V
0.2
BOOST Pin Current
ISW2 = 0.5A
ISW2 = 1.5A
15
30
A/V
3.1
A
mV
5
μA
mA
mA
LDO
LDO Maximum Output Current
VFB3 = 600mV
10
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3570E is guaranteed to meet performance speciﬁcations
from 0°C to 125°C junction temperature. Speciﬁcations over the –40°C
to 125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LT3570I is guaranteed over the full –40°C to 125°C operating junction
temperature range.
Note 3: VIN2 supplies power for the part. VIN1 supplies power only to the
boost converter. VIN3 supplies power only to the LDO Controller.
20
mA
Note 4: Shutdown current is for each individual input current.
Note 5: Current ﬂows into the pin.
Note 6: Current ﬂows out of the pin.
Note 7: Switch current limit and switch VCESAT guaranteed by design
and/or correlation to static test.
Note 8: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed the maximum operating junction temperature
range when overtemperature protection is active. Continuous operation
above the speciﬁed maximum operating junction temperature may impair
device reliability.
TYPICAL PERFORMANCE CHARACTERISTICS
Frequency vs Temperature
Feedback Voltage vs Temperature
RT = 7.87k
0.795
4.8
2000
0.790
0.785
0.780
4.6
CURRENT (μA)
FREQUENCY (kHz)
VOLTAGE (V)
Soft-Start Current vs Temperature
5.0
2500
0.800
1500
RT = 20k
1000
0.770
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3570 G01
0
–50 –25
4.2
4.0
RT = 44.2k
500
0.775
4.4
3.8
0
25 50
75 100 125 150
TEMPERATURE (°C)
3570 G02
3.6
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3570 G03
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LT3570
TYPICAL PERFORMANCE CHARACTERISTICS
VIN1 Quiescent Current
vs Temperature
VIN2 Quiescent Current
vs Temperature
3.5
3.0
4.0
900
3.5
800
700
CURRENT (mA)
2.0
1.5
1.0
0.5
0
–50 –25
0
CURRENT (μA)
3.0
2.5
CURRENT (mA)
VIN3 Quiescent Current
vs Temperature
2.5
2.0
1.5
200
0.5
100
0
1.25
25 50 75 100 125 150
TEMPERATURE (°C)
75
100 125 150
1.00
0.75
0.50
0
–50 –25
SHDN Pin Current vs Voltage
40
30
20
10
0.25
0
50
50
0
0
25 50 75 100 125 150
TEMPERATURE (°C)
3570 G07
0
5
10
15 20
25
VOLTAGE (V)
30
35
40
3570 G09
3570 G08
SW1 Saturation Voltage
vs SW1 Current
SW1 Current Limit vs Duty Cycle
3.0
350
2.5
300
250
2.0
VOLTAGE (mV)
0
–50 –25
25
3570 G06
CURRENT (μA)
SHDN PIN VOLTAGE (V)
2.5
CURRENT (A)
CURRENT (mA)
1.50
0.5
0
TEMPERATURE (°C)
SHDN Pin UVLO vs Temperature
Bias Pin Current vs Temperature
1.0
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
3570 G05
3.0
1.5
400
300
3570 G04
2.0
500
1.0
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
600
1.5
1.0
200
150
100
TJ = 125°C
TJ = 25°C
TJ = –40°C
0.5
0
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
3570 G10
50
0
TJ = 125°C
TJ = 25°C
TJ = –40°C
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
CURRENT (A)
3570 G11
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LT3570
TYPICAL PERFORMANCE CHARACTERISTICS
SW2 Saturation Voltage
vs SW2 Current
3.0
350
2.5
300
250
2.0
VOLTAGE (mV)
CURRENT (A)
SW2 Current Limit vs Duty Cycle
1.5
1.0
200
150
100
TJ = 125°C
TJ = 25°C
TJ = –40°C
0.5
0
0
TJ = 125°C
TJ = 25°C
TJ = –40°C
50
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
CURRENT (A)
3570 G12
3570 G13
BOOST Pin Current
vs Switch Current
NPN_DRV Output Current vs VIN3
30
18
16
25
CURRENT (mA)
CURRENT (mA)
20
TJ = –40°C
15
TJ = 25°C
14
TJ = 25°C
TJ = 125°C
10
TJ = 125°C
12
TJ = –40°C
10
8
6
4
5
2
0
0
0.2
0.4
0.6
CURRENT (A)
0.8
1.0
3570 G14
0
1
5
10
15 20 25
VOLTAGE (V)
30
35
40
3570 G15
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LT3570
PIN FUNCTIONS
(QFN/TSSOP)
VIN2 (Pins 1,2/Pin 14): Input Voltage for the Buck Regulator.
This pin also supplies the current to the internal circuitry
of the LT3570. This pin must be locally bypassed with a
capacitor.
SHDN3 (Pin13/Pin 4): Shutdown Pin. Tie to 1.5V or more
to enable the NPN LDO. Tie all SHDN pins to 0.3V or less
to shutdown the part.
SW2 (Pin 3/Pin15): Switch Node. This pin connects to
the emitter of an internal NPN power switch. Connect a
diode, inductor and boost capacitor to this pin to form
the buck regulator.
SYNC (Pin 14/Pin 5): Synchronization Pin. The SYNC pin
is used to synchronize the internal oscillator to an external signal. The synchronizing range is equal to the initial
operating frequency set by the RT pin up to 1.3 times the
initial operating frequency.
SW1 (Pin 4/Pin16): Switch Node. This pin connects to the
collector of an internal NPN power switch. Connect a diode
and inductor to this pin to form the boost regulator.
RT (Pin 15/Pin 6): Frequency Set Pin. Place a resistor to
GND to set the internal frequency. The range of oscillation
is 500kHz to 2MHz.
GND (Pins 5, 6, 16, 25/Pins 17, 21): Ground. The Exposed
Pad of the package provides both electrical contact to
ground and good thermal contact to the printed circuit
board. The Exposed Pad must be soldered to the circuit
board for proper operation.
SS2 (Pin 17/Pin 7): Soft-Start Pin. Place a soft-start
capacitor here. Upon start-up, a current charges the
capacitor to 2V. This pin ramps the reference voltage of
the buck switcher.
VIN1 (Pin 7/Pin18): Input Voltage for the Boost Regulator.
This pin supplies current to drive the boost NPN transistor
of the LT3570. This pin must be locally bypassed with a
capacitor.
SS1 (Pin 8/Pin 19): Soft-Start Pin. Place a soft-start
capacitor here. Upon start-up, a current charges the
capacitor to 2V. This pin ramps the reference voltage of
the boost switcher.
VC1 (Pin 9/Pin 20): Control Voltage and Compensation Pin
for the Internal Error Ampliﬁer. Connect a series RC from
this pin to ground to compensate the switching regulator
loop for the boost regulator.
FB1 (Pin 10/Pin 1): Feedback Pin. The LT3570 regulates
this pin to 788mV. Connect the feedback resistors to
this pin to set the output voltage for the boost switching
regulator.
SHDN1 (Pin 11/Pin 2): Shutdown Pin. Tie to 1.4V or more
to enable the boost switcher. Tie all SHDN pins to 0.3V or
less to shutdown the part.
SHDN2 (Pin 12/Pin 3): Shutdown Pin. Tie to 1.4V or more
to enable the buck switcher. Tie all SHDN pins to 0.3V or
less to shutdown the part.
VC2 (Pin 18/Pin 8): Control Voltage and Compensation Pin
for the Internal Error Ampliﬁer. Connect a series RC from
this pin to ground to compensate the switching regulator
loop for the buck regulator.
FB2 (Pin 19/Pin 9): Feedback Pin. The LT3570 regulates
this pin to 788mV. Connect the feedback resistors to
this pin to set the output voltage for the buck switching
regulator.
FB3 (Pin 20/Pin 10): Feedback Pin. The LT3570 regulates
this pin to 788mV. Connect the feedback resistors to this
pin to set the output voltage for the LDO controller.
NPN_DRIVE (Pin 21/Pin 11): Base Drive for the External
NPN. This pin provides a bias current to drive the base of
the NPN. This base current is driven from the IN3 supply
voltage.
VIN3 (Pin 22/Pin 12): Input Voltage for the NPN LDO. This
pin supplies current to drive the base of the NPN. This pin
must be locally bypassed with a capacitor.
BIAS (Pin 23): QFN Package Only. This pin supplies current
to the internal circuitry of the LT3570 if greater than 2.5V.
This pin must be locally bypassed with a capacitor.
BOOST (Pin 24/Pin 13): Bias for the Base Drive of the
NPN Switch for the Buck Regulator. This pin provides a
bias voltage higher than VIN2. The voltage on this pin is
charged up through an external Schottky diode.
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8
C2
R2B
R1B
D3
L2
C6B
D2
C5
R4
FB2
SS2
SW2
Q2
BOOST
RT
SYNC
R5
VIN2
R
788mV
4.5μA
A8
Q
S
+
+
–
–
+
A5
A7
A6
–
+
Figure 1. Block Diagram
C4B
C4A
R3A
R3B
–
+
A3
SHDN2
VC1
A9
OSCILLATOR
A10
REGULATOR
SHDN1
VC2
BIAS
Q
A4
R
S
788mV
SHDN3
–
+
A1
A2
A11
+
+
–
–
+
GND
788mV
FB1
4.5μA
SS1
R6
Q1
SW1
VIN1
FB3
NPN_DRV
VIN3
3570 BD
C6A
R2A
R1A
D1
L1
C1
R2C
R1C
Q3
C3
LT3570
BLOCK DIAGRAM
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LT3570
OPERATION
The LT3570 is a constant frequency, current mode, buck
converter and boost converter with an NPN LDO regulator. Operation can be best understood by referring to the
Block Diagram.
If all of the SHDN pins are held low, the LT3570 is shut
down and draws zero quiescent current. When any of the
pins exceed 1.4V the internal bias circuits turn on. Each
regulator will only begin regulating when its corresponding
SHDN pin is pulled high.
Each switching regulator controls the output voltage in
a similar manner. The operation of the switchers can be
understood by looking at the boost regulator. A pulse
from the oscillator sets the RS ﬂip-ﬂop A4 and turns on
the internal NPN bipolar power switch Q1. Current in Q1
and the external inductor L1 begins to increase. When
this current exceeds a level determined by the voltage at
VC1, comparator A3 resets A4, turning off Q1. The current
in L1 ﬂows through the external Schottky diode D1 and
begins to decrease. The cycle begins again at the next
pulse from the oscillator. In this way, the voltage on the
VC1 pin controls the current through the inductor to the
output. The internal error ampliﬁer A1 regulates the output
voltage by continually adjusting the VC1 pin voltage. The
threshold for switching on the VC1 pin is approximately
750mV and an active clamp of 1.15V limits the output
current. The soft-start capacitor C6A allows the part to
slowly start up by ramping the internal reference.
The driver for the buck regulator can operate from either
VIN2 or from the BOOST pin. An external capacitor and
diode are used to generate a voltage at the BOOST pin that
is higher than the input supply. This allows the driver to
saturate the internal bipolar NPN power switch for efﬁcient
operation. The driver for the boost regulator is operated
from VIN1.
The BIAS pin allows the internal circuitry to draw its
current from a lower voltage supply than the input. This
reduces power dissipation and increases efﬁciency. If the
voltage on the BIAS pin falls below 2.5V, then the LT3570
quiescent current will ﬂow from VIN2.
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LT3570
APPLICATIONS INFORMATION
FB Resistor Network
The output voltage is programmed with a resistor divider
(refer to the Block Diagram) between the output and the
FB pin. Choose the resistors according to:
⎞
⎛ V
R1= R2 ⎜ OUT – 1⎟
⎝ 788mV ⎠
Buck Inductor Selection and Maximum Output Current
A good ﬁrst choice for the inductor value is
L=
VOUT2 + VF
for SW2
0.75 • f
where VF is the voltage drop of the catch diode (~0.4V)
and f is the switching frequency. With this inductance
value or greater, the maximum load current will be 1A,
independent of input voltage. The inductor’s RMS current
rating must be greater than the maximum load current and
its saturation current should be at least 30% higher. For
highest efﬁciency, the series resistance (DCR) should be
less than 0.1Ω. Table 1 lists several vendors and types
that are suitable.
Table 1. Inductors
PART NUMBER
Sumida
CDRH4D28-3R3
CDRH4D28-4R7
CDC5D23-2R2
CR43-3R3
CDRH5D28-100
Coilcraft
DO1608C-332
DO1608C-472
MOS6020-332
D03314-103
D03314-222
Toko
(D62F)847FY-2R4M
(D73LF)817FY-2R2M
Coiltronics
TP3-4R7
TP1-2R2
TP4-100
VALUE
(μH)
ISAT
(A)
DCR
(Ω)
HEIGHT
(mm)
3.3
4.7
2.2
3.3
10
1.57
1.32
2.50
1.44
1.3
0.049
0.072
0.03
0.086
0.048
3.0
3.0
2.5
3.5
3.0
3.3
4.7
3.3
10
2.2
2.00
1.50
1.8
0.8
1.6
0.080
0.090
0.046
0.520
0.200
2.9
2.9
2.0
1.4
1.4
The optimum inductor for a given application may differ
from the one indicated by this simple design guide. A
larger value inductor provides a slightly higher maximum
load current and will reduce the output voltage ripple. If
your load is lower than the maximum load current, then
you can relax the value of the inductor and operate with
higher ripple current. This allows you to use a physically
smaller inductor or one with a lower DCR resulting in
higher efﬁciency. Be aware that if the inductance differs
from the simple rule above, then the maximum load current
will depend on input voltage. In addition, low inductance
may result in discontinuous mode operation, which further
reduces maximum load current. For details of maximum
output current and discontinuous mode operation, see
Linear Technology’s Application Note 44. Finally, for duty
cycles greater than 50% (VOUT2/VIN2 > 0.5) a minimum
inductance is required to avoid subharmonic oscillations,
see Application Note 19.
The current in the inductor is a triangle wave with an average
value equal to the load current. The peak switch current
is equal to the output current plus half the peak-to-peak
inductor ripple current. The LT3570 limits its switch current in order to protect itself and the system from overload
faults. Therefore, the maximum output current that the
LT3570 will deliver depends on the switch current limit,
the inductor value and the input and output voltages.
When the switch is off, the potential across the inductor
is the output voltage plus the catch diode drop. This gives
the peak-to-peak ripple current in the inductor:
ΔIL2 =
(1– DC2)( VOUT2 + VF )
L•f
where DC2 is the duty cycle and is deﬁned as:
DC2 =
VOUT2
VIN2
The peak inductor and switch current is:
2.4
2.2
2.5
2.7
0.037
0.03
2.7
3.0
4.7
2.2
10
1.5
1.3
1.5
0.181
0.188
0.146
2.2
1.8
3.0
ISWPK2 =ILPK2 =IOUT2 +
ΔIL2
2
To maintain output regulation, this peak current must be
less than the LT3570’s switch current limit ILIM2. ILIM2 is
at least 1.5A at low duty cycles and decreases linearly
3570fb
10
LT3570
APPLICATIONS INFORMATION
to 1.2A at DC2 = 0.8. The maximum output current is a
function of the chosen inductor value:
IOUT2(MAX) =ILIM2 –
ΔIL2
2
= 1.5 • (1– 0.25 • DC2) –
ΔIL2
2
Choosing an inductor value so that the ripple current is
small will allow a maximum output current near the switch
current limit.
One approach to choosing the inductor is to start with the
simple rule given above, look at the available inductors
and choose one to meet cost or space goals. Then use
these equations to check that the LT3570 will be able to
deliver the required output current. Note again that these
equations assume that the inductor current is continuous. Discontinuous operation occurs when IOUT2 is less
than ΔIL2/2.
Boost Inductor Selection
For most applications the inductor will fall in the range
of 2.2μH to 22μH. Lower values are chosen to reduce
physical size of the inductor. Higher values allow more
output current because they reduce peak current seen by
the power switch, which has a 1.5A current limit. Higher
values also reduce input ripple voltage and reduce core
loss. The following procedure is suggested as a way of
choosing a more optimum inductor.
Assume that the average inductor current for a boost
converter is equal to the load current times VOUT1/VIN1
and decide whether or not the inductor must withstand
continuous overload conditions. If average inductor current at maximum load current is 0.5A, for instance, a 0.5A
inductor may not survive a continuous 1.5A overload
condition. Also be aware that boost converters are not
short-circuit protected, and that under short conditions,
inductor current is limited only by the available current
of the input supply.
Calculate peak inductor current at full load current to ensure that the inductor will not saturate. Peak current can
be signiﬁcantly higher than output current, especially with
smaller inductors and lighter loads, so don’t omit this step.
Powdered iron cores are forgiving because they saturate
softly, whereas ferrite cores saturate abruptly. Other
core materials fall somewhere in between. The following
formula assumes continuous mode operation but it errs
only slightly on the high side for discontinuous mode, so
it can be used for all conditions.
IPEAK1 =
IOUT1 • VOUT1 VIN1 ( VOUT1 – VIN1 )
+
VIN1
2 • f • L • VOUT1
Make sure that IPEAK1 is less than the switch current ILIM1.
ILIM1 is at least 1.5A at low duty cycles and decreases
linearly to 1.2A at DC1 = 0.8. The maximum switch current
limit can be calculated by the following formula:
ILIM1 = 1.5 • (1 – 0.25 • DC1)
where DC1 is the duty cycle and is deﬁned as:
DC1= 1–
VIN1
VOUT1
Remember also that inductance can drop signiﬁcantly with
DC current and manufacturing tolerance. Consideration
should also be given to the DC resistance of the inductor
as this contributes directly to the efﬁciency losses in the
overall converter. Table 1 lists several inductor vendors
and types that are suitable.
Buck Output Capacitor Selection
For 5V and 3.3V outputs, a 10μF, 6.3V ceramic capacitor
(X5R or X7R) at the output results in very low output voltage ripple and good transient response. For lower voltages,
10μF is adequate for ripple requirements but increasing
COUT will improve transient performance. Other types and
values will also work; the following discusses tradeoffs in
output ripple and transient performance.
The output capacitor ﬁlters the inductor current to generate
an output with low voltage ripple. It also stores energy in
order to satisfy transient loads and stabilize the LT3570’s
control loop. Because the LT3570 operates at a high
frequency, minimal output capacitance is necessary. In
addition, the control loop operates well with or without
the presence of output capacitor series resistance (ESR).
Ceramic capacitors, which achieve very low output ripple
3570fb
11
LT3570
APPLICATIONS INFORMATION
and small circuit size, are therefore an option. You can
estimate output ripple with the following equations:
VRIPPLE =
ΔIL2
for ceramic capacitors
8 • f • COUT
and
VRIPPLE = ΔIL2 • ESR for electrolytic capacitors (tantalum
and aluminum)
The RMS content of this ripple is very low so the RMS
current rating of the output capacitor is usually not of
concern. It can be estimated with the formula:
IC(RMS) =
ΔIL2
Table 2. Low ESR Surface Mount Capacitors
VENDOR
TYPE
SERIES
Taiyo Yuden
Ceramic
X5R, X7R
AVX
Ceramic
Tantalum
X5R, X7R
TPS
Kemet
Tantalum
Ta Organic
Al Organic
T491, T494, T495
T520
A700
Sanyo
Ta or Al Organic
POSCAP
Panasonic
Al Organic
SP CAP
TDK
Ceramic
X5R, X7R
Boost Output Capacitor Selection
12
Another constraint on the output capacitor is that it must
have greater energy storage than the inductor; if the stored
energy in the inductor transfers to the output, the resulting
voltage step should be small compared to the regulation
voltage. For a 5% overshoot, this requirement indicates:
⎛ I
⎞
COUT > 10 • L • ⎜ LIM2 ⎟
⎝ VOUT2 ⎠
response for large changes in load current. Table 2 lists
several capacitor vendors.
2
The low ESR and small size of ceramic capacitors make
them the preferred type for LT3570 applications. Not all
ceramic capacitors are the same, however. Many of the
higher value capacitors use poor dielectrics with high
temperature and voltage coefﬁcients. In particular, Y5V
and Z5U types lose a large fraction of their capacitance
with applied voltage and at temperature extremes. Because
loop stability and transient response depend on the value
of COUT, this loss may be unacceptable. Use X7R and X5R
types.
Electrolytic capacitors are also an option. The ESRs of
most aluminum electrolytic capacitors are too large to
deliver low output ripple. Tantalum, as well as newer,
lower ESR organic electrolytic capacitors intended for
power supply use are suitable. Chose a capacitor with a
low enough ESR for the required output ripple. Because
the volume of the capacitor determines its ESR, both the
size and the value will be larger than a ceramic capacitor
that would give similar ripple performance. One beneﬁt
is that the larger capacitance may give better transient
Low ESR capacitors should be used at the output to
minimize the output ripple voltage. Multilayer ceramic
capacitors are the best choice, as they have a very low
ESR and are available in very small packages. Always use
a capacitor with a sufﬁcient voltage rating. Boost regulators have large RMS ripple current in the output capacitor,
which must be rated to handle the current. The formula
to calculate this is:
IRIPPLE(RMS) =IOUT
V
–V
DC1
= IOUT1 OUT1 IN1
VIN1
1– DC1
and is largest when VIN1 is at its minimum value if VOUT1
and IOUT1 are constant. With a 1.5A current limit, the
maximum that the output current ripple can be is ~0.75A.
Table 2 lists several capacitor vendors.
Buck Input Capacitor Selection
Bypass the input of the LT3570 circuit with a 10μF or
higher ceramic capacitor of X7R or X5R type. A lower
value or a less expensive Y5V type will work if there is
additional bypassing provided by bulk electrolytic capacitors, or if the input source impedance is low. The following
paragraphs describe the input capacitor considerations
in more detail.
Step-down regulators draw current from the input supply
in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple
at the LT3570 input and to force this switching current
3570fb
12
LT3570
APPLICATIONS INFORMATION
into a tight local loop, minimizing EMI. The input capacitor must have low impedance at the switching frequency
to do this effectively and it must have an adequate ripple
current rating. The RMS input current is:
IIN2(RMS) =IOUT2 •
VOUT2 ( VIN2 – VOUT2 )
VIN2
<
IOUT2
2
and is largest when VIN2 = 2 • VOUT2 (50% duty cycle).
Considering that the maximum load current is ~1.5A, RMS
ripple current will always be less than 0.75A.
The high frequency of the LT3570 reduces the energy
storage requirements of the input capacitor, so that the
capacitance required is often less than 10μF. The combination of small size and low impedance (low equivalent
series resistance or ESR) of ceramic capacitors makes
them the preferred choice. The low ESR results in very
low voltage ripple. Ceramic capacitors can handle larger
magnitudes of ripple current than other capacitor types
of the same value. Use X5R and X7R types.
An alternative to a high value ceramic capacitor is a lower
value along with a larger electrolytic capacitor, for example
a 1μF ceramic capacitor in parallel with a low ESR tantalum
capacitor. For the electrolytic capacitor, a value larger than
10μF will be required to meet the ESR and ripple current
requirements. Because the input capacitor is likely to see
high surge currents when the input source is applied,
tantalum capacitors should be surge rated. The manufacturer may also recommend operation below the rated
voltage of the capacitor. Be sure to place the 1μF ceramic
as close as possible to the VIN2 and GND pins on the IC
for optimal noise immunity.
A ﬁnal caution is in order regarding the use of ceramic
capacitors at the input. A ceramic input capacitor can
combine with stray inductance to form a resonant tank
circuit. If power is applied quickly (for example by plugging
the circuit into a live power source), this tank can ring,
doubling the input voltage and damaging the LT3570. The
solution is to either clamp the input voltage or dampen the
tank circuit by adding a lossy capacitor in parallel with the
ceramic capacitor. For details, see Application Note 88.
Boost Input Capacitor Selection
The capacitor of a boost converter is less critical due to
the fact that the input current waveform is triangular and
does not contain large squarewave currents as found in
the output capacitor. Capacitors in the range of 10μF to
100μF with an ESR of 0.3Ω or less work well up to the
full 1.5A switch current. Higher ESR capacitors may be
acceptable at low switch currents. Input capacitor ripple
current for boost converters is:
IRIPPLE = 0.3 • VIN1 •
VOUT1 – VIN1
f • L • VOUT1
Buck Diode Selection
The catch diode (D2 from Figure 1) conducts current only
during switch-off time. Average forward current in normal
operation can be calculated from:
ID(AVG) =IOUT1 •
VIN1 – VOUT1
VIN1
The only reason to consider a diode with a larger current
rating than necessary for nominal operation is for the
worst-case condition of shorted output. The diode current
will then increase to the typical peak switch current.
Peak reverse voltage is equal to the regulator input voltage.
Use a diode with a reverse voltage rating greater than the
input voltage. Table 3 lists several Schottky diodes and
their manufacturers.
Table 3. Schottky Diodes
PART NUMBER
VR (V)
IAVE (A)
VF AT 1A (mV)
MBRM120E
20
1
530
MBRM140
40
1
550
B120
20
1
500
B130
30
1
500
30
1
420
On Semiconductor
Diodes Inc.
International Rectiﬁer
10BQ030
3570fb
13
LT3570
APPLICATIONS INFORMATION
Boost Diode Selection
D3
A Schottky diode is recommended for use with the LT3570
inverter/boost regulator. The Microsemi UPS120 is a very
good choice. Where the input to output voltage differential exceeds 20V, use the UPS140 (a 40V diode). These
diodes are rated to handle an average forward current of
1A. For applications where the average forward current
of the diode is less than 0.5A, use an ON Semiconductor
MBR0520L diode.
BOOST
C3
LT3570
VIN
SW
D2
GND
C2
(2a)
D3
BOOST Pin Considerations
BOOST
The capacitor and diode tied to the BOOST pin generate
a voltage that is higher than the input voltage. In most
cases, a 0.1μF capacitor and fast switching diode (such
as the CMDSH-3 or MMSD914LT1) will work well. Figure 2 shows three ways to arrange the boost circuit. The
BOOST pin must be more than 2.5V above the SW pin for
full efﬁciency. For outputs of 3.3V and higher, the standard
circuit (Figure 2a) is best. For outputs between 2.8V and
3.3V, use a small Schottky diode (such as the BAT-54).
For lower output voltages, the boost diode can be tied
to the input (Figure 2b). The circuit in Figure 2a is more
efﬁcient because the BOOST pin current comes from a
lower voltage source. Finally, as shown in Figure 2c, the
anode of the boost diode can be tied to another source
that is at least 3V. For example, if you are generating 3.3V
and 1.8V and the 3.3V is on whenever the 1.8V is on, the
1.8V boost diode can be connected to the 3.3V output. In
any case, be sure that the maximum voltage at the BOOST
pin is less than 60V and the voltage difference between
the BOOST and SW2 pins is less than 25V.
The minimum operating voltage of an LT3570 application
is limited by the undervoltage lockout (2.5V) and by the
maximum duty cycle. The boost circuit also limits the
minimum input voltage for proper start-up. If the input
voltage ramps slowly, or the LT3570 turns on when the
output is already in regulation, the boost capacitor may
not be fully charged. Because the boost capacitor charges
with the energy stored in the inductor, the circuit will rely
on some minimum load current to get the boost circuit
running properly. This minimum load will depend on input
and output voltages, and on the arrangement of the boost
circuit. The minimum load current generally goes to zero
C5
LT3570
VIN
SW
D2
GND
C2
(2b)
D3
VEXT
BOOST
C5
LT3570
VIN
SW
D2
GND
(2c)
C2
3570 F02
Figure 2. Boost Pin Conﬁgurations
once the circuit has started. Even without an output load
current, in many cases the discharged output capacitor will
present a load to the switcher that will allow it to start.
Switcher Frequency Compensation
The LT3570 uses current mode control to regulate the
output. This simpliﬁes loop compensation. In particular, the
LT3570 does not depend on the ESR of the output capacitor for stability so you are free to use ceramic capacitors
to achieve low output ripple and small circuit size.
To compensate the feedback loop of the LT3570, a series
resistor-capacitor network should be connected from
the VC pin to GND. For most applications, a capacitor in
the range of 500pF to 4.7nF will sufﬁce. A good starting
value for the compensation capacitor, CC, is 1nF. The
3570fb
14
LT3570
APPLICATIONS INFORMATION
compensation resistor, RC, is usually in the range of 5k to
50k. A good technique to compensate a new application
is to use a 50k potentiometer in place of RC, and use a
1nF capacitor for CC. By adjusting the potentiometer while
observing the transient response, the optimum value for
RC can be found. Figures 3a to 3c illustrate this process
for the circuit of Figure 1 with load current stepped from
100mA to 500mA for the buck converter. Figure 3a shows
the transient response with RC equal to 1.6k. The phase
margin is poor as evidenced by the excessive ringing in
the output voltage and inductor current. In Figure 3b,
the value of RC is increased to 5.75k, which results in a
more damped response. Figure 3c shows the result when
RC is increased further to 25k. The transient response
is nicely damped and the compensation procedure is
complete. The same procedure is used to compensate
the boost converter.
Soft-Start
The soft-start time is programmed with an external capacitor to ground on SS. An internal current source charges it
with a nominal 4.5μA. The voltage on the soft-start pin is
used to control the feedback voltage. The soft-start time
is determined by the equation:
tSS = 0.2 • CSS
where CSS is in nF and tSS is in ms. In the event of a
commanded shutdown, ULVO on the input or a thermal
shutdown, the capacitor is discharged automatically. The
soft-start will remain low and only charge back up after
the fault goes away and the voltage on SS is less than
approximately 100mV.
IOUT
500mA/DIV
IOUT
500mA/DIV
VOUT
VOUT
3570 F03a
200μs/DIV
200μs/DIV
Figure 3a. Transient Response Shows Excessive Ringing
3570 F03b
Figure 3b. Transient Response is Better
IOUT
500mA/DIV
VOUT
200μs/DIV
3570 F03c
Figure 3c. Transient Response Well Damped
3570fb
15
LT3570
APPLICATIONS INFORMATION
25
The free-running frequency is set through a resistor from
the RT pin to ground. The oscillator frequency vs RT can
be seen in Figure 4. The oscillator can be synchronized
with an external clock applied to the SYNC pin. When
synchronizing the oscillator, the free running frequency
must be set approximately 10% lower than the desired
synchronized frequency.
2250
MINIMUM DUTY CYCLE (%)
Oscillator
20
15
10
5
0
500
2000
750
1000 1250 1500
FREQUENCY (kHz)
1750
FREQUENCY (kHz)
1750
2000
3570 F05
Figure 5. Minimum Duty Cycle vs Frequency
1500
1250
LDO Regulator
1000
750
500
250
5
10
15
20 25 30 35
RESISTANCE (kΩ)
40
45
3570 F04
Figure 4. Frequency vs RT Resistance
Buck Regulator Minimum On-Time
As the input voltage is increased, the LT3570 is required
to turn on for shorter periods of time. Delays associated
with turning off the power switch determine the minimum
on-time that can be achieved and limit the minimum duty
cycle. Figure 5 shows the minimum duty cycle versus
frequency for the LT3570. When the required on-time has
decreased below the minimum on-time of the LT3570 the
inductor current will increase, exceeding the current limit.
If the current through the inductor exceeds the current limit
of the LT3570, the switch is prevented from turning on for
10μs allowing the inductor current to decrease. The 10μs
off-time limits the average current that can be delivered to
the load. To return to normal switching frequency either
the input voltage or load current must decrease.
The LT3570 LDO regulator is capable of delivering up to
10mA of base drive for an external NPN transistor. For
stable operation the total output capacitance can be from
1μF up to 100μF. The regulator has its own independent
supply voltage which allows for the base of the NPN to be
driven from a higher voltage than its collector. This allows
for the NPN regulator to run more efﬁciently. The power
Dissipated in the external NPN is equal to:
PDISS = (VCOL – VOUT3) • ILOAD
where VCOL is the collector voltage of the NPN. The maximum output voltage is limited to:
VIN3 – 1.4V and VCOL – 0.2V or 8V
The short-circuit protection of the NPN regulator is set by
the max output current of the NPN_DRV pin multiplied by
the beta of the NPN.
Thermal Shutdown
An internal temperature monitor will turn off the internal
circuitry and prevent the switches from turning on when
the die temperature reaches approximately 160°C. When
the die temperature has dropped below this value the part
3570fb
16
LT3570
APPLICATIONS INFORMATION
will be enabled again going through a soft-start cycle.
Note: Overtemperature protection is intended to protect the
device during momentary overload conditions. Continuous
operation above the speciﬁed maximum operating junction
temperature may result in device degradation or failure.
to system ground at one location. Additionally, keep
the SW and BOOST nodes as small as possible. This is
implemented in the suggested layout of Figure 8 for the
QFN package which shows the topside metal from the
DC1106A demonstration board.
PCB Layout
Thermal Considerations
For proper operation and minimum EMI, care must be
taken during printed circuit board (PCB) layout. Figure 6
shows the high current paths in the step-down regulator
circuit. Note that in the step-down regulator, large switched
currents ﬂow in the power switch, the catch diode and the
input capacitor.
To deliver the power that the LT3570 is capable of, it
is imperative that a good thermal path be provided to
dissipate the heat generated within the package. This can
be accomplished by taking advantage of the large thermal pad on the underside of the IC. It is recommended
that multiple vias in the printed circuit board be used to
conduct heat away from the IC and into a copper plane
with as much area as possible.
Figure 7 shows the high current paths in the step-up
regulator. In the boost regulator, large switched currents
ﬂow through the power switch, the switching diode, and
the output capacitor.
The loop formed by these large switched current components should be as small as possible. Place these
components on the same side of the circuit board and
connect them on that layer. Place a local, unbroken ground
plane below these components and tie this ground plane
Related Linear Technology Publications
Application notes 19, 35, 44, 76 and 88 contain more
detailed descriptions and design information for buck
regulators and other switching regulators. The LT1375 data
sheet has a more extensive discussion of output ripple,
loop compensation, and stability testing.
LT3570
L2
HIGH
FREQUENCY
CIRCULATING
PATH
CIN
D1 COUT
LOAD
3570 F06
Figure 6. Buck High Speed Switching Path
L2
LT3570
CIN
D1
HIGH
FREQUENCY
SWITCHING
PATH
COUT LOAD
3750 F07
Figure 7. Boost High Speed Switching Path
Figure 8. Suggested Layout
3570fb
17
LT3570
TYPICAL APPLICATIONS
DSL Modem
VIN
8V TO 28V
C9
10μF
VIN1 VIN2 VIN3 BIAS
SHDN1
SHDN2
SHDN3
SHDN1
SHDN2
SHDN3
D3
BOOST
SW2
SW1
C1
10μF
L2
10μH
D2
L1
D1 4.7μH
VOUT1
8V
250mA
C8
100nF
R1
105k
LT3570
FB1
SS1
VC1
R2
11.5k
FB2
SS2
VC2
VOUT2
5V
R3
118k
C2
22μF
C5
10nF
R8
25k
R4
22.1k
C7
1nF
R7
25k
NPN_DRV
C6
1nF
Q1
10nF
VOUT3
3.3V
500mA
R5
34.0k
RT
FB3
GND
SYNC
C3
2.2μF
R6
10.7k
R9
20.0k
3570 TA02
“Dying Gasp” System
VIN
12V
C9
10μF
C10
0.1μF
VIN1 VIN2 VIN3 BIAS
SHDN1
SHDN2
SHDN3
BOOST
SW2
VOUT1
34V
SW1
R1
442k
FB1
SS1
VC1
R2
10.5k
LT3570
FB2
SS2
VC2
L2
22μH
VOUT2
3.3V
200mA
R3
205k
C2
22μF
R8
51k
C5
10nF
R4
64.9k
C7
1nF
R7
20k
C4
10nF
C8
100nF
D2
L1
D1 47μH
C1
10μF
D3
NPN_DRV
C6
1nF
Q1
R5
137k
RT
SYNC
R9
44.2k
FB3
GND
R6
64.9k
VOUT3
2.5V
200mA
C3
2.2μF
3570 TA03
3570fb
18
LT3570
PACKAGE DESCRIPTION
UF Package
24-Lead Plastic QFN (4mm × 4mm)
(Reference LTC DWG # 05-08-1697)
BOTTOM VIEW—EXPOSED PAD
4.00 ± 0.10
(4 SIDES)
0.70 ±0.05
R = 0.115
TYP
0.75 ± 0.05
PIN 1
TOP MARK
(NOTE 6)
PIN 1 NOTCH
R = 0.20 TYP OR
0.35 × 45° CHAMFER
23 24
0.40 ± 0.10
1
2
4.50 ± 0.05
2.45 ± 0.05
3.10 ± 0.05 (4 SIDES)
2.45 ± 0.10
(4-SIDES)
PACKAGE
OUTLINE
(UF24) QFN 0105
0.200 REF
0.25 ±0.05
0.50 BSC
0.25 ± 0.05
0.00 – 0.05
0.50 BSC
NOTE:
1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WGGD-X)—TO BE APPROVED
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE, IF PRESENT
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
FE Package
20-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1663)
Exposed Pad Variation CB
6.40 – 6.60*
(.252 – .260)
3.86
(.152)
3.86
(.152)
20 1918 17 16 15 14 13 12 11
6.60 ±0.10
2.74
(.108)
4.50 ±0.10
6.40
2.74 (.252)
(.108) BSC
SEE NOTE 4
0.45 ±0.05
1.05 ±0.10
0.65 BSC
1 2 3 4 5 6 7 8 9 10
RECOMMENDED SOLDER PAD LAYOUT
4.30 – 4.50*
(.169 – .177)
0.09 – 0.20
(.0035 – .0079)
0.25
REF
0.50 – 0.75
(.020 – .030)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
3. DRAWING NOT TO SCALE
1.20
(.047)
MAX
0° – 8°
0.65
(.0256)
BSC
0.195 – 0.30
(.0077 – .0118)
TYP
0.05 – 0.15
(.002 – .006)
FE20 (CB) TSSOP 0204
4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LT3570
TYPICAL APPLICATION
PDA Core
VIN
4V TO 12V
C9
10μF
VIN1 VIN2 VIN3 BIAS
SHDN1
SHDN2
SHDN3
D1
VOUT1
15V
200mA
SHDN1
SHDN2
SHDN3
C8
100nF
SW2
R1
191k
LT3570
FB1
SS1
VC1
R2
10.7k
FB2
SS2
VC2
VOUT2
3.3V
500mA
R3
34k
C2
22μF
R8
25k
C5
10nF
R4
10.7k
C7
1nF
R7
25k
NPN_DRV
C6
1nF
C4
10nF
L2
8.2μH
D2
L1
12μH
SW1
C1
10μF
D3
BOOST
Q1
R5
13.7k
SYNC
RT
FB3
GND
R6
10.7k
R9
20k
VOUT3
1.8V
500mA
C3
4.7μF
3570 TA04
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1767
1.5A, 1.25MHz Step-Down Switching Regulator
3V to 25V Input, VREF = 1.2V, Synchronizable Up to 2MHz,
MSOP Package
LT1930/LT1930A
1A (ISW), 1.2MHz/2.2MHz, High Efﬁciency Step-Up DC/DC
Converter
VIN: 2.6V to 16V, VOUT(MAX) = 34V, IQ = 4.2mA/5.5mA,
ISD < 1μA, ThinSOT™ Package
LT1939
25V, 2.4MHz Step-Down DC/DC Converter and LDO Controller
VIN: 3V to 40V, VOUT(MIN) = 0.8V, IQ = 2mA, ISD < 1μA,
3mm × 3mm DFN
LT1943
Quad Output, 2.6A Buck, 2.6A Boost, 0.3A Boost, 0.4A Inverter
1.2MHz TFT DC/DC Converter
VIN: 4.5V to 22V, VOUT(MAX) = 40V, IQ = 10mA, ISD < 35μA,
TSSOP28E Package
LT1945
Dual Output Pos/Neg 350mA (ISW), Constant Off-Time,
High Efﬁciency Step-Up DC/DC Converter
VIN: 1.2V to 15V, VOUT(MAX) = ±34V, IQ = 20μA, ISD < 1μA,
10-Pin MS Package
LT3463
Dual Output Pos/Neg 250mA (ISW), Constant Off-Time, High
Efﬁciency Step-Up DC/DC Converter with Integrated Schottkys
VIN: 2.4V to 15V, VOUT(MAX) = ±40V, IQ = 40μA, ISD < 1μA,
3mm × 3mm DFN10 Package
LT3467
1.1A, 1.3MHz Step Up DC/DC Converter with Integrated Soft-Start
VIN: 2.4V to 16V, VOUT(MAX) = 40V, ISD < 1 μA, Low proﬁle
(1mm) SOT-23 Package
LT3500
40V, 2A, 2.4MHz Step-Down DC/DC Converter and LDO Controller
VIN: 3V to 40V, VOUT(MIN) = 0.8V, IQ = 2mA, ISD < 1μA,
3mm × 3mm DFN
LT3507
36V, 2.5MHz Triple (2.4A, 1.5A, 1.5A) Step-Down DC/DC Converter VIN: 4V to 36V, VOUT(MIN) = 0.8V, IQ = 7mA, ISD < 1μA,
5mm × 7mm QFN38 Package
and LDO Controller
ThinSOT is a trademark of Linear Technology Corporation.
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20 Linear Technology Corporation
LT 0309 REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2008

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