LTC1624
High Efficiency SO-8
N-Channel Switching
Regulator Controller
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DESCRIPTION
FEATURES
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The LTC®1624 is a current mode switching regulator
controller that drives an external N-channel power MOSFET
using a fixed frequency architecture. It can be operated in
all standard switching configurations including boost,
step-down, inverting and SEPIC. Burst ModeTM operation
provides high efficiency at low load currents. A maximum
high duty cycle limit of 95% provides low dropout operation
which extends operating time in battery-operated systems.
N-Channel MOSFET Drive
Implements Boost, Step-Down, SEPIC
and Inverting Regulators
Wide VIN Range: 3.5V to 36V Operation
Wide VOUT Range: 1.19V to 30V in Step-Down
Configuration
± 1% 1.19V Reference
Low Dropout Operation: 95% Duty Cycle
200kHz Fixed Frequency
Low Standby Current
Very High Efficiency
Remote Output Voltage Sense
Logic-Controlled Micropower Shutdown
Internal Diode for Bootstrapped Gate Drive
Current Mode Operation for Excellent Line and
Load Transient Response
Available in an 8-Lead SO Package
The operating frequency is internally set to 200kHz, allowing
small inductor values and minimizing PC board space. The
operating current level is user-programmable via an external
current sense resistor. Wide input supply range allows
operation from 3.5V to 36V (absolute maximum).
A multifunction pin (I TH / RUN) allows external
compensation for optimum load step response plus
shutdown. Soft start can also be implemented with the
ITH /RUN pin to properly sequence supplies.
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APPLICATIONS
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Notebook and Palmtop Computers, PDAs
Cellular Telephones and Wireless Modems
Battery-Operated Digital Devices
DC Power Distribution Systems
Battery Chargers
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a trademark of Linear Technology Corporation.
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TYPICAL APPLICATION
1000pF
SENSE –
ITH /RUN
CC
470pF
RC
6.8k
VIN
4.8V TO 28V
VIN
RSENSE
0.05Ω
BOOST
LTC1624
VFB
TG
GND
SW
M1
Si4412DY
CB
0.1µF
100pF
+
D1
MBRS340T3
L1
10µH
R2
35.7k
R1
20k
Figure 1. High Efficiency Step-Down Converter
CIN
22µF
35V
×2
VOUT
3.3V
2A
+
COUT
100µF
10V
×2
1624 F01
1
LTC1624
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SYMBOL
PARAMETER
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ELECTRICAL CHARACTERISTICS
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Input Supply Voltage (VIN)......................... 36V to – 0.3V
Topside Driver Supply Voltage (BOOST)....42V to – 0.3V
Switch Voltage (SW).................................. 36V to – 0.6V
Differential Boost Voltage
(BOOST to SW) ....................................7.8V to – 0.3V
SENSE – Voltage
VIN < 15V .................................. (VIN + 0.3V) to – 0.3V
VIN ≥ 15V .......................... (VIN +0.3V) to (VIN – 15V)
ITH/RUN, VFB Voltages ............................ 2.7V to – 0.3V
Peak Driver Output Current < 10µs (TG) .................... 2A
Operating Temperature Range
LTC1624CS ............................................ 0°C to 70°C
LTC1624IS ......................................... – 40°C to 85°C
Junction Temperature (Note 1)............................. 125°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
ORDER PART
NUMBER
TOP VIEW
SENSE – 1
8
VIN
ITH /RUN 2
7
BOOST
VFB 3
6
TG
GND 4
5
SW
LTC1624CS8
LTC1624IS8
S8 PART MARKING
S8 PACKAGE
8-LEAD PLASTIC SO
1624
1624I
TJMAX = 125°C, θJA = 110°C/ W
Consult factory for Military grade parts.
TA = 25°C, VIN = 15V, unless otherwise noted.
CONDITIONS
MIN
TYP
MAX
UNITS
10
50
nA
1.19
1.2019
V
0.002
0.01
%/V
0.5
– 0.5
0.8
– 0.8
%
%
1.28
1.32
V
550
16
900
30
µA
µA
Main Control Loop
IIN VFB
Feedback Current
(Note 2)
VFB
Feedback Voltage
(Note 2)
∆VLINE REG
Reference Voltage Line Regulation
VIN = 3.6V to 20V (Note 2)
∆VLOAD REG
Output Voltage Load Regulation
(Note 2)
ITH Sinking 5µA
ITH Sourcing 5µA
VOVL
Output Overvoltage Lockout
IQ
Input DC Supply Current
Normal Mode
Shutdown
VITH/RUN
Run Threshold
IITH/RUN
Run Current Source
Run Pullup Current
∆VSENSE(MAX) Maximum Current Sense Threshold
●
●
●
1.24
(Note 3)
VITH/RUN = 0V
0.6
0.8
VITH/RUN = 0.3V
VITH/RUN = 1V
– 0.8
– 50
– 2.5
–160
– 5.0
– 350
µA
µA
VFB = 1.0V
145
160
185
mV
50
50
150
150
ns
ns
175
200
225
kHz
4.8
5.15
5.5
V
3
5
%
TG tr
TG tf
TG Transition Time
Rise Time
Fall Time
fOSC
Oscillator Frequency
VBOOST
Boost Voltage
SW = 0V, IBOOST = 5mA, VIN = 8V
∆VBOOST
Boost Load Regulation
SW = 0V, IBOOST = 2mA to 20mA
CLOAD = 3000pF
CLOAD = 3000pF
●
The ● denotes specifications which apply over the full operating
temperature range.
LTC1624CS: 0°C ≤ TA ≤ 70°C
LTC1624IS: – 40°C ≤ TA ≤ 85°C
Note 1: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
2
1.1781
V
TJ = TA + (PD • 110°C/W)
Note 2: The LTC1624 is tested in a feedback loop which servos VFB to
the midpoint for the error amplifier (VITH = 1.8V).
Note 3: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
LTC1624
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TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Input Voltage
VOUT = 3.3V
Efficiency vs Load Current
VOUT = 3.3V
100
95
90
VIN = 5V
95
VIN = 10V
90
EFFICIENCY (%)
VOUT = 3.3V
RSENSE = 0.033Ω
85
80
ILOAD = 1A
85
ILOAD = 0.1A
80
70
70
0.001
0.01
0.1
1
LOAD CURRENT (A)
10
90
85
80
75
0
20
15
10
INPUT VOLTAGE (V)
5
25
1624 G07
70
0.001
30
100
Input Supply Current vs
Input Voltage
700
0.7
VOUT = 5V
RSENSE = 0.033Ω
RSENSE = 0.033Ω
VOUT DROP OF 5%
VFB = 1.21V
600
SUPPLY CURRENT (µA)
0.6
ILOAD = 1A
0.5
VIN – VOUT (V)
85
ILOAD = 0.1A
80
10
1624 G08
VIN – VOUT Dropout Voltage
vs Load Current
90
0.01
0.1
1
LOAD CURRENT (A)
1624 G09
Efficiency vs Input Voltage
VOUT = 5V
95
VOUT = 5V
VIN = 10V
RSENSE = 0.033Ω
95
75
75
EFFICIENCY (%)
100
VOUT = 3.3V
RSENSE = 0.033Ω
EFFICIENCY (%)
100
EFFICIENCY (%)
Efficiency vs Load Current
VOUT = 5V
0.4
0.3
0.2
SLEEP MODE
500
400
300
200
75
0.1
100
70
0
0
SHUTDOWN
0
5
20
15
10
INPUT VOLTAGE (V)
30
25
0
0.5
1.0
1.5
2.0
LOAD CURRENT (A)
2.5
0
3.0
6
5
5
Boost Voltage vs Temperature
ILOAD = 1mA
VIN = 5V
BOOST VOLTAGE (V)
BOOST VOLTAGE (V)
BOOST VOLTAGE (V)
35
6.0
VIN = 15V
2
30
1624 G05
Boost Load Regulation
Boost Line Regulation
6
3
20
15
25
10
INPUT VOLTAGE (V)
1624 G11
1624 G10
4
5
4
3
2
5.5
5.0
4.5
1
0
1
IBOOST = 1mA
VSW = 0V
0
5
20
15
25
10
INPUT VOLTAGE (V)
VSW = 0V
30
35
1624 G04
0
0
5
20
15
25
10
BOOST LOAD CURRENT (mA)
30
1624 G06
4.0
–40 –15
60
35
85
10
TEMPERATURE (°C)
110
135
1624 G15
3
LTC1624
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IITH vs VITH
200
2.4
IITH (µA)
VITH /RUN (V)
150
ACTIVE
MODE
50
1.2
ACTIVE
MODE
0.8
SHUTDOWN
0
3
0
IOUT(MAX)
SHUTDOWN
0
0
0.8
IOUT
(a)
1.2
VITH (V)
2.4
(b)
1624 G01
300
250
3
ITH /RUN = 0V
150
2
100
1
50
0
–40 –15
100
0
1.25
0
135
170
150
100
50
50
110
Maximum Current Sense
Threshold vs Temperature
CURRENT SENSE THRESHOLD (mV)
FREQUENCY (kHz)
150
60
35
85
10
TEMPERATURE (°C)
1624 G14
200
200
0.75
1.00
0.50
FEEDBACK VOLTAGE
ITH /RUN = 1V
1624 G02
VOUT IN REGULATION
0.25
4
200
250
250
0
5
Operating Frequency vs
Temperature
Frequency vs Feedback Voltage
FREQUENCY (kHz)
ITH/RUN Pin Source Current vs
Temperature
ITH/RUN PIN SOURCE CURRENT WITH VITH = 0V (µA)
VITH vs Output Current
ITH/RUN PIN SOURCE CURRENT WITH VITH = 1V (µA)
TYPICAL PERFORMANCE CHARACTERISTICS
0
–40 –15
VFB = 0V
60
10
85
35
TEMPERATURE (°C)
1624 G03
110
135
1448 G12
168
166
164
162
160
158
156
154
152
150
–40 –15
60
35
10
85
TEMPERATURE (°C)
110
135
1448 G13
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PIN FUNCTIONS
SENSE – (Pin 1): Connects to the (–) input for the current
comparator. Built-in offsets between the SENSE – and VIN
pins in conjunction with RSENSE set the current trip thresholds. Do not pull this pin more than 15V below VIN or more
than 0.3V below ground.
ITH/RUN (Pin 2): Combination of Error Amplifier Compensation Point and Run Control Inputs. The current comparator threshold increases with this control voltage.
Nominal voltage range for this pin is 1.19V to 2.4V. Forcing
this pin below 0.8V causes the device to be shut down. In
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shutdown all functions are disabled and TG pin is held low.
VFB (Pin 3): Receives the feedback voltage from an external resistive divider across the output.
GND (Pin 4): Ground. Connect to the (–) terminal of COUT,
the Schottky diode and the (–) terminal of CIN.
SW (Pin 5): Switch Node Connection to Inductor. In stepdown applications the voltage swing at this pin is from a
Schottky diode (external) voltage drop below ground to
VIN.
LTC1624
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PIN FUNCTIONS
TG (Pin 6): High Current Gate Drive for Top N-Channel
MOSFET. This is the output of a floating driver with a
voltage swing equal to INTVCC superimposed on the
switch node voltage SW.
BOOST (Pin 7): Supply to Topside Floating Driver. The
bootstrap capacitor CB is returned to this pin. Voltage
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OPERATIO
swing at this pin is from INTVCC to VIN + INTVCC in stepdown applications. In non step-down topologies the voltage at this pin is constant and equal to INTVCC if SW = 0V.
VIN (Pin 8): Main Supply Pin and the (+) Input to the
Current Comparator. Must be closely decoupled to ground.
(Refer to Functional Diagram)
Main Control Loop
The LTC1624 uses a constant frequency, current mode
architecture. During normal operation, the top MOSFET is
turned on each cycle when the oscillator sets the RS latch
and turned off when the main current comparator I1 resets
the RS latch. The peak inductor current at which I1 resets
the RS latch is controlled by the voltage on the ITH /RUN
pin, which is the output of error amplifier EA. The VFB pin,
described in the pin functions, allows EA to receive an
output feedback voltage from an external resistive divider.
When the load current increases, it causes a slight
decrease in VFB relative to the 1.19V reference, which in
turn causes the ITH /RUN voltage to increase until the
average inductor current matches the new load current.
While the top MOSFET is off, the internal bottom MOSFET
is turned on for approximately 300ns to 400ns to recharge
the bootstrap capacitor CB.
The top MOSFET driver is biased from the floating bootstrap capacitor CB that is recharged during each off cycle.
The dropout detector counts the number of oscillator
cycles that the top MOSFET remains on and periodically
forces a brief off period to allow CB to recharge.
The main control loop is shut down by pulling the ITH /RUN
pin below its 1.19V clamp voltage. Releasing ITH /RUN
allows an internal 2.5µA current source to charge compensation capacitor CC. When the ITH /RUN pin voltage
reaches 0.8V the main control loop is enabled with the ITH /
RUN voltage pulled up by the error amp. Soft start can be
implemented by ramping the voltage on the ITH /RUN pin
from 1.19V to its 2.4V maximum (see Applications Information section).
Comparator OV guards against transient output overshoots >7.5% by turning off the top MOSFET and keeping
it off until the fault is removed.
Low Current Operation
The LTC1624 is capable of Burst Mode operation in which
the external MOSFET operates intermittently based on
load demand. The transition to low current operation
begins when comparator B detects when the ITH /RUN
voltage is below 1.5V. If the voltage across RSENSE does
not exceed the offset of I2 (approximately 20mV) for one
full cycle, then on following cycles the top and internal
bottom drives are disabled. This continues until the ITH
voltage exceeds 1.5V, which causes drive to be returned to
the TG pin on the next cycle.
INTVCC Power/Boost Supply
Power for the top and internal bottom MOSFET drivers is
derived from VIN. An internal regulator supplies INTVCC
power. To power the top driver in step-down applications
an internal high voltage diode recharges the bootstrap
capacitor CB during each off cycle from the INTVCC supply.
A small internal N-channel MOSFET pulls the switch node
(SW) to ground each cycle after the top MOSFET has
turned off ensuring the bootstrap capacitor is kept fully
charged.
5
R1
CC
RC
R2
VFB
ITH /RUN
3
2
VFB
1.28V
30k
–
+
OV
EA
gm = 1m
180k
1.19V
–
+
–
+
1.19V
1.19V
REF
1.19V
+
8k
I1
–
S
R
OSC
Q
1.5V
ST
4k
200kHz
SLOPE
COMP
+
–
SLOPE
COMP
COSC
200kHz
0.8V
3µA
8k
4k
VIN
I2
B
RUN
3µA
+
–
SENSE –
DROPOUT
DET
–
+
1
1-SHOT
400ns
SWITCH
LOGIC
VIN
INTVCC
5.6V
INTVCC
REG
N-CHANNEL
MOSFET
FLOATING
DRIVER
DB
INTVCC
4
5
6
7
GND
SW
TG
BOOST
CB
D1
L1
N-CHANNEL
MOSFET
+
1624 FD
COUT
VOUT
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CIN
VIN
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FUNCTIONAL DIAGRA
Ω
2.5µA
8
RSENSE
+
LTC1624
(Shown in a step-down application)
LTC1624
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APPLICATIONS INFORMATION
The LTC1624 can be used in a wide variety of switching
regulator applications, the most common being the stepdown converter. Other switching regulator architectures
include step-up, SEPIC and positive-to-negative converters.
The basic LTC1624 step-down application circuit is shown
in Figure 1 on the first page. External component selection
is driven by the load requirement and begins with the
selection of RSENSE. Once RSENSE is known, the inductor
can be chosen. Next, the power MOSFET and D1 are
selected. Finally, CIN and COUT are selected. The circuit
shown in Figure 1 can be configured for operation up to an
input voltage of 28V (limited by the external MOSFETs).
Step-Down Converter: RSENSE Selection for
Output Current
RSENSE is chosen based on the required output current.
The LTC1624 current comparator has a maximum threshold of 160mV/RSENSE. The current comparator threshold
sets the peak of the inductor current, yielding a maximum
average output current IMAX equal to the peak value less
half the peak-to-peak ripple current, ∆IL.
Allowing a margin for variations in the LTC1624 and
external component values yields:
RSENSE =
100mV
IMAX
The LTC1624 works well with values of RSENSE from
0.005Ω to 0.5Ω.
Step-Down Converter: Inductor Value Calculation
With the operating frequency fixed at 200kHz smaller
inductor values are favored. Operating at higher frequencies generally results in lower efficiency because of
MOSFET gate charge losses. In addition to this basic
trade-off, the effect of inductor value on ripple current and
low current operation must also be considered.
The inductor value has a direct effect on ripple current. The
inductor ripple current ∆IL decreases with higher inductance and increases with higher VIN or VOUT:
V
+V
V −V
∆IL = IN OUT OUT D
VIN + VD
f L
( )( )
where VD is the output Schottky diode forward drop.
Accepting larger values of ∆IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆IL = 0.4(IMAX). Remember, the
maximum ∆IL occurs at the maximum input voltage.
The inductor value also has an effect on low current
operation. Lower inductor values (higher ∆IL) will cause
Burst Mode operation to begin at higher load currents,
which can cause a dip in efficiency in the upper range of
low current operation. In Burst Mode operation lower
inductance values will cause the burst frequency to
decrease. In general, inductor values from 5µH to 68µH
are typical depending on the maximum input voltage and
output current. See also Modifying Burst Mode Operation
section.
Step-Down Converter: Inductor Core Selection
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ® cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires
more turns of wire and, therefore, copper losses will
increase.
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
ferrite. A reasonable compromise from the same manufacturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire.
Because they generally lack a bobbin, mounting is more
difficult. However, designs for surface mount that do not
increase the height significantly are available.
Kool Mu is a registered trademark of Magnetics, Inc.
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LTC1624
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APPLICATIONS INFORMATION
Step-Down Converter: Power MOSFET Selection
One external N-channel power MOSFET must be selected
for use with the LTC1624 for the top (main) switch.
characteristics. The constant k = 2.5 can be used to
estimate the contributions of the two terms in the PMAIN
dissipation equation.
The peak-to-peak gate drive levels are set by the INTVCC
voltage. This voltage is typically 5V. Consequently, logic
level threshold MOSFETs must be used in most LTC1624
applications. If low input voltage operation is expected
(VIN < 5V) sublogic level threshold MOSFETs should be
used. Pay close attention to the BVDSS specification for the
MOSFETs as well; many of the logic level MOSFETs are
limited to 30V or less.
Step-Down Converter: Output Diode Selection (D1)
The Schottky diode D1 shown in Figure 1 conducts during
the off-time. It is important to adequately specify the diode
peak current and average power dissipation so as not to
exceed the diode ratings.
Selection criteria for the power MOSFET include the “ON”
resistance RDS(ON), reverse transfer capacitance CRSS,
input voltage and maximum output current. When the
LTC1624 is operating in continuous mode the duty cycle
for the top MOSFET is given by:
+V
V
Main Switch Duty Cycle = OUT D
VIN + VD
The MOSFET power dissipation at maximum output
current is given by:
( )( ) ( )
1.85
k(VIN) (IMAX)(CRSS)(f)
2
V
+ VD
PMAIN = OUT
IMAX 1 + δ RDS ON +
VIN + VD
where δ is the temperature dependency of RDS(ON) and k
is a constant inversely related to the gate drive current.
MOSFETs have I2R losses, plus the PMAIN equation
includes an additional term for transition losses that are
highest at high output voltages. For VIN < 20V the high
current efficiency generally improves with larger MOSFETs,
while for VIN > 20V the transition losses rapidly increase to
the point that the use of a higher RDS(ON) device with lower
CRSS actual provides higher efficiency. The diode losses
are greatest at high input voltage or during a short circuit
when the diode duty cycle is nearly 100%.
The term (1+ δ) is generally given for a MOSFET in the form
of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs. CRSS is usually specified in the MOSFET
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The most stressful condition for the output diode is under
short circuit (VOUT = 0V). Under this condition, the diode
must safely handle ISC(PK) at close to 100% duty cycle.
Under normal load conditions, the average current conducted by the diode is simply:
IDIODE AVG = ILOAD AVG VIN − VOUT
V +V
IN
D
( )
( )
Remember to keep lead lengths short and observe proper
grounding (see Board Layout Checklist) to avoid ringing
and increased dissipation.
The forward voltage drop allowable in the diode is calculated from the maximum short-circuit current as:
VD ≈
VIN + VD
ISC AVG VIN
PD
( )
where PD is the allowable diode power dissipation and will
be determined by efficiency and/or thermal requirements
(see Efficiency Considerations).
Step-Down Converter: CIN and COUT Selection
In continuous mode the source current of the top
N-channel MOSFET is a square wave of approximate duty
cycle VOUT/VIN. To prevent large voltage transients, a low
ESR input capacitor sized for the maximum RMS current
must be used. The maximum RMS capacitor current is
given by:
CIN Required IRMS ≈ IMAX
[V ( V
OUT IN − VOUT
)]
1/ 2
VIN
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is com-
LTC1624
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APPLICATIONS INFORMATION
monly used for design because even significant deviations
do not offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on only 2000 hours
of life. This makes it advisable to further derate the
capacitor, or to choose a capacitor rated at a higher
temperature than required. Several capacitors may also be
paralleled to meet size or height requirements in the
design. Always consult the manufacturer if there is any
question.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR requirement is satisfied the capacitance is adequate for filtering.
The output ripple (∆VOUT) is determined by:
1
∆VOUT ≈ ∆IL ESR +
4 fCOUT
where f = operating frequency, COUT = output capacitance
and ∆IL = ripple current in the inductor. The output ripple
is highest at maximum input voltage since ∆IL increases
with input voltage. With ∆IL = 0.4IOUT(MAX) the output
ripple will be less than 100mV at maximum VIN, assuming:
COUT Required ESR < 2RSENSE
Manufacturers such as Nichicon, United Chemicon and
SANYO should be considered for high performance
through-hole capacitors. The OS-CON semiconductor
dielectric capacitor available from SANYO has the lowest
ESR(size) product of any aluminum electrolytic at a somewhat higher price. Once the ESR requirement for COUT has
been met, the RMS current rating generally far exceeds
the IRIPPLE(P-P) requirement.
In surface mount applications multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in
surface mount configurations. In the case of tantalum it is
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS series of surface mount tantalums, available in case
heights ranging from 2mm to 4mm. Other capacitor types
include SANYO OS-CON, Nichicon WF series and Sprague
595D series and the new ceramics. Ceramic capacitors are
now available in extremely low ESR and high ripple current
ratings that are ideal for input capacitor applications.
Consult the manufacturer for other specific recommendations.
INTVCC Regulator
An internal regulator produces the 5V supply that powers
the drivers and internal circuitry within the LTC1624.
Good VIN bypassing is necessary to supply the high
transient currents required by the MOSFET gate drivers.
High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the LTC1624 to be
exceeded. The supply current is dominated by the gate
charge supply current as discussed in the Efficiency
Considerations section. The junction temperature can be
estimated by using the equations given in Note 1 of the
Electrical Characteristics table. For example, the LTC1624
is limited to less than 17mA from a 30V supply:
TJ = 70°C + (17mA)(30V)(110°C/W) = 126°C
To prevent maximum junction temperature from being
exceeded, the input supply current must be checked
operating in continuous mode at maximum VIN.
Step-Down Converter: Topside MOSFET Driver
Supply (CB, DB)
An external bootstrap capacitor CB connected to the BOOST
pin supplies the gate drive voltage for the topside MOSFET.
Capacitor CB in the functional diagram is charged through
internal diode DB from INTVCC when the SW pin is low.
When the topside MOSFET is to be turned on, the driver
places the CB voltage across the gate to source of the
MOSFET. This enhances the MOSFET and turns on the
topside switch. The switch node voltage SW rises to VIN
and the BOOST pin rises to VIN + INTVCC. The value of the
boost capacitor CB needs to be 50 times greater than the
total input capacitance of the topside MOSFET. In most
applications 0.1µF is adequate.
Significant efficiency gains can be realized by supplying
topside driver operating voltage from the output, since the
VIN current resulting from the driver and control currents
will be scaled by a factor of (Duty Cycle)/(Efficiency). For
5V regulators this simply means connecting the BOOST
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pin through a small Schottky diode (like a Central
CMDSH-3) to VOUT as shown in Figure 10. However, for
3.3V and other lower voltage regulators, additional circuitry is required to derive boost supply power from the
output.
For low input voltage operation (VIN < 7V), a Schottky
diode can be connected from VIN to BOOST to increase the
external MOSFET gate drive voltage. Be careful not to
exceed the maximum voltage on BOOST to SW pins
of 7.8V.
Output Voltage Programming
The output voltage is set by a resistive divider according
to the following formula:
ITH /RUN
ITH /RUN
D1
CC
CC
RC
RC
(a)
(b)
ITH /RUN
R1
D1
CC
C1
RC
(c)
R2
VOUT = 1.19V 1 +
R1
1624 F03
Figure 3. ITH / RUN Pin Interfacing
The external resistive divider is connected to the output as
shown in Figure 2, allowing remote voltage sensing. When
using remote sensing, a local 100Ω resistor should be
connected from L1 to R2 to prevent VOUT from running
away if the sense lead is disconnected.
VOUT
L1
R2
100pF
Soft start can be implemented by ramping the voltage on
ITH /RUN during start-up as shown in Figure 3(c). As the
voltage on ITH/RUN ramps from 1.19V to 2.4V the internal
peak current limit is also ramped at a proportional linear
rate. The peak current limit begins at approximately
10mV/RSENSE (at VITH/RUN = 1.4V) and ends at:
160mV/RSENSE (VITH/RUN = 2.4V)
The output current thus ramps up slowly, charging the
output capacitor. The peak inductor current and maximum
output current are as follows:
VFB
LTC1624
3.3V
OR 5V
R1
GND
1624 F02
IL(PEAK) = (VITH/RUN – 1.3V)/(6.8RSENSE)
Figure 2. Setting the LTC1624 Output Voltage
IOUT(MAX) = ILPEAK – ∆IL / 2
ITH /RUN Function
with ∆IL = ripple current in the inductor.
The ITH /RUN pin is a dual purpose pin that provides the
loop compensation and a means to shut down the LTC1624.
Soft start can also be implemented with this pin. Soft start
reduces surge currents from VIN by gradually increasing
the internal current limit. Power supply sequencing can
also be accomplished using this pin.
During normal operation the voltage on the ITH /RUN pin
will vary from 1.19V to 2.4V depending on the load current.
Pulling the ITH /RUN pin below 0.8V puts the LTC1624 into
a low quiescent current shutdown (IQ < 30µA). This pin can
be driven directly from logic as shown in Figures 3(a)
and 3(b).
An internal 2.5µA current source charges up the external
capacitor CC. When the voltage on ITH /RUN reaches 0.8V
the LTC1624 begins operating. At this point the error
amplifier pulls up the ITH /RUN pin to its maximum of 2.4V
(assuming VOUT is starting low).
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
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what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can be
expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1624 circuits:
1. LTC1624 VIN current
2. I2R losses
3. Topside MOSFET transition losses
4. Voltage drop of the Schottky diode
1. The VIN current is the sum of the DC supply current IQ,
given in the Electrical Characteristics table, and the
MOSFET driver and control currents. The MOSFET
driver current results from switching the gate
capacitance of the power MOSFET. Each time a MOSFET
gate is switched from low to high to low again, a packet
of charge dQ moves from INTVCC to ground. The
resulting dQ/dt is a current out of VIN which is typically
much larger than the control circuit current. In
continuous mode, IGATECHG = f (QT + QB), where QT and
QB are the gate charges of the topside and internal
bottom side MOSFETs.
By powering BOOST from an output-derived source
(Figure 10 application), the additional VIN current
resulting from the topside driver will be scaled by a
factor of (Duty Cycle)/(Efficiency). For example, in a
20V to 5V application, 5mA of INTVCC current results in
approximately 1.5mA of VIN current. This reduces the
midcurrent loss from 5% or more (if the driver was
powered directly from VIN) to only a few percent.
2. I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current shunt. In continuous
mode the average output current flows through L but is
“chopped” between the topside main MOSFET/current
shunt and the Schottky diode. The resistances of the
topside MOSFET and RSENSE multiplied by the duty
cycle can simply be summed with the resistance of L to
obtain I2R losses. (Power is dissipated in the sense
resistor only when the topside MOSFET is on. The I2R
loss is thus reduced by the duty cycle.) For example, at
50% DC, if RDS(ON) = 0.05Ω, RL = 0.15Ω and RSENSE =
0.05Ω, then the effective total resistance is 0.2Ω. This
results in losses ranging from 2% to 8% for VOUT = 5V
as the output current increases from 0.5A to 2A. I2R
losses cause the efficiency to drop at high output
currents.
3. Transition losses apply only to the topside MOSFET(s),
and only when operating at high input voltages (typically
20V or greater). Transition losses can be estimated
from:
Transition Loss = 2.5(VIN)1.85 (IMAX)(CRSS)(f)
4. The Schottky diode is a major source of power loss at
high currents and gets worse at high input voltages.
The diode loss is calculated by multiplying the forward
voltage drop times the diode duty cycle multiplied by
the load current. For example, assuming a duty cycle of
50% with a Schottky diode forward voltage drop of
0.5V, the loss is a relatively constant 5%.
As expected, the I2R losses and Schottky diode loss
dominate at high load currents. Other losses including
CIN and COUT ESR dissipative losses and inductor core
losses generally account for less than 2% total additional
loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in DC (resistive) load
current. When a load step occurs, VOUT immediately shifts
by an amount equal to (∆ILOAD • ESR), where ESR is the
effective series resistance of COUT. ∆ILOAD also begins to
charge or discharge COUT which generates a feedback
error signal. The regulator loop then acts to return VOUT to
its steady-state value. During this recovery time VOUT can
be monitored for overshoot or ringing that would indicate
a stability problem. The ITH external components shown in
the Figure 1 circuit will provide adequate compensation for
most applications.
A second, more severe transient, is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
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with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive so that
the load rise time is limited to approximately (25 • CLOAD).
Thus a 10µF capacitor would require a 250µs rise time,
limiting the charging current to about 200mA.
Automotive Considerations: Plugging into the
Cigarette Lighter
As battery-powered devices go mobile there is a natural
interest in plugging into the cigarette lighter in order to
conserve or even recharge battery packs during operation.
But before you connect, be advised: you are plugging into
the supply from hell. The main battery line in an automobile is the source of a number of nasty potential transients,
including load dump, reverse battery and double battery.
Load dump is the result of a loose battery cable. When the
cable breaks connection, the field collapse in the alternator
can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse battery is
just what it says, while double battery is a consequence of
tow-truck operators finding that a 24V jump start cranks
cold engines faster than 12V.
The network shown in Figure 4 is the most straightforward
approach to protect a DC/DC converter from the ravages
of an automotive battery line. The series diode prevents
current from flowing during reverse battery, while the
transient suppressor clamps the input voltage during load
dump. Note that the transient suppressor should not
conduct during double battery operation, but must still
clamp the input voltage below breakdown of the converter.
Although the LTC1624 has a maximum input voltage of
36V, most applications will be limited to 30V by the
MOSFET BVDSS.
Modifying Burst Mode Operation
The LTC1624 automatically enters Burst Mode operation
at low output currents to boost efficiency. The point when
continuous mode operation changes to Burst Mode operation scales as a function of maximum output current.
The output current when Burst Mode operation commences is approximately 8mV/RSENSE (8% of maximum
output current).
With the additional circuitry shown in Figure 5 the LTC1624
can be forced to stay in continuous mode longer at low
output currents. Since the LTC1624 is not a fully synchronous architecture, it will eventually start to skip cycles as
the load current drops low enough. The point when the
minimum on-time (450ns) is reached determines the load
current when cycle skipping begins at approximately 1%
of maximum output current. Using the circuit in Figure 5
the LTC1624 will begin to skip cycles but stays in regulation when IOUT is less than IOUT(MIN):
2
t
f
ON
MIN
VIN + VD
−
IOUT MIN =
V
V
IN
OUT
V
2L
OUT + VD
( )
( )
The transistor Q1 in the circuit of Figure 5 operates as a
current source developing an 18mV offset across the
VIN
+
1000pF
SENSE –
50A IPK
RATING
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
LTC1624
+
TG
LTC1624
R*
L1
SW
1624 F04
Figure 4. Plugging into the Cigarette Lighter
12
–
CIN
RSENSE
100Ω
18mV
Q1
2N2222
VIN
)
where tON(MIN) = 450ns, f = 200kHz.
VIN
12V
(
*R =
(VOUT – 0.7V)
180µA
D1
MBRS340T3
+
Figure 5. Modifying Burst Mode Operation
VOUT
COUT
1624 F05
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100Ω resistor in series with the SENSE – pin. This offset
cancels the internal offset in current comparator I2 (refer
to Functional Diagram). This comparator in conjunction
with the voltage on the ITH /RUN pin determines when to
enter into Burst Mode operation (refer to Low Current
Operation in Operation section). With the additional external offset present, the drive to the topside MOSFET is
always enabled every cycle and constant frequency operation occurs for IOUT > IOUT(MIN).
Step-Down Converter: Design Example
As a design example, assume VIN = 12V(nominal),
VIN = 22V(max), VOUT = 3.3V and IMAX = 2A. RSENSE can
immediately be calculated:
RSENSE = 100mV/2A = 0.05Ω
Assume a 10µH inductor. To check the actual value of the
ripple current the following equation is used:
V
+V
V −V
∆IL = IN OUT OUT D
f L VIN + VD
( )( )
The highest value of the ripple current occurs at the
maximum input voltage:
∆IL =
22V − 3.3V 3.3V + 0.5V
= 1.58AP-P
200kHz 10µH 22V + 0.5V
(
)
The power dissipation on the topside MOSFET can be
easily estimated. Choosing a Siliconix Si4412DY results
in: RDS(ON) = 0.042Ω, CRSS = 100pF. At maximum input
voltage with T(estimated) = 50°C:
PMAIN =
2
3.3V + 0.5V
2A 1 + 0.005 50°C − 25°C 0.042Ω
22V + 0.5V
( ) [ ( )(
)](
1.85
+ 2.5 (22V) (2A)(100pF)(200kHz) = 62mW
( )( )
VORIPPLE = RESR(∆IL) = 0.03Ω (1.58AP-P) = 47mVP-P
Step-Down Converter: Duty Cycle Limitations
At high input to output differential voltages the on-time
gets very small. Due to internal gate delays and response
times of the internal circuitry the minimum recommended
on-time is 450ns. Since the LTC1624’s frequency is internally set to 200kHz a potential duty cycle limitation exists.
When the duty cycle is less than 9%, cycle skipping may
occur which increases the inductor ripple current but does
not cause VOUT to lose regulation. Avoiding cycle skipping
imposes a limit on the input voltage for a given output
voltage only when VOUT < 2.2V using 30V MOSFETs.
(Remember not to exceed the absolute maximum voltage
of 36V.)
VIN(MAX) = 11.1VOUT + 5V
For DC > 9%
Boost Converter Applications
The LTC1624 is also well-suited to boost converter applications. A boost converter steps up the input voltage to a
higher voltage as shown in Figure 6.
VIN
RSENSE
)
The most stringent requirement for the Schottky diode
occurs when VOUT = 0V (i.e. short circuit) at maximum VIN.
In this case the worst-case dissipation rises to:
VIN
PD = ISC AVG VD
VIN + VD
With the 0.05Ω sense resistor ISC(AVG) = 2A will result,
increasing the 0.5V Schottky diode dissipation to 0.98W.
CIN is chosen for an RMS current rating of at least 1.0A at
temperature. COUT is chosen with an ESR of 0.03Ω for low
output ripple. The output ripple in continuous mode will be
highest at the maximum input voltage. The output voltage
ripple due to ESR is approximately:
+
CIN
VIN
SENSE –
L1
BOOST
D1
VOUT
LTC1624
M1
TG
R2
CB
GND
SW
VFB
+
COUT
R1
1624 F06
Figure 6. Boost Converter
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Boost Converters: Power MOSFET Selection
One external N-channel power MOSFET must be selected
for use with the LTC1624 for the switch. In boost applications the source of the power MOSFET is grounded along
with the SW pin. The peak-to-peak gate drive levels are set
by the INTVCC voltage. The gate drive voltage is equal to
approximately 5V for VIN > 5.6V and a logic level MOSFET
can be used. At VIN voltages below 5V the gate drive
voltage is equal to VIN – 0.6V and a sublogic level MOSFET
should be used.
Selection criteria for the power MOSFET include the “ON”
resistance RDS(ON), reverse transfer capacitance CRSS,
input voltage and maximum output current. When the
LTC1624 is operating in continuous mode the duty cycle
for the MOSFET is given by:
Main Switch Duty Cycle = 1−
VIN
VOUT + VD
The MOSFET power dissipation at maximum output current is given by:
( )
VIN MIN
2
1 + δ RDS ON +
PMAIN = IIN MAX 1 −
V
+
V
OUT
D
( )
(
k VOUT
)
1.85
( )(
( ) ( )
)(
)
C
I
200kHz
IN MAX RSS
VOUT + VD
where IIN MAX = IOUT MAX
VIN MIN
( )
( )
( )
δ is the temperature dependency of RDS(ON) and k is a
constant inversely related to the gate drive current.
MOSFETs have I2R losses, plus the PMAIN equation
includes an additional term for transition losses that are
highest at high output voltages. For VOUT < 20V the high
current efficiency generally improves with larger MOSFETs,
while for VOUT > 20V the transition losses rapidly increase
to the point that the use of a higher RDS(ON) device with
lower CRSS actual provides higher efficiency. For additional information refer to Step-Down Converter: Power
MOSFET Selection in the Applications Information
section.
14
Boost Converter: Inductor Selection
For most applications the inductor will fall in the range of
10µH to 100µH. Higher values reduce the input ripple
voltage and reduce core loss. Lower inductor values are
chosen to reduce physical size.
The input current of the boost converter is calculated at full
load current. Peak inductor current can be significantly
higher than output current, especially with smaller inductors and lighter loads. The following formula assumes
continuous mode operation and calculates maximum peak
inductor current at minimum VIN:
( )
∆IL MAX
V
IL PEAK = IOUT MAX OUT +
2
VIN MIN
(
)
( )
( )
The ripple current in the inductor (∆IL) is typically 20% to
30% of the peak inductor current occuring at VIN(MIN) and
IOUT(MAX).
( )
∆IL P-P =
(
)
VIN VOUT + VD − VIN
(200kHz)(L)(VOUT + VD)
with ∆IL(MAX) = ∆IL(P-P) at VIN = VIN(MIN).
Remember boost converters are not short-circuit protected, and that under output short conditions, inductor
current is limited only by the available current of the input
supply, IOUT(OVERLOAD). Specify the maximum inductor
current to safely handle the greater of IL(PEAK) or
IOUT(OVERLOAD). Make sure the inductor’s saturation current rating (current when inductance begins to fall)
exceeds the maximum current rating set by RSENSE.
Boost Converter: RSENSE Selection for Maximum
Output Current
RSENSE is chosen based on the required output current.
Remember the LTC1624 current comparator has a maximum threshold of 160mV/RSENSE. The current comparator threshold sets the peak of the inductor current, yielding
a maximum average output current IOUT(MAX) equal to
IL(PEAK) less half the peak-to-peak ripple current (∆IL),
divided by the output-input voltage ratio (see equation for
IL(PEAK)).
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Allowing a margin for variations in the LTC1624 (without
considering variation in RSENSE), assuming 30% ripple
current in the inductor, yields:
( )
VIN MIN
RSENSE =
IOUT MAX VOUT + VD
100mV
( )
Boost Converter: Output Diode
The output diode conducts current only during the switch
off-time. Peak reverse voltage for boost converters is
equal to the regulator output voltage. Average forward
current in normal operation is equal to output current.
Remember boost converters are not short-circuit protected. Check to be sure the diode’s current rating exceeds
the maximum current set by RSENSE. Schottky diodes such
as Motorola MBR130LT3 are recommended.
Boost Converter: Output Capacitors
The output capacitor is normally chosen by its effective
series resistance (ESR), because this is what determines
output ripple voltage.
Since the output capacitor’s ESR affects efficiency, use
low ESR capacitors for best performance. Boost regulators have large RMS ripple current in the output capacitor
that must be rated to handle the current. The output
capacitor ripple current (RMS) is:
( )
COUT IRIPPLE RMS ≈ IOUT
VOUT − VIN
VIN
CIN IRIPPLE ≈
( )(
)
(200kHz)(L)(VOUT)
0.3 VIN VOUT − VIN
The input capacitor can see a very high surge current when
a battery is suddenly connected and solid tantalum capacitors can fail under this condition. Be sure to specify surge
tested capacitors.
Boost Converter: Duty Cycle Limitations
The minimum on-time of 450ns sets a limit on how close
VIN can approach VOUT without the output voltage overshooting and tripping the overvoltage comparator. Unless
very low values of inductances are used, this should never
be a problem. The maximum input voltage in continuous
mode is:
VIN(MAX) = 0.91VOUT + 0.5V
For DC = 9%
SEPIC Converter Applications
The LTC1624 is also well-suited to SEPIC (Single Ended
Primary Inductance Converter) converter applications.
The SEPIC converter shown in Figure 7 uses two inductors. The advantage of the SEPIC converter is the input
voltage may be higher or lower than the output voltage.
The first inductor L1 together with the main N-channel
MOSFET switch resemble a boost converter. The second
inductor L2 and output diode D1 resemble a flyback or
buck-boost converter. The two inductors L1 and L2 can be
independent but also can be wound on the same core since
VIN
Output ripple is then simply: VOUT = RESR (∆IL(RMS)).
Boost Converter: Input Capacitors
The input capacitor of a boost converter is less critical due
to the fact that the input current waveform is triangular,
and does not contain large square wave currents as found
in the output capacitor. The input voltage source impedance determines the size of the capacitor that is typically
10µF to 100µF. A low ESR is recommended although not
as critical as the output capacitor and can be on the order
of 0.3Ω. Input capacitor ripple current for the LTC1624
used as a boost converter is:
RSENSE
+
CIN
VIN
SENSE –
L1
C1
BOOST
D1
VOUT
+
LTC1624
M1
TG
L2
CB
GND
SW
R2
+
COUT
VFB
R1
1624 F07
Figure 7. SEPIC Converter
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identical voltages are applied to L1 and L2 throughout the
switching cycle. By making L1 = L2 and wound on the
same core the input ripple is reduced along with cost and
size. All SEPIC applications information that follows
assumes L1 = L2 = L.
SEPIC Converter: Power MOSFET Selection
One external N-channel power MOSFET must be selected
for use with the LTC1624 for the switch. As in boost
applications the source of the power MOSFET is grounded
along with the SW pin. The peak-to-peak gate drive levels
are set by the INTVCC voltage. This voltage is equal to
approximately 5V for VIN > 5.6V and a logic level MOSFET
can be used. At VIN voltages below 5V the INTVCC voltage
is equal to VIN – 0.6V and a sublogic level MOSFET should
be used.
Selection criteria for the power MOSFET include the “ON”
resistance RDS(ON), reverse transfer capacitance CRSS,
input voltage and maximum output current. When the
LTC1624 is operating in continuous mode the duty cycle
for the MOSFET is given by:
Main Switch Duty Cycle =
VOUT + VD
VIN + VOUT + VD
The MOSFET power dissipation and maximum switch
current at maximum output current are given by:
PMAIN =
2
VOUT + VD
I
1 + δ RDS ON +
SW MAX V
+ VOUT + VD
IN
MIN
( )
1.85
k VIN MIN + VOUT
( )
( ) ( )
( )
( )(
)(
)
I
C
200kHz
SW MAX RSS
+V
V
where ISW MAX = IOUT MAX OUT D + 1
VIN MIN
( )
( )
( )
δ is the temperature dependency of RDS(ON) and k is a
constant inversely related to the gate drive current. The
peak switch current is ISW(MAX) + ∆IL.
MOSFETs have I2R losses plus the PMAIN equation
includes an additional term for transition losses that are
16
highest at high total input plus output voltages. For
(VIN + VOUT) < 20V the high current efficiency generally
improves with larger MOSFETs, while for (VIN + VOUT) >
20V the transition losses rapidly increase to the point that
the use of a higher RDS(ON) device with lower CRSS actual
provides higher efficiency. For additional information refer
to the Step-Down Converter: Power MOSFET Selection in
the Applications Information section.
SEPIC Converter: Inductor Selection
For most applications the equal inductor values will fall in
the range of 10µH to 100µH. Higher values reduce the
input ripple voltage and reduce core loss. Lower inductor
values are chosen to reduce physical size and improve
transient response.
Like the boost converter the input current of the SEPIC
converter is calculated at full load current. Peak inductor
current can be significantly higher than output current,
especially with smaller inductors and lighter loads. The
following formula assumes continuous mode operation
and calculates maximum peak inductor current at minimum VIN:
∆I
VOUT
+ L1
IL1 PEAK = IOUT MAX
2
VIN MIN
VIN MIN + VD
∆I
+ L2
IL2 PEAK = IOUT MAX
2
VIN MIN
(
)
( )
(
)
( )
( )
( )
( )
The ripple current in the inductor (∆IL) is typically 20% to
30% of the peak current occuring at VIN(MIN) and IOUT(MAX),
and ∆IL1 = ∆IL2. Maximum ∆IL occurs at maximum VIN.
(VIN)(VOUT + VD)
( ) (200kHz)(L)(VIN + VOUT + VD)
∆IL P-P =
By making L1 = L2 and wound on the same core the value
of inductance in all the above equations are replaced by
2L due to their mutual inductance. Doing this maintains
the same ripple current and inductive energy storage in the
inductors. For example a Coiltronix CTX10-4 is a 10µH
inductor with two windings. With the windings in parallel
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10µH inductance is obtained with a current rating of 4A.
Splitting the two windings creates two 10µH inductors
with a current rating of 2A each. Therefore substitute
(2)(10µH) = 20µH for L in the equations.
Specify the maximum inductor current to safely handle
IL(PEAK). Make sure the inductor’s saturation current rating (current when inductance begins to fall) exceeds the
maximum current rating set by RSENSE.
SEPIC Converter: RSENSE Selection for Maximum
Output Current
RSENSE is chosen based on the required output current.
Remember the LTC1624 current comparator has a maximum threshold of 160mV/RSENSE. The current comparator threshold sets the peak of the inductor current, yielding
a maximum average output current IOUT(MAX) equal to
IL1(PEAK) less half the peak-to-peak ripple current, ∆IL,
divided by the output-input voltage ratio (see equation for
IL1(PEAK)).
Allowing a margin for variations in the LTC1624 (without
considering variation in RSENSE), assuming 30% ripple
current in the inductor, yields:
( )
VIN MIN
RSENSE =
IOUT MAX VOUT + VD
100mV
( )
SEPIC Converter: Output Diode
The output diode conducts current only during the switch
off-time. Peak reverse voltage for SEPIC converters is
equal to VOUT + VIN. Average forward current in normal
operation is equal to output current. Peak current is:
VOUT + VD
ID1 PEAK = I OUT MAX
+ 1 + ∆IL
VIN MIN
(
)
( )
( )
Schottky diodes such as MBR130LT3 are recommended.
SEPIC Converter: Input and Output Capacitors
The output capacitor is normally chosen by its effective
series resistance (ESR), because this is what determines
output ripple voltage. The input capacitor needs to be sized
to handle the ripple current safely.
Since the output capacitor’s ESR affects efficiency, use
low ESR capacitors for best performance. SEPIC regulators, like step-down regulators, have a triangular current
waveform but have maximum ripple at VIN(MAX). The input
capacitor ripple current is:
( )
IRIPPLE RMS =
∆IL
12
The output capacitor ripple current is:
( )
IRIPPLE RMS = IOUT VOUT
VIN
The output capacitor ripple voltage (RMS) is:
VOUT(RIPPLE) = 2(∆IL)(ESR)
The input capacitor can see a very high surge current when
a battery is suddenly connected, and solid tantalum
capacitors can fail under this condition. Be sure to specify
surge tested capacitors.
SEPIC Converter: Coupling Capacitor (C1)
The coupling capacitor C1 in Figure 7 sees a nearly
rectangular current waveform. During the off-time the
current through C1 is IOUT(VOUT/VIN) while approximately
– IOUT flows though C1 during the on-time. This current
waveform creates a triangular ripple voltage on C1:
VOUT
IOUT
∆VC1 =
200kHz C1 VIN + VOUT + VD
(
)( )
The maximum voltage on C1 is then:
VC1(MAX) = VIN + ∆VC1 /2 (typically close to VIN(MAX)).
The ripple current though C1 is:
( )
IRIPPLE C1 = IOUT
VOUT
VIN
The maximum ripple current occurs at IOUT(MAX) and
VIN(MIN). The capacitance of C1 should be large enough so
17
LTC1624
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APPLICATIONS INFORMATION
that the voltage across C1 is constant such that VC1 = VIN
at full load over the entire VIN range. Assuming the enegry
storage in the coupling capacitor C1 must be equal to the
enegry stored in L1, the minimum capacitance of C1 is:
( )
C1 MIN =
( ) (VOUT)
L1 IOUT
2
2
4
( )
VIN MIN
SEPIC Converter: Duty Cycle Limitations
The minimum on-time of 450ns sets a limit on how high
an input-to-output ratio can be tolerated while not skipping cycles. This only impacts designs when very low
output voltages (VOUT < 2.5V) are needed. Note that a
SEPIC converter would not be appropriate at these low
output voltages. The maximum input voltage is (remember not to exceed the absolute maximum limit of 36V):
VIN(MAX) = 10.1VOUT + 5V
For DC > 9%
Positive-to-Negative Converter Applications
The LTC1624 can also be used as a positive-to-negative
converter with a grounded inductor shown in Figure 8.
Since the LTC1624 requires a positive feedback signal
relative to device ground, Pin 4 must be tied to the
regulated negative output. A resistive divider from the
negative output to ground sets the output voltage.
Remember not to exceed maximum VIN ratings VIN +
VOUT ≤ 36V.
1000pF
1
RC
SENSE –
VIN
3
BOOST
ITH /RUN
LTC1624
VFB
TG
4
GND
SW
RSENSE
7
6
5
The external resistive divider is connected to the output as
shown in Figure 8.
Positive-to-Negative Converter: Power
MOSFET Selection
One external N-channel power MOSFET must be selected
for use with the LTC1624 for the switch. As in step-down
applications the source of the power MOSFET is connected to the Schottky diode and inductor. The peak-topeak gate drive levels are set by the INTVCC voltage. The
gate drive voltage is equal to approximately 5V for VIN >
5.6V and a logic level MOSFET can be used. At VIN voltages
below 5V the INTVCC voltage is equal to VIN – 0.6V and a
sublogic level MOSFET should be used.
Selection criteria for the power MOSFET include the “ON”
resistance RDS(ON), reverse transfer capacitance CRSS,
input voltage and maximum output current. When the
LTC1624 is operating in continuous mode the duty cycle
for the MOSFET is given by:
CIN
VOUT + VD
VIN + VOUT + VD
with VOUT being the absolute value of VOUT.
M1
CB
L1
D1
R1
The MOSFET power dissipation and maximum switch
current are given by:
+
COUT
R2
–VOUT
1624 F08
Figure 8. Positive-to-Negative Converter
18
R1
DC
VOUT = 1.19V 1 + ≈ − VIN
R2
1 − DC
Main Switch Duty Cycle =
+
100pF
Setting the output voltage for a positive-to-negative converter is different from other architectures since the feedback voltage is referenced to the LTC1624 ground pin and
the ground pin is referenced to – VOUT. The output voltage
is set by a resistive divider according to the following
formula:
VIN
8
CC
2
Positive-to-Negative Converter: Output Voltage
Programming
PMAIN = ISW(MAX) ×
{I
( )(
( )
)
( )
OUT MAX I + δ RDS ON +
k V IN MAX + VOUT 1.85 CRSS 200kHz
(
) (
)(
){
LTC1624
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APPLICATIONS INFORMATION
V + V
IN
OUT + VD
where: ISW MAX = IOUT MAX
V
IN
( )
( )
δ is the temperature dependency of RDS(ON) and k is a
constant inversely related to the gate drive current. The
maximum switch current occurs at VIN(MIN) and the peak
switch current is ISW(MAX) + ∆IL /2. The maximum voltage
across the switch is VIN(MAX) + VOUT.
MOSFETs have I2R losses plus the PMAIN equation
includes an additional term for transition losses that are
highest at high total input plus output voltages. For
(VOUT+ VIN) < 20V the high current efficiency generally
improves with larger MOSFETs, while for (VOUT+ VIN)
> 20V the transition losses rapidly increase to the point
that the use of a higher RDS(ON) device with lower CRSS
actual provides higher efficiency. For additional information refer to the Step-Down Converter: Power MOSFET
Selection in the Applications Information section.
Positive-to-Negative Converter: Inductor Selection
For most applications the inductor will fall in the range of
10µH to 100µH. Higher values reduce the input and output
ripple voltage (although not as much as step-down converters) and also reduce core loss. Lower inductor values
are chosen to reduce physical size and improve transient
response but do increase output ripple.
Like the boost converter, the input current of the positiveto-negative converter is calculated at full load current.
Peak inductor current can be significantly higher than
output current, especially with smaller inductors (with
high ∆IL values). The following formula assumes continuous mode operation and calculates maximum peak inductor current at minimum VIN:
V + V
∆I
IN
OUT + VD
IL PEAK = IOUT MAX
+ L
VIN
2
(
)
( )
The ripple current in the inductor (∆IL) is typically 20% to
50% of the peak inductor current occuring at VIN(MIN) and
IOUT(MAX) to minimize output ripple. Maximum ∆IL occurs
at minimum VIN.
(VIN)( VOUT + VD)
( ) (200kHz)(L) V + V + V
( IN OUT D)
∆IL P-P =
Specify the maximum inductor current to safely handle
IL(PEAK). Make sure the inductor’s saturation current rating (current when inductance begins to fall) exceeds the
maximum current rating set by RSENSE.
Positive-to-Negative Converter: RSENSE Selection for
Maximum Output Current
RSENSE is chosen based on the required output current.
Remember the LTC1624 current comparator has a maximum threshold of 160mV/RSENSE. The current comparator threshold sets the peak of the inductor current, yielding
a maximum average output current IOUT(MAX) equal to
IL(PEAK) less half the peak-to-peak ripple current with the
remainder divided by the duty cycle.
Allowing a margin for variations in the LTC1624 (without
considering variation in RSENSE) and assuming 30% ripple
current in the inductor, yields:
VIN MIN
100mV
RSENSE =
IOUT MAX VIN MIN + VOUT + VD
( )
( )
( )
Positive-to-Negative Converter: Output Diode
The output diode conducts current only during the switch
off-time. Peak reverse voltage for positive-to-negative
converters is equal to VOUT+ VIN. Average forward
current in normal operation is equal to ID(PEAK) – ∆IL /2.
Peak diode current (occurring at VIN(MIN)) is:
(
)
V
OUT + VD
∆I
ID PEAK = IOUT MAX
+ 1 + L
VIN
2
(
)
( )
Positive-to-Negative Converter: Input and
Output Capacitors
The output capacitor is normally chosen by its effective
series resistance (ESR), because this is what determines
output ripple voltage. Both input and output capacitors
need to be sized to handle the ripple current safely.
19
LTC1624
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APPLICATIONS INFORMATION
Positive-to-negative converters have high ripple current in
both the input and output capacitors. For long capacitor
lifetime, the RMS value of this current must be less than
the high frequency ripple rating of the capacitor.
The following formula gives an approximate value for RMS
ripple current. This formula assumes continuous mode
and low current ripple. Small inductors will give somewhat
higher ripple current, especially in discontinuous mode.
For the exact formulas refer to Application Note 44, pages
28 to 30. The input and output capacitor ripple current
(occurring at VIN(MIN)) is:
( )( )
Capacitor IRMS = ff IOUT
VOUT
VIN
ff = Fudge factor (1.2 to 2.0)
The output peak-to-peak ripple voltage is:
VOUT(P-P) = RESR (ID(MAX))
ITH /RUN pin below 0.8V relative to the LTC1624 ground
pin. With the LTC1624 ground pin referenced to – VOUT,
the nonimal range on the ITH /RUN pin is – VOUT (in
shutdown) to (– VOUT + 2.4V)(at Max IOUT). Referring to
Figure 15, M2, M3 and R3 provide a level shift from typical
TTL levels to the LTC1624 operating as positive-to-negative converter. MOSFET M3 supplies gate drive to M2
during shutdown, while M2 pulls the ITH/RUN pin voltage to
– VOUT, shutting down the LTC1624.
Step-Down Converters: PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1624. These items are also illustrated graphically in
the layout diagram of Figure 9. Check the following in your
layout:
1. Are the signal and power grounds segregated? The
LTC1624 ground (Pin 4) must return to the (–) plate
of COUT.
The input capacitor can also see a very high surge current
when a battery is suddenly connected, and solid tantalum
capacitors can fail under this condition. Be sure to specify
surge tested capacitors.
2. Does the VFB (Pin 3) connect directly to the feedback
resistors? The resistive divider R1, R2 must be connected between the (+) plate of COUT and signal ground.
The 100pF capacitor should be as close as possible to
the LTC1624.
Positive-to-Negative Converter: Duty Cycle
Limitations
The minimum on-time of 450ns sets a limit on how high
of input-to-output ratio can be tolerated while not skipping
cycles. This only impacts designs when very low output
voltages (VOUT< 2.5V) are needed. The maximum input
voltage is:
3. Does the VIN lead connect to the input voltage at the
same point as RSENSE and are the SENSE – and VIN leads
routed together with minimum PC trace spacing? The
filter capacitor between VIN and SENSE – should be as
close as possible to the LTC1624.
VIN(MAX) < 10.1VOUT + 5V
For DC > 9%
VIN(MAX) < 36V –VOUTFor absolute maximum ratings
Positive-to-Negative Converter: Shutdown
Considerations
Since the ground pin on the LTC1624 is referenced to
– VOUT, additional circuitry is needed to put the LTC1624
into shutdown. Shutdown is enabled by pulling the
20
4. Does the (+) plate of CIN connect to RSENSE as closely
as possible? This capacitor provides the AC current to
the MOSFET(s). Also, does CIN connect as close as
possible to the VIN and ground pin of the LTC1624?
This capacitor also supplies the energy required to
recharge the bootstrap capacitor. Adequate input
decoupling is critical for proper operation.
5. Keep the switch node SW away from sensitive smallsignal nodes. Ideally, M1, L1 and D1 should be connected as closely as possible at the switch node.
LTC1624
U
TYPICAL APPLICATIONS
1000pF
+
1
RC
SENSE –
VIN
8
+
CC
2
3
BOOST
ITH /RUN
LTC1624
TG
VFB
–
6
M1
SW
GND
L1
CB
0.1µF
100pF
4
VIN
CIN
RSENSE
7
5
+
R2
D1
+
COUT
VOUT
R1
–
BOLD LINES INDICATE
HIGH CURRENT PATHS
1624 F09
Figure 9. LTC1624 Layout Diagram (See Board Layout Checklist)
VIN
5.3V TO 28V
1
2
CC
560pF
RC
4.7k
SENSE –
VIN
ITH /RUN
BOOST
LTC1624
3
VFB
TG
GND
SW
8
7
5
RSENSE
0.033Ω
0.1µF
6
100pF
4
1000pF
CB
0.1µF
D1
MBRS340T3
M1
Si4412DY
+
CIN
22µF
35V
×2
D2
CMDSH-3
L1*
10µH
*COILTRONICS CTX10-4
R2
35.7k
1%
R1
11k
1%
VOUT
5V
3A
+
COUT
100µF
10V
×2
1624 F10
Figure 10. 5V/3A Converter with Output Derived Boost Voltage
21
LTC1624
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TYPICAL APPLICATIONS
1
SENSE –
2
CC
470pF
VIN
BOOST
ITH /RUN
LTC1624
3
TG
VFB
RC
6.8k
1000pF
8
6
SW
GND
M1
Si6436DY
CB
0.1µF
100pF
4
RSENSE
0.068Ω
0.1µF
7
5
VIN
4.8V TO 22V
CIN
22µF
35V
×2
+
L1*
10µH
D1
MBRS340T3
R2
35.7k
1%
VOUT
1.8V
1.5A
+
R1
69.8k
1%
*SUMIDA CDR105B-100
1624 F11
Figure 11. Wide Input Range 1.8V/1.5A Converter
1
2
CC
470pF
SENSE –
BOOST
ITH /RUN
LTC1624
3
RC
6.8k
VIN
TG
VFB
1000pF
8
7
4
SW
GND
VIN
12.3V TO 28V
6
5
+
RSENSE
0.068Ω
0.1µF
100pF
M1
Si4412DY
CB
0.1µF
CIN
22µF
35V
×2
L1*
47µH
D1
MBRS140T3
R2
35.7k
1%
VOUT
12V
1A
+
R1
3.92k
1%
*SUMIDA CDRH125-470
Figure 12. 12V/1A Low Dropout Converter
1
2
CC
330pF
RC
3.3k
SENSE –
ITH /RUN
VIN
BOOST
LTC1624
3
VFB
TG
GND
SW
8
7
0.1µF
1624 F12
VIN
5.2V TO 11V
+
RSENSE
0.04Ω
L1*
22µH
5
CB
0.1µF
CIN
22µF
D1
35V
× 2 MBRS130LT3
VOUT
12V
0.75A
M1
Si4412DY
R2
35.7k
1%
R1
3.92k
1%
*SUMIDA CDRH125-220
Figure 13. 12V/0.75A Boost Converter
22
COUT
100µF
16V
×2
6
100pF
4
1000pF
COUT
100µF
10V
×2
+
COUT
100µF
16V
×2
1624 F13
LTC1624
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TYPICAL APPLICATIONS
VIN
5V TO 15V
1
2
CC
330pF
RC
4.7k
SENSE –
VIN
ITH /RUN
BOOST
LTC1624
3
TG
VFB
1000pF
8
7
SW
GND
RSENSE
0.068Ω
0.1µF
CIN
22µF
22µF
35V
35V
L1a*
D1
MBRS130LT3
+
6
M1
Si4412DY
CB
0.1µF
100pF
4
+
5
VOUT
12V
0.5A
L1b*
R2
35.7k
1%
COUT
100µF
16V
×2
+
R1
3.92k
1%
*COILTRONICS CTX20-4
Figure 14. 12V/0.4A SEPIC Converter
1624 F14
VIN
5V TO 22V
VCC
VCC
SHUTDOWN
1
CC
RC 1000pF
3.3k
2
SENSE –
BOOST
ITH /RUN
LTC1624
3
M3
TP0610L
VIN
VFB
TG
GND
SW
8
4
6
5
+
RSENSE
0.025Ω
0.1µF
7
100pF
M2
VN2222
1000pF
M1
Si4410DY L1*
33µH
CB
0.1µF
R2
78.7k
1%
D1
MBRS340T3
R3
100k
R1
24.9k
1%
*COILCRAFT DO5022P-333
CIN
22µF
35V
×2
D2
CMDSH-3
+
COUT
100µF
10V
×2
VOUT
–5V
2A
1624 F15
Figure 15. Inverting – 5V/2A Converter
1
2
CC
470pF
RC
6.8k
SENSE –
ITH /RUN
VIN
BOOST
LTC1624
3
VFB
TG
GND
SW
8
7
0.1µF
6
100pF
4
1000pF
5
CB
0.1µF
VIN
3.5V TO 18V
RSENSE
0.068Ω
+
CIN
22µF
35V
×2
M1
Si6426DQ L1*
20µH
D1
MBRS340T3
*COILTRONICS CTX20-4
Figure 16. Low Dropout 3.3V/1.5A Converter
R2
35.7k
1%
R1
20k
1%
VOUT
3.3V
1.5A
+
COUT
100µF
10V
×2
1624 F16
23
LTC1624
U
TYPICAL APPLICATIONS
VIN
3.6V TO 18V
1
2
CC
330pF
RC
6.8k
SENSE –
VIN
BOOST
ITH /RUN
LTC1624
3
VFB
TG
GND
SW
8
7
4
D2
CMDSH-3
5
RSENSE
0.05Ω
0.1µF
L1a*
VOUT
6
100pF
+
1000pF
CIN
22µF
35V
× 2 22µF
35V
+
M1
Si6426DQ
CB
0.1µF
D1
MBRS130LT3
VOUT
5V
1A
L1b*
R2
35.7k
1%
+
R1
11k
1%
* COILTRONICS CTX20-4
1624 F17
Figure 17. 5V/1A SEPIC Converter with Output Derived Boost Voltage
VIN
13V TO
28V
+
CIN1, CIN2
1000µF
35V
×2
COUT
100µF
16V
×2
RSENSE1, 0.015Ω
RSENSE2, 0.015Ω
C4, 0.1µF
LTC1624
CC
100pF
1
2
SENSE –
VIN
ITH /RUN
BOOST
3
RC
20k
R1
11k
1%
4
VFB
TG
GND
SW
8
C5
3.3µF
50V
C7
3.3µF
50V
7
6
5
CB
0.1µF
M1*
VOUT
12V
10A
L1
D1*
+
COUT
2700µF
16V
R5
220Ω
D2
MBR0540
Z1
IN 755
C9
0.1µF
R2
100k
1%
C10
220pF
1624 F18
CIN1, CIN2 = SANYO 35MV1000GX
C5, C7 = WIMA MKS2
COUT = SANYO 16MV2700GX
D1 = MOTOROLA MBR2535CT
L1 = PULSE ENGINEERING PO472
M1 = INTERNATIONAL RECTIFIER IRL3803
RSENSE1, RSENSE2 = IRC LR2010-01-R015-F
* BOTH D1 AND M1 MOUNTED TO SAME
THERMALLOY #6399B HEAT SINK
Figure 18. 24V to 12V/10A Buck Converter with Output-Derived Boost Voltage
24
LTC1624
U
TYPICAL APPLICATIONS
VIN
20V TO
32V
+
RSENSE
0.025Ω
CIN
22µF
35V
L1
47µH
C5
0.1µF
D1
VOUT
90V
0.5A
LTC1624
1
CC
820pF
2
SENSE –
VIN
ITH /RUN
BOOST
3
C3
100pF
R1
13.3k
RC
6.8k
4
VFB
TG
GND
SW
8
7
6
+
CB
0.1µF
M1
COUT
100µF
100V
5
R2, 1M, 1%
1624 F19
L1 = COILCRAFT D05022P-473
M1 = INTERNATIONAL RECTIFIER IRL 540NS
RSENSE = IRC LR2010-01-R025-F
CIN = KEMET T495X226M035AS
COUT = SANYO 100MV100GX
D1 = MOTOROLA MBRS1100
Figure 19. 24V to 90V at 0.5A Boost Converter
VIN
9V TO
15V
+
RSENSE
0.005Ω, 5%
CIN
100µF
16V
L1
10µH
C5
0.1µF
D1*
VOUT
24V
5A
LTC1624
1
CC
4700pF
2
SENSE –
VIN
ITH /RUN
BOOST
3
RC
27k
R1
52.3k
C3
100pF
C4
1500pF
4
VFB
TG
GND
SW
R5
750Ω
0.5W
8
7
6
CB
0.1µF
M1*
+
5
COUT1
1000µF
35V
+
COUT2
1000µF
35V
Z1
IN755
7.5V
R2, 1M, 1%
CIN = KEMET T495X107M016AS
COUT1, COUT2 = SANYO 35MV 1000GX
D1 = MOTOROLA MBR2535CT
L1 = MAGNETICS CORE #55930AZ WINDING = 8T#14BIF
M1 = INTERNATIONAL RECTIFIER IRL 3803
RSENSE = IRC OAR-3, 0.005Ω, 5%
*BOTH D1 AND Q1 MOUNTED ON
THERMALLOY MODEL 6399 HEAT SINK
1624 F20
Figure 20. 12V to 24V/5A Boost Converter
25
LTC1624
U
TYPICAL APPLICATIONS
VIN
13V TO
28V
+
+VIN
CIN1, CIN2
22µF
35V
×2
RSENSE
0.033Ω
C5, 0.1µF
LTC1624
1
CC
330pF
2
SENSE –
VIN
ITH /RUN
BOOST
3
VFB
RC
10k
R1
3.92k
C4
100pF
4
GND
TG
SW
8
7
6
CB
0.1µF
M1
L1
27µH
5
D1
MBRS340
Q2
C10
0.1µF
1
8
2
3
4
SENSE
AVE
IOUT
PROG
LTC1620
GND
VCC
–IN
+IN
C11
0.1µF
+
R6
10k
6
R7
56k
5
CURRENT
ADJ
COUT
100µF
16V
×2
1
C12
1µF
7
C14, 0.01µF
R4
0.025Ω
2
OUT
R2
35.7k
C9
100pF
IN
8
VOUT
12V
3A
+VIN
7
3
NC/ADJ
NC
LT1121-5
6
GND
NC
4
5
NC
SHDN
C13
0.1µF
R8
1M
1624 F21
CIN1, CIN2 = KEMET T495X226M035AS
L1 = SUMIDA CDRH127-270
RSENSE = IRC LR2010-01-R033-F
R4 = IRC LR2010-01-R025-F
M1 = SILICONIX Si4412DY
Q2 = MOTOROLA MMBT A14
Figure 21. 12V/3A Adjustable Current Power Supply for Battery Charger or Current Source Applications
26
LTC1624
U
TYPICAL APPLICATIONS
1
2
CC
680pF
SENSE –
VIN
BOOST
ITH /RUN
LTC1624
3
TG
VFB
RC
3.3k
8
7
SW
GND
1000pF
5
CIN
22µF
35V
×3
+
RSENSE
0.015Ω
0.1µF
6
100pF
4
VIN
4.8V TO 28V
M1**
L1*
8µH
CB
0.1µF
D1
MBRD835L
VOUT
3.3V
6.5A
R2
35.7k
1%
COUT
100µF
10V
×3
+
R1
20k
1%
* PANASONIC 12TS-7ROLB
** SILICONIX SUD50N03-10
1624 F22
Figure 22. High Current 3.3V/6.5A Converter
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 – 0.197*
(4.801 – 5.004)
8
7
6
5
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
0.053 – 0.069
(1.346 – 1.752)
0°– 8° TYP
0.016 – 0.050
0.406 – 1.270
0.014 – 0.019
(0.355 – 0.483)
2
3
4
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
TYP
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
SO8 0996
27
LTC1624
U
TYPICAL APPLICATION
R3, 10Ω
RSENSE
0.0082Ω
C3, 0.033µF
+
C4, 0.1µF
+VIN
4.5V TO
5.5V
CIN
100µF
10V
×4
LTC1624
1
CC
820pF
2
SENSE –
VIN
ITH /RUN
BOOST
3
TG
VFB
RC
6.8k
R1
20k
C2
100pF
4
SW
GND
8
7
6
D2
CB
0.1µF
M1
5
L1
1.68µH
D1
COUT
470µF
6.3V
×2
+VOUT
3.3V
10A
R2
35.7k
+
C8
100pF
VOUT
RTN
Q2
10k
1624 F23
CIN (× 4) = KEMET T495D107M010AS
COUT (× 2) = AVX TPSV477M006R0055
D1 = MOTOROLA MBRB2515L
D2 = MOTOROLA MBR0520
L1 = PULSE ENGINEERING PE53691
M1 = INTERNATIONAL RECTIFIER IRL3803S
Q2 = MOTOROLA MMBTA14LT1
RSENSE = IRC OAR3-R0082
Figure 23. 5V to 3.3V/10A Converter (Surface Mount)
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Adaptive Power is a trademark of Linear Technology Corporation.
28
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417● (408)432-1900
FAX: (408) 434-0507● TELEX: 499-3977 ● www.linear-tech.com
1624f LT/TP 0198 4K • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 1997