Cherry CS5150H Cpu 4-bit synchronous buck controller Datasheet

CS5150H
CS5150H
CPU 4-Bit Synchronous Buck Controller
Features
Description
The CS5150H is a 4-bit synchronous
dual N-Channel buck controller. It
is designed to provide unprecedented transient response for
today’s demanding high-density,
high-speed logic. The regulator
operates using a proprietary control
method, which allows a 100ns
response time to load transients.
The CS5150H is designed to operate
over a 4.25-20V range (VCC) using
12V to power the IC and 5V or 12V
as the main supply for conversion.
The CS5150H is specifically
designed to power Pentium® Pro
processors and other high performance core logic. It includes the following features: on board, 4-bit
DAC, short circuit protection, 1.0%
output tolerance, VCC monitor, and
programmable soft start capability.
The CS5150H is upward compatible
with the 5-bit CS5155H, allowing
the mother board designer the
capability of using either the
CS5150H or the CS5155H with no
change in layout. The CS5150H is
available in 16 pin surface mount.
■ Dual N-Channel Design
■ Excess of 1MHz Operation
■ 100ns Transient Response
■ 4-Bit DAC
■ Upward Compatible with
5-Bit CS5155H/5156H
and Adjustable
CS5120/5121
■ 30ns Gate Rise/Fall Times
■ 1% DAC Accuracy
■ 5V & 12V Operation
■ Remote Sense
■ Programmable Soft Start
■ Lossless Short Circuit
Protection
Application Diagram
■ VCC Monitor
Switching Power Supply for core logic - Pentium® Pro processor
12V
VCC1 VCC2
VID0
VID1
VID2
VID1
VID2
VID3
VID3
1200µF/16V x 3
AlEl
2µH
2.1V to 3.5V @ 13A
■ Overvoltage Protection
Package Options
IRL3103
VGATE(L)
1200µF/16V x 5
AlEl
COFF
PGnd
SS
LGnd
16 Lead SO Narrow
VID0
VID1
VFB
1
COMP
LGnd
VID2
VFB
COMP
0.33µF
■ Current Sharing
IRL3103
VGATE(H)
CS5150H
330pF
0.1µF
■ Adaptive Voltage
Positioning
■ V2™ Control Topology
5V
0.1µF
VID0
■ 25ns FET Nonoverlap Time
VID3
SS
3.3k
VFFB
100pF
VCC1
VGATE(L)
NC
COFF
PGnd
VGATE(H)
VFFB
VCC2
V2 is a trademark of Switch Power, Inc.
Pentium is a registered trademark of Intel Corporation.
Cherry Semiconductor Corporation
2000 South County Trail, East Greenwich, RI 02818
Tel: (401)885-3600 Fax: (401)885-5786
Email: [email protected]
Web Site: www.cherry-semi.com
Rev. 1/21/99
1
A
®
Company
CS5150H
Absolute Maximum Ratings
Pin Name
Max Operating Voltage
Max Current
VCC1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .16V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .25mA DC/1.5A peak
VCC2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .20V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .20mA DC/1.5A peak
SS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-100µA
COMP . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .200µA
VFB . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-0.2µA
COFF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-0.2µA
VFFB . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-0.2µA
VID0 - VID3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-50µA
VGATE(H) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .20V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .100mA DC/1.5A peak
VGATE(L) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .16V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .100mA DC/1.5A peak
LGnd . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .0V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .25mA
PGnd . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .0V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .100mA DC/1.5A peak
Operating Junction Temperature, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .°0 to 150°C
Lead Temperature Soldering
Reflow (SMD styles only) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .60 sec. max above 183°C, 230°C peak
Storage Temperature Range, TS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-65° to 150°C
ESD Susceptibility . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2kV
Electrical Characteristics: 0°C < TA < +70°C; 0°C < TJ < +125°C; 8V < VCC1 < 14V; 5V < VCC2 < 20V; DAC Code: VID2 = VID1 =
VID0 = 1; VID3 = 0; CVGATE(L) and CVGATE(H) = 1nF; COFF = 330pF; CSS = 0.1µF, unless otherwise specified.
PARAMETER
TEST CONDITIONS
■ Error Amplifier
VFB Bias Current
Open Loop Gain
Unity Gain Bandwidth
COMP SINK Current
COMP SOURCE Current
COMP CLAMP Current
COMP High Voltage
COMP Low Voltage
PSRR
VFB = 0V
1.25V < VCOMP < 4V; Note 1
Note 1
VCOMP = 1.5V; VFB = 3V; VSS > 2V
VCOMP = 1.2V; VFB = 2.7V; VSS = 5V
VCOMP = 0V; VFB = 2.7V
VFB = 2.7V; VSS = 5V
VFB =3V
8V < VCC1 < 14V @ 1kHz; Note 1
■ VCC1 Monitor
Start Threshold
Stop Threshold
Hysteresis
Output switching
Output not switching
Start-Stop
■ DAC
Input Threshold
VID0, VID1, VID2, VID3
Input Pull Up Resistance
VID0, VID1, VID2, VID3
Pull Up Voltage
Accuracy (all codes except 1111) Measure VFB = VCOMP, 25°C ≤ TJ ≤ 125°C
VID3
VID2
VID1
VID0
1
1
1
1
1
1
1
0
1
1
0
1
1
1
0
0
1
0
1
1
1
0
1
0
1
0
0
1
1
0
0
0
0
1
1
1
0
1
1
0
2
MIN
50
500
0.4
30
0.4
4.0
60
TYP
0.3
60
3000
2.5
50
1.0
4.3
160
85
MAX
1.0
8.0
80
1.6
5.0
600
UNIT
µA
dB
kH
mA
µA
mA
V
mV
dB
3.75
3.70
3.90
3.85
50
4.05
4.00
V
V
mV
1.00
25
4.85
1.25
50
5.00
2.40
100
5.15
1.0
V
kΩ
V
%
1.2191
2.1186
2.2176
2.3166
2.4156
2.5146
2.6136
2.7126
2.8116
2.9106
1.2440
2.1400
2.2400
2.3400
2.4400
2.5400
2.6400
2.7400
2.8400
2.9400
1.2689
2.1614
2.2624
2.3634
2.4644
2.5654
2.6664
2.7674
2.8684
2.9694
V
V
V
V
V
V
V
V
V
V
VID0 = 1; VID3 = 0; CVGATE(L) and CVGATE(H) = 1nF; COFF = 330pF; CSS = 0.1µF, unless otherwise specified.
PARAMETER
MIN
TYP
MAX
UNIT
3.0096
3.1086
3.2076
3.3066
3.4056
3.5046
3.0400
3.1400
3.2400
3.3400
3.4400
3.5400
3.0704
3.1714
3.2724
3.3734
3.4744
3.5754
V
V
V
V
V
V
1.2
1.0
2.0
1.5
V
V
30
50
ns
30
50
ns
25
50
50
mA
ns
25
50
ns
50
600
100
800
kΩ
mV
3.3
100
3.3
0.95
1.0
2.5
5.0
200
6.0
1.10
1.1
3.0
ms
ms
%
V
V
V
VFFB = 0 to 5V to VGATE(H) = 9V to 1V;
VCC1 = VCC2 = 12V
VFFB = 0V
100
125
ns
■ Supply Current
ICC1
ICC2
Operating ICC1
Operating ICC2
No Switching
No Switching
VFB = COMP = VFFB
VFB = COMP = VFFB
8.5
1.6
8
2
13.5
3.0
13
5
mA
mA
mA
mA
■ COFF
Normal Charge Time
Extension Charge Time
Discharge Current
VFFB = 1.5V; VSS = 5V
VSS = VFFB = 0
COFF to 5V; VFB >1V
1.6
8.0
2.2
11.0
µs
µs
mA
■
DAC: continued
VID3 VID2
VID1
0
1
0
0
1
0
0
0
1
0
0
1
0
0
0
0
0
0
TEST CONDITIONS
VID0
1
0
1
0
1
0
■ VGATE(H) and VGATE(L)
Out SOURCE Sat at 100mA
Out SINK Sat at 100mA
Out Rise Time
Out Fall Time
Shoot-Through Current
Delay VGATE(H) to VGATE(L)
Delay VGATE(L) to VGATE(H)
VGATE(H), VGATE(L) Resistance
VGATE(H), VGATE(L) Schottky
■ Soft Start (SS)
Charge Time
Pulse Period
Duty Cycle
COMP Clamp Voltage
VFFB SS Fault Disable
High Threshold
■ PWM Comparator
Transient Response
VFFB Bias Current
Measure VCC1 – VGATE(L),;VCC2 – VGATE(H)
Measure VGATE(H) – VPGnd;
VGATE(L) – VPGnd
1V < VGATE(H) < 9V; 1V < VGATE(L) < 9V
VCC1 = VCC2 = 12V
9V > VGATE(H) > 1V; 9V > VGATE(L) > 1V
VCC1 = VCC2 = 12V
Note 1
VGATE(H) falling to 2V; VCC1 = VCC2 = 8V
VGATE(L) rising to 2V
VGATE(L) falling to 2V; VCC1 = VCC2 = 8V
VGATE(H) rising to 2V
Resistor to LGnd (Note 1)
20
LGnd to VGATE(H) @ 10mA
LGnd to VGATE(L) @ 10mA
(Charge Time/Pulse Period) × 100
VFB = 0V; VSS = 0
VGATE(H) = Low; VGATE(L) = Low
1.6
25
1.0
0.50
0.9
0.3
1.0
5.0
5.0
3
µA
CS5150H
Electrical Characteristics: 0°C < TA < +70°C; 0°C < TJ < +125°C; 8V < VCC1 < 14V; 5V < VCC2 < 20V; DAC Code: VID2 = VID1 =
CS5150H
Electrical Characteristics: 0°C < TA < +70°C; 0°C < TJ < +125°C; 8V < VCC1 < 14V; 5V < VCC2 < 20V; DAC Code: VID2 = VID1 =
VID0 = 1; VID3 = 0; CVGATE(L) and CVGATE(H) = 1nF; COFF = 330pF; CSS = 0.1µF, unless otherwise specified.
PARAMETER
■ Time Out Timer
Time Out Time
Fault Mode Duty Cycle
TEST CONDITIONS
VFB = VCOMP; VFFB = 2V;
Record VGATE(H) Pulse High Duration
VFFB = 0V
MIN
TYP
MAX
UNIT
10
30
65
µs
35
50
70
%
Note 1: Guaranteed by design, not 100% tested in production.
Package Pin Description
PACKAGE PIN #
PIN SYMBOL
FUNCTION
16L SO Narrow
1,2,3,4
VID0 – VID3
Voltage ID DAC input pins. These pins are internally pulled up to 5V
providing logic ones if left open. The DAC range is 2.14V to 3.54V with
100mV increments. VID0 - VID3 select the desired DAC output voltage.
Leaving all 4 DAC input pins open results in a DAC output voltage of
1.244V, allowing for adjustable output voltage, using a traditional resistor divider.
5
SS
Soft Start Pin. A capacitor from this pin to LGnd in conjunction with
internal 60µA current source provides soft start function for the controller. This pin disables fault detect function during Soft Start. When a
fault is detected, the soft start capacitor is slowly discharged by internal
2µA current source setting the time out before trying to restart the IC.
Charge/discharge current ratio of 30 sets the duty cycle for the IC when
the regulator output is shorted.
6
NC
No connection.
7
COFF
A capacitor from this pin to ground sets the time duration for the on
board one shot, which is used for the constant off time architecture.
8
VFFB
Fast feedback connection to the PWM comparator. This pin is connected
to the regulator output. The inner feedback loop terminates on time.
9
VCC2
Boosted power for the high side gate driver.
10
VGATE(H)
11
PGnd
12
VGATE(L)
13
VCC1
Input power for the IC and low side gate driver.
14
LGnd
Signal ground for the IC. All control circuits are referenced to this pin.
15
COMP
Error amplifier compensation pin. A capacitor to ground should be provided externally to compensate the amplifier.
16
VFB
Error amplifier DC feedback input. This is the master voltage feedback
which sets the output voltage. This pin can be connected directly to the
output or a remote sense trace.
High FET driver pin capable of 1.5A peak switching current. Internal circuit prevents VGATE(H) and VGATE(L) from being in high state simultaneously.
High current ground for the IC. The MOSFET drivers are referenced to
this pin. Input capacitor ground and the source of lower FET should be
tied to this pin.
Low FET driver pin capable of 1.5A peak switching current.
4
VCC2
VCC1
-
VCC1 Monitor
Comparator
5V
+
-
3.90V
3.85V
0.7V
+
2µA
Q
S
Q
PGnd
FAULT
FAULT
Latch
SS High
Comparator
VCC1
-
VID0
VID2
R
FAULT
+
60µA
SS
VID1
VGATE(H)
SS Low
Comparator
VGATE(L)
4 BIT
DAC
+
VID3
2.5V
Error
Amplifier
PGnd
-
PWM
Comparator
-
VFB
COMP
VFFB
R
Maximum
On-Time
Timeout
+
Slow Feedback
S
Extended
Off-Time
Timeout
-
Q
PWM
Latch
Normal
Off-Time
Timeout
Fast Feedback
Q
Off-Time
Timeout
GATE(H) = ON
GATE(H) = OFF
COFF
One Shot
S
+
LGnd
COFF
R
Q
VFFB Low
Comparator
1V
PWM
COMP
Time Out
Timer
(30µs)
Edge Triggered
Applications Information
The V2™ control method is illustrated in Figure 1. The output voltage is used to generate both the error signal and the
ramp signal. Since the ramp signal is simply the output
voltage, it is affected by any change in the output regardless of the origin of that change. The ramp signal also contains the DC portion of the output voltage, which allows
the control circuit to drive the main switch to 0% or 100%
duty cycle as required.
A change in line voltage changes the current ramp in the
inductor, affecting the ramp signal, which causes the V2™
control scheme to compensate the duty cycle. Since the
change in inductor current modifies the ramp signal, as in
current mode control, the V2™ control scheme has the
same advantages in line transient response.
A change in load current will have an affect on the output
voltage, altering the ramp signal. A load step immediately
changes the state of the comparator output, which controls
the main switch. Load transient response is determined
only by the comparator response time and the transition
speed of the main switch. The reaction time to an output
load step has no relation to the crossover frequency of the
error signal loop, as in traditional control methods.
The error signal loop can have a low crossover frequency,
since transient response is handled by the ramp signal loop.
The main purpose of this ‘slow’ feedback loop is to provide
DC accuracy. Noise immunity is significantly improved,
since the error amplifier bandwidth can be rolled off at a low
frequency. Enhanced noise immunity improves remote sens-
Theory of Operation
V2™ Control Method
The V2™ method of control uses a ramp signal that is generated by the ESR of the output capacitors. This ramp is
proportional to the AC current through the main inductor
and is offset by the value of the DC output voltage. This
control scheme inherently compensates for variation in
either line or load conditions, since the ramp signal is generated from the output voltage itself. This control scheme
differs from traditional techniques such as voltage mode,
which generates an artificial ramp, and current mode,
which generates a ramp from inductor current.
PWM
Comparator
+
VGATE(H)
C
–
Ramp
Signal
VGATE(L)
VFFB
VFB
Error
Amplifier
COMP
Error
Signal
Output
Voltage
Feedback
–
E
+
Reference
Voltage
Figure 1: V2™ Control Diagram
5
CS5150H
Block Diagram
CS5150H
Applications Information: continued
ing of the output voltage, since the noise associated with
long feedback traces can be effectively filtered.
Line and load regulation are drastically improved because
there are two independent voltage loops. A voltage mode
controller relies on a change in the error signal to compensate for a deviation in either line or load voltage. This
change in the error signal causes the output voltage to
change corresponding to the gain of the error amplifier,
which is normally specified as line and load regulation. A
current mode controller maintains fixed error signal under
deviation in the line voltage, since the slope of the ramp
signal changes, but still relies on a change in the error signal for a deviation in load. The V2™ method of control
maintains a fixed error signal for both line and load variation, since the ramp signal is affected by both line and load.
set by the time out timer and is approximately equal to the
maximum on time, resulting in a 50% duty cycle. The
GateL pin will then drive low, the GateH pin will drive
high, and the cycle repeats.
When regulator output voltage achieves the 1V level present at the COMP pin, regulation has been achieved and
normal off time will ensue. The PWM comparator terminates the switch on time, with off time set by the COFF
capacitor. The V2™ control loop will adjust switch duty
cycle as required to ensure the regulator output voltage
tracks the output of the error amplifier.
The soft start and COMP capacitors will charge to their
final levels, providing a controlled turn on of the regulator
output. Regulator turn on time is determined by the COMP
capacitor charging to its final value. Its voltage is limited by
the soft start COMP clamp and the voltage on the soft start
pin (see Figures 2 and 3).
Constant Off Time
To maximize transient response, the CS5150H uses a constant off time method to control the rate of output pulses.
During normal operation, the off time of the high side
switch is terminated after a fixed period, set by the COFF
capacitor. To maintain regulation, the V2™ control loop
varies switch on time. The PWM comparator monitors the
output voltage ramp, and terminates the switch on time.
Constant off time provides a number of advantages. Switch
duty cycle can be adjusted from 0 to 100% on a pulse by
pulse basis when responding to transient conditions. Both
0% and 100% duty cycle operation can be maintained for
extended periods of time in response to load or line transients. PWM slope compensation to avoid sub-harmonic
oscillations at high duty cycles is avoided.
Switch on time is limited by an internal 30µs timer, minimizing stress to the power components.
Trace 1 - Regulator Output Voltage (1V/div.)
Trace 2 - Inductor Switching Node (2V/div.)
Trace 3 - 12V input (VCC1 and VCC2) (5V/div.)
Trace 4 - 5V Input (1V/div.)
Programmable Output
The CS5150H is designed to provide two methods for programming the output voltage of the power supply. A four
bit on board digital to analog converter (DAC) is used to
program the output voltage from 2.14V to 3.54V in 100mV
steps, depending on the digital input code. If all four bits
are left open, the CS5150H enters adjust mode. In adjust
mode, the designer can choose any output voltage by using
resistor divider feedback to the VFB and VFFB pins, as in traditional controllers. The CS5150H is specifically designed
to be upwards compatible with the CS5155H, which uses a
five bit DAC code.
Figure 2: CS5150H demonstration board startup in response to increasing 12V and 5V input voltages. Extended off time is followed by normal
off time operation when output voltage achieves regulation to the error
amplifier output.
Start Up
Until the voltage on the VCC1 supply pin exceeds the 3.9V
monitor threshold, the soft start and gate pins are held low.
The FAULT latch is reset (no Fault condition). The output
of the error amplifier (COMP) is pulled up to 1V by the
comparator clamp. When the VCC1 pin exceeds the monitor
threshold, the GateH output is activated, and the soft start
capacitor begins charging. The GateH output will remain
on, enabling the NFET switch, until terminated by either
the PWM comparator, or the maximum on time timer.
If the maximum on time is exceeded before the regulator
output voltage achieves the 1V level, the pulse is terminated. The GateH pin drives low, and the GateL pin drives
high for the duration of the extended off time. This time is
Trace 1 - Regulator Output Voltage (1V/div.)
Trace 3 - COMP Pin (error amplifier output) (1V/div.)
Trace 4 - Soft Start Pin (2V/div.)
Figure 3: CS5150H demonstration board startup waveforms.
6
If the input voltage rises quickly, or the regulator output is
enabled externally, output voltage will increase to the level
set by the error amplifier output more rapidly, usually
within a couple of cycles (see Figure 4).
Trace1 - Regulator Output Voltage (10V/div.)
Trace 2 - Inductor Switching Node (5V/div.)
Figure 6: Peak-to-peak ripple on VOUT = 2.8V, IOUT = 13A (heavy load).
Trace 1 - Regulator Output Voltage (5V/div.)
Trace 2 - Inductor Switching Node (5V/div.)
Transient Response
The CS5150H V2™ control loop’s 100ns reaction time provides unprecedented transient response to changes in
input voltage or output current. Pulse by pulse adjustment
of duty cycle is provided to quickly ramp the inductor current to the required level. Since the inductor current cannot
be changed instantaneously, regulation is maintained by
the output capacitor(s) during the time required to slew the
inductor current.
Overall load transient response is further improved
through a feature called “adaptive voltage positioning”.
This technique pre-positions the output capacitor’s voltage
to reduce total output voltage excursions during changes
in load.
Holding tolerance to 1% allows the error amplifier’s reference voltage to be targeted +40mV high without compromising DC accuracy. A “droop resistor“, implemented
through a PC board trace, connects the error amplifier’s
feedback pin (VFB) to the output capacitors and load and
carries the output current. With no load, there is no DC
drop across this resistor, producing an output voltage
tracking the error amplifier’s, including the +40mV offset.
When the full load current is delivered, an 80mV drop is
developed across this resistor. This results in output voltage being offset -40mV low.
The result of adaptive voltage positioning is that additional
margin is provided for a load transient before reaching the
output voltage specification limits. When load current suddenly increases from its minimum level, the output capacitor is pre-positioned +40mV. Conversely, when load current suddenly decreases from its maximum level, the output capacitor is pre-positioned -40mV (see Figures 7, 8, and
9). For best transient response, a combination of a number
of high frequency and bulk output capacitors are usually
used.
Figure 4: CS5150H demonstration board enable startup waveforms.
Normal Operation
During normal operation, switch off time is constant and
set by the COFF capacitor. Switch on time is adjusted by the
V2™ control loop to maintain regulation. This results in
changes in regulator switching frequency, duty cycle, and
output ripple in response to changes in load and line.
Output voltage ripple will be determined by inductor ripple current working into the ESR of the output capacitors
(see Figures 5 and 6).
Trace 1 - Regulator Output Voltage (10V/div.)
Trace 2 - Inductor Switching Node (5V/div.)
Figure 5: Peak-to-peak ripple on VOUT = 2.8V, IOUT = 0.5A (light load).
7
CS5150H
Applications Information: continued
CS5150H
Applications Information: continued
If the maximum on time is exceeded while responding to a
sudden increase in load current, a normal off time occurs to
prevent saturation of the output inductor.
Trace1 - Regulator Output Voltage (1V/div.)
Trace 2 - Inductor Switching Node (5V/div.)
Trace 3 - Output Current (13 to 0.5 Amps) (20V/div.)
Figure 9: CS5150H demonstration board response to 13A load turn off
(output set for 2.8V). V2™ control topology immediately connects
inductor to ground, providing 0% duty cycle. Regulation is achieved in
less than 10µs.
Trace 1 - Regulator Output Voltage (1V/div.)
Trace 3 - Regulator Output Current (20V/div.)
Figure 7: CS5150H demonstration board response to a 0.5 to 13A load
pulse (output set for 2.8V).
Protection and Monitoring Features
VCC1 Monitor
To maintain predictable startup and shutdown characteristics an internal VCC1 monitor circuit is used to prevent the
part from operating below 3.75V minimum startup. The
VCC1 monitor comparator provides hysteresis and guarantees a 3.70V minimum shutdown threshold.
Short Circuit Protection
A lossless hiccup mode short circuit protection feature is
provided, requiring only the soft start capacitor to implement. If a short circuit condition occurs (VFFB < 1V), the VFFB
low comparator sets the FAULT latch. This causes the top
MOSFET to shut off, disconnecting the regulator from its
input voltage. The soft start capacitor is then slowly discharged by a 2µA current source until it reaches its lower
0.7V threshold. The regulator will then attempt to restart normally, operating in its extended off time mode with a 50%
duty cycle, while the soft start capacitor is charged with a
60µA charge current.
If the short circuit condition persists, the regulator output
will not achieve the 1V low VFFB comparator threshold
before the soft start capacitor is charged to its upper 2.5V
threshold. If this happens the cycle will repeat itself until the
short is removed. The soft start charge/discharge current
ratio sets the duty cycle for the pulses (2µA/60µA = 3.3%),
while actual duty cycle is half that due to the extended off
time mode (1.65%).
Trace 1 - Regulator Output Voltage (1V/div.)
Trace 2 - Inductor Switching Node (5V/div.)
Trace 3 - Output Current (0.5 to 13 Amps) (20V/div.)
Figure 8: CS5150H demonstration board response to 13A load turn on
(output set for 2.8V). Upon completing a normal off time, the V2™ control loop immediately connects the inductor to the input voltage, providing 100% duty cycle. Regulation is achieved in less than 20µs.
8
This protection feature results in less stress to the regulator
components, input power supply, and PC board traces
than occurs with constant current limit protection (see
Figures 10 and 11).
If the short circuit condition is removed, output voltage
will rise above the 1V level, preventing the FAULT latch
from being set, allowing normal operation to resume.
ed, resulting in a “crowbar” action to clamp the output
voltage and prevent damage to the load (see Figures 12 and
13). The regulator will remain in this state until the overvoltage condition ceases or the input voltage is pulled low.
The bottom FET and board trace must be properly
designed to implement the OVP function.
Trace 4 = 5V from PC Power Supply (5V/div.)
Trace1 = Regulator Output Voltage (1V/div.)
Trace 2 = Inductor Switching Node (5V/div.)
Trace 4 - 5V Supply Voltage (2V/div.)
Trace 3 - Soft Start Timing Capacitor (1V/div.)
Trace 2 - Inductor Switching Node (2V/div.)
Figure 12: OVP response to an input-to-output short circuit by immediately providing 0% duty cycle, crow-barring the input voltage to
ground.
Figure 10: CS5150H demonstration board hiccup mode short circuit protection. Gate pulses are delivered while the soft start capacitor charges,
and cease during discharge.
Trace 4 = 5V from PC Power Supply (2V/div.)
Trace 1 = Regulator Output Voltage (1V/div.)
Trace 4 = 5V from PC Power Supply (2V/div.)
Trace 2 = Inductor Switching Node (2V/div.)
Figure 13: OVP response to an input-to-output short circuit by pulling
the input voltage to ground.
Figure 11: Startup with regulator output shorted.
External Output Enable Circuit
On/off control of the regulator can be implemented
through the addition of two additional discrete components (see Figure 14). This circuit operates by pulling the
soft start pin high, and the VFFB pin low, emulating a short
circuit condition.
Overvoltage Protection
Overvoltage protection (OVP) is provided as result of the
normal operation of the V2™ control topology and requires
no additional external components. The control loop
responds to an overvoltage condition within 100ns, causing
the top MOSFET to shut off, disconnecting the regulator
from its input voltage. The bottom MOSFET is then activat-
9
CS5150H
Applications Information: continued
CS5150H
Applications Information: continued
5V
MMUN2111T1 (SOT-23)
5
SS
CS5150H
8 V
FFB
IN4148
Shutdown
Input
Figure 14: Implementing shutdown with the CS5150H.
Trace 3 = 12V Input (VCC1) and VCC2) (10V/div.)
Trace 4 = 5V Input (2V/div.)
Trace 1 = Regulator Output Voltage (1V/div.)
Trace 2 = Power Good Signal (2V/div.)
External Power Good Circuit
An optional Power Good signal can be generated through
the use of four additional external components (see Figure
15). The threshold voltage of the Power Good signal can be
adjusted per the following equation:
VPower Good =
Figure 16: CS5150H demonstration board during power up. Power Good
signal is activated when output voltage reaches 1.70V.
Selecting External Components
The CS5150H can be used with a wide range of external
power components to optimize the cost and performance of
a particular design. The following information can be used
as general guidelines to assist in their selection.
(R1 + R2) × 0.65V
R2
This circuit provides an open collector output that drives
the Power Good output to ground for regulator voltages
less than VPower Good.
NFET Power Transistors
Both logic level and standard MOSFETs can be used. The
reference designs derive gate drive from the 12V supply
which is generally available in most computer systems and
utilize logic level MOSFETs. A charge pump may be easily
implemented to permit use of standard MOSFETs or support 5V or 12V only systems (maximum of 20V). Multiple
MOSFETs may be paralleled to reduce losses and improve
efficiency and thermal management.
Voltage applied to the MOSFET gates depends on the
application circuit used. Both upper and lower gate driver
outputs are specified to drive to within 1.5V of ground
when in the low state and to within 2V of their respective
bias supplies when in the high state. In practice, the MOSFET gates will be driven rail to rail due to overshoot caused
by the capacitive load they present to the controller IC. For
the typical application where VCC1 = VCC2 = 12V and 5V is
used as the source for the regulator output current, the following gate drive is provided;
5V
R3
10k
R1
10k
PN3904
VOUT
CS5150H
Power Good
PN3904
R2
6.2k
Figure 15: Implementing Power Good with the CS5150H.
VGATE(H) = 12V - 5V = 7V, VGATE(L) = 12V (see Figure 17).
10
CS5150H
Applications Information: continued
COFF timing capacitor:
COFF =
Period × (1 - duty cycle)
,
4848.5
where:
Period =
1
switching frequency
Schottky Diode for Synchronous MOSFET
A Schottky diode may be placed in parallel with the synchronous MOSFET to conduct the inductor current upon
turn off of the switching MOSFET to improve efficiency.
The CS5150H reference circuit does not use this device due
to its excellent design. Instead, the body diode of the synchronous MOSFET is utilized to reduce cost and conducts
the inductor current. For a design operating at 200kHz or so,
the low non-overlap time combined with Schottky forward
recovery time may make the benefits of this device not
worth the additional expense (see Figure 6, channel 2). The
power dissipation in the synchronous MOSFET due to body
diode conduction can be estimated by the following equation:
Trace 3 = VGATE(H) (10V/div.)
Math 1= VGATE(H) - 5VIN
Trace 4 = VGATE(L) (10V/div.)
Trace 2 = Inductor Switching Node (5V/div.)
Figure 17: CS5150H gate drive waveforms depicting rail to rail swing.
The most important aspect of MOSFET performance is
RDSON, which effects regulator efficiency and MOSFET
thermal management requirements.
The power dissipated by the MOSFETs may be estimated
as follows;
Switching MOSFET:
Power = ILOAD2 × RDSON × duty cycle
Power = Vbd × ILOAD × conduction time × switching frequency
Where Vbd = the forward drop of the MOSFET body diode.
For the CS5150H demonstration board as shown in Figure 6;
Power = 1.6V × 13A × 100ns × 233kHz = 0.48W
This is only 1.3% of the 36.4W being delivered to the load.
Synchronous MOSFET:
Power = ILOAD2 × RDSON × (1 - duty cycle)
“Droop” Resistor for Adaptive Voltage Positioning
Adaptive voltage positioning is used to reduce output voltage excursions during abrupt changes in load current.
Regulator output voltage is offset +40mV when the regulator is unloaded, and -40mV at full load. This results in
increased margin before encountering minimum and maximum transient voltage limits, allowing use of less capacitance on the regulator output (see Figure 7).
To implement adaptive voltage positioning, a “droop”
resistor must be connected between the output inductor
and output capacitors and load. This is normally implemented by a PC board trace of the following value:
Duty Cycle =
VOUT + (ILOAD × RDSON OF SYNCH FET)
VIN + (ILOAD × RDSON OF SYNCH FET) - (ILOAD × RDSON OF SWITCH FET)
Off Time Capacitor (COFF)
The COFF timing capacitor sets the regulator off time:
TOFF = COFF × 4848.5
When the VFFB pin is less than 1V, the current charging the
COFF capacitor is reduced. The extended off time can be calculated as follows:
TOFF = COFF × 24,242.5.
Off time will be determined by either the TOFF time, or the
time out timer, whichever is longer.
The preceding equations for duty cycle can also be used to
calculate the regulator switching frequency and select the
RDROOP =
80mV
IMAX
Adaptive voltage positioning can be disabled for improved
DC regulation by connecting the VFB pin directly to the load
using a separate, non-load current carrying circuit trace.
11
CS5150H
Applications Information: continued
Input and Output Capacitors
These components must be selected and placed carefully to
yield optimal results. Capacitors should be chosen to provide acceptable ripple on the input supply lines and regulator output voltage. Key specifications for input capacitors
are their ripple rating, while ESR is important for output
capacitors. For best transient response, a combination of
low value/high frequency and bulk capacitors placed close
to the load will be required.
Output Inductor
The inductor should be selected based on its inductance,
current capability, and DC resistance. Increasing the inductor value will decrease output voltage ripple, but degrade
transient response.
Layout Guidelines
1. Place 12V filter capacitor next to the IC and connect
capacitor ground to pin 11 (PGnd).
2. Connect pin 11 (PGnd) with a separate trace to the
ground terminals of the 5V input capacitors.
3. Place fast feedback filter capacitor next to pin 8 (VFFB)
and connect its ground terminal with a separate, wide trace
directly to pin 14 (LGnd).
4. Connect the ground terminals of the Compensation
capacitor directly to the ground of the fast feedback filter
capacitor to prevent common mode noise from effecting
the PWM comparator.
5. Place the output filter capacitor(s) as close to the load as
possible and connect the ground terminal to pin 14 (LGnd).
6. To implement adaptive voltage positioning, connect
both slow and fast feedback pins 16 (VFB) and 8 (VFFB) to
the regulator output right at the inductor terminal. Connect
inductor to the output capacitors via a trace with the following resistance:
Thermal Management
Thermal Considerations for Power MOSFETs and Diodes
In order to maintain good reliability, the junction temperature of the semiconductor components should be kept to a
maximum of 150°C or lower. The thermal impedance (junction to ambient) required to meet this requirement can be
calculated as follows:
Thermal Impedance =
RTRACE =
80mV
IMAX
This causes the output voltage to be +40mV with no load,
and -40mV with a full load, improving regulator transient
response. This trace must be wide enough to carry the full
output current. (Typical trace is 1.0 inch long, 0.17 inch
wide). Care should be taken to minimize any additional
losses after the feedback connection point to maximize regulation.
7. If DC regulation is to be optimized (at the expense of
degraded transient regulation), adaptive voltage positioning can be disabled by connecting to VFB pin directly to the
load with a separate trace (remote sense).
8. Place 5V input capacitors close to the switching MOSFET
and synchronous MOSFET.
Route gate drive signals VGATE(H) (pin 10) and VGATE(L)
(pin 12 when used) with traces that are a minimum of 0.025
inches wide.
TJUNCTION(MAX) - TAMBIENT
Power
A heatsink may be added to TO-220 components to reduce
their thermal impedance. A number of PC board layout
techniques such as thermal vias and additional copper foil
area can be used to improve the power handling capability
of surface mount components.
EMI Management
As a consequence of large currents being turned on and off
at high frequency, switching regulators generate noise as a
consequence of their normal operation. When designing for
compliance with EMI/EMC regulations, additional components may be added to reduce noise emissions. These
components are not required for regulator operation and
experimental results may allow them to be eliminated. The
input filter inductor may not be required because bulk filter
and bypass capacitors, as well as other loads located on the
board will tend to reduce regulator di/dt effects on the circuit board and input power supply. Placement of the
power component to minimize routing distance will also
help to reduce emissions.
VCC
0.1µF
To the negative terminal of the
input capacitors
15
11
1.0µF
VCOMP
8
100pF
VFFB
5
SOFTSTART
OFF TIME
To the negative terminal of the output capacitors
2µH
33Ω
2µH
1200µF x 3/16V
Figure 20: Layout Guidelines
+
1000pF
Figure 18: Filter components
Figure 19: Input Filter
12
CS5150H
Additional Application Circuits
5V
12V
0.1µF
MBRS
120
+
1µF
1µF
100µF/10V x 3
Tantalum
+12V
1N5818
22Ω
1/4W
MBRS120
MBRS120
1N5818
1N4746
18V 1W
0.1µF
1µF
Si4410DY
1µF
VCC2 VGATE(H)
VCC1
3µH
3.3V/10A
VCC2 VGATE(H)
VCC1
VID0
820µF/16V × 4
Aluminum
Electrolytic
FY10AAJ03
1.1µH
3.3V/5A
VID0
VID1
VID2
VID1
CS5150H
VID3
330pF
VID2
Si9410DY
VGATE(L)
COFF
VGATE(L)
COFF
0.1µF
0.33µF
+
100pF
PGnd
COMP
VFFB
LGnd
100µF/10V x 3
Tantalum
1200µF/10V × 2
Aluminum
Electrolytic
FY10AAJ03
SS
3.3k
VFFB
LGnd
+
FY10AAJ03
330pF
VFB
COMP
VFB
CS5150H
VID3
PGnd
SS
0.1µF
+
0.33µF
3.3k
100pF
Figure 23: 12V to 3.3V/5A converter with remote sense.
Figure 21: 5V to 3.3V/10A converter.
5V
3.3V
12V
0.1µF
MBRS
120
1µF
MBRS120
1µF
+
1µF
MBRS120
100µF/10V x 3
Tantalum
Si4410
VCC2 VGATE(H)
VCC1
3µH
+
Remote
Sense
Si9410
3.3V/10A
VID3
VFB
Si9410
COFF
VID3
100µF/10V x 3
Tantalum
VFB
COFF
LGnd
VFFB
0.1µF
3.3k
100pF
PGnd
COMP
LGnd
0.33µF
Connect to
other circuits for
current sharing
Si9410
VGATE(L)
330pF
PGnd
COMP
2.5V/7A
+
CS5150H
SS
SS
0.33µF
+
VGATE(L)
330pF
0.1µF
10Ω
5µH
VID0
VID1
VID2
CS5150H
VCC2 VGATE(H)
VCC1
VID0
VID1
VID2
33µF/25V x 3
Tantalum
VFFB
3.3k
100pF
Figure 24: 3.3V to 2.5V/7A converter with 12V bias.
Figure 22: 5V to 3.3V/10A converter with current sharing.
13
100µF/10V x 2
Tantalum
CS5150H
Package Specification
PACKAGE THERMAL DATA
PACKAGE DIMENSIONS IN mm (INCHES)
D
Lead Count
Metric
Max
Min
10.00
9.80
16L SO Narrow
Thermal Data
English
Max Min
.394 .386
RΘJC
RΘJA
typ
typ
16L
SO Narrow
28
115
˚C/W
˚C/W
Surface Mount Narrow Body (D); 150 mil wide
4.00 (.157)
3.80 (.150)
6.20 (.244)
5.80 (.228)
0.51 (.020)
0.33 (.013)
1.27 (.050) BSC
1.75 (.069) MAX
1.57 (.062)
1.37 (.054)
1.27 (.050)
0.40 (.016)
0.25 (.010)
0.19 (.008)
D
0.25 (0.10)
0.10 (.004)
REF: JEDEC MS-012
Ordering Information
Part Number
CS5150HGD16
CS5150HGDR16
Rev. 1/21/99
Description
16L SO Narrow
16L SO Narrow (tape & reel)
Cherry Semiconductor Corporation reserves the right to
make changes to the specifications without notice. Please
contact Cherry Semiconductor Corporation for the latest
available information.
14
© 1999 Cherry Semiconductor Corporation