LINER LT3800 High-voltage synchronous current mode step-down controller Datasheet

LT3800
High-Voltage Synchronous
Current Mode Step-Down
Controller
DESCRIPTIO
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FEATURES
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Wide 4V to 60V Input Voltage Range
Output Voltages up to 36V
Adaptive Nonoverlap Circuitry Prevents Switch
Shoot-Through
Reverse Inductor Current Inhibit for Discontinuous
Operation Improves Efficiency with Light Loads
Output Slew Rate Controlled Soft-Start with
Auto-Reset
100µA No Load Quiescent Current
Low 10µA Current Shutdown
1% Regulation Accuracy
200kHz Operating Frequency
Standard Gate N-Channel Power MOSFETs
Current Limit Unaffected by Duty Cycle
Reverse Overcurrent Protection
16-Lead Thermally Enhanced TSSOP Package
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APPLICATIO S
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12V and 42V Automotive and Heavy Equipment
48V Telecom Power Supplies
Avionics and Industrial Control Systems
Distributed Power Converters
The LT®3800 is a 200kHz fixed frequency high voltage
synchronous current mode step-down switching regulator controller. The IC drives standard gate N-channel power
MOSFETs and can operate with input voltages from 4V to
60V. An onboard regulator provides IC power directly from
VIN and provides for output-derived power to minimize VIN
quiescent current. MOSFET drivers employ an internal
dynamic bootstrap feature, maximizing gate-source “ON”
voltages during normal operation for improved operating
efficiencies. The LT3800 incorporates Burst Mode® operation, which reduces no load quiescent current to under
100µA. Light load efficiencies are also improved through
a reverse inductor current inhibit, allowing the controller
to support discontinuous operation. Both Burst Mode
operation and the reverse-current inhibit features can be
disabled if desired. The LT3800 incorporates a programmable soft-start that directly controls the voltage slew rate
of the converter output for reduced startup surge currents
and overshoot errors. The LT3800 is available in a 16-lead
thermally enhanced TSSOP package.
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
Burst Mode is a registered trademark of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
Protected by U.S. Patents, including 5481178, 6611131, 6304066, 6498466, 6580258.
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TYPICAL APPLICATIO
12V 75W DC/DC Converter with Reverse Current Inhibit and Input UVLO
+
56µF
×2
VIN
Efficiency and Power Loss
1µF
×3
BOOST
100
1µF
1M
LT3800
82.5k
95
SW
BAS19
200k
CSS
20k
1%
174k
1%
BURST_EN
15µH
1N4148
VCC
1µF
VFB
BG
Si7370DP
B160
90
85
VIN = 36V
5
VIN = 60V
4
VIN = 48V
3
2
80
POWER LOSS (W)
1.5nF
SHDN
6
VIN = 24V
Si7850DP
TG
EFFICIENCY (%)
VIN
20V TO 55V
LOSS (48V)
VC
100pF
82.5k
680pF
PGND
1
75
SENSE–
SENSE+
SGND
70
0.1
0.015Ω
+
10µF
VOUT
12V AT 75W
1
ILOAD (A)
10
0
3800 TA01b
270µF
3800 TA01a
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LT3800
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ABSOLUTE
AXI U RATI GS (Note 1) PI CO FIGURATIO
Supply Voltages
Input Supply Pin (VIN) .............................. –0.3V to 65V
Boosted Supply Pin (BOOST) ................... –0.3V to 80V
Boosted Supply Voltage (BOOST – SW) .. –0.3V to 24V
Boosted Supply Reference Pin (SW) ........... –2V to 65V
Local Supply Pin (VCC) ............................. –0.3V to 24V
Input Voltages
SENSE+, SENSE– ...................................... –0.3V to 40V
SENSE+ – SENSE– ......................................... –1V to 1V
BURST_EN Pin ......................................... –0.3V to 24V
Other Inputs (SHDN, CSS, VFB, VC) .......... –0.3V to 5.0V
Input Currents
SHDN, CSS ............................................... –1mA to 1mA
Maximum Temperatures
Operating Junction Temperature Range (Note 2)
LT3800E (Note 3) ............................. –40°C to 125°C
LT3800I ............................................ –40°C to 125°C
Storage Temperature Range ................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
TOP VIEW
VIN
1
16 BOOST
NC
2
15 TG
SHDN
3
14 SW
CSS
4
BURST_EN
5
12 VCC
VFB
6
11 BG
VC
7
10 PGND
SENSE –
8
9
17
13 NC
SENSE+
FE PACKAGE
16-LEAD PLASTIC TSSOP
TJMAX = 125°C, θJA = 40°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 17) IS SGND
MUST BE SOLDERED TO PCB
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ORDER I FOR ATIO
LEAD FREE FINISH
LT3800EFE#PBF
LT3800IFE#PBF
TAPE AND REEL
LT3800EFE#TRPBF
LT3800IFE#TRPBF
PART MARKING
3800EFE
3800IFE
PACKAGE DESCRIPTION
16-Lead Plastic TSSOP
16-Lead Plastic TSSOP
TEMPERATURE RANGE
–40°C to 125°C
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges.
Consult LTC Marketing for information on nonstandard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 20V, VCC = BOOST = BURST_EN = 10V, SHDN = 2V,
SENSE – = SENSE+ = 10V, SGND = PGND = SW = 0V, CTG = CBG = 3300pF, unless otherwise noted.
SYMBOL
PARAMETER
VIN
Operating Voltage Range (Note 4)
Minimum Start Voltage
UVLO Threshold (Falling)
UVLO Hysteresis
IVIN
VIN Supply Current
VIN Burst Mode Current
VIN Shutdown Current
VBOOST
Operating Voltage
Operating Voltage Range (Note 5)
UVLO Threshold (Rising)
UVLO Hysteresis
CONDITIONS
MIN
●
●
●
VCC > 9V
VBURST_EN = 0V, VFB = 1.35V
VSHDN = 0V
VBOOST – VSW
VBOOST – VSW
VBOOST – VSW
●
4
7.5
3.65
TYP
MAX
60
3.80
670
20
20
8
●
●
3.95
15
75
20
5
0.4
UNITS
V
V
V
mV
µA
µA
µA
V
V
V
V
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LT3800
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 20V, VCC = BOOST = BURST_EN = 10V, SHDN = 2V,
SENSE – = SENSE+ = 10V, SGND = PGND = SW = 0V, CTG = CBG = 3300pF, unless otherwise noted.
SYMBOL
PARAMETER
IBOOST
BOOST Supply Current (Note 6)
BOOST Burst Mode Current
BOOST Shutdown Current
VCC
IVCC
VCC Supply Current (Note 6)
VCC Burst Mode Current
VCC Shutdown Current
Short-Circuit Current
Enable Threshold (Rising)
Threshold Hysteresis
VSENSE
Common Mode Range
Current Limit Sense Voltage
Reverse Protect Sense Voltage
Reverse Current Offset
ISENSE
Input Current
(ISENSE+ + ISENSE–)
fO
Operating Frequency
Error Amp Reference Voltage
MIN
●
●
MAX
●
●
–40
●
1.30
●
●
0
140
VSENSE(CM) = 0V
VSENSE(CM) = 2.75V
VSENSE(CM) > 4V
mA
µA
µA
V
V
V
mV
3
80
20
–120
3.6
mA
µA
µA
mA
1.35
120
1.40
V
mV
150
–150
10
36
175
0.8
–20
–0.3
mV
mV
mV
mA
µA
mA
190
175
200
●
210
220
1.224
1.215
1.231
●
1.238
1.245
Measured at VFB Pin
UNITS
20
8.3
8.0
6.25
500
VBURST_EN = 0V
VSHDN = 0V
VSENSE+ – VSENSE–
VSENSE+ – VSENSE–, VBURST_EN = VCC
VBURST_EN = 0V or VBURST_EN = VFB
TYP
1.4
0.1
0.1
VBURST_EN = 0V
VSHDN = 0V
Operating Voltage (Note 5)
Output Voltage
UVLO Threshold (Rising)
UVLO Hysteresis
VSHDN
VFB
CONDITIONS
kHz
kHz
V
V
IFB
Feedback Input Current
VFB(SS)
Soft-Start Disable Voltage
Soft-Start Disable Hysteresis
ICSS
Soft-Start Capacitor Control Current
gm
Error Amp Transconductance
AV
Error Amp DC Voltage Gain
VC
Error Amp Output Range
1.2
V
IVC
Error Amp Sink/Source Current
±30
µA
VTG,BG
Gate Drive Output On Voltage (Note 7)
Gate Drive Output Off Voltage
9.8
0.1
V
V
tTG,BG
Gate Drive Rise/Fall Time
50
ns
tTG(OFF)
Minimum Off Time
450
ns
tTG(ON)
Minimum On Time
tNOL
Gate Drive Nonoverlap Time
VFB Rising
25
nA
1.185
300
V
mV
µA
2
●
275
350
400
62
Zero Current to Current Limit
10% to 90% or 90% to 10%
●
TG Fall to BG Rise
BG Fall to TG Rise
300
200
150
µmhos
dB
500
ns
ns
ns
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LT3800
ELECTRICAL CHARACTERISTICS
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: This IC includes overtemperature protection that is intended to
protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
Note 3: The LT3800E is guaranteed to meet performance specifications from
0°C to 125°C junction temperature. Specifications over the –40°C to 125°C
operating junction temperature range are assured by design, characterization
and correlation with statistical process controls. The LT3800I is guaranteed
over the full –40°C to 125°C operating junction temperature range.
Note 4: VIN voltages below the start-up threshold (7.5V) are only
supported when VCC is externally driven above 6.5V.
Note 5: Operating range dictated by MOSFET absolute maximum gatesource voltage ratings.
Note 6: Supply current specification does not include switch drive
currents. Actual supply currents will be higher.
Note 7: DC measurement of gate drive output “ON” voltage is typically
8.6V. Internal dynamic bootstrap operation yields typical gate “ON”
voltages of 9.8V during standard switching operation. Standard operation
gate “ON” voltage is not tested but guaranteed by design.
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TYPICAL PERFOR A CE CHARACTERISTICS
Shutdown Threshold (Falling)
vs Temperature
Shutdown Threshold (Rising)
vs Temperature
1.36
1.35
1.34
1.33
–50
0
25
50
75
TEMPERATURE (°C)
–25
100
ICC = 0mA
1.235
8.0
VCC (V)
SHUTDOWN THRESHOLD, FALLING (V)
SHUTDOWN THRESHOLD, RISING (V)
VCC vs Temperature
8.1
1.240
1.37
1.230
7.9
1.225
1.220
–50
125
–25
0
25
50
75
TEMPERATURE (°C)
100
3800 G01
ICC CURRENT LIMIT (mA)
VCC (V)
VCC (V)
6
4
10
15 20 25
ICC(LOAD) (mA)
30
35
40
4
5
6
7
8
9
10
11
12
VIN (V)
3800 G04
125
150
125
100
75
3
5
100
175
5
0
50
75
25
TEMPERATURE (°C)
ICC Current Limit vs Temperature
7
7.90
0
200
ICC = 20mA
TA = 25°C
TA = 25°C
7.95
–25
3800 G03
VCC vs VIN
8
8.00
7.85
7.8
–50
125
3800 G02
VCC vs ICC(LOAD)
8.05
ICC = 20mA
3800 G05
50
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
3800 G06
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LT3800
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TYPICAL PERFOR A CE CHARACTERISTICS
VCC UVLO Threshold (Rising)
vs Temperature
25
ERROR AMP TRANSCONDUCTANCE (µmho)
TA = 25°C
20
6.25
ICC (µA)
VCC UVLO THRESHOLD, RISING (V)
6.30
15
10
6.20
5
6.15
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
0
125
0
2
4
6
8
0
–200
–400
0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0
VSENSE(CM) (V)
0
25
50
75
TEMPERATURE (°C)
125
1.232
210
200
190
180
–50
100
3800 G08
ERROR AMP REFERENCE (V)
200
–25
Error Amp Reference
vs Temperature
220
OPERATING FREQUENCY (kHz)
400
340
Operating Frequency
vs Temperature
TA = 25°C
600
360
3800 G12
I(SENSE+ + SENSE–) vs VSENSE(CM)
800
380
320
–50
10 12 14 16 18 20
VCC (V)
3800 G07
I(SENSE+ + SENSE–) (µA)
Error Amp Transconductance
vs Temperature
ICC vs VCC (SHDN = 0V)
–25
0
25
50
75
TEMPERATURE (°C)
3800 G09
100
125
1.231
1.230
1.229
1.228
1.227
–50
3800 G10
–25
0
25
50
75
TEMPERATURE (°C)
100
125
3800 G11
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VIN (Pin 1): Converter Input Supply.
NC (Pin 2): No Connection.
SHDN (Pin 3): Precision Shutdown Pin. Enable threshold
is 1.35V (rising) with 120mV of input hysteresis. When in
shutdown mode, all internal IC functions are disabled. The
precision threshold allows use of the SHDN pin to incorporate UVLO functions. If the SHDN pin is pulled below
0.7V, the IC enters a low current shutdown mode with
IVIN < 10µA. In low-current shutdown, the IC will sink 20µA
from the VCC pin until that local supply has collapsed.
Typical pin input bias current is <10nA and the pin is
internally clamped to 6V.
CSS (Pin 4): Soft-Start AC Coupling Capacitor Input.
Connect capacitor (CSS) in series with a 200k resistor from
pin to converter output (VOUT). Controls converter startup output voltage slew rate (∆VOUT/∆t). Slew rate corresponds to 2µA average current through the soft-start
coupling capacitor. The capacitor value for a desired
output startup slew rate follows the relation:
CSS = 2µA/(∆VOUT/∆t)
Shorting this pin to SGND disables the soft-start function
BURST_EN (Pin 5): Burst Mode Operation Enable Pin.
This pin also controls reverse-inhibit mode of operation.
When the pin voltage is below 0.5V, Burst Mode operation
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LT3800
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and reverse-current inhibit functions are enabled. When
the pin voltage is above 0.5V, Burst Mode operation is
disabled, but reverse-current inhibit operation is maintained. DC/DC converters operating with reverse-current
inhibit operation (BURST_EN = VFB) have a 1mA minimum
load requirement. Reverse-current inhibit is disabled when
the pin voltage is above 2.5V. This pin is typically shorted
to ground to enable Burst Mode operation and reversecurrent inhibit, shorted to VFB to disable Burst Mode
operation while enabling reverse-current inhibit, and connected to VCC pin to disable both functions. See Applications Information section.
V FB (Pin 6): Error Amplifier Inverting Input. The
noninverting input of the error amplifier is connected to an
internal 1.231V reference. Desired converter output voltage (VOUT) is programmed by connecting a resistive
divider from the converter output to the VFB pin. Values for
the resistor connected from VOUT to VFB (R2) and the
resistor connected from VFB to ground (R1) can be calculated via the following relationship:
⎛V
⎞
R2 = R1 • ⎜ OUT – 1⎟
⎝ 1.231 ⎠
The VFB pin input bias current is 25nA, so use of extremely
high value feedback resistors could cause a converter
output that is slightly higher than expected. Bias current
error at the output can be estimated as:
∆VOUT(BIAS) = 25nA • R2
VC (Pin 7): Error Amplifier Output. The voltage on the VC
pin corresponds to the maximum (peak) switch current
per oscillator cycle. The error amplifier is typically configured as an integrator by connecting an RC network from
this pin to ground. This network creates the dominant pole
for the converter voltage regulation feedback loop. Specific integrator characteristics can be configured to optimize transient response. Connecting a 100pF or greater
high frequency bypass capacitor from this pin to ground
is also recommended. When Burst Mode operation is
enabled (see Pin 5 description), an internal low impedance
clamp on the VC pin is set at 100mV below the burst
threshold, which limits the negative excursion of the pin
voltage. Therefore, this pin cannot be pulled low with a
low-impedance source. If the VC pin must be externally
manipulated, do so through a 1kΩ series resistance.
SENSE– (Pin 8): Negative Input for Current Sense Amplifier. Sensed inductor current limit set at ±150mV across
SENSE inputs.
SENSE+ (Pin 9): Positive Input for Current Sense Amplifier. Sensed inductor current limit set at ±150mV across
SENSE inputs.
PGND (Pin 10): High Current Ground Reference for Synchronous Switch. Current path from pin to negative terminal of VCC decoupling capacitor must not corrupt SGND.
BG (Pin 11): Synchronous Switch Gate Drive Output.
VCC (Pin 12): Internal Regulator Output. Most IC functions are powered from this pin. Driving this pin from an
external source reduces VIN pin current to 20µA. This pin
is decoupled with a low ESR 1µF capacitor to PGND.
In shutdown mode, this pin sinks 20µA until the pin
voltage is discharged to 0V. See Typical Performance
Characteristics.
NC (Pin 13): No Connection.
SW (Pin 14): Reference for VBOOST Supply and High
Current Return for Bootstrapped Switch.
TG (Pin 15): Bootstrapped Switch Gate Drive Output.
BOOST (Pin 16): Bootstrapped Supply – Maximum Operating Voltage (Ground Referred) to 75V. This pin is
decoupled with a low ESR 1µF capacitor to pin SW. The
voltage on the decoupling capacitor is refreshed through
a rectifier from either VCC or an external source.
Exposed Package Backside (SGND) (Pin 17): Low Noise
Ground Reference. SGND connection is made through the
exposed lead frame on back of TSSOP package which
must be soldered to the PCB ground.
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LT3800
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FU CTIO AL DIAGRA
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VIN
UVLO
(<4V)
8V
REG
3.8V
REG
INTERNAL
SUPPLY RAIL
FEEDBACK
REFERENCE
1.231V
+
–
BST
UVLO
VCC
UVLO
(<6V)
VIN 1
16 BOOST
BOOSTED
SWITCH
DRIVER
DRIVE
CONTROL
15 TG
14 SW
NOL
SWITCH
DRIVE
LOGIC
CONTROL
12 VCC
+
SHDN 3
SYNCHRONOUS
SWITCH DRIVER
DRIVE
CONTROL
11 BG
–
10 PGND
+
BURST_EN
5
–
OSCILLATOR
+
–
VFB
6
–
ERROR
AMP
gm
0.5V
7
SOFT-START
DISABLE/BURST
ENABLE
S
R
+
SLOPE COMP
GENERATOR
+
–
1V
R
S
+
REVERSE
CURRENT
INHIBIT
–
+
+
–
Burst Mode
OPERATION
1.185V
2µA
CSS
Q
CURRENT
SENSE
COMPARATOR
–
VC
Q
160mV
4
–
+
GND 17
9
SENSE+
8
SENSE–
10mV
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LT3800
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APPLICATIO S I FOR ATIO
Overview
The LT3800 is a high input voltage range step-down
synchronous DC/DC converter controller IC that uses a
200kHz constant frequency, current mode architecture
with external N-channel MOSFET switches.
The LT3800 has provisions for high efficiency, low load
operation for battery-powered applications. Burst Mode
operation reduces total average input quiescent currents
to 100µA during no load conditions. A low current shutdown
mode can also be activated, reducing quiescent current to
<10µA. Burst Mode operation can be disabled if desired.
The LT3800 also employs a reverse-current inhibit feature, allowing increased efficiencies during light loads
through nonsynchronous operation. This feature disables
the synchronous switch if inductor current approaches
zero. If full time synchronous operation is desired, this
feature can be disabled.
Much of the LT3800’s internal circuitry is biased from an
internal linear regulator. The output of this regulator is the
VCC pin, allowing bypassing of the internal regulator. The
associated internal circuitry can be powered from the
output of the converter, increasing overall converter efficiency. Using externally derived power also eliminates the
IC’s power dissipation associated with the internal VIN to
VCC regulator.
Theory of Operation (See Block Diagram)
The LT3800 senses converter output voltage via the VFB
pin. The difference between the voltage on this pin and an
internal 1.231V reference is amplified to generate an error
voltage on the VC pin which is, in turn, used as a threshold
for the current sense comparator.
During normal operation, the LT3800 internal oscillator
runs at 200kHz. At the beginning of each oscillator cycle,
the switch drive is enabled. The switch drive stays enabled
until the sensed switch current exceeds the VC derived
threshold for the current sense comparator and, in turn,
disables the switch driver. If the current comparator
threshold is not obtained for the entire oscillator cycle, the
switch driver is disabled at the end of the cycle for 450ns.
This minimum off-time mode of operation assures regeneration of the BOOST bootstrapped supply.
Power Requirements
The LT3800 is biased using a local linear regulator to
generate internal operational voltages from the VIN pin.
Virtually all of the circuitry in the LT3800 is biased via an
internal linear regulator output (VCC). This pin is decoupled
with a low ESR 1µF capacitor to PGND.
The VCC regulator generates an 8V output provided there
is ample voltage on the VIN pin. The VCC regulator has
approximately 1V of dropout, and will follow the VIN pin
with voltages below the dropout threshold.
The LT3800 has a start-up requirement of VIN > 7.5V. This
assures that the onboard regulator has ample headroom
to bring the VCC pin above its UVLO threshold. The VCC
regulator can only source current, so forcing the VCC pin
above its 8V regulated voltage allows use of externally
derived power for the IC, minimizing power dissipation in
the IC. Using the onboard regulator for start-up, then
deriving power for VCC from the converter output maximizes conversion efficiencies and is common practice. If
VCC is maintained above 6.5V using an external source, the
LT3800 can continue to operate with VIN as low as 4V.
The LT3800 operates with 3mA quiescent current from the
VCC supply. This current is a fraction of the actual VCC
quiescent currents during normal operation. Additional
current is produced from the MOSFET switching currents
for both the boosted and synchronous switches and are
typically derived from the VCC supply.
Because the LT3800 uses a linear regulator to generate
VCC, power dissipation can become a concern with high
VIN voltages. Gate drive currents are typically in the range
of 5mA to 15mA per MOSFET, so gate drive currents can
create substantial power dissipation. It is advisable to
derive VCC and VBOOST power from an external source
whenever possible.
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LT3800
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APPLICATIO S I FOR ATIO
The onboard VCC regulator will provide gate drive power for
start-up under all conditions with total MOSFET gate charge
loads up to 180nC. The regulator can operate the LT3800
continuously, provided the VIN voltage and/or MOSFET gate
charge currents do not create excessive power dissipation
in the IC. Safe operating conditions for continuous regulator use are shown in Figure 1. In applications where these
conditions are exceeded, VCC must be derived from an
external source after start-up.
Charge Pump Doubler
VOUT
B0520
B0520
VCC
1µF
1µF
LT3800
Si1555DL
BG
Charge Pump Tripler
70
VOUT
B0520
60
1µF
B0520
VIN (V)
50
B0520
40
1µF
VCC
1µF
30
Si1555DL
Si1555DL
LT3800
SAFE
OPERATING
CONDITIONS
20
3800 AI01
BG
10
0
50
100
150
TOTAL FET GATE CHARGE (nC)
200
3800 F01
Figure 1. VCC Regulator Continuous Operating Conditions
In LT3800 converter applications with output voltages in
the 9V to 20V range, back-feeding VCC and VBOOST from
the converter output is trivial, accomplished by connecting diodes from the output to these supply pins. Deriving
these supplies from output voltages greater than 20V will
require additional regulation to reduce the feedback voltage. Outputs lower than 9V will require step-up techniques
to increase the feedback voltage to something greater than
the 8V VCC regulated output. Low power boost switchers
are sometimes used to provide the step-up function, but
a simple charge-pump can perform this function in many
instances.
Inductor Auxiliary Winding
TG
SW
LT3800
VCC
•
N
VOUT
•
BG
3800 AI04
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LT3800
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APPLICATIO S I FOR ATIO
Burst Mode
The LT3800 employs low current Burst Mode functionality
to maximize efficiency during no load and low load conditions. Burst Mode operation is enabled by shorting the
BURST_EN pin to SGND. Burst Mode operation can be
disabled by shorting BURST_EN to either VFB or VCC.
When the required switch current, sensed via the VC pin
voltage, is below 15% of maximum, the Burst Mode
operation is employed and that level of sense current is
latched onto the IC control path. If the output load requires
less than this latched current level, the converter will
overdrive the output slightly during each switch cycle.
This overdrive condition is sensed internally and forces
the voltage on the VC pin to continue to drop. When the
voltage on VC drops 150mV below the 15% load level,
switching is disabled and the LT3800 shuts down most of
its internal circuitry, reducing total quiescent current to
100µA. When the converter output begins to fall, the VC pin
voltage begins to climb. When the voltage on the VC pin
climbs back to the 15% load level, the IC returns to normal
operation and switching resumes. An internal clamp on
the VC pin is set at 100mV below the switch disable
threshold, which limits the negative excursion of the pin
voltage, minimizing the converter output ripple during
Burst Mode operation.
IL
PULSE SKIP MODE
During Burst Mode operation, VIN pin current is 20µA and
VCC current is reduced to 80µA. If no external drive is
provided for VCC, all VCC bias currents originate from the
VIN pin, giving a total VIN current of 100µA. Burst current
can be reduced further when VCC is driven using an output
derived source, as the VCC component of VIN current is
then reduced by the converter buck ratio.
Reverse-Current Inhibit
The LT3800 contains a reverse-current inhibit feature to
maximize efficiency during light load conditions. This
mode of operation allows discontinuous operation, and is
sometimes referred to as “pulse-skipping” mode. Refer to
Figure 2.
This feature is enabled with Burst Mode operation, and can
also be enabled while Burst Mode operation is disabled by
shorting the BURST_EN pin to VFB.
When reverse-current inhibit is enabled, the LT3800 sense
amplifier detects inductor currents approaching zero and
disables the synchronous switch for the remainder of the
switch cycle. If the inductor current is allowed to go
negative before the synchronous switch is disabled, the
switch node could inductively kick positive with a high
dv/dt. The LT3800 prevents this by incorporating a 10mV
positive offset at the sense inputs.
IL
FORCED CONTINUOUS
DECREASING
LOAD
CURRENT
3800 F02
Figure 2. Inductor Current vs Mode
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With the reverse-current inhibit feature enabled, an LT3800
converter will operate much like a nonsynchronous
converter during light loads. Reverse-current inhibit reduces resistive losses associated with inductor ripple
currents, which improves operating efficiencies during
light-load conditions.
An LT3800 DC/DC converter that is operating in reverseinhibit mode has a minimum load requirement of 1mA
(BURST_EN = VFB). Since most applications use outputgenerated power for the LT3800, this requirement is met
by the bias currents of the IC, however, for applications
that do not derive power from the output, this requirement is easily accomplished by using a 1.2k resistor
connected from VFB to ground as one of the converter
output voltage programming resistors (R1). There are no
minimum load restrictions when in Burst Mode operation
(BURST_EN < 0.5V) or continuous conduction mode
(BURST_EN > 2.5V).
Soft-Start
The LT3800 incorporates a programmable soft-start function to control start-up surge currents, limit output overshoot and for use in supply sequencing. The soft-start
function directly monitors and controls output voltage
slew rate during converter start-up.
As the output voltage of the converter rises, the soft-start
circuit monitors δV/δt current through a coupling capacitor and adjusts the voltage on the VC pin to maintain an
average value of 2µA. The soft-start function forces the
programmed slew rate while the converter output rises to
95% regulation, which corresponds to 1.185V on the VFB
pin. Once 95% regulation is achieved, the soft-start circuit
is disabled. The soft-start circuit will re-enable when the VFB
pin drops below 70% regulation, which corresponds to
300mV of control hysteresis on the VFB pin, which allows
for a controlled recovery from a ‘brown-out’ condition.
The desired soft-start rise time (tSS) is programmed via
a programming capacitor CSS1, using a value that
corresponds to 2µA average current during the soft-start
interval. This capacitor value follows the relation:
CSS1 =
2E –6 • tSS
VOUT
RSS is typically set to 200k for most applications.
CSS1
VOUT
A
LT3800
RSS
CSS
3800 AI06
Considerations for Low Voltage Output Applications
The LT3800 CSS pin biases to 220mV during the soft-start
cycle, and this voltage is increased at network node “A” by
the 2µA signal current through RSS, so the output has to
reach this value before the soft-start function is engaged.
The value of this output soft-start start-up voltage offset
(VOUT(SS)) follows the relation:
VOUT(SS) = 220mV + RSS • 2E–6
which is typically 0.64V for RSS = 200k.
In some low voltage output applications, it may be desirable to reduce the value of this soft-start start-up voltage
offset. This is possible by reducing the value of RSS. With
reduced values of RSS, the signal component caused by
voltage ripple on the output must be minimized for proper
soft-start operation.
Peak-to-peak output voltage ripple (∆VOUT) will be imposed on node “A” through the capacitor CSS1. The value
of RSS can be set using the following equation:
RSS =
∆VOUT
1.3E –6
It is important to use low ESR output capacitors for
LT3800 voltage converter designs to minimize this ripple
voltage component. A design with an excessive ripple
component can be evidenced by observing the VC pin
during the start cycle.
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Soft-Start Characteristic Showing Excessive Ripple Component
250µs/DIV
VOUT
VOUT
VOUT(SS)
V(VC)
VOUT(SS)
V(VC)
3800 AI07
The soft-start cycle should be evaluated to verify that the
reduced RSS value allows operation without excessive
modulation of the VC pin before finalizing the design.
If the VC pin has an excessive ripple component during the
soft-start cycle, converter output ripple should be reduced
or RSS increased. Reduction in converter output ripple is
typically accomplished by increasing output capacitance
and/or reducing output capacitor ESR.
External Current Limit Foldback Circuit
An additional start-up voltage offset can occur during the
period before the LT3800 soft-start circuit becomes active. Before the soft-start circuit throttles back the VC pin
in response to the rising output voltage, current as high as
the peak programmed current limit (IMAX) can flow in the
inductor. Switching will stop once the soft-start circuit
takes hold and reduces the voltage on the VC pin, but the
output voltage will continue to increase as the stored
energy in the inductor is transferred to the output capacitor. With IMAX flowing in the inductor, the resulting leading-edge rise on VOUT due to energy stored in the inductor
follows the relationship:
∆VOUT
⎛ L ⎞
= IMAX • ⎜
⎟
⎝ COUT ⎠
1/ 2
Desirable Soft-Start Characteristic
250µs/DIV
3800 AI08
Inductor current typically doesn’t reach IMAX in the few
cycles that occur before soft-start becomes active, but can
with high input voltages or small inductors, so the above
relation is useful as a worst-case scenario.
This energy transfer increase in output voltage is typically
small, but for some low voltage applications with relatively
small output capacitors, it can become significant. The
voltage rise can be reduced by increasing output capacitance, which puts additional limitations on COUT for these
low voltage supplies. Another approach is to add an
external current limit foldback circuit which reduces the
value of IMAX during start-up.
An external current limit foldback circuit can be easily
incorporated into an LT3800 DC/DC converter application
by placing a 1N4148 diode and a 47k resistor from the
converter output (VOUT) to the LT3800’s VC pin. This limits
the peak current to 0.25 • IMAX when VOUT = 0V. A current
limit foldback circuit also has the added advantage of
providing a reduced output current in the DC/DC converter
during short-circuit fault conditions, so a foldback circuit
may be useful even if the soft-start function is disabled.
If the soft-start circuit is disabled by shorting the CSS pin
to ground, the external current limit fold-back circuit must
be modified by adding an additional diode and resistor.
The 2-diode, 2-resistor network shown also provides
0.25 • IMAX when VOUT = 0V.
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Current Limit Foldback Circuit for
Applications That Use Soft-Start
Alternative Current Limit Foldback Circuit for Applications That
Have Soft-Start Disabled
VC
VC
1N4148
1N4148
1N4148
47k
39k
27k
VOUT
3800 AI10
3800 AI09
VOUT
Adaptive Nonoverlap (NOL) Output Stage
Shutdown
The FET driver output stages implement adaptive
nonoverlap control. This feature maintains a constant
dead time, preventing shoot-through switch currents,
independent of the type, size or operating conditions of the
external switch elements.
The LT3800 SHDN pin uses a bandgap generated reference threshold of 1.35V. This precision threshold allows
use of the SHDN pin for both logic-level controlled applications and analog monitoring applications such as power
supply sequencing.
Each of the two switch drivers contains a NOL control
circuit, which monitors the output gate drive signal of the
other switch driver. The NOL control circuits interrupt the
“turn on” command to their associated switch driver until
the other switch gate is fully discharged.
The LT3800 operational status is primarily controlled by a
UVLO circuit on the VCC regulator pin. When the IC is
enabled via the SHDN pin, only the VCC regulator is enabled. Switching remains disabled until the UVLO threshold is achieved at the VCC pin, when the remainder of the
IC is enabled and switching commences.
Antislope Compensation
Most current mode switching controllers use slope compensation to prevent current mode instability. The LT3800
is no exception. A slope-compensation circuit imposes an
artificial ramp on the sensed current to increase the rising
slope as duty cycle increases. Unfortunately, this additional ramp corrupts the sensed current value, reducing
the achievable current limit value by the same amount as
the added ramp represents. As such, current limit is
typically reduced as duty cycles increase. The LT3800
contains circuitry to eliminate the current limit reduction
typically associated with slope compensation. As the
slope-compensation ramp is added to the sensed current,
a similar ramp is added to the current limit threshold
reference. The end result is that current limit is not
compromised, so a LT3800 converter can provide full
power regardless of required duty cycle.
Because an LT3800 controlled converter is a power transfer device, a voltage that is lower than expected on the
input supply could require currents that exceed the sourcing capabilities of that supply, causing the system to lock
up in an undervoltage state. Input supply start-up protection can be achieved by enabling the SHDN pin using a
resistive divider from the VIN supply to ground. Setting the
divider output to 1.35V when that supply is at an adequate
voltage prevents an LT3800 converter from drawing large
currents until the input supply is able to provide the
required power. 120mV of input hysteresis on the SHDN
pin allows for almost 10% of input supply droop before
disabling the converter.
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Programming LT3800 VIN UVLO
VIN
LT3800
RB
3
SHDN
RA
SGND
17
3800 AI02
The UVLO voltage, VIN(UVLO), is set using the following
relation:
VIN(UVLO) – 1.35V
RA = RB •
1.35V
If additional hysteresis is desired for the enable function,
an external positive feedback resistor can be used from the
LT3800 regulator output.
The shutdown function can be disabled by connecting the
SHDN pin to VIN through a large value pull-up resistor.
This pin contains a low impedance clamp at 6V, so the
SHDN pin will sink current from the pull-up resistor (RPU):
V – 6V
ISHDN = IN
RPU
Because this arrangement will pull the SHDN pin to the 6V
clamp voltage, it will violate the 5V absolute maximum
voltage rating of the pin. This is permitted, however, as
long as the absolute maximum input current rating of 1mA
is not exceeded. Input SHDN pin currents of <100µA are
recommended; a 1MΩ or greater pull-up resistor is typically used for this configuration.
Inductor Selection
The primary criterion for inductor value selection in LT3800
applications is ripple current created in that inductor.
Basic design considerations for ripple current are output
voltage ripple, and the ability of the internal slope compensation waveform to prevent current mode instability. Once
the value is determined, an inductor must also have a
saturation current equal to or exceeding the maximum
peak current in the inductor.
Ripple current (∆IL) in an inductor for a given value (L) can
be approximated using the relation:
⎛ V ⎞ V
∆IL = ⎜ 1– OUT ⎟ • OUT
⎝
VIN ⎠ fO • L
The typical range of values for ∆I is 20% to 40% of
IOUT(MAX), where IOUT(MAX) is the maximum converter
output load current. Ripple currents in this range typically
yield a good design compromise between inductor performance versus inductor size and cost, and values in this
range are generally a good starting point. A starting point
inductor value can thus be determined using the relation:
⎛ VOUT ⎞
⎟
⎜ 1–
⎝
VIN ⎠
L = VOUT •
fO • 0.3 • IOUT(MAX)
Use of smaller inductors increase output ripple currents,
requiring more capacitance on the converter output. Also,
with converter operation with duty cycles greater than
50%, the slope compensation criterion, described later,
must be met. Designing for smaller ripple currents requires larger inductor values, which can increase converter cost and/or footprint.
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Some magnetics vendors specify a volt-second product in
their data sheet. If they do not, consult the vendor to make
sure the specification is not being exceeded by your
design. The required volt-second product is calculated as
follows:
Volt - Second ≥
VOUT
fO
⎛ V ⎞
• ⎜ 1 – OUT ⎟
⎝
VIN ⎠
Magnetics vendors specify either the saturation current,
the RMS current, or both. When selecting an inductor
based on inductor saturation current, the peak current
through the inductor, IOUT(MAX) + (∆I/2), is used. When
selecting an inductor based on RMS current the maximum
load current, IOUT(MAX), is used.
The requirement for avoiding current mode instability is
keeping the rising slope of sensed inductor ripple current
(S1) greater than the falling slope (S2). During continuouscurrent switcher operation, the rising slope of the current
waveform in the switched inductor is less than the falling
slope when operating at duty cycles (DC) greater than 50%.
To avoid the instability condition during this operation, a
false signal is added to the sensed current, increasing the
perceived rising slope. To prevent current mode instability, the slope of this false signal (Sx) must be sufficient such
that the sensed rising slope exceeds the falling slope, or
S1 + Sx ≥ S2. This leads to the following relations:
Sx ≥ S2 (2DC – 1)/DC
where:
S2 ~ VOUT/L
Solving for L yields a relation for the minimum inductance
that will satisfy slope compensation requirements:
2DC – 1
LMIN = VOUT •
DC • Sx
The LT3800 maximizes available dynamic range using a
slope compensation generator that continuously increases
the additional signal slope as duty cycle increases. The
slope compensation waveform is calibrated at an 80%
duty cycle, to generate an equivalent slope of at least
1E5 • ILIMIT A/sec, where ILIMIT is the programmed converter current limit. Current limit is programmed by using
a sense resistor (RS) such that ILIMIT = 150mV/RS, so the
equation for the minimum inductance to meet the current
mode instability criterion can be reduced to:
LMIN = (5E–5)(VOUT)(RS)
For example, with VOUT = 5V and RS = 20mΩ:
LMIN = (5E–5)(5)(0.02) = 5µH
After calculating the minimum inductance value, the voltsecond product, the saturation current and the RMS
current for your design, an off the shelf inductor can be
selected from a magnetics vendor. A list of magnetics
vendors can be found at http://www.linear.com/ezone/
vlinks or by contacting the Linear Technology Applications
department.
Output Voltage Programming
Output voltage is programmed through a resistor feedback network to VFB (Pin 6) on the LT3800. This pin is the
inverting input of the error amplifier, which is internally
referenced to 1.231V. The divider is ratioed to provide
1.231V at the VFB pin when the output is at its desired
value. The output voltage is thus set following the relation:
⎛V
⎞
R2 = R1 • ⎜ OUT – 1⎟
⎝ 1.231 ⎠
when an external resistor divider is connected to the
output as shown.
Programming LT3800 Output Voltage
VOUT
LT3800
R2
VFB
6
R1
SGND
17
3800 AI03
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Power MOSFET Selection
IMAIN = (ILOAD)(DC)
External N-channel MOSFET switches are used with the
LT3800. The positive gate-source drive voltage of the
LT3800 for both switches is roughly equivalent to the VCC
supply voltage, for use of standard threshold MOSFETs.
ISYNC = (ILOAD)(1 – DC)
Selection criteria for the power MOSFETs include the “ON”
resistance (RDS(ON)), total gate charge (QG), reverse transfer
capacitance (CRSS), maximum drain-source voltage (VDSS)
and maximum current.
The power FETs selected must have a maximum operating
VDSS exceeding the maximum VIN. VGS voltage maximum
must exceed the VCC supply voltage.
Total gate charge (QG) is used to determine the FET gate
drive currents required. QG increases with applied gate
voltage, so the QG for the maximum applied gate voltage
must be used. A graph of QG vs. VGS is typically provided
in MOSFET datasheets.
In a configuration where the LT3800 linear regulator is
providing VCC and VBOOST currents, the VCC 8V output
voltage can be used to determine QG. Required drive
current for a given FET follows the simple relation:
IGATE = QG(8V) • fO
QG(8V) is the total FET gate charge for VGS = 8V, and f0 =
operating frequency. If these currents are externally derived by backdriving VCC, use the backfeed voltage to
determine QG. Be aware, however, that even in a backfeed
configuration, the drive currents for both boosted and
synchronous FETs are still typically supplied by the LT3800
internal VCC regulator during start-up. The LT3800 can
start using FETs with a combined QG(8V) up to 180nC.
Once voltage requirements have been determined, RDS(ON)
can be selected based on allowable power dissipation and
required output current.
In an LT3800 buck converter, the average inductor current
is equal to the DC load current. The average currents
through the main (bootstrapped) and synchronous
(ground-referred) switches are:
The RDS(ON) required for a given conduction loss can be
calculated using the relation:
PLOSS = ISWITCH2 • RDS(ON)
In high voltage applications (VIN > 20V), the main switch
is required to slew very large voltages. MOSFET transition
losses are proportional to VIN2 and can become the
dominant power loss term in the main switch. This transition loss takes the form:
PTR ≈ (k)(VIN)2(ISWITCH)(CRSS)(fO)
where k is a constant inversely related to the gate drive
current, approximated by k = 2 in LT3800 applications,
and ISWITCH is the converter output current. The power
loss terms for the switches are thus:
PMAIN = (DC)(ISWITCH)2(1 + d)(RDS(ON)) +
2(VIN)2(ISWITCH)(CRSS)(fO)
PSYNC = (1 – DC)(ISWITCH)2(1 + d)(RDS(ON))
The (1 + d) term in the above relations is the temperature
dependency of RDS(ON), typically given in the form of a
normalized RDS(ON) vs Temperature curve in a MOSFET
data sheet.
The CRSS term is typically smaller for higher voltage FETs,
and it is often advantageous to use a FET with a higher VDS
rating to minimize transition losses at the expense of
additional RDS(ON) losses.
In some applications, parasitic FET capacitances couple
the negative going switch node transient onto the bottom
gate drive pin of the LT3800, causing a negative voltage in
excess of the Absolute Maximum Rating to be imposed on
that pin. Connection of a catch Schottky diode from this
pin to ground will eliminate this effect. A 1A current rating
is typically sufficient for the diode.
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Input Capacitor Selection
Output Capacitor Selection
The large currents typical of LT3800 applications require
special consideration for the converter input and output
supply decoupling capacitors. Under normal steady state
buck operation, the source current of the main switch
MOSFET is a square wave of duty cycle VOUT/VIN. Most of
this current is provided by the input bypass capacitor. To
prevent large input voltage transients and avoid bypass
capacitor heating, a low ESR input capacitor sized for the
maximum RMS current must be used. This maximum
capacitor RMS current follows the relation:
The output capacitor in a buck converter generally has
much less ripple current than the input capacitor. Peak-topeak ripple current is equal to that in the inductor (∆IL),
typically a fraction of the load current. COUT is selected to
reduce output voltage ripple to a desirable value given an
expected output ripple current. Output ripple (∆VOUT) is
approximated by:
IRMS =
I MAX ( VOUT ( VIN – VOUT ))
1
2
VIN
which peaks at a 50% duty cycle, when IRMS = IMAX/2.
The bulk capacitance is calculated based on an acceptable
maximum input ripple voltage, ∆VIN, which follows the
relation:
VOUT
VIN
CIN(BULK) = IOUT(MAX) •
∆VIN • fO
∆V is typically on the order of 100mV to 200mV. Aluminum electrolytic capacitors are a good choice for high
voltage, bulk capacitance due to their high capacitance per
unit area.
The capacitor voltage rating must be rated greater than
VIN(MAX). The combination of aluminum electrolytic capacitors and ceramic capacitors is a common approach to
meeting supply input capacitor requirements. Multiple
capacitors are also commonly paralleled to meet size or
height requirements in a design.
Capacitor ripple current ratings are often based on only
2000 hours (three months) lifetime; it is advisable to
derate either the ESR or temperature rating of the capacitor for increased MTBF of the regulator.
∆VOUT ≈ ∆IL(ESR + [(8)(fO) • COUT]–1)
where fO = operating frequency.
∆VOUT increases with input voltage, so the maximum
operating input voltage should be used for worst-case
calculations. Multiple capacitors are often paralleled to
meet ESR requirements. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering
and has the required RMS current rating. An additional
ceramic capacitor in parallel is commonly used to reduce
the effect of parasitic inductance in the output capacitor,
which reduces high frequency switching noise on the
converter output.
Increasing inductance is an option to reduce ESR requirements. For extremely low ∆VOUT, an additional LC filter
stage can be added to the output of the supply. Application Note 44 has information on sizing an additional
output LC filter.
Layout Considerations
The LT3800 is typically used in DC/DC converter designs
that involve substantial switching transients. The switch
drivers on the IC are designed to drive large capacitances
and, as such, generate significant transient currents themselves. Careful consideration must be made regarding
supply bypass capacitor locations to avoid corrupting the
ground reference used by IC.
Typically, high current paths and transients from the input
supply and any local drive supplies must be kept isolated
from SGND, to which sensitive circuits such as the error
amp reference and the current sense circuits are referred.
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Effective grounding can be achieved by considering switch
current in the ground plane, and the return current paths
of each respective bypass capacitor. The VIN bypass
return, VCC bypass return, and the source of the synchronous FET carry PGND currents. SGND originates at the
negative terminal of the VOUT bypass capacitor, and is the
small signal reference for the LT3800.
Don’t be tempted to run small traces to separate ground
paths. A good ground plane is important as always, but
PGND referred bypass elements must be oriented such
that transient currents in these return paths do not
corrupt the SGND reference.
During the dead-time between switch conduction, the
body diode of the synchronous FET conducts inductor
current. Commutating this diode requires a significant
charge contribution from the main switch. At the instant
the body diode commutates, a current discontinuity is
created and parasitic inductance causes the switch node
to fly up in response to this discontinuity. High currents
and excessive parasitic inductance can generate extremely
fast dV/dt rise times. This phenomenon can cause
avalanche breakdown in the synchronous FET body diode, significant inductive overshoot on the switch node,
and shoot-through currents via parasitic turn-on of the
synchronous FET. Layout practices and component orientations that minimize parasitic inductance on this node
is critical for reducing these effects.
such that current paths in the ground plane do not cross
through signal ground areas. Signal ground refers to the
Exposed Pad on the backside of the LT3800 IC. SGND is
referenced to the (–) terminal of the VOUT decoupling
capacitor and is used as the converter voltage feedback
reference. Power ground currents are controlled on the
LT3800 via the PGND pin, and this ground references the
high current synchronous switch drive components, as
well as the local VCC supply. It is important to keep PGND
and SGND voltages consistent with each other, so separating these grounds with thin traces is not recommended.
When the synchronous FET is turned on, gate drive surge
currents return to the LT3800 PGND pin from the FET
source. The BOOST supply refresh surge currents also
return through this same path. The synchronous FET must
be oriented such that these PGND return currents do not
corrupt the SGND reference. Problems caused by the
PGND return path are generally recognized during heavy
load conditions, and are typically evidenced as multiple
switch pulses occurring during a single 5µs switch cycle.
This behavior indicates that SGND is being corrupted and
grounding should be improved. SGND corruption can
often be eliminated, however, by adding a small capacitor
(100pF-200pF) across the synchronous switch FET from
drain to source.
Ringing waveforms in a converter circuit can lead to
device failure, excessive EMI, or instability. In many cases,
you can damp a ringing waveform with a series RC
network across the offending device. In LT3800 applications, any ringing will typically occur on the switch node,
which can usually be reduced by placing a snubber across
the synchronous FET. Use of a snubber network, however,
should be considered a last resort. Effective layout practices typically reduce ringing and overshoot, and will
eliminate the need for such solutions.
The high di/dt loop formed by the switch MOSFETs and the
input capacitor (CIN) should have short wide traces to
minimize high frequency noise and voltage stress from
inductive ringing. Surface mount components are preferred to reduce parasitic inductances from component
leads. Connect the drain of the main switch MOSFET
directly to the (+) plate of CIN, and connect the source of
the synchronous switch MOSFET directly to the (–) terminal of CIN. This capacitor provides the AC current to the
switch MOSFETs. Switch path currents can be controlled
by orienting switch FETs, the switched inductor, and input
and output decoupling capacitors in close proximity to
each other.
Effective grounding techniques are critical for successful
DC/DC converter layouts. Orient power path components
Locate the VCC and BOOST decoupling capacitors in close
proximity to the IC. These capacitors carry the MOSFET
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drivers’ high peak currents. Locate the small-signal components away from high frequency switching nodes (BOOST,
SW, TG, VCC and BG). Small-signal nodes are oriented on
the left side of the LT3800, while high current switching
nodes are oriented on the right side of the IC to simplify
layout. This also helps prevent corruption of the SGND
reference.
Connect the VFB pin directly to the feedback resistors independent of any other nodes, such as the SENSE– pin.
The feedback resistors should be connected between the
(+) and (–) terminals of the output capacitor (COUT).
Locate the feedback resistors in close proximity to the
LT3800 to minimize the length of the high impedance VFB
node.
The SENSE– and SENSE+ traces should be routed together and kept as short as possible.
The LT3800 packaging has been designed to efficiently
remove heat from the IC via the Exposed Pad on the
backside of the package. The Exposed Pad is soldered to
a copper footprint on the PCB. This footprint should be
made as large as possible to reduce the thermal resistance
of the IC case to ambient air.
Orientation of Components Isolates Power Path and PGND Currents,
Preventing Corruption of SGND Reference
VIN
BOOST
SW
TG
LT3800
VCC
SGND PGND
SW
BG
+
SGND
REFERRED
COMPONENTS
+
3800 AI05
VOUT
ISENSE
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19
LT3800
U
TYPICAL APPLICATIO S
6.5V-55V to 5V 10A DC/DC Converter with Charge Pump Doubler VCC Refresh and Current Limit Foldback
VIN
6.5V TO 55V
+
RA
1M
C7
1.5nF
R1
100k
1%
C2
1µF
100V
X7R ×3
C8
56µF
63V
×2
R2
309k
1%
VIN
BOOST
NC
TG
SW
CSS
NC
R4 75k
BURST_EN
R5
47k
L1
5.6µH
VCC
C3 1µF
16V X7R
BG
VC
C10
100pF
M1
Si7850DP
×2
LT3800
SHDN
VFB
R3
62k
C9
470pF
D1
BAS19
C1 1µF
16V X7R
M2
Si7370DP
×2
DS3
B160
×2
PGND
SENSE–
SENSE+
SGND
RS
0.01Ω
D2
1N4148
DS1
MBRO520L
DS2
MBRO520L
M3
1/2 Si1555DL
M4
1/2 Si1555DL
C4
1µF
C6
10µF
6.3V
X7R
+
VOUT
5V AT 10A
C5
220µF
×2
3800 TA02a
C5: SANYO POSCAP 6TP220M
L1: IHLP-5050FD-01
Efficiency and Power Loss
100
12
10
VIN = 24V
VIN = 13.8V
VIN = 48V
90
8
VIN = 55V
85
POWER LOSS
VIN = 48V
80
75
6
4
POWER LOSS
VIN = 13.8V
POWER LOSS (W)
EFFICIENCY (%)
95
2
70
0
0
2
4
6
IOUT (A)
8
10
3800 TA02b
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20
LT3800
U
TYPICAL APPLICATIO S
9V-38V to 3.3V 10A DC/DC Converter with Input UVLO and Burst Mode Operation
No Load I(VIN) = 100µA
+
C8
100µF
50V
×2
RA
RB 1M
187k
R3
82k
C2
330pF
C10
100pF
C3
100pF
BOOST
NC
TG
C5 1µF
16V X7R
C9
4.7µF
50V
X7R ×3
M1
Si7884DP
LT3800
C1 1nF
R1
100k
1%
VIN
R2
169k
1%
SHDN
SW
CSS
NC
R4 39k
BURST_EN
VFB
VCC
BG
VC
D1
MBR520
L1
3.3µH
C4 1µF
16V X7R
M2
Si7884DP
DS1
SS14
×2
PGND
SENSE–
SENSE+
SGND
RS
0.01Ω
C7
10µF
6.3V
X7R
C6: SANYO POSCAP 4TPD470M
L1: IHLP-5050FD-01
+
C6
470µF
×2
VOUT
3.3V AT 10A
3800 TA03a
Efficiency and Power Loss
92
7
VIN = 13.8V
90
6
88
5
86
4
84
3
82
2
80
1
78
0.1
1
POWER LOSS (W)
EFFICIENCY (%)
VIN
9V TO 38V
0
10
ILOAD (A)
3800 TA03b
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21
LT3800
U
TYPICAL APPLICATIO S
9V-38V to 5V 6A DC/DC Converter with All Ceramic Capacitors, Input UVLO,
Burst Mode Operation and Current Limit Foldback
VIN
9V TO 38V
C8
22µF
×3
RA
1M
C9 1nF
VIN
BOOST
NC
TG
R1
49.9k
1%
R2
154k
1%
SHDN
SW
CSS
NC
R4 51k
C6
47pF
BURST_EN
VFB
R5
D2
1N4148 47k
R3
27k
C2
1nF
M1
Si7884DP
LT3800
RB 187k
C1 3.9nF
C5 1µF
10V X7R
VCC
L1
10µH
C4 1µF
10V X7R
BG
VC
C3
100pF
D1
BAS19
M2
Si7884DP
DS1
SS14
PGND
SENSE–
SENSE+
SGND
RS
0.02Ω
C7
100µF
×2
C7: TDK C4532X5R0J107MT
C8: TDK C5750X7R1E226MT
L1: IHLP-5050FD-01
VOUT
5V AT 6A
3800 TA05a
Efficiency and Power Loss
3.20
VIN = 13.8V
95
2.80
90
2.40
85
2.00
80
1.60
75
1.20
70
0.80
65
0.40
60
0.001
POWER LOSS (W)
EFFICIENCY (%)
100
0
0.01
0.1
1
10
ILOAD (A)
3800 TA05b
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22
LT3800
U
PACKAGE DESCRIPTIO
FE Package
16-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1663)
Exposed Pad Variation BC
4.90 – 5.10*
(.193 – .201)
3.58
(.141)
3.58
(.141)
16 1514 13 12 1110
6.60 ±0.10
9
2.94
(.116)
4.50 ±0.10
6.40
2.94
(.252)
(.116)
BSC
SEE NOTE 4
0.45 ±0.05
1.05 ±0.10
0.65 BSC
1 2 3 4 5 6 7 8
RECOMMENDED SOLDER PAD LAYOUT
4.30 – 4.50*
(.169 – .177)
0.09 – 0.20
(.0035 – .0079)
0.50 – 0.75
(.020 – .030)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
3. DRAWING NOT TO SCALE
0.25
REF
1.10
(.0433)
MAX
0° – 8°
0.65
(.0256)
BSC
0.195 – 0.30
(.0077 – .0118)
TYP
0.05 – 0.15
(.002 – .006)
FE16 (BC) TSSOP 0204
4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LT3800
U
TYPICAL APPLICATIO
24V-48V to –12V 75W Inverting DC/DC Converter with VIN UVLO
VIN
24V TO 48V
+
R7
1M
C9
56µF
63V
×2
C10
1µF
100V
X7R
×4
D1
BAS19
R8
1M
VIN
2N3906
BOOST
NC
TG
R6
130k
R2
174k
1%
C1 1nF
SHDN
M1
FDD3570
C2 1µF
16V X7R
LT3800
R3
1M
D2
1N4148
SW
L1
15µH
RS
0.01Ω
R1 200k
CSS
NC
BURST_EN
VFB
R5
20k
1%
M2
FDD3570
BG
VC
R4
39k
C6
1µF
16V
X7R
VCC
C7
150pF
100V
DS1
B180
PGND
SENSE–
SENSE+
SGND
C3
470pF
C8
4.7µF
16V
X7R
C4
100pF
C5
270µF
16V
SPRAGUE SP VOUT
–12V
3800 TA04a
75W
+
L1: COEV MGPWL-00099
Efficiency and Power Loss
12
100
10
90
80
8
VIN = 48V
VIN = 36V
70
6
4
60
POWER LOSS (W)
EFFICIENCY (%)
VIN = 24V
VIN = 36V LOSS
2
50
40
0.1
1
ILOAD (A)
0
10
3800 TA04b
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1735
Synchronous Step-Down Controller
4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 6V, IOUT ≤ 20A
LTC1778
No RSENSETM Synchronous Step-Down Controller
Current Mode Without Using Sense Resistor, 4V ≤ VIN ≤ 36V
LT 1934
Micropower Step-Down Switching Regulator
3.2V ≤ VIN ≤ 34V, 300mA Switch, ThinSOTTM Package
LT1952
Synchronous Single Switch Forward Converter
25W to 500W Isolated Power Supplies, Small Size, High Efficiency
LT1976
60V Switching Regulator
3.2V ≤ VIN ≤ 60V, 1.5A Switch, 16-Lead TSSOP
LT3010
3V to 80V LDO
50mA Output Current, 1.275V ≤ VOUT ≤ 60V
LT3430/LT3431
3A, 60V Switching Regulators
5.5V ≤ VIN ≤ 60V, 200kHz, 16-Lead TSSOP
LTC3703
100V Synchronous Step-Down Controller
Large 1Ω Gate Drivers, No RSENSE
LTC3703-5
60V Synchronous Step-Down Controller
Large 1Ω Gate Drivers, No RSENSE
®
LTC3727-1
High VOUT 2-Phase Dual Step-Down Controller
0.8V ≤ VOUT ≤ 14V, PLL: 250kHz to 550kHz
LTC3728L
2-Phase, Dual Synchronous Step-Down Controller
550kHz, PLL: 250kHz to 550kHz, 4V ≤ VIN ≤ 36V
No RSENSE and ThinSOT are trademarks of Linear Technology Corporation.
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24
Linear Technology Corporation
LT 1007 REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
●
FAX: (408) 434-0507 ● www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2005