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by AN548A/D
SEMICONDUCTOR APPLICATION NOTE
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Prepared by: Glenn Young
power gain of 16 dB and a bandwidth (– 1 dB) of 8 MHz.
Overall efficiency is 48.5% and all harmonics are a minimum
of 20 dB below the fundamental output.
Sections on construction and device handling
considerations are also presented.
INTRODUCTION
This note uses a 25 watt UHF amplifier design as a vehicle
to discuss microstrip design techniques. The design
concentrates on impedance matching and microstrip
construction considerations. A basic knowledge of Smith
chart techniques is helpful in understanding this note.1
The amplifier itself, as shown in Figure 1, provides 25
watts of output power in the 450 – 512 MHz UHF band. It
is designed for 12.5 volt operation which makes it useful for
mobile transmitting equipment. A variety of police, taxi,
trucking and utility maintenance communication systems
operate in this band.
A summary of the performance of the completed amplifier
operating with a 12.5 volt supply at 512 MHz indicates a
MICROSTRIP DESIGN CONSIDERATIONS
Microstrip design was used for this amplifier due to its
inherent superiority over other methods at this frequency.
These techniques not only offer good compatibility with the
Motorola “stripline” package but they also offer very good
reproducibility. Microstrip construction is more efficient than
lumped constant equivalents since microstrip lines are less
lossy than lumped constant components.
!
!
!
!
Ω
Ω "'&
! All Microstrip lines 5.72 mm wide 2.5 cm long
C1,2,3
470 pF feedthru
C4,5,6
1.0 µ Tantalum
C7,8,9
0.1 µF Ceramic
C10,11,13,15,16,17 1.5-20 pF Compression Trimmer ARCO 420
C12,14
10 pF Microwave capacitor ATC type
100-B-10-M-MS or equivalent
L1, L2, L3 – 5 turns #20 Closewound 3/16” I.D.
L4, L5, L6 – 0.15 µh molded choke
L7, L8 – Ferroxcube VK 200 20/4B or equivalent
Ferrite beads are Ferroxcube 56 590 65/3B
or equivalent
Figure 1. Schematic Diagram of 25 W UHF Amplifier
REV 0
MOTOROLA
SEMICONDUCTOR APPLICATION INFORMATION
Motorola, Inc. 1993
1
Microstrip board with Teflon bonded fiberglass dielectric
rather than the higher dielectric constant ceramics was
chosen due to the ease of working with that type of material.
A substrate thickness of 1/16-inch is convenient since a line
of the same width as the transistor leads (0.225 inch)
produces a reasonable characteristic impedance (Zo) of
40.65 ohms. The value of the characteristic impedance is
calculated from:4
377h
Zo =
εr x W 1 + 1.735
εr–.0724
W –.836
h = 62 – (2 x 1.4) = 59.2 mils
(2)
1 oz. copper = 1.4 mils thick
The effective width should be used when the conductor is of
finite thickness.
t
π
ln 2h
t
+ 1
(3)
where t = thickness of the conductor
Weff = 225 + (1.4/π) ln
2 x 59.2
1.4
2N5946
+ 1 = 227.4 mils
1.3 + j1.2 ohms
4.2 – j0.5 ohms
Zin
Zout
2N6136
Zin
Zout
h
εr = dielectric constant
W = width of microstrip line
h = thickness of the dielectric
The h term is equal to the total thickness of the microstrip
board minus the thickness of the copper on both sides. In
this design that term is equal to
1.3 + j1.5 ohms
4.6 – j5.4 ohms
Zin
Zout
(1)
where
Weff = W +
2N5945
1.3 + j4.11 ohms
3.2 + j1.96 ohms
Figure 2. Transistor Complex Input and Output
Impedance at 470 MHz (Series Form)
Smith chart techniques are used to synthesize the
matching networks in the amplifier to be described. The
complex series equivalent input and output impedances as
taken from the data sheets are shown in Figure 2. There
are an infinite number of solutions to the required matching
networks, however, once an initial choice of one of the
components is made, only one solution exists. It is obvious
that all components need to be kept within reasonable limits,
however it would seem that the most critical parameter is
the length of the microstrip line. Using this assumption, the
length of the line is chosen as a starting point. The input
network, shown in Figure 3 will be solved to illustrate the
technique.
(4)
therefore:
377 x .0592
Zo =
2.5 x .2274 1 + 1.735 x 2.5–.0724 x
= 40.65 Ω
227.4 –.836
-
59.2
- (5)
THE AMPLIFIER DESIGN
The first decision in the design was determining the type
of matching networks to be used. The network shown in
Figure 3 was chosen because of its ability to “map” a large
area of complex impedances; this allows a good tuning
margin to compensate for normal variations in transistor
impedances and other peripheral effects. A side benefit of
this network is that the series tuning element provides the
dc blocking function, eliminating the need for coupling
capacitors.
The synthesis of the matching networks utilizes the large
signal impedances of the transistors as specified on the data
sheets. These parameters should not be confused with small
signal 2-port parameters. A complete discussion of large
signal characterization is given in Motorola Application note
AN-282A. The impedance parameters used in this note are
taken from the respective data sheets and were obtained
in the manner described in AN282A.
2
L1 = Microstrip Line 5.72 mm wide 2.5 cm long
CS = 4.08 pF
CP = 16.84 pF
Figure 3. Equivalent Circuit of Input Network
Before proceeding to determine the component values,
the effective wavelength of the desired frequency in the
microstrip line must be known. This is accomplished by first
finding λo, the wavelength in free space:
λo =
c
freq
=
3 x 108
4.7 x 108
= 0.638 meters
(6)
where c = propagation constant, free space
The TEM mode wavelength is determined:
λTEM = λo/(εr)1/2 = 63.8 cm/(2.5)1/2 = 40.37 cm
(7)
MOTOROLA SEMICONDUCTOR APPLICATION INFORMATION
Now as the propagation in microstrip line is not pure TEM
mode, a correction factor must be applied to the last
calculation.4
K=
εr
1 + 0.63 (εr – 1)
1/2
W .1225
h
A source impedance of 50 + j0 is normalized to 1.23 +
j0 and a load impedance of 1.3 + j1.5 is normalized to 0.032
+ j0.0369. The load impedance is plotted at point A in Figure
4 and the source impedance at point F. An arbitrary choice
of 2.5 cm for the line length was made. This is an electrical
length of:
electrical length = line length/λ′
=
2.5
1 + 0.63 (2.5 – 1) (227.4/59.2).1225
1/2
= 1.086
Then:
λ′ = (λtem) (K) = (40.37) (1.086) = 43.85 cm
= 2.5 cm/43.85 cm = 0.057 λ
(8)
(9)
This is the effective wavelength and will be used in all further
calculations. Equation 8 is valid for width to height ratios of
0.6:1 or greater. For ratios less than 0.6:1 alter the (w/h)
factor in the denominator to (w/h).0297.
The source and load impedances must now be
normalized to the 40.65Ω characteristic impedance of the
line and plotted on the Smith chart. It should be noted that
the terms “source” and “load” are used here only in reference
to the Smith chart solution.
(10)
Point A is rotated on a constant VSWR circle 0.057 λ toward
the generator to point B. Reactance must now be added in
parallel with the impedance presented at the end of the line
just plotted. As parallel additions are more easily handled in
admittance form, point B is converted to an admittance by
rotating it one-quarter wavelength on the same constant
VSWR circle. This results in point C in Figure 4. The constant
conductance circle that point C lies on is noted to be 0.23.
The problem now is to move along this circle towards the
generator until the reciprocal of the constant resistance circle
of the source impedance is intercepted. This circle does not
exist on a standard Smith chart and must be constructed.
CHART NOT AVAILABLE ELECTRONICALLY
Figure 4. Smith Chart Solution
MOTOROLA SEMICONDUCTOR APPLICATION INFORMATION
3
This is done by determining the radius of the constant
resistance circle representing the real part of the source
impedance and then constructing a circle of equal radius with
its center on the real axis and its circumference tangent to
the outer radius of the chart at zero resistance. When this
is done the intercept with the 0.23 constant real circle is seen
to lie at point D. The amount of parallel susceptance needed
to move from point C to point D is:
CONSTRUCTION CONSIDERATIONS
As in all RF power applications, solid emitter grounds are
imperative. In microstrip amplifiers gain can be increased
more than 1 dB by grounding both of the emitter leads to
the bottom foil of the microstrip board by wrapping strips of
copper foil through the transistor mounting hole as shown
in Figure 6.
BCP = (BC – BD) (Yo) =
(2.4 – 0.38) (24.6) = 49.72 mmhos
(11)
This is a parallel capacitance of:
CP = BCP/2πf = 49.72/(2π) (470 x 106) = 16.84 pF
(12)
All that remains to finish the solution is to determine the
amount of reactance necessary to reach the source at point
F. To do this, it is first necessary to transpose point D, which
is an admittance, to an impedance. This is accomplished by
rotating point D one-quarter wavelength on a constant VSWR
circle. This moves point D to point E which is on the 2.04
reactance line thus representing a series reactance of:
XCS = (XE) • (Zo) = (2.04) • (40.65) = 82.9 ohms
(13)
A series capacitance with this reactance is:
CS =
1
(2π) (f) (XCS)
=
1
(2π) (470 x 106) (82.9)
Figure 6. Proper Emitter Grounding Method
= 4.08 pF
(14)
This completes the solution for the input network.
The interstage networks as well as the output network
are solved in similar fashion with the following differences.
In the case of the interstage networks when the imaginary
term of the source impedance is other than zero, point F
would be plotted at the complex conjugate of the source
impedance. In the output network solution the “source” is
the output load of the amplifier (50 + j0) and the “load” is
the collector impedance of the output device.
Figure 5 gives details on the performance of the
completed amplifier. The use of the porcelain dielectric chip
capacitors for the series elements in the interstage networks
was found to provide an additional 2.5 to 3.0 dB of gain over
that obtained with compression trimmers as well as reducing
the number of tuning adjustments necessary.
450 MHz
480 MHz
512 MHz
Power Gain
18 db
17.2 db
16 db
Bandwidth (– 1 db)
5 MHz
6 MHz
8 MHz
Overall Efficiency
44.5%
46.5%
48.5%
Harmonics
Stability
Power Output
Burnout
All Harmonics Better Than – 20 db
Amplifier Stable under all Conditions of
Drive Down to VCC = 5.0 volts
25 W
25 W
25 W
No Damage to any Transistor with
Load Open & Shorted with 0 to
± 180° Phase Angle
Figure 5. Typical Performance Specifications
4
Stability under normal operating conditions is essential,
however, stability should be maintained over as wide a range
of supply voltage and drive levels as possible. If amplifier
stability is maintained at all RF drive levels with the supply
voltage reduced to between three and five volts, the designer
can be practically certain that the amplifier will remain stable
under all conditions of load. Maintaining stability is a key
factor in protecting these transistors from damage. In a stable
amplifier that has adequate heat sinking, these transistors
will withstand high VSWR loads including open and shorted
loads without damage. The major controlling factors in
obtaining wide range stability are:
1) Mechanical layout: Good mechanical layout includes
good emitter grounds (as previously described), compact
layout and short ground paths.
2) Biasing: The devices are all zero biased for Class “C”
operation. The use of relatively low Q base chokes with
ferrite beads on the ground side will maintain good base
circuit stability. In some applications, the use of a resistor
in series with the ground side of the base chokes on the
output and driver stages may enhance the stability.
Approximate values of these resistors should be 10 ohms,
1/2 watt for the driver and 1.0 ohms, 1/2 watt for the output
device. The addition of these series resistors will cause a
slight loss in gain; (about 0.1 to 0.2 dB overall).
3) Collector supply feed method: The collector supply feed
system is designed to provide decoupling at or near the
operating frequency and a low collector load impedance at
frequencies much lower than the operating frequency.
4) Heat sinking: In order to protect against burnout under
all conditions of load, adequate heat-sinking must be
provided. In heat sinking the device it is imperative to use
a good grade of thermal compound, such as Dow-Corning
340, on the interface between the device and its heat sink.
Figure 7 shows the microstrip board layout while Figure 8
is a photo of the completed amplifier.
MOTOROLA SEMICONDUCTOR APPLICATION INFORMATION
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Figure 7. Microstrip Board Layout
type can crack or dislodge the cap. This type stress
sometimes occurs due to adverse tolerance build-up in
dimensions when the device is mounted through a microstrip
board onto a heat sink. Many times this type of stress is
applied even in the most carefully thought out designs due
to solder build-up on the copper foil when a device is
replaced. In device replacement care should be taken to flow
all solder away from the mounting area before the stud nut
is torqued. Finally, one must be sure to torque the stud nut
before soldering the device leads. Refer to Motorola
Application Note AN-555 for details on mounting Motorola
“stripline packaged transistors.”
Figure 8. Photograph of Amplifier
DEVICE HANDLING CONSIDERATIONS
Although the Motorola stripline package is a rugged
assembly, some care in its handling should be observed. The
most important mechanical parameter is stud-torque,
specified on the data sheet at 6.5 inch-pounds maximum.
This data sheet specification is an absolute maximum and
should not be exceeded under any circumstances. A good
limit to use in production assembly is 6 inch-pounds and if
for any reason repeated assembly/disassembly is required
torque should be limited to 5 inch-pounds.
Another major precaution to observe is to avoid upward
pressure on the leads near the case body. Stresses of this
REFERENCES
1. P. H. Smith, “Electronic Applications of the Smith Chart”,
McGraw-Hill, 1969.
2. H. A. Wheeler, “Transmission-Line Properties of Parallel
Wide Strips by a Conformal-Mapping Approximation” IEEE
Trans. Microwave Theory and Techniques Vol. MTT-12,
May 1964.
3. H. A. Wheeler, “Transmission-Line Properties of Parallel
Strips Separated by a Dielectric Sheet” IEEE Trans.
Microwave Theory and Techniques Vol. MTT-3, March
1965.
4. H. Sobol “Extending Microwave and Technology to
Microwave Equipment” Electronics, March 20, 1967.
5. Microwave Engineers Technical and Buyers Guide, Edition
of the Microwave Journal, 1969.
MOTOROLA SEMICONDUCTOR APPLICATION INFORMATION
5
NOTES
6
MOTOROLA SEMICONDUCTOR APPLICATION INFORMATION
NOTES
MOTOROLA SEMICONDUCTOR APPLICATION INFORMATION
7
Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty, representation or guarantee regarding
the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit,
and specifically disclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters can and do vary in different
applications. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. Motorola does
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systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of
the Motorola product could create a situation where personal injury or death may occur. Should Buyer purchase or use Motorola products for any such
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against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death
associated with such unintended or unauthorized use, even if such claim alleges that Motorola was negligent regarding the design or manufacture of the part.
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MOTOROLA SEMICONDUCTOR APPLICATION INFORMATION
8
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AN548A/D