LINER LTC3444 Micropower synchronous buck-boost dc/dc converter for wcdma application Datasheet

LTC3444
Micropower Synchronous
Buck-Boost DC/DC Converter
for WCDMA Applications
DESCRIPTIO
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FEATURES
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The LTC®3444 is a highly efficient, fixed frequency, buckboost DC/DC converter, which operates from input voltages above, below, and equal to the output voltage. The
topology incorporated in the IC provides a continuous
transfer function through all operating modes, making the
product ideal for a single Lithium-Ion or multi-cell
applications where the output voltage can vary over a wide
range.
Optimized Features for WCDMA Handsets
Regulated Output with Input Voltages
Above, Below, or Equal to the Output
0.5V to 5V Output Range
Up to 400mA Continuous Output Current From
a Single Lithium-Ion Cell
Minimal External Components
1.5MHz Fixed Frequency Operation
Internal Loop Compensation for Fast Response
<25μs Full Scale Output Slewing; COUT 4.7μF
Output Disconnect in Shutdown
2.75V to 5.5V Input
<1μA Shutdown Current
Internal Soft-Start
Output Overvoltage Protection
Single Inductor, No Schottky Diodes Required
Small, Thermally Enhanced 8-Lead (3mm × 3mm)
DFN Package
The LTC3444 has been optimized for use in 3G WCDMA
applications. A unique design yields high efficiency at
very low output voltages while also eliminating external
components. The high speed error amplifier provides the
fast transient response required to slew the RF power
amplifier from standby to transmit and transmit to stand
by power levels. Output overvoltage protection protects
the RF power amplifier.
Operating frequency is internally set to 1.5MHz to minimize external component size while maximizing efficiency.
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APPLICATIO S
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WCDMA Applications–3G Handsets with High Speed
Data Rate Capability
MP3 Players
Digital Cameras
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
Protected by U.S. Patents including 6404251, 6166527.
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Other features include <1μA shutdown current, internal
soft-start, peak current limit and thermal shutdown. The
LTC3444 is available in a small, thermally enhanced
8-lead (3mm × 3mm) DFN package.
TYPICAL APPLICATIO
2.2μH
LTC3444 Dynamic Response
VOUT
0.8V TO 4.2V
LTC3444
340k
SW1
SW2
VIN
VOUT
3.1V TO 4.2V
FB
GND
VC
VOUT
VCONTROL
4.7μF
Li-Ion
1V/DIV
+
SHDN
4.7μF
205k
267k
10μs/DIV
VIN = 3.6V, VOUT = 0.8V TO 4.2V
VCONTROL = 2.36V TO 0.28V, ILOAD = 100mA
VCONTROL
DAC
3444 G16a
3444 TA01
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LTC3444
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ABSOLUTE
RATI GS
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PACKAGE/ORDER I FOR ATIO
(Note 1)
VIN,VOUT Voltages .......................................... –0.3 to 6V
SW1,SW2 Voltages DC .................................. –0.3 to 6V
Pulsed <100ns ............... –0.3 to 7V
SHDN Voltage ................................................ –0.3 to 6V
Operating Temperature (Note 2) .............. –40°C to 85°C
Maximum Junction Temperature (Note 4) ............ 125°C
Storage Temperature Range .................. –65°C to 125°C
TOP VIEW
SHDN 1
8
SW1 2
7
VC
6
VIN
5
VOUT
GND 3
SW2 4
9
ORDER PART
NUMBER
FB
LTC3444EDD
DD PACKAGE
8-LEAD (3mm × 3mm) PLASTIC DFN
DD PART MARKING
TJMAX = 125°C, θJA = 43°C/W,
4-LAYER BOARD θJC = 2.96°C/W
EXPOSED PAD IS GND (PIN 9)
MUST BE SOLDERED TO PCB
LBVZ
Order Options Tape and Reel: Add #TR
Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF
Lead Free Part Marking: http://www.linear.com/leadfree/
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.
VIN = VOUT = 3.6V unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
2.55
2.65
2.75
V
5
V
1.22
1.25
V
1
50
nA
Input Start-Up Voltage
●
Output Voltage Adjust Range
●
0.5
Feedback Voltage
●
1.19
Feedback Input Current
VFB = 1.22V
Quiescent Current - Shutdown
SD = 0V, VOUT = 0V Not Including Switch Leakage
0.1
1
μA
Quiescent Current - Active
(Note 3)
700
1100
μA
NMOS Switch Leakage
Switches B and C
0.1
7
μA
PMOS Switch Leakage
Switches A and D
0.1
10
μA
NMOS Switch On Resistance
Switches B and C
0.19
Ω
PMOS Switch On Resistance
Switches A and D
0.22
Ω
PMOS Switch On Resistance
Switch D VIN = 3.6, VOUT = 1V
0.4
Ω
3.5
A
Input Current Limit
●
2.5
Reverse Current Limit
●
3
●
●
70
100
Max Duty Cycle
Boost (%Switch C On)
Buck (% Switch A On)
Min Duty Cycle
●
Frequency Accuracy
●
VC = 1.5V, FB = 0V
Error Amp Sink Current
VC = 1.5V, FB = 1.5V
Internal Soft-Start Time
SHDN Going High
Output OV Threshold
82
%
%
0
1.2
Error Amp AVOL
Error Amp Source Current
A
1.5
1.8
5.1
MHz
65
dB
8
μA
230
μA
μs
250
●
%
5.3
5.5
V
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LTC3444
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.
VIN = VOUT = 3.6V unless otherwise noted.
PARAMETER
CONDITIONS
SHDN Threshold (On)
IC is Enabled
●
MIN
SHDN Threshold (Off)
IC is Disabled
●
TYP
MAX
UNITS
1.4
V
0.4
V
SHDN Input Current
VSHDN = 3.6V
0.01
1
μA
VC Output Current
VC = GND
0.5
2
μA
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3444E is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 3: Current measurements are performed when the outputs are not
switching.
Note 4: This IC includes overtemperature protection that is intended to
protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature is active.
Continuous operation above the specified maximum operating junction
temperature may result in device degradation or failure.
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TYPICAL PERFOR A CE CHARACTERISTICS
Li-Ion to 1V Efficiency
Li-Ion to 3.3V Efficiency
90
VOUT = 1.0V
80
75
IOUT = 65mA
70
IOUT = 50mA
60
0.10
40
0.08
VIN = 3.6V
0.06
VIN = 4.4V
PLOSS
10
60
3.1
3.3
3.5
3.7 3.9
VIN (V)
4.1
4.5
4.3
VIN = 3.1V
0
1
3444 G03
0.25
40
0.20
30
0.15
VIN = 4.4V
10
PLOSS
1
10
100
OUTPUT CURRENT (mA)
0.10
0.05
VIN = 3.1V
0
30
0.04
20
0.02
10
PLOSS
VIN = 3.1V
1
10
100
OUTPUT CURRENT (mA)
0
1000
3444 G04
0
1000
3444 G05
Operating Frequency
1.8
1.7
15
13
11
9
7
0.05
VIN = 3.6V
0
FREQUENCY (MHz)
0.30
E/A SOURCE CURRENT (μA)
EFFICIENCY (%)
60
POWER LOSS (W)
0.35
20
0.10
40
17
0.40
VIN = 4.4V
0.15
VIN = 4.4V
50
19
0.45
70
50
0.20
VIN = 3.6V
3444 G06
0.50
VIN = 3.1V
80
60
Error Amp Source Current
VIN = 3.6V
90
VIN = 3.1V
70
0
1000
10
100
OUTPUT CURRENT (mA)
Li-Ion to 4.2V Efficiency
100
0.12
VIN = 4.4V
30
80
0.14
50
20
65
90
0.25
POWER LOSS (W)
EFFICIENCY (%)
70
0.16
POWER LOSS (W)
EFFICIENCY (%)
IOUT = 100mA
VIN = 3.1V
0.18
100
EFFICIENCY (%)
Efficiency vs VIN
85
80
(TA = 25°C unless otherwise specified)
–25
1.5
1.4
1.3
VC = 1V
FB = 0V
5
–55
1.6
35
65
5
TEMPERATURE (°C)
95
125
3444 G07
1.2
–55
–25
0
35
65
TEMPERATURE (°C)
95
125
3444 G08
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LTC3444
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TYPICAL PERFOR A CE CHARACTERISTICS
PMOS RDS(ON)
(TA = 25°C unless otherwise specified)
Boost Maximum Duty Cycle
NMOS RDS(ON)
0.30
0.30
0.25
0.25
90
0.20
85
SWITCH C
0.20
0.15
DUTY CYCLE (%)
RDS(ON) (Ω)
RDS(ON) (Ω)
SWITCH B
75
0.15
0.10
–55
–25
5
35
65
TEMPERATURE (°C)
95
0.10
–55
125
–25
5
35
65
TEMPERATURE (°C)
95
3444 G09
800
370
360
95
700
650
600
1.23
1.22
1.21
1.20
500
–55
125
125
1.24
550
5
35
65
TEMPERATURE (°C)
95
1.25
VIN = VOUT = 3.6V
FEEDBACK VOLTAGE (V)
380
5
35
65
TEMPERATURE (°C)
Feedback Voltage
750
VIN + VOUT CURRENT (μA)
E/A SINK CURRENT (μA)
VIN = VOUT = 3.6V
VC = 2V, FB = 3.6V
390
–25
3444 G11
Active Quiescent Current
400
–25
70
–55
125
3444 G10
Error Amp Sink Current
350
–55
80
–25
5
65
35
TEMPERATURE (°C)
95
3444 G12
125
3444 G13
1.19
–55
–25
5
35
65
TEMERATURE (°C)
95
125
3444 G14
Minimum Start Voltage
2.85
2.80
START VOLTAGE (V)
2.75
2.70
2.65
2.60
2.55
2.50
2.45
2.40
–55
–25
5
35
65
TEMPERATURE (°C)
95
125
3444 G15
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LTC3444
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PI FU CTIO S
SHDN (Pin 1): Shutdown Function. A logic low input shuts
down the IC. A logic high input enables the IC and starts
the internal soft-start function by limiting the rise time of
the internal PWM command.
VOUT (Pin 5): Output of the Synchronous Rectifier. A filter
capacitor is placed from VOUT to GND. A ceramic bypass
capacitor is recommended as close to the VOUT and GND
pins as possible.
SW1 (Pin 2): Switch Pin Where the Internal Switches A
and B are Connected. Connect inductor from SW1 to SW2.
An optional Schottky diode can be connected from ground
to SW1 for a moderate efficiency improvement. Minimize
trace length to minimize EMI.
VIN (Pin 6): Input Supply Pin. Internal VCC for the IC. A
4.7μF ceramic capacitor is recommended as close to VIN
and GND as possible.
GND (Pin 3): Ground Pin for the IC.
SW2 (Pin 4): Switch Pin Where the Internal Switches C
and D are Connected. An optional Schottky diode can be
connected from SW2 to VOUT for a moderate efficiency
improvement. Minimize trace length to keep EMI down.
VC (Pin 7): Error Amp Output. Pull VC to ground to select
internal loop compensation. External compensation may
be connected from VC to FB. Internal compensation will be
disabled if VC is tied to an external compensation network.
FB (Pin 8): Feedback Pin. Connect resistive divider tap
here. The output voltage can be adjusted from 0.5V to 5V.
The feedback reference voltage is typically 1.22V.
GND (Pin 9, Exposed Pad): Solder to Board GND.
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LTC3444
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BLOCK DIAGRA
SW1
SW2
2
2.75V TO 5.5V
VIN
4
A
VOUT
D
6
B
VOUT
5
3A
GATE DRIVERS
AND
ANTI-CROSS
CONDUCTION
C
PEAK
REVERSE
CURRENT
LIMIT
OUTPUT OV
PWM LOGIC
AND
OUTPUT PHASING
335k
+
–
1.22V
100k
+
–
3.5A
+
–
PEAK
CURRENT
LIMIT
1.8V
PWM
COMPARATORS
+
–
+
–
2.65V
EA
UVLO
+
–
1.22V
INTERNAL
COMPENSATION
THERMAL
SHUTDOWN
GND = INTERNAL COMP
FLOAT = EXTERNAL COMP
OSC
SOFTSTART
THERMAL
SHUTDOWN
GND
3
FB
8
UVLO
VC
7
1
INTERNAL
SOFTSTART
VIN
SHDN
VCONTROL
2.5A
+
–
OUTLOW
+
–
+
INPUT
CURRENT
LIMIT
3444 BD
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LTC3444
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OPERATIO
The LTC3444 is a highly efficient, fixed frequency, buckboost DC/DC converter, which operates from input voltages above, below, and equal to the output voltage. The
topology incorporated in the IC provides a continuous
transfer function through all operating modes, making the
product ideal for single Lithium-Ion or multi-cell applications where the output voltage can vary over a wide range.
The LTC3444 is designed to provide dynamic voltage
control in space constrained 3G WCDMA applications.
Due to the high operating frequency and integrated loop
compensation a complete WCDMA application requires
only six additional components; input and output capacitors (ceramic), an inductor, and three resistors. The high
speed error amplifier and integrated loop compensation
provide the fast transient response required to slew the RF
power amplifier’s voltage rail from standby to transmit
and transmit to standby levels in < 25μs while minimizing
output overshoot or undershoot.
Efficiency under low output voltage conditions
(standby mode) is improved by using an N-channel
MOSFET in parallel to P-channel MOSFET switch D.
This parallel MOSFET eliminates the need for an external
Schottky. Output overvoltage protection protects the RF
power amplifier from voltages greater than 5.5V.
When used with the proper inductance and output capacitance, the LTC3444 internal compensation is designed to
be consistent with the transient requirements of a typical
WCDMA application. External compensation can be used
with other combinations of inductance and output capacitance, however, the transient response may not be
consistent with typical WCDMA requirements.
Output voltage programming is accomplished via a summing resistor input to the feedback resistive divider string.
The output voltage varies inversely with the command
voltage. When using the internal loop compensation,
resistor R1 in the feedback resistive divider string must be
340k. There are no constraints on R1 when using external
compensation. However, lower value resistors will decrease the resistance value required for programming the
output voltage. Care must be taken not to load down the
control voltage source.
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LTC3444
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OPERATIO
Error Amp
Internal Current Limit
The LTC3444 error amplifier is a voltage mode amplifier.
The internal loop compensation is designed to optimize
transient response to control input change when the
proper output L-C and R1 values are used. Refer to
Figure 1.
There are two different current limit circuits in the LTC3444.
The two circuits have internally fixed thresholds.
The first circuit sources current out of the FB pin to drop
the output voltage once the peak input current exceeds
2.5A minimum. During conditions where VOUT is near
ground, such as during a short circuit or during startup,
this threshold is cut in half, providing current foldback
protection.
Internal loop compensation is selected by grounding the
VC pin. The loop is designed to exhibit a single pole roll-off
(–20dB/dec) with a crossover frequency of ~100KHz.
External compensation can be used by connecting the
compensation components from FB to VC. The VC pin
must be allowed to float when using external compensation. If external compensation is used the internal compensation is automatically disabled. A Type III compensation network is typically required to meet the output
transient requirements of WCDMA.
The second circuit is a high-speed peak current limit
amplifier that shuts off P-channel MOSFET switch A if the
input current exceeds 3.5A typical. The delay to output for
this amplifier is typically 50ns.
During start-up, the ramp rate of the error amp output is
controlled to provide a soft-start function. Refer to
Figure 2.
VOUT
ERROR AMP
20μA
TO PWM
COMPARATORS
+
–
VC
R1
1.22V
FB
8
R3
VCONTROL
R2
INTERNAL
COMPENSATION
NETWORK
INT
ON
VOUT
VIN
0.5μA
VC
7
GND = INTERNAL
OPEN = EXTERNAL
3444 F01
Figure 1. Error Amplifier with Compensation Select Function
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LTC3444
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OPERATIO
Reverse Current Limit
The LTC3444 always operates in forced continuous conduction mode. The reverse current limit amplifier monitors the inductor current from the output through switch
D. Once the negative inductor current exceeds 3A minimum, the LTC3444 will shut off switch D. The high reverse
current is required to meet the transient slew requirements for WCDMA power amplifiers.
Output Overvoltage Protection
The LTC3444 provides output overvoltage protection. If
the output voltage exceeds 5.3V typical, P-channel MOSFET
switches A and D are turned off and N-channel MOSFET
switches B and C are turned on. Normal switching will
VIN
resume once the output voltage drops below ~5.1V. If the
condition which caused the output overvoltage is still
present the output will charge up to 5.3V again and the
overvoltage cycle will be repeat. Normal output regulation
will resume once the condition responsible for the output
overvoltage is removed.
Soft-Start
The soft-start function is initiated when the SHDN pin is
brought above 1.4V and the LTC3444 is out of UVLO
(above minimum input operating specs). The LTC3444 is
enabled but the PWM duty cycle is clamped via the error
amp output. The soft-start time is internally set to 250μs
to minimize output overshoot. A detailed diagram of this
function is shown in Figure 2.
ERROR AMP
20μA
SOFT-START
CLAMP
+
–
1.22V
FB
VC
8
7
TO PWM VCI
COMPARATORS
ISS
+
–
CSS
SHDN
1
1V
3444 F02
Figure 2. Soft-Start Circuitry
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LTC3444
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OPERATIO
Buck-Boost Four-Switch Control
Figure 3 shows a simplified diagram of how the four
internal switches are connected to the inductor, VIN, VOUT
and GND. Figure 4 shows the regions of operation for the
LTC3444 as a function of the internal control voltage, VCI.
Depending on the control voltage, the LTC3444 will operate in either buck, buck-boost or boost mode. The VCI
VIN
VOUT
6
5
PMOS A
voltage is a level shifted voltage from the output of the
error amp (VC pin) (see Figure 2). The four power switches
are properly phased so the transfer between operating
modes is continuous, smooth and transparent to the user.
The buck-boost region is reached when VIN approaches
VOUT. The conduction time of the four switch region is
typically 125ns. The three operating modes of the four
switch buck-boost converter are described below. Please
refer to Figures 3 and 4.
88% DMAX
BOOST
V4 (~1.16V)
A ON, B OFF
PWM CD
SWITCHES
BOOST REGION
DMIN
BOOST
PMOS D
V3 (~0.73V)
FOUR SWITCH PWM
SW1
SW2
2
4
NMOS B
DMAX
BUCK
NMOS C
0%
DUTY
CYCLE
3444 F03
D ON, C OFF
PWM AB
SWITCHES
BUCK-BOOST
REGION
V2 (~0.49V)
BUCK REGION
V1 (OV)
INTERNAL
CONTROL
VOLTAGE, VCI
3444 F04
Figure 3. Simplified Diagram of Output Switches
Figure 4. Switch Control vs Internal Control Voltage, VCI
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LTC3444
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OPERATIO
Buck Region (VIN > VOUT)
Buck-Boost or Four Switch (VIN ~ VOUT)
Switch D is always on and switch C is always off during this
mode. When the internal control voltage, VCI, is above
voltage V1, Switch A is on. During the off time of switch A,
synchronous switch B turns on for the remainder of the
time. Switches A and B will alternate similar to a typical
synchronous buck regulator. As the control voltage increases, the duty cycle of switch A increases until the
maximum duty cycle of the converter in buck mode
reaches DMAX_BUCK, given by:
When the internal control voltage, VCI, is above voltage V2,
but below V3, switch pair AD remain on for duty cycle
DMAX_BUCK, and the switch pair AC begins to phase in.
As switch pair AC phases in, switch pair BD phases out
accordingly. When the VCI voltage reaches the edge of the
buck-boost range, at voltage V3, the AC switch pair
completely phase out the BD pair, and the boost phase
begins at duty cycle D4SW. The input voltage, VIN, where
the four switch region begins is given by:
DMAX_BUCK = 100% – D4SW
where D4SW = duty cycle % of the four switch range.
D4SW = (125ns • f) • 100 %
VIN =
VOUT
V
1– (125ns • f)
where f = operating frequency, Hz.
Beyond this point the “four switch,” or Buck-Boost region
is reached.
The point at which the four switch region ends is given by:
VIN = VOUT(1–D) = VOUT(1–125ns • f) V
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LTC3444
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OPERATIO
Boost Region (VIN < VOUT)
Switch A is always on and switch B is always off during this
mode. When the internal control voltage, VCI, is above
voltage V3, switch pair CD will alternately switch to provide a boosted output voltage. This operation is typical to
a synchronous boost regulator. The maximum duty cycle
of the converter is limited to 82% typical and is reached
when VCI is above V4.
control voltage range. When using the internal loop compensation, VC = GND, R1 must be 340k. For external
compensation R1 should be chosen first and R2 and R3
calculated from the following equations.
The resistor values are given by:
R3 =
( VCON(MAX ) – VCON(MIN) )
VO(MAX ) – VO(MIN)
• R1 Ω
CONTROLLING THE OUTPUT VOLTAGE
The output voltage is controlled via a summing resistor
input at the feedback (FB) resistive divider string. Refer to
Figure 1. The output voltage has an inverse relation to the
control voltage as shown in Figure 5. The resistor values
are dependent on the desired output voltage range and the
R2 =
1.22
Ω
( VCON(MAX ) – 1.22) (1.22 – VO(MIN) )
–
R3
R1
4.5
4
3.5
VOUT
3
2.5
2
1.5
1
0.5
0
0.5
1
1.5
VCONTROL
2
2.5
3444 G01
Figure 5. VOUT vs VCONTROL with R1 = 340k, R2 = 249k, and
R3 = 182k, VCONTROL = 0.5V to 2.5V
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LTC3444
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OPERATIO
Table 1. Shows some typical resistor value combinations
for several VCONTROL vs VOUT voltage ranges. One percent
(1%) resistor tolerances were assumed.
COMPONENT SELECTION
Recommended Component Placement
Figure 6. Shows a recommended component placement.
Traces carrying high current should be made short and
wide. Trace area at FB and VC pins should be minimized.
Lead lengths to the battery should be kept short. VOUT and
VIN ceramic capacitors should be placed close to the IC
pins. Multiple vias should be used between layers.
Table 1. Typical Resistor Values for VOUT vs VCONTROL
VCONTROL(V)
VOUT (V)
RESISTANCE (kΩ)
MIN
MAX
MIN
MAX
R1
R2
R3
0.35
2.4
0.8
4.2
340
271
205
0.35
2.5
0.5
5.0
340
210
162
0.8
2.35
0.8
4.2
340
200
154
0.5
2.5
0.5
4.2
340
249
182
VCONTROL
LTC3444
VIN
1
SHDN
FB
8
2
SW1
VC
7
VIN
3
GND
VIN
6
4
SW2
VOUT
5
VOUT
3444 F06
MULTIPLE VIAS
Figure 6. Recommended Component Placement
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LTC3444
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OPERATIO
Inductor Selection
The high frequency operation of the LTC3444 allows the
use of small surface mount inductors. The internal loop
compensation is designed to work with a 2.2μH inductor
(1.5μH for VIN < 3.1V). The 2.2μH inductor was selected to
optimize the transient response to the control input. The
use of a 2.2μH inductor pushes out the right half plane
(RHP) zero frequency and allows the loop crossover to
occur at frequencies higher than the output L-C double
pole.
For external compensation the inductor selection is based
on the desired inductor ripple current. The inductor ripple
current is typically set to 20% to 40% of the average
inductor current. Increased inductance results in lower
ripple current, however, higher inductance pulls in the
RHP zero frequency and limits the maximum crossover
frequency possible. Refer to Closing the Feedback Loop
for more information on the RHP zero. For a given ripple
the inductance terms are given as follows:
LBOOST >
VIN(MIN) • ( VOUT – VIN(MIN) )
f • IOUT(MAX ) • ΔIL • VOUT
H
LBUCK >
VOUT • ( VIN(MAX) – VOUT )
f • IOUT(MAX) • ΔIL • VIN(MAX)
H
where f = operating frequency, Hz
ΔIL = inductor ripple current, A
VIN(MIN) = minimum input voltage, V
VIN(MAX) = maximum input voltage, V
VOUT = output voltage, V
IOUT(MAX) = maximum output load current
In most cases, the boost configuration will be used to
determine the minimum inductance allowed for a given
ripple current.
For high efficiency, choose a ferrite inductor with a high
frequency core material to reduce core loses. The inductor
should have low ESR (equivalent series resistance) to
reduce the I2R losses, and must be able to handle the peak
inductor current without saturating. To minimize radiated
noise, use a shielded inductor. See Table 2 for a suggested
list of inductor suppliers.
Table 2. Inductor Vendor Information
SUPPLIER
PHONE
FAX
WEB SITE
Coilcraft
(847) 639-6400
(847) 639-1469
www.coilcraft.com
CoEv Magnetics
(800) 227-7040
(650) 361-2508
www.circuitprotection.com/magnetics.asp
COOPER Bussmann
(636) 394-2877
1-800-544-2570
www.coooperET.com
Murata
(814) 237-1431
(800) 831-9172
(814) 238-0490
www.murata.com
Sumida
USA: (847) 956-0666
Japan: 81(3) 3607-5111
USA: (847) 956-0702
Japan: 81(3) 3607-5144
www.sumida.com
TDK
(847) 803-6100
(847) 803-6296
www.component.tdk.com
TOKO
(847) 297-0070
(847) 699-7864
www.tokoam.com
3444fb
14
LTC3444
U
OPERATIO
Output Capacitor Selection
A 4.7μF, X5R or X7R type ceramic capacitor should be
used when using the internal loop compensation. When
using external compensation, larger values of output
capacitance can be used, however, larger output capacitance will increase the time needed to slew the output
voltage as required in typical WCDMA applications. The
bulk value of the output filter capacitor is set to reduce the
ripple due to charge into the capacitor each cycle. The
steady state ripple due to charge is given by:
% RIPPLE _ BOOST =
IOUT • ( VOUT – VIN(MIN) ) • 100
2
The other component of ripple is due to the ESR (equivalent series resistance) of the output capacitor. Low ESR
capacitors should be used to minimize output voltage
ripple. For surface mount applications, Taiyo Yuden or
TDK ceramic capacitors, AVX TPS series tantalum capacitors or Sanyo POSCAP are recommended. See Table 3 for
contact information.
%
COUT • VOUT • f
% RIPPLE _ BUCK =
IOUT(MAX) • ( VIN(MAX) – VOUT ) • 100
COUT • VIN(MAX) • VOUT • f
In a typical application the output capacitance may be
many times larger than that calculated above in order to
handle the transient load response requirements of the
converter. For a rule of thumb, the ratio of the operating
frequency to the unity-gain bandwidth of the converter is
the amount the output capacitance will have to increase
from the above calculations in order to maintain the
desired transient response. However, in WCDMA applications the output capacitance should be kept at a minimum
to maximize the output slew rate. Refer to the Loop
Compensation Networks section of this datasheet.
Ceramic output capacitors should use case size 1206 or
larger. Smaller case sizes have a larger voltage coefficient
that can greatly reduce the output capacitance value at
higher output voltages.
%
where C OUT = output filter capacitor in farads
f = switching frequency in Hz.
Input Capacitor Selection
Since the VIN pin is the supply voltage for the LTC3444, as
well as the input to the power stage of the converter, it is
recommended to place at least a 4.7μF, X5R or X7R
ceramic bypass capacitor close to the VIN and GND pins.
It is also important to minimize any stray resistance from
the converter to the battery or other power source.
Table 3. Capacitor Vendor Information
SUPPLIER
PHONE
FAX
WEB SITE
AVX
(803) 448-9411
(803) 448-1943
www.avxcorp.com
Sanyo
(619) 661-6322
(619) 661-1055
www.sanyovideo.com
Taio Yuden
(408) 573-4150
(408) 573-4159
www.t-yuden.com
TDK
(847) 803-6100
(847) 803-6296
www.component.tdk.com
3444fb
15
LTC3444
U
OPERATIO
Optional Schottky Diodes
A troublesome problem when operating in boost mode is
dealing with the right-half plane zero (RHP), given by:
Schottky diodes across the synchronous switches B and
D are not required, but provide a lower drop during the
break-before-make time (typically 15ns) of the NMOS to
PMOS transition, improving efficiency. Use a surface
mount Schottky diode such as an MBRM120T3 or equivalent. Do not use ordinary rectifier diodes, since the slow
recovery times will compromise efficiency.
f RHPZ =
The RHP zero has a +20dB/dec gain typical of a zero but
the –90° phase lag of a pole. This causes the loop gain to
flatten out while the phase margin decreases. The only way
to combat a RHP zero is to roll off the loop well before the
RHP zero frequency.
Closing the Feedback Loop
The LTC3444 incorporates voltage mode PWM control.
The control to output gain varies with operation region
(buck, boost, buck-boost), but is usually ~20dB. The
output filter exhibits a double pole response, as given by:
f FILTER_ POLE =
LOOP COMPENSATION NETWORKS
A simple Type I compensation network, refer to Figure 7,
can be incorporated to stabilize the loop, but at a cost of
reduced bandwidth and slower transient response. To
ensure proper phase margin using Type I compensation,
the loop must be crossed over at least a decade before the
output LC double pole frequency. The unity-gain frequency of the error amplifier with the Type I compensation
is given by:
1
Hz
2 • π • L • C OUT
(in buck mod e )
VIN
Hz
2 • VOUT • π • L • C OUT
(in boost mod e )
f FILTER_ POLE =
f UG=
The output filter zero is given by:
1
2 • π • RESR • C OUT
1
Hz
2 • π • R1• C 2
WCDMA applications demand an improved transient response to the input control voltage. In other applications,
the output capacitor can be increased to meet help meet
the load transient requirements.
where L is in Henries and COUT is in farads.
f FILTER_ ZERO =
VIN2
Hz
2 • π • IOUT • L • VOUT
Hz
where RESR is the equivalent series resistance of the
output cap.
C2
–
7
FB
R1
VOUT
8
VC
VREF
R2
+
3444 F07
Figure 7. Error Amplifier with Type I Compensation
3444fb
16
LTC3444
U
OPERATIO
7
1
=
Hz
2 • π • R5 • C 2
FB
R4
R1
VOUT
8
VC
VREF
R2
3444 F08
Figure 8, Error Amplifier with Type III Compensation
GAIN (db)
1
f ZERO1 =
Hz
2 • π • R1• C1
f ZERO2
C1
1
Hz
2 • π • R5 • C 3
1
=
Hz
2 • π • R4 • C1
R5
80
360
60
270
40
180
20
90
fUO
0
0
–20
–90
–40
–180
–60
–270
–80
1e
1e1
1e2
1e3 1e4 1e5 1e6
FREQUENCY (Hz)
1e7
PHASE (DEG)
f POLE2
C2
+
f POLE1 ≅
C3
–
However, due to the output voltage slewing requirements
found in WCDMA applications the output filter capacitor
must be minimized. To maximize the transient response,
while minimizing the output capacitance, a higher bandwidth, Type III compensation is required. A Type III
compensation network, refer to Figure 8, has a double zero
to cancel the double pole of the output LC filter and a
double pole to compensate for the ESR zero and RHP zero
of the boost topology. In addition to the double poles,
the Type III network also has a single pole at DC. The
Type III compensation provides a maximum 135° phase
boost and allows the loop crossover to occur at frequencies higher than the output LC. Refer to Figure 9. Referring
to Figure 8, the location of the poles and zeros are given by:
Assume C2 >> C3, R1 >> R4.
–360
1e8
3444 G02
And the unity gain frequency (fUG) of the Type III compensation is given by:
f UG =
Figure 9. Frequency Response for LTC3444 Error
Amplifier with a Typical Type III Compensation Network
1
Hz
2 • π • R1• C2
where resistance is in ohms and capacitance is in farads.
Note: Bias resistor, R2, does not affect the Pole/Zero
placement.
3444fb
17
LTC3444
U
TYPICAL APPLICATIO S
Example of Internal Compensation Transient Response for a
Command Voltage Change
LTC3444 Dynamic Response
LTC3444 Dynamic Response
VOUT
1V/DIV
1V/DIV
VOUT
VCONTROL
VCONTROL
10μs/DIV
10μs/DIV
VIN = 3.6V, VOUT = 0.8V TO 4.2V
VCONTROL = 2.36V TO 0.28V, ILOAD = 100mA
VIN = 3.6V, VOUT = 4.2V TO 0.8V
VCONTROL = 0.28V TO 2.36V, ILOAD = 100mA
!""" /%
3444 G16a
Internally Compensated WCDMA Application. Singe Cell, 2.7V to
4.2V Input, 0.8V to 4.2V at 400mA Output.
1.5μH
L1
VOUT
0.8V TO 4.2V
LTC3444
SW1
SW2
VIN
VOUT
R1
340k
2.7V TO 4.2V
+
CIN
4.7μF
SHDN
FB
GND
VC
Li-Ion
COUT
4.7μF
R2
267k
R3
205k
VCONTROL
DAC
3444 TA02
CIN = MURATA:GRM31CR61C475K
COUT = MURATA:GRM31CR61C475K
L1 = COOPER BUSSMAN SD12-2R2
3444fb
18
LTC3444
U
TYPICAL APPLICATIO S
Single Li-Ion, 3.1V to 4.2V Input, 3.3V at 400mA
Output with Internal Compensation
2.2μH
L1
VOUT
3.3V AT 400mA
LTC3444
SW1
SW2
VIN
VOUT
R1
340k
3.1V TO 4.4V
+
CIN
4.7μF
SHDN
FB
GND
VC
Li-Ion
COUT
4.7μF
R2
200k
3444 TA04
CIN = MURATA:GRM31CR61C475K
COUT = MURATA:GRM31CR61C475K
L1 = COOPER BUSSMAN SD12-2R2
U
PACKAGE DESCRIPTIO
DD Package
8-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1698)
R = 0.115
TYP
5
0.38 ± 0.10
8
0.675 ±0.05
3.5 ±0.05
1.65 ±0.05
2.15 ±0.05 (2 SIDES)
3.00 ±0.10
(4 SIDES)
PACKAGE
OUTLINE
1.65 ± 0.10
(2 SIDES)
PIN 1
TOP MARK
(NOTE 6)
(DD) DFN 1203
0.25 ± 0.05
0.200 REF
0.50
BSC
2.38 ±0.05
(2 SIDES)
0.75 ±0.05
0.00 – 0.05
4
0.25 ± 0.05
1
0.50 BSC
2.38 ±0.10
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-1)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON TOP AND BOTTOM OF PACKAGE
3444fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LTC3444
U
TYPICAL APPLICATIO
Externally Compensated WCDMA Application. Singe Cell,
3.1V to 4.2V Input, 0.8V to 4.2V at 400mA Output.
3.3μH
L1
VOUT
0.8V TO 4.2V
LTC3444
SW1
SW2
VIN
VOUT
R4
47.5k
3.1V TO 4.2V
+
CIN
4.7μF
Li-Ion
SHDN
FB
GND
VC
C1
10pF
R5
47.5k
C2
220pF
C3
10pF
DAC
R1
340k
VCONTROL
R2
267k
COUT
4.7μF
R3
205k
3444 TA03
CIN = MURATA:GRM31CR61C475K
COUT = MURATA:GRM31CR61C475K
L1 = COOPER BUSSMAN SD12-3R3
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC3403
1.5MHz, 600mA, Synchronous Step-Down Regulator
with Bypass Transistor
96% Efficiency, VIN: 2.5V to 5V, VOUT: 0.3V to 3.5V,
ISD <1μA, (3mm × 3mm) DFN Package
LTC3408
1.5MHz, 600mA, Synchronous Step-Down Regulator
with Bypass Transistor
96% Efficiency, VIN: 2.5V to 5V, VOUT: 0.3V to 3.5V,
ISD <1μA, (3mm × 3mm) DFN Package
LTC3440
Up to 2MHz, 600μA, Synchronous Buck-Boost
DC/DC Converter
95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 2.5V,
ISD <1μA, IQ = 25μA, 10-Lead MS Package
LTC3441
1MHz, 1.2A, Synchronous Buck-Boost
DC/DC Converter
95% Efficiency, VIN: 2.5V to 5.5V VOUT(MIN) = 2.5V,
ISD <1μA, IQ = 25μA, 12-Lead (4mm × 3mm) DFN Package
LTC3442
Up to 2MHz, 1.2A, Synchronous Buck-Boost
DC/DC Converter
95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 2.5V,
ISD <1μA, IQ = 25μA, 12-Lead (4mm × 3mm) DFN Package
LTC3443
600MHz, 1.2A Synchronous Buck-Boost
DC/DC Converter
95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 2.5V,
ISD <1μA, IQ = 25μA, 12-Lead (4mm × 3mm) DFN Package
3444fb
20
Linear Technology Corporation
LT 0507 REV B • PRINTED IN THE USA
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