LINER LTC1474 1.5mhz, 600ma synchronous step-down egulator in thinsot Datasheet

LTC3406
LTC3406-1.5/LTC3406-1.8
1.5MHz, 600mA
Synchronous Step-Down
Regulator in ThinSOT
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FEATURES
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DESCRIPTIO
The LTC ®3406 is a high efficiency monolithic synchronous buck regulator using a constant frequency, current
mode architecture. The device is available in an adjustable
version and fixed output voltages of 1.5V and 1.8V. Supply
current during operation is only 20µA and drops to ≤1µA
in shutdown. The 2.5V to 5.5V input voltage range makes
the LTC3406 ideally suited for single Li-Ion battery-powered applications. 100% duty cycle provides low dropout
operation, extending battery life in portable systems.
Automatic Burst Mode® operation increases efficiency at
light loads, further extending battery life.
High Efficiency: Up to 96%
Very Low Quiescent Current: Only 20µA
During Operation
600mA Output Current
2.5V to 5.5V Input Voltage Range
1.5MHz Constant Frequency Operation
No Schottky Diode Required
Low Dropout Operation: 100% Duty Cycle
0.6V Reference Allows Low Output Voltages
Shutdown Mode Draws ≤ 1µA Supply Current
Current Mode Operation for Excellent Line and
Load Transient Response
Overtemperature Protected
Low Profile (1mm) ThinSOTTM Package
Switching frequency is internally set at 1.5MHz, allowing
the use of small surface mount inductors and capacitors.
The internal synchronous switch increases efficiency and
eliminates the need for an external Schottky diode. Low
output voltages are easily supported with the 0.6V feedback reference voltage. The LTC3406 is available in a low
profile (1mm) ThinSOT package.
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APPLICATIO S
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Cellular Telephones
Personal Information Appliances
Wireless and DSL Modems
Digital Still Cameras
MP3 Players
Portable Instruments
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a registered trademark of Linear Technology Corporation.
ThinSOT is a trademark of Linear Technology Corporation.
Protected by U.S. Patents, including 6580258, 5481178.
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TYPICAL APPLICATIO
95
90
4
CIN**
4.7µF
CER
VIN
SW
3
2.2µH*
COUT†
10µF
CER
LTC3406-1.8
1
RUN
VOUT
5
3406 F01a
GND
2
*MURATA LQH32CN2R2M33
**TAIYO YUDEN JMK212BJ475MG
†
TAIYO YUDEN JMK316BJ106ML
VOUT
1.8V
600mA
85
EFFICIENCY (%)
VIN
2.7V
TO 5.5V
VIN = 2.7V
VIN = 3.6V
80
VIN = 4.2V
75
70
65
60
0.1
VOUT = 1.8V
1
10
100
OUTPUT CURRENT (mA)
1000
3406 F01b
Figure 1a. High Efficiency Step-Down Converter
Figure 1b. Efficiency vs Load Current
3406fa
1
LTC3406
LTC3406-1.5/LTC3406-1.8
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W W
W
ABSOLUTE
AXI U
RATI GS
(Note 1)
Input Supply Voltage .................................. – 0.3V to 6V
RUN, VFB Voltages ..................................... – 0.3V to VIN
SW Voltage .................................. – 0.3V to (VIN + 0.3V)
P-Channel Switch Source Current (DC) ............. 800mA
N-Channel Switch Sink Current (DC) ................. 800mA
Peak SW Sink and Source Current ........................ 1.3A
Operating Temperature Range (Note 2) .. – 40°C to 85°C
Junction Temperature (Note 3) ............................ 125°C
Storage Temperature Range ................ – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
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PACKAGE/ORDER I FOR ATIO
ORDER PART
NUMBER
TOP VIEW
RUN 1
5 VFB
LTC3406ES5
GND 2
SW 3
4 VIN
S5 PACKAGE
5-LEAD PLASTIC TSOT-23
RUN 1
TJMAX = 125°C, θJA = 250°C/ W, θJC = 90°C/ W
LTA5
5 VOUT
LTC3406ES5-1.5
LTC3406ES5-1.8
GND 2
SW 3
S5 PART MARKING
ORDER PART
NUMBER
TOP VIEW
4 VIN
S5 PART MARKING
S5 PACKAGE
5-LEAD PLASTIC TSOT-23
LTD6
LTC4
TJMAX = 125°C, θJA = 250°C/ W, θJC = 90°C/ W
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.
VIN = 3.6V unless otherwise specified.
SYMBOL
PARAMETER
IVFB
Feedback Current
VFB
Regulated Feedback Voltage
CONDITIONS
MIN
TYP
MAX
UNITS
±30
nA
0.5880
0.5865
0.5850
0.6
0.6
0.6
0.6120
0.6135
0.6150
V
V
V
0.04
0.4
1.455
1.746
1.500
1.800
1.545
1.854
●
LTC3406 (Note 4) TA = 25°C
LTC3406 (Note 4) 0°C TA ≤ 85°C
LTC3406 (Note 4) –40°C ≤ TA ≤ 85°C
●
∆VFB
Reference Voltage Line Regulation
VIN = 2.5V to 5.5V (Note 4)
●
VOUT
Regulated Output Voltage
LTC3406-1.5, IOUT = 100mA
LTC3406-1.8, IOUT = 100mA
●
●
∆VOUT
Output Voltage Line Regulation
VIN = 2.5V to 5.5V
●
IPK
Peak Inductor Current
VIN = 3V, VFB = 0.5V or VOUT = 90%,
Duty Cycle < 35%
VLOADREG
Output Voltage Load Regulation
VIN
Input Voltage Range
IS
Input DC Bias Current
Active Mode
Sleep Mode
Shutdown
(Note 5)
VFB = 0.5V or VOUT = 90%, ILOAD = 0A
VFB = 0.62V or VOUT = 103%, ILOAD = 0A
VRUN = 0V, VIN = 4.2V
fOSC
Oscillator Frequency
VFB = 0.6V or VOUT = 100%
VFB = 0V or VOUT = 0V
RPFET
RDS(ON) of P-Channel FET
RNFET
ILSW
0.75
V
V
0.04
0.4
%/V
1
1.25
A
0.5
●
%/V
%
5.5
V
300
20
0.1
400
35
1
µA
µA
µA
1.5
210
1.8
MHz
kHz
ISW = 100mA
0.4
0.5
Ω
RDS(ON) of N-Channel FET
ISW = –100mA
0.35
0.45
Ω
SW Leakage
VRUN = 0V, VSW = 0V or 5V, VIN = 5V
±0.01
±1
●
2.5
1.2
µA
3406fa
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LTC3406
LTC3406-1.5/LTC3406-1.8
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.
VIN = 3.6V unless otherwise specified.
SYMBOL
PARAMETER
CONDITIONS
VRUN
RUN Threshold
●
IRUN
RUN Leakage Current
●
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LTC3406E is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
MIN
TYP
MAX
UNITS
0.3
1
1.5
V
±0.01
±1
µA
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
LTC3406: TJ = TA + (PD)(250°C/W)
Note 4: The LTC3406 is tested in a proprietary test mode that connects
VFB to the output of the error amplifier.
Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
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TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure1a Except for the Resistive Divider Resistor Values)
Efficiency vs Input Voltage
95
IOUT = 100mA
95
Efficiency vs Output Current
Efficiency vs Output Current
100
IOUT = 10mA
95
VOUT = 1.2V
VIN = 2.7V
90
85
IOUT = 600mA
80
75
70
IOUT = 0.1mA
65
VIN = 2.7V
85
EFFICIENCY (%)
EFFICIENCY (%)
EFFICIENCY (%)
90
85 IOUT = 1mA
VOUT = 1.5V
90
VIN = 4.2V
80
VIN = 3.6V
75
VIN = 4.2V
80
VIN = 3.6V
75
70
70
65
65
60
55
50
VOUT = 1.8V
3
2
4
5
INPUT VOLTAGE (V)
60
0.1
6
1
10
100
OUTPUT CURRENT (mA)
VIN = 3.6V
85
VIN = 4.2V
75
70
1.60
FREQUENCY (MHz)
REFERENCE VOLTAGE (V)
EFFICIENCY (%)
VIN = 3.6V
1.65
0.609
90
0.604
0.599
0.594
10
100
1
OUTPUT CURRENT (mA)
1000
3406 G04
1.55
1.50
1.45
1.40
0.589
65
60
0.1
1.70
VIN = 3.6V
VIN = 2.7V
80
Oscillator Frequency vs
Temperature
0.614
VOUT = 2.5V
1000
3406 G03
Reference Voltage vs
Temperature
Efficiency vs Output Current
95
1
10
100
OUTPUT CURRENT (mA)
3406 G02
3406 G01
100
60
0.1
1000
0.584
–50 –25
1.35
50
25
75
0
TEMPERATURE (°C)
100
125
3406 G05
1.30
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
3406 G06
3406fa
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LTC3406
LTC3406-1.5/LTC3406-1.8
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TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure1a Except for the Resistive Divider Resistor Values)
Oscillator Frequency vs
Supply Voltage
1.6
1.5
1.4
1.3
0.7
VIN = 3.6V
1.834
0.6
1.824
0.5
RDS(ON) (Ω)
1.7
OUTPUT VOLTAGE (V)
OSCILLATOR FREQUENCY (MHz)
1.8
1.2
1.814
1.804
3
4
5
SUPPLY VOLTAGE (V)
1.784
0.1
0
6
RDS(ON) vs Temperature
VIN = 2.7V
VIN = 3.6V
0.2
30
25
20
15
0
6
4
3
5
SUPPLY VOLTAGE (V)
2
15
0
–50 –25
VIN = 5.5V
RUN = 0V
SWITCH LEAKAGE (pA)
150
100
MAIN SWITCH
RUN = 0V
SW
5V/DIV
SYNCHRONOUS
SWITCH
80
60
VOUT
100mV/DIV
AC COUPLED
MAIN
SWITCH
IL
200mA/DIV
40
20
VIN = 3.6V
VOUT = 1.8V
ILOAD = 50mA
SYNCHRONOUS SWITCH
0
125
3406 G13
125
Burst Mode Operation
100
250
100
3406 G12
Switch Leakage vs Input Voltage
200
50
25
0
75
TEMPERATURE (°C)
3406 G11
120
100
20
5
Switch Leakage vs Temperature
50
25
75
0
TEMPERATURE (°C)
25
10
125
300
0
–50 –25
30
5
3406 G10
50
35
10
MAIN SWITCH
SYNCHRONOUS SWITCH
7
6
VIN = 3.6V
45 VOUT = 1.8V
= 0A
I
40 LOAD
VOUT = 1.8V
ILOAD = 0A
35
0.1
100
5
4
2
3
INPUT VOLTAGE (V)
Supply Current vs Temperature
SUPPLY CURRENT (µA)
SUPPLY CURRENT (µA)
RDS(ON) (Ω)
0.3
1
3406 G09
40
0.4
50
25
75
0
TEMPERATURE (°C)
0
50
45
0.5
0
–50 –25
0
100 200 300 400 500 600 700 800 900
LOAD CURRENT (mA)
Supply Current vs Supply Voltage
50
VIN = 4.2V
SYNCHRONOUS
SWITCH
3406 G08
0.7
0.6
0.3
0.2
1.774
2
MAIN
SWITCH
0.4
1.794
3406 G07
SWITCH LEAKAGE (nA)
RDS(ON) vs Input Voltage
Output Voltage vs Load Current
1.844
0
1
2
3
4
INPUT VOLTAGE (V)
5
4µs/DIV
3406 G15
6
3406 G14
3406fa
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LTC3406
LTC3406-1.5/LTC3406-1.8
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TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure 1a Except for the Resistive Divider Resistor Values)
Start-Up from Shutdown
Load Step
RUN
2V/DIV
VOUT
2V/DIV
Load Step
VOUT
100mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
ILOAD
500mA/DIV
VIN = 3.6V
VOUT = 1.8V
ILOAD = 600mA
40µs/DIV
3406 G16
IL
500mA/DIV
IL
500mA/DIV
ILOAD
500mA/DIV
ILOAD
500mA/DIV
VIN = 3.6V
20µs/DIV
VOUT = 1.8V
ILOAD = 0mA TO 600mA
Load Step
VIN = 3.6V
20µs/DIV
VOUT = 1.8V
ILOAD = 50mA TO 600mA
3406 G18
Load Step
VOUT
100mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
IL
500mA/DIV
IL
500mA/DIV
ILOAD
500mA/DIV
ILOAD
500mA/DIV
VIN = 3.6V
20µs/DIV
VOUT = 1.8V
ILOAD = 100mA TO 600mA
3406 G17
3406 G19
VIN = 3.6V
20µs/DIV
VOUT = 1.8V
ILOAD = 200mA TO 600mA
3406 G20
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PI FU CTIO S
RUN (Pin 1): Run Control Input. Forcing this pin above
1.5V enables the part. Forcing this pin below 0.3V shuts
down the device. In shutdown, all functions are disabled
drawing <1µA supply current. Do not leave RUN floating.
GND (Pin 2): Ground Pin.
SW (Pin 3): Switch Node Connection to Inductor. This pin
connects to the drains of the internal main and synchronous power MOSFET switches.
VIN (Pin 4): Main Supply Pin. Must be closely decoupled
to GND, Pin 2, with a 2.2µF or greater ceramic capacitor.
VFB (Pin 5) (LTC3406): Feedback Pin. Receives the feedback voltage from an external resistive divider across the
output.
VOUT (Pin 5) (LTC3406-1.5/LTC3406-1.8): Output Voltage Feedback Pin. An internal resistive divider divides the
output voltage down for comparison to the internal reference voltage.
3406fa
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LTC3406
LTC3406-1.5/LTC3406-1.8
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FU CTIO AL DIAGRA
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SLOPE
COMP
0.65V
OSC
OSC
4 VIN
FREQ
SHIFT
–
VFB /VOUT
+
5
LTC3406-1.5
R1 + R2 = 550k
0.6V
R1
FB
LTC3406-1.8
R1 + R2 = 540k
–
+
– EA
0.4V
R2
SLEEP
–
+
BURST
S
Q
R
Q
RS LATCH
VIN
RUN
SWITCHING
LOGIC
AND
BLANKING
CIRCUIT
ANTISHOOTTHRU
3 SW
0.6V REF
+
1
5Ω
+
ICOMP
SHUTDOWN
IRCMP
2 GND
–
3406 BD
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OPERATIO (Refer to Functional Diagram)
Main Control Loop
Burst Mode Operation
The LTC3406 uses a constant frequency, current mode
step-down architecture. Both the main (P-channel
MOSFET) and synchronous (N-channel MOSFET) switches
are internal. During normal operation, the internal top
power MOSFET is turned on each cycle when the oscillator
sets the RS latch, and turned off when the current comparator, ICOMP, resets the RS latch. The peak inductor
current at which ICOMP resets the RS latch, is controlled by
the output of error amplifier EA. When the load current
increases, it causes a slight decrease in the feedback
voltage, FB, relative to the 0.6V reference, which in turn,
causes the EA amplifier’s output voltage to increase until
the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is
turned on until either the inductor current starts to reverse,
as indicated by the current reversal comparator IRCMP, or
the beginning of the next clock cycle.
The LTC3406 is capable of Burst Mode operation in which
the internal power MOSFETs operate intermittently based
on load demand.
In Burst Mode operation, the peak current of the inductor
is set to approximately 200mA regardless of the output
load. Each burst event can last from a few cycles at light
loads to almost continuously cycling with short sleep
intervals at moderate loads. In between these burst events,
the power MOSFETs and any unneeded circuitry are turned
off, reducing the quiescent current to 20µA. In this sleep
state, the load current is being supplied solely from the
output capacitor. As the output voltage droops, the EA
amplifier’s output rises above the sleep threshold signaling the BURST comparator to trip and turn the top MOSFET
on. This process repeats at a rate that is dependent on the
load demand.
3406fa
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LTC3406
LTC3406-1.5/LTC3406-1.8
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OPERATIO (Refer to Functional Diagram)
When the output is shorted to ground, the frequency of the
oscillator is reduced to about 210kHz, 1/7 the nominal
frequency. This frequency foldback ensures that the inductor current has more time to decay, thereby preventing
runaway. The oscillator’s frequency will progressively
increase to 1.5MHz when VFB or VOUT rises above 0V.
Dropout Operation
As the input supply voltage decreases to a value approaching the output voltage, the duty cycle increases toward the
maximum on-time. Further reduction of the supply voltage
forces the main switch to remain on for more than one cycle
until it reaches 100% duty cycle. The output voltage will then
be determined by the input voltage minus the voltage drop
across the P-channel MOSFET and the inductor.
An important detail to remember is that at low input supply
voltages, the RDS(ON) of the P-channel switch increases
(see Typical Performance Characteristics). Therefore, the
user should calculate the power dissipation when the
LTC3406 is used at 100% duty cycle with low input voltage
(See Thermal Considerations in the Applications Information section).
Low Supply Operation
The LTC3406 will operate with input supply voltages as
low as 2.5V, but the maximum allowable output current is
reduced at this low voltage. Figure 2 shows the reduction
in the maximum output current as a function of input
voltage for various output voltages.
Slope Compensation and Inductor Peak Current
Slope compensation provides stability in constant frequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally by
adding a compensating ramp to the inductor current
signal at duty cycles in excess of 40%. Normally, this
results in a reduction of maximum inductor peak current
for duty cycles > 40%. However, the LTC3406 uses a
patent-pending scheme that counteracts this compensating ramp, which allows the maximum inductor peak
current to remain unaffected throughout all duty cycles.
1200
MAXIMUM OUTPUT CURRENT (mA)
Short-Circuit Protection
1000
800
600
VOUT = 1.8V
VOUT = 2.5V
VOUT = 1.5V
400
200
0
2.5
3.0
3.5
4.0
4.5
SUPPLY VOLTAGE (V)
5.0
5.5
3406 F02
Figure 2. Maximum Output Current vs Input Voltage
3406fa
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LTC3406
LTC3406-1.5/LTC3406-1.8
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APPLICATIO S I FOR ATIO
The basic LTC3406 application circuit is shown in Figure 1.
External component selection is driven by the load requirement and begins with the selection of L followed by CIN and
COUT.
inductor to use often depends more on the price vs size
requirements and any radiated field/EMI requirements
than on what the LTC3406 requires to operate. Table 1
shows some typical surface mount inductors that work
well in LTC3406 applications.
Inductor Selection
For most applications, the value of the inductor will fall in
the range of 1µH to 4.7µH. Its value is chosen based on the
desired ripple current. Large value inductors lower ripple
current and small value inductors result in higher ripple
currents. Higher VIN or VOUT also increases the ripple
current as shown in equation 1. A reasonable starting point
for setting ripple current is ∆IL = 240mA (40% of 600mA).
∆IL =
⎛ V ⎞
1
VOUT ⎜ 1 − OUT ⎟
( f)(L) ⎝ VIN ⎠
(1)
The DC current rating of the inductor should be at least
equal to the maximum load current plus half the ripple
current to prevent core saturation. Thus, a 720mA rated
inductor should be enough for most applications (600mA
+ 120mA). For better efficiency, choose a low DC-resistance inductor.
The inductor value also has an effect on Burst Mode
operation. The transition to low current operation begins
when the inductor current peaks fall to approximately
200mA. Lower inductor values (higher ∆IL) will cause this
to occur at lower load currents, which can cause a dip in
efficiency in the upper range of low current operation. In
Burst Mode operation, lower inductance values will cause
the burst frequency to increase.
Inductor Core Selection
Different core materials and shapes will change the size/
current and price/current relationship of an inductor.
Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with
similar electrical characteristics. The choice of which style
Table 1. Representative Surface Mount Inductors
PART
NUMBER
VALUE
(µH)
DCR
(Ω MAX)
MAX DC
SIZE
CURRENT (A) W × L × H (mm3)
Sumida
CDRH3D16
1.5
2.2
3.3
4.7
0.043
0.075
0.110
0.162
1.55
1.20
1.10
0.90
3.8 × 3.8 × 1.8
Sumida
CMD4D06
2.2
3.3
4.7
0.116
0.174
0.216
0.950
0.770
0.750
3.5 × 4.3 × 0.8
Panasonic
ELT5KT
3.3
4.7
0.17
0.20
1.00
0.95
4.5 × 5.4 × 1.2
Murata
LQH32CN
1.0
2.2
4.7
0.060
0.097
0.150
1.00
0.79
0.65
2.5 × 3.2 × 2.0
CIN and COUT Selection
In continuous mode, the source current of the top MOSFET
is a square wave of duty cycle VOUT/VIN. To prevent large
voltage transients, a low ESR input capacitor sized for the
maximum RMS current must be used. The maximum
RMS capacitor current is given by:
1/ 2
VOUT (VIN − VOUT )]
[
CIN required IRMS ≅ IOMAX
VIN
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations
do not offer much relief. Note that the capacitor
manufacturer’s ripple current ratings are often based on
2000 hours of life. This makes it advisable to further derate
the capacitor, or choose a capacitor rated at a higher
temperature than required. Always consult the manufacturer if there is any question.
3406fa
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LTC3406
LTC3406-1.5/LTC3406-1.8
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APPLICATIO S I FOR ATIO
The selection of COUT is driven by the required effective
series resistance (ESR).
Typically, once the ESR requirement for COUT has been
met, the RMS current rating generally far exceeds the
IRIPPLE(P-P) requirement. The output ripple ∆VOUT is determined by:
⎛
1 ⎞
∆VOUT ≅ ∆IL ⎜ ESR +
⎟
⎝
8fC OUT ⎠
where f = operating frequency, COUT = output capacitance
and ∆IL = ripple current in the inductor. For a fixed output
voltage, the output ripple is highest at maximum input
voltage since ∆IL increases with input voltage.
Aluminum electrolytic and dry tantalum capacitors are
both available in surface mount configurations. In the case
of tantalum, it is critical that the capacitors are surge tested
for use in switching power supplies. An excellent choice is
the AVX TPS series of surface mount tantalum. These are
specially constructed and tested for low ESR so they give
the lowest ESR for a given volume. Other capacitor types
include Sanyo POSCAP, Kemet T510 and T495 series, and
Sprague 593D and 595D series. Consult the manufacturer
for other specific recommendations.
induce ringing at the input, VIN. At best, this ringing can
couple to the output and be mistaken as loop instability. At
worst, a sudden inrush of current through the long wires
can potentially cause a voltage spike at VIN, large enough
to damage the part.
When choosing the input and output ceramic capacitors,
choose the X5R or X7R dielectric formulations. These
dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size.
Output Voltage Programming (LTC3406 Only)
In the adjustable version, the output voltage is set by a
resistive divider according to the following formula:
⎛ R2 ⎞
VOUT = 0.6 V ⎜ 1 + ⎟
⎝ R1⎠
(2)
The external resistive divider is connected to the output,
allowing remote voltage sensing as shown in Figure 3.
0.6V ≤ VOUT ≤ 5.5V
R2
VFB
LTC3406
R1
GND
Using Ceramic Input and Output Capacitors
Higher values, lower cost ceramic capacitors are now
becoming available in smaller case sizes. Their high ripple
current, high voltage rating and low ESR make them ideal
for switching regulator applications. Because the
LTC3406’s control loop does not depend on the output
capacitor’s ESR for stable operation, ceramic capacitors
can be used freely to achieve very low output ripple and
small circuit size.
However, care must be taken when ceramic capacitors are
used at the input and the output. When a ceramic capacitor
is used at the input and the power is supplied by a wall
adapter through long wires, a load step at the output can
3406 F03
Figure 3. Setting the LTC3406 Output Voltage
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
3406fa
9
LTC3406
LTC3406-1.5/LTC3406-1.8
U
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APPLICATIO S I FOR ATIO
Although all dissipative elements in the circuit produce
losses, two main sources usually account for most of the
losses in LTC3406 circuits: VIN quiescent current and I2R
losses. The VIN quiescent current loss dominates the
efficiency loss at very low load currents whereas the I2R
loss dominates the efficiency loss at medium to high load
currents. In a typical efficiency plot, the efficiency curve at
very low load currents can be misleading since the actual
power lost is of no consequence as illustrated in Figure 4.
1
POWER LOSS (W)
0.1
RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC)
The RDS(ON) for both the top and bottom MOSFETs can
be obtained from the Typical Performance Charateristics
curves. Thus, to obtain I2R losses, simply add RSW to
RL and multiply the result by the square of the average
output current.
VOUT = 1.2V
VOUT = 1.5V
VOUT = 1.8V
VOUT = 2.5V
0.01
Other losses including CIN and COUT ESR dissipative
losses and inductor core losses generally account for less
than 2% total additional loss.
0.001
0.0001
0.00001
0.1
2. I2R losses are calculated from the resistances of the
internal switches, RSW, and external inductor RL. In
continuous mode, the average output current flowing
through inductor L is “chopped” between the main
switch and the synchronous switch. Thus, the series
resistance looking into the SW pin is a function of both
top and bottom MOSFET RDS(ON) and the duty cycle
(DC) as follows:
Thermal Considerations
1
10
100
LOAD CURRENT (mA)
1000
3406 F04
Figure 4. Power Lost vs Load Current
1. The VIN quiescent current is due to two components:
the DC bias current as given in the electrical characteristics and the internal main switch and synchronous
switch gate charge currents. The gate charge current
results from switching the gate capacitance of the
internal power MOSFET switches. Each time the gate is
switched from high to low to high again, a packet of
charge, dQ, moves from VIN to ground. The resulting
dQ/dt is the current out of VIN that is typically larger than
the DC bias current. In continuous mode, IGATECHG =
f(QT + QB) where QT and QB are the gate charges of the
internal top and bottom switches. Both the DC bias and
gate charge losses are proportional to VIN and thus
their effects will be more pronounced at higher supply
voltages.
In most applications the LTC3406 does not dissipate
much heat due to its high efficiency. But, in applications
where the LTC3406 is running at high ambient temperature with low supply voltage and high duty cycles, such
as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If the junction
temperature reaches approximately 150°C, both power
switches will be turned off and the SW node will become
high impedance.
To avoid the LTC3406 from exceeding the maximum
junction temperature, the user will need to do some
thermal analysis. The goal of the thermal analysis is to
determine whether the power dissipated exceeds the
maximum junction temperature of the part. The temperature rise is given by:
TR = (PD)(θJA)
where PD is the power dissipated by the regulator and θJA
is the thermal resistance from the junction of the die to the
ambient temperature.
3406fa
10
LTC3406
LTC3406-1.5/LTC3406-1.8
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W
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APPLICATIO S I FOR ATIO
The junction temperature, TJ, is given by:
T J = TA + TR
where TA is the ambient temperature.
As an example, consider the LTC3406 in dropout at an
input voltage of 2.7V, a load current of 600mA and an
ambient temperature of 70°C. From the typical performance graph of switch resistance, the RDS(ON) of the
P-channel switch at 70°C is approximately 0.52Ω. Therefore, power dissipated by the part is:
PD = ILOAD2 • RDS(ON) = 187.2mW
For the SOT-23 package, the θJA is 250°C/ W. Thus, the
junction temperature of the regulator is:
TJ = 70°C + (0.1872)(250) = 116.8°C
which is below the maximum junction temperature of
125°C.
Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)).
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to (∆ILOAD • ESR), where ESR is the effective series
resistance of COUT. ∆ILOAD also begins to charge or
discharge COUT, which generates a feedback error signal.
The regulator loop then acts to return VOUT to its steadystate value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability
problem. For a detailed explanation of switching control
loop theory, see Application Note 76.
A second, more severe transient is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive so that
the load rise time is limited to approximately (25 • CLOAD).
Thus, a 10µF capacitor charging to 3.3V would require a
250µs rise time, limiting the charging current to about
130mA.
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC3406. These items are also illustrated graphically in
Figures 5 and 6. Check the following in your layout:
1. The power traces, consisting of the GND trace, the SW
trace and the VIN trace should be kept short, direct and
wide.
2. Does the VFB pin connect directly to the feedback
resistors? The resistive divider R1/R2 must be connected between the (+) plate of COUT and ground.
3. Does the (+) plate of CIN connect to VIN as closely as
possible? This capacitor provides the AC current to the
internal power MOSFETs.
4. Keep the switching node, SW, away from the sensitive
VFB node.
5. Keep the (–) plates of CIN and COUT as close as possible.
3406fa
11
LTC3406
LTC3406-1.5/LTC3406-1.8
U
W
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APPLICATIO S I FOR ATIO
1
RUN
VFB
5
LTC3406
2
–
3
+
L1
SW
RUN
LTC3406-1.8
COUT
VOUT
1
R1
R2
GND
VIN
2
CFWD
4
GND VOUT
–
COUT
VOUT
CIN
5
3
+
L1
VIN
SW
VIN
4
CIN
VIN
3406 F05b
3406 F05a
BOLD LINES INDICATE HIGH CURRENT PATHS
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 5a. LTC3406 Layout Diagram
Figure 5b. LTC3406-1.8 Layout Diagram
VIA TO GND
R1
L1
PIN 1
CFWD
LTC3406
LTC3406-1.8
VOUT
SW
L1
COUT
SW
COUT
CIN
CIN
GND
GND
3406 F06b
3406 F06a
Figure 6a. LTC3406 Suggested Layout
Figure 6b. LTC3406-1.8 Suggested Layout
Design Example
As a design example, assume the LTC3406 is used in a
single lithium-ion battery-powered cellular phone
application. The VIN will be operating from a maximum of
4.2V down to about 2.7V. The load current requirement
is a maximum of 0.6A but most of the time it will be in
standby mode, requiring only 2mA. Efficiency at both low
and high load currents is important. Output voltage is
2.5V. With this information we can calculate L using
equation (1),
L=
⎛ V ⎞
1
VOUT ⎜ 1 − OUT ⎟
( f)(∆IL ) ⎝ VIN ⎠
VIN
VIA TO VOUT
R2
PIN 1
VOUT
VIA TO VOUT
VIA TO VIN
VIN
VIA TO VIN
(3)
Substituting VOUT = 2.5V, VIN = 4.2V, ∆IL = 240mA and
f = 1.5MHz in equation (3) gives:
L=
2.5V
⎛ 2.5V ⎞
⎜1 −
⎟ = 2.81µH
1.5MHz(240mA) ⎝ 4.2V ⎠
A 2.2µH inductor works well for this application. For best
efficiency choose a 720mA or greater inductor with less
than 0.2Ω series resistance.
CIN will require an RMS current rating of at least 0.3A ≅
ILOAD(MAX)/2 at temperature and COUT will require an ESR
of less than 0.25Ω. In most cases, a ceramic capacitor will
satisfy this requirement.
3406fa
12
LTC3406
LTC3406-1.5/LTC3406-1.8
U
W
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APPLICATIO S I FOR ATIO
For the feedback resistors, choose R1 = 316k. R2 can
then be calculated from equation (2) to be:
Figure 7 shows the complete circuit along with its efficiency curve.
⎛V
⎞
R2 = ⎜ OUT − 1⎟ R1 = 1000k
⎝ 0.6
⎠
100
95
VOUT = 2.5V
VIN = 2.7V
VIN
2.7V
TO 4.2V
4
CIN†
2.2µF
CER
VIN
SW
2.2µH*
3
VOUT
2.5V
22pF
COUT**
10µF
CER
LTC3406
1
VFB
RUN
EFFICIENCY (%)
90
5
VIN = 3.6V
85
80
VIN = 4.2V
75
70
1M
GND
65
316k
2
60
0.1
3406 F07a
1
10
100
OUTPUT CURRENT (mA)
*MURATA LQH32CN2R2M33
** TAIYO YUDEN JMK316BJ106ML
†
TAIYO YUDEN LMK212BJ225MG
1000
3406 F07b
Figure 7b
Figure 7a
U
TYPICAL APPLICATIO S
Single Li-Ion 1.5V/600mA Regulator for
High Efficiency and Small Footprint
VIN
2.7V
TO 4.2V
4
CIN**
4.7µF
CER
VIN
SW
3
2.2µH*
COUT1†
LTC3406-1.5
1
10µF
CER
RUN
VOUT
GND
VOUT
1.5V
5
3406 TA05
2
*MURATA LQH32CN2R2M33
**TAIYO YUDEN JMK212BJ475MG
†
TAIYO YUDEN JMK316BJ106ML
95
VOUT = 1.5V
90
VOUT
100mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
VIN = 2.7V
EFFICIENCY (%)
85
VIN = 4.2V
80
IL
500mA/DIV
IL
500mA/DIV
ILOAD
500mA/DIV
ILOAD
500mA/DIV
VIN = 3.6V
75
70
VIN = 3.6V
20µs/DIV
VOUT = 1.5V
ILOAD = 0A TO 600mA
65
60
0.1
1
10
100
OUTPUT CURRENT (mA)
3406 TA07
VIN = 3.6V
20µs/DIV
VOUT = 1.5V
ILOAD = 200mA TO 600mA
3406 TA08
1000
3406 TA06
3406fa
13
LTC3406
LTC3406-1.5/LTC3406-1.8
U
TYPICAL APPLICATIO S
Single Li-Ion 1.2V/600mA Regulator for High Efficiency and Small Footprint
VIN
2.7V
TO 4.2V
4
CIN†
2.2µF
CER
VIN
3
SW
2.2µH*
COUT**
10µF
CER
LTC3406
1
5
VFB
RUN
301k
GND
301k
2
3406 TA09
95
VOUT = 1.2V
90
VOUT
1.2V
22pF
*MURATA LQH32CN2R2M33
** TAIYO YUDEN JMK316BJ106ML
†
TAIYO YUDEN LMK212BJ225MG
VOUT
100mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
VIN = 2.7V
EFFICIENCY (%)
85
VIN = 4.2V
80
IL
500mA/DIV
IL
500mA/DIV
ILOAD
500mA/DIV
ILOAD
500mA/DIV
VIN = 3.6V
75
70
60
0.1
3406 TA11
VIN = 3.6V
20µs/DIV
VOUT = 1.2V
ILOAD = 0mA TO 600mA
65
1
10
100
OUTPUT CURRENT (mA)
VIN = 3.6V
20µs/DIV
VOUT = 1.2V
ILOAD = 200mA TO 600mA
3406 TA12
1000
3406 TA10
Tiny 3.3V/600mA Buck Regulator
VIN
5V
4
†
CIN
4.7µF
CER
VIN
SW
3
2.2µH*
VOUT
3.3V
600mA
22pF
COUT**
10µF
CER
LTC3406
1
VFB
RUN
GND
2
5
301k
66.5k
3406 TA13
*MURATA LQH32CN2R2M33
** TAIYO YUDEN JMK316BJ106ML
†
TAIYO YUDEN JMK212BJ475MG
100
VIN = 5V
95 VOUT = 3.3V
VOUT
100mV/DIV
AC COUPLED
EFFICIENCY (%)
90
IL
500mA/DIV
85
80
ILOAD
500mA/DIV
75
70
VIN = 5V
20µs/DIV
VOUT = 3.3V
ILOAD = 200mA TO 600mA
65
60
0.1
10
100
1
OUTPUT CURRENT (mA)
3406 TA15
1000
3406 TA14
3406fa
14
LTC3406
LTC3406-1.5/LTC3406-1.8
U
PACKAGE DESCRIPTIO
S5 Package
5-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1635)
0.62
MAX
0.95
REF
2.90 BSC
(NOTE 4)
1.22 REF
1.4 MIN
3.85 MAX 2.62 REF
2.80 BSC
1.50 – 1.75
(NOTE 4)
PIN ONE
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45 TYP
5 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.20 BSC
0.01 – 0.10
1.00 MAX
DATUM ‘A’
0.30 – 0.50 REF
0.09 – 0.20
(NOTE 3)
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
1.90 BSC
S5 TSOT-23 0302
3406fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LTC3406
LTC3406-1.5/LTC3406-1.8
U
TYPICAL APPLICATIO
Single Li-Ion 1.8V/600mA Regulator for Low Output Ripple and Small Footprint
VIN
2.7V
TO 4.2V
4
CIN**
4.7µF
CER
VIN
3
SW
VOUT
1.8V
+
LTC3406-1.8
1
4.7µH*
RUN
5
VOUT
GND
COUT1†
100µF
3406 TA01
2
*MURATA LQH32CN4R7M34
**TAIYO YUDEN CERAMIC JMK212BJ475MG
†
SANYO POSCAP 4TPB100M
95
VOUT = 1.8V
VIN = 2.7V
VIN = 3.6V
EFFICIENCY (%)
85
VOUT
100mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
90
80
VIN = 4.2V
IL
500mA/DIV
IL
500mA/DIV
ILOAD
500mA/DIV
ILOAD
500mA/DIV
75
70
VIN = 3.6V
40µs/DIV
VOUT = 1.8V
ILOAD = 0mA TO 600mA
65
60
0.1
1
10
100
OUTPUT CURRENT (mA)
3406 TA03
3406 TA04
VIN = 3.6V
40µs/DIV
VOUT = 1.8V
ILOAD = 200mA TO 600mA
1000
3406 TA02
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1474/LTC1475
250mA (IOUT) Low Quiescent Current Step-Down
DC/DC Converters
VIN: 3V to 18V, Constant Off-Time, IQ = 10µA, MS8 Package
LT1616
1.4MHz, 600mA Step-Down DC/DC Converter
VIN: 3.6V to 25V, IQ = 1.9mA, ThinSOT Package
LTC1701
1MHz, 500mA (IOUT) Step-Down DC/DC Converter
VIN: 2.5V to 5.5V, Constant Off-Time, IQ = 135µA, ThinSOT Package
LTC1767
1.5A, 1.25MHz Step-Down Switching Regulator
VIN: 3V to 25V, IQ = 1mA, MS8/E Packages
LTC1779
550kHz, 250mA (IOUT) Step-Down Switching Regulator
VIN: 2.5V to 9.8V, IQ = 135µA, ThinSOT Package
LTC1875
550kHz, 1.2A (IOUT) Synchronous Step-Down Regulator
VIN: 2.7V to 6V, IQ = 15µA, TSSOP-16 Package
LTC1877
550kHz, 600mA (IOUT) Synchronous Step-Down Regulator
VIN: 2.65V to 10V, IQ = 10µA, MS8 Package
LTC1878
550kHz, 600mA (IOUT) Synchronous Step-Down Regulator
VIN: 2.65V to 6V, IQ = 10µA, MS8 Package
LTC1879
550kHz, 1.2A (IOUT) Synchronous Step-Down Regulator
VIN: 2.7V to 10V, IQ = 15µA, TSSOP-16 Package
LTC3404
1.4MHz, 600mA (IOUT) Synchronous Monolithic
Step-Down Regulator
Up to 95% Efficiency, VIN: 2.65V to 6V, IQ = 10µA, MS8 Package
LTC3405/LTC3405A
LTC3405A-1.5
LTC3405A-1.8
1.5MHz, 300mA (IOUT) Synchronous Monolithic
Step-Down Regulators
Up to 95% Efficiency, VIN: 2.5V to 5.5V, IQ = 20µA,
Fixed Output Voltages Available, ThinSOT Package
LTC3406B
LTC3406B-1.5
LTC3406B-1.8
1.5MHz, 600mA (IOUT) Synchronous Monolithic
Step-Down Regulators
Up to 95% Efficiency, with Pulse Skipping Mode
Fixed Output Voltages Available, ThinSOT Package
LTC3411
4MHz, 1.25A (IOUT) Synchronous Monolithic
Step-Down Regulator
Up to 95% Efficiency, VIN: 2.5V to 5.5V, IQ = 60µA, MS Package
LTC3412
4MHz, 2.5A (IOUT) Synchronous Monolithic
Step-Down Regulator
Up to 95% Efficiency, VIN: 2.5V to 5.5V, IQ = 60µA, TSSOP Package
3406fa
16
Linear Technology Corporation
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●
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