AD AD9251-80EBZ 1.8 v dual analog-to-digital converter Datasheet

14-Bit, 20 MSPS/40 MSPS/65 MSPS/80 MSPS,
1.8 V Dual Analog-to-Digital Converter
AD9251
Data Sheet
FEATURES
FUNCTIONAL BLOCK DIAGRAM
1.8 V analog supply operation
1.8 V to 3.3 V output supply
SNR
74.3 dBFS at 9.7 MHz input
71.5 dBFS at 200 MHz input
SFDR
93 dBc at 9.7 MHz input
80 dBc at 200 MHz input
Low power
33 mW per channel at 20 MSPS
73 mW per channel at 80 MSPS
Differential input with 700 MHz bandwidth
On-chip voltage reference and sample-and-hold circuit
2 V p-p differential analog input
DNL = ±0.45 LSB
Serial port control options
Offset binary, gray code, or twos complement data format
Optional clock duty cycle stabilizer
Integer 1-to-8 input clock divider
Data output multiplex option
Built-in selectable digital test pattern generation
Energy-saving power-down modes
Data clock out with programmable clock and data
alignment
AVDD
Communications
Diversity radio systems
Multimode digital receivers
GSM, EDGE, W-CDMA, LTE, CDMA2000, WiMAX, TD-SCDMA
I/Q demodulation systems
Smart antenna systems
Battery-powered instruments
Hand held scope meters
Portable medical imaging
Ultrasound
Radar/LIDAR
Rev. B
ADC
VREF
SENSE
AD9251
REF
SELECT
RBIAS
VIN–B
ADC
VIN+B
DIVIDE
1 TO 8
CLK+ CLK–
SYNC
D13A
D0A
DCOA
DRVDD
CMOS
OUTPUT BUFFER
VCM
MUX OPTION
VIN–A
ORA
DUTY CYCLE
STABILIZER
MODE
CONTROLS
DCS
PDWN DFS OEB
ORB
D13B
D0B
DCOB
07938-001
PROGRAMMING DATA
VIN+A
CMOS
OUTPUT BUFFER
SPI
Figure 1.
PRODUCT HIGHLIGHTS
1.
2.
APPLICATIONS
SDIO SCLK CSB
GND
3.
4.
The AD9251 operates from a single 1.8 V analog power
supply and features a separate digital output driver supply
to accommodate 1.8 V to 3.3 V logic families.
The patented sample-and-hold circuit maintains excellent
performance for input frequencies up to 200 MHz and is
designed for low cost, low power, and ease of use.
A standard serial port interface supports various product
features and functions, such as data output formatting,
internal clock divider, power-down, DCO/DATA timing
and offset adjustments, and voltage reference modes.
The AD9251 is packaged in a 64-lead RoHS compliant
LFCSP that is pin compatible with the AD9268 16-bit
ADC, the AD9258 14-bit ADC, the AD9231 12-bit ADC,
and the AD9204 10-bit ADC, enabling a simple migration
path between 10-bit and 16-bit converters sampling from
20 MSPS to 125 MSPS.
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Last Content Update: 02/23/2017
COMPARABLE PARTS
REFERENCE MATERIALS
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Product Selection Guide
EVALUATION KITS
Technical Articles
• AD9251 Evaluation Board
• Designing 1-ppm DAC Accuracy into Instrumentation
Applications - Part 1
DOCUMENTATION
• Designing 1-ppm DAC Accuracy into Instrumentation
Applications - Part 2
• RF Source Booklet
Application Notes
• AN-1142: Techniques for High Speed ADC PCB Layout
• AN-742: Frequency Domain Response of SwitchedCapacitor ADCs
• AN-807: Multicarrier WCDMA Feasibility
• AN-808: Multicarrier CDMA2000 Feasibility
• AN-812: MicroController-Based Serial Port Interface (SPI)
Boot Circuit
• AN-827: A Resonant Approach to Interfacing Amplifiers to
Switched-Capacitor ADCs
• Improve The Design Of Your Passive Wideband ADC
Front-End Network
• MS-2210: Designing Power Supplies for High Speed ADC
DESIGN RESOURCES
• AD9251 Material Declaration
• PCN-PDN Information
• Quality And Reliability
• Symbols and Footprints
• AN-878: High Speed ADC SPI Control Software
• AN-905: Visual Analog Converter Evaluation Tool Version
1.0 User Manual
• AN-935: Designing an ADC Transformer-Coupled Front
End
Data Sheet
• AD9251: 14-Bit, 20 MSPS/40 MSPS/65 MSPS/80 MSPS, 1.8
V Dual Analog-to-Digital Converter Data Sheet
User Guides
• UG-003: Evaluating the AD9650/AD9268/AD9258/
AD9251/AD9231/AD9204 Analog-to-Digital Converters
TOOLS AND SIMULATIONS
• Visual Analog
DISCUSSIONS
View all AD9251 EngineerZone Discussions.
SAMPLE AND BUY
Visit the product page to see pricing options.
TECHNICAL SUPPORT
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number.
DOCUMENT FEEDBACK
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• AD9251 IBIS Models
• AD9204/AD9231/AD9251 S-Parameter Data
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AD9251
Data Sheet
TABLE OF CONTENTS
Features .............................................................................................. 1
Voltage Reference ....................................................................... 23
Applications ....................................................................................... 1
Clock Input Considerations ...................................................... 24
Functional Block Diagram .............................................................. 1
Channel/Chip Synchronization ................................................ 26
Product Highlights ........................................................................... 1
Power Dissipation and Standby Mode .................................... 26
Revision History ............................................................................... 2
Digital Outputs ........................................................................... 27
General Description ......................................................................... 3
Timing.......................................................................................... 27
Specifications..................................................................................... 4
Built-In Self-Test (BIST) and Output Test .................................. 28
DC Specifications ......................................................................... 4
Built-In Self-Test (BIST) ............................................................ 28
AC Specifications.......................................................................... 5
Output Test Modes ..................................................................... 28
Digital Specifications ................................................................... 6
Serial Port Interface (SPI) .............................................................. 29
Switching Specifications .............................................................. 7
Configuration Using the SPI ..................................................... 29
Timing Specifications .................................................................. 8
Hardware Interface ..................................................................... 30
Absolute Maximum Ratings .......................................................... 10
Configuration Without the SPI ................................................ 30
Thermal Characteristics ............................................................ 10
SPI Accessible Features .............................................................. 30
ESD Caution ................................................................................ 10
Memory Map .................................................................................. 31
Pin Configuration and Function Descriptions ........................... 11
Reading the Memory Map Register Table............................... 31
Typical Performance Characteristics ........................................... 13
Open Locations .......................................................................... 31
AD9251-80 .................................................................................. 13
Default Values ............................................................................. 31
AD9251-65 .................................................................................. 15
Memory Map Register Table ..................................................... 32
AD9251-40 .................................................................................. 16
Memory Map Register Descriptions ........................................ 34
AD9251-20 .................................................................................. 17
Applications Information .............................................................. 35
Equivalent Circuits ......................................................................... 18
Design Guidelines ...................................................................... 35
Theory of Operation ...................................................................... 20
Outline Dimensions ....................................................................... 36
ADC Architecture ...................................................................... 20
Ordering Guide .......................................................................... 36
Analog Input Considerations.................................................... 20
REVISION HISTORY
9/2016—Rev. A to Rev. B
Changes to Figure 3 .......................................................................... 8
10/2009—Rev. 0 to Rev. A
Changes to Features.......................................................................... 1
Change to Table 1 ............................................................................. 4
Moved Timing Diagrams................................................................. 8
Deleted Table 11; Renumbered Sequentially .............................. 22
Changes to Internal Reference Connection Section .................. 23
Moved Channel/Chip Synchronization Section......................... 26
Change to Table 15 ......................................................................... 30
Changes to Reading the Memory Map Register
Table Section ................................................................................... 31
Changes to Table 16 ....................................................................... 32
7/2009—Revision 0: Initial Version
Rev. B | Page 2 of 36
Data Sheet
AD9251
GENERAL DESCRIPTION
The AD9251 is a monolithic, dual-channel, 1.8 V supply,
14-bit, 20 MSPS/40 MSPS/65 MSPS/80 MSPS analog-to-digital
converter (ADC). It features a high performance sample-andhold circuit and on-chip voltage reference.
A differential clock input controls all internal conversion cycles.
An optional duty cycle stabilizer (DCS) compensates for wide
variations in the clock duty cycle while maintaining excellent
overall ADC performance.
The product uses multistage differential pipeline architecture
with output error correction logic to provide 14-bit accuracy at
80 MSPS data rates and to guarantee no missing codes over the
full operating temperature range.
The digital output data is presented in offset binary, gray code,
or twos complement format. A data output clock (DCO) is
provided for each ADC channel to ensure proper latch timing
with receiving logic. Both 1.8 V and 3.3 V CMOS levels are
supported and output data can be multiplexed onto a single
output bus.
The ADC contains several features designed to maximize
flexibility and minimize system cost, such as programmable
clock and data alignment and programmable digital test pattern
generation. The available digital test patterns include built-in
deterministic and pseudorandom patterns, along with custom
user-defined test patterns entered via the serial port interface (SPI).
The AD9251 is available in a 64-lead RoHS Compliant LFCSP
and is specified over the industrial temperature range (−40°C
to +85°C).
Rev. B | Page 3 of 36
AD9251
Data Sheet
SPECIFICATIONS
DC SPECIFICATIONS
AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS,
DCS disabled, unless otherwise noted.
Table 1.
Parameter
RESOLUTION
ACCURACY
No Missing Codes
Offset Error
Gain Error 1
Differential Nonlinearity (DNL) 2
Integral Nonlinearity (INL)2
MATCHING CHARACTERISTICS
Offset Error
Gain Error1
TEMPERATURE DRIFT
Offset Error
INTERNAL VOLTAGE REFERENCE
Output Voltage (1 V Mode)
Load Regulation Error at 1.0 mA
INPUT-REFERRED NOISE
VREF = 1.0 V
ANALOG INPUT
Input Span, VREF = 1.0 V
Input Capacitance 3
Input Common-Mode Voltage
Input Common-Mode Range
REFERENCE INPUT RESISTANCE
POWER SUPPLIES
Supply Voltage
AVDD
DRVDD
Supply Current
IAVDD2
IDRVDD2 (1.8 V)
IDRVDD2 (3.3 V)
POWER CONSUMPTION
DC Input
Sine Wave Input2 (DRVDD = 1.8 V)
Sine Wave Input2 (DRVDD = 3.3 V)
Standby Power 4
Power-Down Power
Temp
Full
AD9251-20/AD9251-40
Min
Typ
Max
14
Full
Full
Full
Full
25°C
Full
25°C
Guaranteed
±0.1
±0.70
−1.5
±0.60
±0.3
±1.75
±0.6
25°C
25°C
±0.0
±0.2
Full
±2
Full
Full
0.981
Min
14
AD9251-65
Typ
Max
Min
14
Guaranteed
±0.1
±0.50
−1.5
±0.75
±0.45
±1.75
±0.6
±0.65
±0.0
±0.2
AD9251-80
Typ
Max
Guaranteed
±0.1
±0.70
−1.5
±0.70
±0.45
±2.50
±1.0
±0.65
±0.0
±0.2
±2
0.993
2
1.005
0.981
0.993
2
±0.65
±2
1.005
0.981
0.993
2
Unit
Bits
% FSR
% FSR
LSB
LSB
LSB
LSB
% FSR
% FSR
ppm/°C
1.005
V
mV
25°C
0.98
0.98
0.98
LSB rms
Full
Full
Full
Full
Full
2
6
0.9
2
6
0.9
2
6
0.9
V p-p
pF
V
V
kΩ
Full
Full
0.5
1.3
0.5
7.5
1.7
1.7
1.3
0.5
7.5
1.8
1.9
3.6
Full
Full
Full
36.5/49.5
3.4/5.6
6.3/10.6
39.4/52.8
Full
Full
Full
Full
Full
66/89
71.8/99
86.5/124
37
2.2
77.0/105.5
1.7
1.7
1.8
1.9
3.6
69.0
8.4
16.0
72.9
125
139.0
176.7
37
2.2
1.7
1.7
146.5
Measured with 1.0 V external reference.
Measured with a 10 MHz input frequency at rated sample rate, full-scale sine wave, with approximately 5 pF loading on each output bit.
Input capacitance refers to the effective capacitance between one differential input pin and AGND.
4
Standby power is measured with a dc input and the CLK active.
1
2
3
Rev. B | Page 4 of 36
1.3
7.5
1.8
1.9
3.6
V
V
80.5
10.3
19.5
85.5
mA
mA
mA
145
163.4
209
37
2.2
173
mW
mW
mW
mW
mW
Data Sheet
AD9251
AC SPECIFICATIONS
AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS,
DCS disabled, unless otherwise noted.
Table 2.
Parameter 1
SIGNAL-TO-NOISE RATIO (SNR)
fIN = 9.7 MHz
fIN = 30.5 MHz
fIN = 70 MHz
fIN = 200 MHz
SIGNAL-TO-NOISE-AND-DISTORTION (SINAD)
fIN = 9.7 MHz
fIN = 30.5 MHz
fIN = 70 MHz
fIN = 200 MHz
EFFECTIVE NUMBER OF BITS (ENOB)
fIN = 9.7 MHz
fIN = 30.5 MHz
fIN = 70 MHz
fIN = 200 MHz
WORST SECOND OR THIRD HARMONIC
fIN = 9.7 MHz
fIN = 30.5 MHz
fIN = 70 MHz
fIN = 200 MHz
SPURIOUS-FREE DYNAMIC RANGE (SFDR)
fIN = 9.7 MHz
fIN = 30.5 MHz
fIN = 70 MHz
fIN = 200 MHz
WORST OTHER (HARMONIC OR SPUR)
fIN = 9.7 MHz
fIN = 30.5 MHz
fIN = 70 MHz
fIN = 200 MHz
TWO-TONE SFDR
fIN = 30.5 MHz (−7 dBFS), 32.5 MHz (−7 dBFS)
CROSSTALK 2
ANALOG INPUT BANDWIDTH
1
2
Temp
25°C
25°C
Full
25°C
Full
25°C
25°C
25°C
Full
25°C
Full
25°C
AD9251-20/AD9251-40
Min
Typ
Max
Min
74.7
74.4
AD9251-65
Typ
Max
Min
74.5
74.3
73.6
AD9251-80
Typ
Max
74.3
74.1
dBFS
dBFS
dBFS
dBFS
dBFS
dBFS
73.6
73.7
73.7
73.6
72.5
71.5
71.5
71.5
74.6
74.3
74.4
74.2
74.1
74.0
Unit
70.0
70.0
70.0
dBFS
dBFS
dBFS
dBFS
dBFS
dBFS
25°C
25°C
25°C
25°C
12.0
12.0
11.9
11.3
12.0
12.0
11.9
11.3
12.0
12.0
11.9
11.3
Bits
Bits
Bits
Bits
25°C
25°C
Full
25°C
Full
25°C
−95
−95
−95
−95
−93
−93
dBc
dBc
dBc
dBc
dBc
dBc
25°C
25°C
Full
25°C
Full
25°C
73.4
73.4
73.6
73.6
73.5
72.4
−81
−81
−94
−94
−92
−80
−80
−80
95
94
95
94
93
93
−81
81
dBc
dBc
dBc
dBc
dBc
dBc
81
93
93
92
81
80
80
80
−98
−98
−98
−98
−97
−97
25°C
25°C
Full
25°C
Full
25°C
−98
−98
−96
−95
−95
−95
dBc
dBc
dBc
dBc
dBc
dBc
25°C
Full
25°C
90
−110
700
90
−110
700
90
−110
700
dBc
dBc
MHz
−90
−90
−89
See the AN-835 Application Note, Understanding High Speed ADC Testing and Evaluation, for a complete set of definitions.
Crosstalk is measured at 100 MHz with −1.0 dBFS on one channel and no input on the alternate channel.
Rev. B | Page 5 of 36
AD9251
Data Sheet
DIGITAL SPECIFICATIONS
AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS,
DCS disabled, unless otherwise noted.
Table 3.
Parameter
DIFFERENTIAL CLOCK INPUTS (CLK+, CLK−)
Logic Compliance
Internal Common-Mode Bias
Differential Input Voltage
Input Voltage Range
High Level Input Current
Low Level Input Current
Input Resistance
Input Capacitance
LOGIC INPUTS (SCLK/DFS, SYNC, PDWN) 1
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Resistance
Input Capacitance
LOGIC INPUTS (CSB) 2
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Resistance
Input Capacitance
LOGIC INPUTS (SDIO/DCS)2
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Resistance
Input Capacitance
DIGITAL OUTPUTS
DRVDD = 3.3 V
High Level Output Voltage, IOH = 50 µA
High Level Output Voltage, IOH = 0.5 mA
Low Level Output Voltage, IOL = 1.6 mA
Low Level Output Voltage, IOL = 50 µA
DRVDD = 1.8 V
High Level Output Voltage, IOH = 50 µA
High Level Output Voltage, IOH = 0.5 mA
Low Level Output Voltage, IOL = 1.6 mA
Low Level Output Voltage, IOL = 50 µA
1
2
Temp
Full
Full
Full
Full
Full
Full
Full
Min
AD9251-20/AD9251-40/AD9251-65/AD9251-80
Typ
Max
CMOS/LVDS/LVPECL
0.9
0.2
GND − 0.3
−10
−10
8
Full
Full
Full
Full
Full
Full
1.2
0
−50
−10
Full
Full
Full
Full
Full
Full
1.2
0
−10
40
Full
Full
Full
Full
Full
Full
1.2
0
−10
40
Full
Full
Full
Full
3.29
3.25
Full
Full
Full
Full
1.79
1.75
10
4
3.6
AVDD + 0.2
+10
+10
12
V
V
µA
µA
kΩ
pF
DRVDD + 0.3
0.8
+10
135
V
V
µA
µA
kΩ
pF
DRVDD + 0.3
0.8
+10
130
V
V
µA
µA
kΩ
pF
26
2
26
5
Rev. B | Page 6 of 36
V
V p-p
V
µA
µA
kΩ
pF
DRVDD + 0.3
0.8
−75
+10
30
2
Internal 30 kΩ pull-down.
Internal 30 kΩ pull-up.
Unit
0.2
0.05
V
V
V
V
0.2
0.05
V
V
V
V
Data Sheet
AD9251
SWITCHING SPECIFICATIONS
AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS,
DCS disabled, unless otherwise noted.
Table 4.
Parameter
CLOCK INPUT PARAMETERS
Input Clock Rate
Conversion Rate 1
CLK Period—Divide-by-1 Mode (tCLK)
CLK Pulse Width High (tCH)
Aperture Delay (tA)
Aperture Uncertainty (Jitter, tJ)
DATA OUTPUT PARAMETERS
Data Propagation Delay (tPD)
DCO Propagation Delay (tDCO)
DCO to Data Skew (tSKEW)
Pipeline Delay (Latency)
Wake-Up Time 2
Standby
OUT-OF-RANGE RECOVERY TIME
1
2
Temp
Full
Full
Full
AD9251-20/AD9251-40
Min
Typ
Max
625
20/40
Min
AD9251-65
Typ
Max
AD9251-80
Typ
Max
Full
Full
25.0/12.5
1.0
0.1
7.69
1.0
0.1
6.25
1.0
0.1
Full
Full
Full
Full
Full
Full
Full
3
3
0.1
9
350
600/400
2
3
3
0.1
9
350
300
2
3
3
0.1
9
350
260
2
ns
ns
ns
Cycles
µs
ns
Cycles
50/25
Conversion rate is the clock rate after the CLK divider.
Wake-up time is dependent on the value of the decoupling capacitors.
Rev. A | Page 7 of 36
3
15.38
625
80
Unit
MHz
MSPS
ns
ns
ns
ps rms
3
625
65
Min
3
12.5
AD9251
Data Sheet
TIMING SPECIFICATIONS
Table 5.
Parameter
SYNC TIMING REQUIREMENTS
tSSYNC
tHSYNC
SPI TIMING REQUIREMENTS
tDS
tDH
tCLK
tS
tH
tHIGH
tLOW
tEN_SDIO
tDIS_SDIO
Conditions
Min
SYNC to rising edge of CLK setup time
SYNC to rising edge of CLK hold time
Typ
Max
0.24
0.40
Setup time between the data and the rising edge of SCLK
Hold time between the data and the rising edge of SCLK
Period of the SCLK
Setup time between CSB and SCLK
Hold time between CSB and SCLK
SCLK pulse width high
SCLK pulse width low
Time required for the SDIO pin to switch from an input to an
output relative to the SCLK falling edge
Time required for the SDIO pin to switch from an output to an
input relative to the SCLK rising edge
ns
ns
2
2
40
2
2
10
10
10
ns
ns
ns
ns
ns
ns
ns
ns
10
ns
Timing Diagrams
N–1
N+4
tA
N+5
N
N+3
VIN
N+1
tCH
N+2
tCLK
CLK+
CLK–
tDCO
DCOA/DCOB
N–9
N–8
N–7
N–6
N–5
tPD
07938-002
tSKEW
CH A/CH B DATA
Figure 2. CMOS Output Data Timing
N–1
N+4
tA
N+5
N
N+3
VIN
N+1
tCH
N+2
tCLK
CLK+
CLK–
tDCO
DCOA/DCOB
CH A
N–9
CH B
N–9
CH A
N–8
CH B
N–8
CH A
N–7
CH B
N–7
CH A
N–6
CH B
N–6
tPD
Figure 3. CMOS Interleaved Output Timing, Output as Appears on Channel A Output Pins
Rev. A | Page 8 of 36
CH A
N–5
07938-003
tSKEW
CH A/CH B DATA
AS APPEARS ON
CH A OUTPUT PINS
Unit
Data Sheet
AD9251
CLK+
tHSYNC
07938-004
tSSYNC
SYNC
Figure 4. SYNC Input Timing Requirements
Rev. A | Page 9 of 36
AD9251
Data Sheet
ABSOLUTE MAXIMUM RATINGS
THERMAL CHARACTERISTICS
Table 6.
Parameter
AVDD to AGND
DRVDD to AGND
VIN+A, VIN+B, VIN−A, VIN−B to AGND
CLK+, CLK− to AGND
SYNC to AGND
VREF to AGND
SENSE to AGND
VCM to AGND
RBIAS to AGND
CSB to AGND
SCLK/DFS to AGND
SDIO/DCS to AGND
OEB to AGND
PDWN to AGND
D0A/D0B through D13A/D13B to AGND
DCOA/DCOB to AGND
Operating Temperature Range (Ambient)
Maximum Junction Temperature
Under Bias
Storage Temperature Range (Ambient)
Rating
−0.3 V to +2.0 V
−0.3 V to +3.9 V
−0.3 V to AVDD + 0.2 V
−0.3 V to AVDD + 0.2 V
−0.3 V to DRVDD + 0.3 V
−0.3 V to AVDD + 0.2 V
−0.3 V to AVDD + 0.2 V
−0.3 V to AVDD + 0.2 V
−0.3 V to AVDD + 0.2 V
−0.3 V to DRVDD + 0.3 V
−0.3 V to DRVDD + 0.3 V
−0.3 V to DRVDD + 0.3 V
−0.3 V to DRVDD + 0.3 V
−0.3 V to DRVDD + 0.3 V
−0.3 V to DRVDD + 0.3 V
−0.3 V to DRVDD + 0.3 V
−40°C to +85°C
150°C
The exposed paddle is the only ground connection for the chip.
The exposed paddle must be soldered to the AGND plane of the
user’s circuit board. Soldering the exposed paddle to the user’s
board also increases the reliability of the solder joints and
maximizes the thermal capability of the package.
Typical θJA is specified for a 4-layer PCB with a solid ground
plane. As shown in Table 7, airflow improves heat dissipation,
which reduces θJA. In addition, metal in direct contact with the
package leads from metal traces, through holes, ground, and
power planes, reduces the θJA.
Table 7. Thermal Resistance
Package Type
64-Lead LFCSP
(CP-64-4)
Airflow
Velocity
(m/sec)
0
1.0
2.5
θJA1, 2
23
20
18
θJC1, 3
2.0
θJB1, 4
12
Per JEDEC 51-7, plus JEDEC 25-5 2S2P test board.
Per JEDEC JESD51-2 (still air) or JEDEC JESD51-6 (moving air).
3
Per MIL-Std 883, Method 1012.1.
4
Per JEDEC JESD51-8 (still air).
1
2
−65°C to +150°C
Stresses at or above those listed under Absolute Maximum
Ratings may cause permanent damage to the product. This is a
stress rating only; functional operation of the product at these
or any other conditions above those indicated in the operational
section of this specification is not implied. Operation beyond
the maximum operating conditions for extended periods may
affect product reliability.
ESD CAUTION
Rev. A | Page 10 of 36
Unit
°C/W
°C/W
°C/W
Data Sheet
AD9251
64
63
62
61
60
59
58
57
56
55
54
53
52
51
50
49
AVDD
AVDD
VIN+B
VIN–B
AVDD
AVDD
RBIAS
VCM
SENSE
VREF
AVDD
AVDD
VIN–A
VIN+A
AVDD
AVDD
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
PIN 1
INDICATOR
AD9251
TOP VIEW
(Not to Scale)
48
47
46
45
44
43
42
41
40
39
38
37
36
35
34
33
PDWN
OEB
CSB
SCLK/DFS
SDIO/DCS
ORA
D13A (MSB)
D12A
D11A
D10A
D9A
DRVDD
D8A
D7A
D6A
D5A
NOTES
1. NC = NO CONNECT
2. THE EXPOSED PADDLE MUST BE SOLDERED TO THE PCB GROUND
TO ENSURE PROPER HEAT DISSIPATION, NOISE, AND MECHANICAL
STRENGTH BENEFITS.
07938-005
D10B
D11B
DRVDD
D12B
(MSB) D13B
ORB
DCOB
DCOA
NC
NC
(LSB) D0A
DRVDD
D1A
D2A
D3A
D4A
17
18
19
20
21
22
23
24
25
26
27
28
29
30
31
32
CLK+
CLK–
SYNC
NC
NC
(LSB) D0B
D1B
D2B
D3B
DRVDD
D4B
D5B
D6B
D7B
D8B
D9B
Figure 5. Pin Configuration
Table 8. Pin Function Description
Pin No.
0
1, 2
3
4, 5, 25, 26
6 to 9, 11 to 18, 20, 21
10, 19, 28, 37
22
23
24
27, 29 to 36, 38 to 42
43
44
Mnemonic
GND
CLK+, CLK−
SYNC
NC
D0B to D13B
DRVDD
ORB
DCOB
DCOA
D0A to D13A
ORA
SDIO/DCS
45
SCLK/DFS
46
47
CSB
OEB
48
PDWN
49, 50, 53, 54, 59, 60, 63, 64
51, 52
AVDD
VIN+A, VIN−A
Description
Exposed paddle is the only ground connection for the chip. Must be connected to PCB AGND.
Differential Encode Clock. PECL, LVDS, or 1.8 V CMOS inputs.
Digital Input. SYNC input to clock divider. 30 kΩ internal pull-down.
Do Not Connect.
Channel B Digital Outputs. D13B = MSB.
Digital Output Driver Supply (1.8 V to 3.3 V).
Channel B Out-of-Range Digital Output.
Channel B Data Clock Digital Output.
Channel A Data Clock Digital Output.
Channel A Digital Outputs. D13A = MSB.
Channel A Out-of-Range Digital Output.
SPI Data Input/Output (SDIO). Bidirectional SPI Data I/O in SPI mode. 30 kΩ internal pulldown in SPI mode.
Duty Cycle Stabilizer (DCS). Static enable input for duty cycle stabilizer in non-SPI mode.
30 kΩ internal pull-up in non-SPI (DCS) mode.
SPI Clock (SCLK) Input in SPI mode. 30 kΩ internal pull-down.
Data Format Select (DFS). Static control of data output format in non-SPI mode. 30 kΩ internal
pull-down.
DFS high = twos complement output.
DFS low = offset binary output.
SPI Chip Select. Active low enable; 30 kΩ internal pull-up.
Digital Input. Enable Channel A and Channel B digital outputs if low, tristate outputs if high.
30 kΩ internal pull-down.
Digital Input. 30 kΩ internal pull-down.
PDWN high = power-down device.
PDWN low = run device, normal operation.
1.8 V Analog Supply Pins.
Channel A Analog Inputs.
Rev. A | Page 11 of 36
AD9251
Pin No.
55
56
57
58
61, 62
Data Sheet
Mnemonic
VREF
SENSE
VCM
RBIAS
VIN−B, VIN+B
Description
Voltage Reference Input/Output.
Reference Mode Selection.
Analog output voltage at midsupply to set common mode of the analog inputs.
Sets Analog Current Bias. Connect to 10 kΩ (1% tolerance) resistor to ground.
Channel B Analog Inputs.
Rev. A | Page 12 of 36
Data Sheet
AD9251
TYPICAL PERFORMANCE CHARACTERISTICS
AD9251-80
AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS, DCS
disabled, unless otherwise noted.
0
0
80MSPS
9.7MHz @ –1dBFS
SNR = 73.4dB (74.4dBFS)
SFDR = 94.4dBc
–15
–15
–30
AMPLITUDE (dBFS)
–45
–60
–75
–45
–60
–75
–90
3
3
2
–105
5
6
6
4
–120
–120
8
12
16
20
24
28
FREQUENCY (MHz)
32
4
07938-033
4
36
8
12
16
20
24
28
FREQUENCY (MHz)
32
36
Figure 9. Single-Tone FFT with fIN = 30.5 MHz
Figure 6. Single-Tone FFT with fIN = 9.7 MHz
0
0
80MSPS
70.3MHz @ –1dBFS
SNR = 72.1dB (73.1dBFS)
SFDR = 93.5dBc
–15
–15
–30
80MSPS
200MHz @ –1dBFS
SNR = 70.5dB (71.5dBFS)
SFDR = 80.2dBc
–30
AMPLITUDE (dBFS)
AMPLITUDE (dBFS)
2
5
–105
4
07938-034
AMPLITUDE (dBFS)
–30
–90
80MSPS
30.5MHz @ –1dBFS
SNR = 73.2dB (74.2dBFS)
SFDR = 93.6dBc
–45
–60
–75
–90
2
3
6
–105
–60
–75
2
3
–90
4
5
–45
–105
4 6
5
–120
12
16
20
24
28
FREQUENCY (MHz)
32
36
–120
4
Figure 7. Single-Tone FFT with fIN = 70.3 MHz
8
12
16
20
24
28
FREQUENCY (MHz)
32
07938-036
8
07938-062
4
36
Figure 10. Single-Tone FFT with fIN = 200 MHz
0
–15
80MSPS
30.5MHz @ –7dBFS
32.5MHz @ –7dBFS
SFDR = 89.5dBc (96.5dBFS)
0
–20
SFDR/IMD3 (dBc/dBFS)
–45
–60
–75
–90
2F1 + F2
2F2 + F1
F1 + F2
F2 – F1
2F1 – F2
2F2 – F1
SFDR (dBc)
–40
IMD3 (dBc)
–60
–80
–105
SFDR (dBFS)
–100
–120
8
12
28
16
20
24
FREQUENCY (MHz)
32
36
–120
–90
Figure 8. Two-Tone FFT with fIN1 = 30.5 MHz and fIN2 = 32.5 MHz
–78
–66
–54
–42
–30
INPUT AMPLITUDE (dBFS)
–18
–6
07938-054
IMD3 (dBFS)
4
07938-200
AMPLITUDE (dBFS)
–30
Figure 11. Two-Tone SFDR/IMD3 vs. Input Amplitude (AIN) with fIN1 =
30.5 MHz and fIN2 = 32.5 MHz
Rev. A | Page 13 of 36
AD9251
Data Sheet
AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS, DCS
disabled, unless otherwise noted.
120
100
SFDR (dBc)
90
SFDRFS
100
70
SNR (dBFS)
SNR/SFDR (dBFS)
SNR/SFDR (dBFS/dBc)
80
60
50
40
80
SNRFS
60
SFDR
40
30
SNR
20
20
0
50
100
150
INPUT FREQUENCY (MHz)
200
0
–90
07938-057
0
Figure 12. SNR/SFDR vs. Input Frequency (AIN) with 2 V p-p Full Scale
–80
–60
–40
INPUT AMPLITUDE (dBFS)
–20
0
07938-061
10
Figure 15. SNR/SFDR vs. Input Amplitude (AIN) with fIN = 9.7 MHz
120
450,000
400,000
SFDR (dBc)
100
SNR (dBFS)
80
NUMBER OF HITS
SNR/SFDR (dBFS/dBc)
350,000
60
40
300,000
250,000
200,000
150,000
100,000
20
20
30
40
50
60
SAMPLE RATE (MSPS)
70
80
0
07938-055
0
10
N
N+1 N+2 N+3 N+4
N–4 N–3 N–2 N–1
OUTPUT CODE
Figure 13. SNR/SFDR vs. Sample Rate with AIN = 9.7 MHz
07938-048
50,000
Figure 16. Grounded Input Histogram
0.5
2.0
0.4
1.5
0.3
1.0
INL ERROR (LSB)
0.1
0
–0.1
0.5
0
–0.5
–0.2
–1.0
–0.3
–0.5
0
2048
4096
6144
8192 10,240 12,288 14,336 16,384
OUTPUT CODE
–2.0
0
2048
4096
6144
8192 10,240 12,288 14,336 16,384
OUTPUT CODE
Figure 17. INL with fIN = 9.7 MHz
Figure 14. DNL Error with fIN = 9.7 MHz
Rev. A | Page 14 of 36
07938-037
–1.5
–0.4
07938-038
DNL ERROR (LSB)
0.2
Data Sheet
AD9251
AD9251-65
AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS, DCS
disabled, unless otherwise noted.
0
120
65MSPS
9.7MHz @ –1dBFS
SNR = 73.5dB (74.5dBFS)
SFDR = 97.7dBc
–15
SFDRFS
100
SNR/SFDR (dBFS)
AMPLITUDE (dBFS)
–30
–45
–60
–75
80
SNRFS
60
SFDR
40
–90
SNR
5
6
–105
2
4
3
20
21
15
18
12
FREQUENCY (MHz)
9
24
27
0
–90
07938-030
6
3
30
Figure 18. Single-Tone FFT with fIN = 9.7 MHz
90
SNR/SFDR (dBFS/dBc)
–60
–75
2
4
5
6
6
9
12
15
18
21
FREQUENCY (MHz)
24
27
30
50
40
30
0
0
Figure 19. Single-Tone FFT with fIN = 70.3 MHz
65MSPS
30.5MHz @ –1dBFS
SNR = 73.3dB (74.3dBFS)
SFDR = 99.3dBc
–45
–60
–75
–90
3
5
2
–105
6
4
6
9
12
15
18
21
FREQUENCY (MHz)
24
27
30
07938-031
–120
3
50
100
150
INPUT FREQUENCY (MHz)
Figure 22. SNR/SFDR vs. Input Frequency (AIN) with
2 V p-p Full Scale
–30
AMPLITUDE (dBFS)
SNR (dBFS)
60
20
07938-032
3
–15
70
10
–120
0
SFDR (dBc)
Figure 20. Single-Tone FFT with fIN = 30.5 MHz
Rev. A | Page 15 of 36
200
07938-056
AMPLITUDE (dBFS)
–45
3
0
80
–30
–105
–20
100
65MSPS
70.3MHz @ –1dBFS
SNR = 72.6dB (73.6dBFS)
SFDR = 94.1dBc
–90
–60
–40
INPUT AMPLITUDE (dBFS)
Figure 21. SNR/SFDR vs. Input Amplitude (AIN) with fIN = 9.7 MHz
0
–15
–80
07938-060
–120
AD9251
Data Sheet
AD9251-40
AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS, DCS
disabled, unless otherwise noted.
0
–15
120
40MSPS
9.7MHz @ –1dBFS
SNR = 73.5dB (74.5dBFS)
SFDR = 95.4dBc
SFDRFS
100
SNR/SFDR (dBFS)
AMPLITUDE (dB)
–30
–45
–60
–75
–90
–105
80
SNRFS
60
SFDR
40
SNR
4
5
3
6
2
20
4
6
12
14
8
10
FREQUENCY (MHz)
18
16
0
–90
07938-028
2
Figure 23. Single-Tone FFT with fIN = 9.7 MHz
0
–15
40MSPS
30.5MHz @ –1dBFS
SNR = 73.2dB (74.2dBFS)
SFDR = 95.7dBc
–60
–75
–90
5
3
2
6
–120
2
4
6
14
12
10
8
FREQUENCY (MHz)
16
18
07938-029
AMPLITUDE (dBFS)
–45
4
–60
–40
INPUT AMPLITUDE (dBFS)
–20
0
Figure 25. SNR/SFDR vs. Input Amplitude (AIN) with fIN = 9.7 MHz
–30
–105
–80
07938-059
–120
Figure 24. Single-Tone FFT with fIN = 30.5 MHz
Rev. A | Page 16 of 36
Data Sheet
AD9251
AD9251-20
AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS, DCS
disabled, unless otherwise noted.
0
–15
120
20MSPS
9.7MHz @ –1dBFS
SNR = 73.5dBFS (74.5dBFS)
SFDR = 97.2dBc
SFDR (dBFS)
100
SNR/SFDR (dBc/dBFS)
AMPLITUDE (dBFS)
–30
–45
–60
–75
–90
–105
2
4
5 3
6
SNR (dBFS)
80
60
SFDR (dBc)
40
SNR (dBc)
20
0
–100
07938-024
950k 1.90 2.85 3.80 4.75 5.70 6.65 7.60 8.55 9.50
FREQUENCY (MHz)
Figure 26. Single-Tone FFT with fIN = 9.7 MHz
20MSPS
30.5MHz @ –1dBFS
SNR = 73.2dB (74.2dBFS)
SFDR = 98.1dBc
–45
–60
–75
–90
2
4
6
5
3
–120
950k 1.90 2.85 3.80 4.75 5.70 6.65 7.60 8.55 9.50
FREQUENCY (MHz)
07938-026
AMPLITUDE (dBFS)
–30
–105
–80
–70 –60 –50 –40 –30
INPUT AMPLITUDE (dBFS)
–20
–10
0
Figure 28. SNR/SFDR vs. Input Amplitude (AIN) with fIN = 9.7 MHz
0
–15
–90
07938-058
–120
Figure 27. Single-Tone FFT with fIN = 30.5 MHz
Rev. A | Page 17 of 36
AD9251
Data Sheet
EQUIVALENT CIRCUITS
DRVDD
AVDD
07938-042
07938-039
VIN±x
Figure 29. Equivalent Analog Input Circuit
Figure 32. Equivalent Digital Output Circuit
5Ω
CLK+
15kΩ
0.9V
15kΩ
DRVDD
5Ω
CLK–
350Ω
30kΩ
Figure 30. Equivalent Clock Input Circuit
07938-043
07938-040
SCLK/DFS, SYNC,
OEB, AND PDWN
Figure 33. Equivalent SCLK/DFS, SYNC, OEB, and PDWN Input Circuit
AVDD
AVDD
DRVDD
30kΩ
SDIO/DCS
350Ω
30kΩ
375Ω
07938-044
07938-041
RBIAS
AND VCM
Figure 31. Equivalent SDIO/DCS Input Circuit
Figure 34. Equivalent RBIAS, VCM Circuit
Rev. A | Page 18 of 36
Data Sheet
AD9251
DRVDD
AVDD
AVDD
30kΩ
375Ω
VREF
07938-045
7.5kΩ
Figure 35. Equivalent CSB Input Circuit
Figure 37. Equivalent VREF Circuit
AVDD
375Ω
07938-046
SENSE
Figure 36. Equivalent SENSE Circuit
Rev. A | Page 19 of 36
07938-047
350Ω
CSB
AD9251
Data Sheet
THEORY OF OPERATION
ANALOG INPUT CONSIDERATIONS
The analog input to the AD9251 is a differential switchedcapacitor circuit designed for processing differential input
signals. This circuit can support a wide common-mode range
while maintaining excellent performance. By using an input
common-mode voltage of midsupply, users can minimize
signal-dependent errors and achieve optimum performance.
H
In nondiversity applications, the AD9251 can be used as a baseband or direct downconversion receiver, where one ADC is
used for I input data and the other is used for Q input data.
CPAR
H
VIN+x
CSAMPLE
Synchronization capability is provided to allow synchronized
timing between multiple channels or multiple devices.
S
S
S
S
CSAMPLE
VIN–x
Programming and control of the AD9251 is accomplished using
a 3-bit SPI-compatible serial interface.
H
CPAR
H
ADC ARCHITECTURE
The AD9251 architecture consists of a multistage, pipelined ADC.
Each stage provides sufficient overlap to correct for flash errors in
the preceding stage. The quantized outputs from each stage are
combined into a final 14-bit result in the digital correction logic.
The pipelined architecture permits the first stage to operate with a
new input sample while the remaining stages operate with
preceding samples. Sampling occurs on the rising edge of
the clock.
Each stage of the pipeline, excluding the last, consists of a low
resolution flash ADC connected to a switched-capacitor DAC
and an interstage residue amplifier (for example, a multiplying
digital-to-analog converter (MDAC)). The residue amplifier
magnifies the difference between the reconstructed DAC output
and the flash input for the next stage in the pipeline. One bit of
redundancy is used in each stage to facilitate digital correction
of flash errors. The last stage simply consists of a flash ADC.
The output staging block aligns the data, corrects errors, and
passes the data to the CMOS output buffers. The output buffers
are powered from a separate (DRVDD) supply, allowing
adjustment of the output voltage swing. During power-down,
the output buffers go into a high impedance state.
07938-006
The AD9251 dual ADC design can be used for diversity
reception of signals, where the ADCs are operating identically
on the same carrier but from two separate antennae. The ADCs
can also be operated with independent analog inputs. The user
can sample any fS/2 frequency segment from dc to 200 MHz,
using appropriate low-pass or band-pass filtering at the ADC
inputs with little loss in ADC performance. Operation to
300 MHz analog input is permitted but occurs at the expense
of increased ADC noise and distortion.
Figure 38. Switched-Capacitor Input Circuit
The clock signal alternately switches the input circuit between
sample-and-hold mode (see Figure 38). When the input circuit
is switched to sample mode, the signal source must be capable
of charging the sample capacitors and settling within one-half
of a clock cycle. A small resistor in series with each input can
help reduce the peak transient current injected from the output
stage of the driving source. In addition, low Q inductors or ferrite
beads can be placed on each leg of the input to reduce high
differential capacitance at the analog inputs and, therefore,
achieve the maximum bandwidth of the ADC. Such use of low
Q inductors or ferrite beads is required when driving the converter
front end at high IF frequencies. Either a shunt capacitor or two
single-ended capacitors can be placed on the inputs to provide a
matching passive network. This ultimately creates a low-pass
filter at the input to limit unwanted broadband noise. See the
AN-742 Application Note, the AN-827 Application Note, and the
Analog Dialogue article “Transformer-Coupled Front-End for
Wideband A/D Converters” (Volume 39, April 2005) for more
information. In general, the precise values depend on the
application.
Rev. A | Page 20 of 36
Data Sheet
AD9251
VIN
The analog inputs of the AD9251 are not internally dc-biased.
Therefore, in ac-coupled applications, the user must provide a
dc bias externally. Setting the device so that VCM = AVDD/2 is
recommended for optimum performance, but the device can
function over a wider range with reasonable performance, as
shown in Figure 39 and Figure 40.
An on-board, common-mode voltage reference is included in
the design and is available from the VCM pin. The VCM pin
must be decoupled to ground by a 0.1 μF capacitor, as described
in the Applications Information section.
76.8Ω
33Ω
10pF
ADA4938
0.1µF
VIN–x
AVDD
90Ω
120Ω
33Ω
ADC
VCM
VIN+x
200Ω
07938-007
200Ω
Input Common Mode
Figure 41. Differential Input Configuration Using the ADA4938-2
For baseband applications below ~10 MHz where SNR is a key
parameter, differential transformer-coupling is the recommended
input configuration. An example is shown in Figure 42. To bias
the analog input, the VCM voltage can be connected to the
center tap of the secondary winding of the transformer.
100
SFDR (dBc)
2V p-p
49.9Ω
C
R
80
ADC
VIN–x
VCM
07938-008
SNR/SFDR (dBFS/dBc)
90
0.1µF
SNR (dBFS)
Figure 42. Differential Transformer-Coupled Configuration
70
The signal characteristics must be considered when selecting
a transformer. Most RF transformers saturate at frequencies
below a few megahertz (MHz). Excessive signal power can also
cause core saturation, which leads to distortion.
60
0.6
0.7
0.8
0.9
1.0
1.1
INPUT COMMON-MODE VOLTAGE (V)
1.2
1.3
At input frequencies in the second Nyquist zone and above, the
noise performance of most amplifiers is not adequate to achieve
the true SNR performance of the AD9251. For applications above
~10 MHz where SNR is a key parameter, differential double balun
coupling is the recommended input configuration (see Figure 44).
07938-049
50
0.5
Figure 39. SNR/SFDR vs. Input Common-Mode Voltage,
fIN = 32.1 MHz, fS = 80 MSPS
100
An alternative to using a transformer-coupled input at frequencies
in the second Nyquist zone is to use the AD8352 differential driver.
An example is shown in Figure 45. See the AD8352 data sheet
for more information.
SFDR (dBc)
90
SNR/SFDR (dBFS/dBc)
VIN+x
R
In any configuration, the value of Shunt Capacitor C is dependent
on the input frequency and source impedance and may need to
be reduced or removed. Table 9 displays the suggested values to set
the RC network. However, these values are dependent on the
input signal and should be used only as a starting guide.
80
SNR (dBFS)
70
60
50
0.5
0.6
0.7
0.8
0.9
1.0
1.1
INPUT COMMON-MODE VOLTAGE (V)
1.2
1.3
07938-050
Table 9. Example RC Network
Frequency Range (MHz)
0 to 70
70 to 200
Figure 40. SNR/SFDR vs. Input Common-Mode Voltage,
fIN = 10.3 MHz, fS = 20 MSPS
Differential Input Configurations
Optimum performance is achieved while driving the AD9251 in a
differential input configuration. For baseband applications, the
AD8138, ADA4937-2, and ADA4938-2 differential drivers provide
excellent performance and a flexible interface to the ADC.
The output common-mode voltage of the ADA4938-2 is easily
set with the VCM pin of the AD9251 (see Figure 41), and the
driver can be configured in a Sallen-Key filter topology to
provide band limiting of the input signal.
Rev. A | Page 21 of 36
R Series
(Ω Each)
33
125
C Differential (pF)
22
Open
AD9251
Data Sheet
AVDD
10µF
1kΩ
A single-ended input can provide adequate performance in
cost-sensitive applications. In this configuration, SFDR and
distortion performance degrade due to the large input commonmode swing. If the source impedances on each input are matched,
there should be little effect on SNR performance. Figure 43
shows a typical single-ended input configuration.
1V p-p
49.9Ω
R
VIN+x
1kΩ
AVDD
ADC
C
1kΩ
R
VIN–x
10µF
0.1µF
1kΩ
Figure 43. Single-Ended Input Configuration
0.1µF
0.1µF
0.1µF
R
VIN+x
2V p-p
25Ω
S
S
P
ADC
C
0.1µF
25Ω
0.1µF
R
VIN–x
VCM
07938-010
PA
Figure 44. Differential Double Balun Input Configuration
VCC
ANALOG INPUT
0Ω
16
1
8, 13
11
RD
RG
3
ANALOG INPUT
0.1µF 0Ω
VIN+x
C
AD8352
10
4
5
R
200Ω
2
CD
0.1µF
0.1µF
0.1µF
200Ω
ADC
R
VIN–x
14
0.1µF
0.1µF
Figure 45. Differential Input Configuration Using the AD8352
Rev. A | Page 22 of 36
VCM
07938-011
0.1µF
07938-009
Single-Ended Input Configuration
Data Sheet
AD9251
Internal Reference Connection
A comparator within the AD9251 detects the potential at the
SENSE pin and configures the reference into two possible
modes, which are summarized in Table 10. If SENSE is grounded,
the reference amplifier switch is connected to the internal resistor
divider (see Figure 46), setting VREF to 1.0 V.
0
VIN+A/VIN+B
–0.5
–1.0
INTERNAL VREF = 0.993V
–1.5
–2.0
–2.5
–3.0
VIN–A/VIN–B
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
LOAD CURRENT (mA)
Figure 47. VREF Accuracy vs. Load Current
ADC
CORE
VREF
1.0µF
0.1µF
SELECT
LOGIC
SENSE
ADC
07938-012
0.5V
Figure 46. Internal Reference Configuration
Table 10. Reference Configuration Summary
Selected Mode
Fixed Internal Reference
Fixed External Reference
SENSE Voltage (V)
AGND to 0.2
AVDD
Resulting VREF (V)
1.0 internal
1.0 applied to external VREF pin
Rev. B | Page 23 of 36
Resulting Differential Span (V p-p)
2.0
2.0
07938-014
A stable and accurate 1.0 V voltage reference is built into the
AD9251. The VREF can be configured using either the internal
1.0 V reference or an externally applied 1.0 V reference voltage.
The various reference modes are summarized in the sections that
follow. The Reference Decoupling section describes the best
practices PCB layout of the reference.
If the internal reference of the AD9251 is used to drive multiple
converters to improve gain matching, the loading of the reference
by the other converters must be considered. Figure 47 shows
how the internal reference voltage is affected by loading.
REFERENCE VOLTAGE ERROR (%)
VOLTAGE REFERENCE
AD9251
Data Sheet
External Reference Operation
Clock Input Options
The use of an external reference may be necessary to enhance
the gain accuracy of the ADC or improve thermal drift characteristics. Figure 48 shows the typical drift characteristics of the
internal reference in 1.0 V mode.
The AD9251 has a very flexible clock input structure. The clock
input can be a CMOS, LVDS, LVPECL, or sine wave signal.
Regardless of the type of signal being used, clock source jitter is
of the most concern, as described in the Jitter Considerations
section.
4
Figure 50 and Figure 51 show two preferred methods for clocking the AD9251 (at clock rates up to 625 MHz). A low jitter clock
source is converted from a single-ended signal to a differential
signal using either an RF transformer or an RF balun.
3
2
0
The RF balun configuration is recommended for clock frequencies
between 125 MHz and 625 MHz, and the RF transformer is recommended for clock frequencies from 10 MHz to 200 MHz. The
back-to-back Schottky diodes across the transformer/balun
secondary limit clock excursions into the AD9251 to
approximately 0.8 V p-p differential.
–2
–3
–4
–6
–40
–20
0
20
40
TEMPERATURE (°C)
60
80
07938-052
–5
Figure 48. Typical VREF Drift
When the SENSE pin is tied to AVDD, the internal reference is
disabled, allowing the use of an external reference. An internal
reference buffer loads the external reference with an equivalent
7.5 kΩ load (see Figure 37). The internal buffer generates the
positive and negative full-scale references for the ADC core.
Therefore, the external reference must be limited to a maximum
of 1.0 V.
This limit helps prevent the large voltage swings of the clock
from feeding through to other portions of the AD9251 while
preserving the fast rise and fall times of the signal that are critical
to a low jitter performance.
Mini-Circuits®
ADT1-1WT, 1:1 Z
0.1µF
CLOCK
INPUT
CLK+
100Ω
50Ω
ADC
0.1µF
SCHOTTKY
DIODES:
HSMS2822
0.1µF
Figure 50. Transformer-Coupled Differential Clock (Up to 200 MHz)
1nF
CLOCK
INPUT
AVDD
0.1µF
CLK+
50Ω
ADC
0.1µF
1nF
CLK–
SCHOTTKY
DIODES:
HSMS2822
0.9V
CLK+
Figure 51. Balun-Coupled Differential Clock (Up to 625 MHz)
CLK–
2pF
07938-016
2pF
0.1µF
CLK–
CLOCK INPUT CONSIDERATIONS
For optimum performance, clock the AD9251 sample clock
inputs, CLK+ and CLK−, with a differential signal. The signal
is typically ac-coupled into the CLK+ and CLK− pins via a
transformer or capacitors. These pins are biased internally
(see Figure 49) and require no external bias.
XFMR
07938-017
–1
Figure 49. Equivalent Clock Input Circuit
Rev. B | Page 24 of 36
07938-018
VREF ERROR (mV)
VREF ERROR (mV)
1
Data Sheet
AD9251
If a low jitter clock source is not available, another option is to
ac couple a differential PECL signal to the sample clock input
pins, as shown in Figure 52. The AD9510/AD9511/AD9512/
AD9513/AD9514/AD9515/AD9516/AD9517 clock drivers offer
excellent jitter performance.
0.1µF
CLOCK
INPUT
CLK+
0.1µF
50kΩ
50kΩ
AD951x
PECL DRIVER
240Ω
100Ω
0.1µF
ADC
CLK–
07938-019
CLOCK
INPUT
0.1µF
240Ω
Figure 52. Differential PECL Sample Clock (Up to 625 MHz)
A third option is to ac couple a differential LVDS signal to the
sample clock input pins, as shown in Figure 53. The AD9510/
AD9511/AD9512/AD9513/AD9514/AD9515/AD9516/AD9517
clock drivers offer excellent jitter performance.
0.1µF
CLOCK
INPUT
CLK+
0.1µF
50kΩ
AD951x
LVDS DRIVER
100Ω
0.1µF
ADC
CLK–
07938-020
CLOCK
INPUT
0.1µF
50kΩ
Figure 53. Differential LVDS Sample Clock (Up to 625 MHz)
In some applications, it may be acceptable to drive the sample
clock inputs with a single-ended 1.8 V CMOS signal. In such
applications, drive the CLK+ pin directly from a CMOS gate, and
bypass the CLK− pin to ground with a 0.1 μF capacitor (see
Figure 54).
CLOCK
INPUT
50Ω 1
1kΩ
AD951x
CMOS DRIVER
OPTIONAL
0.1µF
100Ω
1kΩ
The AD9251 contains an input clock divider with the ability
to divide the input clock by integer values between 1 and 8.
Optimum performance is obtained by enabling the internal
duty cycle stabilizer (DCS) when using divide ratios other than
1, 2, or 4.
The AD9251 clock divider can be synchronized using the
external SYNC input. Bit 1 and Bit 2 of Register 0x100 allow the
clock divider to be resynchronized on every SYNC signal or
only on the first SYNC signal after the register is written. A
valid SYNC causes the clock divider to reset to its initial state.
This synchronization feature allows multiple parts to have their
clock dividers aligned to guarantee simultaneous input sampling.
Clock Duty Cycle
Typical high speed ADCs use both clock edges to generate
a variety of internal timing signals and, as a result, may be
sensitive to clock duty cycle. Commonly, a ±5% tolerance is
required on the clock duty cycle to maintain dynamic
performance characteristics.
The AD9251 contains a duty cycle stabilizer (DCS) that retimes
the nonsampling (falling) edge, providing an internal clock
signal with a nominal 50% duty cycle. This allows the user to
provide a wide range of clock input duty cycles without affecting
the performance of the AD9251. Noise and distortion performance are nearly flat for a wide range of duty cycles with the DCS
on, as shown in Figure 55.
Jitter in the rising edge of the input is still of concern and is not
easily reduced by the internal stabilization circuit. The duty
cycle control loop does not function for clock rates less than
20 MHz, nominally. The loop has a time constant associated
with it that must be considered in applications in which the
clock rate can change dynamically. A wait time of 1.5 μs to 5 μs
is required after a dynamic clock frequency increase or decrease
before the DCS loop is relocked to the input signal.
VCC
0.1µF
Input Clock Divider
CLK+
ADC
80
DCS ON
CLK–
RESISTOR IS OPTIONAL.
70
SNR (dBFS)
Figure 54. Single-Ended 1.8 V CMOS Input Clock (Up to 200 MHz)
65
DCS OFF
60
55
50
45
40
10
20
30
40
50
60
POSITIVE DUTY CYCLE (%)
Figure 55. SNR vs. DCS On/Off
Rev. B | Page 25 of 36
70
80
07938-053
150Ω
75
07938-021
0.1µF
AD9251
Data Sheet
Jitter Considerations
The maximum DRVDD current (IDRVDD) can be calculated as
High speed, high resolution ADCs are sensitive to the quality
of the clock input. The degradation in SNR from the low frequency SNR (SNRLF) at a given input frequency (fINPUT) due to
jitter (tJRMS) can be calculated by
SNRHF = −10 log[(2π × fINPUT × tJRMS)2 + 10 ( − SNRLF /10) ]
In the previous equation, the rms aperture jitter represents the
clock input jitter specification. IF undersampling applications
are particularly sensitive to jitter, as illustrated in Figure 56.
80
75
where N is the number of output bits (30, in the case of the
AD9251).
This maximum current occurs when every output bit switches
on every clock cycle, that is, a full-scale square wave at the Nyquist
frequency of fCLK/2. In practice, the DRVDD current is established by the average number of output bits switching, which
is determined by the sample rate and the characteristics of the
analog input signal.
Reducing the capacitive load presented to the output drivers can
minimize digital power consumption. The data in Figure 57 was
taken using the same operating conditions as those used for the
Typical Performance Characteristics, with a 5 pF load on each
output driver.
0.05ps
70
0.2ps
65
150
0.5ps
55
ANALOG CORE POWER (mW)
1.0ps
1.5ps
50
3.0ps
45
1
10
2.0ps
2.5ps
100
FREQUENCY (MHz)
1k
Figure 56. SNR vs. Input Frequency and Jitter
The clock input should be treated as an analog signal in cases
where aperture jitter may affect the dynamic range of the AD9251.
To avoid modulating the clock signal with digital noise, keep
power supplies for clock drivers separate from the ADC output
driver supplies. Low jitter, crystal-controlled oscillators make the
best clock sources. If the clock is generated from another type of
source (by gating, dividing, or another method), it should be
retimed by the original clock at the last step.
See the AN-501 Application Note and the AN-756 Application
Note available on www.analog.com for more information.
CHANNEL/CHIP SYNCHRONIZATION
The AD9251 has a SYNC input that offers the user flexible
synchronization options for synchronizing sample clocks
across multiple ADCs. The input clock divider can be enabled
to synchronize on a single occurrence of the SYNC signal or on
every occurrence. The SYNC input is internally synchronized
to the sample clock; however, to ensure there is no timing
uncertainty between multiple parts, the SYNC input signal should
be externally synchronized to the input clock signal, meeting the
setup and hold times shown in Table 5. Drive the SYNC input
using a single-ended CMOS-type signal.
POWER DISSIPATION AND STANDBY MODE
As shown in Figure 57, the analog core power dissipated by
the AD9251 is proportional to its sample rate. The digital
power dissipation of the CMOS outputs are determined
primarily by the strength of the digital drivers and the load
on each output bit.
130
AD9251-80
110
AD9251-65
90
AD9251-40
70
AD9251-20
50
0
10
20
30
40
50
CLOCK RATE (MSPS)
60
70
80
07938-051
60
07938-022
SNR (dBFS)
IDRVDD = VDRVDD × CLOAD × fCLK × N
Figure 57. Analog Core Power vs. Clock Rate
The AD9251 is placed in power-down mode either by the SPI
port or by asserting the PDWN pin high. In this state, the ADC
typically dissipates 2.2 mW. During power-down, the output
drivers are placed in a high impedance state. Asserting the
PDWN pin low returns the AD9251 to its normal operating
mode. Note that PDWN is referenced to the digital output
driver supply (DRVDD) and should not exceed that supply
voltage.
Low power dissipation in power-down mode is achieved by
shutting down the reference, reference buffer, biasing networks,
and clock. Internal capacitors are discharged when entering powerdown mode and then must be recharged when returning to normal
operation. As a result, wake-up time is related to the time spent
in power-down mode, and shorter power-down cycles result in
proportionally shorter wake-up times.
When using the SPI port interface, the user can place the ADC
in power-down mode or standby mode. Standby mode allows
the user to keep the internal reference circuitry powered when
faster wake-up times are required. See the Memory Map section
for more details.
Rev. B | Page 26 of 36
Data Sheet
AD9251
DIGITAL OUTPUTS
The AD9251 output drivers can be configured to interface with
1.8 V to 3.3 V CMOS logic families. Output data can also be
multiplexed onto a single output bus to reduce the total number of
traces required.
The CMOS output drivers are sized to provide sufficient output
current to drive a wide variety of logic families. However, large
drive currents tend to cause current glitches on the supplies and
may affect converter performance.
Applications requiring the ADC to drive large capacitive loads
or large fanouts may require external buffers or latches.
The output data format can be selected to be either offset binary
or twos complement by setting the SCLK/DFS pin when operating
in the external pin mode (see Table 11).
As detailed in the AN-877 Application Note, Interfacing to High
Speed ADCs via SPI, the data format can be selected for offset
binary, twos complement, or gray code when using the SPI control.
Table 11. SCLK/DFS Mode Selection (External Pin Mode)
Voltage at Pin
AGND
DRVDD
SCLK/DFS
Offset binary (default)
Twos complement
SDIO/DCS
DCS disabled
DCS enabled (default)
Digital Output Enable Function (OEB)
The AD9251 has a flexible three-state ability for the digital
output pins. The three-state mode is enabled using the OEB pin
or through the SPI interface. If the OEB pin is low, the output
data drivers and DCOs are enabled. If the OEB pin is high, the
output data drivers and DCOs are placed in a high impedance
state. This OEB function is not intended for rapid access to the
data bus. Note that OEB is referenced to the digital output
driver supply (DRVDD) and should not exceed that supply
voltage.
When using the SPI interface, the data outputs and DCO of
each channel can be independently three-stated by using the
output disable (OEB) bit (Bit 4) in Register 0x14.
TIMING
The AD9251 provides latched data with a pipeline delay of
9 clock cycles. Data outputs are available one propagation delay
(tPD) after the rising edge of the clock signal.
Minimize the length of the output data lines and loads placed
on them to reduce transients within the AD9251. These
transients can degrade converter dynamic performance.
The lowest typical conversion rate of the AD9251 is 3 MSPS. At
clock rates below 3 MSPS, dynamic performance can degrade.
Data Clock Output (DCO)
The AD9251 provides two data clock output (DCO) signals
intended for capturing the data in an external register. The CMOS
data outputs are valid on the rising edge of the DCO, unless the
DCO clock polarity has been changed via the SPI. See Figure 2
and Figure 3 for a graphical timing description.
Table 12. Output Data Format
Input (V)
VIN+ − VIN−
VIN+ − VIN−
VIN+ − VIN−
VIN+ − VIN−
VIN+ − VIN−
Condition (V)
< −VREF − 0.5 LSB
= −VREF
=0
= +VREF − 1.0 LSB
> +VREF − 0.5 LSB
Offset Binary Output Mode
00 0000 0000 0000
00 0000 0000 0000
10 0000 0000 0000
11 1111 1111 1111
11 1111 1111 1111
Rev. B | Page 27 of 36
Twos Complement Mode
10 0000 0000 0000
10 0000 0000 0000
00 0000 0000 0000
01 1111 1111 1111
01 1111 1111 1111
OR
1
0
0
0
1
AD9251
Data Sheet
BUILT-IN SELF-TEST (BIST) AND OUTPUT TEST
The AD9251 includes a built-in test feature designed to enable
verification of the integrity of each channel, as well as to
facilitate board level debugging. A built-in self-test (BIST) feature
that verifies the integrity of the digital datapath of the AD9251
is included. Various output test options are also provided to place
predictable values on the outputs of the AD9251.
BUILT-IN SELF-TEST (BIST)
The BIST is a thorough test of the digital portion of the selected
AD9251 signal path. Perform the BIST test after a reset to ensure
the part is in a known state. During BIST, data from an internal
pseudorandom noise (PN) source is driven through the digital
datapath of both channels, starting at the ADC block output. At
the datapath output, CRC logic calculates a signature from the
data. The BIST sequence runs for 512 cycles and then stops.
Once completed, the BIST compares the signature results with a
pre-determined value. If the signatures match, the BIST sets Bit 0
of Register 0x24, signifying the test passed. If the BIST test fails,
Bit 0 of Register 0x24 is cleared. The outputs are connected
during this test, so the PN sequence can be observed as it runs.
Writing the value 0x05 to Register 0x0E runs the BIST. This enables
the Bit 0 (BIST enable) of Register 0x0E and resets the PN sequence
generator, Bit 2 (BIST INIT) of Register 0x0E. At the completion of
the BIST, Bit 0 of Register 0x24 is automatically cleared. The PN
sequence can be continued from its last value by writing a 0 in
Bit 2 of Register 0x0E. However, if the PN sequence is not reset,
the signature calculation does not equal the predetermined
value at the end of the test. At that point, the user needs to rely
on verifying the output data.
OUTPUT TEST MODES
The output test options are described in Table 16 at Address
0x0D. When an output test mode is enabled, the analog section
of the ADC is disconnected from the digital back-end blocks
and the test pattern is run through the output formatting block.
Some of the test patterns are subject to output formatting, and
some are not. The PN generators from the PN sequence tests
can be reset by setting Bit 4 or Bit 5 of Register 0x0D. These
tests can be performed with or without an analog signal (if
present, the analog signal is ignored), but they do require an
encode clock. For more information, see the AN-877
Application Note, Interfacing to High Speed ADCs via SPI.
Rev. B | Page 28 of 36
Data Sheet
AD9251
SERIAL PORT INTERFACE (SPI)
The AD9251 serial port interface (SPI) allows the user to configure
the converter for specific functions or operations through a
structured register space provided inside the ADC. The SPI
gives the user added flexibility and customization, depending
on the application. Addresses are accessed via the serial port
and can be written to or read from via the port. Memory is
organized into bytes that can be further divided into fields,
which are documented in the Memory Map section. For
detailed operational information, see AN-877 Application
Note, Interfacing to High Speed ADCs via SPI.
The falling edge of CSB, in conjunction with the rising edge of
SCLK, determines the start of the framing. An example of the
serial timing and its definitions can be found in Figure 58 and
Table 5.
CONFIGURATION USING THE SPI
During an instruction phase, a 16-bit instruction is transmitted.
Data follows the instruction phase, and its length is determined
by the W0 and W1 bits as shown in Figure 58.
Other modes involving the CSB are available. The CSB can be
held low indefinitely, which permanently enables the device;
this is called streaming. The CSB can stall high between bytes to
allow for additional external timing. When CSB is tied high, SPI
functions are placed in high impedance mode. This mode turns
on any SPI pin secondary functions.
Three pins define the SPI of this ADC: the SCLK, the SDIO, and
the CSB (see Table 13). The SCLK (a serial clock) is used to
synchronize the read and write data presented from and to the
ADC. The SDIO (serial data input/output) is a dual-purpose
pin that allows data to be sent and read from the internal ADC
memory map registers. The CSB (chip select bar) is an activelow control that enables or disables the read and write cycles.
All data is composed of 8-bit words. The first bit of the first byte in
a multibyte serial data transfer frame indicates whether a read
command or a write command is issued. This allows the serial
data input/output (SDIO) pin to change direction from an input
to an output at the appropriate point in the serial frame.
In addition to word length, the instruction phase determines
whether the serial frame is a read or write operation, allowing
the serial port to be used both to program the chip and to read
the contents of the on-chip memory. If the instruction is a readback
operation, performing a readback causes the serial data input/
output (SDIO) pin to change direction from an input to an output
at the appropriate point in the serial frame.
Table 13. Serial Port Interface Pins
Pin
SCLK
SDIO
CSB
Function
Serial Clock. The serial shift clock input, which is used to
synchronize serial interface reads and writes.
Serial Data Input/Output. A dual-purpose pin that
typically serves as an input or an output, depending on
the instruction being sent and the relative position in the
timing frame.
Chip Select Bar. An active-low control that gates the read
and write cycles.
tHIGH
tDS
tS
tDH
Data can be sent in MSB-first mode or in LSB-first mode. MSB
first is the default on power-up and can be changed via the SPI
port configuration register. For more information about this
and other features, see the AN-877 Application Note, Interfacing
to High Speed ADCs via SPI.
tCLK
tH
tLOW
CSB
SCLK DON’T CARE
R/W
W1
W0
A12
A11
A10
A9
A8
A7
D5
D4
D3
D2
D1
D0
DON’T CARE
07938-023
SDIO DON’T CARE
DON’T CARE
Figure 58. Serial Port Interface Timing Diagram
Rev. B | Page 29 of 36
AD9251
Data Sheet
HARDWARE INTERFACE
The pins described in Table 13 constitute the physical interface
between the programming device of the user and the serial port
of the AD9251. The SCLK pin and the CSB pin function as
inputs when using the SPI interface. The SDIO pin is
bidirectional, functioning as an input during write phases and
as an output during readback.
feature control. In this mode, connect the CSB chip select to
AVDD, which disables the serial port interface.
Table 14. Mode Selection
Pin
SDIO/DCS
SCLK/DFS
The SPI interface is flexible enough to be controlled by
either FPGAs or microcontrollers. One method for SPI
configuration is described in detail in the AN-812 Application Note, Microcontroller-Based Serial Port Interface
(SPI) Boot Circuit.
OEB
PDWN
The SPI port should not be active during periods when the full
dynamic performance of the converter is required. Because the
SCLK signal, the CSB signal, and the SDIO signal are typically
asynchronous to the ADC clock, noise from these signals can
degrade converter performance. If the on-board SPI bus is used for
other devices, it may be necessary to provide buffers between
this bus and the AD9251 to prevent these signals from transitioning at the converter inputs during critical sampling periods.
SDIO/DCS and SCLK/DFS serve a dual function when the
SPI interface is not being used. When the pins are strapped to
AVDD or ground during device power-on, they are associated
with a specific function. The Digital Outputs section describes
the strappable functions supported on the AD9251.
CONFIGURATION WITHOUT THE SPI
In applications that do not interface to the SPI control registers,
the SDIO/DCS pin, the SCLK/DFS pin, the OEB pin, and the
PDWN pin serve as standalone CMOS-compatible control pins.
When the device is powered up, it is assumed that the user
intends to use the pins as static control lines for the duty cycle
stabilizer, output data format, output enable, and power-down
External
Voltage
AVDD (default)
AGND
AVDD
AGND (default)
AVDD
AGND (default)
AVDD
AGND (default)
Configuration
Duty cycle stabilizer enabled
Duty cycle stabilizer disabled
Twos complement enabled
Offset binary enabled
Outputs in high impedance
Outputs enabled
Chip in power-down or standby
Normal operation
SPI ACCESSIBLE FEATURES
Table 15 provides a brief description of the general features that
are accessible via the SPI. These features are described in detail
in the AN-877 Application Note, Interfacing to High Speed ADCs
via SPI. The AD9251 part-specific features are described in
detail in Table 16.
Table 15. Features Accessible Using the SPI
Feature
Mode
Clock
Offset
Test I/O
Output Mode
Output Phase
Output Delay
Rev. B | Page 30 of 36
Description
Allows the user to set either power-down mode
or standby mode
Allows the user to access the DCS via the SPI
Allows the user to digitally adjust the
converter offset
Allows the user to set test modes to have known
data on output bits
Allows the user to set up outputs
Allows the user to set the output clock polarity
Allows the user to vary the DCO delay
Data Sheet
AD9251
MEMORY MAP
READING THE MEMORY MAP REGISTER TABLE
DEFAULT VALUES
Each row in the memory map register table (see Table 16) has
eight bit locations. The memory map is roughly divided into
four sections: the chip configuration registers (Address 0x00 to
Address 0x02); the device index and transfer registers (Address
0x05 and Address 0xFF); the program registers, including setup,
control, and test (Address 0x08 to Address 0x2E); and the
digital feature control registers (Address 0x100 and Address
0x101).
After the AD9251 is reset, critical registers are loaded with
default values. The default values for the registers are given in
the memory map register table (see Table 16).
Table 16 documents the default hexadecimal value for each
hexadecimal address shown. The column with the heading Bit 7
(MSB) is the start of the default hexadecimal value given. For
example, Address 0x05, the Channel Index register, has a
hexadecimal default value of 0x03. This means that in Address
0x05 Bit[7:2] = 0, and the remaining Bits[1:0] = 1. This setting
is the default channel index setting. The default value results in
both ADC channels receiving the next write command. For
more information on this function and others, see the AN-877
Application Note, Interfacing to High Speed ADCs via SPI. This
document details the functions controlled by Register 0x00 to
register 0xFF. The remaining registers, Register 0x100 and
Register 0x101, are documented in the Memory Map Register
Descriptions section following Table 16.
•
OPEN LOCATIONS
All address and bit locations that are not included in the SPI
map are not currently supported for this device. Unused bits of
a valid address location should be written with 0s. Writing to these
locations is required only when part of an address location is
open (for example, Address 0x05). If the entire address location
is open, it is omitted from the SPI map (for example, Address 0x13)
and should not be written.
Logic Levels
An explanation of logic level terminology follows:
•
“Bit is set” is synonymous with “bit is set to Logic 1” or
“writing Logic 1 for the bit.”
“Clear a bit” is synonymous with “bit is set to Logic 0” or
“writing Logic 0 for the bit.”
Transfer Register Map
Address 0x08 to Address 0x18 are shadowed. Writes to these
addresses do not affect part operation until a transfer command
is issued by writing 0x01 to Address 0xFF, setting the transfer bit.
This allows these registers to be updated internally and simultaneously when the transfer bit is set. The internal update takes
place when the transfer bit is set, and then the bit autoclears.
Channel-Specific Registers
Some channel setup functions can be programmed differently
for each channel. In these cases, channel address locations are
internally duplicated for each channel. These registers and bits
are designated in the memory map register table as local. These
local registers and bits can be accessed by setting the appropriate
Channel A (Bit 0) or Channel B (Bit 1) bits in Register 0x05.
If both bits are set, the subsequent write affects the registers of
both channels. In a read cycle, set only Channel A or Channel B
to read one of the two registers. If both bits are set during an
SPI read cycle, the part returns the value for Channel A.
Registers and bits designated as global in the memory map
register table affect the entire part or the channel features for
which independent settings are not allowed between channels.
The settings in Register 0x05 do not affect the global registers
and bits.
Rev. B | Page 31 of 36
AD9251
Data Sheet
MEMORY MAP REGISTER TABLE
All address and bit locations that are not included in Table 16 are not currently supported for this device.
Table 16.
Address Register
(Hex)
Name
Chip Configuration Registers
SPI port
0x00
configuration
(global)
Bit 7
(MSB)
0x01
Chip ID (global)
8-bit chip ID bits [7:0]
AD9251 = 0x23
0x02
Chip grade
(global)
Open
0
Default
Value
(Hex)
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
(LSB)
LSB
first
Soft reset
1
1
Soft
reset
LSB first
0
0x18
The nibbles are
mirrored so
that LSB- or
MSB-first mode
registers
correctly,
regardless of
shift mode
Unique chip ID
used to differentiate
devices; read
only
Unique speed
grade ID used
to differentiate
devices; read
only
Bits are set to
determine
which device
on chip
receives the
next write
command; the
default is all
devices on chip
Synchronously
transfers data
from the
master shift
register to the
slave
Speed grade ID 6:4
20 MSPS = 000
40 MSPS = 001
65 MSPS = 010
80 MSPS = 011
Open
Device Index and Transfer Registers
0x05
Channel index
Open
Open
Open
Open
Open
Open
ADC B
default
ADC A
default
0x03
0xFF
Open
Open
Open
Open
Open
Open
Transfer
0x00
Open
Open
00 = chip run
01 = full powerdown
10 = standby
11 = chip wide
digital reset
(local)
Duty
Open
cycle
stabilize
Clock divider [2:0]
Clock divide ratio
000 = divide by 1
001 = divide by 2
010 = divide by 3
011 = divide by 4
100 = divide by 5
101 = divide by 6
110 = divide by 7
111 = divide by 8
0x80
Transfer
Open
Program Registers (May or May Not Be Indexed by Device Index)
External External pin function
0x08
Modes
0x00 full powerpowerdown
down
0x01 standby
enable
(local)
(local)
0x09
Clock (global)
Open
0x0B
Clock divide
(global)
Open
Open
Open
Open
Rev. B | Page 32 of 36
Comments
Determines
various generic
modes of chip
operation
0x00
0x00
The divide ratio
is the value
plus 1
Data Sheet
AD9251
Register
Name
Test mode (local)
Bit 7
(MSB)
Bit 6
User test mode
(local)
00 = single
01 = alternate
10 = single once
11 = alternate
once
0x0E
BIST enable
Open
0x10
0x14
Offset adjust
(local)
Output mode
8-bit device offset adjustment [7:0] (local)
Offset adjust in LSBs from +127 to −128 (twos complement format)
Output
Output
00 = 3.3 V CMOS
Output mux
Open
invert
disable
10 = 1.8 V CMOS
enable
(local)
(interleaved) (local)
0x15
OUTPUT_ADJUST
1.8 V DCO
drive strength
00 = 1 stripe
01 = 2 stripes
10 = 3 stripes (default)
11 = 4 stripes
0x16
OUTPUT_PHASE
3.3 V DCO
drive strength
00 = 1 stripe
(default)
01 = 2 stripes
10 = 3 stripes
11 = 4 stripes
DCO
output
polarity
0=
normal
1=
inverted
(local)
0x17
OUTPUT_DELAY
Enable
DCO
delay
Enable
data
delay
0x19
USER_PATT1_LSB
B7
B6
B5
B4
B3
B2
B1
B0
0x00
0x1A
USER_PATT1_MSB
B15
B14
B13
B12
B11
B10
B9
B8
0x00
0x1B
USER_PATT2_LSB
B7
B6
B5
B4
B3
B2
B1
B0
0x00
0x1C
USER_PATT2_MSB
B15
B14
B13
B12
B11
B10
B9
B8
0x00
0x24
MISR_LSB
Open
Open
Open
Open
Open
Open
Open
B0
0x00
Bit 5
Reset PN
long gen
Bit 4
Reset PN
short
gen
Bit 0
(LSB)
Default
Value
(Hex)
0x00
Address
(Hex)
0x0D
Bit 3
Bit 2
Bit 1
Output test mode [3:0] (local)
0000 = off (default)
0001 = midscale short
0010 = positive FS
0011 = negative FS
0100 = alternating checkerboard
0101 = PN 23 sequence
0110 = PN 9 sequence
0111 = one/zero word toggle
1000 = user input
1001 = 1-/0-bit toggle
1010 = 1x sync
1011 = one bit high
1100 = mixed bit frequency
BIST
BIST
Open
Open
INIT
enable
0x00
00 = offset binary
01 = twos
complement
10 = gray code
11 = offset binary
(local)
1.8 V data
3.3 V data
drive strength
drive strength
00 = 1 stripe
00 = 1 stripe
01 = 2 stripes
(default)
10 = 3 stripes
01 = 2 stripes
(default)
10 = 3 stripes
11 = 4 stripes
11 = 4 stripes
Input clock phase adjust [2:0]
(Value is number of input clock
cycles of phase delay)
000 = no delay
001 = 1 input clock cycle
010 = 2 input clock cycles
011 = 3 input clock cycles
100 = 4 input clock cycles
101 = 5 input clock cycles
110 = 6 input clock cycles
111 = 7 input clock cycles
0x00
When Bit 0 is
set, the BIST
function is
initiated
Device offset
trim
Configures the
outputs and
the format of
the data
0x22
Determines
CMOS output
drive strength
properties
0x00
On devices that
utilize global
clock divide,
determines
which phase of
the divider
output is used
to supply the
output clock;
internal
latching is
unaffected
This sets the
fine output
delay of the
output clock
but does not
change internal
timing
0x00
DCO/Data delay [2:0]
000 = 0.56 ns
001 = 1.12 ns
010 = 1.68 ns
011 = 2.24 ns
100 = 2.80 ns
101 = 3.36 ns
110 = 3.92 ns
111 = 4.48 ns
Rev. B | Page 33 of 36
0x00
Comments
When set, the
test data is
placed on the
output pins in
place of normal
data
User-defined
pattern, 1 LSB
User-defined
pattern, 1 MSB
User-defined
pattern, 2 LSB
User-defined
pattern, 2 MSB
Least
significant byte
of MISR; read
only
AD9251
Data Sheet
Register
Name
Features
Bit 7
(MSB)
Open
Bit 6
Open
Bit 5
Open
Bit 4
Open
Bit 3
Open
Bit 2
Open
Bit 1
Open
0x2E
Output assign
Open
Open
Open
Open
Open
Open
Open
0 = ADC A
1 = ADC B
(local)
Ch A =
0x00
Ch B =
0x01
Open
Open
Open
Open
Open
Clock
divider
sync
enable
Master
sync
enable
0x01
Enable
OEB
Pin 47
(local)
Open
Open
Open
Enable
GCLK
detect
Clock
divider
next
sync
only
Run
GCLK
Open
Disable
SDIO pulldown
0x88
Digital Feature Control
Sync control
0x100
(global)
0x101
USR2
Bit 0
(LSB)
OR OE
(local)
Default
Value
(Hex)
0x01
Address
(Hex)
0x2A
Comments
Disable the OR
pin for the
indexed
channel
Assign an ADC
to an output
channel
Enables
internal
oscillator for
clock rates <
5 MHz
MEMORY MAP REGISTER DESCRIPTIONS
USR2 (Register 0x101)
For additional information about functions controlled in
Register 0x00 to Register 0xFF, see the AN-877 Application
Note, Interfacing to High Speed ADCs via SPI.
Bit 7—Enable OEB Pin 47
Normally set high, this bit allows Pin 47 to function as the
output enable. If it is set low, it disables Pin 47.
Sync Control (Register 0x100)
Bits[7:3]—Reserved
Bit 3—Enable GCLK Detect
Bit 2—Clock Divider Next Sync Only
If the master sync enable bit (Address 0x100, Bit 0) and the
clock divider sync enable bit (Address 0x100, Bit 1) are high,
Bit 2 allows the clock divider to sync to the first sync pulse it
receives and to ignore the rest. The clock divider sync enable
bit (Address 0x100, Bit 1) resets after it syncs.
Bit 1—Clock Divider Sync Enable
Bit 1 gates the sync pulse to the clock divider. The sync signal
is enabled when Bit 1 and Bit 0 are high and the device is
operating in continuous sync mode as long as Bit 2 of the
sync control is low.
Normally set high, this bit enables a circuit that detects encode
rates below about 5 MSPS. When a low encode rate is detected,
an internal oscillator, GCLK, is enabled ensuring the proper
operation of several circuits. If set low, the detector is disabled.
Bit 2—Run GCLK
This bit enables the GCLK oscillator. For some applications
with encode rates below 10 MSPS, it may be preferable to set
this bit high to supersede the GCLK detector.
Bit 0—Disable SDIO Pull-Down
This bit can be set high to disable the internal 30 kΩ pull-down
on the SDIO pin, which can be used to limit the loading when
many devices are connected to the SPI bus.
Bit 0—Master Sync Enable
Bit 0 must be high to enable any of the sync functions.
Rev. B | Page 34 of 36
Data Sheet
AD9251
APPLICATIONS INFORMATION
DESIGN GUIDELINES
Before starting design and layout of the AD9251 as a system,
it is recommended that the designer become familiar with these
guidelines, which discuss the special circuit connections and
layout requirements needed for certain pins.
Power and Ground Recommendations
When connecting power to the AD9251, it is strongly
recommended that two separate supplies be used. Use one 1.8 V
supply for analog (AVDD); use a separate 1.8 V to 3.3 V supply for
the digital output supply (DRVDD). If a common 1.8 V AVDD
and DRVDD supply must be used, the AVDD and DRVDD
domains must be isolated with a ferrite bead or filter choke and
separate decoupling capacitors. Several different decoupling
capacitors can be used to cover both high and low frequencies.
Locate these capacitors close to the point of entry at the PCB
level and close to the pins of the part, with minimal trace length.
A single PCB ground plane should be sufficient when using the
AD9251. With proper decoupling and smart partitioning of the
PCB analog, digital, and clock sections, optimum performance
is easily achieved.
Exposed Paddle Thermal Heat Sink Recommendations
The exposed paddle (Pin 0) is the only ground connection for
the AD9251; therefore, it must be connected to analog ground
(AGND) on the PCB of the customer. To achieve the best
electrical and thermal performance, mate an exposed (no solder
mask) continuous copper plane on the PCB to the AD9251
exposed paddle, Pin 0.
The copper plane should have several vias to achieve the
lowest possible resistive thermal path for heat dissipation to
flow through the bottom of the PCB. Fill or plug these vias
with nonconductive epoxy.
To maximize the coverage and adhesion between the ADC and
the PCB, a silkscreen should be overlaid to partition the continuous
plane on the PCB into several uniform sections. This provides
several tie points between the ADC and the PCB during the reflow
process. Using one continuous plane with no partitions guarantees
only one tie point between the ADC and the PCB. For detailed
information about packaging and PCB layout of chip scale
packages, see the AN-772 Application Note, A Design and
Manufacturing Guide for the Lead Frame Chip Scale Package
(LFCSP), at www.analog.com.
VCM
The VCM pin should be decoupled to ground with a 0.1 μF
capacitor, as shown in Figure 42.
RBIAS
The AD9251 requires that a 10 kΩ resistor be placed between
the RBIAS pin and ground. This resistor sets the master current
reference of the ADC core and should have at least a 1% tolerance.
Reference Decoupling
The VREF pin should be externally decoupled to ground with a
low ESR, 1.0 μF capacitor in parallel with a low ESR, 0.1 μF
ceramic capacitor.
SPI Port
The SPI port should not be active during periods when the full
dynamic performance of the converter is required. Because the
SCLK, CSB, and SDIO signals are typically asynchronous to the
ADC clock, noise from these signals can degrade converter
performance. If the on-board SPI bus is used for other devices,
it may be necessary to provide buffers between this bus and the
AD9251 to keep these signals from transitioning at the converter
inputs during critical sampling periods.
Rev. B | Page 35 of 36
AD9251
Data Sheet
OUTLINE DIMENSIONS
9.10
9.00 SQ
8.90
0.60
0.42
0.24
0.60
0.42
0.24
0.30
0.23
0.18
64 1
49
48
PIN 1
INDICATOR
PIN 1
INDICATOR
8.85
8.75 SQ
8.65
0.50
BSC
0.50
0.40
0.30
0.25 MIN
7.50 REF
0.80 MAX
0.65 NOM
0.05 MAX
0.02 NOM
SEATING
PLANE
PKG-1184
16
17
0.20 REF
FOR PROPER CONNECTION OF
THE EXPOSED PAD, REFER TO
THE PIN CONFIGURATION AND
FUNCTION DESCRIPTIONS
SECTION OF THIS DATA SHEET.
COMPLIANT TO JEDEC STANDARDS MO-220-VMMD-4
01-22-2015-D
12° MAX
33
32
BOTTOM VIEW
TOP VIEW
1.00
0.85
0.80
6.35
6.20 SQ
6.05
EXPOSED
PAD
Figure 59. 64-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
9 mm × 9 mm Body, Very Thin Quad (CP-64-4)
Dimensions shown in millimeters
ORDERING GUIDE
Model 1, 2
AD9251BCPZ-80
AD9251BCPZRL7-80
AD9251BCPZ-65
AD9251BCPZRL7-65
AD9251BCPZ-40
AD9251BCPZRL7-40
AD9251BCPZ-20
AD9251BCPZRL7-20
AD9251-80EBZ
AD9251-65EBZ
AD9251-40EBZ
AD9251-20EBZ
1
2
Temperature Range
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
Package Description
64-Lead Lead Frame Chip Scale Package (LFCSP_VQ)
64-Lead Lead Frame Chip Scale Package (LFCSP_VQ)
64-Lead Lead Frame Chip Scale Package (LFCSP_VQ)
64-Lead Lead Frame Chip Scale Package (LFCSP_VQ)
64-Lead Lead Frame Chip Scale Package (LFCSP_VQ)
64-Lead Lead Frame Chip Scale Package (LFCSP_VQ)
64-Lead Lead Frame Chip Scale Package (LFCSP_VQ)
64-Lead Lead Frame Chip Scale Package (LFCSP_VQ)
Evaluation Board
Evaluation Board
Evaluation Board
Evaluation Board
Z = RoHS Compliant Part.
The exposed paddle (Pin 0) is the only GND connection on the chip and must be connected to the PCB AGND.
©2009–2016 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D07938-0-9/16(B)
Rev. B | Page 36 of 36
Package Option
CP-64-4
CP-64-4
CP-64-4
CP-64-4
CP-64-4
CP-64-4
CP-64-4
CP-64-4