Burr-Brown OPA693IDR Ultra-wideband, fixed gain video buffer amplifier with disable Datasheet

OPA
693
OPA6
93
OPA693
SBOS285 – OCTOBER 2003
Ultra-Wideband, Fixed Gain
Video BUFFER AMPLIFIER with Disable
FEATURES
DESCRIPTION
● VERY HIGH BANDWIDTH (G = +2): 700MHz
● FLEXIBLE SUPPLY RANGE:
+5V to +12V Single Supply
±2.5V to ±6V Dual Supplies
● INTERNALLY FIXED GAIN: +2 or ±1
● LOW SUPPLY CURRENT: 13mA
● LOW DISABLED CURRENT: 120µA
● HIGH OUTPUT CURRENT: ±120mA
● OUTPUT VOLTAGE SWING: ±4.1V
● SOT23-6 AVAILABLE
The OPA693 provides an easy to use, broadband, fixed gain
buffer amplifier. Depending on the external connections, the
internal resistor network may be used to provide either a
fixed gain of +2 video buffer or a gain of ±1 voltage buffer.
Operating on a low 13mA supply current, the OPA693 offers
a slew rate (2500V/µs) and bandwidth (> 700MHz) normally
associated with a much higher supply current. A new output
stage architecture delivers high output current with a minimal
headroom and crossover distortion. This gives exceptional
single-supply operation. Using a single +5V supply, the
OPA693 can deliver a 2.5VPP swing with over 90mA drive
current and 500MHz bandwidth at a gain of +2. This combination of features makes the OPA693 an ideal RGB line
driver or single-supply undersampling Analog-to-Digital Converter (ADC) input driver.
APPLICATIONS
●
●
●
●
●
●
The OPA693’s low 13mA supply current is precisely trimmed
at 25°C. This trim, along with low drift over temperature,
ensures lower maximum supply current than competing
products that report only a room temperature nominal supply
current. System power may be further reduced by using the
optional disable control pin. Leaving this disable pin open, or
holding it HIGH, gives normal operation. This optional disable allows the OPA693 to fit into existing video buffer
layouts where the disable pin is unconnected to get improved
performance with no board changes. If pulled LOW, the
OPA693 supply current drops to less than 170µA while the
output goes into a high impedance state. This feature may be
used for power savings.
BROADBAND VIDEO LINE DRIVERS
MULTIPLE LINE VIDEO DA
PORTABLE INSTRUMENTS
ADC BUFFERS
HIGH FREQUENCY ACTIVE FILTERS
HFA1112 IMPROVED DROP-IN
OPA693 RELATED PRODUCTS
SINGLES
DUALS
TRIPLES
OPA690
OPA2690
OPA3690
Current Feedback
OPA691
OPA2691
OPA3691
Fixed Gain
OPA692
—
OPA3692
> 900MHz
OPA695
—
—
Voltage Feedback
The low gain stable current-feedback architecture used in the
OPA693 is particularly suitable for high full-power bandwidth
cable driving requirements. Where the additional flexibility of
an op amp is required, consider the OPA695 ultra-wideband
current feedback op amp. Where a unity gain stable voltage
feedback op amp with very high slew rate is required,
consider the OPA690.
OPA693
1
300Ω
8
DIS
75Ω
300Ω
Video
In
2
7
3
6
4
5
Video
Out
+5V
RG-59
75Ω
75Ω
−5V
75Ω
Video
Out
RG-59
75Ω
SO-8
G = +2
700MHz, 2-Output Component Video DA
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
Copyright © 2002-2003, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
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ELECTROSTATIC
DISCHARGE SENSITIVITY
ABSOLUTE MAXIMUM RATINGS(1)
Power Supply ............................................................................... ±6.5VDC
Internal Power Dissipation(2) ............................ See Thermal Information
Differential Input Voltage .................................................................. ±1.2V
Input Voltage Range ............................................................................ ±VS
Storage Temperature Range: D, DVB ........................... –40°C to +125°C
Lead Temperature (soldering, 10s) .............................................. +300°C
Junction Temperature (TJ ) ........................................................... +150°C
ESD Rating (Human Body Model) .................................................. 2000V
(Charge Device Model) ............................................... 1000V
(Machine Model) ............................................................ 100V
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling
and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits
may be more susceptible to damage because very small
parametric changes could cause the device not to meet its
published specifications.
NOTES:: (1) Stresses above these ratings may cause permanent damage.
Exposure to absolute maximum conditions for extended periods may degrade
device reliability. These are stress ratings only, and functional operation of the
device at these or any other conditions beyond those specified is not implied.
(2) Packages must be derated based on specified θJA. Maximum TJ must be
observed.
PACKAGE/ORDERING INFORMATION
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
OPA693ID
OPA693IDR
OPA693IDBVT
OPA693IDBVR
Rails, 100
Tape and Reel, 2500
Tape and Reel, 250
Tape and Reel, 3000
PRODUCT
PACKAGE-LEAD
PACKAGE
DESIGNATOR(1)
OPA693
SO-8
D
–40°C to +85°C
OPA693D
"
"
"
"
"
OPA693
SOT23-6
DBV
–40°C to +85°C
C59
"
"
"
"
"
NOTE: (1) For the most current specifications and package information, refer to our web site at www.ti.com.
PIN CONFIGURATION
Top View
SO
Top View
SOT23
Output
1
6
+VS
5
DIS
4
−IN
RF
300Ω
NC
1
RG
−IN
2
+IN
3
RF
300Ω
8
2
+IN
3
DIS
7
+VS
6
Output
300Ω
−VS
RG
300Ω
6
−VS
4
5
5
4
NC
C59
NC = No Connection
1
2
3
Pin Orientation/Package Marking
2
OPA693
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SBOS285
ELECTRICAL CHARACTERISTICS: VS = ±5V
Boldface limits are tested at 25°C.
G = +2 (–IN grounded) and RL = 100Ω (see Figure 1 for AC performance only), unless otherwise noted.
OPA693ID, IDBV
MIN/MAX OVER TEMPERATURE(1)
TYP
PARAMETER
+25°C
0°C to
70°C
–40°C to
+85°C
MHz
typ
700
510
490
480
MHz
min
B
G = –1
700
510
490
480
MHz
typ
C
B
CONDITIONS
+25°C
G = +1
1400
G = +2
UNITS
MIN/ TEST
MAX LEVEL(2 )
AC PERFORMANCE (see Figure 1)
Small-Signal Bandwidth (VO < 1.0VPP)
Bandwidth for 0.2dB Gain Flatness
C
G = +2, VO < 1.0VPP, RL = 150Ω
400
122
112
108
MHz
min
Peaking at a Gain of +1
VO < 1.0VPP
2.5
3.8
4.8
5.2
dB
max
B
Large-Signal Bandwidth
G = +2, VO = 4VPP
400
MHz
typ
C
Slew Rate
Rise-and-Fall Time
Settling Time to 0.02%
Settling Time to 0.1%
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
G = +2, 4V Step
2500
2200
2100
2000
V/µs
min
B
G = +2, VO = 0.5V Step
0.6
0.8
0.8
0.8
ns
max
B
G = +2, VO = 5V Step
1.2
1.3
1.3
1.4
ns
max
B
G = +2, VO = 2V Step
16
ns
typ
C
G = +2, VO = 2V Step
12
ns
typ
C
G = +2, f = 10MHz, VO = 2VPP
RL = 100Ω
–69
–66
–65
–64
dBc
max
B
RL ≥ 500Ω
–82
–80
–79
–78
dBc
max
B
RL = 100Ω
–83
–80
–70
–69
dBc
max
B
RL ≥ 500Ω
–96
–86
–85
–82
dBc
max
B
Input Voltage Noise
f > 1MHz
1.8
2
2.7
2.9
nV/√Hz
max
B
Noninverting Input Current Noise
f > 1MHz
18
19
21
22
pA/√Hz
max
B
Inverting Input Current Noise (internal)
f > 1MHz
22
24
26
27
pA/√Hz
max
B
NTSC, RL = 150Ω
0.03
%
typ
C
NTSC, RL = 37.5Ω
0.03
%
typ
C
NTSC, RL = 150Ω
0.01
deg
typ
C
NTSC, RL = 37.5Ω
0.1
deg
typ
C
G = +1
±0.2
%
typ
C
G = +2
±0.3
±0.9
±1.0
±1.1
%
max
A
G = –1, RS = 0Ω
±0.2
±0.8
±0.9
±1.0
%
max
B
VO = ±2, RL = 100Ω, G = +2
0.0016
%
typ
C
Differential Gain
Differential Phase
DC PERFORMANCE(3)
Gain Error
DC Linearity
Internal RF and RG
Maximum
300
341
345
346
Ω
max
A
Minimum
300
264
260
259
Ω
min
A
0.03
0.03
%/C°
max
B
±2.3
±2.5
mV
max
A
±5
±8
µV/°C
max
B
Average Drift
Input Offset Voltage
VCM = 0V
Average Offset Voltage Drift
VCM = 0V
Noninverting Input Bias Current
VCM = 0V
Average Noninverting Input Bias Current Drift
VCM = 0V
Inverting Input Bias Current (internal)
VCM = 0V
Average Inverting Input Bias Current Drift
VCM = 0V
±0.3
+15
±20
±2.0
±35
±50
±43
±45
µA
max
A
170
170
nA/°C
max
B
±52
±54
µA
max
A
50
60
nA°C
max
B
±3.2
±3.2
INPUT
±3.4
Common-Mode Input Range
Noninverting Input Impedance
±3.3
300 || 1.2
V
min
B
kΩ || pF
typ
C
OUTPUT
No Load
±4.1
±3.8
V
min
A
±3.8
±3.9
±3.7
±3.9
100Ω Load
±3.7
±3.6
V
min
A
Current Output, Sourcing
+120
+90
+80
+70
mA
min
A
Current Output, Sinking
–120
–90
–80
–70
mA
min
A
Ω
typ
C
Voltage Output Swing
Closed-Loop Output Impedance
G = +2, f = 100kHz
0.18
(1) Junction temperature = ambient temperature for low temperature limit and +25°C specifications. Junction temperature = ambient temperature +20°C at high
temperature limit specifications.
(2) Test Levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation.
(B) Limits set by characterization and simulation.
(C) Typical value only for information.
(3) Current is considered positive out-of-node. VCM is the input common-mode voltage.
OPA693
SBOS285
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3
ELECTRICAL CHARACTERISTICS: VS = ±5V (Cont.)
Boldface limits are tested at 25°C.
G = +2 (–IN grounded) and RL = 100Ω (see Figure 1 for AC performance only), unless otherwise noted.
OPA693ID, IDBV
MIN/MAX OVER TEMPERATURE(1)
TYP
PARAMETER
CONDITIONS
+25°C
+25°C
0°C to
70°C
–40°C to
+85°C
UNITS
–170
–186
–192
µA
max
A
µs
typ
C
MIN/ TEST
MAX LEVEL(2 )
DISABLE/POWER DOWN (DIS Pin)
Power-Down Supply Current (+VS)
VDIS = 0V
–70
VIN = +1VDC
3
Enable Time
VIN = +1VDC
25
ns
typ
C
Off Isolation
G = +2, 5MHz
70
dB
typ
C
4
pF
typ
C
Turn-On Glitch
G = +2, RL = 150Ω, VIN = 0VDC
±100
mV
typ
C
Turn-Off Glitch
G = +2, RL = 150Ω, VIN = 0VDC
±20
mV
typ
C
Enable Voltage
+VS = +5V
3.3
3.5
3.6
3.7
V
min
A
Disable Voltage
+VS = +5V
1.8
1.7
1.6
1.5
V
max
A
Control Pin Input Bias Current
VDIS = 0V
75
130
143
149
µA
max
A
V
typ
C
±6
±6
±6
V
max
A
Disable Time
Output Capacitance in Disable
POWER SUPPLY
±5
Specified Operating Voltage
Maximum Operating Voltage Range
Max Quiescent Current
VS = ±5V
13
13.3
13.7
14.1
mA
max
A
Min Quiescent Current
VS = ±5V
13
12.5
11.6
11.0
mA
min
A
Input Referred
58
54
52
51
dB
min
A
–40 to +85
°C
typ
C
125
°C/W
typ
C
150
°C/W
typ
C
Power-Supply Rejection Ratio (+PSRR)
TEMPERATURE RANGE
Specification: D, DBV
Thermal Resistance, θJA
D
SO-8
DBV SOT23-6
(1) Junction temperature = ambient temperature for low temperature limit and +25°C specifications. Junction temperature = ambient temperature +20°C at high
temperature limit specifications.
(2) Test Levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation.
(B) Limits set by characterization and simulation.
(C) Typical value only for information.
(3) Current is considered positive out-of-node. VCM is the input common-mode voltage.
4
OPA693
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SBOS285
ELECTRICAL CHARACTERISTICS: VS = +5V
Boldface limits are tested at +25°C.
G = +2 (–IN grounded though 0.001µF) and RL = 100Ω to VS/2 (see Figure 4 for AC performance only), unless otherwise noted.
OPA693ID, IDBV
TYP
PARAMETER
CONDITIONS
+25°C
MIN/MAX OVER TEMPERATURE
+25°C(1)
0°C to
70°C
–40°C to
+85°C
UNITS
MHz
typ
400
390
380
MHz
min
B
MHz
typ
C
B
MIN/ TEST
MAX LEVEL(2 )
AC PERFORMANCE (see Figure 4)
Small-Signal Bandwidth (VO < 1.0VPP)
Bandwidth for 0.2dB Gain Flatness
G = +1
634
G = +2
526
G = –1
512
C
G = +2, VO < 1.0VPP
210
110
100
96
MHz
min
Peaking at a Gain of +1
VO < 1.0VPP
1.9
2.6
3.6
3.9
dB
max
B
Large-Signal Bandwidth
G = +2, VO = 2VPP
400
MHz
typ
C
Slew Rate
Rise-and-Fall Time
G = +2, 2V Step
1500
V/µs
min
B
G = +2, VO = 0.5V Step
0.8
1200
1100
1000
ns
typ
C
G = +2, VO = 2V Step
1.0
ns
typ
C
Settling Time to 0.02%
G = +2, VO = 2V Step
16
ns
typ
C
Settling Time to 0.1%
G = +2, VO = 2V Step
12
ns
typ
C
Harmonic Distortion
2nd-Harmonic
G = +2, f = 10MHz, VO = 2VPP
RL = 100Ω to VS /2
–66
–62
–62
–61
dBc
max
B
RL ≥ 500Ω to VS /2
–75
–69
–68
–68
dBc
max
B
RL = 100Ω to VS /2
–70
–64
–63
–62
dBc
max
B
RL ≥ 500Ω to VS /2
–69
–63
–62
–61
dBc
max
B
Input Voltage Noise
f > 1MHz
1.8
2
2.7
2.9
nV/√Hz
max
B
Noninverting Input Current Noise
f > 1MHz
18
19
21
22
pA/√Hz
max
B
Inverting Input Current Noise
f > 1MHz
22
24
26
27
pA/√Hz
max
B
G = +1
±0.2
%
typ
C
G = +2
±0.5
±1.2
±1.3
±1.4
%
max
A
G = –1
±0.4
±1.1
±1.2
±1.3
%
max
B
Maximum
300
341
345
346
Ω
max
B
Minimum
300
264
260
259
Ω
min
B
0.03
0.03
0.03
%/C°
max
B
±0.3
±2.5
±2.8
±3.0
mV
max
A
±5
±8
µV/°C
max
B
+5
±25
±33
±35
µA
max
A
±170
±170
nA/°C
max
B
±52
±54
µA
max
A
±50
±60
nA°C
max
B
3rd-Harmonic
DC PERFORMANCE(3)
Gain Error
Internal RF and RG
Average Drift
Input Offset Voltage
VCM = 2.5V
Average Offset Voltage Drift
VCM = 2.5V
Noninverting Input Bias Current
VCM = 2.5V
Average Noninverting Input Bias Current Drift
VCM = 2.5V
Inverting Input Bias Current
VCM = 2.5V
Average Inverting Input Bias Current Drift
VCM = 2.5V
±20
±50
INPUT
Least Positive Input Voltage
1.6
1.7
1.8
1.8
V
max
B
Most Positive Input Voltage
3.4
3.3
3.2
3.2
V
min
B
kΩ || pF
typ
C
Noninverting Input Impedance
300 || 1.2
OUTPUT
Most Positive Output Voltage
Least Positive Output Voltage
No Load
4.1
3.9
3.9
3.8
V
min
A
RL = 100Ω
3.9
3.8
3.8
3.7
V
min
A
No Load
0.9
1.1
1.1
1.2
V
max
A
RL = 100Ω
1.1
1.2
1.2
1.3
V
max
A
Current Output, Sourcing
+120
+90
+80
+70
mA
min
A
Current Output, Sinking
–120
–90
–80
–70
mA
min
A
Ω
typ
C
Output Impedance
G = +2, f = 100kHz
0.18
(1) Junction temperature = ambient temperature for low temperature limit and +25°C specifications. Junction temperature = ambient temperature +10°C at high
temperature limit specifications.
(2) Test Levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation.
(B) Limits set by characterization and simulation.
(C) Typical value only for information.
(3) Current is considered positive out-of-node. VCM is the input common-mode voltage.
OPA693
SBOS285
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5
ELECTRICAL CHARACTERISTICS: VS = +5V (Cont.)
Boldface limits are tested at +25°C.
G = +2 (–IN grounded though 0.001µF) and RL = 100Ω to VS/2 (see Figure 4 for AC performance only), unless otherwise noted.
OPA693ID, IDBV
TYP
PARAMETER
MIN/MAX OVER TEMPERATURE
CONDITIONS
+25°C(1)
+25°C
0°C to
70°C
–40°C to
+85°C
–155
–172
–180
UNITS
MIN/ TEST
MAX LEVEL(2 )
DISABLE/POWER DOWN (DIS Pin)
Power-Down Supply Current (+VS)
µA
typ
A
dB
typ
C
pF
typ
C
mV
typ
B
VDIS = 0
–95
G = +2, 5MHz
65
4
Turn-On Glitch
G = +2, RL = 150Ω, VIN = 2.5V
±20
Turn-Off Glitch
G = +2, RL = 150Ω, VIN = 2.5V
±20
mV
typ
B
V
min
B
Off Isolation
Output Capacitance in Disable
Enable Voltage
3.3
Disable Voltage
Control Pin Input Bias Current (DIS )
VDIS = 0
3.5
3.6
3.7
1.8
1.7
1.6
1.5
V
max
B
80
137
153
160
µA
typ
A
V
typ
C
+12
+12
+12
V
max
A
POWER SUPPLY
Specified Single-Supply Operating Voltage
5
Maximum Single-Supply Operating Voltage
Maximum Quiescent Current
VS = +5V
11.5
12.0
12.5
12.9
mA
max
A
Minimum Quiescent Current
VS = +5V
11.5
11.0
9.5
9.2
mA
min
A
Input Referred
57
dB
typ
C
–40 to +85
°C
typ
C
125
°C/W
typ
C
150
°C/W
typ
C
Power-Supply Rejection Ratio (+PSRR)
TEMPERATURE RANGE
Specification: D, DBV
Thermal Resistance, θJA
D
SO-8
DBV SOT23-6
(1) Junction temperature = ambient temperature for low temperature limit and +25°C specifications. Junction temperature = ambient temperature +10°C at high
temperature limit specifications.
(2) Test Levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation.
(B) Limits set by characterization and simulation.
(C) Typical value only for information.
(3) Current is considered positive out-of-node. VCM is the input common-mode voltage.
6
OPA693
www.ti.com
SBOS285
TYPICAL CHARACTERISTICS: VS = ±5V
At TA = +25°C, G = +2, and RL = 100Ω, unless otherwise specified.
LARGE-SIGNAL FREQUENCY RESPONSE
SMALL-SIGNAL FREQUENCY RESPONSE
8
3
VO = 1VPP
G = +1
G = +2
RL = 100Ω
7
6
1
G = +2
0
−1
G = −1
−2
4
2
−4
1
−5
0
−6
−1
100
1000
VO = 2VPP
3
−3
10
VO = 1VPP
5
Gain (dB)
Normalized Gain (dB)
2
VO = 7VPP
VO = 4VPP
See Figure 1
0
2000
200
Frequency (MHz)
FREQUENCY RESPONSE FLATNESS vs LOAD
Normalized Gain (dB)
0.1
RL = 200Ω
0
−0.1
RL = 75Ω
RL = 100Ω
−0.3
See Figure 1
−0.4
RL = 100Ω
0.75
G = +1
0.50
G = −1
0.25
0
−0.25
G = +2
−0.50
−0.75
−1.00
0
100
200
Frequency (MHz)
300
400
0
50
GAIN OF +2 PULSE RESPONSE
150
200
GAIN OF +1 PULSE RESPONSE
3
RL = 100Ω
RL = 100Ω
Large Signal
2
Large Signal
2
1
1
Output (V)
Small Signal
0
−1
Small Signal
0
−1
−2
−2
See Figure 1
See Figure 2
−3
−3
Time (2ns/div)
Time (2ns/div)
OPA693
SBOS285
100
Frequency (MHz)
3
Output (V)
1000
DEVIATION FROM LINEAR PHASE
RL = 150Ω
−0.2
800
1.00
G = +2
VO = 1VPP
Deviation from Linear Phase (°)
0.2
400
600
Frequency (MHz)
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7
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
At TA = +25°C, G = +2, and RL = 100Ω, unless otherwise specified.
10MHz HARMONIC DISTORTION
vs LOAD RESISTANCE
−60
−60
Harmonic Distortion (dBc)
−65
−70
2nd Harmonic
−75
−80
−85
3rd Harmonic
G = +2
f = 10MHz
VO = 2VPP
See Figure 1
−90
−95
−100
50
3rd Harmonic
−80
−85
G = +2
RL = 100Ω
VO = 2VPP
See Figure 1
−90
500
2.5
3.0
3.5
10MHz HARMONIC DISTORTION
vs OUTPUT VOLTAGE
−60
G = +2
RL = 100Ω
f = 10MHz
See Figure 1
−65
2nd Harmonic
−80
3rd Harmonic
−85
−90
−70
See Figure 1
−100
0.5
6.0
2nd Harmonic
−75
−80
3rd Harmonic
−85
−90
1
10
−100
0.5
50
1
G = –1 HARMONIC DISTORTION vs FREQUENCY
G = +1 HARMONIC DISTORTION vs FREQUENCY
RL = 100Ω
VO = 2VPP
See Figure 2
−50
2nd Harmonic
3rd Harmonic
−75
RL = 100Ω
VO = 2VPP
See Figure 3
−55
Harmonic Distortion (dBc)
−60
5
Output Voltage (VPP)
Frequency (MHz)
Harmonic Distortion (dBc)
5.5
−95
−95
−80
−85
−90
−95
−60
2nd Harmonic
−65
−70
3rd Harmonic
−75
−80
−85
−90
−95
−100
−100
0.5
1
10
0.5
50
Frequency (MHz)
8
5.0
G = +2 HARMONIC DISTORTION vs FREQUENCY
−75
−70
4.5
Supply Voltage (±V)
−70
−65
4.0
Load Resistance (Ω)
RL = 100Ω
VO = 2VPP
−65
−75
−100
100
−60
2nd Harmonic
−70
−95
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
−65
Harmonic Distortion (dBc)
10MHz HARMONIC DISTORTION
vs SUPPLY VOLTAGE
1
10
50
Frequency (MHz)
OPA693
www.ti.com
SBOS285
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
At TA = +25°C, G = +2, and RL = 100Ω, unless otherwise specified.
2-TONE, 3RD-ORDER
INTERMODULATION INTERCEPT
INPUT VOLTAGE vs CURRENT NOISE DENSITY
60
100
+5V
Inverting Current Noise (internal)
22pA/√Hz
Noninverting Current Noise
17.8pA/√Hz
10
Voltage Noise
50Ω
Intercept Point (+dBm)
Voltage Noise (nV/√Hz)
Current Noise (pA/√Hz)
PI
PO
OPA693
RF
300Ω
50
500Ω
RG
300Ω
−5V
40
+5V
PI
30
50Ω
PO
OPA693
50Ω
RF
300Ω
1.8nV/√Hz
50Ω
RG
300Ω
−5V
20
1
100
1k
10k
100k
1M
0
10M
25
50
Frequency (MHz)
−10
−10
−20
VSWR < 1.2:1
G = −1
See Figure 3
−40
−50
G = +2
See Figure 1
125
150
175
200
No Output
Trim Capacitor
−20
VSWR < 1.2:1
−30
−40
50Ω
OPA693
−50
G = +2
See Figure 1
−60
−70
With
Trim Capacitor
−60
10
60
100
1000
10
SMALL-SIGNAL FREQUENCY RESPONSE
vs CAPACITIVE LOAD
9
Gain to Cap. Load (dB)
RS
VO
OPA693
50Ω
300Ω
CL
1kΩ
3
CL = 100pF
CL = 10pF
0
CL = 50pF
CL = 20pF
−3
−6
300Ω
1kΩ is optional
−9
0
10
100
Capacitive Load (pF)
OPA693
SBOS285
G = +2
Optimized RS
6
VIN
1
1000
RECOMMENDED RS vs CAPACITIVE LOAD
30
10
100
Frequency (MHz)
40
20
S22
1.8pF
Frequency (MHz)
G = +2
< 0.1dB Peaking
50
RS (Ω)
100
OUTPUT RETURN LOSS vs FREQUENCY (S22)
0
Return Loss (dB)
Return Loss (dB)
INPUT RETURN LOSS vs FREQUENCY (S11)
0
−30
75
Frequency (MHz)
www.ti.com
10
100
Frequency (MHz)
1000
9
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
At TA = +25°C, G = +2, and RL = 100Ω, unless otherwise specified.
CLOSED-LOOP OUTPUT IMPEDANCE
PSRR vs FREQUENCY
10
–PSRR
+5V
60
+PSRR
55
Output Impedance (Ω)
50
45
40
35
30
OPA693
50Ω
–5V 300Ω
1
300Ω
25
0.1
20
1k
10k
100k
1M
10M
10k
100M
100k
1M
OUTPUT VOLTAGE AND CURRENT LIMITATIONS
150
1W Internal
Power
Boundary
2
50Ω Load Line
1
20Ω Load Line
0
−1
−2
−3
1W Internal
Power
Boundary
−4
−5
−250 −200 −150 −100 −50
0
50
IO (mA)
130
120
14
Supply Current
Right Scale
13
Sourcing Output Current
Left Scale
12
Sinking Output Current
110
11
100
100
150
10
–50
200 250
–25
0
25
50
75
100
125
Ambient Temperature (°C)
NONINVERTING OVERDRIVE RECOVERY
INVERTING OVERDRIVE RECOVERY
6
4
15
140
100Ω Load Line
Output Current (mA)
3
100M
SUPPLY AND OUTPUT CURRENT vs TEMPERATURE
5
4
10M
Frequency (Hz)
Frequency (Hz)
VO (V)
ZO
6
G = +2
RL = 100Ω
4
G = −1
RL = 100Ω
Input/Output (V)
Input/Output (V)
Output
2
Input
0
−2
−4
2
Output
Input
0
−2
−4
See Figure 1
See Figure 3
−6
−6
Time (50ns/div)
10
Time (50ns/div)
OPA693
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SBOS285
Supply Current (mA)
Power-Supply Rejection Ratio (dB)
65
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
At TA = +25°C, G = +2, and RL = 100Ω, unless otherwise specified.
SETTLING TIME
DISABLED FEEDTHRU vs FREQUENCY
−20
20
G = +2
RL = 100Ω
2V → 0V
Output Step
10
−30
−40
5
Gain (dB)
Input/Output (5mV/div)
15
Input
0
−5
G = +2
RL = 100Ω
VDIS = 0V
Output
Forward and Reverse
−50
−60
−70
−80
−10
−15
−90
See Figure 1
−100
−20
0
2
4
6
8 10 12
Time (2ns/div)
14
16
18
20
See Figure 1
10
100
Frequency (MHz)
COMMON-MODE INPUT AND OUTPUT SWING
vs SUPPLY VOLTAGE
TYPICAL DC DRIFT OVER TEMPERATURE
1.0
6
16
8
0
0
VIO
−8
IB− (internal)
Input/Output Range (±V)
0.5
Input Bias Currents (µA)
Input Offset Voltage (mV)
IB+
−0.5
−16
–25
5
4
Output
3
Input
2
1
−1.0
–50
0
25
50
75
100
0
125
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
6.5
Supply Voltages (±V)
Ambient Temperature (°C)
LARGE-SIGNAL DISABLE/ENABLE RESPONSE
COMPOSITE VIDEO dG/dP
7
0.12
No Pull-Down
With 1.0kΩ Pull-Down
+5V
Video In
DIS
5
OPA693
75Ω
0.08
dP
Optional
1.0kΩ
Pull-Down
−5V
0.06
6
Video Loads
4
VDIS/VOUT (V)
0.10
dG/dP (%/°)
1000
dP
0.04
2
1
VOUT
0
dG
G = +2
VIN = 1VDC
RL = 100Ω
−1
dG
0.02
VDIS
3
−2
See Figure 1
−3
0
1
2
3
Time (500ns/div)
4
Number of 150Ω Loads
OPA693
SBOS285
www.ti.com
11
TYPICAL CHARACTERISTICS: VS = +5V
At TA = +25°C, G = +2, and RL = 100Ω to VS/2, unless otherwise specified.
SMALL-SIGNAL FREQUENCY RESPONSE
3
G = +1
1
6
G = +2
−1
5
Gain (dB)
0
G = −1
−2
2
1
−5
0
−6
−1
10
100
1000
VO = 2VPP
See Figure 4
0
200
SMALL-SIGNAL BANDWIDTH
vs SINGLE-SUPPLY VOLTAGE
800
RL = 150Ω
−0.1
−0.2
RL = 75Ω
700
650
600
550
500
450
RL = 100Ω
See Figure 4
See Figure 4
−0.4
400
0
50
100
150
4
200
5
6
7
8
9
10
11
12
Single-Supply Voltage (V)
Frequency (MHz)
GAIN OF +2 PULSE RESPONSE
GAIN OF +1 PULSE RESPONSE
4.5
4.5
RL = 100Ω
4.0
RL = 100Ω
Large Signal
Large Signal
3.5
3.5
Small Signal
3.0
Small Signal
Output (V)
Output (V)
1000
G = +2
VO = 0.5VPP
RL = 100Ω
750
−0.3
2.5
2.0
1.5
3.0
2.5
2.0
1.5
1.0
See Figure 4
0.5
See Figure 5
0.5
Time (2ns/div)
12
800
FREQUENCY RESPONSE FLATNESS vs LOAD
RL = 200Ω
1.0
600
Frequency (MHz)
0
4.0
400
Frequency (MHz)
G = +2
VO = 1VPP
0.1
VO = 3VPP
3
−4
1
VO = 1VPP
4
−3
0.2
G = +2
RL = 100Ω
7
Small-Signal BW (MHz)
Normalized Gain (dB)
2
Normalized Gain (dB)
LARGE-SIGNAL FREQUENCY RESPONSE
8
VO = 1VPP
Time (2ns/div)
OPA693
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SBOS285
TYPICAL CHARACTERISTICS: VS = +5V (Cont.)
At TA = +25°C, G = +2, and RL = 100Ω to VS/2, unless otherwise specified.
HARMONIC DISTORTION vs FREQUENCY
G = +2
RL = 100Ω
VO = 2VPP
Harmonic Distortion (dBc)
−60
−65
2nd Harmonic
−75
−80
−85
−65
−70
2nd Harmonic
−75
−80
−85
3rd Harmonic
−90
−90
See Figure 4
−95
0.5
See Figure 4
−95
1
10
−55
0.1
50
1
3
Frequency (MHz)
Output Voltage (VPP)
HARMONIC DISTORTION
vs LOAD RESISTANCE
2-TONE, 3RD-ORDER
INTERMODULATION INTERCEPT
50
2nd Harmonic
−65
G = +2
f = 10MHz
1kΩ
PI
−70
3rd Harmonic
−75
+5V
45
Intercept Point (+dBm)
−60
Harmonic Distortion (dBc)
G = +2
RL = 100Ω
f = 10MHz
−60
3rd Harmonic
−70
HARMONIC DISTORTION vs OUTPUT VOLTAGE
−55
Harmonic Distortion (dBc)
−55
−80
−85
−90
40
50Ω
35
1kΩ
PO
OPA693
RF
300Ω
500Ω
RG
300Ω
30
+5V
1kΩ
25
PI
50Ω
20
50Ω
1kΩ
PO
OPA693
RF
300Ω
50Ω
RG
300Ω
15
See Figure 4
−95
10
50
100
500
25
50
75
100
125
150
175
200
Frequency (MHz)
Load Resistance (Ω)
OPA693
SBOS285
0
www.ti.com
13
APPLICATION INFORMATION
WIDEBAND BUFFER OPERATION
The OPA693 gives the exceptional AC performance of a
wideband current-feedback op amp with a highly linear
output stage. It features internal RF and RG resistors, making
it a simple matter to select a gain of +2, +1 or –1 with no
external resistors. Requiring only 13mA supply current, the
OPA693’s output swings to within 1V of either supply with
> 700MHz small signal bandwidth and > 300MHz delivering
7VPP into a 100Ω load. This low output headroom in a very
high-speed amplifier gives remarkable single +5V operation.
The OPA693 delivers 2VPP swing with > 500MHz bandwidth
operating on a single +5V supply. The primary advantage of
a current-feedback fixed gain video buffer, as opposed to a
slew-enhanced low-gain stable voltage-feedback implementation, is a higher slew rate with lower quiescent power and
output noise.
Figure 1 shows the DC-coupled, gain of +2V/V, dual powersupply circuit configuration used as the basis for the ±5V
Electrical Characteristics table and Typical Characteristics
curves. For test purposes, the input impedance is set to 50Ω
with a resistor to ground and the output impedance is set to
50Ω with a series output resistor. Voltage swings reported in
the specifications are taken directly at the input and output
pins while load powers (dBm) are defined at a matched 50Ω
load. For the circuit of Figure 1, the total effective load will be
100Ω || 600Ω = 85.7Ω. The disable control line (DIS) is
typically left open to ensure normal amplifier operation. In
addition to the usual power supply decoupling capacitors to
ground, a 0.01µF capacitor can be included between the two
power-supply pins. This optional added capacitor will typically improve the 2nd harmonic distortion performance by
3dB to 6dB.
+5V
+
0.1µF
6.8µF
50Ω Source
DIS
VI
50Ω
50Ω
OPA693
VO
50Ω Load
RF
300Ω
RG
300Ω
0.1µF
+
6.8µF
−5V
Figure 1. DC-Coupled, G = +2, Bipolar-Supply, Specification
and Test Circuit.
14
Figure 2 shows the DC-coupled, gain of +1V/V buffer configuration used as a starting point for the gain of +1V/V
Typical Characteristic curves. In this case, the inverting input
resistor, RG, is left open giving a very broadband gain of +1
performance. While the test circuit shows a 50Ω input resistor, a buffer application is typically transforming from a
source that cannot drive a heavy load to a 100Ω load, such
as shown in Figure 2. The noninverting input impedance of
the OPA693 is typically 100kΩ || 2pF. Driving directly into
the noninverting input will provide this very light load to the
source. However, the source must still provide the noninverting
input bias current required by the input stage to operate. An
alternative approach to a gain of +1 buffer is described in the
Wideband Unity Gain Buffers section of this data sheet.
+5V
+
0.1µF
6.8µF
50Ω Source
DIS
VI
50Ω
50Ω
OPA693
VO
50Ω Load
RF
300Ω
RG
300Ω
0.1µF
+
6.8µF
Open
−5V
Figure 2. DC-Coupled, G = +1V/V, Bipolar-Supply, Specification and Test Circuit.
Figure 3 shows the DC-coupled, gain of –1V/V buffer configuration used as a starting point for the gain of –1V/V
Typical Characteristic curves. The input impedance is set to
50Ω using the parallel combination of an external 60.4Ω
resistor and the internal 300Ω RG resistor. The noninverting
input is tied directly to ground. Since the internal design for
the OPA693 is current-feedback, trying to get improved DC
accuracy by including a resistor on the noninverting input to
ground is ineffective. Using a direct short to ground on the
noninverting input reduces both the contribution of the DC
bias current and noise current to the output error. While the
external 60.4Ω is used here to match to the 50Ω source from
the test equipment, the maximum input impedance in this
configuration is limited to the 300Ω RG resistor even with the
RM resistor removed. Unlike the noninverting unity gain
buffer application, removing RM does not strongly impact the
DC operating point because the short on the noninverting
input of Figure 3 provides the DC operating voltage. This
application of the OPA693 provides a very broadband, highoutput, signal inverter.
OPA693
www.ti.com
SBOS285
+5V
+VS
0.1µF
+
6.8µF
50Ω Source
DIS
VO
OPA693
+5V
RG
300Ω
6.8µF
604Ω
1000pF
DIS
VI
50Ω
60.4Ω
VO
604Ω
100Ω
OPA693
50Ω Load
50Ω Source
+
0.1µF
VS/2
RF
300Ω
RF
300Ω
VI
RG
300Ω
RM
60.4Ω
0.1µF
−5V
+
6.8µF
1000pF
Figure 3. DC-Coupled, G = –1V/V, Bipolar-Supply Specification and Test Circuit.
Figure 4. AC-Coupled, G = +2V/V, Single-Supply Specification and Test Circuit.
SINGLE-SUPPLY OPERATION
While the circuit of Figure 4 shows +5V single-supply operation, this same circuit may be used for single supplies
ranging as high as +12V nominal. The noninverting input bias
resistors are relatively low in Figure 4 to minimize output DC
offset due to noninverting input bias current. At higher signalsupply voltage, these should be increased to limit the added
supply current drawn through this path.
The OPA693 may be used over a single-supply range of +5V
to +12V. Though not a rail-to-rail output design, the OPA693
requires minimal input and output voltage headroom compared to other very-wideband video buffer amplifiers. As
shown in the single +5V Typical Characteristic curves, the
OPA693 provides > 300MHz bandwidth driving a 3VPP swing
into a 100Ω load. The key requirement of broadband singlesupply operation is to maintain input and output signal
swings within the useable voltage ranges at both the input
and the output.
The circuit of Figure 4 shows the AC-coupled, gain of
+2V/V, video buffer circuit used as the basis for the Electrical
Characteristics table and Typical Characteristics curves. The
circuit of Figure 4 establishes an input midpoint bias using a
simple resistive divider from the +5V supply (two 604Ω
resistors). The input signal is then AC-coupled into this
midpoint voltage bias. The input voltage can swing to within
1.7V of either supply pin, giving a 1.6VPP input signal range
centered between the supply pins. The input impedance
matching resistor (60.4Ω) used for testing is adjusted to give
a 50Ω input match when the parallel combination of the
biasing divider network is included. The gain resistor (RG) is
AC-coupled, giving the circuit a DC gain of +1, which puts the
input DC bias voltage (2.5V) on the output as well. Again, on
a single +5V supply, the output voltage can swing to within
1V of either supply pin while delivering more than 90mA
output current. A demanding 100Ω load to a midpoint bias is
used in this characterization circuit. The new output stage
used in the OPA693 can deliver large bipolar output current
into this midpoint load with minimal crossover distortion, as
shown by the +5V supply, 3rd-harmonic distortion plots.
Figure 5 shows the AC-coupled, G = +1V/V, single-supply
specification and test circuit. In this case, the gain setting
resistor, RG, is simply left open to get a gain of +1V for AC
signals. Once again, the noninverting input is DC biased at
mid-supply, putting that same VS/2 at the output pin. The
signal is AC-coupled into this midpoint with an added termination resistor on the source side of the blocking capacitor.
VS
+
0.1µF
50Ω Source
1000pF
6.8µF
604Ω
DIS
VI
60.4Ω
604Ω
OPA693
VO 100Ω
VS/2
RF
300Ω
RG
300Ω
Open
Figure 5. AC-Coupled, G = +1V/V, Single-Supply Specification and Test Circuit.
OPA693
SBOS285
+5V
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15
SINGLE-SUPPLY ADC INTERFACE
Most modern, high-performance ADCs (such as the Texas
Instruments ADS8xx series) operate on a single +5V (or
lower) power supply. It has been a considerable challenge
for single-supply op amps to deliver a low distortion input
signal at the ADC input for signal frequencies exceeding
5MHz. The high slew rate, exceptional output swing, and
high linearity of the OPA693 make it an ideal single-supply
ADC driver. Figure 6 shows an example input interface to a
very high-performance, 10-bit, 75MSPS CMOS converter.
The OPA693 in the circuit of Figure 6 provides > 500MHz
bandwidth at an operating gain of +2V/V delivering 1VPP at the
output for a 0.5VPP input. This broad bandwidth provides
adequate margin to deliver low distortion to the maximum
20Mhz analog input frequency intended for the circuit of Figure
6. A 40MHz low-pass filter is provided as part of the converter
interface to both limit broadband noise and reduce harmonics
as the signal frequency exceeds 15MHz. The noninverting input
bias voltage is referenced to the midpoint of the ADC signal
range by dividing off the top and bottom of the internal ADC
reference ladder.
This circuit creates an additional input offset voltage as the
difference in the two input bias currents times the impedance
to ground at VI. Figure 8 shows a comparison of small-signal
frequency response for the unity gain buffer of Figure 2
compared to the improved approach shown in Figure 7.
+5V
DIS
OPA693
RG
300Ω
VO
RO
50Ω
RF
300Ω
VI
RM
50Ω
−5V
Figure 7. Improved Unity Gain Buffer.
2
G = +1, Figure 2
WIDEBAND UNITY GAIN BUFFER WITH IMPROVED
FLATNESS
Normalized Gain (dB)
1
As shown in the Typical Characteristic curves, the unity gain
buffer configuration of Figure 2 shows a peaking in the frequency response exceeding 2dB. This gives the slight amount
of overshoot and ringing apparent in the gain of +1V/V pulse
response curves. A similar circuit that holds a flatter frequency
response, giving improved pulse fidelity, is shown in Figure 7.
This circuit removes the peaking by bootstrapping out any
parasitic effects on RG. The input impedance is still set by RM
as the apparent impedance looking into RG is very high. RM
may be increased to show a higher input impedance, but
larger values will start to impact DC output offset voltage.
0
−1
G = +1, Figure 7
−2
−3
−4
−5
−6
10
100
1000
Frequency (MHz)
Figure 8. Buffer Frequency Response Comparison.
+5V
+5V
RF
300Ω
RG
300Ω
1000pF
Clock
50Ω
Input
OPA693
1VPP
0.5VPP
100pF
ADS828
10-Bit
75MSPS
Input
1000pF
CM
DIS
2kΩ
+3.5V
REFT
0.1µF
+2.5V DC Bias
2kΩ
+1.5V
REFB
0.1µF
Figure 6. Wideband, AC-Coupled, Single-Supply ADC Driver.
16
OPA693
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SBOS285
WIDEBAND, DC-COUPLED,
SINGLE-TO-DIFFERENTIAL CONVERSION
The frequency response shown in Figure 7 for the improved
gain of +1V/V buffer closely matches the inverting gain of
–1V/V frequency response. Combining two OPA693s to give
a +1 and –1 response will give a very broadband, DCcoupled, single-ended input to differential output conversion.
Figure 9 shows this implementation where the input match is
now set by RM in parallel with the RG resistor of the inverting
stage. This circuit is essentially providing a DC to 700MHz
1:1 transformer operation. A 50Ω input match is shown, but
this may be increased by increasing RM. For instance,
targeting a 200Ω input impedance requires an RM = 600Ω to
get the parallel combination of RM and RG = 200Ω.
shows an example gain of +2 line driver using the OPA693
that incorporates a 40MHz low-pass Butterworth response
with just a few external components. The filter resistor values
have been adjusted slightly here from an ideal filter analysis
to account for parasitic effects.
+5V
22pF
100Ω
226Ω
VI
50Ω
22pF
VO
OPA693
0Ω
Source
RF
300Ω
RG
300Ω
+5V
50Ω
−5V
DIS
OPA693
RG
300Ω
+VI
Figure 10. Line Driver with 40 MHz Low-Pass Active Filter.
RF
300Ω
This type of filter depends on a low output impedance from
the amplifier through very high frequencies to continue to
provide an increasing attenuation with frequency. As the
amplifier output impedance rises with frequency, any input
signal or noise starts to feed directly through to the output via
the feedback capacitor. Since the OPA693 used in Figure 10
has a 700MHz bandwidth, the active filter will continue to roll
off through frequencies exceeding 200MHz. Figure 11 shows
the frequency response for the filter of Figure 10, where the
desired 40MHz cutoff is achieved and a 40dB/dec rolloff is
held through very high frequencies.
VI
RM
60.4Ω
−5V
RG
300Ω
+5V
2VI
RF
300Ω
−VI
OPA693
DIS
−5V
3
0
−3
Figure 9. DC → 700MHz, Single-to-Differential Conversion.
The extremely wide bandwidth of the OPA693 allows a wide
range of active filter topologies to be implemented with
minimal amplifier bandwidth interaction in the filter shape.
Sallen-Key filters, using either a gain of 1 or gain of 2
amplifier, may be easily implemented with no external gain
setting elements. In general, given a desired filter WO, the
amplifier should have at least 20X that WO to minimize filter
interaction with the amplifier frequency response. Figure 10
Gain (dB)
HIGH-FREQUENCY ACTIVE FILTERS
−6
−15
−18
−21
−24
−27
−30
1
10
100
1000
Frequency (MHz)
Figure 11. 40MHz Low-Pass Active Filter Response.
OPA693
SBOS285
−9
−12
www.ti.com
17
DESIGN-IN TOOLS
DEMONSTRATION BOARDS
Two printed circuit (PC) boards are available to assist in the
initial evaluation of the circuit performance using the OPA693 in
its two package styles. Both are available free as unpopulated
PC boards delivered with descriptive documentation. The summary information for these boards is shown in Table I.
PRODUCT
PACKAGE
DEMO BOARD
PART
NUMBER
LITERATURE
REQUEST
NUMBER
SO-8
DEM-OPA68xU
SBOU009
SOT23-6
DEM-OPA6xxN
SBOU010
OPA693ID
OPA693IDBV
TABLE I. Demo Board Ordering Information.
To request either of these boards, check the Texas Instruments web site at www.ti.com.
OPERATING SUGGESTIONS
GAIN SETTING
Setting the gain for the OPA693 is very easy. For a gain of +2,
ground the –IN pin and drive the +IN pin with the signal. For
a gain of +1, either leave the –IN pin open and drive the +IN
pin or drive both the +IN and –IN pins as shown in Figure 7.
For a gain of –1, ground the +IN pin and drive the –IN pin with
the input signal. An external resistor may be used in series
with the –IN pin to reduce the gain. However, since the internal
resistors (RF and RG) have a tolerance and temperature drift
different than the external resistor, the absolute gain accuracy
and gain drift over temperature will be relatively poor compared to the previously described standard gain connections
using no external resistor.
The minimum specified output voltage and current specifications over temperature are set by worst-case simulations at
the cold temperature extreme. Only at cold startup will the
output current and voltage decrease to the numbers shown
in the over-temperature min/max specifications. As the output transistors deliver power, their junction temperatures
increase, which decreases their VBE’s (increasing the available output voltage swing) and increases their current gains
(increasing the available output current). In steady state
operation, the available output voltage and current will always be greater than that shown in the over-temperature
characteristics since the output stage junction temperatures
will be higher than the minimum specified operating ambient.
To maintain maximum output stage linearity, no output shortcircuit protection is provided. This will not normally be a
problem, since most applications include a series matching
resistor at the output that limits the internal power dissipation
if the output side of this resistor is shorted to ground.
However, shorting the output pin directly to an adjacent
positive power supply pin will, in most cases, destroy the
amplifier. If additional protection to a power-supply short is
required, consider a small series resistor in the power supply
leads. Under heavy output loads, this will reduce the available output voltage swing. A 5Ω series resistor in each
supply lead will limit the internal power dissipation to < 1W for
an output short while decreasing the available output voltage
swing only 0.5V, for up to 100mA desired load currents.
Always place the 0.1µF power supply decoupling capacitors
after these supply current limiting resistors directly on the
device supply pins.
DRIVING CAPACITIVE LOADS
OUTPUT CURRENT AND VOLTAGE
The OPA693 provides output voltage and current capabilities
that can easily support multiple video loads and/or 100Ω
loads with very low distortion. Under no-load conditions at
25°C, the output voltage typically swings to 1V of either
supply rail; the tested swing limit is within 1.2V of either rail.
Into a 15Ω load (the minimum tested load), it is tested to
deliver more than ±90mA.
The specifications described above, though familiar in the
industry, consider voltage and current limits separately. In
many applications, it is the voltage × current, or V-I product,
which is more relevant to circuit operation. Refer to the
Output Voltage and Current Limitations plot in the Typical
Characteristics. The X and Y axes of this graph show the
zero-voltage output current limit and the zero-current output
voltage limit, respectively. The four quadrants give a more
detailed view of the OPA693’s output drive capabilities,
noting that the graph is bounded by a “Safe Operating Area”
of 1W maximum internal power dissipation. Superimposing
resistor load lines onto the plot shows that the OPA693 can
drive ±3.4V into 20Ω or ±3.7V into 50Ω without exceeding
18
either the output capabilities or the 1W dissipation limit. A
100Ω load line (the standard test-circuit load) shows full
±3.8V output swing capability, as shown in the Typical
Characteristics.
One of the most demanding, and yet very common, load
conditions for an op amp is capacitive loading. Often, the
capacitive load is the input of an ADC, including additional
external capacitance, which may be recommended to improve
ADC linearity. A high-speed, high open-loop gain, amplifier like
the OPA693 can be very susceptible to decreased stability
and may give closed-loop response peaking when a capacitive load is placed directly on the output pin. When the
amplifier’s open loop output resistance is considered, this
capacitive load introduces an additional pole in the signal path
that can decrease the phase margin. Several external solutions to this problem have been suggested. When the primary
considerations are frequency response flatness, pulse response fidelity and/or distortion, the simplest and most effective solution is to isolate the capacitive load from the feedback
loop by inserting a series isolation resistor between the amplifier output and the capacitive load. This does not eliminate the
pole from the loop response, but rather shifts it and adds a
zero at a higher frequency. The additional zero acts to cancel
the phase lag from the capacitive load pole, thus increasing
the phase margin and improving stability.
OPA693
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SBOS285
DISTORTION PERFORMANCE
The OPA693 provides good distortion performance into a
100Ω load on ±5V supplies. Relative to alternative solutions,
the OPA693 holds much lower distortion at higher frequencies
(> 20Mhz) than alternative solutions. Generally, until the
fundamental signal reaches very high frequency or power
levels, the 2nd harmonic will dominate the distortion with a
negligible 3rd harmonic component. Focusing then on the 2nd
harmonic, increasing the load impedance improves distortion
directly. Remember that the total load includes the feedback
network—in the noninverting configuration (see Figure 1) this
is the sum of RF + RG, while in the inverting configuration it is
just RF (see Figure 3). Also, providing an additional supply decoupling capacitor (0.01µF) between the supply pins (for
bipolar operation) improves the 2nd-order distortion slightly
(3dB to 6dB).
The OPA693 has an extremely low 3rd-order harmonic
distortion. This also produces a high 2-tone, 3rd-order intermodulation intercept. Two graphs for this intercept are given
in the in the Typical Characteristics; one for ±5V and one for
+5V. The lower curve shown in each graph is defined at the
50Ω load when driven through a 50Ω matching resistor, to
allow direct comparisons to RF MMIC devices. The higher
curve in each graph shows the intercept if the output is taken
directly at the output pin with a 500Ω load, to allow prediction
of the 3rd-order spurious level when driving a lighter load,
such as an ADC input. The output matching resistor attenuates the voltage swing from the output pin to the load by 6dB.
If the OPA693 drives directly into the input of a highimpedance device, such as an ADC, this 6dB attenuation is
not taken and the intercept will increase a minimum of 6dB,
as shown in the 500Ω load typical characteristic.
The intercept is used to predict the intermodulation spurious
levels for two closely-spaced frequencies. If the two test frequencies (f1 and f2) are specified in terms of average and delta
frequency, fO = (f1 + f2)/2 and ∆f = |f2 – f1|/2, then the two, 3rdorder, close-in spurious tones will appear at fO ±3 × ∆f. The
difference between two equal test tone power levels and these
GAIN ACCURACY AND LINEARITY
The OPA693 provides improved absolute gain accuracy and
DC linearity over earlier fixed gain of two line drivers. Operating at a gain of +2V/V by tying the –IN pin to ground, the
OPA693 shows a maximum gain error of ±0.9% at 25°C. The
DC gain will therefore lie between 1.982V/V and 2.018V/V at
room temperature. Over the specified temperature ranges,
this gain tolerance expands only slightly due to the matched
temperature drift for RF and RG. Achieving this gain accuracy
requires a very low impedance ground at –IN. Typical production lots show a much tighter distribution in gain than this
±0.9% specification. Figure 12 shows a typical distribution in
measured gain at the gain of +2V/V configuration, in this
case showing a slight drop in the mean (0.25%) from the
nominal but with a very tight distribution.
600
400
300
200
100
0
Gain(V/V)
Figure 12. Typical +2V/V Gain Distribution.
OPA693
SBOS285
Mean = 1.995
σ = 0.005
500
1.980
1.982
1.984
1.986
1.988
1.990
1.992
1.994
1.996
1.998
2.000
2.002
2.004
2.006
2.008
2.010
2.012
2.014
2.016
2.018
2.020
The criterion for setting this RS resistor is a maximum
bandwidth, flat frequency response at the load (< 0.1dB
peaking). For the OPA693 operating in a gain of +2, the
frequency response at the output pin is very flat to begin with,
allowing relatively small values of RS to be used for low
capacitive loads.
intermodulation spurious power levels is given by
∆dBc = 2 × (IM3 – PO), where IM3 is the intercept taken from
the Typical Characteristics and PO is the power level in dBm at
the 50Ω load for one of the two closely-spaced test frequencies.
For instance, at 50MHz, the OPA693 at a gain of +2 has an
intercept of 44dBm at a matched 50Ω load. If the full envelope
of the two frequencies needs to be 2VPP at this load, this
requires each tone to be 4dBm (1VPP). The 3rd-order intermodulation spurious tones will then be 2 × (44 – 4) = 80dBc
below the test tone power level (–76dBm). If this same 2VPP
2-tone envelope were delivered directly into a lighter 500Ω load,
the intercept would increase to the 52dBm shown in the Typical
Characteristics. With the same output signal and gain conditions, but now driving directly into a light load with no matching
loss, the 3rd-order spurious tones will then be at least
2 × (52 – 4) = 96dBc below the 4dBm test tone power levels
centered on 50MHz (–92dBm). We are still using a 4dBm for the
1VPP output swing into this 500Ω load. While not strictly correct
from a power standpoint, this does give the correct prediction for
spurious level. The class AB output stage for the OPA693 is
much more voltage swing dependent on output distortion than
strictly power dependent. To use the 500Ω intercept curve, use
the single-tone voltage swing as if it were driving a 50Ω load to
compute the PO used in the intercept equation.
Number of Units
The Typical Characteristics show a Recommended RS vs
Capacitive Load curve to help the designer pick a value to
give < 0.1dB peaking to the load. The resulting frequency
response curves show a 0.1dB peaked response for several
selected capacitive loads and recommended RS combinations. Parasitic capacitive loads greater than 2pF can begin
to degrade the performance of the OPA693. Long PC board
traces, unmatched cables, and connections to other amplifier
inputs can easily exceed this value. Always consider this
effect carefully, and add the recommended series resistor as
close as possible to the OPA693 output pin (see the Board
Layout Guidelines section).
www.ti.com
19
The exceptionally linear output stage (as illustrated by the
high 3rd-order intermodulation intercept) and low thermal
gradient induced errors for the OPA693 give an extremely
linear output over large voltage swings and heavy loads.
Figure 13 shows the tested deviation (in % of peak to peak)
from linearity for a range of symmetrical output swings and
loads. Below 4VPP, for either a 100Ω or a 500Ω load, the
OPA693 delivers > 14-bit linear output response.
The total output spot noise voltage can be computed as the
square root of the sum of all squared output noise voltage
contributors. Equation 1 shows the general form for the
output noise voltage using the terms shown in Figure 14.
(1)
2
2
EO =  ENI2 + (IBNR S ) + 4kTRS  NG2 + (IBIRF ) + 4kTRFNG
Dividing this expression through by noise gain (NG = 1 + RF/RG)
will give the equivalent input-referred spot noise voltage at the
non-inverting input, as shown in Equation 2.
0.0200
Figure 1 Test Circuit
0.0175
(2)
% Deviation
0.0150
2
4kTRF
2
I R 
EN = ENI2 + (IBNR S ) + 4kTRS +  BI F  +
 NG 
NG
0.0125
RL = 100Ω
0.0100
0.0075
Evaluating the output noise and input noise expressions for
the two noninverting gain configurations, and with two different values for the noninverting source impedance, gives
output and input referred spot noise voltages of Table II.
0.0050
RL = 500Ω
0.0025
0
2
3
4
5
6
7
8
RS
(Ω)
OUTPUT
SPOT NOISE
EO
(nV/ √Hz )
TOTAL INPUT
SPOT NOISE
EN
(nV/ √Hz )
4.15
VO (peak to peak)
Figure 13. DC Linearity vs Output Swing and Loads.
CONFIGURATION
NOISE PERFORMANCE
G = +2 (Figure 1)
25
8.3
G = +2 (Figure 1)
300
14
7
The OPA693 offers an excellent balance between voltage and
current noise terms to achieve a low output noise under a
variety of operating conditions. The inverting node noise
current (internal) will appear at the output multiplied by the
relatively low 300Ω feedback resistor. The input noise voltage
(1.8nV/√Hz) is extremely low for a unity gain stable amplifier.
This low input voltage noise was achieved at the price of
higher noninverting input current noise (17.8pA/√Hz). As long
as the AC source impedance looking out of the noninverting
input is less than 100Ω, this current noise will not contribute
significantly to the total output noise. The op amp input voltage
noise and the two input current noise terms combine to give
low output noise for the each of the three gain settings
available using the OPA693. Figure 14 shows the op amp
noise analysis model with all of the noise terms included. In
this model, all noise terms are taken to be noise voltage or
current density terms in either nV/√Hz or pA/√Hz.
G = +1 (Figure 2)
25
7.3
7.3
G = +1 (Figure 2)
300
9.2
9.2
ENI
EO
OPA693
RS
IBN
ERS
RF
√4kTRS
4kT
RG
RG
Figure 14. Op Amp Noise Model.
20
IBI
√4kTRF
4kT = 1.6E –20J
at 290°K
TABLE II. Total Output and Input Referred Noise.
The output noise is being dominated by the inverting current
noise times the internal feedback resistor. This gives a total
input referred noise voltage that exceeds the 1.8nV voltage
term for the amplifier itself.
DC ACCURACY AND OFFSET CONTROL
A current-feedback op amp like the OPA693 provides exceptional bandwidth and slew rate giving fast pulse settling but
only moderate DC accuracy. The Electrical Characteristics
show an input offset voltage comparable to high-speed voltage-feedback amplifiers. However, the two input bias currents
are somewhat higher and are unmatched. Whereas bias
current cancellation techniques are very effective with most
voltage-feedback op amps, they do not generally reduce the
output DC offset for wideband current-feedback op amps.
Since the two input bias currents are unrelated in both magnitude and polarity, matching the source impedance looking
out of each input to reduce their error contribution to the output
is ineffective. Evaluating the configuration of Figure 1, using
worst case +25°C input offset voltage and the two input bias
currents, gives a worst-case output offset range equal to:
±(NG × VOS) + (IBN × RS/2 × NG) ± (IBI × RF)
= ±(2 × 2.0mV) ± (35µA × 25Ω × 2) ± (50µA × 300Ω)
= ±4mV ± 1.75mV ± 15mV
= ±30.75mV
where NG = noninverting signal gain.
OPA693
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SBOS285
Minimizing the resistance seen by the noninverting input will
minimize the output DC error. For improved DC precision in
a wideband low-gain amplifier, consider the OPA842 where
a bipolar input is acceptable (low source resistance) or the
OPA656 where a JFET input is required.
DISABLE OPERATION
The OPA693 provides an optional disable feature that can be
used to reduce system power. If the V DIS control pin is left
unconnected, the OPA693 will operate normally. This shutdown is intended only as a power-savings feature. Forward
path isolation when disabled is very good for small signals for
gains of +1 or +2. Large-signal isolation is not ensured. Using
this feature to multiplex two or more outputs together is not
recommended. Large signals applied to the disabled output
stages can turn on parasitic devices degrading signal linearity for the desired channel.
Turn-on time is very quick from the shutdown condition,
typically < 60ns. Turn-off time is strongly dependent on the
selected gain configuration and load, but is typically 3µs for
the circuit of Figure 1.
To shutdown, the control pin must be asserted low. This logic
control is referenced to the positive supply, as shown in the
simplified circuit of Figure 15.
+VS
15kΩ
Q1
The OPA693 does not require heatsinking or airflow in most
applications. Maximum desired junction temperature sets the
maximum allowed internal power dissipation as described
here. In no case should the maximum junction temperature
be allowed to exceed 150°C.
Operating junction temperature (TJ) is given by TA + PD × θJA.
The total internal power dissipation (PD) is the sum of
quiescent power (PDQ) and additional power dissipated in the
output stage (PDL) to deliver load power. Quiescent power is
simply the specified no-load supply current times the total
supply voltage across the part. PDL will depend on the
required output signal and load but would, for a grounded
resistive load, be at a maximum when the output is fixed at
a voltage equal to 1/2 either supply voltage (for equal bipolar
supplies). Under this worst-case condition, PDL = VS2/(4 × RL)
where RL includes feedback network loading. This is the
absolute highest power that can be dissipated for a given RL.
All actual applications will dissipate less power in the output
stage.
As a worst-case example, compute the maximum TJ using an
OPA693IDBV (SOT23-6 package) in the circuit of Figure 1
operating at the maximum specified ambient temperature of
+85°C and driving a grounded 100Ω load. Maximum internal
power is:
25kΩ
IS
Control
THERMAL ANALYSIS
Note that it is the power in the output stage and not into the
load that determines internal power dissipation.
110kΩ
VDIS
The shutdown feature for the OPA693 is a positive supply
referenced, current-controlled, interface. Open collector (or
drain) interfaces are most effective, as long as the controlling
logic can sustain the resulting voltage (in the open mode)
that will appear at the V DIS pin. That voltage will be one diode
below the positive supply voltage applied to the OPA693. For
voltage output logic interfaces, the on/off voltage levels
described in the Electrical Characteristics apply only for a
+5V positive supply on the OPA693. An open-drain interface
is recommended for shutdown operation using a higher
positive supply for the OPA693 and/or logic families with
inadequate high-level voltage swings.
–VS
PD = 10V × 14.1mA + 52 /(4 × (100Ω+|| 600Ω)) = 214mW
Figure 15. Simplified Disable Control Circuit.
Maximum TJ = +85°C + (0.21W × 150°C/W) = 117°C.
In normal operation, base current to Q1 is provided through
the 110kΩ resistor while the emitter current through the 15kΩ
resistor sets up a voltage drop that is inadequate to turn on
the two diodes in Q1’s emitter. As V DIS is pulled LOW,
additional current is pulled through the 15kΩ, eventually
turning on these two diodes (≈80µA). At this point, any further
current pulled out of VDIS goes through those diodes holding
the emitter-base voltage of Q1 at approximately 0V. This
shuts off the collector current out of Q1, turning the amplifier
off. The supply current in the shutdown mode is only that
required to operate the circuit of Figure 15.
All actual applications will operate at a lower junction temperature than the 117°C computed above. Compute your
actual output stage power to get an accurate estimate of
maximum junction temperature, or use the results shown
here as an absolute maximum.
OPA693
SBOS285
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21
BOARD LAYOUT GUIDELINES
Achieving optimum performance with a high-frequency amplifier
like the OPA693 requires careful attention to PC board layout
parasitics and external component types. Recommendations
that will optimize performance include:
a) Minimize parasitic capacitance to any AC ground for all
of the signal I/O pins. Parasitic capacitance on the output can
cause instability; on the noninverting input, it can react with the
source impedance to cause unintentional bandlimiting. To
reduce unwanted capacitance, create a window around the
signal I/O pins in all of the ground and power planes around
those pins. Otherwise, ground and power planes should be
unbroken elsewhere on the board.
b) Minimize the distance (< 0.25”) from the power supply pins
to high frequency 0.1µF decoupling capacitors. At the device
pins, the ground and power plane layout should not be in close
proximity to the signal I/O pins. Avoid narrow power and ground
traces to minimize inductance between the pins and the decoupling capacitors. The power supply connections should always
be decoupled with these capacitors. Larger (2.2µF to 6.8µF)
decoupling capacitors, effective at lower frequency, should also
be used on the supply pins. These may be placed somewhat
farther from the device and may be shared among several
devices in the same area of the PC board.
c) Careful selection and placement of external components will preserve the high frequency performance of the
OPA693. Use resistors that have low reactance at high
frequencies. Surface-mount resistors work best and allow a
tighter overall layout. Metal film and carbon composition axially-leaded resistors can also provide good high-frequency
performance. Again, keep their leads and PC board trace
length as short as possible. Never use wirewound type resistors in a high-frequency application. Since the output pin and
inverting input pin are the most sensitive to parasitic capacitance, always position the series output resistor, if any, as
close as possible to the output pin. Since the inverting input
node is internal for the OPA693, it is more robust to layout
issues than amplifiers with similar speed but external feedback
and gain resistors. Other network components, such as
noninverting input termination resistors, should also be placed
close to the package. Good axial metal film or surface mount
resistors have approximately 0.2pF in shunt with the resistor.
For resistor values > 2.0kΩ, this parasitic capacitance can add
a pole and/or zero below 400MHz that can effect circuit
operation. Keep resistor values as low as possible consistent
with load driving considerations.
d) Connections to other wideband devices on the PC board
may be made with short direct traces or through onboard
transmission lines. For short connections, consider the trace
and the input to the next device as a lumped capacitive load.
Relatively wide traces (50 to 100mils) should be used, preferably with ground and power planes opened up around them.
Estimate the total capacitive load and set RS from the plot of
Recommended RS vs Capacitive Load. Low parasitic capacitive
loads (< 4pF) may not need an RS since the OPA693 is
nominally compensated to operate with a 2pF parasitic load. If
a long trace is required, and the 6dB signal loss intrinsic to a
doubly-terminated transmission line is acceptable, implement a
22
matched impedance transmission line using microstrip or stripline
techniques (consult an ECL design handbook for microstrip and
stripline layout techniques). A 50Ω environment is normally not
necessary on board, and in fact, a higher impedance environment will improve distortion, as shown in the distortion versus
load plots. With a characteristic board trace impedance defined
based on board material and trace dimensions, a matching
series resistor into the trace from the output of the OPA693 is
used, as well as a terminating shunt resistor at the input of the
destination device. Remember also that the terminating impedance will be the parallel combination of the shunt resistor and
the input impedance of the destination device; this total effective
impedance should be set to match the trace impedance. If the
6dB attenuation of a doubly-terminated transmission line is
unacceptable, a long trace can be series-terminated at the
source end only. Treat the trace as a capacitive load in this case
and set the series resistor value as shown in the plot of
Recommended RS vs Capacitive Load. This will not preserve
signal integrity as well as a doubly-terminated line. If the input
impedance of the destination device is low, there will be some
signal attenuation due to the voltage divider formed by the
series output into the terminating impedance.
e) Socketing a high-speed part like the OPA693 is not
recommended. The additional lead length and pin-to-pin
capacitance introduced by the socket can create an extremely
troublesome parasitic network, which can make it almost
impossible to achieve a smooth, stable frequency response.
Best results are obtained by soldering the OPA693 directly
onto the board.
INPUT AND ESD PROTECTION
The OPA693 is built using a very high-speed complementary
bipolar process. The internal junction breakdown voltages are
relatively low for these very small geometry devices. These
breakdowns are reflected in the Absolute Maximum Ratings
table. All device pins are protected with internal ESD protection diodes to the power supplies, as shown in Figure 16.
+V CC
External
Pin
Internal
Circuitry
–V CC
Figure 16. Internal ESD Protection.
These diodes provide moderate protection to input overdrive
voltages above the supplies as well. The protection diodes can
typically support 30mA continuous current. Where higher
currents are possible (for example, in systems with ±15V
supply parts driving into the OPA693), current limiting series
resistors may be added on the noninverting input. Keep this
resistor value as low as possible since high values degrade
both noise performance and frequency response. The inverting input already has a 300Ω resistor from the external pin to
the internal summing junction for the op amp. This provides
considerable protection for that node.
OPA693
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SBOS285
MECHANICAL DATA
MSOI002B – JANUARY 1995 – REVISED SEPTEMBER 2001
D (R-PDSO-G**)
PLASTIC SMALL-OUTLINE PACKAGE
8 PINS SHOWN
0.020 (0,51)
0.014 (0,35)
0.050 (1,27)
8
0.010 (0,25)
5
0.008 (0,20) NOM
0.244 (6,20)
0.228 (5,80)
0.157 (4,00)
0.150 (3,81)
Gage Plane
1
4
0.010 (0,25)
0°– 8°
A
0.044 (1,12)
0.016 (0,40)
Seating Plane
0.010 (0,25)
0.004 (0,10)
0.069 (1,75) MAX
PINS **
0.004 (0,10)
8
14
16
A MAX
0.197
(5,00)
0.344
(8,75)
0.394
(10,00)
A MIN
0.189
(4,80)
0.337
(8,55)
0.386
(9,80)
DIM
4040047/E 09/01
NOTES: A.
B.
C.
D.
All linear dimensions are in inches (millimeters).
This drawing is subject to change without notice.
Body dimensions do not include mold flash or protrusion, not to exceed 0.006 (0,15).
Falls within JEDEC MS-012
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
IMPORTANT NOTICE
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amplifier.ti.com
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dataconverter.ti.com
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