Intersil ISL6726AAZ Active clamp forward pwm controller Datasheet

Active Clamp Forward PWM Controller
ISL6726
Features
The ISL6726 is a highly featured single-ended PWM controller
intended for applications using the active clamp forward
converter topology in either n- or p-channel active clamp
configurations, the asymmetric half-bridge topology, and the
standard forward topologies with synchronous rectification. It is a
current-mode PWM controller with many features designed to
simplify its use. Among its many features are a precision
oscillator which allows accurate control of the deadtime and
maximum duty cycle, bi-directional synchronization with 180°
phase shift for interleaving applications, adjustable soft-start and
soft-stop, a low power disable mode, and average current limit
for “brick-wall” overcurrent protection.
• Precision Maximum Duty Cycle and Deadtime Control
This advanced BiCMOS design features low start-up and
operating currents, adjustable switching frequency to greater
than 1MHz, high current FET drivers, and very low propagation
delays for a fast response to overcurrent faults.
• Supports N- and P-Channel Active Clamp FETs
Applications
• Programmable Undervoltage Lock-Out (UV)
• 125µA Typical Start-up Current
• Adjustable Peak and Average Current Limit Protection
• Programmable Oscillator Frequency
• Bi-Directional Synchronization with 180° Phase Shift for
Interleaved Converter Applications
• Adjustable Soft-Start and Selectable Soft-Stop
• Selectable Minimum Duty Cycle Clamp for Synchronous
Rectifier Applications
• Programmable Slope Compensation
• Programmable Switch Timing Between Main and Active Clamp
Outputs
• Input Voltage Dependent Duty Cycle Clamp
• Telecom and Datacom Power Supplies
• ENABLE Input with Low Power Disable
• AC/DC Power Supplies
• Internal Over-Temperature Protection
• Battery Chargers
• Pb-Free (RoHS Compliant)
+VIN
OUTM
+VIN
LEVEL
SHIFT
LEVEL
SHIFT
OUTAC
OUTAC
OUTM
OUTAC
-VIN
-VIN
N- or P- CHANNEL ACTIVE CLAMP FORWARD
January 31, 2011
FN7654.0
1
ASYMMETRIC HALF-BRIDGE
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas Inc. 2011. All Rights Reserved
Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries.
All other trademarks mentioned are the property of their respective owners.
ISL6726
Pin Configuration
ISL6726
(20 LD QSOP)
TOP VIEW
SYNC
1
20
SS
DCLIM
2
19
MODE
UV
3
18
DELAY
ENABLE
4
17
VREF
RTC
5
16
GND
CT
6
15
OUTM
ISET
7
14
VDD
VERR
8
13
OUTAC
FB
9
12
SLOPE
CS 10
11
IOUT
Pin Descriptions
PIN #
SYMBOL
DESCRIPTION
1
SYNC
A bi-directional edge-sensitive signal used to synchronize multiple devices together. If the SYNC pins of two units are connected, they
will synchronize 180 degrees out of phase with each other. This feature facilitates the design of interleaved topologies. If more than
two units are connected, one will be the master unit and the rest will be slave units. All of the slave units will synchronize 180 degrees
out-of-phase with the master. The master designation is not fixed or predetermined and is self-arbitrating. The master is determined
by the fastest running oscillator on a dynamic basis. SYNC may also be used to synchronize to an external clock.
2
DCLIM
Used in conjunction with UV, DCLIM creates a duty cycle clamp that is dependent on the input voltage. As the input voltage increases,
the maximum allowed duty cycle decreases. This feature is necessary in the active clamp forward to help prevent transformer core
saturation during transients. A resistor divider from VREF sets the threshold of DCLIM.
3
UV
Sets the user programmable undervoltage threshold. Placing a resistor divider from the input voltage to ground and set to 1.00V
determines the minimum operating voltage. The amount of hysteresis is determined by an internal current source and set by the
external impedance of the divider. The current source is active when UV is below 1V.
4
ENABLE
A logic level signal used to enable the IC. When the input is open, the IC is enabled and a soft-start cycle begins if no fault conditions
are present. When pulled low, the outputs are disabled and the IC enters a low power sleep state. If soft-stop is enabled, a logic “0”
on ENABLE forces a soft-stop prior to entering the low power sleep state.
5
RTC
6
CT
The oscillator timing capacitor charge/discharge current control pin. A resistor is connected between this pin and GND and
determines the magnitude of the charge and discharge current. The charge current is nominally 2x the current flowing into the
resistor. The discharge current is nominally 8x the current flowing into the resistor. The ratio of the charge to discharge current is
fixed and sets the maximum duty cycle at 80%.
The oscillator timing capacitor is connected between this pin and GND.
7
ISET
Controls the peak and average current limit thresholds. A voltage up to 1.0V may be applied to ISET.
8
VERR
The error voltage input to the PWM comparator and the compensation connection for the average current loop control. VERR
requires an external pull-up resistor to VREF. A typical application connects the photo-transistor output of an opto-coupler between
VERR and GND.
9
FB
FB is the inverting input to the average current error amplifier (IEA). The amplifier is used as the error amplifier for the average
current limit control loop. If the amplifier is not used, FB should be grounded. The amplifier is normally configured as an integrator.
10
CS
The current sense input to the IC. Provides information to the PWM, the peak overcurrent protection comparators, and the average
current limit circuitry. The CS pin is shorted to GND when the PWM output pulse terminates. Depending on the current sensing
source impedance, a series input resistor may be required due to the delay between the internal logic and the turn off of the external
power switch.
11
IOUT
Output of the sample and hold buffer amplifier that captures and averages the CS signal. With a nominal 4x multiplier and the ability
to scale the signal externally with a resistor divider, the average current limit can be set independently of the peak current limit.
2
FN7654.0
January 31, 2011
ISL6726
Pin Descriptions
(Continued)
PIN #
SYMBOL
DESCRIPTION
12
SLOPE
A slope compensation capacitor is connected between SLOPE and GND. A current source of 100µA charges the capacitor during
the On time and discharges it during the Off time. The amplitude of the signal is multiplied by a gain of 0.2 and summed with the
CS input.
13
OUTAC
The Active Clamp output for driving an external power switch. OUTAC is capable of driving either a p- or n- channel clamp device and
is configured by DELAY.
14
VDD
VDD is the power connection for the IC. To optimize noise immunity, bypass VDD to GND with a ceramic capacitor as close to the
VDD and GND pins as possible. VDD is monitored for undervoltage (UVLO). When VDD is below the UVLO threshold, the IC is disabled
and the reference voltage, VREF, is turned off.
15
OUTM
The main PWM output for driving an external power switch.
16
GND
Logic and power ground for this device. Due to high peak currents and high frequency operation, a low impedance layout is
necessary. Ground planes and short traces are highly recommended.
17
VREF
The 5.00V reference voltage output having a -2/+1.5% tolerance over line, load and operating temperature. Bypass to GND
with a 0.1µF to 2.2µF low ESR capacitor. VREF can source up to 10mA.
18
DELAY
The DELAY pin configures OUTAC for either n-channel or p-channel drive compatibility by setting the phase and the duration when
both the main and active clamp outputs are off. A resistor from DELAY to VREF sets an out-of-phase (non-overlap) relationship for
an n-channel clamp device with adjustable deadtime. A resistor from DELAY to GND sets an in-phase (overlap) relationship for a
p-channel clamp device with an adjustable symmetric non-overlap duration between OUTM and OUTAC.
19
MODE
The MODE pin configures the IC for standard or synchronous rectification operation. If MODE is connected to VREF, standard
rectification operation is selected. Soft-stop and the minimum duty cycle clamp are disabled. If MODE is connected to GND,
synchronous rectification operation is enabled allowing soft-stop and the minimum duty cycle clamp to function.
20
SS
Connect the soft-start timing capacitor between this pin and GND to control the duration of soft-start and soft-stop. The value of the
SS capacitor determines the rate of increase and decrease of the duty cycle during start-up and soft-stop. Soft-stop is
enabled/disabled by MODE.
Ordering Information
PART NUMBER
(Notes 1, 2, 3)
ISL6726AAZ
PART MARKING
ISL6726 AAZ
TEMP.
RANGE (°C)
-40 to +105
PACKAGE
(Pb-free)
20 Ld QSOP
PKG. DWG.
#
M20.15
NOTES:
1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte
tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil
Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page for ISL6726. For more information on MSL, please see Technical Brief TB363.
3
FN7654.0
January 31, 2011
Internal Architecture
VREF
SYNC
VDD
VREF
5.00 V
1%
VDD
UVLOBIAS
OUT
4
INTERNAL
OT SHUTDOWN
130 - 150 C
OUTM
CLK
ON
UVLO
PWM
DISABLE
SYNC
OUTPUT DELAY CONTROL
AND
STEERING LOGIC
DELAY
DELAY
OUTCLAMP
OUTAC
GND
1.00 V
UV
Bi-Directional
SYNC Circuit
+
-
INHIBIT
EXT. SYNC
SYNC OUT
Phase-Shifted
180 º
CLK
IRTC
OSCILLATOR
RTC
CS
CT
IOUT
IOUT
ISL6726
____
CLK
AVERAGE
CURRENT LIMIT
CLK
I CH= 2 x IRTC
VFF = 1.0 - 5.0 V
VREF
ICH
ON
CURRENT
AMPLIFIER
-
I DCH = 8 x IRTC
VFF = 1.0 - 5.0 V
CT
+
FB
ISET
IDCH
+
-
DCLIM
ON
LEADING EDGE BLANKING
INHIBIT
ISET
+
-
OC DETECT
DISABLE
CS
ENABLE
SOFT-START/
SOFT-STOP
CS
OC
SS
DUTY LIMIT
SLOPE
SLOPE
VREF
PWM OUT
VREF
PWM
FN7654.0
January 31, 2011
VERR
UVLOBIAS
VERROR
SOFT-START
MODE
MODE
Typical Application Using ISL6726 - Active Clamp Forward with Synchronous Rectification
T1
VIN+
L1
T2
+Vout
Q3
Q4
+
C13
5
R1
RETURN
CR1
C12
R22
Q1
C1
Q2
R22
Q5
R14
CR2
VIN-
R24
R23
1 SYNC
SS 20
2 DCLIM U1 MODE 19
SYNC
6 CT
C5
ISL6726
5 RTC
7 ISET
R8
R5
C9
C14
CR3
T2
VREF 17
GND 16
ISL6726
4 ENABLE
ON/OFF
R13
DELAY 18
3 UV
C15
OUTM 15
R25
VDD 14
8 VERR
OUTAC 13
9 FB
SLOPE 12
10 CS
IOUT 11
R16
R15
R4
R19
R18
C11
R7
C2
R17
VCC (+10V)
R21
R10
R2
R3
C3
R6
R9
R11
C4
C6
C7
R12
C10
U2
C8
VR1
U4
R20
FN7654.0
January 31, 2011
Typical Application Using ISL6726 - Active Clamp Forward with Diode Rectification
T1
VIN+
L1
+Vout
CR3
T2
CR4
+
C13
RETURN
6
CR1
C12
R1
Q2
Q1
C1
R22
R14
CR2
VIN-
SS 20
2 DCLIM U1 MODE 19
1 SYNC
SYNC
6 CT
C5
ISL6726
5 RTC
7 ISET
R8
R5
C9
DELAY 18
4 ENABLE
ON/OFF
R13
ISL6726
3 UV
VREF 17
GND 16
OUTM 15
VDD 14
8 VERR
OUTAC 13
9 FB
SLOPE 12
10 CS
IOUT 11
R15
R16
R19
R18
R4
C11
R7
C2
R17
VCC (+10V)
R21
R10
C10
U2
R2
R3
C3
R6
R9
C6
C7
C8
R11
C4
R12
VR1
U4
R20
FN7654.0
January 31, 2011
ISL6726
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +22.0V
OUTM, OUTAC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to VDD
Signal Pins. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to 5V
Peak GATE Current, OUTM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3A
Peak GATE Current, OUTAC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2A
ESD Rating
Human Body Model (Tested per JESD22-A114) . . . . . . . . . . . . . . . . . 3kV
Machine Model (Tested per JESD22-A115). . . . . . . . . . . . . . . . . . . 250V
Charged Device Model (Tested per JESD-C101E) . . . . . . . . . . . . . . 1.5kV
Latch Up (Tested per JESD-78B; Class2, Level A) . . . . . . . . . . . . . . . 100mA
Thermal Resistance (Typical)
θJA (°C/W) θJC (°C/W)
20 Lead QSOP (Notes 4, 5) . . . . . . . . . . . . .
86
38
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . .-55°C to +150°C
Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°C
Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Operating Conditions
Temperature Range
ISL6726Axx. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +105°C
Supply Voltage Range (Typical). . . . . . . . . . . . . . . . . . . . . . . 9VDC to 16VDC
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTES:
4. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
5. For θJC, the “case temp” location is taken at the package top center.
6. All voltages are to be measured with respect to GND, unless otherwise specified.
Electrical Specifications Recommended operating conditions unless otherwise noted. Refer to the Block Diagram on page 4 and the
Typical Application schematics on pages 5 and 6. 8V < VD < 20V, RTC = 10.0kΩ, CT = 470pF, TA = -40°C to +105°C, Typical values are at TA = +25°C.
Boldface limits apply over the operating temperature range, -40°C to +105°C.
PARAMETER
TEST CONDITIONS
MIN
(Note 7)
TYP
MAX
(Note 7)
UNITS
12
20
V
125
400
µA
mA
SUPPLY VOLTAGE
Supply Voltage
Start-Up Current, IDD
VDD < START Threshold
50
Operating Current, IDD
COUTM, OUTAC = 0nF, VDD = 12V
10
11
COUTM, OUTAC = 0nF, VDD = 20V
12
13
COUTM, OUTAC = 1nF, VDD = 12V
18
20
mA
UVLO START Threshold
7.40
7.65
8.00
V
UVLO STOP Threshold
6.00
6.23
7.00
V
Hysteresis
1.00
1.40
2.00
V
4.900
5.00
5.075
V
VOLTAGE REFERENCE
Overall Accuracy
IVREF = 0 to -10mA
Long Term Stability
TA = +125°C, 1000 hours
3
Operational Current (Source)
-10
Current Limit
-25
mV
mA
-100
mA
0.35
V
CURRENT SENSE
Current Limit Threshold, ISET Minimum
Current Limit Threshold, ISET Maximum
1.000
CS to OUT Delay
CS to PWM Comparator Offset
VSLOPE = 0V
90
CS Discharge Device, rDS(ON)
1.125
V
100
ns
100
110
mV
15
31
Ω
CS Input Bias Current
-1
1
µA
ISET Input Bias Current
-1
1
µA
Leading Edge Blanking (LEB) Duration
75
130
ns
7
100
FN7654.0
January 31, 2011
ISL6726
Electrical Specifications Recommended operating conditions unless otherwise noted. Refer to the Block Diagram on page 4 and the
Typical Application schematics on pages 5 and 6. 8V < VD < 20V, RTC = 10.0kΩ, CT = 470pF, TA = -40°C to +105°C, Typical values are at TA = +25°C.
Boldface limits apply over the operating temperature range, -40°C to +105°C. (Continued)
PARAMETER
TEST CONDITIONS
IOUT Buffer Gain
TA = +25°C
IOUT VOH
ΔVOH, (VOHILOAD=0µA - VOHILOAD=-300µA), CS = 0.5V,
ISET = 1.00V
IOUT VOL
ILOAD = 100µA, CS = 0V, ISET = 1.00V
MIN
(Note 7)
TYP
MAX
(Note 7)
UNITS
3.90
4.00
4.12
V/V
0.05
0.2
V
0.05
0.1
0.15
V
-90
-100
-110
µA
0.190
0.200
0.210
V/V
50
Ω
2.5
V
0
ns
SLOPE COMPENSATION
Charge Current
SLOPE = 2V
Slope Compensation Gain
Fraction of slope voltage added to CS
SLOPE Discharge Device, rDS(ON)
SLOPE Range, Linear Response
0
PULSE WIDTH MODULATOR
Minimum Duty Cycle
MODE = 5V, VERR < 0.6V
MODE = 0V, VERR < 0.6V
Maximum Duty Cycle
4.8 < VERR < VREF, UV = 4.2V, DCLIM > 4.0V
270
300
350
ns
76
80
84
%
RTC = 25.5kΩ, CT = 220pF
80
%
SS to PWM Comparator Input Gain
0.23
0.25
0.27
V/V
VERR to PWM Comparator Input Gain
0.28
0.30
0.32
V/V
VERR to PWM Comparator Input Offset
0.60
0.80
1.00
V
Overlap, TA = +25°C
1.54
1.83
2.11
ns/kΩ
Non-Overlap, TA = +25°C
1.54
1.79
2.11
ns/kΩ
500
ns
OUTM TO OUTAC DELAY TIMING
DELAY Gain
DELAY Range
50
DELAY Disable, High
4.9
V
DELAY Disable, Low
0.100
V
338
349
kHz
OSCILLATOR
Frequency Accuracy
TA = +25°C
Frequency Variation with VDD
TA = +105°C, |(F20V - F8V)/F8V|, UV = 2.00V (Note 7)
0.1
0.3
%
TA = +25°C, |(F20V - F8V)/F8V|, UV = 2.00V
0.2
0.6
%
TA = -40°C, |(F20V - F8V)/F8V|, UV = 2.00V (Note 7)
0.6
8.0
%
TA = +25°C, |(F4.25V - F2.00V)/F2.00V|
VDD = 8V
0.1
1.5
VDD = 20V
0.3
5.2
%
Temperature Stability
UV = 2.0V, VDD = 8V
0.5
1.5
%
Charge Current Gain
RTC = 10.0kΩ, 100kΩ
1.88
2.0
2.12
µA/µA
Discharge Current Gain
RTC = 10.0kΩ, 100kΩ
6.0
7.2
8.2
µA/µA
CT Valley Voltage
Static operation
0.75
0.80
0.85
V
CT Peak Voltage
Static operation
UV = 2.00V
2.30
2.40
2.50
V
UV = 4.00V
3.80
4.00
4.20
V
Frequency Variation with UV
8
326
%
FN7654.0
January 31, 2011
ISL6726
Electrical Specifications Recommended operating conditions unless otherwise noted. Refer to the Block Diagram on page 4 and the
Typical Application schematics on pages 5 and 6. 8V < VD < 20V, RTC = 10.0kΩ, CT = 470pF, TA = -40°C to +105°C, Typical values are at TA = +25°C.
Boldface limits apply over the operating temperature range, -40°C to +105°C. (Continued)
MIN
(Note 7)
TYP
MAX
(Note 7)
UNITS
RLOAD = Open
UV = 2.00V
1.55
1.60
1.65
V
UV = 4.00V
3.10
3.20
3.30
V
ISS Charge Current
SS = 2V
-45
-55
-65
µA
ISS Discharge Current, Absolute Value
MODE = 0V, SS = 2V
-45
-55
-65
µA
4.5
4.6
4.7
V
0.15
0.20
0.25
V
PARAMETER
TEST CONDITIONS
RTC Voltage
SOFT-START
SS Clamp Voltage
Reset Threshold Voltage
SS decreasing
SS Discharge Current
MODE = 5V, SS = 2V
10.0
mA
UV UNDERVOLTAGE
Input Voltage Low/Inhibit Threshold
0.97
1.00
1.03
V
Hysteresis, Switched Current Amplitude
6.2
10
14
µA
Input High Clamp Voltage
4.8
V
1
MΩ
Input Impedance
Maximum Control Voltage
4.20
VREF
V
OUTPUT OUTM, OUTAC
High Level Output Voltage (VOH)
OUTM
VDD - VOUTM or VOUTAC
0.5
V
OUTAC
IOUT = -100mA
1.0
V
IOUT = 100mA
0.5
V
1.0
V
15
30
ns
30
60
ns
10
20
ns
20
40
ns
1.5
V
-1
1
µA
4.00
VREF
V
Low Level Output Voltage (VOL)
OUTM
OUTAC
Rise Time
OUTM
CGATE = 1nF, VDD = 8V
OUTAC
Fall Time
OUTM
CGATE = 1nF, VDD = 8V
OUTAC
UVLO Output Voltage Clamp
VDD = 5V
OUTM, OUTAC ILOAD = 1mA
DCLIM
Input Bias Current
Maximum Control Voltage
MODE
High Level Input Voltage (VIH)
2
V
Low Level Input Voltage (VIL)
0.8
Pull-up Resistance, Internal
100
9
V
kΩ
FN7654.0
January 31, 2011
ISL6726
Electrical Specifications Recommended operating conditions unless otherwise noted. Refer to the Block Diagram on page 4 and the
Typical Application schematics on pages 5 and 6. 8V < VD < 20V, RTC = 10.0kΩ, CT = 470pF, TA = -40°C to +105°C, Typical values are at TA = +25°C.
Boldface limits apply over the operating temperature range, -40°C to +105°C. (Continued)
PARAMETER
TEST CONDITIONS
MIN
(Note 7)
TYP
MAX
(Note 7)
UNITS
ENABLE
High Level Input Voltage (VIH)
2
V
Low Level Input Voltage (VIL)
0.8
Pull-up Resistance, Internal
275
V
kΩ
ERROR AMPLIFIER
Operating Input Range
VERR = FB, ISET = 0V, 4V
0
Unity Gain Band-Width Product
4.00
8
FB Bias Current
-1
VERR VOL
ILOAD = 5mA, FB = VREF, ISET = 1V
VERR Pull-up Current Source
VERR = FB = 0V, ISET = 1V
V
MHz
1
µA
0.40
V
100
µA
SYNCHRONIZATION
VIL
0.8
VIH
2.0
VOL
ILOAD = 10μA
VOH
ILOAD = -1.0mA
Source Current
Sink Current
V
100
4.0
V
mV
4.5
V
VOH > 2.0V
-10
mA
VOL < 2.5V
10
mA
Output Duration
200
Input Duration, Minimum
100
ns
2
MHz
Maximum Frequency, Input
575
ns
THERMAL PROTECTION
Thermal Shutdown
145
°C
Thermal Shutdown Clear
130
°C
15
°C
Hysteresis, Internal Protection
NOTE:
7. Compliance to datasheet limits is assured by one or more methods: production test, characterization and/or design.
10
FN7654.0
January 31, 2011
ISL6726
Typical Performance Curves
1.010
1000
NORMALIZED FREQUENCY
FREQUENCY (kHz)
RTC = 10kΩ
RTC = 25kΩ
100
RTC = 50kΩ
10
0.1
1
CT (nF)
1.005
1.000
0.995
0.990
-40
10
-25
-10
5
20
35
50
65
80
95
110
TEMPERATURE (°C)
FIGURE 1. OSCILLATOR FREQUENCY vs CT and RTC
FIGURE 2. OSCILLATOR FREQUENCY vs TEMPERATURE
600
1.05
1.04
OVERLAP
RDELAY = 20k
1.03
NORMALIZED DELAY
DELAY TIME (ns)
500
400
300
NON-OVERLAP
200
1.02
1.01
RDELAY = 120k
1.00
0.99
0.98
RDELAY = 200k
0.97
100
0.96
0
0
0.95
-40
20 40 60 80 100 120 140 160 180 200 220 240 260 280 300
-25
-10
5
20
35
50
65
80
95
110
TEMPERATURE (°C)
DELAY RESISTANCE (kΩ)
FIGURE 3. DELAY TIME vs RESISTANCE
FIGURE 4. DELAY TIME vs TEMPERATURE (OVERLAP)
1.05
1.010
1.04
RDELAY = 20k
1.005
1.02
1.01
NORMALIZED VREF
NORMALIZED DELAY
1.03
RDELAY = 120k
1.00
0.99
0.98
RDELAY = 200k
1.000
0.995
0.97
0.96
0.95
-40
-25
-10
5
20
35
50
65
80
95
TEMPERATURE (°C)
FIGURE 5. DELAY TIME vs TEMPERATURE (NON-OVERLAP)
11
110
0.990
-40
-25
-10
5
20
35
50
65
80
95
110
TEMPERATURE (°C)
FIGURE 6. VREF vs TEMPERATURE
FN7654.0
January 31, 2011
ISL6726
Functional Description
The timing component tolerance directly effects the oscillator
accuracy. A NPO/COG dielectric ceramic capacitor or better is
suggested for CT. RTC should be 1% tolerance or better.
Features
The ISL6726 PWM is an excellent choice for low cost high
performance applications requiring D (duty cycle) and 1-D
control signals. This includes active clamp forward, asymmetric
half-bridge, and synchronous rectified (SR) standard forward and
flyback topologies. Among its many features are:
• High current FET drivers
• Adjustable soft-start and soft-stop
Figure 1 graphically portrays the oscillator frequency as function
of the timing components. The minimum deadtime is fixed at
20% of the period allowing an 80% maximum duty cycle. This
limits the maximum voltage stress on the power MOSFETs to 5x
the input voltage. For applications that cannot tolerate this
voltage stress, the maximum duty cycle can be reduced using
the DCLIM feature. The peak voltage stress for an active clamp
topology is approximately VIN/(1-D).
• Slope compensation
Soft-Start/Soft-Stop Operation
• Programmable deadtime control
The ISL6726 features a soft-start using an external capacitor in
conjunction with an internal current source. Soft-start reduces
stresses and surge currents during start-up. Soft-stop reduces
electrical stresses during shutdown when synchronous rectifiers
(SRs) are used and prevents polarity reversal of the converter
output. Soft-stop may be inhibited with MODE for applications
not using SRs.
• Overlapping and non-overlapping output configuration for both
n-channel and p-channel clamp configurations
• Peak and average overcurrent protection
• Internal thermal protection
• Minimum duty cycle clamp
• Input voltage dependent maximum duty cycle clamp
Supply Currents
The total supply current, IDD, will be dependent on the load applied
to outputs OUTM and OUTAC. Total IDD current is the sum of the
quiescent current and the average output current. Knowing the
operating frequency (FSW) and the output loading capacitance
charge (Q) per output, the average output current can be calculated
from Equation 1:
I OUT = 2 • Q • F SW
(EQ. 1)
Oscillator
The ISL6726 oscillator has a programmable frequency range to
2MHz, and can be set with one resistor and one capacitor. The
use of two timing elements, RTC, and CT allow great flexibility
and precision when setting the oscillator frequency.
The switching period is the sum of the timing capacitor charge
and discharge durations. The charge and discharge duration is
determined by RTC and CT.
t C ≈ 0.5 • RTC • CT
t D ≈ 0.125 • RTC • CT
1
t SW = T C + T D = ----------F SW
(EQ. 2)
S
The soft-start feature clamps the duty cycle for the duration of
soft-start. The duty cycle is initially forced to zero and allowed to
linearly increase until the control loop takes control. At the
beginning of a soft-start cycle, the SS capacitor is discharged. If
ENABLE is open and there is no UVLO fault on VDD, a current
source charges the soft-start capacitor. Taking into account the
internal gains and offsets of VERR and SS, soft-start limits the
peak current amplitude as long as it remains below VERR. As
the SS voltage increases, the peak current amplitude is allowed
to increase. The output pulse width increases accordingly, until
the SS voltage exceeds VERR and the control loop takes over.
The SS voltage will continue to increase until it reaches its clamp
voltage of 4.6V even though soft-start is actually finished when
the control loop takes over. The duty cycle increases from zero to
its steady state operating point during the soft-start period. The
soft-start waveform is shown in Figure 7 for the non-overlap
configuration, appropriate for the active clamp forward with a
n-channel clamp FET or the asymmetric half-bridge topology. For
the active clamp topology using a p-channel clamp FET, the
overlap configuration is required and the OUTAC waveform
shown in Figure 7 would be inverted. The non-overlap
configuration is shown for clarity.
(EQ. 3)
S
Soft-start begins
VSS=ISS*t/CSS
1.2(VERR-VOFFSET)
(EQ. 4)
S
0.27V
Soft-start ends
0.0V
Where tC and tD are the charge and discharge times,
respectively, tSW is the oscillator free running period, and FSW is
the oscillator frequency. The actual times will be slightly longer
than calculated due to internal propagation delays of
approximately 10ns/transition. This delay adds directly to the
switching duration, but also causes overshoot of the timing
capacitor peak and valley voltage thresholds, effectively
increasing the peak-to-peak voltage on the timing capacitor.
Additionally, if very low charge and discharge currents are used,
there will be increased error due to the input impedance of the
CT pin.
12
OUTM
OUTAC
CS
FIGURE 7. SOFT-START FUNCTION (IDELAY POSITIVE)
FN7654.0
January 31, 2011
ISL6726
1.2(VERR-VOFFSET)
VSS=5V-ISS*t/CSS
Soft-stop ends
Soft-stop begins
0.27V
0.0V
avalanche when the duty cycle becomes non-zero. When the
forward SR turns on, the inductor current will reflect to the
primary and stress the components there as well. With the
minimum duty cycle clamp feature, the forward rectifier turns on
for ~300ns each cycle and prevents the large negative current in
the output inductor.
Gate Drive
OUTM
OUTAC
CS
FIGURE 8. SOFT-STOP FUNCTION (IDELAY POSITIVE)
The soft-stop function is enabled when MODE=0. The ISL6726
enables a soft-stop when UV falls below 1V or when ENABLE is
pulled low (disable), causing a controlled discharge of the SS
capacitor at the rate equal to and opposite of soft-start. Soft-stop
will not occur for a UVLO fault on VDD regardless of the MODE
setting. Soft-stop continues until the SS pin voltage drops below
~0.25V, even if the fault condition is removed before the
threshold is reached.
Using soft-stop forces an orderly shutdown of a converter that
uses synchronous rectification (SR). It prevents the output
voltage from going negative by controlling the rate at which the
output voltage is discharged through the output inductor. It also
prevents the SRs from being avalanched if SR operation is
stopped when the inductor current is negative.
If a self-driven SR method is used, the behavior during turn-off is
improved as well. During soft-stop, the forward rectifier pulse
width is slowly decreased to its minimum while the
free-wheeling rectifier pulse width is slowly increased to its
maximum. The active clamp capacitor voltage, VIN/(1-D),
approaches VIN as the duty cycle approaches zero. The
freewheeling rectifier gate voltage is VIN D/n(1-D), where n is the
transformer turns ratio Np/Ns, and decreases with decreasing
duty cycle. At some point the voltage applied to the gate is
insufficient to turn on the SR FET and negative inductor current
is prevented.
The ISL6726 has two outputs, OUTM and OUTAC. OUTM is
capable of sourcing 1A and sinking 1.5A peak current, and
OUTAC is capable of sourcing 0.5A and sinking 0.75A peak
current. OUTAC is configured using the DELAY input for either
overlap or non-overlap phasing relative to OUTM. When
configured for non-overlap phasing, OUTAC operates at 1-D with
deadtime, where D is the duty cycle of OUTM. This configuration
is useful for the n-channel active clamp and asymmetric halfbridge topologies. When configured for overlap phasing, OUTAC
has symmetric rising edge advance and falling edge delays
relative to OUTM. This configuration is useful for the p-channel
active clamp topology.
Two typical active clamp converter configurations are shown in
Figures 9 and 10, with overlap or non-overlap delay time
accurately set by a programming resistor. The rising edge
overlap and the falling edge overlap time (or rising edge
deadtime, and falling edge deadtime) are equal and
independent of the operating frequency or duty cycle.
To limit the peak current through the IC, an external resistor may
be placed in series between an output and the gate of the
MOSFET. The resistor also dampens any oscillation caused by
the resonant tank of the parasitic inductance of the PWB traces
and the FET gate input capacitance. The overlap/non-overlap
delay between OUTAC and OUTM prevents simultaneous
conduction of the main and clamp switches in an active clamp
converter, or the upper and lower switches in an asymmetric
half- bridge converter.
Table 1 shows the combinations of the settings with the
corresponding features for different topologies.
Tx
A hard-stop with self-driven SRs results in oscillation of the SRs
because the output voltage can provide gate voltage through the
output inductor and secondary winding.
In addition to soft-stop when MODE=0, the minimum pulse width
of OUTM is clamped to ~300ns independent of the PWM
modulator. Higher duty cycles are obviously allowed depending on
the operating conditions, but shorter duty cycles are not. In SR
applications, this feature prevents excessive negative output
inductor current if the output should experience a large and
sudden reduction in load, such as occurs during a 100% to 0%
load transient. A sudden load dump can cause the control loop
error voltage to drop sufficiently to command 0% duty cycle. This
sets the forward rectifier to 0% duty cycle and the free-wheeling
rectifier to 100% duty cycle. This condition allows the inductor
current to ramp to a large negative amplitude until the duty cycle
again becomes non-zero. Due to the normal deadtime allowed
for proper switching of the SRs, the forward rectifier will
13
+VOUT
Lm
+
Vdc
Minimum Duty Cycle Clamp
VX
VY
OUTM
OUTM
OUTAC
Vout = Vin*D*Ns/Np
Td = K1*Rdelay
Td = K1*Rdelay
D
OUTAC
FIGURE 9. OUTPUT TIMING DIAGRAM FOR P-CHANNEL ACTIVE
CLAMP
FN7654.0
January 31, 2011
ISL6726
TABLE 1. MODE AND DELAY SETTINGS FOR TYPICAL TOPOLOGIES
TOPOLOGY
MODE
DELAY
N-FET Active Clamp with Diode Rectification
HIGH
R to VREF
Non-OverLap
Disabled
Disabled
P-FET Active Clamp with Diode Rectification
HIGH
R to GND
OverLap
Disabled
Disabled
N-FET Active Clamp with SR Rectification
LOW
R to VREF
Non-OverLap
Enabled
Enabled
P-FET Active Clamp with SR Rectification
LOW
R to GND
OverLap
Enabled
Enabled
Standard Forward with Diode Rectification
HIGH
= 0V,
= VREF
OverLap,
Non-Overlap
Disabled
Disabled
Asymmetric Half-Bridge
LOW
R to VREF
Non-OverLap
Enabled
Enabled
Tx
SOFT-STOP
MINIMUM D CLAMP
Overcurrent Operation
VX
+VOUT
Lm
+
Vdc
OUTM
The ISL6726 has two mechanisms for current limit. The peak
current limit function provides cycle-by-cycle overcurrent
protection. The protection threshold is set by a voltage applied to
ISET. If the peak current at CS exceeds ISET, the OUTM pulse is
terminated for the remainder of the switching cycle.
Peak current limit has some shortcomings that discourage its
use as the only current limit mechanism. First, there is the slope
compensation ramp that adds to the current feedback signal. Its
contribution to the CS signal varies with duty cycle, and at high
duty cycles it has a larger contribution than at lower duty cycles.
As an overload condition causes the duty cycle to decrease, the
portion of the current feedback contributed by the slope
compensation decreases and the amount contributed by the
current feedback increases. The result is that the maximum
output current will increase as the output voltage decreases.
VY
Vout = Vin*D*Ns/Np
OUTM
Td = K2*Rdelay
PHASING
Td = K2*Rdelay
OUTAC
IL
IS2
Another phenomenon occurs when the duty cycle is reduced to
the minimum pulse width the IC controller is capable of
producing. If the output voltage is reduced below the value
corresponding to this duty cycle, current tail-out occurs. There is
a certain amount of energy delivered to the output on each
switching cycle that must correspond to voltage and current at
the load. If the voltage is very low due to a shorted output, large
currents can result.
IS1
IS
IMAG
FIGURE 10. OUTPUT TIMING DIAGRAM FOR N-CHANNEL ACTIVE
CLAMP
Overlap phasing results when a resistor is connected between
DELAY and GND. Non-overlap phasing results when a resistor is
connected between DELAY and VREF. The resistor value
determines the magnitude of the delay. The delay feature may
be disabled by connecting DELAY directly to GND or VREF,
depending on which configuration is desired, overlap or
non-overlap. The non-overlap time in the overlap mode can be
calculated using Equation 5.
ns
t DELAY = 1.83 -------- ⋅ R DELAY ( kΩ ) + 13ns
kΩ
(EQ. 5)
The deadtime in non-overlapping mode can be calculated using
Equation 6.
ns
t DELAY = 1.79 -------- ⋅ R DELAY ( kΩ ) + 9ns
kΩ
See Figure 3 for typical DELAY gain curves.
14
(EQ. 6)
Some controllers solve the problem by allowing the converter to
cycle on and off (hic-cup operation) to lower the average short
circuit current. This works acceptably for some applications, but
not when redundancy or parallel operation is required. Such
behavior can prevent a successful fault recovery when the short
is removed. The paralleled or redundant units will not hic-cup in
unison, and each will experience an overload condition each
time a restart is attempted.
An ideal current limiting method requires a constant value
regardless of the output voltage, the so-called “brick-wall”
current limit. The output current remains constant from current
limit inception to a short circuit. The ISL6726 provides this
behavior with the average current limit function.
The average current limit feature uses a patented circuit that
samples the current feedback signal and creates a signal
proportional to the average value of the output inductor current.
The signal, analogous to the voltage feedback signal of voltage
control loop, becomes the feedback signal for the current error
amplifier and produces a current error signal. The voltage
feedback and current feedback share a common control node
FN7654.0
January 31, 2011
ISL6726
(VERR) used by the pulse width modulator. Whichever error
signal, voltage or current, that commands the lower duty cycle is
in control. If the average current is lower than the average
current limit threshold, the current error amplifier has no impact
on VERR and the voltage loop is in control. If the average current
limit threshold is exceeded, however, the current error amplifier
will lower VERR to regulate the output current. The voltage loop
loses control as it must increase the duty cycle to maintain the
output voltage in regulation.
inductor current becomes discontinuous (DCM operation), IOUT
represents one half of the peak inductor current rather than the
average current. This occurs because the sample and hold
circuitry is active only during the on time of the switching cycle
and cannot determine when the inductor current becomes
discontinuous. It is unable to detect when the inductor current
reaches zero during the off time. This behavior does not affect
the average current limit function, but does have an impact if
IOUT is used for current monitoring functions.
After a 100ns leading edge blanking (LEB) delay, the current
sense signal is sampled for the duration of the on time, the
average current is determined, and the result is amplified by 4x
and output to the IOUT pin at the termination of the OUTM pulse.
Due to the sampling algorithm used, if an RC filter is placed on
the CS input, its time constant should not exceed ~30ns or error
may be introduced on IOUT.
IOUT may be used with the available error amplifier (EA) of the
ISL6726 as shown in Figure 13. The error amplifier is typically
configured as an integrator. As shown in Figure 13, IOUT is
attenuated by resistors R1 and R2 so that the average current
limit threshold can be set independently of the peak current
limit threshold. The integrator bandwidth is determined by R and
C. The current error amplifier is similar to the voltage EA found in
most PWM controllers, except it cannot source current. VERR
requires an external pull-up resistor.
20
1
2
RPULL-UP
ISL6726
19
3
18
4
VREF 17
5
16
6
+
7 ISET
8 VERR
14
13
12
9 FB
10 CS
15
S&H 4x
IOUT 11
C
Channel 1: OUTM
Channel 2: IOUT
Channel 4: CS
FIGURE 11. CS INPUT vs IOUT
R
R1
R2
FIGURE 13. AVERAGE CURRENT CONFIGURATION
Channel 1: OUTM
Channel 2: IOUT
Channel 4: CS
FIGURE 12. DYNAMIC BEHAVIOR OF CS AND IOUT
The average current signal on IOUT produces an accurate
representation of the output current provided the converter
operates in continuous conduction mode (CCM). Once the
15
The IEA is configured as an integrating (Type I) amplifier using
ISET as the reference. The voltage applied at FB is integrated
against the ISET reference. The resulting signal, VERR, is applied
to the PWM comparator where it is compared to the current
signal CS. If FB is less than ISET, the IEA will be open loop (can’t
source current), VERR will be at a level determined by the
voltage loop, and the duty cycle is unaffected. As the output load
increases, IOUT will increase, and the voltage applied to FB will
increase until it reaches ISET. At this point the IEA will control
VERR as required to maintain the output current at the level that
corresponds to the ISET reference. When the output current
again drops below the average current limit threshold, the IEA
returns to an open loop condition, and the duty cycle is again
controlled by the voltage loop. The average current control loop
behaves much the same as the voltage control loop found in
typical power supplies except it regulates current rather than
voltage.
FN7654.0
January 31, 2011
ISL6726
The average current loop bandwidth is normally set much lower
than the switching frequency, typically less than 5kHz and
maybe as slow as a few hundred hertz, depending on the
application requirements. This is especially useful if the
application experiences large surges. The average current loop
can be set to the steady state overcurrent threshold and have a
time response that is longer than the required transient.
Under some conditions it will be necessary to clamp the FB pin
with a Schottky diode to signal ground. If the voltage loop causes
a fast decreasing transient on VERR, the feedback capacitor
between VERR and FB can cause a negative voltage on FB and
violate the absolute maximum rating.
DUTY CYCLE
MAXIMUM DUTY CYCLE CLAMP
DUTY CYCLE CLAMPED BY UV
UV = k*Vin
1V 1.2V
MINIMUM DUTY CYCLE CLAMP
MODE = GND
Duty Cycle Clamp
This condition occurs when the input voltage is rapidly
decreased, or when the output load is rapidly increased. Both of
these conditions result in a rapidly increasing duty cycle. If the
duty cycle can increase more quickly than the clamp capacitor
voltage can respond, the core will not be properly reset. One or
the other of these transients can be mitigated by the sizing of
the clamp capacitor value. Smaller values favor input voltage
transient behavior whereas larger values favor load transient
behavior. Most designs favor load transient behavior. In either
case, the maximum duty cycle clamp prevents large duty cycle
increases and limits transformer flux density and FET voltage
stress.
The main output PWM is controlled by the current and voltage
feedback signals. When the feedback loop demands maximum
duty cycle, the duty cycle is limited by the lesser of the input
voltage-dependent duty cycle limiter or the maximum duty cycle
limit of the controller, which is 80% by design.
5V
FIGURE 14. DUTY CYCLE CLAMP
It is very important to control the maximum duty cycle of an
active clamp reset forward converter. The clamp capacitor and
drain-source voltage of the main switch is related to the duty
cycle D by Equation 7. If the duty cycle is not clamped, the FET
drain-source voltage can become quite high and overstress the
FET.
Vin
Vds = -----------------(1 – D)
During transients the situation is particularly bad, not only
because of the voltage stress on the power FETs, but also
because the clamp capacitor voltage is not at the steady state
voltage required to properly reset the transformer. The active
clamp forward topology is also know as the optimum reset
topology because the steady state clamp capacitor voltage is
exactly the value required to reset the core during the off time.
However, it can take many switching cycles before the clamp
capacitor voltage reaches a new steady state value after a
change in operating point. If the clamp capacitor voltage is lower
than required, the transformer core is not reset completely and
can lead to transformer saturation after a few switching cycles.
The input voltage dependent duty cycle limit is inversely
proportional to the input voltage, as shown in Figure 15. The
voltage applied to UV determines the amplitude of the CT
sawtooth waveform, where CTPEAK = 0.8 + 0.8 * UV. Since the
UV turn-on threshold is 1.00V, the minimum amplitude of CT is
1.60V. At UV = 4.00V, the amplitude of CT is 4.00V. The
maximum duty cycle clamp is determined by the voltage applied
to DCLIM and the amplitude of CT. If DCLIM is set to 1.60V or
greater, the maximum duty cycle is 80%. The maximum duty
cycle as a function of UV and DCLIM is:
(EQ. 7)
D MAX
Without the input voltage dependent maximum duty cycle clamp
it is possible to have both high input voltage and high duty cycle
during input voltage or load transients. The duty cycle clamp
reduces the maximum duty cycle as the input voltage increases.
Whereas the maximum duty cycle at minimum input voltage is
large, it is not necessary, nor is it advantageous, to have the
same maximum duty cycle at maximum input voltage. The duty
cycle clamp allows the designer to provide a constant margin of
duty cycle headroom above the steady state operating point to
allow for adequate dynamic response without allowing so much
headroom that it can result in excessive voltage stress on the
FET.
16
= 0.8, DCLIM > 0.8 ⋅ UV + 0.8
DCLIM – 0.8
= ---------------------------------, DCLIM ≤ 0.8 ⋅ UV + 0.8
UV
(EQ. 8)
For most applications the maximum duty cycle will be set for the
minimum operating input voltage, and for which UV is set to
1.00V.
Consequently, the actual duty cycle of the main output, OUTM, is
the minimum of the current mode PWM comparator, the
maximum 80% duty cycle clamp of the controller, or the input
voltage dependent duty cycle clamp.
FN7654.0
January 31, 2011
ISL6726
Vpk=0.8UV+0.8
1.6UV/(CTC*RTC)
UV
DCLIM
T1
0.8V
T2
Intrinsic
Maximum D
T1=1/2CTC*RTC
T2=1/8CTC*RTC
Maximum D
Set by DCLIM
TDC= VDC/Vpk*T1
CS
FIGURE 15. MAXIMUM DUTY CYCLE CLAMP USING DCLIM
Synchronization
ISL6726 provides a single I/O pin synchronization function that
allows synchronization to an external clock or to self-synchronize
to another unit at ~180 degrees out-of-phase for interleaved
applications. When using an external clock, the clock pulse width
must be a minimum of 100ns. The clock frequency must be
higher than the free running frequency of the oscillator.
Multiple units may be synchronized together simply by
connecting the SYNC pins together as shown in Figure 16. In this
configuration all of the devices will synchronize out-of-phase with
the master. The master is usually the unit with the fastest freerunning oscillator, but may not be due to intentional hysteresis
within the arbitration circuitry. Synchronization occurs on the
leading edge of the SYNC signal. However, no unit will accept a
SYNC pulse while its oscillator ramp voltage is less than 3/8 of
the timing capacitor voltage peak voltage. This prevents short
cycling of the period.
If the SYNC pins of multiple devices are connected together, the
first SYNC signal that asserts will reset the oscillator RAMP of all
other devices. Further arbitration may occur if there is a higher
frequency unit present. All slave controllers will operate out-ofphase with the master. Multiple devices may be synchronized in
this fashion, but the number will depend on the distance and
capacitance of the SYNC signal path. Care should be taken to
ensure the ground potential difference between devices is
17
minimized. In most cases an external clock is used to
synchronize more than two units.
1 SYNC
20
1 SYNC
20
2
19
2
19
18
3
4
17
4
17
5
GND 16
5
GND 16
3
CT1 6 CT
ISL6726
OUTM 15 OUTM1
CT2 6 CT
ISL6726
18
OUTM 15 OUTM2
7
14
7
14
8
13
8
13
9
12
9
12
10
11
10
11
OUTM1
OUTM2
CT1
CT2
FIGURE 16. SYNCHRONIZING TWO UNITS
FN7654.0
January 31, 2011
ISL6726
Configuring UV
The UV input is used for input source undervoltage lockout. If the
UV node voltage falls below 1.00V, a UV shutdown fault occurs.
This may be caused by low source voltage or by intentional
grounding of the pin to disable the outputs. There is a nominal
10µA switched current source used to create hysteresis. The
current source is active only during an UV/Inhibit fault; otherwise,
it is inactive and does not affect the UV threshold voltage. The
magnitude of the hysteresis is a function of the external resistor
divider impedance. If the resistor divider impedance results in too
little hysteresis, a series resistor between the UV pin and the
divider may be used to increase the hysteresis. A soft-start cycle
begins when the UV/Inhibit fault clears. The voltage hysteresis
created by the switched current source and the external
impedance is generally small due to the large resistor divider ratio
required to scale the input voltage down to the UV threshold level.
VIN
R1
1.00V
+
-
R3
10µA
R2
ON
must be placed in parallel with R2, and the capacitor and R3
must be physically close to the UV pin.
UV may also be used as an inhibit signal by externally pulling it
below the 1V threshold. However, caution must be exercised as
the maximum duty cycle limit controlled by DCLIM will be
defeated. The peak amplitude of CT will be reduced to ~1.6V
when UV decreases below the 1V turn-off threshold, and the
maximum duty cycle allowed will increase to 80%.
Slope Compensation
For applications where the maximum duty cycle is less than 50%,
slope compensation may be used to improve noise immunity,
particularly at lighter loads. The amount of slope compensation
required for noise immunity is determined empirically, but is
generally about 10% of the full scale current feedback signal.
For applications where the duty cycle is greater than 50%, slope
compensation is required to prevent instability, referred to as
sub-harmonic oscillation. Slope compensation is a technique in
which the current feedback signal is modified by adding slope,
that is, adding a linearly increasing voltage as a function of time.
The minimum amount of slope compensation required
corresponds to 1/2 the inductor downslope, as it would appear
referred to the CS input. See Figure 18. More may be added, but
increasing the slope compensation arbitrarily results in a control
loop that transitions into voltage mode as the slope
compensation begins to dominate the current feedback signal.
The minimum amount of capacitance to place at the SLOPE pin is:
t ON
C SLOPE = 18 • -------------------V SLOPE
FIGURE 17. UV HYSTERESIS
Referring to Figure 17, as VIN decreases to a UV condition, the
threshold level is:
R1 + R2
V IN ( DOWN ) = ---------------------R2
V
(EQ. 9)
Where tON is the maximum ON time in seconds, and VSLOPE is the
amount of voltage to be added as slope compensation to the
current feedback signal at the CS pin. In general, the amount of
slope compensation added is 2 to 3 times the minimum required.
–5
R1 + R2
• 〈 R1 + R3 • ⎛ ----------------------⎞ 〉
⎝ R2 ⎠
V
(EQ. 10)
Setting R3 equal to zero results in the minimum hysteresis, and
yields:
ΔV = 10
–5
• R1
V
CURRENT SENSE SIGNAL
(EQ. 11)
As VIN increases from a UV condition, the threshold level is:
V IN ( UP ) = V IN ( DOWN ) + ΔV
ISENSE SIGNAL (V)
The hysteresis voltage, ΔV, is:
ΔV = 10
V
(EQ. 12)
Although the current hysteresis provides great flexibility in setting
the magnitude of the hysteresis voltage, it is susceptible to noise
on the signal. If the hysteresis was implemented as a fixed
voltage instead, the signal could be filtered with a small
capacitor placed between the UV pin and signal ground. This
technique does not work well when the hysteresis is a current
source because a current source takes time to charge the filter
capacitor. There is no instantaneous change in the threshold
level thereby rendering the current hysteresis ineffective. To
remedy the situation the filter capacitor must be separated from
the UV pin by a resistor. Referring to Figure 17, the filter capacitor
18
(EQ. 13)
μF
DOWNSLOPE
TIME
FIGURE 18. DOWNSLOPE
It should be noted that the power transformer magnetizing
inductance contributes to slope compensation and should be
considered when determining the amount of slope
compensation required.
Example:
Assume the inductor current signal presented at the CS pin
decreases 125mV during the Off period, and:
Switching Frequency, Fsw = 250kHz
Duty Cycle, D = 60%
tON = D/Fsw = 0.6/250E3 = 2.4µs
tOFF = (1 - D)/Fsw = 1.6µs
FN7654.0
January 31, 2011
ISL6726
Determine the downslope:
Ground Plane Requirements
Downslope = 0.125V/1.6µs = 78mV/µs. Now determine the
amount of voltage that must be added to the current sense
signal by the end of the On time.
Careful layout is essential for satisfactory operation of the device.
A good ground plane must be employed. Use a ground layer if
possible. The power ground should be connected to the control
ground at one point. VDD should be bypassed directly to GND with
good high frequency capacitance, such as a ceramic capacitor. A
small ceramic capacitor is also recommended for DCLIM.
1
V SLOPE = --- • 0.078 • 2.4 = 94mV
2
(EQ. 14)
Therefore,
–6
C SLOPE ( MIN ) = 18 ×10
–6
2.4 ×10
• ------------------------ ≈ 470pF
0.094
(EQ. 15)
An appropriate slope compensation capacitance for this example
would be 1/2 to 1/3 the calculated value, or between 150pF and
220pF.
Using MODE
The MODE pin configures the IC for standard or synchronous
rectification compatibility. If MODE is connected to VREF, standard
rectification compatibility is selected. Soft-stop and the minimum
duty cycle clamp are disabled. If MODE is connected to GND,
synchronous rectification compatibility is selected, and soft-stop
and the minimum duty cycle clamp are enabled.
Thermal Protection
An internal temperature sensor protects the device should the
junction temperature exceed +145°C. There is approximately
+15°C of hysteresis.
The OUTM, and OUTAC of ISL6726 are very fast signals, and
should have very short direct paths to the power MOSFETs in
order to minimize inductance in the PC board traces. The return
path should be as short as possible. The components at the Pins
of SS, DCLIM, UV, DELAY, CT, and RTC should be as physically
close as possible to the IC. Proximity to high di/dt loops and high
dv/dt nodes should be avoided.
The CS signal requires proper filtering and the PWB layout is
critical for normal operation of the current related functions. A
RC filter may be required. The time constant should be no greater
than 25ns to prevent incorrect average current information. If a
current sense transformer is used, both leads of the secondary
winding should be routed to the CS filter components and to the
IC pins. The transformer return should be connected via a
dedicated PC board trace to the GND pin rather than through the
ground plane.
If a current sense resistor in series with the switching FET source
is used, a low inductance resistor is recommended. The low level
signals must avoid the high current path.
Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make
sure you have the latest revision.
DATE
REVISION
January 31, 2011
FN7654.0
CHANGE
Initial Release.
Products
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19
FN7654.0
January 31, 2011
ISL6726
Package Outline Drawing
M20.15
20 LEAD QUARTER SIZE OUTLINE PLASTIC PACKAGE (QSOP)
Rev 2, 1/11
20
INDEX
AREA
1
2
0.244 (6.19)
0.157 (3.98) 0.228 (5.80)
0.150 (3.81)
4
3
GAUGE
PLANE
TOP VIEW
6
0.050 (1.27)
0.25
0.010
SEATING PLANE
3
0.069 (1.75)
0.053 (1.35)
0.344 (8.74)
0.337 (8.56)
0.016 (0.41)
0.0196 (0.49)
5
0.0099 (0.26)
8°
0°
0.012 (0.30)
0.008 (0.20)
0.025
(0.635 BSC)
8
0.010 (0.25)
0.004 (0.10)
0.061 MAX (1.54 MIL)
SIDE VIEW
0.010 (0.25)
0.007 (0.18)
DETAIL "X"
NOTES:
0.015 (0.38) x 20
0.025 (0.64) x 18
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of
Publication Number 95.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
20
0.060 (1.52) x 20
3. Dimension does not include mold flash, protrusions or gate burrs.
Mold flash, protrusion and gate burrs shall not exceed 0.15mm
(0.006 inch) per side.
4. Dimension does not include interlead flash or protrusions. Interlead
flash and protrusions shall not exceed 0.25mm (0.010 inch) per
side.
0.220(5.59)
5. The chamfer on the body is optional. If it is not present, a visual index
feature must be located within the crosshatched area.
6. Length of terminal for soldering to a substrate.
7. Terminal numbers are shown for reference only.
1
2
3
TYPICAL RECOMMENDED LAND PATTERN
8. Dimension does not include dambar protrusion. Allowable
dambar protrusion shall be 0.10mm (0.004 inch) total in excess of
dimension at maximum material condition.
9. Controlling dimension: INCHES. Converted millimeter dimensions
are not necessarily exact.
20
FN7654.0
January 31, 2011
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