Micrel MIC9131BM High-voltage, high-speed telecom dc-to-dc controller Datasheet

MIC9131
Micrel
MIC9131
High-Voltage, High-Speed Telecom DC-to-DC Controller
Final Information
General Description
Features
The MIC9131 is a current-mode PWM controller that efficiently converts –48V telecom voltages to logic levels. The
MIC9131 features a high voltage start-up circuit that allows
the device to be connected to input voltages as high as 180V.
The high input voltage capability protects the MIC9131 from
line transients that are common in telecom systems. The
start-up circuitry also saves valuable board space and simplifies designs by integrating several external components.
The MIC9131 is capable of high speed operation. Typically
the MIC9131 can control a sub-25ns pulse width on the gate
out pin. Its internal oscillator can operate over 2.5MHz, with
even higher frequencies available through synchronisation.
The high speed operation of the MIC9131 is made safe by the
very fast, 34ns response from current sense to output,
minimizing power dissipation in a fault condition.
The MIC9131 allows for the designs of high efficiency power
supplies. It can achieve efficiencies over 90% at high output
currents. Its low 1.3mA quiescent current allows high efficiency even at light loads.
The MIC9131 has a maximum duty cycle of 75%. For designs
requiring a maximum duty cycle of 50%, refer to the MIC9130.
The MIC9131 is available in a 16-pin SOP and 16-pin QSOP
package options. The junction temperature range is from
–40°C to +125°C.
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
Input voltages up to 180V
Internal oscillator capable of > 2.5MHz operation
Accurate 75% maximum duty cycle
Synchronisation capability to 6MHz
Current sense delay of 34ns
Minimum pulse width of <25ns
90% efficiency
1.3mA quiescent current
1µA shutdown current
Soft-start
Resistor programmable current sense threshold
Selectable soft-start retry
4Ω sink, 12Ω source output driver
Programmable under-voltage lockout
Constant-frequency PWM current-mode control
16-pin SOIC and 16-pin QSOP
Applications
•
•
•
•
•
•
Telecom power supplies
Line cards
ISDN network terminators
Micro- and pico-cell base stations
Low power (< 100W) dc-dc converters
DSL line cards
Ordering Information
Part Number
Max. Duty Cycle
Junction Temp. Range
Package
MIC9131BM
75%
–40°C to +125°C
16-Pin SOP
MIC9131BQS
75%
–40°C to +125°C
16-Pin QSOP
Typical Application
10Ω
12V
1N5818
10µF
25V
4:1
VIN
36V to 72V
VOUT
3.3V @ 20A
6:1
0.1µF
L = 530µH
1MΩ
1.5µH
Si4884DY
(x2)
B320A
100µF
37.4kΩ
7
10
FB
OUT
10kΩ
6
3
12
8
562kΩ
9
0.1µF
Si4884DY
(x2)
EN
VCC
2
LINE
1
UVLO
13
COMP
RBIAS
ISNS
SS
16
IRFS3IN20D
SLOPE
COMPENSATION
MIC9131
14
8.2kΩ
CPWR
VBIAS
OSC SYNC AGND PGND
4
5
11
0.1Ω
1W
15
2.61kΩ
10nF
270pF
OPTO
FEEDBACK
90% Efficient Telecommunications Power Supply
Micrel, Inc. • 1849 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 944-0970 • http://www.micrel.com
July 2001
1
MIC9131
MIC9131
Micrel
Pin Configuration
LINE 1
16 OUT
VCC 2
15 PGND
14 ISNS
RBIAS 3
13 UVLO
OSC 4
SYNC 5
12 SS
COMP 6
11 AGND
10 EN
FB 7
9 VBIAS
CPWR 8
16-Pin SOP (M)
16-Pin QSOP (QS)
Pin Description
Pin Number
Pin Name
1
LINE
Line (Input): 180Vdc maximum supply input. May be floated if unused.
2
VCC
Supply (Input): MIC9131 internal supply input.
3
RBIAS
4
OSC
5
SYNC
Synchronization (Input): External oscillator input for slave operation of
controller. See OSC. Do not float.
6
COMP
Compensation (External Components): Error amplifier output for external
compensation network connection.
7
FB
8
CPWR
Current Limit Selection (Input): When CPWR is high, an over-current
condition at the ISNS input will terminate the gate drive and reset the softstart latch. If the CPWR pin is low, an over-current condition at the ISNS
input will terminate the gate drive signal, but will not cause a reset of the
soft-start circuit.
9
VBIAS
Reference (Output): Internal 5V supply. Will source 5mA maximum.
10
EN
11
AGND
12
SS
Soft-Start (External Components): Connect external capacitor to slowly ramp
up duty cycle during startup and over-current conditions.
13
UVLO
Undervoltage Lockout (External Components): Connect to unbiased resistive
divider network to set controller’s minimum operating voltage. Connect to
VBIAS if not needed.
14
ISNS
Current Sense (Input): Connect between external switching MOSFET source
and switch current sense resistor.
15
PGND
Power Ground (Return)
16
OUT
MIC9131
Pin Function
Bias Resistor (External Component): Connect 562KΩ to ground.
Oscillator RC Network (External Components): Connect external resistorcapacitor network to set oscillator frequency.
Feedback (Input): Error amplifier inverting input.
Enable (Input): Logic level enable/shutdown input; logic high = enabled (on),
logic low = shutdown (off).
Analog Ground (Return)
Switch Drive Output (Output): Connect to gate of external switching
MOSFET.
2
July 2001
MIC9131
Micrel
Absolute Maximum Ratings (Note 1)
Operating Ratings (Note 2)
Line Input Voltage (VLINE) ......................................... +190V
VCC Input Voltage (VCC) ............................................. +19V
Current Sense Input Voltage (VISNS) ............. –0.3 to +5.3V
Enable Voltage (VEN) ............................ –0.3 to VCC + 0.3V
Feedback Input Voltage (VFB) ........................ –0.3 to +5.3V
Sync Input Voltage (VSYNC) ........................... –0.3 to +5.3V
Soft-Start Voltage (VSS) ................................. –0.3 to +5.3V
UVLO Voltage (VUVLO) .................................. –0.3 to +5.3V
Storage Temperature (TS) ....................... –65°C to +150°C
Power Dissipation (PD) .......................................................
16-pin SOP .................................. 400mW @ TA = +85°C
16-pin QSOP ............................... 245mW @ TA = +85°C
ESD Rating, Note 3
Line Input Voltage (VLINE) ................ VCC to +180V, Note 4
VCC Input Voltage (VCC) ................................. +9V to +18V
Junction Temperature Range (TJ) ........... –40°C to +125°C
Package Thermal Resistance
16-pin SOP (θJA) ............................................... 100°C/W
16-pin QSOP (θJA) ............................................ 163°C/W
Electrical Characteristics
TA = 25°C, VLINE = 48V, VCC = 10V, Rt = 9.47KΩ, Ct = 470pF, RBIAS = 562kΩ, VEN = 10V, VISNS = 0V, VUVLO = 2V, VSYNC = 0V, unless
otherwise noted. Bold values indicate –40 °C ≤TJ ≤ +125°C.
Parameter
Condition
Min
Typ
Max
Units
IVBIAS = 0mA; VOSC = 0V (Oscillator OFF)
4.7
4.85
5.0
V
5.1
V
Bias Regulator
Output Voltage
4.6
Line Regulation
9V ≤ VCC ≤18V, IVBIAS = 0mA; VOSC = 0V
24
40
mV
Load Regulation
0mA ≤ IVBIAS ≤ 5mA; VOSC = 0V
5
30
mV
200
220
kHz
Oscillator Section
Initial Accuracy (fOSC)
Rt = 9.47KΩ, Ct = 470pF
180
Oscillator Output Frequency
Maximum Duty Cycle
fOSC/4
kHz
75
%
Voltage Stability (∆f/f)
9V ≤ VCC ≤18V
2.5
%
Temperature Stability
–40°C ≤ TJ ≤ 125°C
100
ppm/°C
Maximum Sync Frequency
Note 5
6
MHz
Sync Threshold Level
2.5
V
Sync Hysteresis
0.7
V
Sync Minimum Pulse Width
50
ns
Error Amp Section
FB Voltage
VCOMP = VFB
2.475
2.45
2.5
2.525
2.55
V
Open Loop Voltage Gain, AVOL
90
dB
Unity Gain Bandwidth
4
MHz
60
dB
PSRR
9V ≤ VCC ≤ 18V
COMP Sink Current
VFB = 2.7V; VCOMP = 5V
80
100
µA
COMP Source Current
VFB = 2.3V; VCOMP = 0V
1
2.5
mA
VCOMP Low
VFB = 2.7V; ICOMP = –50µA
VCOMP High
VFB = 2.3V; ICOMP = +500µA
Input Bias Current (IFB)
Slew Rate
July 2001
115
3.5
300
mV
4
V
VFB = VCOMP
160
nA
SINK
1.5
V/µs
SOURCE
1.5
V/µs
3
MIC9131
MIC9131
Parameter
Micrel
Condition
Min
Typ
Max
Units
0.1
10
µA
Preregulator
Input Leakage Current
VLINE = 180V, VCC = 10V
VCC Gate Lockout (VGLO(ON))
VLINE = 48V
7.2
7.5
V
VCC Gate Lockout Hysteresis
(∆VGLO)
VLINE = 48V
700
800
mV
VCC Pre-Regulator Off (VPR(OFF))
VLINE = 48V
VGLO(ON)
V
7.7
VCC Pre-Regulator Hysteresis
(∆VPR)
VLINE = 48V
Start-up Current
VLINE = 48V, VCC = 7.5V, Note 4
+0.5V
500
700
mV
9
12
mA
Supply
Supply Current, IVCC
Pin 16 (OUT) = OPEN
Enable Input Current
VEN = 0V ,10V; VLINE = 48V
Shutdown Supply Current
VEN = 0V ; VCC = 18V
–10
1
1.3
mA
0.1
10
µA
0.1
10
µA
0.83
0.888
V
Protection and Control
Current Limit Threshold Voltage
0.772
Current Limit Delay to Output
VISNS = 0V to 5V
Current Limit Source Current
VISNS = 0V
34
Enable Input Threshold (Turn-on)
30
40
50
µA
1
1.6
2.2
V
Enable Input Hysteresis
CPWR Input Current
150
VCPWR = 5V, 0V
–1
CPWR Threshold
Soft-Start Current
ns
mV
µA
+1
1.6
VSS = 0V
Line UVLO Threshold (Turn-on)
V
2.5
4
6
µA
1.16
1.22
1.28
V
Line UVLO Threshold Hysteresis
140
mV
Thermal Shutdown
145
°C
Thermal Shutdown Hysteresis
25
°C
ns
MOSFET Driver
Output Minimum On-Time
VISNS = 5V
0
Output Driver Impedance
SOURCE ; ISOURCE = 200mA
8
12
Ω
SINK ; ISINK = 200mA
4
6
Ω
Rise Time
COUT = 500pF
12
ns
Fall Time
COUT = 500pF
8
ns
Note 1.
Exceeding the absolute maximum rating may damage the device.
Note 2.
The device is not guaranteed to function outside its operating rating.
Note 3.
Devices are ESD sensitive. Handling precautions recommended.
Note 4.
If a substained DC voltage >150V is applied to the LINE pin, a current-limiting 1.8kΩresistor should be used in series with the LINE pin. This
condition does not apply for transient conditions over 150V.
Note 5.
For oscillator frequencies above 2.5MHz it may be necessary to power to VBIAS pin from an external power source due to the current
limitations of the internal 5V regulator. See Applications Information for details.
MIC9131
4
July 2001
MIC9131
Micrel
Typical Characteristics
-1
-1.5
-2
-2.5
-40
Error Amp Reference Voltage
vs. Temperature
160
THRESHOLD (V)
1.210
1.205
1.200
1.195
1.190
1.185
1.6
1.4
1.2
8
12
14
VCC (V)
16
18
VCC = 10V
Rt = 9.53K
Ct = 470pf
3
2.5
fOSC = 200kHz
2
1.5
July 2001
1.4
1.38
VCC = 10V
RBIAS = 560K
Rt = 9.47K
Ct = 470pF
1.36
1.34
1.32
1.3
-40 -20 0 20 40 60 80 100120140
TEMPERATURE (°C)
90
200 400 600 800 1000 1200
RBIAS (kΩ)
Ct = 470pF
5
4
3
2
Ct = 100pF
1
0
ISNS to Gate Output Delay
vs. Overdrive
180
160
70
60
50
40
30
0
0
7
6
200
80
10
0
9
8
GATE DRIVE FREQUENCY (kHz)
ISNS to Gate Output Delay
vs. RBIAS
20
1
0.5
1.44
1.42
Quiescent Current
vs. RBIAS
3.5
QUIESCENT CURRENT (mA)
10
160
10
1.48
1.46
0
50
1.8
0
40
80
120
TEMPERATURE (°C)
Quiescent Current
vs. Frequency
1.5
= 560K
Rt = 9.47K
C = 470pF
t
2.0
1.18
-40
DELAY (ns)
BIAS
QUIESCENT CURRENT (mA)
QUIESCENT CURRENT (mA)
R
2.2
8 9 10 11 12 13 14 15 16 17 18
VCC (V)
1.19
Quiescent Current
vs. Temperature
Quiescent Current
vs. VCC Voltage
2.4
1.0
1.180
1.2
200 400 600 800 1000 1200
RBIAS(kΩ)
5
500
2.480
-40 -20 0 20 40 60 80 100120140
TEMPERATURE (°C)
1.21
RBIAS=160K
140
120
100
80
RBIAS=560K
RBIAS=360K
60
40
20
0
2000
2.485
1.22
400
450
2.490
VCC=10V
RBIAS=560K
1.23
1600
1800
2.495
Line UVLO Threshold
vs. Temperature
300
350
2.500
8 9 10 11 12 13 14 15 16 17 18
VCC (V)
1.24
UVLO THRESHOLD (V)
1.215
2.505 R
BIAS = 560K
2.499
Line UVLO Threshold
vs. VCC
1.220
VCC = 10V
DELAY (ns)
REFERENCE VOLTAGE (V)
2.510
0
40
80
120
TEMPERATURE (°C)
2.500
1200
1400
Ct=470pF
8 9 10 11 12 13 14 15 16 17 18
VCC (V)
-0.5
2.501
0
200
-0.5
Ct = 470pF
0
QUIESCENT CURRENT (mA)
-0.4
0.5
RBIAS = 560K
200
250
FOSC(NOM)=200kHz
Rt=9.47K
1
800
1000
-0
-0.2
VCC = 10V
RBIAS = 560K
Rt = 9.47K
1.5
100
150
0.0
Error Amp Reference Voltage
vs. VCC Voltage
2.502
400
600
0.1
OSC FREQ. VARIATION (%)
OSC FREQ. VARIATION (%)
0.2
-0.3
Oscillator Frequency
vs. Temperature
2
REFERENCE VOLTAGE (V)
Oscillator Frequency
vs. VCC Voltage
0.3
OVERDRIVE (mV)
MIC9131
MIC9131
Micrel
V
BIAS
5.012
vs. V
CC
5.06
5
4.96
5.000
VCC=10V
7.2
RBIAS=560K
7
6.8
Vcc GLO Off
6.6
6.4
-40 -20 0 20 40 60 80 100120140
TEMPERATURE (°C)
821.0
820.5
820.0
819.5
819.0
818.5
818.0
8 9 10 11 12 13 14 15 16 17 18
VCC (V)
Enable Threshold
vs. VCC
2.5
1.95
1.9
1.85
1.8
1.75
1.7
1.65
1.6
1.55
1.5
8 9 10 11 12 13 14 15 16 17 18
VCC (V)
2
SINK
1.5
1
SOURCE
0.5
0
8 9 10 11 12 13 14 15 16 17 18
VCC (V)
Peak Short Circuit Depletion
FET Current vs. VLINE
85
16
80
14
–40°C
65
25°C
CURRENT (mA)
75
70
60
55
125°C
50
45
40
0
VCC = 0V
40
80
120
VLINE (V)
160
200
Gate Drive Current
vs. VCC
0
1
2
3
IBIAS (mA)
4
5
VCC= 10V
RBIAS= 560K
832
830
828
826
824
822
820
818
816
-40
0
40
80
120
TEMPERATURE (°C)
160
Peak Short Circuit Depletion
FET Current vs. Temperature
Depletion FET Current
vs. VLINE
85
180V Line
80
VCC = 0V
75
70
65
60
55
48V Line
50
45
40
-40
12
–40°C
0
40
80
120
TEMPERATURE (°C)
160
Depletion FET Current
vs. Low VLINE Voltage
–40°C
10
12
25°C
10
125°C
8
6
4
2
0
0
4.96
836
834
THRESHOLD (mV)
THRESHOLD (mV)
THRESHOLD (V)
7.4
2
RBIAS=560K
VCC = 10V
821.5
4.97
ISNS Current Limit Threshold
vs. Temperature
822.0
Vcc GLO On
7.6
SINK/SOURCE CURRENT (A)
7.8
4.98
4.94
ISNS Current Limit Threshold
vs. VCC Voltage
vs. Temperature
4.99
4.95
4.94
-40 -20 0 20 40 60 80 100120140
TEMPERATURE (°C)
8 9 10 11 12 13 14 15 16 17 18
VCC (V)
VCC Turn On/Off Thresholds
THRESHOLD VOLTAGE (V)
VCC = 10V
RBIAS = 560K
Rt = 9.47K
Ct = 470pF
4.98
SHORT CIRCUIT CURRENT (mA)
5.002
5.02
40
VCC=7.5V
RBIAS=560K
En=7.5V
80
120 160 200
VLINE (V)
6
CURRENT (mA)
5.004
VCC = 10V
5
VBIAS VOLTAGE (V)
BIAS VOLTAGE (V)
VBIAS (V)
5.006
SHORT CIRCUIT CURRENT (mA)
5.01
5.04
5.008
MIC9131
VBIAS Load Regulation
RBIAS = 560K
5.010
4.998
VBIAS Voltage
vs. Temperature
25°C
8
125°C
6
4
2
0
7
7.5
8
8.5
9
VLINE (V)
9.5
10
July 2001
MIC9131
ISNS CURRENT (µA)
41.5
ISNS Pin Source Current
vs. VCC
45
RBIAS=560K
ISNS CURRENT (µA)
42
Micrel
41
40.5
40
39.5
39
38.5
38
8
July 2001
10
12
14
VCC (V)
16
18
ISNS Pin Source Current
vs. Temperature
44 RBIAS=560K
43 VCC=10V
42
41
40
39
38
37
36
35
-40 -20 0 20 40 60 80 100120140
TEMPERATURE (°C)
7
MIC9131
MIC9131
Micrel
Oscillator Frequency vs.
RC Values
1000000
RESISTOR VALUE (Ω)
47pF
100pF
100000 220pF
470pF
680pF
1000pF
10000
2200pF
1000
10000
Note:
*See applications section for higher
switching frequencies
100000 1000000 10000000
FREQUENCY (Hz)
Output switching frequency is 1/4 the oscillator frequency.
Functional Block Diagram
FB
COMP
OSC
6
7
SYNC
4
5
Oscillator
2
VCC
1.2V
SR Latch
2
R
2.5V
Error
Amplifier
ISNS
40µA
S1
14
EN
RBIAS
9 5V
OUT
15
PGND
11
AGND
0.82V
BIAS
REG
10
3
Peak
Current Limit
1.21V
5V
MAXIMUM DUTY CYCLE
4µA
SS
16
S2
PWM
VBIAS
Q
12
Max.
Duty Cycle
CPWR
Current
Limit
Selection
8
Q
R1
R2
VCC
2
S
1-Shot
VCC
UVLO
LINE
1
UNDERVOLTAGE LOCKOUT
Thermal
Shutdown
LINE
UVLO
13
UVLO
Figure 1
MIC9131
8
July 2001
MIC9131
Micrel
• Control loop operation
• Current sensing & overcurrent protection
• Slope compensation
• Error amplifier
High Voltage Start Up Circuit
Many conventional Off-Line and Telecom power supplies use
an external bias resistor and zener diode to supply the initial
start-up voltage for the control IC. The control IC gets its
supply voltage from a bias winding once the power supply is
running. This method has the disadvantages of extra components (diode and power resistor), continuous power dissipation in the resistor and a large bias capacitor, used to supply
the IC until the bias winding takes over.
The MIC9131 eliminates these problems by using an internal
depletion mode MOSFET as a pre-regulator to provide the
start-up bias voltage from the high voltage input of the power
supply. This approach eliminates the need for external start
up components and reduces the size of the controller’s bias
supply capacitor. The MOSFET is turned off once the external bias winding takes over, which eliminates power dissipation in the start-up circuit. In some cases, the MIC9131 may
be run directly from the input voltage rail, eliminating the need
for an external bias winding.
Functional Description
Micrel’s MIC9131 is a high voltage, high speed current mode
switching power supply controller. It uses a BiC/DMOS
process to achieve a high voltage input, low quiescent current
and very fast internal delay times. The MIC9131 is designed
to drive an external low side N-channel MOSFET, which
makes it suitable for controlling Boost, Flyback and Forward
converter topologies. The high voltage startup pin eliminates
the requirement for an external start up circuit. This makes
it ideal for use with Telecom converters.
A block diagram of the MIC9131 is shown in Figure 1. The
description of the controller is divided into 6 basic functions:
• Power and bias circuitry
• High voltage start-up circuit
• VCC and Bias supplies
• Enable and Undervoltage monitoring circuits
• VCC and VIN UVLO
• Enable
• Oscillator and sync circuitry
• Soft-start and soft-start reset circuits
• MOSFET gate drive circuits
Transformer
Bias
Winding
MIC9131
Internal
Circuitry
VCC
2
VIN
Line
1
1.21V
180V
DEPLETION
FET
VCC
UVLO
THERMAL
SHUTDOWN
VPR(OFF) Depletion FET Pre-Regultor turn-off threshold
∆VPR
∆VGLO VCC Gate Lockout Hysteresis
Depletion FET turn-on threshold
VGLO(ON) VCC gate lockout turn on threshold
VCC voltage when powered from VLINE
Figure 2
July 2001
9
MIC9131
MIC9131
Micrel
topologies since the variation is small (equal to the ∆VPR
hysteresis). The bias regulator in the MIC9131 buffers the
internal circuits from VCC variations.
The pre-regulator FET is protected by a thermal shutdown
circuit, which turns the MOSFET off if its temperature exceeds approximately 150°C.
When operating at input voltages greater than 150V, a fast
input voltage risetime during turn-on (which may occur during
a hot plug operation)may cause a high peak current to flow
through the depletion FET, damaging the MIC9131. A 1.8kΩ
resistor in series between the input voltage and the Line pin
(pin 1) is recommended when operating at input voltages
greater than 150V. This resistor limits the maximum peak
current to 100mA (at 180VIN) and protects the part.
The depletion mode MOSFET contains an internal parasitic
diode. The VIN pin voltage must be greater than the VCC
voltage or the VCC voltage will be clamped to a diode drop
greater than the VIN voltage. Excessive power dissipation in
the parasitic diode will destroy the IC.
VCC and Bias Supplies
The power for the controller and gate drive circuitry is supplied through the VCC pin. The gate drive current is returned
to ground through the power ground pin (PGND). The rest of
the supply current is returned to ground through the analog
ground pin (AGND). The two ground pins must be connected
together through the PCB ground plane.
High frequency decoupling is provided at the VCC pin to
supply the gate drive’s peak current requirements. Turn-on
of the external MOSFET causes a voltage glitch on the VCC
pin. If the glitch is excessive, this disruption can appear as
noise or jitter in the oscillator circuit or the gate drive waveform. The decoupling capacitor must be able to supply the
MOSFET gate with the charge required to turn it on. A 0.1µF
ceramic capacitor is usually sufficient for most MOSFETs.
Larger FETs, with a higher gate charge requirement may
require a 0.22µF ceramic capacitor or a ceramic capacitor
paralleled with a 2.2µF tantalum or 4.7uF aluminum electrolytic. It is recommend that if VLINE is greater than 150V DC
than the maximum capacitor recommended on VCC is
2.2µF.The capacitor must be located next to the VCC pin of
the MIC9131. The ground end of the capacitor should be
connected to the ground plane, making a low impedance
connection to the power ground pin (pin 15).
The internal bias regulator block provides several internal
and external bias voltages. Referring to Figure 1, a 2.5V
reference is used for the internal error amplifier, a 0.82V bias
is used by the current limit comparator and a 1.21V reference
is used by the Line UVLO circuit. An external 5V bias voltage
(VBIAS) powers the oscillator circuit and may be used as a
reference voltage for other external components. The VBIAS
pin requires a minimum 0.1µF capacitor to ground for
decoupling.
Enable and Undervoltage Monitoring Circuits
The two undervoltage lockout circuits in the MIC9131 are
shown in Figure 4. One monitors the VCC voltage and the
other monitors the input line voltage. These signals are OR’d
Start-up circuit operation is illustrated in Figure 2. VIN is
applied and the depletion FET, which is normally enabled
allows current from VIN to charge the VCC bias capacitor.
Once the VCC voltage reaches the VCC enable threshold,
VGLO(ON) , the gate drive is enabled and the MIC9131 starts
switching. Vcc continues to increase until the Pre-Regulator
turn-off threshold, (VPR(OFF)), is reached and the depletion
FET is turned off. The Vcc voltage decreases as energy from
the bias capacitor is used to supply the controller. The
depletion FET is turned back on when the pre-regulator turnon threshold is reached. A bias winding derived supply
voltage, set higher than the FET turn-off threshold, VPR(OFF),
raises the VCC voltage over the threshold and prevents the
FET from turning on.
In certain designs the MIC9131 may be powered directly from
the Line voltage, eliminating the need for an extra transformer
bias winding. When operating in this fashion the designer
must insure the power dissipation in the IC does not cause the
die temperature to exceed the 125°C maximum. Power
dissipation is calculated by:
PDISS = (VIN − VCC ) × IVCC
Where :
VIN is the line input voltage
VCC is the average VCC voltage (typically 8.5V)
IVCC is the total current drawn by the IC
IVCC is the sum of the operating current of the MIC9131 at a
given frequency and the average current required to drive the
external switching MOSFET. A plot of typical operating
current vs. frequency is given in Figure 3. The average
MOSFET gate drive current is calculated in the “MOSFET
GATE DRIVE” section of this specification.
10
9
8
7
6
5
Ct = 470pF
4
3
2
1
500
400
450
300
350
200
250
150
0
0
Ct = 100pF
50
100
QUIESCENT CURRENT (mA)
Quiescent Current
vs. Frequency
GATE DRIVE FREQUENCY (kHz)
Figure 3
The die junction temperature is calculated by
TJ = TA + PDISS × θ JA
Where: TJ is the die junction temperature
TA is the ambient temperature of the circuit
θJA is the junction to ambient thermal resistance of
the MIC9131 (listed in the operating ratings
section of the specification.
When powered directly from the Line voltage, the VCC voltage
will vary between the upper and lower pre-regulator thresholds. The amplitude of the output gate drive voltage will vary
with the VCC voltage. This should not be a problem for most
MIC9131
10
July 2001
MIC9131
Micrel
5V
MIC9131
4µA
12
SS
SET
S
VCC
Q
R
2
VCC
UVLO
1.21V
/Q
RESET
UVLO
VIN
AGND
11
UVLO
13
16
OUT
R1
LINE
UVLO
R2
15
PGND
Figure 4: UVLO and Soft Start Circuits
together and either one can disable the gate drive pin and
discharge the voltage on the soft start capacitor.
VCC Undervoltage Lockout
The VCC voltage is internally divided down and compared to
a 1.21V internal bandgap reference. As VCC rises above the
turn-on threshold, it disables the VCC undervoltage lockout
circuit. Once above the turn-on threshold, hysteresis prevents the lockout circuit from disabling the IC until the VCC
voltage falls below the lower threshold.
Line Undervoltage Circuit (UVLO)
The line voltage is monitored by an external resistor divider
and fed into the negative input of the line UVLO comparator.
As the comparator trip point is exceeded, the line UVLO
circuit is disabled. Hysteresis built into the comparator prevents the circuit from toggling on an off in the presence of
noise or a high input line impedance.
The line voltage turn-on trip point is:
(VTHRESHOLD – VHYST ) ×
R1+ R2
R2
Enable
A low level on the enable pin turns off all the functions of the
MIC9131 and places it in a low quiescent current state. The
output driver is in a low state. When the enable pin is pulled
high, the MIC9131 goes through its normal start up sequence
including undervoltage lock out and soft start. When not used,
the pin should be connected to VCC.
Oscillator Block
An external resistor and capacitor set the oscillator frequency. The MIC9131 contains an internal divide-by-four
circuit that limits the maximum duty cycle at the gate drive to
75%. The oscillator frequency of the MIC9131 is four times
the output switching frequency.
Oscillator Pin
The operation of the oscillator is shown in Figure 5. The
voltage waveform at the OSC pin is a sawtooth whose
amplitude increases as capacitor Cosc is charged up through
ROSC from the 5VBIAS. When the OSC pin voltage reaches
the internal comparator upper threshold, COSC is quickly
discharged to zero volts by an internal MOSFET. After a brief
delay, typically 75ns, the internal MOSFET is turned off and
the COSC charges, repeating the cycle. Figure 5 show the
relationship between the oscillator and gate drive waveforms.
The delays in the IC force the duty cycle of the gate drive
signal to be slightly less than 75% duty cycle.
For VBIAS = 5V and a peak oscillator waveform voltage of 3V,
the design equations simplify to:
Charging
R1+ R2
R2
where: VTHRESHOLD is the voltage level of the internal
comparator reference, typically 1.21V.
The line hysteresis is equal to:
VLINE_ON= VTHRESHOLD ×
R1 + R2
R2
where: VHYST is the internal hysteresis level, typically
75mV.
VHYSTERESIS is the hysteresis of the line input
voltage
The MIC9131 will be disabled when the line voltage drops
back down to:
VHYSTERESIS = VHYST ×
t CHARGE = 0.92 × R t × C t
VLINE_OFF = VLINE_ON − VHYTERESIS =
Discharging
tDISCHARGE ≈ 40 × C t
July 2001
11
MIC9131
MIC9131
Micrel
TP_OSCILLATOR = t CHARGE + tDISCHARGE + tDELAY
Where tDELAY = 75ns
fS _ OSCILLATOR =
1F
1
TP _ OSCILLATOR
2N3904
VCC
2
SYNC
5
VBIAS
9
3V
4.7 F
ROSC
1.6k
1
× fS _ OSCILLATOR
4
The timing capacitor, COSC, should be an NPO ceramic or a
temperature stable film capacitor. Care must be taken when
using capacitor values less than 47pF. The high impedance
of a small value capacitor makes the OSC pin more susceptible to switching noise. Also, the input capacitance of the
OSC pin and the stray capacitance of the board will have a
noticeable effect on the oscillator frequency.
fS _ OUTPUT =
SYNC
5
VBIAS
9
ROSC
OSC
OSC
4
4.7 F
COSC
33pF
AGND
11
75ns
1-shot
Figure 5a
Oscillator Synchronization
The switching frequency of the MIC9131 can be synchronized to an external oscillator or frequency source. Figure 6
shows the relationship between the sync input, oscillator
waveform and gate drive output. The external frequency
should be set at least 15% greater than the free running
oscillator frequency to account for tolerances in the oscillator
circuit and external components. The positive edge of the
sync signal resets the oscillator. The sync pulse frequency,
like the oscillator, is four times the gate drive frequency.
When an external sync signal is applied, the peak amplitude
of the oscillator signal (pin 4) is less than when it is free
running because the oscillator signal is terminated before it
reaches its 3V (typical) amplitude. When not used, the sync
pin should be connected to ground to prevent noise from
erroneously resetting the oscillator.
3V
4
COSC
75ns
1-shot
11
AGND
Sync Input
(pin 5)
VOSC
Oscillator
Waveform
(pin 4)
Gate Drive
(pin 16)
Gate Drive
(pin 16)
tON
tPERIOD
Figure 5
Higher Switching Frequencies
The MIC9131 is capable of very high switching frequencies.
One of the limitations on the maximum frequency is the
current capability of the 5V regulator supplying the oscillator
and VBIAS. By powering VBIAS with an external source, e.g.
linear regulator much higher switching frequencies can be
achieved. A simple way of using an external current source
is to set an NPN as an emitter follower. Figure 5b shows the
MIC9131 oscillator frequency set to 4MHz using an external
NPN. The emitter followerj circuit allows the current to be
supplied by VCC while the voltage is regulated to a diode drop
below VBIAS. This configuration is quite stable over temperature and voltage variations.
MIC9131
TIME (500ns/div)
Figure 6. Sync Waveform
Soft Start Circuit
The soft start is programmed by a capacitor on the soft start
pin. A 4µA current source charges up the capacitor. At power
up, the SS pin is discharged. Once the UVLO and enable
functions release the soft start circuit, the voltage of the
capacitor increases. The active voltage range of the soft start
pin is from typically from 0.9V to 1.7V. The internal current
source increases the voltage on the soft start capacitor to
approximately 4V. The soft start pin and the current sense
voltage are connected to a comparator in the MIC9131. The
voltage from the soft start pin effectively limits the peak
current through the current sense resistor by prematurely
terminating the on-time of the gate drive output. Referring to
Figure 1, with the soft start voltage low, the duty cycle of the
output is at a minimum. As the soft start voltage increases,
12
July 2001
MIC9131
Micrel
A resistor placed in series with the gate drive output attenuates ringing in the etch connection between the MIC9131 and
the MOSFET. Figure 8 shows a single resistor in series
between the driver output and the gate of the MOSFET. The
zener value should be greater than the gate drive voltage to
prevent excessive power dissipation, but less than the maximum gate to source voltage rating.
the duty cycle of the gate drive output increases until the error
amplifier takes control of the duty cycle. The soft start
capacitor is discharged by an internal MOSFET in the
MIC9131.
The soft start circuit is activated by the following events:
1. Line undervoltage pin less than the 1.21V threshold
2. VCC becomes less than the pre-regulator voltage turn
off threshold.
3. The current limit comparator threshold is exceeded.
This can be disabled with a low level on the CPWR
pin.
4. A low level on the enable pin.
Calculating the soft capacitor depends on many parameters
such as the current limit of the circuit input voltage, output
power and output loading. A starting value of capacitor should
be chosen and the value can be adjusted later in the design.
Recommended starting values of soft start capacitance is
typically 10nF to 100nF. Values below 1nF may be ineffective
in slowing the output voltage turn on time.
CPWR Current Limit Selection
This pin controls whether the soft start circuit is reset if the
voltage on the Isns pin exceeds the overcurrent threshold.
When the CPWR pin is high, an overcurrent condition at the
ISNS pin will terminate the on-time of the gate drive pulse and
discharge the soft start capacitor to 0V. This delay in start up
contributes to a reduction in the average output current during
an overcurrent or short circuit condition. A smaller MOSFET
may be used since the power dissipation in the MOSFET is
minimized under short circuit or overcurrent conditions.
If the CPWR pin is low an overcurrent or short circuit conditions will not trip the soft start circuit. The pulse-by-pulse
current limit, inherent in current mode control, provides a
“brick wall” or constant current limit. With the power supply
operating in this mode, a smaller soft start capacitor can be
used to increase the turn on speed of the supply.
If the CPWR in is held low during the initial turn on at power
up and then raised high, the power supply can maximize the
turn-on time at start up and still provide a high level of
overcurrent and short circuit protection. The circuit shown in
Figure 7 performs this function.
Gate Drive
Output
GND
Figure 8
The circuitry shown in figure 9 allow different rise and fall
times. R1 and the input capacitance of the MOSFET determine the rise-time of the gate voltage and therefore the turnon time of the MOSFET. The diode, D1 is reversed biased,
which removes R2 from the circuit. At turn-off, D1 is forward
biased and the parallel combination of R1 and R2 controls the
turn-off time of the MOSFET. The turn on-time is slower,
which reduces switching noise and ringing during turn-on.
The turn-off time is faster, which minimizes switching losses
during turn-off and improves efficiency. If the turn-on time is
to be faster than the turn-off time, the diode should be
reversed.
R2
GND
Figure 9
A gate drive transformer is used where an increase in drive
voltage, isolation and/or voltage level shifting are required.
Gate drive transformers can have multiple windings and drive
multiple MOSFETs, including MOSFETs that require a drive
signal 180°C out of phase with the ICs drive signal.
Figure 10 shows a gate drive transformer circuit. The capacitor, C1 removes DC from the drive circuit and prevents
transformer saturation. R1 provides damping to eliminate
ringing in the circuit. R1 is usually in the 5 to 20Ω range,
depending on the amount of damping necessary. D1 and D2
form a clamp circuit, which prevents the voltage from exceeding the VGMAX level. If the gate drive is well damped, the
diodes may be removed R2 is used to allow the transformer
to reset properly.
R1
CPWR
C1
AGND
Figure 7
MOSFET Gate Drive Output
The MIC9131 has the capability to directly drive the gate of a
MOSFET. The output driver consists of a complimentary
P-channel and N-channel pair. The typical switching time of
the output is dependent on the IC supply voltage and the gate
charge required to turn the MOSFET on and off.
July 2001
R1
Gate Drive
Output
MIC9131
VREF
D1
D1
13
MIC9131
MIC9131
Micrel
C1
T1
Current Sense Circuit
The current sense input of the MIC9131 has three unique
features, which are advantageous in a high speed, high
efficiency power supply.
1. The overcurrent threshold is nominally 0.82V instead
of the typical 1.0V found in most switching control
ICs.
2. The current sense pin sources a nominal 40µA of
current out of the pin. This is used to raise the current
limit threshold of the pin, which allows a smaller
current sense resistor to be used. This improves the
efficiency of the power supply, especially in lower
current applications.
3. The delay from the current sense input to the output
is typically 50ns.
The current limit threshold of the ISNS pin was set at 0.82V,
allowing the use of a smaller current sense resistor. A stable,
bandgap derived 40µA current is sourced from the ISNS pin.
A voltage drop across a series resistor placed between the
pin and the current sense resistor level increases the current
sense signal at the ISNS pin. This allows the use of a smaller
current sense resistor if the full 0.82V peak to peak current
signal is not required. Decreasing the value of the current
sense resistor decreases the power dissipation in the resistor, which improves the efficiency of the power supply.
The delay between the input of the overcurrent comparator
and the output gate drive is nominally 50ns. This very fast
response time allows the MIC9131 to operate at higher
frequencies and still have adequate overcurrent protection.
The operation of the current sense input is as follows. The
sensed current in the power supply is converted to a voltage
by a resistor or current sense transformer. Referring to Figure
1, this voltage is compared to the output of the error amplifier,
which sets the duty cycle of the gate drive output. The current
signal is also connected to an Imax comparator. Comparing
the current sense signal to the reference voltage sets a
maximum current limit. If the maximum amplitude of the
current sense signal exceeds the reference, the comparator
terminates the gate drive output pulse. It aslo discharges the
soft start capacitor when the CPWR pin is high.
Leading Edge Current Spike
The current signal in a power circuit will often have a leading
edge spike caused by leakage inductance, parasitic inductance and capacitance, diode reverse recovery effects and
snubbers. These spikes can cause premature termination of
the switching cycle if they are not eliminated.
A resistor may be added in series between the current sense
resistor and the Isns input. The input and board trace capacitance of the ISNS pin (pin 14) is approximately 25pF. A 1k
resistor is a good choice, since it attenuates most of the ripple
without distorting the current sense waveform. It has a
minimal effect on level, offsetting the current sense signal by
only 40mV.
A typical rule of thumb is the bandwidth of the RC filter should
be at least 6 times the switching frequency. This avoids
distorting the current sense waveform and adding excessive
delays in the current loop that will interfering with overcurrent
protection. For a 100kHz switcher, the maximum series
R1
Gate Drive
Output
D2
R2
GND
D1
1:N
Figure 10
The gate impedance of a MOSFET is capacitive and the
power required to drive the gate is proportional to the charge
required to turn on the MOSFET, the peak gate voltage and
the switching frequency. Assuming the total gate charge for
turn on and turn off is equal, the power used to switch the
MOSFET on and off is:
PDRIVE = Q G × VGS × fS
where: QG is the total gate charge at VGS
VGS is the gate to source voltage of the MOSFET
usually equal to VCC
fS is the output switching frequency
The power required to drive the MOSFET is dissipated in the
drive circuitry of the MIC9131. This power must not cause the
die temperature to exceed the maximum rated junction
temperature of 125°C.
MOSFET Driver IC’s are used when the drive requirement for
the MOSFETs is greater than the capability of the MIC9131
gate drive output. While the peak current of the MIC9131 gate
drive is typically 1.2A at VIN =12V, a gate driver ICs will sink
or source between 1.2A and 12A of peak current. The higher
peak current allows faster rise and fall times for larger
MOSFETs.
The drive requirements for selecting a MOSFET driver are
determined using the following equation:
QG
t
where: QG is the total gate charge required to turn on the
MOSFET at a specified ID, VG and VDS. This
information is usually given in the MOSFET
specification sheet.
t is the gate voltage transition time (risetime or fall
time)
IPK is the peak current requirement of the
MOSFET driver IC.
For example, if a MOSFET is chosen with a QG of 60nC and
it is desired to have a 50nS gate to source risetime/falltime,
the peak current requirement of the MOSFET driver is:
IPK = 2 ×
2 × 60nC
= 2.4A
50ns
A driver such as the MIC4424 will meet this requirement. For
more information on choosing a MOSFET driver, see the
Micrel application note AN-24, “Designing with Low Side
MOSFET Drivers.”
IPK =
MIC9131
14
July 2001
MIC9131
Micrel
resistance is 10K, for a 500kHz switcher, the maximum series
resistance is 2K.
Sensing Current with a Resistor
The fast transition times of the current signal prohibit the use
of inductive resistors. Standard wire wound power resistors
will not work. Carbon composition or metal film resistors or
low inductance power resistors may be used. The overcurrent
range of the power supply and component tolerances must
be considered when selecting the current sense resistor
value. The power supply specification may call for an
overcurrent limit, which must be accounted for when selecting the current sense resistor value. The relationship between the peak primary current and the current sense resistor
is:
the current loop. This limits the switching frequency to the
range of 100kHz.
Sensing Current with a Current Sense Transformer
At higher power levels, the power dissipation in a current
sense resistor is excessive. A current sense transformer can
be used to sense the current while minimizing power dissipation. See Figure 11. The schematic shows the circuitry
necessary when using a current sense transformer. The
resistor, R1, provides a path to reset the current sense
transformer. The resistor, R2, converts the scaled down
current to a voltage, which is sent to the ISNS pin.
VIN
VISNS = IP × RISENSE + IISNS × R f
where: Ip is the current in the sense resistor
RISENSE is the current sense resistance
IISNS is the current sourced from the ISNS pin
(40µA)
Rf is the series resistor between the ISNS pin and
the current sense resistor.
The current sense resistor must not be too small or the current
sense signal will be susceptible to noise. If noise is a problem,
the current signal level should be increased.
An example is illustrated below.
The maximum peak current, IPMAX= 1A at 120% overcurrent
and minimum input voltage
The maximum rms current, IRMS=0.65A
The desired current sense signal amplitude is 500mV at 1A
output current.
The current sense resistor value and power dissipation is:
RSENSE =
R2
Figure 11
The voltage at the ISNS pin is calculated by:
IP
× R2 + IISNS × R f
N
where: IP is the current in the primary of the current sense
transformer
R2 is the current sense resistance at the
secondary of the current sense transformer
N is the turns ratio of the current sense
transformer (N=Nsec/Npri)
IISNS is the current sourced from the ISNS pin
(40µA)
Rf is the series resistor between the ISNS pin and
the current sense resistor.
Current Transformer Example:
The maximum peak current, IPMAX = 5A at 120% overcurrent
and minimum input voltage
The maximum rms current, IRMS = 3.25A
The full 0.82V peak signal a the ISNS input can be used since
very little power is dissipation in the secondary side sense
resistor. The maximum peak to peak voltage at the sense pin
(pin 14) is 0.82V at the 5A maximum output current.
The current sense resistor value and power dissipation is:
VISNS =
A 0.5 ohm, non inductive resistor with at least a 1/2W rating
should be selected.
The series resistor is calculated to allow the 500mV-peak
signal to reach 0.82V.
IISNS
=
0.82 − (1× 0.5)
= 10.25kΩ
40µA
The next lower value of 10kΩ is selected.
The bandwidth of the 10K resistor and the 25pF input capacitance is calculated. The resistor value must be lowered if the
bandwidth is too low for the switching frequency.
BW =
R2 =
1
= 630kHz
2 × π × 10k × 25pF
VSENSE × N 0.82 × 100
=
= 16.4Ω
5
IP
2
2
I

 3.25 
PDISS =  PRMS  × R2 = 
 × 16.4 = 17.4mW
 100 
 N 
The maximum switching frequency of this power supply
should be approximately six times less than the BW to
prevent current waveform distortion and excessive delays in
July 2001
IPRI
OUT
(pin 16)
VSENSE 0.5
=
= 0.5Ω
ISENSE
1
VISNS − (IP × RISENSE )
R1
MIC9130
PDISS = IRMS 2 × RSENSE = 0.65 2 × 0.5 = 0.21W
Rf
Current Sense
Transformer
Rf
ISNS
(pin 14)
A 16.2 ohm, 1%, non inductive resistor with at least a 50mW
rating should be selected. A good choice would be an 0805
15
MIC9131
MIC9131
Micrel
size metal film or a 1/8 watt leaded metal film resistor. A series
resistor between the current sense transformer and the Isns
input is not necessary unless it is used for low pass filtering.
If the current sense transformer were not used, the sense
resistor would dissipate 1.7 watts.
RSENSE
secondary winding inductance for the flyback
topology)
M2 is the inductor current downslope
For a boost topology, the inductor downslope is:
di VOUT − VIN + VD
=
dt
L
In a transformer isolated topology, the downslope must be
reflected back to the primary by the turns ratio of the transformer. The reflected downslope is:
M2 =
V
0.82
= SENSE =
= 0.164Ω
5
ISENSE
PDISS = IRMS 2 × RSENSE = 3.25 2 × 0.164 = 1.7W
Slope Compensation
Power supplies using peak current mode control techniques
require slope compensation when they are operating in
continuous mode and have a duty cycle greater than 50%.
Without slope compensation, the duty cycle of the power
supply will alternate wide and narrow pulses commonly
referred to as subharmonic oscillations. Even though the
MIC9131 operates below a 50% duty cycle, slope compensation adds the benefits of improved transient response and
greater noise immunity in the current sense loop (especially
when the current ramp is shallow). Slope compensation can
be implemented by adding an optimum 1/2 of the inductor
current downslope, reflected back to the current sense input.
In real world applications, 2/3 of the inductor current downslope
is used to allow for component tolerances.
Slope compensation at the ISNS input may be implemented
by using a resistor and capacitor as shown in Figure 12. The
rectangular waveshape of the gate drive output is integrated
by the resistor/capacitor filter, which results in a ramp used for
the slope compensation signal. When the gate drive and the
current signal at the sense resistor goes low, the capacitor is
discharged to 0V.
M2REFLECTED = M2 ×
where : Ns/Np is the turns ratio of the secondary winding
to the primary winding.
M2REFLECTED is the inductor curent downslope
reflected to the secondary side of the current
sense transformer.
The reflected downslope is multiplied by the current sense
resistor to obtain the downslope at the current sense input pin
(ISNS).
ISNS _ SLOPE = M2REFLECTED × RS
where Rs is the value of the current sense
resistor.
The required downslope of the compensation ramp at the
ISNS input is:
M3 = ISNS _ SLOPE × 0.67
R1 is know if a value for the resistor between the current
sense resistor and the Isns pin, has already been selected.
If not chose a value of 1k, which will minimize any offset and
signal degradation at the ISNS pin. Select a value of C1 to
minimize signal degradation from the cutoff frequency of R1/
C1. The bandwidth should be at least six times the switching
frequency.
Gate Drive
(pin 16)
R2
MIC9130
ISNS
(pin 14)
R1
C1
Ns
Np
C1 =
RSENSE
1
2 × π × fS × R1
where: fS is the switching frequency of the power
supply (not the oscillator frequency)
The slope of the generated compensation ramp is:
Figure 12
R1
1
×
R2 + R1 R2 × C1
Solving for R2 and assuming R2 is much greater than R1.
M3 = VGATE_DRIVE ×
The procedure outlined below demonstrates how to calculate
the component values.
Compute the inductor current downslope as seen at the
current sense input.
For a flyback, buck or forward mode topology the
inductor downslope is equal to:
M2 =
R2 =
VGATE _ DRIVE × R1
M3 × C1
where: VGATE_DRIVE is the amplitude of the gate
drive waveform
di VO + VD
=
dt
L
where :
VO is the output voltage
VD is the forward voltage drop of the rectifier diode
L is the inductance of the output inductor (or the
MIC9131
16
July 2001
MIC9131
Micrel
Error Amplifier
The error amplifier is part of the voltage control loop of the
power supply. The FB pin is the inverting input to the error
amplifier. The non-inverting input is internally connected to
a 2.5V reference. The output of the error amplifier, COMP, is
connected to the PWM comparator. The error amplifier
July 2001
provides the reference to limit and control the peak current of
the power supply. There is a 1.2V level shift between the
output of the error amplifier and the PWM comparator. This
allows the output of the error amplifier to operate in a linear
region and prevents loading on the COMP pin from interfering
with proper control of the current signal.
17
MIC9131
MIC9131
Micrel
Package Information
PIN 1
0.157 (3.99)
0.150 (3.81)
DIMENSIONS:
INCHES (MM)
0.020 (0.51)
REF
0.020 (0.51)
0.013 (0.33) 0.0098 (0.249)
0.0040 (0.102)
0.050 (1.27)
BSC
0.0648 (1.646)
0.0434 (1.102)
0.394 (10.00)
0.386 (9.80)
45°
0°–8°
0.050 (1.27)
0.016 (0.40)
SEATING
PLANE
0.244 (6.20)
0.228 (5.79)
16-Pin SOP (M)
PIN 1
DIMENSIONS:
INCHES (MM)
0.157 (3.99)
0.150 (3.81)
0.009 (0.2286)
REF
0.025 (0.635)
BSC
0.0098 (0.249)
0.0040 (0.102)
SEATING 0.0688 (1.748)
PLANE 0.0532 (1.351)
0.012 (0.30)
0.008 (0.20)
0.0098 (0.249)
0.0075 (0.190)
0.196 (4.98)
0.189 (4.80)
45°
8°
0°
0.050 (1.27)
0.016 (0.40)
0.2284 (5.801)
0.2240 (5.690)
16-Pin QSOP (QS)
MIC9131
18
July 2001
MIC9131
July 2001
Micrel
19
MIC9131
MIC9131
Micrel
MICREL INC. 1849 FORTUNE DRIVE SAN JOSE, CA 95131
TEL
+ 1 (408) 944-0800
FAX
+ 1 (408) 944-0970
WEB
USA
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This information is believed to be accurate and reliable, however no responsibility is assumed by Micrel for its use nor for any infringement of patents or
other rights of third parties resulting from its use. No license is granted by implication or otherwise under any patent or patent right of Micrel Inc.
© 2001 Micrel Incorporated
MIC9131
20
July 2001