LTC1435 High Efficiency Low Noise Synchronous Step-Down Switching Regulator U DESCRIPTION FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ The LTC®1435 is a synchronous step-down switching regulator controller that drives external N-channel power MOSFETs using a fixed frequency architecture. Burst ModeTM operation provides high efficiency at low load currents. A maximum duty cycle limit of 99% provides low dropout operation which extends operating time in battery-operated systems. Dual N-Channel MOSFET Synchronous Drive Programmable Fixed Frequency Wide VIN Range: 3.5V to 36V Operation Ultrahigh Efficiency Very Low Dropout Operation: 99% Duty Cycle Low Standby Current Secondary Feedback Control Programmable Soft Start Remote Output Voltage Sense Logic Controlled Micropower Shutdown: IQ < 25µA Foldback Current Limiting (Optional) Current Mode Operation for Excellent Line and Load Transient Response Output Voltages from 1.19V to 9V Available in 16-Lead Narrow SO and SSOP Packages The operating frequency is set by an external capacitor allowing maximum flexibility in optimizing efficiency. A secondary winding feedback control pin, SFB, guarantees regulation regardless of load on the main output by forcing continuous operation. Burst Mode operation is inhibited when the SFB pin is pulled low which reduces noise and RF interference. U APPLICATIONS ■ ■ ■ ■ Notebook and Palmtop Computers, PDAs Cellular Telephones and Wireless Modems Portable Instruments Battery-Operated Devices DC Power Distribution Systems , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation. U ■ Soft start is provided by an external capacitor which can be used to properly sequence supplies. The operating current level is user-programmable via an external current sense resistor. Wide input supply range allows operation from 3.5V to 30V (36V maximum). TYPICAL APPLICATION COSC 68pF CSS 0.1µF CC 330pF VIN 4.5V TO 28V COSC VIN RUN/SS TG ITH SW M1 Si4412DY BOOST 100pF VOSENSE SENSE – BG RSENSE 0.033Ω VOUT 2.9V/3.5A R1 32.4k CB 0.1µF INTVCC SGND CIN 22µF 35V ×2 L1 10µH DB CMDSH-3 LTC1435 RC 10k + + 4.7µF M2 Si4412DY R2 22.1k D1 MBRS140T3 COUT + 100µF 10V ×2 PGND SENSE + 1000pF 1435 F01 Figure 1. High Efficiency Step-Down Converter 1 LTC1435 U W W W SYMBOL PARAMETER Main Control Loop IIN VOSENSE Feedback Current VOSENSE Feedback Voltage ∆VLINEREG Reference Voltage Line Regulation ∆VLOADREG Output Voltage Load Regulation VSFB ISFB VOVL IQ Secondary Feedback Threshold Secondary Feedback Current Output Overvoltage Lockout Input DC Supply Current Normal Mode Shutdown VRUN/SS Run Pin Threshold IRUN/SS Soft Start Current Source ∆VSENSE(MAX) Maximum Current Sense Threshold TG Transition Time Rise Time TG t r TG t f Fall Time BG Transition Time Rise Time BG tr BG t f Fall Time Internal VCC Regulator VINTVCC Internal VCC Voltage VLDO INT INTVCC Load Regulation VLDO EXT EXTVCC Voltage Drop VEXTVCC EXTVCC Switchover Voltage Oscillator fOSC Oscillator Frequency 2 U ELECTRICAL CHARACTERISTICS W Input Supply Voltage (VIN)......................... 36V to – 0.3V Topside Driver Supply Voltage (Boost) ......42V to – 0.3V Switch Voltage (SW)............................. VIN + 5V to – 5V EXTVCC Voltage ........................................ 10V to – 0.3V Sense+, Sense– Voltages ......... INTVCC + 0.3V to – 0.3V ITH, VOSENSE Voltages .............................. 2.7V to – 0.3V SFB, Run/SS Voltages .............................. 10V to – 0.3V Peak Driver Output Current < 10µs (TG, BG) ............. 2A INTVCC Output Current ........................................ 50mA Operating Ambient Temperature Range LTC1435C............................................... 0°C to 70°C LTC1435I............................................ – 40°C to 85°C Junction Temperature (Note 1)............................. 125°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C U ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION ORDER PART NUMBER TOP VIEW COSC 1 RUN/SS 2 16 TG 15 BOOST ITH 3 14 SW SFB 4 13 VIN SGND 5 LTC1435CG LTC1435CS LTC1435IG LTC1435IS 12 INTVCC VOSENSE 6 11 BG SENSE– 7 10 PGND SENSE+ 8 9 EXTVCC G PACKAGE S PACKAGE 16-LEAD PLASTIC SSOP 16-LEAD PLASTIC SO TJMAX = 125°C, θJA = 130°C/ W (G) TJMAX = 125°C, θJA = 110°C/ W (S) Consult factory for Military grade parts. TA = 25°C, VIN = 15V, VRUN/SS = 5V unless otherwise noted. CONDITIONS TYP MAX UNITS 1.24 10 1.19 0.002 0.5 – 0.5 1.19 –1 1.28 50 1.202 0.01 0.8 – 0.8 1.22 –2 1.32 nA V %/V % % V µA V 0.8 1.5 130 260 16 1.3 3 150 25 2 4.5 180 µA µA V µA mV CLOAD = 3000pF CLOAD = 3000pF 50 50 150 150 ns ns CLOAD = 3000pF CLOAD = 3000pF 50 40 150 150 ns ns 5.2 –1 230 V % mV V 138 kHz (Note 2) (Note 2) VIN = 3.6V to 20V (Note 2) ITH Sinking 5µA (Note 2) ITH Sourcing 5µA VSFB Ramping Negative VSFB = 1.5V MIN ● 1.178 ● ● ● 1.16 EXTVCC = 5V (Note 3) 3.6V < VIN < 30V VRUN/SS = 0V, 3.6V < VIN < 15V ● VRUN/SS = 0V VOSENSE = 0V, 5V 6V < VIN < 30V, VEXTVCC = 4V IINTVCC = 15mA, VEXTVCC = 4V IINTVCC = 15mA, VEXTVCC = 5V IINTVCC = 15mA, VEXTVCC Ramping Positive COSC = 100pF (Note 4) ● 4.8 ● 4.5 5.0 – 0.2 130 4.7 112 125 LTC1435 ELECTRICAL CHARACTERISTICS TA = 25°C, VIN = 15V, VRUN/SS = 5V unless otherwise noted. The ● denotes specifications which apply over the full operating temperature range. LTC1435CG/LTC1435CS: 0°C ≤ TA ≤ 70°C LTC1435IG/LTC1435IS: – 40°C ≤ TA ≤ 85°C Note 1: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: LTC1435CG/LTC1435IG: TJ = TA + (PD)(130°C/W) LTC1435CS/LTC1435IS: TJ = TA + (PD)(110°C/W) Note 2: The LTC1435 is tested in a feedback loop which servos VOSENSE to the balance point for the error amplifier (VITH = 1.19V). Note 3: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information. Note 4: Oscillator frequency is tested by measuring the COSC charge and discharge currents and applying the formula: ( )( ) 8.4(108) 1 + 1 –1 fOSC (kHz) = C (pF) + 11 I OSC CHG IDIS U W TYPICAL PERFORMANCE CHARACTERISTICS Efficiency vs Input Voltage VOUT = 3.3V Efficiency vs Input Voltage VOUT = 5V VOUT = 3.3V VOUT = 5V 90 ILOAD = 1A ILOAD = 1A 85 ILOAD = 100mA 80 90 EFFICIENCY (%) EFFICIENCY (%) 90 ILOAD = 100mA 85 80 75 75 70 70 VIN = 10V VOUT = 5V RSENSE = 0.033Ω 95 95 95 EFFICIENCY (%) Efficiency vs Load Current 100 100 100 85 80 75 Burst Mode OPERATION 70 CONTINUOUS MODE 65 60 55 0 5 10 15 20 INPUT VOLTAGE (V) 25 30 0 5 10 15 20 INPUT VOLTAGE (V) 25 30 VIN – VOUT Dropout Voltage vs Load Current Load Regulation VITH Pin Voltage vs Output Current 3.0 RSENSE = 0.033Ω ∆VOUT (%) 0.4 0.3 0.2 0.1 – 0.25 2.5 – 0.50 2.0 VITH (V) RSENSE = 0.033Ω VOUT DROP OF 5% – 0.75 0.5 1.0 1.5 2.0 LOAD CURRENT (A) 2.5 3.0 1435 G04 1.5 Burst Mode OPERATION –1.00 1.0 –1.25 0.5 –1.50 0 0 10 1435 G03 0 0.5 1 0.01 0.1 LOAD CURRENT (A) 1435 G02 1435 G01 VIN – VOUT (V) 50 0.001 CONTINUOUS MODE 0 0 0.5 1.0 1.5 2.0 LOAD CURRENT (A) 2.5 3.0 1435 G05 0 10 20 30 40 50 60 70 80 90 100 OUTPUT CURRENT (%) 1435 G06 3 LTC1435 U W TYPICAL PERFORMANCE CHARACTERISTICS Input Supply and Shutdown Current vs Input Voltage 100 VOUT = 5V EXTVCC = VOUT 60 VOUT = 3.3V EXTVCC = OPEN 1.0 40 0.5 20 200 VEXTVCC = 0V 180 70°C 0 25°C – 0.3 0 5 10 15 20 INPUT VOLTAGE (V) 140 25°C 120 100 – 55°C 80 60 40 20 SHUTDOWN 0 70°C 160 0.3 EXTVCC – INTVCC (mV) SUPPLY CURRENT (mA) 80 SHUTDOWN CURRENT (µA) 2.0 0.5 ∆INTVCC (%) 2.5 1.5 EXTVCC Switch Drop vs INTVCC Load Current INTVCC Regulation vs INTVCC Load Current 25 30 0 – 0.5 0 0 10 15 5 INTVCC LOAD CURRENT (mA) 1435 G07 20 0 2 4 6 8 10 12 14 16 18 20 INTVCC LOAD CURRENT (mA) 1435 G09 1435 G08 RUN/SS Pin Current vs Temperature Normalized Oscillator Frequency vs Temperature 10 4 5 3 SFB Pin Current vs Temperature 0 fO –5 SFB CURRENT (µA) RUN/SS CURRENT (µA) FREQUENCY (%) – 0.25 2 – 0.50 – 0.75 –1.00 1 –1.25 –10 – 40 –15 60 35 85 10 TEMPERATURE (°C) 110 135 0 – 40 –15 85 10 35 60 TEMPERATURE (°C) 110 135 –1.50 – 40 –15 60 35 85 10 TEMPERATURE (°C) 1435 G11 1435 G10 Maximum Current Sense Threshold Voltage vs Temperature 110 135 1435 G12 Transient Response Transient Response CURRENT SENSE THRESHOLD (mV) 154 152 VOUT 50mV/DIV VOUT 50mV/DIV 150 148 ILOAD = 50mA to 1A 146 – 40 –15 85 10 35 60 TEMPERATURE (°C) 110 135 1435 G13 4 1435 G14 ILOAD = 1A to 3A 1435 G15 LTC1435 U W TYPICAL PERFORMANCE CHARACTERISTICS Soft Start: Load Current vs Time Burst Mode Operation VOUT 20mV/DIV RUN/SS 5V/DIV INDUCTOR CURRENT 1A/DIV VITH 200mV/DIV ILOAD = 50mA 1435 G16 1435 G17 U U U PIN FUNCTIONS COSC (Pin 1): External capacitor COSC from this pin to ground sets the operating frequency. RUN/SS (Pin 2): Combination of Soft Start and Run Control Inputs. A capacitor to ground at this pin sets the ramp time to full current output. The time is approximately 0.5s/µF. Forcing this pin below 1.3V causes the device to be shut down. In shutdown all functions are disabled. ITH (Pin 3): Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. Nominal voltage range for this pin is 0V to 2.5V. SFB (Pin 4): Secondary Winding Feedback Input. Normally connected to a feedback resistive divider from the secondary winding. This pin should be tied to: ground to force continuous operation; INTVCC in applications that don’t use a secondary winding; and a resistive divider from the output in applications using a secondary winding. SGND (Pin 5): Small-Signal Ground. Must be routed separately from other grounds to the (–) terminal of COUT. VOSENSE (Pin 6): Receives the feedback voltage from an external resistive divider across the output. SENSE – (Pin 7): The (–) Input to the Current Comparator. SENSE + (Pin 8): The (+) Input to the Current Comparator. Built-in offsets between SENSE– and SENSE+ pins in conjunction with RSENSE set the current trip thresholds. EXTVCC (Pin 9): Input to the Internal Switch Connected to INTVCC. This switch closes and supplies VCC power when- ever EXTVCC is higher than 4.7V. See EXTVCC connection in Applications Information section. Do not exceed 10V on this pin. Connect to VOUT if VOUT ≥ 5V. PGND (Pin 10): Driver Power Ground. Connects to source of bottom N-channel MOSFET and the (–) terminal of CIN. BG (Pin 11): High Current Gate Drive for Bottom N-Channel MOSFET. Voltage swing at this pin is from ground to INTVCC. INTVCC (Pin 12): Output of the Internal 5V Regulator and EXTVCC Switch. The driver and control circuits are powered from this voltage. Must be closely decoupled to power ground with a minimum of 2.2µF tantalum or electrolytic capacitor. VIN (Pin 13): Main Supply Pin. Must be closely decoupled to the IC’s signal ground pin. SW (Pin 14): Switch Node Connection to Inductor. Voltage swing at this pin is from a Schottky diode (external) voltage drop below ground to VIN. BOOST (Pin 15): Supply to Topside Floating Driver. The bootstrap capacitor is returned to this pin. Voltage swing at this pin is from INTVCC to VIN + INTVCC. TG (Pin 16): High Current Gate Drive for Top N-Channel MOSFET. This is the output of a floating driver with a voltage swing equal to INTVCC superimposed on the switch node voltage SW. 5 LTC1435 W FUNCTIONAL DIAGRA U U VIN COSC + CIN 1 COSC 4 SFB 13 VIN SGND 5 INTVCC 1.19V REF 1µA DB BOOST 15 – 1.19V TG 16 SHUTDOWN OSC + CB + DROP OUT DET OV S Q R – 1.28V 0.6V SWITCH LOGIC + – SW 14 VOSENSE 6 VFB – – I1 EA + 1.19V Ω R2 gm = 1m + 180k I2 – 4k D1 + VIN + VSEC INTVCC INTVCC CSEC + 12 + – SHUTDOWN R1 5V LDO REG 3µA 6V RUN SOFT START 30k + 4.8V BG 11 8k VOUT – RC 2 RUN/SS CSS 3 ITH CC DFB* SENSE+ 8 7 SENSE – 9 EXTVCC COUT PGND 10 + RSENSE 1435 • FD * FOLDBACK CURRENT LIMITING OPTION 6 LTC1435 U OPERATION (Refer to Functional Diagram) Main Control Loop Low Current Operation The LTC1435 uses a constant frequency, current mode step-down architecture. During normal operation, the top MOSFET is turned on each cycle when the oscillator sets the RS latch, and turned off when the main current comparator I1 resets the RS latch. The peak inductor current at which I1 resets the RS latch is controlled by the voltage on the ITH pin , which is the output of error amplifier EA. The VOSENSE pin, described in the Pin Functions section, allows EA to receive an output feedback voltage VFB from an external resistive divider. When the load current increases, it causes a slight decrease in VFB relative to the 1.19V reference, which in turn causes the ITH voltage to increase until the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by current comparator I2, or the beginning of the next cycle. The LTC1435 is capable of Burst Mode operation in which the external MOSFETs operate intermittently based on load demand. The transition to low current operation begins when comparator I2 detects current reversal and turns off the bottom MOSFET. If the voltage across RSENSE does not exceed the hysteresis of I2 (approximately 20mV) for one full cycle, then on following cycles the top and bottom drives are disabled. This continues until an inductor current peak exceeds 20mV/RSENSE or the ITH voltage exceeds 0.6V, either of which causes drive to be returned to the TG pin on the next cycle. The top MOSFET driver is biased from floating bootstrap capacitor CB, which normally is recharged during each off cycle. However, when VIN decreases to a voltage close to VOUT, the loop may enter dropout and attempt to turn on the top MOSFET continuously. The dropout detector counts the number of oscillator cycles that the top MOSFET remains on and periodically forces a brief off period to allow CB to recharge. The main control loop is shut down by pulling the RUN/SS pin low. Releasing RUN/SS allows an internal 3µA current source to charge soft start capacitor CSS. When CSS reaches 1.3V, the main control loop is enabled with the ITH voltage clamped at approximately 30% of its maximum value. As CSS continues to charge, ITH is gradually released allowing normal operation to resume. Two conditions can force continuous synchronous operation, even when the load current would otherwise dictate low current operation. One is when the common mode voltage of the SENSE+ and SENSE – pins is below 1.4V and the other is when the SFB pin is below 1.19V. The latter condition is used to assist in secondary winding regulation as described in the Applications Information section. INTVCC/EXTVCC Power Power for the top and bottom MOSFET drivers and most of the other LTC1435 circuitry is derived from the INTVCC pin. The bottom MOSFET driver supply pin is internally connected to INTVCC in the LTC1435. When the EXTVCC pin is left open, an internal 5V low dropout regulator supplies INTVCC power. If EXTVCC is taken above 4.8V, the 5V regulator is turned off and an internal switch is turned on to connect EXTVCC to INTVCC. This allows the INTVCC power to be derived from a high efficiency external source such as the output of the regulator itself or a secondary winding, as described in the Applications Information section. Comparator OV guards against transient overshoots > 7.5% by turning off the top MOSFET and keeping it off until the fault is removed. 7 LTC1435 U W U U APPLICATIONS INFORMATION 300 The basic LTC1435 application circuit is shown in Figure 1, High Efficiency Step-Down Converter. External component selection is driven by the load requirement and begins with the selection of RSENSE. Once RSENSE is known, COSC and L can be chosen. Next, the power MOSFETs and D1 are selected. Finally, CIN and COUT are selected. The circuit shown in Figure 1 can be configured for operation up to an input voltage of 28V (limited by the external MOSFETs). COSC VALUE (pF) 250 200 150 100 50 0 RSENSE Selection for Output Current RSENSE is chosen based on the required output current. The LTC1435 current comparator has a maximum threshold of 150mV/RSENSE and an input common mode range of SGND to INTVCC. The current comparator threshold sets the peak of the inductor current, yielding a maximum average output current IMAX equal to the peak value less half the peak-to-peak ripple current ∆IL. Allowing a margin for variations in the LTC1435 and external component values yields: RSENSE = 100mV IMAX The LTC1435 works well with values of RSENSE from 0.005Ω to 0.2Ω. COSC Selection for Operating Frequency The LTC1435 uses a constant frequency architecture with the frequency determined by an external oscillator capacitor COSC. Each time the topside MOSFET turns on, the voltage COSC is reset to ground. During the on-time, COSC is charged by a fixed current. When the voltage on the capacitor reaches 1.19V, COSC is reset to ground. The process then repeats. The value of COSC is calculated from the desired operating frequency: 1.37(104 ) – 11 COSC (pF) = Frequency (kHz) A graph for selecting COSC vs frequency is given in Figure 2. As the operating frequency is increased the gate charge 8 0 100 200 300 400 OPERATING FREQUENCY (kHz) 500 LTC1435 • F02 Figure 2. Timing Capacitor Value losses will be higher, reducing efficiency (see Efficiency Considerations). The maximum recommended switching frequency is 400kHz. Inductor Value Calculation The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of smaller inductor and capacitor values. So why would anyone ever choose to operate at lower frequencies with larger components? The answer is efficiency. A higher frequency generally results in lower efficiency because of MOSFET gate charge losses. In addition to this basic trade-off, the effect of inductor value on ripple current and low current operation must also be considered. The inductor value has a direct effect on ripple current. The inductor ripple current ∆IL decreases with higher inductance or frequency and increases with higher VIN or VOUT: ∆IL = V 1 VOUT 1– OUT VIN ( f)(L) Accepting larger values of ∆IL allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is ∆IL = 0.4(IMAX). Remember, the maximum ∆IL occurs at the maximum input voltage. The inductor value also has an effect on low current operation. The transition to low current operation begins when the inductor current reaches zero while the bottom LTC1435 U W U U APPLICATIONS INFORMATION MOSFET is on. Lower inductor values (higher ∆IL) will cause this to occur at higher load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to decrease. The Figure 3 graph gives a range of recommended inductor values vs operating frequency and VOUT. 60 VOUT = 5.0V VOUT = 3.3V VOUT = 2.5V INDUCTOR VALUE (µH) 50 40 30 20 10 0 0 100 150 200 250 50 OPERATING FREQUENCY (kHz) 300 1435 F03 Figure 3. Recommended Inductor Values Inductor Core Selection Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool Mµ® cores. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Molypermalloy (from Magnetics, Inc.) is a very good, low loss core material for toroids, but it is more expensive than Kool Mµ is a registered trademark of Magnetics, Inc. ferrite. A reasonable compromise from the same manufacturer is Kool Mµ. Toroids are very space efficient, especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more difficult. However, designs for surface mount are available which do not increase the height significantly. Power MOSFET and D1 Selection Two external power MOSFETs must be selected for use with the LTC1435: an N-channel MOSFET for the top (main) switch and an N-channel MOSFET for the bottom (synchronous) switch. The peak-to-peak gate drive levels are set by the INTVCC voltage. This voltage is typically 5V during start-up (see EXTVCC Pin Connection). Consequently, logic level threshold MOSFETs must be used in most LTC1435 applications. The only exception is applications in which EXTVCC is powered from an external supply greater than 8V (must be less than 10V), in which standard threshold MOSFETs (VGS(TH) < 4V) may be used. Pay close attention to the BVDSS specification for the MOSFETs as well; many of the logic level MOSFETs are limited to 30V or less. Selection criteria for the power MOSFETs include the “ON” resistance RSD(ON), reverse transfer capacitance CRSS, input voltage and maximum output current. When the LTC1435 is operating in continuous mode the duty cycles for the top and bottom MOSFETs are given by: V Main Switch Duty Cycle = OUT VIN (V − V ) Synchronous Switch Duty Cycle = IN OUT VIN The MOSFET power dissipations at maximum output current are given by: V 2 PMAIN = OUT (IMAX ) (1 + δ )RDS(ON) + VIN k(VIN ) 1.85 (IMAX )(CRSS )( f) V −V 2 PSYNC = IN OUT (IMAX ) (1 + δ )RDS(ON) VIN 9 LTC1435 U W U U APPLICATIONS INFORMATION where δ is the temperature dependency of RDS(ON) and k is a constant inversely related to the gate drive current. Both MOSFETs have I2R losses while the topside N-channel equation includes an additional term for transition losses, which are highest at high input voltages. For VIN < 20V the high current efficiency generally improves with larger MOSFETs, while for VIN > 20V the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CRSS actual provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage or during a short circuit when the duty cycle in this switch is nearly 100%. Refer to the Foldback Current Limiting section for further applications information. The term (1 + δ) is generally given for a MOSFET in the form of a normalized RDS(ON) vs Temperature curve, but δ = 0.005/°C can be used as an approximation for low voltage MOSFETs. CRSS is usually specified in the MOSFET characteristics. The constant k = 2.5 can be used to estimate the contributions of the two terms in the main switch dissipation equation. The Schottky diode D1 shown in Figure 1 conducts during the dead-time between the conduction of the two large power MOSFETs. This prevents the body diode of the bottom MOSFET from turning on and storing charge during the dead-time, which could cost as much as 1% in efficiency. A 1A Schottky is generally a good size for 3A regulators. CIN and COUT Selection In continuous mode, the source current of the top N-channel MOSFET is a square wave of duty cycle VOUT/ VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: CIN required IRMS ≈ IMAX [V (V OUT IN − VOUT )] 1/ 2 VIN This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations 10 do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor or to choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied the capacitance is adequate for filtering. The output ripple (∆VOUT) is approximated by: 1 ∆VOUT ≈ ∆IL ESR + 4 fC OUT where f = operating frequency, COUT = output capacitance and ∆IL= ripple current in the inductor. The output ripple is highest at maximum input voltage since ∆IL increases with input voltage. With ∆IL = 0.4IOUT(MAX) the output ripple will be less than 100mV at max VIN assuming: COUT required ESR < 2RSENSE Manufacturers such as Nichicon, United Chemicon and Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR(size) product of any aluminum electrolytic at a somewhat higher price. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. In surface mount applications multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum, available in case heights ranging from 2mm to 4mm. Other capacitor types include Sanyo OS-CON, Nichicon PL series and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. LTC1435 U W U U APPLICATIONS INFORMATION INTVCC Regulator An internal P-channel low dropout regulator produces the 5V supply which powers the drivers and internal circuitry within the LTC1435. The INTVCC pin can supply up to 15mA and must be bypassed to ground with a minimum of 2.2µF tantalum or low ESR electrolytic. Good bypassing is necessary to supply the high transient currents required by the MOSFET gate drivers. High input voltage applications, in which large MOSFETs are being driven at high frequencies, may cause the maximum junction temperature rating for the LTC1435 to be exceeded. The IC supply current is dominated by the gate charge supply current when not using an output derived EXTVCC source. The gate charge is dependent on operating frequency as discussed in the Efficiency Considerations section. The junction temperature can be estimated by using the equations given in Note 1 of the Electrical Characteristics. For example, the LTC1435 is limited to less than 17mA from a 30V supply: TJ = 70°C + (17mA)(30V)(100°C/W) = 126°C To prevent maximum junction temperature from being exceeded, the input supply current must be checked when operating in continuous mode at maximum VIN. EXTVCC Connection The LTC1435 contains an internal P-channel MOSFET switch connected between the EXTVCC and INTVCC pins. The switch closes and supplies the INTVCC power whenever the EXTVCC pin is above 4.8V, and remains closed until EXTVCC drops below 4.5V. This allows the MOSFET driver and control power to be derived from the output during normal operation (4.8V < VOUT < 9V) and from the internal regulator when the output is out of regulation (start-up, short circuit). Do not apply greater than 10V to the EXTVCC pin and ensure that EXTVCC < VIN. Significant efficiency gains can be realized by powering INTVCC from the output, since the VIN current resulting from the driver and control currents will be scaled by a factor of Duty Cycle/Efficiency. For 5V regulators this supply means connecting the EXTVCC pin directly to VOUT. However, for 3.3V and other lower voltage regulators, additional circuitry is required to derive INTVCC power from the output. The following list summarizes the four possible connections for EXTVCC: 1. EXTVCC left open (or grounded). This will cause INTVCC to be powered from the internal 5V regulator resulting in an efficiency penalty of up to 10% at high input voltages. 2. EXTVCC connected directly to VOUT. This is the normal connection for a 5V regulator and provides the highest efficiency. 3. EXTVCC connected to an output-derived boost network. For 3.3V and other low voltage regulators, efficiency gains can still be realized by connecting EXTVCC to an output-derived voltage which has been boosted to greater than 4.8V. This can be done with either the inductive boost winding as shown in Figure 4a or the capacitive charge pump shown in Figure 4b. The charge pump has the advantage of simple magnetics. 4. EXTVCC connected to an external supply. If an external supply is available in the 5V to 10V range (EXTVCC ≤ VIN), it may be used to power EXTVCC providing it is compatible with the MOSFET gate drive requirements. When driving standard threshold MOSFETs, the external supply must always be present during operation to prevent MOSFET failure due to insufficient gate drive. + VIN CIN 1N4148 VIN OPTIONAL EXT VCC CONNECTION 5V ≤ VSEC ≤ 9V TG N-CH VOUT COUT SW BG R5 SGND 1µF + LTC1435 SFB + RSENSE EXTVCC R6 L1 1:N VSEC N-CH PGND LTC1435 • F04a Figure 4a. Secondary Output Loop and EXTVCC Connection 11 LTC1435 U U W U APPLICATIONS INFORMATION + 1.19V ≤ VOUT ≤ 9V + VIN 1µF CIN BAT85 0.22µF R2 VOSENSE BAT85 TG SGND BAT85 N-CH EXTVCC R1 LTC1435 • F05 VN2222LL L1 RSENSE VOUT + LTC1435 Figure 5. Setting the LTC1435 Output Voltage COUT SW BG 100pF LTC1435 VIN 3.3V OR 5V N-CH RUN/SS RUN/SS D1 PGND CSS LTC1435 • F04b Figure 4b. Capacitive Charge Pump for EXTVCC CSS LTC1435 • F06 Figure 6. RUN/SS Pin Interfacing Topside MOSFET Driver Supply (CB, DB) An external bootstrap capacitor CB connected to the Boost pin supplies the gate drive voltage for the topside MOSFET. Capacitor CB in the Functional Diagram is charged through diode DB from INTVCC when the SW pin is low. When the topside MOSFET is to be turned on, the driver places the CB voltage across the gate source of the MOSFET. This enhances the MOSFET and turns on the topside switch. The switch node voltage SW rises to VIN and the Boost pin rises to VIN + INTVCC. The value of the boost capacitor CB needs to be 100 times greater than the total input capacitance of the topside MOSFET. In most applications 0.1µF is adequate. The reverse breakdown on DB must be greater than VIN(MAX). Output Voltage Programming The output voltage is set by a resistive divider according to the following formula: R2 VOUT = 1.19V 1 + R1 The external resistor divider is connected to the output as shown in Figure 5 allowing remote voltage sensing. Run/ Soft Start Function The RUN/SS pin is a dual purpose pin which provides the soft start function and a means to shut down the LTC1435. 12 Soft start reduces surge currents from VIN by gradually increasing the internal current limit. Power supply sequencing can also be accomplished using this pin. An internal 3µA current source charges up an external capacitor CSS. When the voltage on RUN/SS reaches 1.3V the LTC1435 begins operating. As the voltage on RUN/SS continues to ramp from 1.3V to 2.4V, the internal current limit is also ramped at a proportional linear rate. The current limit begins at approximately 50mV/RSENSE (at VRUN/SS = 1.3V) and ends at 150mV/RSENSE (VRUN/SS > 2.7V). The output current thus ramps up slowly, charging the output capacitor. If RUN/SS has been pulled all the way to ground there is a delay before starting of approximately 500ms/µF, followed by an additional 500ms/µF to reach full current. tDELAY = 5(10 5)CSS Seconds Pulling the RUN/SS pin below 1.3V puts the LTC1435 into a low quiescent current shutdown (IQ < 25µA). This pin can be driven directly from logic as shown in Figure 6. Diode D1 in Figure 6 reduces the start delay but allows CSS to ramp up slowly for the soft start function; this diode and CSS can be deleted if soft start is not needed. The RUN/SS pin has an internal 6V Zener clamp (See Functional Diagram). LTC1435 U W U U APPLICATIONS INFORMATION Foldback Current Limiting Efficiency Considerations As described in Power MOSFET and D1 Selection, the worst-case dissipation for either MOSFET occurs with a short-circuited output, when the synchronous MOSFET conducts the current limit value almost continuously. In most applications this will not cause excessive heating, even for extended fault intervals. However, when heat sinking is at a premium or higher RDS(ON) MOSFETs are being used, foldback current limiting should be added to reduce the current in proportion to the severity of the fault. The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Foldback current limiting is implemented by adding diode DFB between the output and the ITH pin as shown in the Functional Diagram. In a hard short (VOUT = 0V) the current will be reduced to approximately 25% of the maximum output current. This technique may be used for all applications with regulated output voltages of 1.8V or greater. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC1435 circuits. LTC1435 VIN current, INTVCC current, I2R losses, and topside MOSFET transition losses. SFB Pin Operation When the SFB pin drops below its ground referenced 1.19V threshold, continuous mode operation is forced. In continuous mode, the large N-channel main and synchronous switches are used regardless of the load on the main output. In addition to providing a logic input to force continuous synchronous operation, the SFB pin provides a means to regulate a flyback winding output. Continuous synchronous operation allows power to be drawn from the auxiliary windings without regard to the primary output load. The SFB pin provides a way to force continuous synchronous operation as needed by the flyback winding. The secondary output voltage is set by the turns ratio of the transformer in conjunction with a pair of external resistors returned to the SFB pin as shown in Figure 4a. The secondary regulated voltage, VSEC, in Figure 4a is given by: R6 VSEC ≈ (N + 1)VOUT > 1.19 1 + R5 where N is the turns ratio of the transformer and VOUT is the main output voltage sensed by VOSENSE. Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. 1. The VIN current is the DC supply current given in the electrical characteristics which excludes MOSFET driver and control currents. VIN current results in a small (< 1%) loss which increases with VIN. 2. INTVCC current is the sum of the MOSFET driver and control currents. The MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from INTVCC to ground. The resulting dQ/dt is a current out of INT VCC which is typically much larger than the control circuit current. In continuous mode, IGATECHG = f(QT + QB), where QT and QB are the gate charges of the topside and bottom side MOSFETs. By powering EXTVCC from an output-derived source, the additional VIN current resulting from the driver and control currents will be scaled by a factor of Duty Cycle/Efficiency. For example, in a 20V to 5V application, 10mA of INTVCC current results in approximately 3mA of VIN current. This reduces the midcurrent loss from 10% or more (if the driver was powered directly from VIN) to only a few percent. 3. I2R losses are predicted from the DC resistances of the MOSFET, inductor and current shunt. In continuous mode the average output current flows through L and RSENSE, but is “chopped” between the topside main 13 LTC1435 U W U U APPLICATIONS INFORMATION MOSFET and the synchronous MOSFET. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one MOSFET can simply be summed with the resistances of L and RSENSE to obtain I2R losses. For example, if each RDS(ON) = 0.05Ω, RL = 0.15Ω, and RSENSE = 0.05Ω, then the total resistance is 0.25Ω. This results in losses ranging from 3% to 10% as the output current increases from 0.5A to 2A. I2R losses cause the efficiency to drop at high output currents. 4. Transition losses apply only to the topside MOSFET(s), and only when operating at high input voltages (typically 20V or greater). Transition losses can be estimated from: Transition Loss = 2.5 (VIN)1.85(IMAX)(CRSS)(f) Other losses, including CIN and COUT ESR dissipative losses, Schottky conduction losses during dead-time, and inductor core losses, generally account for less than 2% total additional loss. Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in DC (resistive) load current. When a load step occurs, VOUT immediately shifts by an amount equal to (∆ILOAD)(ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT which generates a feedback error signal. The regulator loop then acts to return VOUT to its steady-state value. During this recovery time VOUT can be monitored for overshoot or ringing which would indicate a stability problem. The ITH external components shown in the Figure 1 circuit will provide adequate compensation for most applications. A second, more severe transient is caused by switching in loads with large (>1µF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The 14 only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25)(CLOAD). Thus a 10µF capacitor would require a 250µs rise time, limiting the charging current to about 200mA. Automotive Considerations: Plugging into the Cigarette Lighter As battery-powered devices go mobile, there is a natural interest in plugging into the cigarette lighter in order to conserve or even recharge battery packs during operation. But before you connect, be advised: you are plugging into the supply from hell. The main battery line in an automobile is the source of a number of nasty potential transients, including load dump, reverse battery and double battery. Load dump is the result of a loose battery cable. When the cable breaks connection, the field collapse in the alternator can cause a positive spike as high as 60V which takes several hundred milliseconds to decay. Reverse battery is just what it says, while double battery is a consequence of tow truck operators finding that a 24V jump start cranks cold engines faster than 12V. The network shown in Figure 7 is the most straightforward approach to protect a DC/DC converter from the ravages of an automotive battery line. The series diode prevents current from flowing during reverse battery, while the transient suppressor clamps the input voltage during load dump. Note that the transient suppressor should not conduct during double battery operation, but must still clamp the input voltage below breakdown of the converter. Although the LT1435 has a maximum input voltage of 36V, most applications will be limited to 30V by the MOSFET BVDSS. 12V 50A IPK RATING VIN TRANSIENT VOLTAGE SUPPRESSOR GENERAL INSTRUMENT 1.5KA24A LTC1435 1435 F07 Figure 7. Automotive Application Protection LTC1435 U W U U APPLICATIONS INFORMATION Design Example As a design example, assume VIN = 12V(nominal), VIN = 22V(max), VOUT = 3.3V, IMAX = 3A and f = 250kHz, RSENSE and COSC can immediately be calculated: RSENSE = 100mV/3A = 0.033Ω COSC = 1.37(104)/250 – 11 = 43pF Referring to Figure 3, a 10µH inductor falls within the recommended range. To check the actual value of the ripple current the following equation is used: V V ∆IL = OUT 1– OUT ( f)(L) VIN The highest value of the ripple current occurs at the maximum input voltage: ∆IL = 3.3V 3.3V 1– = 1.12A 250kHz(10µH) 22V The power dissipation on the topside MOSFET can be easily estimated. Choosing a Siliconix Si4412DY results in: RDS(ON) = 0.042Ω, CRSS = 100pF. At maximum input voltage with T(estimated) = 50°C: ( ) [ ( )( )]( ) 1.85 + 2.5 (22V ) (3A )(100pF )(250kHz) = 122mW PMAIN = 3.3V 2 3 1 + 0.005 50°C − 25°C 0.042Ω 22V The most stringent requirement for the synchronous N-channel MOSFET occurs when VOUT = 0 (i.e. short circuit). In this case the worst-case dissipation rises to: ( PSYNC = ISC( AVG) ) (1+ δ ) RDS(ON) 2 highest at the maximum input voltage. The output voltage ripple due to ESR is approximately: VORIPPLE = RESR(∆IL) = 0.03Ω(1.112A) = 34mVP-P PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1435. These items are also illustrated graphically in the layout diagram of Figure 8. Check the following in your layout: 1. Are the signal and power grounds segregated? The LTC1435 signal ground pin must return to the (–) plate of COUT. The power ground connects to the source of the bottom N-channel MOSFET, anode of the Schottky diode, and (–) plate of CIN, which should have as short lead lengths as possible. 2. Does the VOSENSE pin connect directly to the feedback resistors? The resistive divider R1, R2 must be connected between the (+) plate of COUT and signal ground. The 100pF capacitor should be as close as possible to the LTC1435. 3. Are the SENSE – and SENSE + leads routed together with minimum PC trace spacing? The filter capacitor between SENSE + and SENSE – should be as close as possible to the LTC1435. 4. Does the (+) plate of CIN connect to the drain of the topside MOSFET(s) as closely as possible? This capacitor provides the AC current to the MOSFET(s). 5. Is the INTVCC decoupling capacitor connected closely between INTVCC and the power ground pin? This capacitor carries the MOSFET driver peak currents. With the 0.033Ω sense resistor ISC(AVG) = 4A will result, increasing the Si4412DY dissipation to 950mW at a die temperature of 105°C. 6. Keep the switching node SW away from sensitive smallsignal nodes. Ideally the switch node should be placed at the furthest point from the LTC1435. CIN is chosen for an RMS current rating of at least 1.5A at temperature. COUT is chosen with an ESR of 0.03Ω for low output ripple. The output ripple in continuous mode will be 7. SGND should be exclusively used for grounding external components on COSC, ITH, VOSENSE and SFB pins. 15 LTC1435 U U W U APPLICATIONS INFORMATION + M1 1 CSS 2 TG RUN/SS BOOST 16 CIN 15 VIN CC1 RC COSC + COSC 3 CC2 4 5 ITH SW LTC1435 SFB VIN INTVCC SGND 14 13 VOSENSE BG 7 SENSE – PGND 8 SENSE + EXTVCC D1 12 – 100pF 6 CB 0.1µF DB 11 + M2 4.7µF 10 1000pF 9 L1 – R1 + R2 COUT VOUT RSENSE BOLD LINES INDICATE HIGH CURRENT PATHS + LTC1435 • F08 Figure 8. LTC1435 Layout Diagram U TYPICAL APPLICATIONS Dual Output 5V and Synchronous 12V Application VIN 5.4V TO 28V COSC 68pF + 1 CSS 0.1µF RC 10k CC1 470pF CC2 51pF 2 3 4 COSC TG RUN/SS BOOST ITH SW SFB LTC1435 5 VIN 7 BG VOSENSE SENSE – PGND SENSE + EXTVCC IRLL014 4.7k 14 T1 10µH 1:1.42 13 0.1µF 12 + 11 4.7µF M2 Si4412DY MBRS140T3 + RSENSE 0.033Ω CSEC 3.3µF 35V VOUT 5V/3.5A R1 35.7k 1% COUT 100µF 10V ×2 + 10 1000pF 8 0.01µF 15 100pF 6 M1 Si4412DY CMDSH-3 INTVCC SGND 16 CIN 22µF 35V ×2 9 100Ω R2 20k 1% SGND 100Ω 11.3k 1% 16 100k 1% T1: DALE LPE6562-A236 LTC1435 • TA04 VOUT2 12V 120mA LTC1435 U TYPICAL APPLICATIONS 3.3V/4.5A Converter with Foldback Current Limiting VIN 4.5V TO 28V COSC 68pF 1 CSS 0.1µF RC 10k 2 CC1 330pF CC2 51pF 3 4 INTVCC COSC TG RUN/SS LTC1435 5 14 SW SFB ITH PIN 3 13 VIN IN4148 7 SENSE – PGND SENSE + EXTVCC M2 Si4410DY 9 + MBRS140T3 10 100pF R2 20k 1% 1000pF 8 VOUT 3.3V/4.5A R1 35.7k 1% 4.7µF 11 BG VOSENSE RSENSE 0.025Ω + 100pF 6 L1 10µH 0.1µF CMDSH-3 12 INTVCC SGND M1 Si4410DY 15 BOOST ITH CIN 22µF 35V ×2 + 16 OPTIONAL: CONNECT TO 5V COUT 100µF 10V ×2 SGND (PIN 5) LTC1435 • TA01 Dual Output 5V and 12V Application VIN 5.4V TO 28V COSC 68pF + 1 CSS 0.1µF RC 10k CC1 510pF CC2 51pF 2 3 4 COSC TG RUN/SS BOOST ITH SW SFB LTC1435 5 VIN 15 7 BG VOSENSE SENSE – PGND SENSE + EXTVCC 100Ω 24V T1 10µH 1:2.2 13 + 0.1µF CSEC 3.3µF 25V VOUT 5V/3.5A 12 + 11 RSENSE 0.033Ω 4.7µF M2 IRF7403 MBRS140T3 R1 35.7k 1% COUT 100µF 10V ×2 + 10 1000pF 8 MBRS1100T3 14 100pF 6 M1 IRF7403 CMDSH-3 INTVCC SGND 16 CIN 22µF 35V ×2 9 R2 20k 1% SGND 100Ω 10k 90.9k T1: DALE LPE6562-A092 LTC1435 • TA02 VOUT2 12V 17 LTC1435 U TYPICAL APPLICATIONS Constant-Current/Constant-Voltage High Efficiency Battery Charger E1 VIN + C1* 22µF 35V E3 GND E3 SHDN + C2* 22µF 35V 1 2 3 C14 1000pF 4 5 C9 100pF U2 LT1620 1 2 3 4 AVG SENSE PROG GND VCC NIN 6 7 8 C15 0.1µF IOUT PIN C4 0.1µF C11 56pF C12 0.1µF C13 0.033µF R5 1k R7 1.5M COSC TG RUN/SS BOOST C5 0.1µF 16 Q1 Si4412DY 15 D1 14 U1 SW LTC1435 13 VIN SFB 12 SGND INTVCC 11 VOSENSE BG 10 SENSE – PGND 9 + SENSE EXTVCC ITH C3 22µF 35V Q2 Si4412DY + 7 C8 100pF C7 4.7µF 16V 6 R2 1M 0.1% 5 R3 105k 0.1% JP1A C16 0.33µF R6 10k 1% C17 0.01µF R1 0.025Ω + C6 0.33µF D2 C10 100pF 8 L1 27µH C18 0.1µF R4 76.8k 0.1% JP1B DC133 F01 E5 GND RPROG E4 IPROG *CONSULT CAPACITOR MANUFACTURER FOR RECOMMENDED ESR RATING FOR CONTINUOUS 4A OPERATION Current Programming Equation )(R6) – 0.04 (I IBATT = PROG 10(R1) Efficiency 100 VIN = 24V VBATT = 16V 95 EFFICIENCY (%) VBATT = 12V 90 VBATT = 6V 85 80 75 0 1 3 4 2 BATTERY CHARGE CURRENT (A) 5 1435 TA05 18 E6 BATT E7 GND LTC1435 U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. G Package 16-Lead Plastic SSOP (0.209) (LTC DWG # 05-08-1640) 0.239 – 0.249* (6.07 – 7.33) 16 15 14 13 12 11 10 9 0.205 – 0.212** (5.20 – 5.38) 0.068 – 0.078 (1.73 – 1.99) 0.301 – 0.311 (7.65 – 7.90) 0° – 8° 0.0256 (0.65) BSC 0.022 – 0.037 (0.55 – 0.95) 0.005 – 0.009 (0.13 – 0.22) *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 0.002 – 0.008 (0.05 – 0.21) 0.010 – 0.015 (0.25 – 0.38) 1 2 3 4 5 6 7 8 G16 SSOP 0795 S Package 16-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.386 – 0.394* (9.804 – 10.008) 16 15 14 13 12 11 10 9 0.150 – 0.157** (3.810 – 3.988) 0.228 – 0.244 (5.791 – 6.197) 1 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 2 3 4 5 6 0.053 – 0.069 (1.346 – 1.752) 0.014 – 0.019 (0.355 – 0.483) 8 0.004 – 0.010 (0.101 – 0.254) 0° – 8° TYP 0.016 – 0.050 0.406 – 1.270 7 0.050 (1.270) TYP S16 0695 *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 19 LTC1435 U TYPICAL APPLICATION Low Dropout 2.9V/3A Converter VIN 3.5V TO 25V COSC 68pF 1 CSS 0.1µF RC 10k 2 CC1 330pF CC2 51pF INTVCC 3 4 COSC TG RUN/SS BOOST ITH SW SFB LTC1435 5 SGND VIN 16 14 13 CMDSH-3 INTVCC 7 VOSENSE SENSE – BG PGND CIN 22µF 35V ×2 + 15 L1 10µH 0.1µF 12 RSENSE 0.033Ω VOUT 2.9V/3A + 100pF 6 M1 1/2 Si9925DY 11 4.7µF M2 1/2 Si9925DY 100pF MBRS140T3 SENSE + EXTVCC + 10 R2 24.9k 1% 1000pF 8 R1 35.7k 1% 9 OPTIONAL: CONNECT TO 5V COUT 100µF 10V ×2 SGND LTC1435 • TA03 L1: SUMIDA CDRH125-10 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1142HV/LTC1142 Dual High Efficiency Synchronous Step-Down Switching Regulators Dual Synchronous, VIN ≤ 20V LTC1148HV/LTC1148 High Efficiency Sychronous Step-Down Switching Regulator Controllers Synchronous, VIN ≤ 20V LTC1159 High Efficiency Synchronous Step-Down Switching Regulator Synchronous, VIN ≤ 40V, For Logic Threshold FETs LT®1375/LT1376 1.5A, 500kHz Step-Down Switching Regulators High Frequency, Small Inductor, High Efficiency Switchers, 1.5A Switch LTC1430 High Power Step-Down Switching Regulator Controller High Efficiency 5V to 3.3V Conversion at Up to 15A LTC1436/LTC1436-PLL/ High Efficiency Low Noise Synchronous Step-Down LTC1437 Switching Regulators Full-Featured Single Controller LTC1438/LTC1439 Dual High Efficiency, Low Noise, Synchronous Step-Down Switching Regulators Full-Featured Dual Controllers LT1510 Constant-Voltage/ Constant-Current Battery Charger 1.3A, Li-Ion, NiCd, NiMH, Pb-Acid Charger LTC1538-AUX Dual High Efficiency, Low Noise, Synchronous Step-Down Switching Regulator 5V Standby in Shutdown LTC1539 Dual High Efficiency, Low Noise, Synchronous Step-Down Switching Regulator 5V Standby in Shutdown 20 Linear Technology Corporation LT/GP 0896 7K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● TELEX: 499-3977 LINEAR TECHNOLOGY CORPORATION 1996