Microchip MCP16311-E/MS 30v input, 1a output, high-efficiency, integrated synchronous switch step-down regulator Datasheet

MCP16311/2
30V Input, 1A Output, High-Efficiency,
Integrated Synchronous Switch Step-Down Regulator
Features
General Description
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The MCP16311/2 is a compact, high-efficiency, fixed
frequency, synchronous step-down DC-DC converter in
an 8-pin MSOP, or 2 x 3 TDFN package that operates
from input voltage sources up to 30V. Integrated
features include a high-side and a low-side switch, fixed
frequency peak current mode control, internal
compensation, peak current limit and overtemperature
protection. The MCP16311/2 provides all the active
functions for local DC-DC conversion, with fast transient
response and accurate regulation.
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Up to 95% Efficiency
Input Voltage Range: 4.4V to 30V
1A Output Current Capability
Output Voltage Range: 2.0V to 24V
Qualification: AEC-Q100 Rev. G, Grade 1 (-40°C
to 125°C)
Integrated N-Channel High-Side and Low-Side
Switches:
- 170 m, Low Side
- 300 m, High Side
Stable Reference Voltage: 0.8V
Automatic Pulse Frequency Modulation/PulseWidth Modulation (PFM/PWM) Operation
(MCP16311):
- PFM Operation Disabled (MCP16312)
- PWM Operation: 500 kHz
Low Device Shutdown Current: 3 µA typical
Low Device Quiescent Current:
- 44 µA (non-switching, PFM Mode)
Internal Compensation
Internal Soft-Start: 300 µs (EN low-to-high)
Peak Current Mode Control
Cycle-by-Cycle Peak Current Limit
Undervoltage Lockout (UVLO):
- 4.1V typical to start
- 3.6V typical to stop
Overtemperature Protection
Thermal Shutdown:
- +150°C
- +25°C Hysteresis
Applications
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PIC®/dsPIC® Microcontroller Bias Supply
24V Industrial Input DC-DC Conversion
General Purpose DC-DC Conversion
Local Point of Load Regulation
Automotive Battery Regulation
Set-Top Boxes
Cable Modems
Wall Transformer Regulation
Laptop Computers
Networking Systems
AC-DC Digital Control Bias
Distributed Power Supplies
 2013-2014 Microchip Technology Inc.
High converter efficiency is achieved by integrating the
current-limited, low-resistance, high-speed high-side
and low-side switches and associated drive circuitry.
The MCP16311 is capable of running in PWM/PFM
mode. It switches in PFM mode for light load
conditions and for large buck conversion ratios. This
results in a higher efficiency over all load ranges. The
MCP16312 runs in PWM-only mode, and is
recommended for noise-sensitive applications.
The MCP16311/2 can supply up to 1A of continuous
current while regulating the output voltage from 2V to
12V. An integrated, high-performance peak current
mode architecture keeps the output voltage tightly
regulated, even during input voltage steps and output
current transient conditions common in power systems.
The EN input is used to turn the device on and off.
While off, only a few micro amps of current are
consumed from the input.
Output voltage is set with an external resistor divider.
The MCP16311/2 is offered in small MSOP-8 and 2 x 3
TDFN surface mount packages.
Package Type
MCP16311/2
2x3 TDFN*
MCP16311/2
MSOP
VFB 1
VCC 2
EN 3
VIN 4
8
7
6
5
VFB
AGND
BOOST VCC
SW
EN
PGND
VIN
1
2
3
4
8 AGND
EP
9
7 BOOST
6 SW
5 PGND
* Includes Exposed Thermal Pad (EP); see Table 3-1.
DS20005255B-page 1
MCP16311/2
Typical Applications
VIN
4.5V to 30V
CBOOST L1
100 nF 15 µH
BOOST
SW
VIN
CIN
2 x 10 µF
CVCC
1 µF
Vin
6V to 30V
31.6 k
VFB
VCC
GND
10 k
VIN
CVCC
1 µF
COUT
2 x 10 µF
EN
CBOOST L1
100 nF 22 µH
BOOST
CIN
2 x 10 µF
VOUT
3.3V @ 1A
VOUT
5V, @ 1A
SW
COUT
2 x 10 µF
EN
52.3 k
VFB
VCC
GND
10 k
100
V OUT = 5V
90
VOUT = 3.3V
Efficiency (%)
80
70
60
50
40
30
VIN = 12V
PWM ONLY
PWM/PFM
20
10
0
1
10
100
1000
IOUT (mA)
DS20005255B-page 2
 2013-2014 Microchip Technology Inc.
MCP16311/2
1.0
ELECTRICAL
CHARACTERISTICS
Absolute Maximum Ratings †
VIN, SW ............................................................... -0.5V to 32V
BOOST – GND ................................................... -0.5V to 38V
BOOST – SW Voltage........................................ -0.5V to 6.0V
VFB Voltage ........................................................ -0.5V to 6.0V
EN Voltage ............................................. -0.5V to (VIN + 0.3V)
Output Short-Circuit Current ................................. Continuous
Power Dissipation ....................................... Internally Limited
Storage Temperature ....................................-65°C to +150°C
Ambient Temperature with Power Applied ....-40°C to +125°C
Operating Junction Temperature...................-40°C to +150°C
ESD Protection on All Pins:
HBM ..................................................................... 1 kV
MM ......................................................................200V
† Notice: Stresses above those listed under “Maximum
Ratings” may cause permanent damage to the device.
This is a stress rating only and functional operation of
the device at those or any other conditions above those
indicated in the operational sections of this
specification is not intended. Exposure to maximum
rating conditions for extended periods may affect
device reliability.
DC CHARACTERISTICS
Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VIN = VEN = 7V, VBOOST - VSW = 5.0V,
VOUT = 5.0V, IOUT = 100 mA, L = 22 µH, COUT = CIN = 2 x 10 µF X7R Ceramic Capacitors.
Boldface specifications apply over the TA range of -40°C to +125°C.
Parameters
Sym.
Min.
Typ.
Max.
Units
Conditions
Input Voltage
VIN
4.4
—
30
V
Note 1
Quiescent Current
IQ
—
44
60
µA
Nonswitching,
VFB = 0.9V
Quiescent Current PFM Mode
IQ_PFM
—
85
—
µA
Switching,
IOUT = 0 (MCP16311)
Quiescent Current PWM Mode
IQ_PWM
—
3.8
8
mA
Switching,
IOUT = 0 (MCP16312)
Quiescent Current Shutdown
IQ_SHDN
—
3
9
µA
VOUT = EN = 0V
VIN Supply Voltage
VIN Undervoltage Lockout
Undervoltage Lockout Start
UVLOSTRT
—
4.1
4.4
V
VIN Rising
Undervoltage Lockout Stop
UVLOSTOP
3.18
3.6
—
V
VIN Falling
Undervoltage Lockout
Hysteresis
UVLOHYS
0.2
0.5
1
V
VFB
0.784
0.800
0.816
V
IOUT = 5 mA
VOUT
2.0
—
24
V
Note 2, Note 3
Feedback Voltage
Line Regulation
VFB/VFB)/VIN
-0.15
0.01
0.15
%/V
VIN = 7V to 30V,
IOUT = 50 mA
Feedback Voltage
Load Regulation
VFB / VFB
—
0.25
—
%
Output Characteristics
Feedback Voltage
Output Voltage
Adjust Range
Note 1:
2:
3:
4:
IOUT = 5 mA to 1A,
MCP16312
The input voltage should be greater than the output voltage plus headroom voltage; higher load currents
increase the input voltage necessary for regulation. See characterization graphs for typical input-to-output
operating voltage range.
For VIN < VOUT, VOUT will not remain in regulation; for output voltages above 12V, the maximum current
will be limited to under 1A.
Determined by characterization, not production tested.
This is ensured by design.
 2013-2014 Microchip Technology Inc.
DS20005255B-page 3
MCP16311/2
DC CHARACTERISTICS (CONTINUED)
Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VIN = VEN = 7V, VBOOST - VSW = 5.0V,
VOUT = 5.0V, IOUT = 100 mA, L = 22 µH, COUT = CIN = 2 x 10 µF X7R Ceramic Capacitors.
Boldface specifications apply over the TA range of -40°C to +125°C.
Parameters
Sym.
Min.
Typ.
Max.
Units
Conditions
Feedback Input
Bias Current
IFB
—
10
250
nA
Output Current
IOUT
1
—
—
A
Switching Frequency
fSW
425
500
575
kHz
Maximum Duty Cycle
DCMAX
85
94
—
%
Note 3
Minimum Duty Cycle
DCMIN
—
2
—
%
Note 4
RDS(ON)
—
0.3
—

VBOOST – VSW = 5V,
Note 3
I(MAX)
—
1.8
—
A
VBOOST – VSW = 5V,
Note 3
RDS(ON)
—
0.17
—

Note 3
EN Input Logic High
VIH
1.85
—
—
V
EN Input Logic Low
VIL
—
—
0.4
V
IENLK
—
0.1
1
µA
VEN = 5V
tSS
—
300
—
µs
EN Low-to-High,
90% of VOUT
TSD
—
150
—
°C
Note 3
TSDHYS
—
25
—
°C
Note 3
Notes 1 to 3, Figure 2-7
Switching Characteristics
High-Side NMOS Switch-On
Resistance
Buck NMOS Switch
Current Limit
Synchronous NMOS SwitchOn Resistance
EN Input Characteristics
EN Input Leakage Current
Soft-Start Time
Thermal Characteristics
Thermal Shutdown
Die Temperature
Die Temperature Hysteresis
Note 1:
2:
3:
4:
The input voltage should be greater than the output voltage plus headroom voltage; higher load currents
increase the input voltage necessary for regulation. See characterization graphs for typical input-to-output
operating voltage range.
For VIN < VOUT, VOUT will not remain in regulation; for output voltages above 12V, the maximum current
will be limited to under 1A.
Determined by characterization, not production tested.
This is ensured by design.
TEMPERATURE CHARACTERISTICS
Electrical Specifications: Unless otherwise indicated, TA = +25°C, VIN = VEN = 7V, VBOOST - VSW = 5.0V,
VOUT = 5.0V.
Parameters
Sym.
Min.
Typ.
Max.
Units
Conditions
Operating Junction Temperature Range
TJ
-40
—
+125
°C
Storage Temperature Range
TA
-65
—
+150
°C
Maximum Junction Temperature
TJ
—
—
+150
°C
Thermal Resistance, 8L-MSOP
JA
—
211
—
°C/W
EIA/JESD51-3 Standard
Thermal Resistance, 8L-2x3 TDFN
JA
—
52.5
—
°C/W
EIA/JESD51-3 Standard
Temperature Ranges
Steady State
Transient
Package Thermal Resistances
DS20005255B-page 4
 2013-2014 Microchip Technology Inc.
MCP16311/2
2.0
TYPICAL PERFORMANCE CURVES
The graphs and tables provided following this note are a statistical summary based on a limited number of
samples and are provided for informational purposes only. The performance characteristics listed herein
are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified
operating range (e.g., outside specified power supply range) and therefore outside the warranted range.
Note:
Note: Unless otherwise indicated, VIN = EN = 7V, COUT = CIN = 2 x 10 µF, L = 22 µH, VOUT = 5.0V, ILOAD = 100 mA,
TA = +25°C, 8L-MSOP package.
100
100
VIN = 6V
VIN = 12V
90
IOUT = 800 mA
80
70
VIN = 24V
60
VIN = 30V
50
40
30
Efficiency (%)
Efficiency (%)
80
20
10
100
40
PWM/PFM option
0
1
0
1000
5
IOUT (mA)
FIGURE 2-1:
IOUT.
3.3V VOUT Efficiency vs.
15
VIN (V)
20
25
30
3.3V VOUT Efficiency vs.VIN.
FIGURE 2-4:
IOUT = 800 mA
90
VIN = 12V
80
70
Efficiency (%)
80
Efficiency (%)
10
100
100
VIN = 24V
60
VIN = 30V
50
40
30
20
IOUT = 200 mA
IOUT = 10 mA
60
40
20
PWM/PFM
PWM ONLY
10
PWM/PFM option
0
0
1
10
100
1000
6
10
14
IOUT (mA)
FIGURE 2-2:
IOUT.
5.0V VOUT Efficiency vs.
18
VIN (V)
22
26
30
5.0V VOUT Efficiency vs.VIN.
FIGURE 2-5:
100
100
90
VIN = 15V
80
70
60
VIN = 24V
50
IOUT = 800 mA
80
VIN = 30V
Efficiency (%)
Efficiency (%)
IOUT = 10 mA
20
PWM/PFM
PWM ONLY
10
IOUT = 200 mA
60
40
IOUT = 200 mA
60
IOUT = 10 mA
40
30
20
20
PWM/PFM
PWM ONLY
10
0
1
FIGURE 2-3:
IOUT.
10
IOUT (mA)
100
1000
12.0V VOUT Efficiency vs.
 2013-2014 Microchip Technology Inc.
PWM/PFM option
0
12
14
FIGURE 2-6:
VIN.
16
18
20 22
VIN (V)
24
26
28
30
12.0V VOUT Efficiency vs.
DS20005255B-page 5
MCP16311/2
Note: Unless otherwise indicated, VIN = EN = 7V, COUT = CIN = 2 x 10 µF, L = 22 µH, VOUT = 5.0V, ILOAD = 100 mA,
TA = +25°C, 8L-MSOP package.
1600
5
VOUT = 3.3V
1400
Input Voltage (V)
VOUT = 5V
1200
IOUT (mA)
1000
VOUT = 12V
800
600
400
4.6
UVLO START
4.2
3.8
UVLO STOP
3.4
200
3
0
0
5
10
FIGURE 2-7:
15
VIN (V)
20
25
-40 -25 -10
30
Max IOUT vs.VIN.
FIGURE 2-10:
Temperature.
0.798
VIN =7V
VOUT = 3.3V
IOUT = 100 mA
0.796
0.794
0.792
Enable Voltage (V)
Feedback Voltage (V)
20 35 50 65 80 95 110 125
Temperature (°C)
Undervoltage Lockout vs.
1.4
0.8
VIN = 12V
VOUT = 3.3V
IOUT = 200 mA
1.3
1.2
HIGH
1.1
LOW
1
0.79
-40 -25 -10 5
FIGURE 2-8:
VOUT = 3.3V.
0.9
20 35 50 65 80 95 110 125
Temperature (°C)
VFB vs. Temperature;
-40 -25 -10
5
FIGURE 2-11:
vs. Temperature.
0.5
20 35 50 65 80 95 110 125
Temperature (°C)
Enable Threshold Voltage
5.03
0.45
5.02
0.4
0.35
High Side
0.3
0.25
0.2
Low Side
0.15
VIN = 12V
VOUT = 5V
IOUT = 500 mA
0.1
0.05
0
Output Voltage (V)
Switch RDSON (:)
5
VIN = 12V
VOUT = 5V
IOUT = 100 mA
5.01
5
4.99
4.98
4.97
-40 -25 -10
5
FIGURE 2-9:
Temperature.
DS20005255B-page 6
20 35 50 65 80 95 110 125
Temperature (°C)
Switch RDSON vs.
-40 -25 -10
FIGURE 2-12:
5
20 35 50 65 80 95 110 125
Temperature (°C)
VOUT vs. Temperature.
 2013-2014 Microchip Technology Inc.
MCP16311/2
Note: Unless otherwise indicated, VIN = EN = 7V, COUT = CIN = 2 x 10 µF, L = 22 µH, VOUT = 5.0V, ILOAD = 100 mA,
TA = +25°C, 8L-MSOP package.
1.8
VIN = 12V
VOUT = 5V
VOUT = 3.3V
40
Input Current (mA)
Quiescent Current (μA)
60
Non-Swithcing
20
1.6
1.4
1.2
Shutdown
0
1
-40 -25 -10
FIGURE 2-13:
Temperature.
5
20 35 50 65 80
Temperature (°C)
95 110 125
Input Quiescent Current vs.
5
15
20
VIN (V)
30
Output Current (mA)
150
Non-Switching
40
VOUT = 3.3V
30
20
10
Shutdown
5
10
15
20
125
VOUT = 3.3V
100
VOUT = 5V
75
50
25
0
25
VOUT = 12V
0
30
5
10
15
Input Voltage (°C)
FIGURE 2-14:
Input Voltage.
20
25
30
VIN (V)
Input Quiescent Current vs.
FIGURE 2-17:
vs. VIN.
PFM/PWM IOUT Threshold
50
120
VOUT = 3.3V
Output Current (mA)
No Load Input Current (μA)
25
FIGURE 2-16:
PWM No Load Input Current
vs.VIN, MCP16312.
50
Quiescent Current (μA)
10
100
80
60
40
VOUT = 3.3V
30
20
VOUT = 5V
10
VOUT = 12V
0
40
5
10
15
20
Input Voltage (V)
25
30
FIGURE 2-15:
PFM No Load Input Current
vs. Input Voltage, MCP16311.
 2013-2014 Microchip Technology Inc.
5
10
15
20
25
30
VIN (V)
FIGURE 2-18:
Skipping/PWM IOUT
Threshold vs. Input Voltage.
DS20005255B-page 7
MCP16311/2
Note: Unless otherwise indicated, VIN = EN = 7V, COUT = CIN = 2 x 10 µF, L = 22 µH, VOUT = 5.0V, ILOAD = 100 mA,
TA = +25°C, 8L-MSOP package.
4.5
VIN (V)
VOUT = 3.3V
To Start
VOUT
2 V/div
4
VIN
5 V/div
To Stop
3.5
0
200
400
600
800
Output Current (mA)
1000
FIGURE 2-19:
Typical Minimum Input
Voltage vs. Output Current.
200 µs/div
Start-Up From VIN.
FIGURE 2-22:
Switching Frequency (kHz)
525
VOUT
2 V/div
500
IL
500 mA/div
475
IOUT
2 A/div
VIN = 12V
VOUT = 3.3V
IOUT = 200 mA
450
-40 -25 -10
FIGURE 2-20:
Temperature.
5
20 35 50 65 80 95 110 125
Temperature (°C)
Switching Frequency vs.
10 µs/div
FIGURE 2-23:
Short-Circuit Response.
Load Step from
100 mA to 800 mA
VOUT
2 V/div
IOUT
500 mA/div
EN
2 V/div
VOUT
100 mV/div
AC Coupled
80 µs/div
FIGURE 2-21:
DS20005255B-page 8
Start-Up From Enable.
200 µs/div
FIGURE 2-24:
Load Transient Response.
 2013-2014 Microchip Technology Inc.
MCP16311/2
Note: Unless otherwise indicated, VIN = EN = 7V, COUT = CIN = 2 x 10 µF, L = 22 µH, VOUT = 5.0V, ILOAD = 100 mA,
TA = +25°C, 8L-MSOP package.
VOUT
50 mV/div
AC Coupled
VIN = 12V
VOUT = 5V
IOUT = 800 mA
IL
200 mA/div
VIN Step from 7V to 12V
SW
10 V/div
VIN
5 V/div
VOUT
50 mV/div
AC Coupled
400 µs/div
FIGURE 2-25:
Line Transient Response.
FIGURE 2-28:
Waveforms.
2 µs/div
Heavy Load Switching
VIN = 24V
IOUT = 25 mA
SW
10 V/div
IL
200 mA/div
VOUT
100 mV/div
AC Coupled
FIGURE 2-26:
Waveforms.
SW
10 V/div
20 µs/div
PFM Light Load Switching
VOUT
100 mV/div
AC Coupled
VIN = 12V
VOUT = 5V
Load Current
50 mA/div
SW
5 V/div
400 µs/div
FIGURE 2-29:
PFM to PWM Transition;
Load Step from 5 mA to 100 mA.
VIN = 24V
IOUT = 15 mA
IL
100 mA/div
VOUT
10 mV/div
AC Coupled 1 µs/div
FIGURE 2-27:
Waveforms.
PWM Light Load Switching
 2013-2014 Microchip Technology Inc.
DS20005255B-page 9
MCP16311/2
NOTES:
DS20005255B-page 10
 2013-2014 Microchip Technology Inc.
MCP16311/2
3.0
PIN DESCRIPTIONS
The descriptions of the pins are listed in Table 3-1.
TABLE 3-1:
PIN FUNCTION TABLE
MCP16311/2
2 x 3 TDFN
MCP16311/2
MSOP
Symbol
Description
1
1
VFB
Output Voltage Feedback pin. Connect VFB to an external resistor
divider to set the output voltage.
2
2
VCC
Internal Regulator Output pin. Bypass Capacitor is required on this
pin to provide high peak current for gate drive.
3
3
EN
Enable pin. Logic high enables the operation. Do not allow this pin to
float.
4
4
VIN
Input Supply Voltage pin for power and internal biasing.
5
5
PGND
6
6
SW
Output Switch Node pin, connects to the inductor and the bootstrap
capacitor.
7
7
BOOST
8
8
AGND
Boost Voltage pin that supplies the driver used to control the highside NMOS switch. A bootstrap capacitor is connected between the
BOOST and SW pins.
Signal Ground pin
9
—
EP
3.1
Power Ground pin
Exposed thermal pad
Feedback Voltage Pin (VFB)
The VFB pin is used to provide output voltage regulation
by using a resistor divider. The VFB voltage will be
0.800V typical with the output voltage in regulation.
3.2
Internal Bias Pin (VCC)
The VCC internal bias is derived from the input voltage
VIN. VCC is set to 5.0V typical. The VCC is used to provide a stable low bias voltage for the upper and lower
gate drive circuits. This output should be decoupled to
AGND with a 1 µF capacitor, X7R. This capacitor should
be connected as close as possible to the VCC and
AGND pin.
3.3
Enable Pin (EN)
The EN pin is a logic-level input used to enable or
disable the device and lower the quiescent current
while disabled. A logic high (> 1.3V) will enable the regulator output. A logic low (< 1V) will ensure that the regulator is disabled.
3.4
Power Supply Input Voltage Pin
(VIN)
Connect the input voltage source to VIN. The input
source should be decoupled to GND with a
4.7 µF-20 µF capacitor, depending on the impedance
of the source and output current. The input capacitor
provides current for the switch node and a stable voltage source for the internal device power. This capacitor
should be connected as close as possible to the VIN
and GND pins. For light-load applications, a 2.2 µF
X7R or X5R ceramic capacitor can be used.
 2013-2014 Microchip Technology Inc.
3.5
Analog Ground Pin (AGND)
This ground is used by most internal circuits, such as
the analog reference, control loop and other circuits.
3.6
Power Ground Pin (PGND)
This is a separate ground connection used for the lowside synchronous switch.The length of the trace from
the input cap return, output cap return and GND pin
should be made as short as possible to minimize the
noise in the system. The power ground and the analog
ground should be connected in a single point.
3.7
Switch Node Pin (SW)
The switch node pin is connected internally to the lowside and high-side switch, and externally to the SW
node, consisting of the inductor and boost capacitor.
The SW node can rise very fast as a result of the
internal switch turning on.
3.8
Boost Pin (BOOST)
The high side of the floating supply used to turn the
integrated N-Channel high-side MOSFET on and off is
connected to the boost pin.
3.9
Exposed Thermal Pad Pin (EP)
There is an internal electrical connection between the
EP and the PGND and AGND pins.
DS20005255B-page 11
MCP16311/2
NOTES:
DS20005255B-page 12
 2013-2014 Microchip Technology Inc.
MCP16311/2
4.0
DETAILED DESCRIPTION
4.1.3
4.1
Device Overview
An integrated precise 0.8V reference combined with an
external resistor divider sets the desired converter
output voltage. The resistor divider range can vary
without affecting the control system gain. High-value
resistors consume less current, but are more
susceptible to noise. Consult typical applications for the
recommended resistors value.
The MCP16311/2 is a high input voltage step-down
regulator, capable of supplying 1A typical to a regulated
output voltage from 2.0V to 12V. Internally, the trimmed
500 kHz oscillator provides a fixed frequency, while the
peak current mode control architecture varies the duty
cycle for output voltage regulation. An internal floating
driver is used to turn the high-side integrated
N-Channel MOSFET on and off. The power for this
driver is derived from an external boost capacitor
whose energy is replenished when the low-side NChannel MOSFET is turned on.
4.1.1
PWM/PFM MODE OPTION
The MCP16311 selects the best operating switching
mode (PFM or PWM) for high efficiency across a wide
range of load currents. Switching to PFM mode at lightload currents results in a low quiescent current. During
the sleep period (between two packets of switching
cycles), the MCP16311 draws 44 µA (typical) from the
supply line. The switching pulse packets represent a
small percentage of the total running cycle, and the
overall average current drawn from power line is small.
The disadvantages of PWM/PFM mode are higher
output ripple voltage and variable PFM mode frequency.
The PFM mode threshold is a function of the input
voltage, output voltage and load (see Figure 2-17).
4.1.2
PWM-ONLY MODE OPTION
In the MCP16312 devices, the PFM mode is disabled
and the part runs only in PWM over the entire load
range. During normal operation, the MCP16312
continues to operate at a constant 500 kHz switching
frequency, keeping the output ripple voltage lower than
in PFM mode. At lighter loads, the MCP16312 devices
begin to skip pulses. Figure 2-18 represents the input
voltage versus load current for the pulse skipping
threshold in PWM-only mode.
Because the MCP16312 has very low output voltage
ripple, it is recommended for noise-sensitive applications.
TABLE 4-1:
Part Number
PART NUMBER SELECTION
PWM/PFM
PWM
MCP16311
X
—
MCP16312
—
X
4.1.4
INTERNAL REFERENCE VOLTAGE
(VFB)
INTERNAL BIAS REGULATOR (VCC)
An internal Low Dropout Voltage Regulator (LDO) is
used to supply 5.0V to all the internal circuits. The LDO
regulates the input voltage (VIN) and can supply
enough current (up to 50 mA) to sustain the drivers and
internal bias circuitry. The VCC pin must be decoupled
to ground with a 1 µF capacitor. In the event of a
thermal shut down, the LDO will shut down. There is a
short-circuit protection for the VCC pin, with a threshold
set at 150 mA.
In PFM switching mode, during sleep periods, the VCC
regulator enters Low Quiescent Current mode to avoid
unnecessary power dissipation.
Avoid driving any external load using the VCC pin.
4.1.5
INTERNAL COMPENSATION
All control system components necessary for stable
operation over the entire device operating range are
integrated, including the error amplifier and inductor
current slope compensation. To add the proper amount
of slope compensation, the inductor value changes
along with the output voltage (see Table 5-1).
4.1.6
EXTERNAL COMPONENTS
External components consist of:
•
•
•
•
Input capacitor
Output filter (inductor and capacitor)
Boost capacitor
Resistor divider
The selection of the external inductor, output capacitor
and input capacitor is dependent upon the output voltage and the maximum output current.
4.1.7
ENABLE INPUT
The enable input (EN) is used to disable the device. If
disabled, the device consumes a minimum current from
the input. Once enabled, the internal soft start controls
the output voltage rate of rise, preventing high-inrush
current and output voltage overshoot.
There is no internal pull-up or pull-down resistor. To
enable the converter, the EN pin must be pulled high.
To disable the converter, the EN pin must be pulled low.
 2013-2014 Microchip Technology Inc.
DS20005255B-page 13
MCP16311/2
4.1.8
SOFT START
The internal reference voltage rate of rise is controlled
during start-up, minimizing the output voltage
overshoot and the inrush current.
4.1.9
UNDERVOLTAGE LOCKOUT
An integrated Undervoltage Lockout (UVLO) prevents
the converter from starting until the input voltage is high
enough for normal operation. The converter will
typically start at 4.1V and operate down to 3.6V.
Hysteresis is added to prevent starting and stopping
during start-up as a result of loading the input voltage
source.
4.1.10
OVERTEMPERATURE
PROTECTION
Overtemperature protection limits the silicon die
temperature to +150°C by turning the converter off. The
normal switching resumes at +125°C.
VREG
VIN
VCC
VCC
C VCC
BG
REF
CIN
BOOST
VOUT
SS OTEMP
VREF
RTOP
+
Amp
-
FB
RBOT
RCOMP
VREF CCOMP
CBOOST
500 kHz OSC
VOUT
S
Comp
+
PWM
Latch
HS
Drive
SW
COUT
R
UVLO
Overtemp
CS
PFM
RSENSE
PFM
CTR
+
+
VREF
EN
+
-
VCC
Slope
Comp
LS
Drive
SHDN all blocks
AGND
FIGURE 4-1:
DS20005255B-page 14
PGND
MCP16311/2 Block Diagram.
 2013-2014 Microchip Technology Inc.
MCP16311/2
4.2
Functional Description
L
4.2.1
STEP-DOWN OR BUCK
CONVERTER
IL
The MCP16311/2 is a synchronous step-down or buck
converter capable of stepping input voltages ranging
from 4.4V to 30V down to 2.0V to 24V for VIN > VOUT.
The integrated high-side switch is used to chop or
modulate the input voltage using a controlled duty
cycle. The integrated low-side switch is used to
freewheel current when the high-side switch is turned
off. High efficiency is achieved by using low-resistance
switches and low equivalent series resistance (ESR)
inductors and capacitors. When the high-side switch is
turned on, a DC voltage is applied to the inductor (VIN –
VOUT), resulting in a positive linear ramp of inductor
current. When the high-side switch turns off and the
low-side switch turns on, the applied inductor voltage is
equal to –VOUT, resulting in a negative linear ramp of
inductor current. In order to ensure there is no shootthrough current, a dead time where both switches are
off is implemented between the high-side switch
turning off and the low-side switch turning on, and the
low-side switch turning off and the high-side switch
turning on.
For steady-state, continuous inductor current
operation, the positive inductor current ramp must
equal the negative current ramp in magnitude. While
operating in steady state, the switch duty cycle must be
equal to the relationship of VOUT/VIN for constant
output voltage regulation, under the condition that the
inductor current is continuous or never reaches zero.
For discontinuous inductor current operation, the
steady-state duty cycle will be less than VOUT/VIN to
maintain voltage regulation. When the inductor current
reaches zero, the low-side switch is turned off so that
current does not flow in the reverse direction, keeping
the efficiency high. The average of the chopped input
voltage or SW node voltage is equal to the output
voltage, while the average inductor current is equal to
the output current.
 2013-2014 Microchip Technology Inc.
VOUT
S1
VIN
COUT
S2
IL
IOUT
VIN
SW
VOUT
S1 ON
S2 ON
Continuous Inductor Current Mode
IL
IOUT
VIN
SW
S2 Both
ON OFF
Discontinuous Inductor Current Mode
S1 ON
FIGURE 4-2:
Converter.
Synchronous Step-Down
DS20005255B-page 15
MCP16311/2
4.2.2
PEAK CURRENT MODE CONTROL
The MCP16311/2 integrates a peak current mode
control architecture, resulting in superior AC regulation
while minimizing the number and size of voltage loop
compensation components for integration. Peak
current mode control takes a small portion of the
inductor current, replicates it, and compares this
replicated current sense signal with the error voltage. In
practice, the inductor current and the internal switch
current are equal during the switch-on time. By adding
this peak current sense to the system control, the stepdown power train system can be approximated by a
first order system rather than a second order system.
This reduces the system complexity and increases its
dynamic performance.
For Pulse-Width Modulation (PWM) duty cycles that
exceed 50%, the control system can become bimodal,
where a wide pulse followed by a short pulse repeats
instead of the desired fixed pulse width. To prevent this
mode of operation, an internal compensating ramp is
summed into the current sense signal.
4.2.3
PULSE-WIDTH MODULATION
The internal oscillator periodically starts the switching
period, which in the MCP16311/2’s case occurs every
2 µs or 500 kHz. With the high-side integrated
N-Channel MOSFET turned on, the inductor current
ramps up until the sum of the current sense and slope
compensation ramp exceeds the integrated error
amplifier output. Once this occurs, the high-side switch
turns off and the low-side switch turns on. The error
amplifier output slews up or down to increase or
decrease the inductor peak current feeding into the
output LC filter. If the regulated output voltage is lower
than its target, the inverting error amplifier output rises.
This results in an increase in the inductor current to
correct for errors in the output voltage. The fixed
frequency duty cycle is terminated when the sensed
inductor peak current, summed with the internal slope
compensation, exceeds the output voltage of the error
amplifier. The PWM latch is set by turning off the highside internal switch and preventing it from turning on
until the beginning of the next cycle.
When working close to the boundary conduction
threshold, a jitter on the SW node may occur, reflecting
in the output voltage. Although the low-frequency
output component is very small, it may be desirable to
completely eliminate this component. To achieve this,
an RC Snubber between the SW node and GND is
used.
Typical values for the snubber are: 680 pF and 430.
Using such a snubber completely eliminates the jitter
on the SW node, but slightly decreases the overall
efficiency of the converter.
4.2.4
PFM MODE OPERATION
The MCP16311 devices are capable of automatic
operation in normal PWM or PFM mode to maintain
high efficiency at all loads. In PFM mode, the output
ripple has a variable frequency component that
changes with the input voltage and output current. With
no load, the quiescent current drawn from the output is
very low.
There are two comparators that decide when device
starts switching in PFM mode. One of the comparators
is monitoring the output voltage and has a reference of
810 mV with 10 mV hysteresis. If the load current is
low, the output rises and triggers the comparator, which
will put the logic control of the drivers and other block
circuitry (including the internal regulator VCC) in Sleep
mode to minimize the power consumption during the
switching cycle’s off period. When the output voltage
drops below its nominal value, PFM operation pulses
one or several times to bring the output back into
regulation (Figure 2-26). The second comparator fixes
the minimum duty cycle for PFM mode. Minimum duty
cycle in PFM mode depends on the sensed peak
current and input voltage. As a result, the PFM-to-PWM
mode threshold depends on load current and value of
the input voltage (Figure 2-17). If the output load
current rises above the upper threshold, the
MCP16311 transitions smoothly into PWM mode.
The MCP16312 devices will operate in PWM-only
mode even during periods of light load operation. By
operating in PWM-only mode, the output ripple remains
low and the frequency is constant (Figure 2-28).
Operating in fixed PWM mode results in lower
efficiency during light-load operation (when compared
to PFM mode (MCP16311)).
DS20005255B-page 16
 2013-2014 Microchip Technology Inc.
MCP16311/2
4.2.5
HIGH-SIDE DRIVE
The MCP16311/2 features an integrated high-side
N-Channel MOSFET for high-efficiency step-down
power conversion. An N-Channel MOSFET is used for
its low resistance and size (instead of a P-Channel
MOSFET). The N-Channel MOSFET gate must be
driven above its source to fully turn on the device, resulting in a gate-drive voltage above the input to turn on the
high-side N-Channel. The high-side N-channel source
is connected to the inductor and boost cap or switch
node. When the high-side switch is off and the low-side
switch is on, the inductor current flows through the lowside switch, providing a path to recharge the boost cap
from the boost voltage source. The voltage for the boost
cap is supplied from the internal regulator (VCC). An
internal boost blocking diode is used to prevent current
flow from the boost cap back into the regulator during
the internal switch-on time. If the boost voltage
decreases significantly, the low side will be forced low
for 90 ns in order to charge the boost capacitor.
 2013-2014 Microchip Technology Inc.
DS20005255B-page 17
MCP16311/2
NOTES:
DS20005255B-page 18
 2013-2014 Microchip Technology Inc.
MCP16311/2
5.0
APPLICATION INFORMATION
5.1
Typical Applications
The MCP16311/2 synchronous step-down converter
operates over a wide input range, up to 30V maximum.
Typical applications include generating a bias or VDD
voltage for PIC® microcontrollers, digital control system
bias supply for AC-DC converters and 12V industrial
input and similar applications.
5.2
Adjustable Output Voltage
Calculations
To calculate the resistor divider values for the
MCP16311/2 adjustable version, use Equation 5-1.
RTOP is connected to VOUT, RBOT is connected to
AGND, and both are connected to the VFB input pin.
EQUATION 5-1:
RESISTOR DIVIDER
CALCULATION
V OUT
R TOP = R BOT   ------------- – 1
V FB
EXAMPLE 5-1:
3.3V RESISTOR DIVIDER
5.3
General Design Equations
The step-down converter duty cycle can be estimated
using Equation 5-2 while operating in Continuous
Inductor Current mode. This equation accounts for the
forward drop of the two internal N-Channel MOSFETS.
As load current increases, the voltage drop in both
internal switches will increase, requiring a larger PWM
duty cycle to maintain the output voltage regulation.
Switch voltage drop is estimated by multiplying the
switch current times the switch resistance or RDSON.
EQUATION 5-2:
CONTINUOUS INDUCTOR
CURRENT DUTY CYCLE
V OUT +  I LSW  R DSONL 
D = ------------------------------------------------------------V IN –  I HSW  R DSONH 
The MCP16311/2 device features an integrated slope
compensation to prevent bimodal operation of the
PWM duty cycle. Internally, half of the inductor current
down slope is summed with the internal current sense
signal. For the proper amount of slope compensation,
it is recommended to keep the inductor down-slope
current constant by varying the inductance with VOUT,
where K = 0.22 V/µH.
EQUATION 5-3:
VOUT = 3.3V
VFB = 0.8V
K = V OUT  L
RBOT = 10 k
RTOP = 31.25 k (standard value = 31.6 k)
VOUT = 3.328V (using standard value)
EXAMPLE 5-2:
5.0V RESISTOR DIVIDER
TABLE 5-1:
VOUT = 5.0V
VFB = 0.8V
RBOT = 10 k
RTOP = 52.5 k (standard value = 52.3 k)
VOUT = 4.984V (using standard values)
EXAMPLE 5-3:
For example, for VOUT = 3.3V, an inductance of 15 µH
is recommended.
12.0V RESISTOR DIVIDER
VOUT = 12.0V
VFB = 0.8V
RECOMMENDED INDUCTOR
VALUES
VOUT
K
LSTANDARD
2.0V
0.20
10 µH
3.3V
0.22
15 µH
5.0V
0.23
22 µH
12V
0.21
56 µH
15V
0.22
68 µH
24V
0.24
100 µH
RBOT = 10 k
RTOP = 140 k (standard value = 140 k)
The error amplifier is internally compensated to ensure
loop stability. External resistor dividers, inductance and
output capacitance all have an impact on the control
system and should be selected carefully and evaluated
for stability. A 10 kΩ bottom resistor is recommended as a
good trade-off for quiescent current and noise immunity.
 2013-2014 Microchip Technology Inc.
DS20005255B-page 19
MCP16311/2
5.4
Input Capacitor Selection
5.6
Inductor Selection
The step-down converter input capacitor must filter the
high-input ripple current that results from pulsing or
chopping the input voltage. The MCP16311/2 input
voltage pin is used to supply voltage for the power train
and as a source for internal bias. A low equivalent
series resistance (ESR), preferably a ceramic
capacitor,
is
recommended.
The
necessary
capacitance is dependent upon the maximum load
current and source impedance. Three capacitor
parameters to keep in mind are the voltage rating,
equivalent series resistance and the temperature
rating. For wide temperature range applications, a
multi-layer X7R dielectric is recommended, while for
applications with limited temperature range, a
multi-layer X5R dielectric is acceptable. Typically, input
capacitance between 10 µF and 20 µF is sufficient for
most applications. For applications with 100 mA to
200 mA load, a 4.7 µF to 2.2 µF X7R capacitor can be
used, depending on the input source and its
impedance. In case of an application with high
variations of the input voltage, a higher capacitor value
is recommended. The input capacitor voltage rating
must be VIN plus margin.
The MCP16311/2 is designed to be used with small
surface-mount inductors. Several specifications should
be considered prior to selecting an inductor. To
optimize system performance, low DCR inductors
should be used.
Table 5-2 contains the recommended range for the
input capacitor value.
EQUATION 5-5:
EQUATION 5-4:
V
EXAMPLE 5-4:
VIN = 12V
VOUT = 3.3V
IOUT = 800 mA
I
I LPK = -------L- + I OUT
2
TABLE 5-2:
CAPACITOR VALUE RANGE
Parameter
Min.
Max.
CIN
2.2 µF
None
COUT
20 µF
None
DS20005255B-page 20
Where:
Inductor ripple current
=
319 mA
Inductor peak current
=
960 mA
For this example, an inductor with a current saturation
rating of minimum 960 mA is recommended. Low DCR
inductors result in higher system efficiency. A trade-off
between size, cost and efficiency is made to achieve
the desired results.
TABLE 5-3:
Part Number
MCP16311/2 RECOMMENDED
3.3V VOUT INDUCTORS
ISAT (A)
The output voltage capacitor rating should be a
minimum of VOUT plus margin.
INDUCTOR PEAK
CURRENT
DCR ()
The amount and type of output capacitance and
equivalent series resistance will have a significant
effect on the output ripple voltage and system stability.
The range of the output capacitance is limited due to
the integrated compensation of the MCP16311/2. See
Table 5-2 for the recommended output capacitor range.
–V
L
IN
OUT
-  t ON
 IL = ---------------------------
Output Capacitor Selection
The output capacitor provides a stable output voltage
during sudden load transients and reduces the output
voltage ripple. As with the input capacitor, X5R and
X7R ceramic capacitors are well suited for this
application. For typical applications, the output
capacitance can be as low as 10 µF ceramic and as
high as 100 µF electrolytic. In a typical application, a
20 µF output capacitance usage will result in a 10 mV
output ripple.
INDUCTOR RIPPLE
CURRENT
Value
(µH)
5.5
To optimize system performance, the inductance value
is determined by the output voltage (Table 5-1) so the
inductor ripple current is somewhat constant over the
output voltage range.
Size
WxLxH
(mm)
Coilcraft
XAL4040
15
0.109
2.8
4.0x4.0x2.1
LPS6235
15
0.125
2.00
6.0x6.0x3.5
6.1x6.1x3.2
MSS6132
15
0.135
1.56
XAL6060
15
0.057
1.78 6.36x6.5x6.1
MSS7341
15
0.057
1.78
7.3x7.3x4.1
 2013-2014 Microchip Technology Inc.
MCP16311/2
Size
WxLxH
(mm)
74408943150
15
0.118
1.7
4.8x4.8x3.8
744062150
15
0.085
1.1
6.8x6.8x2.3
Part Number
Value
(µH)
ISAT (A)
MCP16311/2 RECOMMENDED
3.3V VOUT INDUCTORS
DCR ()
TABLE 5-3:
Wurth Elektronik®
744778115
15
0.1
1.75
7.3x7.3x3.2
7447779115
15
0.07
2.2
7.3x7.3x4.5
15
0.095
1.08
5.2x5.2x2.5
14.1 0.103
1.1
6.0x6.0x3.0
Another important aspect when creating such an
application is the value of the inductor. The value of the
inductor needs to follow Equation 5-3 or, as a guideline,
Table 5-1, where the output voltage is approximated as
the sum of the forward voltages of the LEDs and a 0.8V
headroom for the sense resistor. A typical application is
shown in Figure 5-3.
The following equations are used to determine the
value and the losses for the sense resistor:
EQUATION 5-6:
VFB
RB = ----------ILED
Coiltronics®
SD25
SD6030
PLOSSES = V FB  I LED
®
Where:
TDK - EPC
B82462G4153M
15
0.097
1.05
6.0x6.0x3.0
B82462A4153K
15
0.21
1.5
6.0x6.0x3.0
5.7
Boost Capacitor
The boost capacitor is used to supply current for the
internal high-side drive circuitry that is above the input
voltage. The boost capacitor must store enough energy to
completely drive the high-side switch on and off. A 100 nF
X5R or X7R capacitor is recommended for all
applications. The boost capacitor maximum voltage is 5V.
VFB = Feedback Voltage
EXAMPLE 5-5:
ILED = 400 mA
VFB = 0.8V
VF = 1 x 3.2V (one white LED is used)
RB = 2
PLOSSES = 0.32 W (sense resistor losses)
L = 22 µH
5.8
Vcc Capacitor
The VCC internal bias regulates at 5V. The VCC pin is
current limited to 50 mA and protected from a shortcircuit condition at 150 mA load. The VCC regulator
must sustain all load and line transients because it
supplies the internal drivers for power switches. For
stability reasons, the VCC capacitor must be at least
1 µF X7R ceramic for extended temperature range, or
X5R for limited temperature range.
5.9
MCP16312 – LED Constant
Current Driver
MCP16312 can be used to drive an LED or a string of
LEDs. The process of transforming the MCP16312
from a constant voltage source into a constant current
source is simple. It implies that the sensing/feedback
for the current is on the low side by adding a resistor in
series with the string of LEDs.
When using the MCP16312 as an LED driver, care must
be taken when selecting the sense resistor. Due to the
high feedback voltage of 0.8V, there will be significant
losses on the sense resistor, so a larger package with
better power dissipation must be selected.
 2013-2014 Microchip Technology Inc.
5.10
Thermal Calculations
The MCP16311/2 is available in MSOP-8 and DFN-8
packages. By calculating the power dissipation and
applying the package thermal resistance (θJA), the
junction temperature is estimated. The maximum
continuous junction temperature rating for the
MCP16311/2 is +125°C.
To quickly estimate the internal power dissipation for
the switching step-down regulator, an empirical
calculation using measured efficiency can be used.
Given the measured efficiency, the internal power
dissipation is estimated in Equation 5-7. This power
dissipation includes all internal and external
component losses. For a quick internal estimate,
subtract the estimated inductor DCR loss from the PDIS
calculation in Equation 5-7.
EQUATION 5-7:
TOTAL POWER
DISSIPATION ESTIMATE
V OUT  I OUT
P DIS = ------------------------------- –  V OUT  I OUT 
Efficiency
DS20005255B-page 21
MCP16311/2
The difference between the first term, input power, and
the second term, power delivered, is the total system
power dissipation. The inductor losses are estimated
by PL = IOUT2 x LDCR.
EXAMPLE 5-6:
POWER DISSIPATION –
MCP16311/2 MSOP
PACKAGE
VIN =
12V
VOUT =
5.0V
IOUT =
0.8A
Efficiency
=
92.5%
Total System Dissipation
=
324 mW
LDCR =
0.15 
PL =
96 mW
MCP16311/2 internal power dissipation estimate:
PDIS – PL =
228 mW
JA =
Estimated Junction =
Temperature Rise
211°C/W
EXAMPLE 5-7:
+48.1°C
POWER DISSIPATION –
MCP16311/2 DFN
PACKAGE
VIN =
12V
VOUT =
3.3V
IOUT =
0.8A
Efficiency
=
90%
Total System Dissipation
=
293 mW
LDCR =
0.15 
PL =
96 mW
MCP16311 internal power dissipation estimate:
PDIS – PL =
197 mW
JA =
68°C/W
Estimated Junction
Temperature Rise
DS20005255B-page 22
=
+13.4°C
 2013-2014 Microchip Technology Inc.
MCP16311/2
5.11
supplying high-frequency switch current, the input
capacitor also provides a stable voltage source for the
internal MCP16311/2 circuitry. Unstable PWM operation can result if there are excessive transients or ringing on the VIN pin of the MCP16311/2 device. In
Figure 5-1, the input capacitors are placed close to the
VIN pins. A ground plane on the bottom of the board
provides a low-resistive and low-inductive path for the
return current. The next priority in placement is the
freewheeling current loop formed by output capacitors
and inductance (L1), while strategically placing the output capacitor ground return close to the input capacitor
ground return. Then, CBOOST should be placed
between the boost pin and the switch node pin. This
leaves space close to the MCP16311/2 VFB pin to place
RTOP and RBOT. The feedback loop must be routed
away from the switch node, so noise is not coupled into
the high-impedance VFB input.
Printed Circuit Board (PCB)
Layout Information
Good PCB layout techniques are important to any
switching circuitry, and switching power supplies are no
different. When wiring the switching high-current paths,
short and wide traces should be used. Therefore, it is
important that the input and output capacitors be
placed as close as possible to the MCP16311/2 to
minimize the loop area.
The feedback resistors and feedback signal should be
routed away from the switching node and the switching
current loop. When possible, ground planes and traces
should be used to help shield the feedback signal and
minimize noise and magnetic interference.
A good MCP16311/2 layout starts with the placement of
the input capacitor, which supplies current to the input
of the circuit when the switch is turned on. In addition to
CBOOST
VIN
12V
BOOST
VIN
CIN
CVCC
FIGURE 5-1:
SW
EN
REN
L1
VFB
VCC
GND
Component
Value
CIN
2 x 10 µF
COUT
2 x 10 µF
L1
22 µH
RT
52.3 k
RB
10 k
REN
1 M
CVCC
1 µF
CBOOST
0.1 µF
VOUT
5V @ 1A
COUT
RT
RB
MSOP-8 Recommended Layout, 5V Output Design.
 2013-2014 Microchip Technology Inc.
DS20005255B-page 23
MCP16311/2
CBOOST
VIN
12V
BOOST
VIN
L1
SW
CIN
EN
REN
CVCC
FIGURE 5-2:
DS20005255B-page 24
VFB
VCC
GND
Component
Value
CIN
2 x 10 µF
COUT
2 x 10 µF
L1
15 µH
RT
31.2 k
RB
10 k
REN
1 M
CVCC
1 µF
CBOOST
0.1 µF
VOUT
3.3V @ 1A
COUT
RT
RB
DFN Recommended Layout, 3.3V Output Design.
 2013-2014 Microchip Technology Inc.
MCP16311/2
CBOOST
VIN
12V
BOOST
VIN
ILED = 400 mA
L1
SW
COUT
LED
CIN
EN
REN
CVCC
FIGURE 5-3:
VFB
VCC
GND
RB
Component
Value
CIN
2 x 10 µF
COUT
2 x 10 µF
L1
15 µH
RB
2
REN
1 M
CVCC
1 µF
CBOOST
0.1 µF
LED
1 x White LED
V FB
RB = ----------ILED
MCP16312 - Typical LED Driver Application: 400 mA Output.
 2013-2014 Microchip Technology Inc.
DS20005255B-page 25
MCP16311/2
NOTES:
DS20005255B-page 26
 2013-2014 Microchip Technology Inc.
MCP16311/2
6.0
PACKAGING INFORMATION
6.1
Package Marking Information
8-Lead MSOP (3x3 mm)
Example
16311E
309256
8-Lead TDFN (2x3)
Example
Part Number
Legend: XX...X
Y
YY
WW
NNN
e3
*
Note:
Code
MCP16311T-E/MNY
ABM
MCP16312T-E/MNY
ABU
ABM
309
25
Customer-specific information
Year code (last digit of calendar year)
Year code (last 2 digits of calendar year)
Week code (week of January 1 is week ‘01’)
Alphanumeric traceability code
Pb-free JEDEC® designator for Matte Tin (Sn)
This package is Pb-free. The Pb-free JEDEC designator ( e3)
can be found on the outer packaging for this package.
In the event the full Microchip part number cannot be marked on one line, it will
be carried over to the next line, thus limiting the number of available
characters for customer-specific information.
 2013-2014 Microchip Technology Inc.
DS20005255B-page 27
MCP16311/2
Note:
For the most current package drawings, please see the Microchip Packaging Specification located at
http://www.microchip.com/packaging
DS20005255B-page 28
 2013-2014 Microchip Technology Inc.
MCP16311/2
Note:
For the most current package drawings, please see the Microchip Packaging Specification located at
http://www.microchip.com/packaging
 2013-2014 Microchip Technology Inc.
DS20005255B-page 29
MCP16311/2
Note:
For the most current package drawings, please see the Microchip Packaging Specification located at
http://www.microchip.com/packaging
DS20005255B-page 30
 2013-2014 Microchip Technology Inc.
MCP16311/2
Note:
For the most current package drawings, please see the Microchip Packaging Specification located at
http://www.microchip.com/packaging
 2013-2014 Microchip Technology Inc.
DS20005255B-page 31
MCP16311/2
Note:
For the most current package drawings, please see the Microchip Packaging Specification located at
http://www.microchip.com/packaging
DS20005255B-page 32
 2013-2014 Microchip Technology Inc.
MCP16311/2
(
!""#$%&'
! "# $% &"' "" ($ ) %
*++&&&! !+ $
 2013-2014 Microchip Technology Inc.
DS20005255B-page 33
MCP16311/2
NOTES:
DS20005255B-page 34
 2013-2014 Microchip Technology Inc.
MCP16311/2
APPENDIX A:
REVISION HISTORY
Revision B (November 2014)
The following is the list of modifications:
1.
2.
3.
4.
5.
6.
Added AEC-Q100 qualification information.
Updated the Typical Applications section.
Updated the DC Characteristics table.
Updated Section 4.2.2 “Peak Current Mode
Control”.
Updated the standard values in Example 5-1.
Added a 24V option in Table 5-1.
Revision A (December 2013)
• Original Release of this Document.
 2013-2014 Microchip Technology Inc.
DS20005255B-page 35
MCP16311/2
NOTES:
DS20005255B-page 36
 2013-2014 Microchip Technology Inc.
MCP16311/2
PRODUCT IDENTIFICATION SYSTEM
To order or obtain information, e.g., on pricing or delivery, refer to the factory or the listed sales office.
PART NO.
X
/XX
Device
Temperature
Range
Package
Device:
MCP16311:
MCP16311T:
MCP16312:
MCP16312T:
Temperature
Range:
E
Package:
MNY* =
MS
=
*Y
Examples:
a) MCP16311-E/MS:
High-Efficiency, PFM/PWM Integrated
Synchronous Switch Step-Down Regulator
(MSOP only)
High-Efficiency, PFM/PWM Integrated
Synchronous Switch Step-Down Regulator
(Tape and Reel) (MSOP and TDFN)
High-Efficiency, PFM Integrated Synchronous
Switch Step-Down Regulator
(MSOP only)
High-Efficiency, PWM Integrated Synchronous
Switch Step-Down Regulator (Tape and Reel)
(MSOP and TDFN)
= -40°C to +125°C
(Extended)
Extended Temperature,
8LD MSOP package
b) MCP16311T-E/MS: Tape and Reel,
Extended Temperature,
8LD MSOP package
c) MCP16311T-E/MNY: Tape and Reel,
Extended Temperature,
8LD 2 x 3 TDFN package
a) MCP16312-E/MS:
Extended Temperature,
8LD MSOP package
b) MCP16312T-E/MS: Tape and Reel,
Extended Temperature,
8LD MSOP package
c) MCP16312T-E/MNY: Tape and Reel,
Extended Temperature,
8LD 2 x 3 TDFN package
Plastic Micro Small Outline Package
Plastic Dual Flat, No Lead Package 2 x 3 x 0.75 mm Body
= Nickel palladium gold manufacturing designator.
 2013-2014 Microchip Technology Inc.
DS20005255B-page 37
MCP16311/2
NOTES:
DS20005255B-page 38
 2013-2014 Microchip Technology Inc.
Note the following details of the code protection feature on Microchip devices:
•
Microchip products meet the specification contained in their particular Microchip Data Sheet.
•
Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the
intended manner and under normal conditions.
•
There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our
knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data
Sheets. Most likely, the person doing so is engaged in theft of intellectual property.
•
Microchip is willing to work with the customer who is concerned about the integrity of their code.
•
Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not
mean that we are guaranteeing the product as “unbreakable.”
Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our
products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts
allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.
Information contained in this publication regarding device
applications and the like is provided only for your convenience
and may be superseded by updates. It is your responsibility to
ensure that your application meets with your specifications.
MICROCHIP MAKES NO REPRESENTATIONS OR
WARRANTIES OF ANY KIND WHETHER EXPRESS OR
IMPLIED, WRITTEN OR ORAL, STATUTORY OR
OTHERWISE, RELATED TO THE INFORMATION,
INCLUDING BUT NOT LIMITED TO ITS CONDITION,
QUALITY, PERFORMANCE, MERCHANTABILITY OR
FITNESS FOR PURPOSE. Microchip disclaims all liability
arising from this information and its use. Use of Microchip
devices in life support and/or safety applications is entirely at
the buyer’s risk, and the buyer agrees to defend, indemnify and
hold harmless Microchip from any and all damages, claims,
suits, or expenses resulting from such use. No licenses are
conveyed, implicitly or otherwise, under any Microchip
intellectual property rights.
Trademarks
The Microchip name and logo, the Microchip logo, dsPIC,
FlashFlex, flexPWR, JukeBlox, KEELOQ, KEELOQ logo, Kleer,
LANCheck, MediaLB, MOST, MOST logo, MPLAB,
OptoLyzer, PIC, PICSTART, PIC32 logo, RightTouch, SpyNIC,
SST, SST Logo, SuperFlash and UNI/O are registered
trademarks of Microchip Technology Incorporated in the
U.S.A. and other countries.
The Embedded Control Solutions Company and mTouch are
registered trademarks of Microchip Technology Incorporated
in the U.S.A.
Analog-for-the-Digital Age, BodyCom, chipKIT, chipKIT logo,
CodeGuard, dsPICDEM, dsPICDEM.net, ECAN, In-Circuit
Serial Programming, ICSP, Inter-Chip Connectivity, KleerNet,
KleerNet logo, MiWi, MPASM, MPF, MPLAB Certified logo,
MPLIB, MPLINK, MultiTRAK, NetDetach, Omniscient Code
Generation, PICDEM, PICDEM.net, PICkit, PICtail,
RightTouch logo, REAL ICE, SQI, Serial Quad I/O, Total
Endurance, TSHARC, USBCheck, VariSense, ViewSpan,
WiperLock, Wireless DNA, and ZENA are trademarks of
Microchip Technology Incorporated in the U.S.A. and other
countries.
SQTP is a service mark of Microchip Technology Incorporated
in the U.S.A.
Silicon Storage Technology is a registered trademark of
Microchip Technology Inc. in other countries.
GestIC is a registered trademarks of Microchip Technology
Germany II GmbH & Co. KG, a subsidiary of Microchip
Technology Inc., in other countries.
All other trademarks mentioned herein are property of their
respective companies.
© 2013-2014, Microchip Technology Incorporated, Printed in
the U.S.A., All Rights Reserved.
ISBN: 978-1-63276-806-3
QUALITY MANAGEMENT SYSTEM
CERTIFIED BY DNV
== ISO/TS 16949 ==
 2013-2014 Microchip Technology Inc.
Microchip received ISO/TS-16949:2009 certification for its worldwide
headquarters, design and wafer fabrication facilities in Chandler and
Tempe, Arizona; Gresham, Oregon and design centers in California
and India. The Company’s quality system processes and procedures
are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping
devices, Serial EEPROMs, microperipherals, nonvolatile memory and
analog products. In addition, Microchip’s quality system for the design
and manufacture of development systems is ISO 9001:2000 certified.
DS20005255B-page 39
Worldwide Sales and Service
AMERICAS
ASIA/PACIFIC
ASIA/PACIFIC
EUROPE
Corporate Office
2355 West Chandler Blvd.
Chandler, AZ 85224-6199
Tel: 480-792-7200
Fax: 480-792-7277
Technical Support:
http://www.microchip.com/
support
Web Address:
www.microchip.com
Asia Pacific Office
Suites 3707-14, 37th Floor
Tower 6, The Gateway
Harbour City, Kowloon
Hong Kong
Tel: 852-2943-5100
Fax: 852-2401-3431
India - Bangalore
Tel: 91-80-3090-4444
Fax: 91-80-3090-4123
Austria - Wels
Tel: 43-7242-2244-39
Fax: 43-7242-2244-393
Denmark - Copenhagen
Tel: 45-4450-2828
Fax: 45-4485-2829
Australia - Sydney
Tel: 61-2-9868-6733
Fax: 61-2-9868-6755
Atlanta
Duluth, GA
Tel: 678-957-9614
Fax: 678-957-1455
China - Beijing
Tel: 86-10-8569-7000
Fax: 86-10-8528-2104
Austin, TX
Tel: 512-257-3370
China - Chengdu
Tel: 86-28-8665-5511
Fax: 86-28-8665-7889
Boston
Westborough, MA
Tel: 774-760-0087
Fax: 774-760-0088
Chicago
Itasca, IL
Tel: 630-285-0071
Fax: 630-285-0075
Cleveland
Independence, OH
Tel: 216-447-0464
Fax: 216-447-0643
Dallas
Addison, TX
Tel: 972-818-7423
Fax: 972-818-2924
Detroit
Novi, MI
Tel: 248-848-4000
Houston, TX
Tel: 281-894-5983
Indianapolis
Noblesville, IN
Tel: 317-773-8323
Fax: 317-773-5453
Los Angeles
Mission Viejo, CA
Tel: 949-462-9523
Fax: 949-462-9608
New York, NY
Tel: 631-435-6000
San Jose, CA
Tel: 408-735-9110
Canada - Toronto
Tel: 905-673-0699
Fax: 905-673-6509
DS20005255B-page 40
China - Chongqing
Tel: 86-23-8980-9588
Fax: 86-23-8980-9500
China - Hangzhou
Tel: 86-571-8792-8115
Fax: 86-571-8792-8116
China - Hong Kong SAR
Tel: 852-2943-5100
Fax: 852-2401-3431
China - Nanjing
Tel: 86-25-8473-2460
Fax: 86-25-8473-2470
China - Qingdao
Tel: 86-532-8502-7355
Fax: 86-532-8502-7205
China - Shanghai
Tel: 86-21-5407-5533
Fax: 86-21-5407-5066
China - Shenyang
Tel: 86-24-2334-2829
Fax: 86-24-2334-2393
China - Shenzhen
Tel: 86-755-8864-2200
Fax: 86-755-8203-1760
China - Wuhan
Tel: 86-27-5980-5300
Fax: 86-27-5980-5118
China - Xian
Tel: 86-29-8833-7252
Fax: 86-29-8833-7256
India - New Delhi
Tel: 91-11-4160-8631
Fax: 91-11-4160-8632
France - Paris
Tel: 33-1-69-53-63-20
Fax: 33-1-69-30-90-79
India - Pune
Tel: 91-20-3019-1500
Japan - Osaka
Tel: 81-6-6152-7160
Fax: 81-6-6152-9310
Germany - Dusseldorf
Tel: 49-2129-3766400
Germany - Munich
Tel: 49-89-627-144-0
Fax: 49-89-627-144-44
Japan - Tokyo
Tel: 81-3-6880- 3770
Fax: 81-3-6880-3771
Germany - Pforzheim
Tel: 49-7231-424750
Korea - Daegu
Tel: 82-53-744-4301
Fax: 82-53-744-4302
Italy - Milan
Tel: 39-0331-742611
Fax: 39-0331-466781
Korea - Seoul
Tel: 82-2-554-7200
Fax: 82-2-558-5932 or
82-2-558-5934
Italy - Venice
Tel: 39-049-7625286
Malaysia - Kuala Lumpur
Tel: 60-3-6201-9857
Fax: 60-3-6201-9859
Netherlands - Drunen
Tel: 31-416-690399
Fax: 31-416-690340
Malaysia - Penang
Tel: 60-4-227-8870
Fax: 60-4-227-4068
Poland - Warsaw
Tel: 48-22-3325737
Philippines - Manila
Tel: 63-2-634-9065
Fax: 63-2-634-9069
Singapore
Tel: 65-6334-8870
Fax: 65-6334-8850
Taiwan - Hsin Chu
Tel: 886-3-5778-366
Fax: 886-3-5770-955
Spain - Madrid
Tel: 34-91-708-08-90
Fax: 34-91-708-08-91
Sweden - Stockholm
Tel: 46-8-5090-4654
UK - Wokingham
Tel: 44-118-921-5800
Fax: 44-118-921-5820
Taiwan - Kaohsiung
Tel: 886-7-213-7830
Taiwan - Taipei
Tel: 886-2-2508-8600
Fax: 886-2-2508-0102
Thailand - Bangkok
Tel: 66-2-694-1351
Fax: 66-2-694-1350
China - Xiamen
Tel: 86-592-2388138
Fax: 86-592-2388130
China - Zhuhai
Tel: 86-756-3210040
Fax: 86-756-3210049
03/25/14
 2013-2014 Microchip Technology Inc.