AD AD823AR-REEL Dual, 16 mhz, rail-to-rail fet input amplifier Datasheet

a
FEATURES
Single Supply Operation
Output Swings Rail to Rail
Input Voltage Range Extends Below Ground
Single Supply Capability from +3 V to +36 V
High Load Drive
Capacitive Load Drive of 500 pF, G = +1
Output Current of 15 mA, 0.5 V from Supplies
Excellent AC Performance on 2.6 mA/Amplifier
–3 dB Bandwidth of 16 MHz, G = +1
350 ns Settling Time to 0.01% (2 V Step)
Slew Rate of 22 V/ms
Good DC Performance
800 mV Max Input Offset Voltage
2 mV/8C Offset Voltage Drift
25 pA Max Input Bias Current
Low Distortion
–108 dBc Worst Harmonic @ 20 kHz
Low Noise
16 nV/√Hz @ 10 kHz
No Phase Inversion with Inputs to the Supply Rails
Dual, 16 MHz, Rail-to-Rail
FET Input Amplifier
AD823
CONNECTION DIAGRAM
8-Pin Plastic Mini-DIP
and
8-Lead SOIC
OUT1
1
8 +VS
–IN1
2
7 OUT2
+IN1
3
6 –IN2
–VS
4
AD823
5 +IN2
The AD823 is available over the industrial temperature range of
–40°C to +85°C and is offered in both 8-pin plastic DIP and
SOIC packages.
RL = 100kΩ
CL = 50pF
VS = +3V
3V
APPLICATIONS
Battery Powered Precision Instrumentation
Photodiode Preamps
Active Filters
12- to 16-Bit Data Acquisition Systems
Medical Instrumentation
GND
PRODUCT DESCRIPTION
500mV
The AD823 is a dual precision, 16 MHz, JFET input op amp
that can operate from a single supply of +3.0 V to +36 V, or
dual supplies of ± 1.5 V to ± 18 V. It has true single supply
capability with an input voltage range extending below ground
in single supply mode. Output voltage swing extends to within
50 mV of each rail for IOUT ≤ 100 µA providing outstanding output dynamic range.
This combination of ac and dc performance, plus the outstanding load drive capability results in an exceptionally versatile amplifier for applications such as A/D drivers, high-speed active
filters, and other low voltage, high dynamic range systems.
Figure 1. Output Swing, VS = +3 V, G = +1
2
1
0
–1
OUTPUT – dB
Offset voltage of 800 µV max, offset voltage drift of 2 µV/°C,
input bias currents below 25 pA and low input voltage noise
provide dc precision with source impedances up to a Gigohm.
16 MHz, –3 dB bandwidth, –108 dB THD @ 20 kHz and
22 V/µs slew rate are provided with a low supply current of
2.6 mA per amplifier. The AD823 drives up to 500 pF of direct
capacitive load as a follower, and provides an output current of
15 mA, 0.5 V from the supply rails. This allows the amplifier to
handle a wide range of load conditions.
200µs
–2
–3
–4
–5
VS = +5V
G = +1
–6
–7
–8
1k
10k
100k
1M
FREQUENCY – Hz
10M
Figure 2. Small Signal Bandwidth, G = +1
REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
© Analog Devices, Inc., 1995
One Technology Way, P.O. Box 9106, Norwood. MA 02062-9106, U.S.A.
Tel: 617/329-4700
Fax: 617/326-8703
AD823–SPECIFICATIONS (@ T = +25°C, V = +5 V, R = 2 kΩ to +2.5 V, unless otherwise noted)
A
Parameter
DYNAMIC PERFORMANCE
–3 dB Bandwidth, VO ≤ 0.2 V p-p
Full Power Response
Slew Rate
Settling Time
to 0.1%
to 0.01%
NOISE/DISTORTION PERFORMANCE
Input Voltage Noise
Input Current Noise
Harmonic Distortion
Crosstalk
f = 1 kHz
f = 1 MHz
DC PERFORMANCE
Initial Offset
Max Offset Over Temperature
Offset Drift
Input Bias Current
at TMAX
Input Offset Current
at TMAX
Open-Loop Gain
S
L
Conditions
Min
G = +1
VO = 2 V p-p
G = –1, VO = 4 V Step
G = –1, VO = 2 V Step
12
OUTPUT CHARACTERISTICS
Output Voltage Swing
IL = ± 100 µA
IL = ± 2 mA
IL = ± 10 mA
Output Current
Short Circuit Current
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current
Power Supply Rejection Ratio
Max
Units
16
3.5
22
MHz
MHz
V/µs
320
350
ns
ns
f = 10 kHz
f = 1 kHz
RL = 600 Ω to 2.5 V, VO = 2 V p-p,
f = 20 kHz
16
1
–108
nV/√Hz
fA/√Hz
dBc
RL = 5 kΩ
RL = 5 kΩ
–130
–93
dB
dB
VCM = 0 V to +4 V
0.2
0.3
2
3
0.5
2
0.5
VO = 0.2 V to 4 V
RL = 2 kΩ
TMIN to TMAX
INPUT CHARACTERISTICS
Input Common-Mode Voltage Range
Input Resistance
Input Capacitance
Common-Mode Rejection Ratio
AD823A
Typ
VCM = 0 V to 3 V
14
20
20
25
5
20
45
V
Ω
pF
dB
0.025 to 4.975
0.08 to 4.92
0.25 to 4.75
16
40
30
500
+3
70
5.2
80
mV
mV
µV/°C
pA
nA
pA
nA
V/mV
V/mV
–0.2 to 3 –0.2 to 3.8
1013
1.8
60
76
VOUT = 0.5 V to 4.5 V
Sourcing to 2.5 V
Sinking to 2.5 V
G = +1
TMIN to TMAX, Total
VS = +5 V to +15 V, TMIN to TMAX
0.8
2.0
V
V
V
mA
mA
mA
pF
+36
5.6
V
mA
dB
Specification subject to change without notice.
–2–
REV. 0
AD823
SPECIFICATIONS
(@ TA = +25°C, VS = +3.3 V, RL = 2 kΩ to +1.65 V, unless otherwise noted)
Parameter
DYNAMIC PERFORMANCE
–3 dB Bandwidth, VO ≤ 0.2 V p-p
Full Power Response
Slew Rate
Settling Time
to 0.1%
to 0.01%
NOISE/DISTORTION PERFORMANCE
Input Voltage Noise
Input Current Noise
Harmonic Distortion
Crosstalk
f = 1 kHz
f = 1 MHz
DC PERFORMANCE
Initial Offset
Max Offset Over Temperature
Offset Drift
Input Bias Current
at TMAX
Input Offset Current
at TMAX
Open-Loop Gain
Conditions
Min
G = +1
VO = 2 V p-p
G = –1, VO = 2 V Step
G = –1, VO = 2 V Step
12
OUTPUT CHARACTERISTICS
Output Voltage Swing
IL = ± 100 µA
IL = ± 2 mA
IL = ± 10 mA
Output Current
Short Circuit Current
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current
Power Supply Rejection Ratio
Units
MHz
MHz
V/µs
250
300
ns
ns
f = 10 kHz
f = 1 kHz
RL = 100 Ω, VO = 2 V p-p, f = 20 kHz
16
1
–93
nV/√Hz
fA/√Hz
dBc
RL = 5 kΩ
RL = 5 kΩ
–130
–93
dB
dB
VCM = 0 V to +2 V
0.2
0.5
2
3
0.5
2
0.5
VO = 0.2 V to 2 V
RL = 2 kΩ
VCM = 0 V to 1 V
13
15
12
–3–
25
5
20
30
70
5.0
80
mV
mV
µV/°C
pA
nA
pA
nA
V/mV
V/mV
V
Ω
pF
dB
0.025 to 3.275
0.08 to 3.22
0.25 to 3.05
15
40
30
500
+3
TMIN to TMAX, Total
VS = +3.3 V to +15 V, TMIN to TMAX
1.5
2.5
–0.2 to 1 –0.2 to 1.8
1013
1.8
54
70
VOUT = 0.5 V to 2.5 V
Sourcing to 1.5 V
Sinking to 1.5 V
G = +1
Specification subject to change without notice.
REV. 0
Max
15
3.2
20
TMIN to TMAX
INPUT CHARACTERISTICS
Input Common-Mode Voltage Range
Input Resistance
Input Capacitance
Common-Mode Rejection Ratio
AD823A
Typ
V
V
V
mA
mA
mA
pF
+36
5.7
V
mA
dB
AD823–SPECIFICATIONS
Parameter
DYNAMIC PERFORMANCE
–3 dB Bandwidth, VO ≤ 0.2 V p-p
Full Power Response
Slew Rate
Settling Time
to 0.1%
to 0.01%
NOISE/DISTORTION PERFORMANCE
Input Voltage Noise
Input Current Noise
Harmonic Distortion
Crosstalk
f = 1 kHz
f = 1 MHz
DC PERFORMANCE
Initial Offset
Max Offset Over Temperature
Offset Drift
Input Bias Current
at TMAX
Input Offset Current
at TMAX
Open-Loop Gain
(@ TA = +25°C, VS = ±15 V, RL = 2 kΩ to 0 V, unless otherwise noted)
Conditions
Min
G = +1
VO = 2 V p-p
G = –1, VO = 10 V Step
G = –1, VO = 10 V Step
12
OUTPUT CHARACTERISTICS
Output Voltage Swing
IL = ± 100 µA
IL = ± 2 mA
IL = ± 10 mA
Output Current
Short Circuit Current
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current
Power Supply Rejection Ratio
Max
Units
16
4
25
MHz
MHz
V/µs
550
650
ns
ns
f = 10 kHz
f = 1 kHz
RL = 600 Ω, VO = 10 V p-p,
f = 20 kHz
16
1
–90
nV/√Hz
fA/√Hz
dBc
RL = 5 kΩ
RL = 5 kΩ
–130
–93
dB
dB
17
0.7
1.0
2
5
60
0.5
2
0.5
VCM = 0 V
VCM = –10 V
VCM = 0 V
VO = +10 V to –10 V
RL = 2 kΩ
TMIN to TMAX
INPUT CHARACTERISTICS
Input Common-Mode Voltage Range
Input Resistance
Input Capacitance
Common-Mode Rejection Ratio
AD823A
Typ
VCM = –15 V to +13 V
30
30
3.5
7
30
5
20
60
V/mV
V/mV
–15.2 to 13 –15.2 to 13.8
1013
1.8
66
82
VOUT = –14.5 V to +14.5 V
Sourcing to 0 V
Sinking to 0 V
G = +1
–14.95 to +14.95
–14.92 to +14.92
–14.75 to +14.75
17
80
60
500
+3
TMIN to TMAX, Total
VS = +5 V to +15 V, TMIN to TMAX 70
7.0
80
mV
mV
µV/°C
pA
pA
nA
pA
nA
V
Ω
pF
dB
V
V
V
mA
mA
mA
pF
+36
8.4
V
mA
dB
Specification subject to change without notice.
–4–
REV. 0
AD823
ABSOLUTE MAXIMUM RATINGS 1
MAXIMUM POWER DISSIPATION – Watts
2.0
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +36 V
Internal Power Dissipation2
Plastic Package (N) . . . . . . . . . . . . . . . . . . . . . . . . 1.3 Watts
Small Outline Package (R) . . . . . . . . . . . . . . . . . . . 0.9 Watts
Input Voltage (Common Mode) . . . . . . . . . . . . . . . . . . . . ± VS
Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . ± 1.2 V
Output Short Circuit Duration
. . . . . . . . . . . . . . . . . . . . . . Observe Power Derating Curves
Storage Temperature Range N, R . . . . . . . . . –65°C to +125°C
Operating Temperature Range . . . . . . . . . . . . –40°C to +85°C
Lead Temperature Range (Soldering 10 sec) . . . . . . . . +300°C
NOTES
1
Stresses above those listed under “Absolute Maximum Ratings” may cause
permanent damage to the device. This is a stress rating only and functional
operation of the device at these or any other conditions above those indicated in the
operational section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
2
Specification is for device in free air:
8-Pin Plastic Package: θJA = 90°C/Watt
8-Pin SOIC Package: θJA = 160°C/Watt
8-PIN MINI-DIP PACKAGE
TJ = +150°C
1.5
1.0
8-PIN SOIC PACKAGE
0.5
0
–50 –40 –30 –20 –10 0 10 20 30 40 50 60
AMBIENT TEMPERATURE – °C
70
80 90
Figure 3. Maximum Power Dissipation vs. Temperature
ORDERING GUIDE
Model
Temperature Range
Package Description
Package Option
AD823AN
AD823AR
AD823AR-REEL
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
8-Pin Plastic DIP
8-Pin Plastic SOIC
SOIC on Reel
N-8
SO-8
SO-8
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD823 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
REV. 0
–5–
WARNING!
ESD SENSITIVE DEVICE
AD823–Typical Characteristics
80
100
60
VS = +5V
317 UNITS
s = 0.4pA
90
VS = +5V
314 UNITS
s = 40µV
70
80
70
60
UNITS
UNITS
50
40
50
40
30
30
20
20
10
0
–200
10
–150
–100
–50
0
50
100
INPUT OFFSET VOLTAGE – µV
150
0
200
Figure 4. Typical Distribution of Input Offset Voltage
1
2
3
4
5
6
7
INPUT BIAS CURRENT – pA
8
9
10
Figure 7. Typical Distribution of Input Bias Current
22
10k
VS = +5V
–55°C TO +125°C
103 UNITS
20
18
VS = +5V
VCM = 0V
1k
INPUT BIAS CURRENT – pA
16
14
UNITS
0
12
10
8
6
4
100
10
1
2
0
–6
–5
–4
1
2
–3 –2 –1 0
3
4
5
INPUT OFFSET VOLTAGE DRIFT – µV/°C
6
0.1
7
Figure 5. Typical Distribution of Input Offset Voltage Drift
50
75
TEMPERATURE – °C
100
125
Figure 8. Input Bias Current vs. Temperature
3
1k
VS = ±15V
VS = +5V
INPUT BIAS CURRENT – pA
2
INPUT BIAS CURRENT – pA
25
0
1
0
–1
–2
100
10
1
–3
–4
–5
–4
–3
–2
–1
0
1
2
3
COMMON MODE VOLTAGE – Volts
4
0.1
–16
5
Figure 6. Input Bias Current vs. Common-Mode Voltage
–12
–8
–4
0
4
8
COMMON MODE VOLTAGE – Volts
12
16
Figure 9. Input Bias Current vs. Common-Mode Voltage
–6–
REV. 0
AD823
95
110
VS = +5V
RL = 2kΩ
94
93
OPEN-LOOP GAIN – dB
OPEN-LOOP GAIN – dB
100
VS = ± 2.5V
90
80
92
91
90
89
88
70
87
60
100
1k
100k
10k
LOAD RESISTANCE – Ω
86
–55
500k
Figure 10. Open-Loop Gain vs. Load Resistance
80
80
RL = 1kΩ
10
RL = 100Ω
1
60
60
40
40
GAIN
20
20
RL = 2kΩ
CL = 20pF
0
0.1
–2.5
–2.0
–1.5
0.5
1.0
–1.0 –0.5
0
OUTPUT VOLTAGE – Volts
1.5
2.0
–20
100
2.5
Figure 11. Open-Loop Gain vs. Output Voltage, VS = ±2.5 V
1k
PHASE MARGIN – Degrees
PHASE
OPEN-LOOP GAIN – dB
OPEN-LOOP GAIN – k V
V
125
100
100
0
10k
100k
1M
FREQUENCY – Hz
10M
–20
100M
Figure 14. Open-Loop Gain and Phase vs. Frequency
100
–40
–60
VS = +3V
VOUT = 2Vp-p
RL = 100Ω
–70
–80
VS = ±2.5V
VOUT = 2Vp-p
RL = 1kΩ
ALL
OTHERS
VS = ±15V
VOUT = 10Vp-p,
RL = 600Ω
VS = +3V,
VOUT = 2Vp-p,
RL = 5kΩ
VS = +5V
VOUT = 2Vp-p
RL = 5kΩ
–100
–110
100
INPUT VOLTAGE NOISE – nV/ √ Hz
RL = 600Ω
–50
–90
95
100
RL = 10kΩ
THD – dB
5
35
65
TEMPERATURE – °C
Figure 13. Open-Loop Gain vs. Temperature
1k
1k
10k
FREQUENCY – Hz
100k
VS = +5V
30
10
3
1M
Figure 12. Total Harmonic Distortion vs. Frequency
REV. 0
–25
10
100
1k
10k
FREQUENCY – Hz
100k
1M
Figure 15. Input Voltage Noise vs. Frequency
–7–
AD823–Typical Characteristics
90
5
G = +1
CL = 20pF
RL = 2kΩ
4
CLOSED-LOOP GAIN – dB
3
VS = ±15V
80
VS = +5V
70
2
CMRR – dB
1
0
+27°C
–1
–55°C
60
50
+125°C
–2
40
–3
30
–4
–5
0.3
3.27
6.24
9.21 12.18 15.15 18.12 21.09 24.06 27.03
FREQUENCY – MHz
20
10
30
Figure 16. Closed Loop Gain vs. Frequency
OUTPUT SATURATION VOLTAGE – Volts
OUTPUT RESISTANCE – Ω
VS = +5V
GAIN = +1
0.1
1k
10k
100k
FREQUENCY – Hz
1M
10M
1
VS – VOH
+25°C
0.1
VOL
+25°C
VOL
+25°C
1
10
LOAD CURRENT – mA
100
Figure 20. Output Saturation Voltage vs. Load Current
10
10
VS = ±15V
CL = 20pF
1%
0.1%
0.01%
4
2
0
–2
–4
1%
0.1%
+125°C
8
6
SUPPLY CURRENT – mA
OUTPUT STEP SIZE FROM 0V TO VSHOWN – Volts
1M
VS = +5V
0.01
0.1
10M
Figure 17. Output Resistance vs. Frequency, VS = 5 V,
Gain = +1
8
10k
100k
FREQUENCY – Hz
10
1.0
0.01
100
1k
Figure 19. Common-Mode Rejection vs. Frequency
100
10
100
0.01%
+25°C
6
–55°C
4
2
–6
–8
–10
100
0
200
300
400
500
SETTLING TIME – ns
600
700
0
5
10
15
SUPPLY VOLTAGE – ±Volts
20
Figure 21. Quiescent Current vs. Supply Voltage
Figure 18. Inverter Settling Time vs. Output Step Size
–8–
REV. 0
AD823
100
21
RS
VIN
18
VS = +5V
CL
80
SERIES RESISTANCE – Ω
POWER SUPPLY REJECTION – dB
90
70
+PSRR
60
50
40
–PSRR
30
15
VS = +5V
12
9
fM = 45°
6
fM = 20°
20
3
10
0
100
1k
10k
100k
FREQUENCY – Hz
1M
0
10M
Figure 22. Power Supply Rejection vs. Frequency
1
0
2
3
4
5
6
7
CAPACITOR – pF 3 1000
8
9
10
Figure 25. Capacitive Load vs. Series Resistance
30
–30
RL = 2kΩ
G = +1
VS = +5V
–40
–60
20
CROSSTALK – dB
OUTPUT VOLTAGE – Vp-p
–50
VS = ±15V
10
–70
–80
–90
–100
VS = +5V
–110
–120
VS = +3V
0
10k
100k
1M
FREQUENCY – Hz
–130
1k
10M
Figure 23. Large Signal Frequency Response
10k
100k
FREQUENCY – Hz
Figure 26. Crosstalk vs. Frequency
VS = +3V
VIN = 2.9Vp-p
G = –1
RL = 100kΩ
CL = 50pF
VS = +3V
3V
500mV
GND
500mV
10M
1M
10µs
100k
200µs
100k
+3V
VIN = 2.9V p-p
VOUT
50Ω
Figure 24. Output Swing, VS = + 3 V, G = +1
100k
50pF
Figure 27. Output Swing, VS = +3 V, G = –1
REV. 0
–9–
AD823–Typical Characteristics
5V
RL = 300Ω
CL = 50pF
RF = RG = 2kΩ
500mV
VS = ±15V
VIN = 20Vp-p
G=1
200µs
20µs
5V
GND
+15V
Figure 28. Output Swing, VS = +5 V, G = –1
20kHz, 20Vp-p
–15V
604Ω
50pF
Figure 31. Output Swing, VS = ± 15 V, G = +1
5V
RL = 2kΩ
CL = 50pF
VS = +3V
VIN = 100mV STEP
G =+1
1.55V
1.45V
25mV
500mV
50ns
100ns
GND
Figure 32. Pulse Response, VS = +5 V, G = +1
Figure 29. Pulse Response, VS = +3 V, G = +1
5V
VS = +5V
G = +1
RL = 2kΩ
CL = 470pF
VS = +5V
G =+2
RL = 2kΩ
CL = 50pF
500mV
50mV
100ns
200ns
GND
Figure 33. Pulse Response, VS = +5 V, G = +1, CL = 470 pF
Figure 30. Pulse Response, VS = +5 V , G = +2
–10–
REV. 0
AD823
RL = 100kΩ
CL = 50pF
10V
–10V
5V
500ns
Figure 34. Pulse Response, VS = ± 15 V, G = +1
of 1 mV max and offset drift of 2 µV/°C are achieved through
the use of Analog Devices’ advanced thin-film trimming
techniques.
THEORY OF OPERATION
This AD823 is fabricated on Analog Devices’ proprietary
complementary bipolar (CB) process that enables the construction of pnp and npn transistors with similar fTs in the 600 MHz
to 800 MHz region. In addition, the process also features
N-channel JFETs, which are used in the input stage of the AD823.
These process features allow the construction of high frequency,
low distortion op amps with picoampere input currents. This
design uses a differential-output input stage to maximize bandwidth and headroom (see Figure 35). The smaller signal swings
required on the S1P, S1N outputs reduce the effect of nonlinear
currents due to junction capacitances and improve the distortion
performance. With this design harmonic distortion of better
than –91 dB @ 20 kHz into 600 Ω with VOUT = 4 V p-p on a
single 5 volt supply is achieved. The complementary commonemitter design of the output stage provides excellent load drive
without the need for emitter followers, thereby improving the
output range of the device considerably with respect to conventional op amps. The AD823 can drive 20 mA with the outputs
within 0.6 V of the supply rails. The AD823 also offers outstanding precision for a high speed op amp. Input offset voltages
A “Nested Integrator” topology is used in the AD823 (see smallsignal schematic shown in Figure 36). The output stage can be
modeled as an ideal op amp with a single-pole response and a
unity-gain frequency set by transconductance gm2 and capacitor
C2. R1 is the output resistance of the input stage; gm is the input transconductance. C1 and C5 provide Miller compensation
for the overall op amp. The unity gain frequency will occur at
gm/C5. Solving the node equations for this circuit yields:
V OUT
=
Vi
A0
 g  
( sR1[C1( A2 + 1)] + 1) ×  s  m2  + 1
  C2  
where:
A0 = gmgm2R2R1 (Open Loop Gain of Op Amp)
A2 = gm2R2 (Open Loop Gain of Output Stage)
VCC
R42
R37
VBE + 0.3V V1
Q43
I5
Q55
Q44
A=1
I6
Q57
A=19
Q61
Q72
J1
Q58
Q49
Q18
Q46
J6
VINP
R44
S1N
S1P
VOUT
Q54
Q21
VINN
C2
R28
Q62
Q60
VCC
C1
VB
Q48
Q53
I1
C6
R33
Q35
I2
Q17
A=19
R43
I3
Q56
Q52
VEE
Figure 35. Simplified Schematic
REV. 0
–11–
I4
Q59
A=1
AD823
The first pole in the denominator is the dominant pole of the
amplifier, and occurs at about 18 Hz. This equals the input
stage output impedance R1 multiplied by the Miller-multiplied
value of C1. The second pole occurs at the unity-gain bandwidth of the output stage, which is 23 MHz. This type of architecture allows more open loop gain and output drive to be
obtained than a standard two-stage architecture would allow.
APPLICATION NOTES
INPUT CHARACTERISTICS
OUTPUT IMPEDANCE
The AD823 does not exhibit phase reversal for input voltages
up to and including +VS. Figure 37a shows the response of an
AD823 voltage follower to a 0 V to +5 V (+VS) square wave
input. The input and output are superimposed. The output
polarity tracks the input polarity up to +VS—no phase reversal.
The reduced bandwidth above a 4 V input causes the rounding
of the output wave form. For input voltages greater than +VS, a
resistor in series with the AD823’s plus input will prevent phase
reversal, at the expense of greater input voltage noise. This is illustrated in Figure 37b.
In the AD823, n-channel JFETs are used to provide a low
offset, low noise, high impedance input stage. Minimum input
common-mode voltage extends from 0.2 V below –VS to 1 V
less than +VS. Driving the input voltage closer to the positive
rail will cause a loss of amplifier bandwidth and increased
common-mode voltage error.
The low frequency open loop output impedance of the
common-emitter output stage used in this design is approximately 30 kΩ. While this is significantly higher than a typical
emitter follower output stage, when connected with feedback
the output impedance is reduced by the open loop gain of the
op amp. With 109 dB of open loop gain the output impedance
is reduced to less than 0.2 Ω. At higher frequencies the output
impedance will rise as the open loop gain of the op amp drops;
however, the output also becomes capacitive due to the integrator capacitors C1 and C2. This prevents the output impedance
from ever becoming excessively high (see Figure 17), which can
cause stability problems when driving capacitive loads. In fact,
the AD823 has excellent cap-load drive capability for a high frequency op amp. Figure 33 shows the AD823 connected as a follower while driving 470 pF direct capacitive load. Under these
conditions the phase margin is approximately 20°. If greater
phase margin is desired a small resistor can be used in series
with the output to decouple the effect of the load capacitance
from the op amp (see Figure 25). In addition, running the part
at higher gains will also improve the capacitive load drive capability of the op amp.
1V
2µs
100
90
10
GND
0%
1V
a. Response with RP = 0; VIN from 0 to VS
S1N
gmVI
C1
1V
R1
VOUT
S1P
+VS
C2
gmVI
R1
1V
10µs
100
90
C5
R2
gm2
10
GND
0%
Figure 36. Small Signal Schematic
1V
R
+5V
P
VIN
AD823
VOUT
b. VIN = 0 to +VS + 200 mV; VOUT = 0 to +VS; RP = 49.9 kΩ
Figure 37. AD823 Input Response
–12–
REV. 0
AD823
Figure 38 shows a schematic of an AD823 being used to drive
both the input and reference input of an AD1672, a 12-bit
3 MSPS single supply A/D converter. One amplifier is configured as a unity gain follower to drive the analog input of the
AD1672 which is configured to accept an input voltage that
ranges from 0 to 2.5 V.
Since the input stage uses n-channel JFETs, input current during normal operation is negative; the current flows out from the
input terminals. If the input voltage is driven more positive than
+VS – 0.4 V, the input current will reverse direction as internal
device junctions become forward biased. This is illustrated in
Figure 6.
A current limiting resistor should be used in series with the input of the AD823 if there is a possibility of the input voltage exceeding the positive supply by more than 300 mV, or if an input
voltage will be applied to the AD823 when ± VS = 0. The amplifier will be damaged if left in that condition for more than 10
seconds. A 1 kΩ resistor allows the amplifier to withstand up to
10 volts of continuous overvoltage, and increases the input voltage noise by a negligible amount.
+5VA
+5VA
2
8
3
VIN
49.9Ω
0.1
µF
28 19
+VCC +VDD
20
REFOUT
21
AIN1
22
AIN2
AD823
AD1672
5
VREF
(1.25V)
7
23
REFIN
24
IN COM
25
NCOMP2
26
NCOMP1
6
4
1k
The AD823 is designed for 16 nV/√Hz wideband input voltage
noise and maintains low noise performance to low frequencies
(refer to Figure 15). This noise performance, along with the
AD823’s low input current and current noise means that the
AD823 contributes negligible noise for applications with source
resistances greater than 10 kΩ and signal bandwidths greater
than 1 kHz.
1k
27
CLOCK
+5VD
0.1
µF
10µF
1
Input voltages less than –VS are a completely different story.
The amplifier can safely withstand input voltages 20 volts below
the minus supply voltage as long as the total voltage from the
positive supply to the input terminal is less than 36 volts. In
addition, the input stage typically maintains picoamp level input
currents across that input voltage range.
16
ACOM
15
13
14
12
11
10
9
8
7
6
5
4
3
2
1
10
µF
0.1µF
OTR
BIT1 (MSB)
BIT2
BIT3
BIT4
BIT5
BIT6
BIT7
BIT8
BIT9
BIT10
BIT11
BIT12 (LSB)
REF
D
COM COM
19
18
Figure 38. AD823 Driving Input and Reference of the
AD1672, a 12-Bit 3 MSPS A/D Converter
OUTPUT CHARACTERISTICS
The AD823’s unique bipolar rail-to-rail output stage swings
within 25 mV of the supplies with no external resistive load. The
AD823’s approximate output saturation resistance is 25 Ω
sourcing and sinking. This can be used to estimate output saturation voltage when driving heavier current loads. For instance,
when driving 5 mA, the saturation voltage to the rails will be approximately 125 mV.
If the AD823’s output is driven hard against the output saturation voltage, it will recover within 250 ns of the input returning
to the amplifier’s linear operating region.
A/D Driver
The rail-to-rail output of the AD823 makes it useful as an A/D
driver in a single supply system. Because it is a dual op amp, it
can be used to drive both the analog input of the A/D along with
its reference input. The high impedance FET input of the
AD823 is well suited for minimally loading of high output impedance devices.
REV. 0
10µF
0.1µF
+5VD
The other amplifier is configured as a gain of two to drive the
reference input from a 1.25 V reference. Although the AD1672
has its own internal reference, there are systems that require
greater accuracy than the internal reference provides. On the
other hand, if the AD1672 internal reference is used, the second
AD823 amplifier can be used to buffer the reference voltage for
driving other circuitry while minimally loading the reference
source.
The circuit was tested with a 500 kHz sine wave input that was
heavily low pass filtered (60 dB) to minimize the harmonic content at the input to the AD823. The digital output of the
AD1672 was analyzed by performing an FFT.
During the testing, it was observed that at 500 kHz, the output
of the AD823 cannot go below about 350 mV (operating with
negative supply at ground) without seriously degrading the second harmonic distortion. Another test was performed with a
200 Ω pull-down resistor to ground that allowed the output to
go as low as 200 mV without seriously affecting the second harmonic distortion. There was, however, a slight increase in the
third harmonic term with the resistor added, but it was still less
than the second harmonic.
–13–
AD823
+3V
Figure 39 is an FFT plot of the results of driving the AD1672
with the AD823 with no pull-down resistor. The input amplitude was 2.15 V p-p and the lower voltage excursion was
350 mV. The input frequency was 490 kHz, which was chosen
to spread the location of the harmonics.
95.3kΩ
CHANNEL 1
1µF
MYLAR
The distortion analysis is important for systems requiring good
frequency domain performance. Other systems may require
good time domain performance. The noise and settling time
performance of the AD823 will provide the necessary information for its applicability for these systems.
0.1µF
95.3k
8
3
1/2
47.5k
AD823
2
95.3k
1
0.1µF
+
500µF
L
4.99k
10k
HEADPHONES
32Ω IMPEDANCE
10k
R
4.99k
1
6
VIN = 2.15Vp-p
G = +1
FI = 490kHz
1µF
1/2
AD823
5
4
47.5k
CHANNEL 2
500µF
7
+
15dB/DIV
MYLAR
2
Figure 40. 3 Volt Single Supply Stereo Headphone Driver
4
5
6
9
7
3
8
Second Order Low-Pass Filter
Figure 41 depicts the AD823 configured as a second order
Butterworth low-pass filter. With the values as shown, the corner frequency will be 200 kHz. The equations for component
selection are shown below:
R1 = R2 = user selected (typical values: 10 kΩ to 100 kΩ).
Figure 39. FFT of AD1672 Output Driven by AD823
C1( farads ) =
3 Volt, Single Supply Stereo Headphone Driver
The AD823 exhibits good current drive and THD+N performance, even at 3 V single supplies. At 20 kHz, total harmonic
distortion plus noise (THD+N) equals –62 dB (0.079%) for a
300 mV p-p output signal. This is comparable to other single
supply op amps which consume more power and cannot run on
3 V power supplies.
In Figure 40, each channel’s input signal is coupled via a 1 µF
Mylar capacitor. Resistor dividers set the dc voltage at the noninverting inputs so that the output voltage is midway between
the power supplies (+1.5 V). The gain is 1.5. Each half of the
AD823 can then be used to drive a headphone channel. A 5 Hz
high-pass filter is realized by the 500 µF capacitors and the
headphones, which can be modeled as 32 ohm load resistors to
ground. This ensures that all signals in the audio frequency
range (20 Hz–20 kHz) are delivered to the headphones.
1.414
0.707
; C2 =
2 πfcutoff R1
2 πfcutoff R1
C2
56pF
R1
20k
+5V
C3
0.1µF
R2
20k
VIN
C1
28pF
1/2
AD823
VOUT
50pF
C4
0.1µF
–5V
Figure 41. Second Order Low-Pass Filter
A plot of the filter is shown below; better than 50 dB of high frequency rejection is provided.
–14–
REV. 0
AD823
HIGH FREQUENCY REJECTION – dB
0
put at node C is then a full-wave rectified version of the input.
Node B is a buffered half-wave rectified version of the input.
Input voltage supply to ± 18 volts can be rectified, depending on
the voltage supply used.
–10
VDB – VOUT
–20
R1
100kΩ
R2
100kΩ
–30
+VS
–40
A
–50
–60
1k
VIN
2
100k
1M
FREQUENCY – Hz
10M
6
5
1
A1
4
10k
0.01µF
8
3
1/2
AD823
A2
7
1/2
AD823
100M
C
FULL-WAVE
RECTIFIED OUTPUT
B
HALF-WAVE
RECTIFIED OUTPUT
Figure 42. Frequency Response of Filter
Single-Supply Half-Wave and Full-Wave Rectifiers
An AD823 configured as a unity gain follower and operated
with a single supply can be used as a simple half-wave rectifier.
The AD823’s inputs maintain picoamp level input currents even
when driven well below the minus supply. The rectifier puts that
behavior to good use, maintaining an input impedance of over
1011 Ω for input voltages from 1 volt from the positive supply to
20 volts below the negative supply.
The full- and half-wave rectifier shown in Figure 43 operates as
follows: when VIN is above ground, R1 is bootstrapped through
the unity gain follower A1 and the loop of amplifier A2. This
forces the inputs of A2 to be equal, thus no current flows
through R1 or R2, and the circuit output tracks the input. When
VIN is below ground, the output of A1 is forced to ground. The
noninverting input of amplifier A2 sees the ground level output
of A1, therefore, A2 operates as a unity gain inverter. The out-
REV. 0
–15–
2V
2V
200µs
100
A
90
B
10
C
0%
2V
Figure 43. Single Supply Half- and Full-Wave Rectifier
AD823
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
8-Lead Plastic DIP
(N-8)
5
0.25
(6.35)
PIN 1
1
C2035–7.5–5/95
8
0.31
(7.87)
4
0.30 (7.62)
REF
0.39 (9.91) MAX
0.035±0.01
(0.89±0.25)
0.165±0.01
(4.19±0.25)
0.011±0.003
(0.28±0.08)
0.18±0.03
(4.57±0.76)
0.125
(3.18)
MIN
0.018±0.003
(0.46±0.08)
0.10
(2.54)
BSC
0.033
(0.84)
NOM
15 °
0°
SEATING
PLANE
8-Lead Plastic SOIC
(SO-8)
5
8
0.1574 (4.00)
0.1497 (3.80)
PIN 1
0.2440 (6.20)
0.2284 (5.80)
4
1
0.1968 (5.00)
0.1890 (4.80)
0.0196 (0.50)
x 45 °
0.0099 (0.25)
0.0688 (1.75)
0.0532 (1.35)
0.0500
(1.27)
BSC
0.0192 (0.49)
0.0138 (0.35)
0.0098 (0.25)
0.0075 (0.19)
8°
0°
0.0500 (1.27)
0.0160 (0.41)
PRINTED IN U.S.A.
0.0098 (0.25)
0.0040 (0.10)
–16–
REV. 0