LINER LT3797IUKGPBF Triple output led driver controller Datasheet

LT3797
Triple Output LED Driver
Controller
DESCRIPTION
FEATURES
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Three Independent LED Driver Channels
Wide Input Voltage Range: 2.5V to 40V
VIN Transient Ride-Through Up to 60V
Rail-to-Rail LED Current Sense: 0V to 100V
3000:1 PWM Dimming
TG Drivers for PMOS LED Disconnection
Operates in Boost, Buck Mode, Buck-Boost Mode,
SEPIC or Flyback Topology
Open-LED Protection
Short-Circuit Protected Boost Capable
Fault Flags for Independent Channels
Programmable VIN Undervoltage and Overvoltage
Lockout
Adjustable Switching Frequency: 100kHz to 1MHz
Synchronizeable to an External Clock
CTRL Pins Provide Analog Dimming
Programmable Soft-Start
52-Lead QFN Package
The LT®3797 is a triple output DC/DC controller designed
to drive three strings of LEDs. The fixed frequency, current
mode architecture results in stable operation over a wide
range of supply and output voltages. The LT3797 includes
an integrated DC/DC converter to produce a regulated 7.5V
supply for the N-channel MOSFET gate drivers of the three
channels. This high efficiency converter enables the part to
operate from a wide input voltage range from 2.5V to 40V.
The LT3797 is designed so that each converter can use the
most suitable configuration to drive its LED load, whether
step-up, step-down or a combination. Two key features
enable this flexibility: first the LT3797 can sense output
current at the high side of the LED string; and second, the
voltage feedback pin, FBH, is referred to the ISP current
sensing input. The CTRL inputs provide output current
analog dimming capability. The TG drivers level shift the
PWM signals to drive the gates of external LED-disconnect
P-channel MOSFETs, allowing high PWM dimming range,
and providing LED overcurrent protection and short-circuit
protected boost capability.
APPLICATIONS
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Automotive and Industrial Lighting
RGB Lighting
Billboards and Large Displays
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear
Technology Corporation. All other trademarks are the property of their respective owners.
Protected by U.S. Patents, including 7199560, 7321203, 7746300.
TYPICAL APPLICATION
VIN
2.5V TO 40V
(60V TRANSIENT,
41V INTERNAL
OVLO PROTECTION)
Triple Boost LED Driver
4.7µF
×3
10µH
10µH
4.7µF
100V
×2
ISP1
250mΩ
ISP2
250mΩ
ISN1
GATE1 SENSEP1 SENSEN1 TG1
VIN
1µF
250mΩ
GATE2 SENSEP2 SENSEN2 TG2
4.7µF
100V
×2
ISN3
1A
50V
0.1µF 8mΩ
47.5k
0.1µF 8mΩ
GATE3 SENSEP3 SENSEN3 TG3
1A
50V
ISN1-3
ISP1-3
20.5k
LT3797
EN/UVLO
49.9k
ISP3
ISN2
1A
50V
0.1µF 8mΩ
10µH
4.7µF
100V
×2
FBH1-3
OVLO VREF CTRL1-3 PWM1-3
VIN
RT
SYNC SS1-3
48.7k
300kHz
487k
FLT1-3 SW1 SW2 BOOST
0.1µF
47µH
75k
INTVCC GND
10µF
0.1µF
VC1-3
4.7k
1M
10nF
3797 TA01
3797f
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1
LT3797
SS3
SENSEN3
SENSEP3
GATE3
INTVCC
INTVCC
SW2
BOOST
SW1
VIN
EN/UVLO
TOP VIEW
OVLO
52 51 50 49 48 47 46 45 44 43 42 41
FLT1 1
40 VC3
FLT2 2
FLT3 3
38 FBH3
PWM1 4
37 ISP3
PWM2 5
36 ISN3
PWM3 6
35 TG3
VREF 7
53
GND
CTRL1 8
33 TG2
CTRL2 9
32 ISN2
CTRL3 10
31 ISP2
RT 11
30 FBH2
SYNC 12
28 VC2
27 SS2
SENSEN2
SENSEP2
GATE2
GATE1
SENSEP1
SENSEN1
19 20 21 22 23 24 25 26
SS1
15 16 17
VC1
TG1 14
FBH1
VIN, EN/UVLO.............................................................60V
INTVCC, SYNC, OVLO, PWM1, PWM2, PWM3.............8V
ISN1........................................................ISP1-1.5V, 100V
ISN2........................................................ISP2-1.5V, 100V
ISN3........................................................ISP3-1.5V, 100V
FBH1....................................................... ISP1 ±6V, 100V
FBH2....................................................... ISP2 ±6V, 100V
FBH3....................................................... ISP3 ±6V, 100V
VC1, VC2, VC3, VREF, SS1, SS2, SS3...........................3V
CTRL1, CTRL2, CTRL3, FLT1, FLT2, FLT3...................12V
RT.............................................................................1.5V
SENSEP1, SENSEP2, SENSEP3, SENSEN1,
SENSEN2, SENSEN3,..............................................±0.3V
SW1, SW2, BOOST, TG1, TG2, TG3, GATE1,
GATE2, GATE3 ................................................... (Note 2)
Operating Ambient Temperature Range
(Note 3)....................................................... –40 to 125°C
Maximum Junction Temperature........................... 125°C
Storage Temperature Range................... –65°C to 150°C
PIN CONFIGURATION
ISP1
(Note 1)
ISN1
ABSOLUTE MAXIMUM RATINGS
UKG PACKAGE
VARIATION: UKG52(47)
52-LEAD (7mm × 8mm) PLASTIC QFN
θJA = 28°C/W
EXPOSED PAD (PIN 53) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3797EUKG#PBF
LT3797EUKG#TRPBF
LT3797UKG
52-Lead (7mm × 8mm) Plastic QFN
–40°C to 125°C
LT3797IUKG#PBF
LT3797IUKG#TRPBF
LT3797UKG
52-Lead (7mm × 8mm) Plastic QFN
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3797f
2
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LT3797
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V; EN/UVLO = 24V; CTRL1, CTRL2, CTRL3, PWM1, PWM2,
PWM3 = 2V; SENSEN1, SENSEN2, SENSEN3 = 0V, unless otherwise noted.
PARAMETER
CONDITIONS
VIN Minimum Operation Voltage
MIN
TYP
MAX
2.5
V
40
41
1
42.5
V
V
0.1
1
15
µA
µA
l
VIN Overvoltage Lockout
Rising VIN
Falling Hysteresis
VIN Shutdown IQ
EN/UVLO = 0V
EN/UVLO = 1.15V
l
UNITS
VIN Operating IQ (Not Switching)
PWM1, PWM2, PWM3 = 0V, INTVCC = 8V
0.5
0.75
mA
INTVCC Operating IQ (Not Switching)
PWM1, PWM2, PWM3 = 0V, INTVCC = 8V
2.4
3
mA
VREF Voltage
0µA ≤ IVREF ≤ 450µA
2.00
2.035
VREF Line Regulation
2.5V ≤ VIN ≤ 40V
SENSEP1-SENSEN1, SENSEP2-SENSEN2,
SENSEP2-SENSEN2 Current Limit Threshold
l
1.955
l
100
0.001
110
V
%/V
120
mV
SENSEP1, SENSEP2, SENSEP3 Input Bias Current Current Out of Pin, SENSEP1, SENSEP2,
SENSEP3 = 0V
55
μA
SENSEN1, SENSEN2, SENSEN3 Input Bias Current Current Out of Pin
210
μA
Integrated INTVCC Power Supply
INTVCC Regulation Voltage
l
INTVCC Undervoltage Lockout Threshold
Falling INTVCC
Hysteresis
INTVCC Line Regulation (ΔVINTVCC/ΔVIN)
2.5V < VIN < 40V
7.15
7.5
7.75
V
5.15
5.25
0.4
5.4
V
V
0.001
0.02
%
243
238
250
250
257
272
mV
mV
Error Amplifiers
LED Current Sense Threshold (ISP1-ISN1,
ISP2-ISN2, ISP3-ISN3)
ISP1, ISP2, ISP3, FBH1, FBH2, FBH3 = 48V
ISN1, ISN2, ISN3, FBH1, FBH2, FBH3 = 0V
8/10th LED Current Sense Threshold (ISP1-ISN1,
ISP2-ISN2, ISP3-ISN3)
CTRL1, CTRL2, CTRL3=1.1V, ISP1, ISP2, ISP3 = 48V l
CTRL1, CTRL2, CTRL3=1.1V, ISN1, ISN2, ISN3 = 0V l
194.5
192
200
200
203.5
218
mV
mV
1/10th LED Current Sense Threshold (ISP1-ISN1,
ISP2-ISN2, ISP3-ISN3)
CTRL1, CTRL2, CTRL3=0.3V, ISP1, ISP2, ISP3 = 48V l
CTRL1, CTRL2, CTRL3=0.3V, ISN1, ISN2, ISN3 = 0V l
17
15
25
25
29
34
mV
mV
l
0.2
1.2
V
50
100
nA
150
20
170
mV
mV
100
V
CTRL1, CTRL2, CTRL3 Range for Linear Current
Sense Threshold Adjustment
CTRL1, CTRL2, CTRL3 Input Bias Current
Current Out of Pin, CTRL1, CTRL2, CTRL3 = 0.3V
CTRL1, CTRL2, CTRL3 Idle Mode Threshold
Falling
Hysteresis
LED Current Sense Amplifier Input Common Mode
Range (ISN1, ISN2, ISN3)
l
l
135
l
0
LED Overcurrent Protection Threshold
(ISP1-ISN1, ISP2-ISN2, ISP3-ISN3)
ISP1, ISP2, ISP3, FBH1, FBH2, FBH3 = 48V
1000
mV
ISP1, ISP2, ISP3 Input Bias Current (Active)
ISP1, ISP2, ISP3, ISN1, ISN2, ISN3 = 48V
ISP1, ISP2, ISP3, ISN1, ISN2, ISN3 = 0V
630
–100
µA
nA
ISP1, ISP2, ISP3 Input Bias Current (Idle)
PWM1, PWM2, PWM3=0V , ISP1, ISP2, ISP3, ISN1,
ISN2, ISN3 = 48V
2
µA
PWM1, PWM2, PWM3, ISP1, ISP2, ISP3, ISN1,
ISN2, ISN3 = 0V
–40
nA
ISP1, ISP2, ISP3, ISN1, ISN2, ISN3 = 48V
ISP1, ISP2, ISP3, ISN1, ISN2, ISN3 = 0V
20
–100
µA
nA
ISN1, ISN2, ISN3 Input Bias Current (Active)
3797f
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LT3797
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V; EN/UVLO = 24V; CTRL1, CTRL2, CTRL3, PWM1, PWM2,
PWM3 = 2V; SENSEN1, SENSEN2, SENSEN3 = 0V, unless otherwise noted.
PARAMETER
CONDITIONS
ISN1, ISN2, ISN3 Input Bias Current (Idle)
PWM1, PWM2, PWM3=0V , ISP1, ISP2, ISP3, ISN1,
ISN2, ISN3 = 48V
LED Current Sense Amplifier gm
MIN
MAX
0
1
UNITS
µA
PWM1, PWM2, PWM3, ISP1, ISP2, ISP3, ISN1,
ISN2, ISN3 = 0V
–20
nA
ISP1-ISN1, ISP2-ISN2, ISP3-ISN3 = 250mV
250
μS
FBH1, FBH2, FBH3 Regulation Voltage “FBH(REG)” ISP1, ISP2, ISP3, ISN1, ISN2, ISN3 = 48V
(|ISP1-FBH1, ISP2-FBH2, ISP3-FBH3|)
FBH1, FBH2, FBH3 Pin Input Bias Current
ISP1-FBH1, ISP2-FBH2, ISP3-FBH3 = 1.25V
ISP1-FBH1, ISP2-FBH2, ISP3-FBH3 = –1.25V
FBH1, FBH2, FBH3 Amplifier gm
|ISP1-FBH1|, |ISP2-FBH2|, |ISP3-FBH3| = 1.25V
FBH1, FBH2, FBH3 Open-LED Threshold
(|ISP1-FBH1|, |ISP2-FBH2|, |ISP3-FBH3|) Voltage
Rising (Note 4)
FBH1, FBH2, FBH3 Overvoltage Threshold
(|ISP1-FBH1|, |ISP2-FBH2|, |ISP3-FBH3|) Voltage
Rising (Note 4)
l
1.225
1.250
1.275
V
2
40
2.4
100
3
nA
µA
480
μS
FBH(REG) FBH(REG) FBH(REG)
– 0.07
– 0.05
– 0.04
V
Hysteresis
20
mV
FBH(REG) FBH(REG) FBH(REG)
+ 0.05
+ 0.06
+ 0.085
Hysteresis
VC1, VC2, VC3 Output Impedance
VC1, VC2, VC3 Standby Input Bias Current
TYP
PWM1, PWM2, PWM3 = 0V
CTRL1, CTRL2, CTRL3 = 0V
25
mV
10
MΩ
–20
–20
VC1, VC2, VC3 Current Mode Gain –ΔVVC/ΔVSENSE
V
20
20
nA
nA
4
V/V
10.5
µA
ISP1, ISP2, ISP3, FBH1, FBH2, FBH3 = 48V,
ISN1, ISN2, ISN3 = 47.7V
12
µA
ISP1, ISP2, ISP3, ISN1, ISN2, ISN3 = 48V,
FBH1, FBH2, FBH3 = 46.7V
32
µA
VC1, VC2, VC3 Source Current
ISP1, ISP2, ISP3, ISN1, ISN2, ISN3, FBH1, FBH2,
FBH3 = 48V, Current Out of Pin
VC1, VC2, VC3 Sink Current
Oscillator
Switching Frequency
RT = 140kΩ
RT = 34.0kΩ
RT = 10.7kΩ
l
95
375
950
RT Voltage
100
400
1000
107
425
1050
1.05
kHz
kHz
kHz
V
GATE1, GATE2, GATE3 Minimum Off-Time
CGATE = 3300pF
200
270
ns
GATE1, GATE2, GATE3 Minimum On-Time
CGATE = 3300pF
220
300
ns
0.4
V
SYNC Input Low
l
SYNC Input High
l
1.5
SYNC Resistance to GND
V
200
kΩ
Logic Inputs/Outputs
EN/UVLO Threshold Voltage Falling
l
1.180
EN/UVLO Rising Hysteresis
1.250
20
EN/UVLO Input Low Voltage
IVIN Drops Below 1µA
EN/UVLO Pin Bias Current Low
EN/UVLO = 1.15V
EN/UVLO Pin Bias Current High
EN/UVLO = 1.33V
l
1.5
OVLO Pin Input Bias Current
OVLO Threshold Voltage
1.220
Rising
Hysteresis
l
1.225
V
mV
0.4
V
2
2.6
µA
40
100
nA
20
100
nA
1.250
125
1.275
V
mV
3797f
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LT3797
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V; EN/UVLO = 24V; CTRL1, CTRL2, CTRL3, PWM1, PWM2,
PWM3 = 2V; SENSEN1, SENSEN2, SENSEN3 = 0V, unless otherwise noted.
PARAMETER
CONDITIONS
MIN
PWM1, PWM2, PWM3 Input High Voltage
l
PWM1, PWM2, PWM3 Input Low Voltage
l
0.6
PWM1, PWM2, PWM3 Resistance to GND
TYP
MAX
1.1
1.4
UNITS
V
0.9
V
200
kΩ
FLT1, FLT2, FLT3 Output Low
IFLT =1mA
SS1, SS2, SS3 Sourcing Current
SS1, SS2, SS3 = 1V, Current Out of Pin
28
300
mV
µA
SS1, SS2, SS3 Sinking Current
SS1, SS2, SS3 = 1V, OVLO =1.3V
2.8
µA
SS1, SS2, SS3 Soft-Start Reset Threshold
Falling, Measured on SS1, SS2, SS3
Hysteresis
160
30
mV
mV
SS1, SS2, SS3 Fault Reset Threshold
Measured on SS1, SS2, SS3
1.7
V
GATE1, GATE2, GATE3 Output Rise Time (tr)
CGATE = 3300pF (Note 5)
25
ns
GATE1, GATE2, GATE3 Output Fall Time (tf)
CGATE = 3300pF (Note 5)
25
NMOS Gate Drivers
Gate Output Low (VOL)
ns
0.1
Gate Output High (VOH)
V
INTVCC –
0.05
V
PMOS Gate Drivers
TG1, TG2, TG3 Turn-On Time
CTG = 1000pF (Note 6)
200
ns
TG1, TG2, TG3 Turn-Off Time
CTG = 1000pF (Note 6)
70
ns
6.5
V
PMOS Gate On Voltage
(ISP1-TG1, ISP2-TG2, ISP3-TG3)
PMOS Gate Off Voltage
(ISP1-TG1, ISP2-TG2, ISP3-TG3)
V
0.3
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: Do not apply a positive or negative voltage or current source
to SW1, SW2, GATE1, GATE2, GATE3, TG1, TG2, TG3 pins, otherwise
permanent damage may occur.
Note 3: The LT3797E is guaranteed to meet performance specifications
from the 0°C to 125°C junction temperature. Specifications over the
–40°C to 125°C operating junction temperature range are assured by
design, characterization and correlation with statistical process controls.
The LT3797I is guaranteed over the full –40°C to 125°C operating junction
temperature range.
Note 4: FBH(REG) denotes the regulation voltage (|ISP-FBH|) of the
corresponding FBH pin.
Note 5: Rise and fall times are measured at 10% and 90% levels.
Note 6: Gate turn-on/turn-off time is measured from 50% level of PWM
voltage to 90% level of gate on/off voltage.
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LT3797
TYPICAL PERFORMANCE CHARACTERISTICS
VISP-ISN Threshold vs VCTRL
200
150
100
50
254
254
253
253
V(ISP-ISN) THRESHOLD (mV)
V(ISP-ISN) THRESHOLD (mV)
250
VISP-ISN THRESHOLD (mV)
VISP-ISN Full-Scale Threshold
vs Temperature
VISP-ISN Threshold vs VISP
300
0
TA = 25°C unless otherwise noted.
252
251
250
249
248
0.2
0.4
0.6 0.8 1.0
VCTRL (V)
1.2
1.4
246
1.6
0
20
40
60
VISP (V)
80
3797 G01
203
700
198
ISP
600
500
400
300
200
100
197
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
248
300
0
125
V(ISP-ISN) THRESHOLD (mV)
ISP, ISN BIAS CURRENT (µA)
VISP-ISN THRESHOLD (mV)
199
249
75
50
25
TEMPERATURE (°C)
0
ISN
0
20
40
60
VISP, VISN (V)
80
3797 F04
2.04
1.260
125
VISP = 48V
250
200
150
100
50
0
100
VISP-ISN Threshold vs V|ISP-FBH|
1.1
1.15
1.2
V|ISP-FBH| (V)
1.25
1.3
3797 G06
3797 G05
|ISP-FBH| Regulation Voltage
vs Temperature, VISP
100
3797 G03
800
200
250
246
–50 –25
100
ISP/ISN Input Bias Current
vs VISP, VISN
201
251
3797 G02
VISP-ISN Threshold at
CTRL = 0.7V vs Temperature
202
252
247
247
0
VISP = 48V
VREF Voltage vs Temperature
VREF Voltage vs VIN
2.010
2.03
2.02
VISP = 48V
2.005
2.01
VREF (V)
V|ISP-FBH| (V)
VISP = 4.5V
1.250
VISP = 100V
IREF = 0µA
2.00
IREF = 450µA
1.99
1.245
VREF (V)
1.255
2.000
1.995
1.98
1.97
1.240
–50
–25
75
0
25
50
TEMPERATURE (°C)
100
125
3797 G07
1.96
–50 –25
75
50
25
TEMPERATURE (°C)
0
100
125
3797 G08
1.990
0
5
10
15
20 25
VIN (V)
30
35
40
3797 G09
3797f
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LT3797
TYPICAL PERFORMANCE CHARACTERISTICS
TA = 25°C unless otherwise noted.
Switching Frequency
vs Temperature
RT vs Switching Frequency
VIN, INTVCC Quiescent Current
vs VIN
420
VIN, INTVCC QUIESCENT CURRENT (mA)
2.5
SWITCHING FREQUENCY (kHz)
415
RT (kΩ)
100
410
405
400
395
390
385
380
–50 –25
0 100 200 300 400 500 600 700 800 900 1000
SWITCHING FREQUENCY (kHz)
75
50
25
TEMPERATURE (°C)
100
0
3797 G10
EN/UVLO (V)
OVLO (V)
1.21
1.19
1.17
1.15
1.23
EN/UVLO RISING THRESHOLD
1.22
EN/UVLO FALLING THRESHOLD
1.21
1.20
–25
0
25
50
75
TEMPERATURE (°C)
100
125
1.19
–50 –25
50
25
75
0
TEMPERATURE (°C)
10
15
20 25
VIN (V)
30
35
100
2.0
1.8
1.6
–50
125
–25
75
0
25
50
TEMPERATURE (°C)
100
SENSE Current Limit Threshold
vs Duty Cycle
115
2.5
125
3797 G15
SENSE Current Limit Threshold
vs Temperature
EN/UVLO Current vs Voltage
40
2.2
3797 G14
3797 G13
115
114
2.0
1.5
1.0
0.5
V(SENSEP-SENSEN) (mV)
113
V(SENSEP-SENSEN) (mV)
EN/UVLO CURRENT (µA)
5
2.4
OVLO FALLING THRESHOLD
1.11
0
EN/UVLO Hysteresis Current
vs Temperature
1.24
OVLO RISING THRESHOLD
1.09
–50
IVIN
0.5
3797 G12
1.25
1.13
1.0
EN/UVLO Falling/Rising
Threshold vs Temperature
1.27
1.23
1.5
3797 G11
OVLO Threshold vs Temperature
1.25
IINTVCC
2.0
0
125
EN/UVLO HYSTERESIS CURRENT (µA)
10
PWM = 0V
112
111
110
109
108
107
0
110
105
100
106
–0.5
0.122
105
–50
1.22
12.2
EN/UVLO VOLTAGE (V)
3797 G16
–25
50
25
0
75
TEMPERATURE (°C)
100
125
3797 G17
95
0
20
60
40
DUTY CYCLE
80
100
3797 G18
3797f
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LT3797
TYPICAL PERFORMANCE CHARACTERISTICS
INTVCC vs Temperature, VIN
INTVCC CURRENT LIMIT IINTVCC_LMT (mA)
INTVCC (V)
VIN = 24V
7.500
VIN = 40V
7.475
–25
75
0
25
50
TEMPERATURE (°C)
100
125
225
L = 47µH
1200
800kHz 700kHz >900kHz
200
175
600kHz
150
500kHz
400kHz
125
300kHz
100
75
200kHz
50
800
FALL TIME
600
400
200
RISE TIME
100kHz
25
0
1000
TIME (ns)
250
7.525
7.450
–50
Top Gate (PMOS) Rise/Fall Time
vs Capacitance
INTVCC Current Limit vs VIN, fSW
7.550
VIN = 2.5V
TA = 25°C unless otherwise noted.
0 3 6 9 12 15 18 21 24 27 30 33 36 39
VIN (V)
0
0
1
2
3 4 5 6 7
CAPACITANCE (nF)
3739 G20
3797 G19
8
9
10
3797 G21
PIN FUNCTIONS
FLT1, FLT2, FLT3 (Pins 1, 2, 3): Open-Collector Pull-Downs
on FLT Pins Report The Fault Conditions:
1. VIN > 41V (typical)
2. Overtemperature (TJ > 165°C)
3. INTVCC < 5.2V (typical)
4. OVLO > 1.25V (typical)
5. LED Overcurrent
6. Open LED
7. Output Overvoltage
CTRL1, CTRL2, CTRL3 (Pins 8, 9, 10): Current Sense
Threshold Adjustment Pins. Sets voltage across external
sense resistor between ISP and ISN pins of the respective
converter:
VISP-ISN = 0V, when VCTRL < 0.2V
VISP-ISN = (VCTRL – 0.2V)/4, when 0.2V < VCTRL < 1.2V
VISP-ISN = 250mV, when VCTRL >1.2V
PWM1, PWM2, PWM3 (Pins 4, 5, 6): Pulse Width Modulated Input Pins. Signal low causes the respective converter
to go into idle mode which means it stops switching, the
TG pin transitions high, the quiescent currents are reduced,
and the VC becomes high impedance. If not used, connect
to the REF pin.
VREF (Pin 7): Reference Output Pin. Can supply up to
450µA. This pin drives a resistor divider for the CTRL1,
CTRL2, CTRL3 pins, either for analog dimming or for
temperature limit/compensation of LED loads. The normal
output voltage is 2V.
Connect CTRL pins to VREF for the 250mV default threshold.
When VCTRL < 150mV (typical), the respective converter
goes into idle mode, which is the same as PWM pin being
pulled low. Do not leave these pins open.
RT (Pin 11): Switching Frequency Adjustment Pin. Set
the frequency using a resistor to GND. Do not leave the
RT pin open.
SYNC (Pin 12): The SYNC pin is used to synchronize the
internal oscillator to an external logic-level signal. The RT
resistor should be chosen to program an internal switching
frequency 20% slower than the SYNC pulse frequency. Gate
turn-on occurs at a 0.2µs (typical) delay after the rising
edge of SYNC. Tie SYNC to GND if not used.
3797f
8
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LT3797
PIN FUNCTIONS
TG1, TG2, TG3 (Pins 14, 33, 35): Top Gate Driver Output Pins for Driving LED Loads Disconnect P-Channel
MOSFETs (PMOSs). One for each channel. An inverted
PWM signal drives an external PMOS gate of the respective converter between VISP and (VISP – 6.5V). Leave TG
pins unconnected if not used.
ISN1, ISN2, ISN3 (Pins 15, 32, 36): Connection Points for
the Negative Terminals of the Current Feedback Resistors.
ISP1, ISP2, ISP3 (Pins 16, 31, 37): Connection Points
for the Positive Terminals of the Current Feedback Resistors. Also serves as positive rails for TG pin drivers and
the reference point for FBH.
FBH1, FBH2, FBH3 (Pins 17, 30, 38): Voltage Loop Feedback Pins. The output feedback voltage VFB is measured
between the ISP pin and the FBH pin (absolute value):
VFB = |ISP – FBH|. The FBH pin is intended for constantvoltage regulation or for LED protection/open-LED detection for each channel. In an open-LED event, the internal
amplifier with output VC regulates VFB to 1.25V (typical)
through the respective converter. If VFB is above the overvoltage threshold (typical 1.3V), the TG pin of the same
channel is driven high to disconnect the external PMOS to
protect the LEDs from an overvoltage event. Either openLED or overvoltage event signals a fault condition. Do
not leave the FBH pins open. It requires ISP to be no less
than 4.5V to maintain an accurate VFB1 voltage sense. If
ISP falls below 4.5V, the voltage regulation is deactivated
and the ISP-ISN current regulation dominates regardless
of the |ISP-FBH| value. If not used, connect the FBH pin
to the ISP pin of the same channel.
VC1, VC2, VC3 (Pins 19, 28, 40): Error Amplifier Compensation Pins. Connect a series RC from each VC pin to
GND. In each channel, the VC pin is high impedance when
the PWM pin is low, or the CTRL pin is below 150mV. This
feature allows the VC pin to store the demand current
state variable for the next PWM or CTRL high transition.
SS1, SS2, SS3 (Pins 20, 27, 41): Soft-Start Pins. Each
SS pin modulates compensation VC pin voltage of the
respective channel. Each of the soft-start intervals is set
with an external capacitor.
SENSEN1, SENSEN2, SENSEN3 (Pins 21, 26, 42): The
Negative Current Sense Inputs for the Control Loops. Kelvin
connect the SENSEN pin to the negative terminal of the
switch current sense resistor (which connects to the GND
plane) of the respective converter. Connect SENSEN pin
to SENSEP pin of the same channel with a 0.1µF ceramic
capacitor placed close to pins.
SENSEP1, SENSEP2, SENSEP3 (Pins 22, 25, 43): The
Positive Current Sense Inputs for the Control Loops.
Kelvin connect the SENSEP pin to the positive terminal
of the switch current sense resistor in the source of the
external N-channel MOSFET (NMOS) switch of the respective converter. Connect SENSEP pin to SENSEN pin of
the same channel with a 0.1µF ceramic capacitor placed
close to pins.
GATE1, GATE2, GATE3 (Pins 23, 24, 44): N-Channel
MOSFET Gate Driver Outputs. Switch between INTVCC and
GND. Driven to GND during shutdown, fault or idle states.
INTVCC (Pins 45, 46): INTVCC pins are the integrated power
supply output voltage nodes that provide supply for control
circuits and NMOS gate drivers. The two INTVCC pins are
internally shorted. Must be bypassed with a 10µF ceramic
capacitor placed close to the pins.
SW2 (Pin 47): Integrated Power Supply Switch Node.
Connect this pin to one side of the integrated power supply inductor.
BOOST (Pin 48): Connect this pin to SW1 pin through a
0.1µF ceramic capacitor.
SW1 (Pin 49): Integrated Power Supply Switch Node.
Connect this pin to the other side of the integrated power
supply inductor, and to the BOOST pin with a 0.1µF ceramic capacitor.
VIN (Pin 50): Input Supply Pin. If VIN is over 41V (typical),
the integrated INTVCC power supply is turned off. All three
channels are also turned off (including pulling the GATE
pins to GND and TG pins to ISP) and the soft-starts are
reset. Must be locally bypassed with low ESR capacitors
placed close to the pin.
3797f
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9
LT3797
PIN FUNCTIONS
EN/UVLO (Pin 51): Enable and Undervoltage Lockout
Pin. An accurate 1.22V falling threshold with externally
programmable hysteresis detects when power is OK to enable the integrated INTVCC power supply and each channel
switching. Rising hysteresis is generated by the external
resistor divider and an accurate internal 2μA pull-down
current. Above the 1.24V (typical) rising threshold (but
below 2.5V), EN/UVLO input bias current is sub-μA. Below
the 1.22V (typical) falling threshold, a 2μA pull-down current is enabled so the user can define the hysteresis with
the external resistor selection. An undervoltage condition
turns off the integrated INTVCC power supply and all the
three channels and resets the soft-starts. Tie to 0.4V, or
less, to disable the device and reduce VIN quiescent current below 1μA.
OVLO (Pin 52): Overvoltage Lockout Pin. An accurate
1.25V rising threshold with 125mV hysteresis detects an
overvoltage condition. An overvoltage condition turns off
all three channels (including pulling the GATE pins to GND
and TG pins to ISP) and resets the soft-starts. Tie OVLO
to GND if not used.
GND (Exposed Pad Pin 53): Ground. Solder the exposed
pad directly to ground plane.
3797f
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LT3797
BLOCK DIAGRAM
D1
VOUT
R6
RSW_SEN
R5
–
VFB1
+
17
16 15
|ISP1 – FBH1|
3V
CSS
8
20
OVFB
12µA
S1
1.2V
–
+
+
A8
A9
1.25V
–
+
A11
150mV
+
–
CTRL1
SS1
12µA AT
A8+ = A8–
12µA AT
A9+ = A9–
Q4
51
S
G2
O
GATE1
M1
23
SET
1mA
SHDN
FLT1
A13
VISENSE1
2.5µA
+
–
110mV
SENSEP1
+
–
SENSEN1
FAULT
PROTECTION
AND REPORT
Q1
FLT2
CH1 FLT
CH2 FLT
CH3 FLT
INTVCC
UVLO
Q2
FLT3
EN/UVLO
1.22V
IS1
2µA
+
–
A1
22
CSEN
RSW_SEN
21
A3
+ –
SET
VIN
VIN
RAMP
5.7V
50kHz TO 1MHz
OSCILLATOR
INTVCC SET
1.05V
+
–
BOOST
SW1
RAMP
GENERATOR
2V
A2
SHDN
41V
165°C THERMAL
SHUTDOWN
Q3
R2
R
REPLICATED FOR EACH CHANNEL
VIN
R1
G1
A14
CTRL_ON
INTVCC
SR1
A12
S2
+
–
3
+
–
25µA
+
–
2
PROTECTION
3V
VIN
OVLO
1
CTRL_ON
PWMON
CH1 FLT
VC1
RC
CC
1.3V
A16
–
+
CVIN2
ISP1-6.5V
G3
CH1 SOFT-START
AND
FAULT PROTECTION
L1
PWM1
OVI
x4 + 0.2V
PWMON
19
A7
G4
ISP1
PWM1 SS1 SHDN
VIN
4
TG1
+ –
A6
COUT
14
ISN1
FBH1 ISP1
LED
STRING
M2
INTEGRATED
POWER
SUPPLY
48
49
VIN
CVIN1
CBOOST
LPWR
SW2
47
INTVCC
45, 46
GND
A5
50
CVCC
53
A4
– +
1.25V
OVLO
52
R4
VREF
7
R3
RT
11
SYNC
12
SHARED COMPONENTS
3797 F01
VIN
RT
Figure 1. LT3797 Block Diagram Working in Boost Configuration (for Simplicity, Only Channel 1 Is Shown)
3797f
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11
LT3797
OPERATION
The LT3797 uses a fixed frequency, current mode control
scheme to provide excellent line and load regulation. It
contains three independent switching regulators. Operation
can be best understood by referring to the Block Diagram
in Figure 1. The oscillator, internal power supply etc., are
shared among the three converters. The LED current control
circuitry, gate drivers etc., are replicated for each of the
three converters. For simplicity, Figure 1 shows the shared
circuits and the channel specific circuits for converter 1.
The LT3797 provides the constant voltage regulation mode
to allow the users to accurately program the output regulation voltage in an open-LED event. In voltage regulation
mode, the operation is similar to that described above,
except the VC1 voltage is set by A9 and is an amplified
version of the difference between the internal reference of
1.25V (typical) and the output feedback voltage, VFB1, which
is measured between ISP1 and FBH1 (the absolute value):
The LED current regulation can be understood by following
the operation of converter 1. The start of each oscillator
cycle sets the SR latch SR1 and turns on the external
power MOSFET switch M1 through gate driver G2 (the
three converters share the same oscillator, which means
if all the three channels are enabled the GATE pins of all
the three channels transition high at the same instant). The
switch current flows through the external current sensing
resistor RSW_SEN1 and generates a voltage proportional to
the switch current. This current sense voltage (amplified
by A14) is added to a stabilizing slope compensation ramp
and the resulting sum VISENSE1 is fed into the negative
terminal of the PWM comparator A12. The current in the
external inductor L1 increases steadily during the time
the switch is on. When VISENSE1 exceeds the level at the
negative input of A12 (VC1), SR1 is reset, turning off the
power switch. During the switch-off phase, L1 current
decreases.
The LED current sense feedback interacts with the FBH1
voltage feedback so that the sense voltage between ISP1
and ISN1 does not exceed the threshold set by the CTRL1
pin, and VFB1 does not exceed 1.25V (typical).
Through this repetitive action, the PWM control algorithm
establishes a switch duty cycle to regulate a current in the
LED string. The VC1 voltage is set by the error amplifier
A8 and is an amplified version of the difference between
the LED current sense voltage, measured between ISP1
and ISN1, and the target difference voltage set by the
CTRL1 pin. In this manner, the error amplifier sets the
correct switch peak current level to keep the LED current
in regulation.
The LT3797 has a switch current limit function. The switch
current sense signal is input to the current limit comparator
A13. If the current sense voltage is higher than the sense
current limit threshold, VSENSE(MAX) (typical 110mV), A13
will reset SR1 and turn off M1 immediately.
VFB1 = |ISP1-FBH1|
For accurate current or voltage regulation, it is necessary
to be sure that under normal operating conditions, the appropriate loop is dominant. To deactivate the voltage loop
entirely, FBH1 can be connected to ISP1. To deactivate the
LED current loop entirely, the ISP1 and ISN1 should be
tied together and the CTRL1 input tied to VREF.
It requires ISP to be no less than 4.5V to maintain an
accurate VFB1 voltage sense. If ISP falls below 4.5V, the
voltage regulation is deactivated and the current regulation
dominates regardless of the |ISP1-FBH1| value.
Two LED driver specific functions featured on the LT3797
are controlled by the voltage feedback pin FBH1. First,
when the VFB1 exceeds a voltage 50mV lower (–4%) than
the VFB1 regulation voltage (typical 1.25V), it indicates that
the LED may be disconnected and the constant-voltage
feedback loop is taking control of the switching regulator.
FLT1 is pulled low to report a fault condition. Second, when
VFB1 exceeds the VFB1 regulation voltage by 60mV (5%
typical), it indicates an output overvoltage fault. In this
condition, TG1 pin is driven high by G3 and G4, turning off
the external PMOS M2. This action disconnects the LED
load from the power path, preventing excessive current
from damaging the LEDs. FLT1 is kept low to report the
fault condition.
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LT3797
APPLICATIONS INFORMATION
Switching Frequency and Synchronization
Duty Cycle Considerations
The RT frequency adjust pin allows the user to program
the switching frequency (fSW) from 100kHz to 1MHz to
optimize efficiency/performance or external component
size. Higher frequency operation yields smaller component size but increases switching losses and gate driving
current, and may not allow sufficiently high or low duty
cycle operation. Lower frequency operation gives higher
efficiency, achieves higher maximum duty cycle or lower
minimum duty cycle at the cost of larger external component size. An external resistor from the RT pin to GND is
required—do not leave this pin open. For an appropriate
RT resistor value see Table 1.
Switching duty cycle is a key variable defining converter
operation, therefore, its limits must be considered when
programming the switching frequency for a particular application. The minimum duty cycle of the switch is limited
by the fixed minimum on-time (200ns maximum) and the
switching frequency (fSW). The maximum duty cycle of the
switch is limited by the fixed minimum off-time (200ns
maximum) and fSW. The following equations express the
minimum/maximum duty cycle:
Table 1. Switching Frequency (fSW) vs RT Value
fSW (kHz)
RT (kΩ)
fSW (kHz)
RT (kΩ)
100
154
600
22.6
150
102
650
20.5
200
75.0
700
17.4
250
59.0
750
19.1
300
48.7
800
16.2
350
41.2
850
15.0
400
35.7
900
14.0
450
31.6
950
13.3
500
28.0
1000
12.4
550
24.9
The operating frequency of the LT3797 can be synchronized
to an external clock source. By providing a digital clock
signal into the SYNC pin, the LT3797 will operate at the
SYNC clock frequency. If this feature is used, an RT resistor
should be chosen to program a switching frequency 20%
slower than SYNC pulse frequency. Tie the SYNC pin to
GND if this feature is not used.
Minimum Duty Cycle = 200ns • fSW
Maximum Duty Cycle = 1 – 200ns • fSW
Besides the limitation by the minimum off-time, it is also
recommended to choose the maximum duty cycle below
95%.
PWM Dimming Control
The LED of each channel can be dimmed with pulse width
modulation using the PWM pin. Figure 1 shows the channel 1 driver. If the PWM1 pin is pulled high, M2 is turned
on by G3 and G4. Converter 1 operates normally. G4 limits
ISP1-TG1 to 6.5V to protect the gate of M2. If the PWM1
pin is pulled low, the external NMOS M1 is turned off
through G1 etc, and converter 1 stops operating. M2 is
turned off through the TG1 pin, disconnecting LED1 and
stopping current drawing from output capacitor, COUT. The
VC1 pin is also disconnected from the internal circuitry
through S1. The capacitors CC and COUT store the state
of the LED string current until PWM1 is pulled up again.
This leads to a highly linear relationship between PWM
duty cycle and output light (brightness), and allows for
a large and accurate dimming range. The PWM dimming
3797f
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LT3797
APPLICATIONS INFORMATION
range can be maximized by using the PWM pin for dimming and the CTRL pin for linearly adjusting the current
sense threshold.
In the applications where the operation frequency of the
LT3797 is synchronized to an external clock source applied
to the SYNC pin, it is recommended to synchronize the
rising edge of the external clock and the rising edge of
the PWM signal of each of the three channels, as shown
in Figure 2.
Besides analog dimming, the CTRL pin can also be used
for PWM dimming control. Refer to Figure 1 for channel 1
operation. If CTRL1 falls below 150mV, the CTRL_ON signal is pulled low by comparator A11. Since CTRL_ON is
connected to one of G3’s inputs, channel 1 has the same
operation as PWM1 being pulled low (such as disconnecting LED1 from COUT and disconnecting CC from VC1, etc).
Therefore, the CTRL pin can be used for a combination
of linear and PWM dimming control if it is connected to
a PWM signal whose low level is below 150mV and high
level is between 0.2V and 1.3V. Connect the PWM pins to
the VREF pin if the CTRL pins are used for PWM dimming
or no PWM dimming is used.
Do not use a low VTH PMOS for LED disconnection. The
PMOS with a minimum VTH of –1V to –2V is recommended.
In the applications where accurate PWM dimming is not
required, the P-channel MOSFETs can be omitted to reduce
cost. In these conditions, the TG pins should be left open.
SYNC PIN
INPUT SIGNAL
PWM PIN
INPUT SIGNAL
3797 F02
Figure 2. Synchronize the SYNC Pin Input Signal
and the PWM Pin Input Signal
Programming the LED Current
The LED current of each channel is programmed by connecting an external sense resistor, RLED_SEN, in series with
the LED load, and setting the voltage regulation threshold
across RLED_SEN using CTRL input. The ISP and ISN sense
node traces should run parallel to each other to a Kelvin
connection on the positive and negative terminals of
RLED_SEN. Typically, sensing of the current should be done
at the top of the LED string. If this option is not available,
then the current may be sensed at the bottom of the LED
string. The CTRL pin should be tied to a voltage higher
than 1.3V to get the full-scale 250mV (typical) threshold
across the sense resistor. The CTRL pin can also be used
to dim the LED current from full scale to zero, although
relative accuracy decreases with the decreasing voltage
sense threshold. When the CTRL pin voltage is less than
1.1V and higher than 0.2V, the LED current is:
ILED =
VCTRL – 200mV
RLED _ SEN • 4
When the CTRL pin voltage is between 1.1V and 1.3V the
LED current varies with CTRL, but departs from the equation
above by an increasing amount as CTRL voltage increases.
Ultimately, above CTRL = 1.3V the LED current no longer
varies with CTRL. The typical (ISP-ISN) threshold vs CTRL
voltage when CTRL is close to 1.2V is listed in Table 2.
Table 2. (ISP-ISN) Threshold vs CTRL
When CTRL Is Close to 1.2V
VCRTL (V)
(ISP-ISN) THRESHOLD (mV)
1.1
225
1.15
236
1.2
244.5
1.25
248.5
1.3
250
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LT3797
APPLICATIONS INFORMATION
When CTRL is higher than 1.3V, the LED current is regulated to:
ILED =
250mV
RLED _ SEN
1.25V, therefore the output regulation voltage can be set
according to the following equation:
VOUT = 1.25V •
The LED current is regulated to 0A when CTRL is lower
than 200mV (typical).
The CTRL pin should not be left open (tie to VREF if not
used). The CTRL pin can also be used in conjunction with
a thermistor to provide overtemperature protection for the
LED load, or with a resistor divider to VIN to reduce output
power and limit peak switching current when VIN is low.
The presence of a time varying differential voltage signal
(ripple) across ISP and ISN at the switching frequency
is expected. The amplitude of this signal is increased
by high LED load current, low switching frequency and/
or a smaller value output filter capacitor. Some level of
ripple signal is acceptable: the compensation capacitor
on the VC pin filters the signal so the average difference
between ISP and ISN is regulated to the user-programmed
value. Ripple voltage amplitude (peak-to-peak) in excess
of 50mV should not cause misoperation, but may lead
to noticeable offset between the average value and the
user-programmed value.
Programming Output Regulation Voltage
for the Open-LED Event
The output voltage of each channel in the open-LED event
can be programmed by selecting two external sense
resistors. Figure 3 shows the sense resistor connection
of channel 1. In the open-LED event, VFB1 is regulated to
R5+R6
R5
Since the output voltage is directly measured between
ISP1 and LED1–, the Figure 3 approach works well for the
converter topologies where LED1– is connected to GND
(such as boost, SEPIC, flyback), as well as the topologies
where LED1– is connected to an inductor (such as buck
mode, buck-boost mode LED drivers).
Typically, the current sense resistor RLED_SEN1 and disconnect PMOS M2 are connected to the top of the LED
string (LED1+), as shown in Figure 3. If this option is not
available (for example some multi-string LED modules
are built with a common anode configuration), then the
current may be sensed at the bottom of the LED string
as shown in Figure 4. In this configuration, the FBH pin
draws 2µA (typical) current. Therefore, the output regulation voltage in the open-LED event can be set according
to the following equation:
VOUT = 1.25V •
R5+R6
+ 2µA •R6
R5
Under normal operating conditions, the LED current
regulation loop is dominant. Therefore, the output regulation voltage (VOUT) in the open-LED event should be
programmed so that VFB1 (VFB1 = |ISP1-FBH1|) should
never exceed 1.1V when LED1 is connected. The only
way for VFB1 to be within 50mV of the regulation voltage
(1.25V) is for an open-LED event to occur.
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LT3797
APPLICATIONS INFORMATION
ISP1
LT3797
FBH1
+
–
VFB1
R5
RLED_SEN1
ISN1
ISN1
ISN1
+
M2
R6
LED1+
VIN(FALLING) = 1.22V •
LED1
TG1
TG1
COUT1
VOUT
TG1
LED1–
–
3797 F03
Figure 3. Output Voltage Sense Resistor Connection
FBH1
LT3797
ISP1
+
–
2µA
R6
VFB1
R5
+
LED1+
LED1
VOUT
LED1–
–
COUT1
RLED_SEN1
ISN1
TG1
active. The purpose of this current is to allow the user
to program the rising hysteresis. The falling threshold
voltage and rising threshold voltage can be calculated by
the following equations:
M2
3797 F04
Figure 4. Output Voltage Sense Resistor Connection When
RLED_SEN1 and M2 Are Connected to the Bottom of the LED String
Programming Enable and Undervoltage Lockout with
the EN/UVLO Pin
EN/UVLO pin controls whether the LT3797 is enabled or is
in shutdown state. As shown in Figure 1, a 1.22V reference,
a comparator, A1, and a controllable current source, IS1,
allow the user to accurately program the supply voltage
at which the IC turns on and off. The falling value can be
accurately set by the resistor divider R1 and R2. When
EN/UVLO is above 0.4V and below the 1.22V threshold,
the small pull-down current source, IS1 (typical 2µA), is
R1+R2
R2
VIN(RISING) = VIN(FALLING) + 2µA •R1
For applications where the EN/UVLO pin is to be used
only as a logic input, the EN/UVLO pin can be connected
directly to the input voltage, VIN, for “always on” operation.
Programming Overvoltage Lockout Threshold
with the OVLO Pin
The LT3797 provides an OVLO pin that allows user-programmable overvoltage lockout. A 1.25V (typical) rising
threshold with 125mV hysteresis detects the overvoltage
condition. The OVLO pin can be used to monitor VIN or
other voltages against overvoltage conditions.
Figure 1 shows OVLO connecting to VIN through a voltage divider to protect against VIN overvoltage. The rising
threshold voltage and falling threshold voltage can be
calculated by the following equations:
VOV(RISING) = 1.25V •
R3+R4
R4
VOV(FALLING) = 1.125V •
R3+R4
R4
An overvoltage condition turns off all three channels
(including pulling the GATE pins to GND and TG pins to
ISP) and resets the soft-starts.
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LT3797
APPLICATIONS INFORMATION
Loop Compensation
Loop compensation determines the stability and transient
performance. The LT3797 uses current mode control to
regulate the output which simplifies loop compensation.
The optimum values depend on the converter topology, the
component values and the operating conditions (including the input voltage, LED current switching frequency).
To compensate the feedback loop of the LT3797, a series
resistor-capacitor network is usually connected from the VC
pin to GND. Figure 1 shows the typical VC compensation
network. For most applications, the capacitor should be
in the range of 2.2nF to 22nF, and the resistor should be
in the range of 2k to 25k. A practical approach to designing the compensation network is to start with one of the
circuits in this data sheet that is similar to your application, and tune the compensation network to optimize the
performance. Stability should then be checked across all
operating conditions, including LED current, input voltage
and temperature. Application Note 76 is a good reference
for loop compensation.
Soft-Start and Fault Protection
The LT3797 has identical soft-start and fault protection
functions for each channel. The soft-start feature is designed to limit peak switch currents and output voltage
(VOUT) overshoot during start-up or recovery from a fault
condition. Figure 4 shows the state diagram of the soft-start
and fault protection of channel 1. Also refer to Figure 1
for channel 1 operation. In soft-start mode, the soft-start
capacitor is charged up by the 25µA current source. The
SS1 pin gradually increases the peak switch current al-
lowed in M1 by clamping the VC1 voltage through Q4. In
this way the SS1 pin allows the output capacitor, COUT,
voltage to be charged gradually toward its final value while
limiting M1 current overshoot. The soft-start interval is
set by the soft-start capacitor selection according to the
following equation:
tSS =
1.2V
•C
25µA SS
The discharge time of the soft-start capacitor is controlled
by a 2.5µA current source. Therefore, the SS1 pin is also
used as an adjustable timer in the FAULT2 protection modes
(see Figure 5) to prevent thermal runaway problems on the
external components and/or the LEDs. In some fault conditions, the soft-start capacitor is charged and discharged
repetitively, referred to as the hiccup mode operation. A
typical hiccup mode operation occurs when an LT3797 LED
driver has an output short-circuit fault. Figure 5 shows that
if an output short-circuit fault causes LT3797 overcurrent
(sensed by ISP1-ISN1) in the normal operation mode, the
LT3797 moves to FAULT2 protection mode, where TG1 is
pulled high, turning off the external PMOS and isolating
the output. As a result, the overcurrent condition is cleared.
When SS1 is discharged below 200mV, the LT3797 moves
to soft-start mode, where TG1 is pulled low to turn on
the external PMOS. If the short-circuit fault still exists,
the LT3797 senses an overcurrent fault again and moves
to FAULT2 protection mode: SS1 charged up and a new
cycle starts. In this manner, the soft-start capacitor is kept
charging and discharging between 200mV and 1.7V until
the short-circuit fault is cleared.
3797f
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17
LT3797
APPLICATIONS INFORMATION
EN/UVLO < 1.22V (TYPICAL) OR
VIN < 2.2V (TYPICAL)
SHUTDOWN MODE
•
•
•
•
INTEGRATED INTVCC POWER SUPPLY OFF
GATE1 LOW
TG1 HIGH
IQ < 1µA
EN/UVLO > 1.22V (TYPICAL) AND
VIN > 2.2V (TYPICAL)
FAULT1
INTEGRATED INTVCC POWER SUPPLY START-UP MODE
•
•
•
•
•
SS1 PULLED LOW BY A 1mA CURRENT SOURCE
FLT1 LOW
TG1 HIGH
GATE1 LOW
INTEGRATED INTVCC POWER SUPPLY ENABLED
AND INTVCC CHARGED UP
FAULT1
CLEARED
FAULT1 PROTECTION MODE
•
•
•
•
INTEGRATED INTVCC POWER SUPPLY OFF
FLT1 LOW
TG1 HIGH
GATE1 LOW
INTVCC > 5.7V (TYPICAL) AND
CTRL1 > 0.2V (TYPICAL) AND
PWM1 = HIGH AND
SS1 < 0.2V AND
OVLO < 1.25V
FAULT2 PROTECTION MODE: SS1 CHARGED UP
SOFT-START MODE
• SS1 CHARGED UP BY THE 25µA
CURRENT SOURCE
• FLT1 HIGH
• TG1 LOW
• GATE1 SWITCHING TO RAMP UP OUTPUT
LED CURRENT
FAULT2
• SS1 CHARGED UP BY THE 25µA
CURRENT SOURCE
• FLT1 LOW
• TG1 HIGH
• GATE1 LOW
SS1 > 1.7V (TYPICAL)
FAULT2 PROTECTION MODE: SS1 DISCHARGED
CONDITION1
•
•
•
•
SS1 DISCHARGED BY A 2.5µA CURRENT SOURCE
FLT1 LOW
TG1 HIGH
GATE1 LOW
3797 F05
SS1 > 1.7V (TYPICAL)
NORMAL OPERATION MODE
FAULT2
• NORMAL OPERATION
NOTES:
FAULT1 = VIN > 41V (TYPICAL) OR
OVER TEMPERATURE (TJ > 165°C)
FAULT2 = VIN > 41V (TYPICAL) OR
OVER TEMPERATURE (TJ > 165°C) OR
INTVCC < 5.2V (TYPICAL) OR
OVLO > 1.25V OR
OUTPUT OVER CURRENT
CONDITION1 = FAULT2 CLEARED AND
CTRL1 > 0.2V (TYPICAL) AND
PWM1 = HIGH AND
SS1 < 0.2V
Figure 5. State Diagram of the Soft-Start and Fault Protection of Channel 1
3797f
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LT3797
APPLICATIONS INFORMATION
The LT3797 fault protection can be configured as the latchoff mode by connecting a 470k resistor between the SS
pin and VREF pin. The FAULT2 conditions (see Figure 4)
cause the LT3797 latch off. The LT3797 does not retry a
soft-start even if the fault condition is cleared, since the SS
pin is not able to fall below 0.2V by the 2.5µA pull-down
current to reset the latch, due to the pulling up of the 470k
resistor. The latch-off can only be cleared by toggling the
EN/UVLO pin low to high.
The open-LED fault and the output overvoltage fault are
not included in FAULT2 in Figure 5. These two faults do
not affect the soft-start status. The open-LED fault in
channel 1 causes FLT1 low. The output overvoltage fault
in channel 1 causes FLT1 low and TG1 high to disconnect
the LED load from power path.
than the LED voltage. The switch duty cycles of different
topologies in continuous conduction mode (CCM) are:
DBOOST =
VLED – VIN
VLED
VLED
VIN
DBUCK =
DBUCK-BOOST =
DSEPIC =
VLED
VLED + VIN
VLED
VLED + VIN
The maximum duty cycle (DMAX) occurs when the converter
has the minimum input voltage (VIN(MIN)).
APPLICATION CIRCUIT DESIGN GUIDELINE
Inductor Selection
The LT3797 contains three independent switching regulators. The following sections describe the LT3797 LED
driver design guideline for the key parameters calculation
and external components selection. The design guideline
applies to each of the switching regulators.
Figure 6 shows a typical inductor current waveform
when the LED driver has maximum output current at the
minimum input voltage. ∆IL and IL_AVG(MAX) denote the
inductor ripple current and the maximum average inductor
current respectively.
Switch Duty Cycle
The LT3797 can be configured with different topologies.
The boost LED driver is used for the applications where the
LED voltage is higher than the input voltage. The LT3797
can be configured as a buck mode LED driver for the applications where the LED voltage is lower than the input
voltage. The buck-boost mode and the SEPIC LED driver
allow for the input voltage to be higher, equal to or lower
∆IL
IL
IL_AVG(MAX)
IL(PEAK)
t
3739 F06
Figure 6. A Typical Inductor Waveform
3797f
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LT3797
APPLICATIONS INFORMATION
The IL_AVG(MAX) of boost, buck mode, and buck-boost
mode LED drivers in CCM are:
IL _ AVG(MAX)_ BUCK =ILED(MAX)
IL _ AVG(MAX)_ BOOST =ILED(MAX) •
1
1–DMAX
IL _ AVG(MAX)_ BUCK-BOOST =ILED(MAX) •
1
1–DMAX
The primary and secondary maximum average inductor
current of the SEPIC LED driver are:
IL1_ AVG(MAX)_ SEPIC = ILED(MAX) •
DMAX
1–DMAX
IL2 _ AVG(MAX)_ SEPIC = ILED(MAX)
where ILED(MAX) is the maximum LED current.
The inductor ripple current ∆IL has a direct effect on the
choice of the inductor value. Choosing smaller values of
∆IL requires large inductances and reduces the current loop
gain (the converter will approach voltage mode). Accepting
larger values of ∆IL provides fast transient response and
allows the use of low inductances, but results in higher
input current ripple and greater core losses.
The inductor ripple percentage of the boost, buck mode,
and buck-boost mode LED drivers is:
Given an operating input voltage range, and having chosen
the operating frequency, f, and ripple current ∆IL in the
inductor, the inductor values of the boost, buck mode, and
buck-boost mode LED drivers can be determined using
the following equations:
LBUCK =
IL(MAX)
For the SEPIC converter, ∆IL of the primary inductor is
equal to ∆IL of the secondary inductor. The inductor ripple
percentage can be calculated as:
(
VLED
• 1–DMAX)
∆IL • f
LBOOST =
VIN(MIN)
∆IL • f
LBUCK-BOOST =
)
•DMAX
VIN(MIN)
∆IL • f
•DMAX
The primary and secondary inductor values of the SEPIC
LED driver are:
L1= L2 =
VIN(MIN)
∆IL • f
•DMAX
By making L1 = L2, and winding them on the same core, the
value of inductance in the preceding equation is replaced
by 2L, due to mutual inductance:
L=
∆IL
2 • ∆IL
IL1(MAX) +IL2(MAX)
The user should choose an appropriate ∆IL based on the
trade-offs to optimize the LED driver performance. It is
recommended that the ripple current percentage fall within
the range of 20% to 60% at DMAX.
VIN(MIN)
2 • ∆IL • f
•DMAX
The inductor peak current and RMS current in continuous
mode operation can be calculated based on IL(MAX) and ∆IL.
IL(PEAK) = IL(MAX) + 0.5 • ∆IL
IL(RMS) ≈ IL(MAX)
Based on the preceding equations, the user should choose
the inductors having sufficient saturation and RMS current ratings.
3797f
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LT3797
APPLICATIONS INFORMATION
Switch Current Sense Resistors Selection
Sense Voltage Ripple Verification
The LT3797 measures each channel’s power N-channel
MOSFET current by using a sense resistor (see RSW_SEN in
Figure 1) between GND and the MOSFET source. Figure 7
shows a typical waveform of the sense voltage (VSW_SENSE)
across the sense resistor in CCM. The placement of the
sense resistor RSW_SEN should be close to the source of the
MOSFET and GND. The SENSEP and SENSEN sense node
traces should run parallel to each other to a Kelvin connection on the positive and negative terminals of RSW_SEN.
After the inductor ripple current and the switch current
sense resistor value have been selected according to the
previous sections, the sense voltage ripple ∆VSW_SENSE
(refer to Figure 7) of the boost, buck, and buck-boost LED
drivers can be determined using the following equation:
Due to the current limit function of the power switch current control, RSW_SEN should be selected to guarantee that
the peak current sense voltage VSW_SENSE(PEAK) during
steady-state normal operation is lower than the SENSE
current limit threshold (100mV minimum). It is recommended to give a 20% margin and set VSW_SENSE(PEAK)
to be 80mV. Then, the switch current sense resistor value
can be calculated as:
RSW _ SEN =
80mV
ISW(PEAK)
where ISW(PEAK) is the peak switch current. ISW(PEAK) of
the boost, buck mode and buck-boost mode LED driver is:
ISW(PEAK) = IL(PEAK)
∆VSW_SENSE = ∆IL • RSW_SEN
∆VSW_SENSE of the SEPIC LED driver can be determined
using the following equation:
∆VSW_SENSE = 2 • ∆IL • RSW_SEN
The LT3797 has internal slope compensation to stabilize
the control loop against sub-harmonic oscillation. When
the LT3797 operates at a duty cycle greater than 0.66
in CCM, the sense voltage ripple, ∆VSW_SENSE (refer to
Figure 7), needs to be limited to ensure the internal slope
compensation is sufficient to stabilize the control loop.
Figure 8 shows the maximum ∆VSW_SENSE over the duty
cycle. It is recommended to check and ensure ∆VSW_SENSE
is below this curve at the highest duty cycle. If ∆VSW_SENSE
is above the maximum ∆VSW_SENSE curve at the highest
duty cycle, the ∆IL needs to be reduced and the parameters
in the previous two sections need to be recalculated until
the optimized values are obtained.
ISW(PEAK) of the SEPIC LED driver is:
110
ISW(PEAK) = IL1(PEAK) + IL2(PEAK)
90
MAX ∆VSW_SENSE (mV)
100
∆VSW_SENSE
VSW_SENSE
70
60
50
40
30
20
VSW_SENSE(PEAK)
10
t
D/f
80
3797 F07
0
0.5 0.55 0.6 0.65 0.7 0.75 0.8 0.85 0.9 0.95
DUTY CYCLE
1/f
3797 F08
Figure 7. The Sense Voltage Across the Sense Resistor in CCM
Figure 8. The Maximum Sense Voltage Ripple
vs Duty Cycle for CCM
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LT3797
APPLICATIONS INFORMATION
Power MOSFET Selection
The selection criteria for the power MOSFET includes the
drain-source breakdown voltage (BVDSS), the threshold
voltage (VGS(TH)), the on-resistance (RDS(ON)), the total
gate charge (QG), the maximum drain current (ID(MAX)) and
the MOSFET ’s thermal resistances (RθJC and RθJA), etc.
The required power MOSFET BVDSS rating of different
topologies can be estimated using the following equations.
Add a diode forward voltage, and any additional ringing
across its drain-to-source during its off-time.
BVDSS_BOOST > VLED
BVDSS_BUCK > VIN(MAX)
BVDSS_BUCK-BOOST > VIN(MAX) + VLED
BVDSS_SEPIC > VIN(MAX) + VLED
The power dissipated by the MOSFET in a boost, buck
mode, or buck-boost mode LED driver is:
PFET = IL(MAX)2 • RDS(ON) • DMAX + 2 • VSW(PEAK) •
IL(MAX) • CRSS • f/1.5A
The power dissipated by the MOSFET in a SEPIC LED
driver is:
PFET = (IL1(MAX) + IL2(MAX))2 • RDS(ON) • DMAX + 2 •
VSW(PEAK) • (IL1(MAX) + IL2(MAX)) • CRSS • f/1.5A
The first terms in the preceding equations represent the
conduction losses in the devices, and the second terms, the
switching losses. CRSS is the reverse transfer capacitance,
which is usually specified in the MOSFET characteristics.
For maximum efficiency, RDS(ON) and QG should be
minimized. From a known power dissipated in the power
MOSFET, its junction temperature can be obtained using
the following equation:
TJ = TA + PFET • θJA = TA + PFET • (θJC + θCA)
TJ must not exceed the MOSFET maximum junction
temperature rating. It is recommended to measure the
MOSFET temperature in steady state to ensure that absolute
maximum ratings are not exceeded.
Schottky Rectifier Selection
The power Schottky diode conducts current during the
interval when the switch is turned off. In an LT3797 LED
driver, the Schottky diode should have the same voltage
rating as the power N-channel MOSFET in the same channel.
Refer to the power MOSFET BVDSS rating in the previous
section for the peak reverse voltage rating selection. If using
the PWM feature for dimming, it is important to consider
diode leakage, which increases with the temperature, from
the output during the PWM low interval. Choose a Schottky
diode with sufficiently low leakage current.
The power dissipated by the diode in a boost, buck, or
buck-boost converter in CCM is:
PD = IL_AVG(MAX) • VD • (1 – DMAX)
where VD is the diode forward voltage drop.
The power dissipated by the diode in a SEPIC converter is:
PD = (IL1_AVG(MAX) + IL2_AVG(MAX)) • VD • (1 – DMAX)
and the diode junction temperature is:
TJ = TA + PD • (θJC + θCA)
TJ must not exceed the diode maximum junction temperature rating.
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LT3797
APPLICATIONS INFORMATION
High Side PMOS Disconnect Switch Selection
Output Capacitor Selection
A PMOS with a minimum VGS(TH) of –1V to –2V is recommended for the high side disconnect switch in most
LT3797 applications to improve the PWM dimming ratio
and protect the LED array from excessive heating during
fault conditions. The PMOS BVDSS rating must be higher
than the open-LED regulation voltage set by the FBH pin.
The maximum continuous drain current ID(MAX) rating
should be higher than the maximum LED current.
The output filter capacitors should be sized to attenuate
the LED current ripple. Use of X5R or X7R type ceramic
capacitors is recommended. To achieve the same LED
ripple current, the required filter capacitor is smaller in the
buck mode applications than that in the boost, buck-boost
mode and SEPIC applications. This is due to the fact that,
in the buck converter, the inductor is in series with the
output and the ripple current flowing through the output
capacitor is continuous. Lower operating frequencies will
require proportionately higher capacitor values.
Input Capacitor Selection
The input capacitor CIN supplies the AC ripple current to
the power inductor of the converter and must be placed
and sized according to the transient current requirements.
The switching frequency, output current and tolerable input
voltage ripple are key inputs to estimating the required
capacitor value. The X5R or X7R type ceramic capacitors
are usually good choices since they have small variation
with temperature and DC bias. Typically, the boost or SEPIC
converter requires a lower value input capacitor than the
buck mode or buck-boost mode converter, due to the fact
that its inductor is in series with the input, and the input
current waveform is continuous. The input capacitor value
can be estimated based on the inductor ripple ∆IL (refer to
Inductor Selection section), the switching frequency, and
the acceptable input voltage ripple ∆VIN on CIN. CIN value
of the boost and SEPIC converter can be calculated by:
CIN = 0.125 •
∆IL
∆VIN • f
(
VLED • VIN(MIN) – VLED
VIN(MIN)2 • ∆VIN • f
The DC voltage rating of the DC coupling capacitor, CDC,
connected between the primary and secondary inductors
should be larger than the maximum input voltage:
VCDC > VIN(MAX)
CDC has nearly a rectangular current waveform in CCM.
During the switch off-time, the current through CDC is IVIN,
while approximately –ILED flows during the on-time. The
CDC voltage ripple causes distortions on the primary and
secondary inductor current waveforms. The CDC should
be sized to limit its voltage ripple. The power loss on the
CDC ESR reduces the LED driver efficiency. Therefore, the
sufficient low ESR ceramic capacitors should be selected.
The X5R or X7R ceramic capacitor is recommended for CDC.
Integrated INTVCC Power Supply
CIN value of the buck mode and buck-boost mode LED
driver can be calculated by:
CIN =ILED •
The DC Coupling Capacitor Selection for
SEPIC LED Driver
)
The LT3797 includes an internal switch mode DC/DC converter to generate a regulated 7.5V INTVCC power supply
to power the NMOS gate drivers of the three channels
(IDRIVE). This INTVCC power supply can also be used to
drive external circuits (IEXT). This INTVCC power supply
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23
LT3797
APPLICATIONS INFORMATION
• CVCC is a 10μF/10V ceramic capacitor used to bypass
INTVCC to GND immediately adjacent to the pins.
• CBOOST is a 0.1μF/10V ceramic capacitor connected
between the BOOST pin and the SW1 pin.
• Select a 47µH inductor with the saturation current rating of 0.6A or greater and RMS current rating of 0.4A
or greater for LPWR.
The INTVCC power supply has an output current limit function to protect itself from excessive electrical and thermal
stress. Figure 9 shows the INTVCC output limit (IINTVCC_LMT)
vs VIN and switching frequency. Make sure the sum of the
IDRIVE and IEXT is always lower than the IINTVCC_LMT across
the whole VIN range of the application circuit:
IDRIVE + IEXT < IINTVCC_LMT
where:
250
INTVCC CURRENT LIMIT IINTVCC_LMT (mA)
has two major advantages over the traditional internal
LDO regulators. It is able to generate 7.5V INTVCC voltage
from a VIN voltage as low as 2.5V, allowing the LT3797 to
drive high threshold MOSFETs in the low VIN applications.
It is also able to deliver large current from a VIN voltage
as high as 40V without overheating the package, due to
its high efficiency (over 70% at full load). This integrated
DC/DC converter requires three external components (CVCC,
CBOOST and LPWR) for operation, as shown in Figure 1.
Select these three components based on the following
guidelines:
800kHz 700kHz >900kHz
225
200
175
600kHz
150
500kHz
400kHz
125
300kHz
100
75
200kHz
50
100kHz
25
0
0 3 6 9 12 15 18 21 24 27 30 33 36 39
VIN (V)
3739 F09
Figure 9. INTVCC Current Limit vs VIN, fSW
Board Layout
The high speed operation of the LT3797 demands careful
attention to board layout and component placement. The
exposed pad of the package is the only GND terminal of
the IC, and is important for thermal management of the
IC. Therefore, it is crucial to achieve a good electrical and
thermal contact between the exposed pad and the ground
planes of the board. For the LT3797 to deliver its full output
power, it is imperative that a good thermal path be provided to dissipate the heat generated within the package.
It is recommended that multiple vias in the printed circuit
board be used to conduct heat away from the IC and into
copper planes with as much area as possible.
IDRIVE = (QG_CH1 + QG_CH2 + QG_CH3) • fSW
QG_CH1-3 is the total gate charge of the NMOS of the three
channels at VGS = 0V to 7.5V.
3797f
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LT3797
APPLICATIONS INFORMATION
The compensation networks (VC1-3) and other DC control
signals (such as SS1-3, RT, EN/UVLO, OVLO and CTRL1‑3)
should have separate signal ground (SGND) from the power
stage ground (PGND). Connect SGND and PGND only at
the LT3797 exposed GND pad (Pin 53). Do not extensively
route high impedance signals such as FBH and VC, as they
may pick up switching noise.
and SENSEN1-3 pins. Figure 10 shows a example of a PCB
layout of the decoupling capacitors and ground separation.
To reduce electromagnetic interference (EMI) and high
frequency resonance problems, proper layout of the LT3797
LED driver power stage is essential, especially the power
paths with high di/dt. Figures 11-14 show the simplified
power stage circuits of boost, buck mode, buck-boost
mode and SEPIC topologies with the high di/dt loops highlighted. The high di/dt loops of different topologies should
be kept as tight as possible to reduce inductive ringing.
Figures 15-16 shows the examples of the high di/dt loop
layout of the different topologies shown in Figures 11-14.
The decoupling capacitors that connect VIN, INTVCC,
SENSEP1-3 and SENSEN1-3 for the LT3797 should be
physically close to their pins. The small footprint size (0201
or 0402) ceramic capacitors are recommended for the
decoupling capacitor connecting between the SENSEP1-3
VIN
PGND
C
C
RSW_SEN3
SS3
PGND
C
SGND
C
3
SENSEP3
2
SENSEN3
1
INTVCC
VIN
INTVCC
52 51 50 49 48 47 46 45 44 43 42 41
40 VC3
R
C
R
C
38
37
4
5
36
53
EXPOSED PAD GND
6
35
7
15 16 17
SENSEN2
14
SENSEP2
30
12
SENSEP1
31
11 RT
SENSEN1
32
10
SS1
SGND
33
VC1
R
8
9
28 VC2
27 SS2
19 20 21 22 23 24 25 26
C
3797 F10
C
SGND
C
R
RSW_SEN1
C
RSW_SEN2
C
SGND
Figure 10. Decoupling Capacitors and Ground Separation
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25
LT3797
APPLICATIONS INFORMATION
L
D
SW
M
+
–
C
VIN
R
LED
STRING
3797 F11
PGND
Figure 11. The Simplified Boost LED Driver Power Stage
with the High di/dt Loop Highlighted
LED STRING
D
L
SW
M
+
–
VIN C
R
PGND
3797 F12
Figure 12. The Simplified Buck Mode LED Driver Power
Stage with the High di/dt Loop Highlighted
LED STRING
L
D
SW
M
+
–
C
VIN
R
3797 F13
PGND
Figure 13. The Simplified Buck-Boost Mode LED Driver
Power Stage with the High di/dt Loop Highlighted
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LT3797
APPLICATIONS INFORMATION
L1
C2
SW1
SW2 D
M
+
–
L2
VIN
C1
R
LED
STRING
3797 F14
PGND
Figure 14. The Simplified SEPIC LED Driver Power Stage
with the High di/dt Loop Highlighted
SW
D
M
R
C
3797 F15
PGND
Figure 15. A Layout Example of the High di/dt Loop of
the Boost, Buck Mode, Buck-Boost Mode LED Drivers
SW2
SW1
C2
D
M
R
C1
3797 F16FF
PGND
Figure 16. A Layout Example of the High di/dt Loop of
the SEPIC Drivers
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LT3797
APPLICATIONS INFORMATION
Check the stress on the power MOSFETs by measuring
the drain-to-source voltage directly across the terminals
of each device (reference the ground of a single scope
probe directly to the source pad on the PC board). Beware
of inductive ringing, which can exceed the maximum
specified voltage rating of the MOSFET. If this ringing cannot be avoided, and exceeds the maximum rating of the
device, either choose a higher voltage device or specify
an avalanche rated power MOSFET.
The LT3797 LED driver circuit can be implemented in a
2-layer PCB board. However, a well designed 4-layer or
6-layer PCB board provides extra ground plane shielding and larger area of electrical and thermal conduction
path, therefore provides better electrical and thermal
performance.
Recommended Component Manufacturers
Some of the recommended component manufacturers
are listed in Table 3.
Table 3. Recommended Component Manufacturers
VENDOR
COMPONENTS
WEB ADDRESS
AVX
Capacitors
avx.com
BH Electronics
Inductors, Transformers
bhelectronics.com
Central
Semiconductor
Diodes
centralsemi.com
Coilcraft
Inductors
coilcraft.com
Coiltronics
Inductors
cooperindustries.com
Diodes, Inc
MOSFETs, Diodes
diodes.com
Fairchild
MOSFETs, Diodes
fairchildsemi.com
International Rectifier
MOSFETs, Diodes
irf.com
IRC
Sense Resistors
irctt.com
Kemet
Capacitors
kemet.com
Murata
Inductors, Capacitors
murata.com
Nichicon
Capacitors
nichicon.com
On Semiconductor
MOSFETs, Diodes
onsemi.com
Panasonic, Industrial
Capacitors, Resistors
panasonic.com
Sumida
Inductors
sumida.com
Taiyo Yuden
Inductors, Capacitors
t-yuden.com
TDK
Inductors, Capacitors
component.tdk.com
United Chemicon
Electrolytic Capacitors
chemi-con.com
Vishay
MOSFETs, Diodes,
Inductors, Capacitors,
Sense Resistors
vishay.com
Würth-Midcom
Inductors
katalog.we-online.de
3797f
28
For more information www.linear.com/LT3797
LT3797
TYPICAL APPLICATIONS
Triple Boost LED Driver
VIN
2.5V TO 40V
(60V TRANSIENT,
41V INTERNAL
OVLO PROTECTION)
CIN1-3
4.7µF
×3
L1
10µH
L2
10µH
D1
COUT1
4.7µF
100V
×2
ISP1
R2
250mΩ
L3
10µH
D2
ISP2
R4
250mΩ
ISN1
M2
D4*
VLED1+
CIN3
1µF
R7
47.5k
1A
50V
GATE1 SENSEP1 SENSEN1 TG1
VIN
0.1µF
M5
1A
50V
R3
8mΩ
GATE2 SENSEP2 SENSEN2 TG2
EN/UVLO
R8
OVLO VREF
49.9k
SYNC
RT
48.7k CSS1-3
300kHz 0.1µF
487k
SS1-3
FLT1-3
SS1-3
SW1 SW2
FLT1-3
LPWR
47µH
100k
INTVCC
RSS1-3**
470k
EFFICIENCY (%)
CBST
0.1µF
INTVCC
GND
INTVCC
CVCC
10µF
VC1-3
RC1-3
4.7k
CC1-3
10nF
R12-R14
1M
3797 TA02
*D4-6: OPTION FOR SHORT LED PROTECTION
**RSS1-3: OPTIONAL FOR FAULT LATCHOFF
1.4
1.2
OUTPUT CURRENT
1.0
70
0.8
65
0.6
60
0.4
55
0.2
15
20 25
VIN (V)
PWM
5V/DIV
30
35
OUTPUT CURRENT (A)
85
10
R9-R11
20.5k
500:1 PWM Dimming at 120Hz
1.6
80
5
BOOST
VREF
1.8
EFFICIENCY
0
ISN1-3
ISP1-3
2.0
PWM1-3 = 2V
90
50
1A
50V
R5
8mΩ
GATE3 SENSEP3 SENSEN3 TG3
Efficiency and Output Current vs VIN
75
0.1µF
FBH1-3
CTRL1-3 PWM1-3 RT
75k
D1-D3: DIODES INC. PDS5100
D4-D6: VISHAY SILICONIX ES1C
L1-L3: COILTRONICS HC9-100-R
LPWR: COILTRONICS SD25-470
M1, M3, M5: INFINEON BSC123N08NS3-G
M2, M4, M6: VISHAY SILICONIX Si7113DN
95
ISN3
M6
D6*
VLED3+
LT3797
VREF V
IN
100
R6
250mΩ
M3
R1
8mΩ
COUT3
4.7µF
100V
×2
ISP3
ISN2
M4
D5*
VLED2+
M1
0.1µF
D3
COUT2
4.7µF
100V
×2
IL
5A/DIV
ILED
1A/DIV
VIN = 12V
5µs/DIV
3797 TA02c
0
40
3797 TA02b
Fault (Short LED) Protection without RSS1-3: Hiccup Mode
Fault (Short LED) Protection without RSS1-3: Latchoff Mode
SS1-3
2V/DIV
SS1-3
2V/DIV
IM2,4,6
10A/DIV
IM2,4,6
10A/DIV
VLED1-3+
50V/DIV
VLED1-3+
100V/DIV
FLT1-3
10V/DIV
FLT1-3
10V/DIV
50ms/DIV
3797 TA02d
50ms/DIV
3797 TA02e
3797f
For more information www.linear.com/LT3797
29
LT3797
TYPICAL APPLICATIONS
3V to 5V Input, Triple Boost LED Driver
VIN
3V TO 5V
CIN1-3
47µF
×3
L1
1.0µH
L2
1.0µH
D1
L3
1.0µH
D2
COUT1
22µF
ISP1
ISP2
0.125Ω
ISN1
0.1µF
CIN4
10µF
ISN2
M3
2A
8V
R1
0.01Ω
0.1µF
ISN3
M6
M5
2A
8V
R3
0.01Ω
0.1µF
GATE2 SENSEP2 SENSEN2 TG2
59k
TG3 ISN1-3
ISP1-3
GATE3 SENSEP3 SENSEN3
80.6k
LT3797
FBH1-3
OVLO VREF CTRL1-3 PWM1-3
22.6k
D1-D3: VISAHY SILICONIX 30BQ015
L1-L3: VISAHY SILICONIX IHLP-2525CZ-01
LPWR: COILTRONICS SD25-470
M1, M3, M5: INFINEON BSC050N03LSG
M2, M4, M6: VISHAY SILICONIX Si7619DN
RT
SYNC SS1-3
RT
12.4k
1MHz
FLT1-3 SW1 SW2
BOOST
LPWR
47µH
0.1µF
0.1µF
INTVCC
GND
10µF
VC1-3
499k
4.7k
10nF
3797 TA03
Efficiency vs VIN
1000:1 PWM Dimming at 120Hz
CTRL1-3 = 2V
PWM1-3 = 2V
95
2A
8V
R5
0.01Ω
EN/UVLO
100
COUT3
22µF
0.125Ω
M4
GATE1 SENSEP1 SENSEN1 TG1
VIN
30.1k
EFFICIENCY (%)
ISP3
0.125Ω
M2
M1
D3
COUT2
22µF
PWM
5V/DIV
90
IL
5A/DIV
85
ILED
2A/DIV
80
75
70
2µs/DIV
3
3.5
4
VIN (V)
4.5
3797 TA03c
5
3797 TA03b
3797f
30
For more information www.linear.com/LT3797
LT3797
TYPICAL APPLICATIONS
Triple Buck Mode LED Driver
PVIN
24V TO 80V
CIN1-3
4.7µF
×3
ISP1
1M
0.25Ω
OVLO
14.3k
TG1
ISN1
TG3
COUT1
4.7µF
LED1+
1A
20V
1A
20V
COUT2
4.7µF
20k
348k
FBH2
D2
D1
0.068Ω
L3
22µH
L2
33µH
M3
GATE3 SENSEP3 SENSEN3
D3
TG1-3
ISN1-3
LT3797
ISP1-3
EN/UVLO
FBH1-3
OVLO VREF CTRL1-3 PWM1-3
OVLO
D1-D3: VISHAY SILICONIX VS-10BQ100
L1-L3: WÜRTH 74437349330
LPWR: COILTRONICS SD25-470
M1, M3, M5: VISHAY SILICONIX Si4100DY
M2, M4, M6: VISHAY SILICONIX Si7113DN
RT
SYNC SS1-3
35.7k
400kHz
FLT1-3 SW1 SW2
LPWR
47µH
0.1µF
BOOST
INTVCC
0.1µF
GND
VC1-3
10k
10µF
10nF
3797 TA04
Efficiency vs PVIN
1000:1 PWM Dimming at 120Hz
CTRL1-3 = 2V
PWM1-3 = 2V
95
20k
348k
FBH1
0.068Ω
0.1µF
0.1µF
SENSEN2 SENSEP2 GATE2
200k
100
COUT3
4.7µF
M5
0.068Ω
GATE1 SENSEP1 SENSEN1
VIN
43.2k
EFFICIENCY (%)
M6
LED1+
20k
348k
FBH1
M1
CIN4
4.7µF
TG2
M4
1A
20V
12V
ISN3
ISN2
LED1+
0.1µF
0.25Ω
0.25Ω
M2
L1
33µH
ISP3
ISP2
PWM
5V/DIV
90
IL1
1A/DIV
85
ILED
1A/DIV
80
75
70
VIN = 48V
20
30
40
50
60
PVIN (V)
70
2µs/DIV
3797 TA04c
80
3797 TA04b
3797f
For more information www.linear.com/LT3797
31
LT3797
TYPICAL APPLICATIONS
Triple Buck Mode LED Driver for Common Anode LEDs
CIN1-3
22µF LED1
RED
×6
34k
FBH1
VIN
–15V TO –9V
5A
ISP1
ISP2
ISN1
ISN2
COUT2
22µF ×2 M4
D1
D2
0.01Ω
L3
4.7µH
M5
M3
0.01Ω
VIN
VIN
VIN
SENSEN2 SENSEP2 GATE2
LT3797
FBH1-3
OVLO CTRL1-3
VREF
PWM1-3
RT
SYNC SS1-3
35.7k
400kHz
8.45k
VIN
TG1-3
ISP1-3
EN/UVLO
17.3k
FLT1-3 SW1 SW2
BOOST
LPWR
47µH
0.1µF
10k
RT1
NTC
10k
INTVCC
INTVCC
GND
VC1-3
4.7k
INTVCC
10nF
10µF
0.1µF
3797 TA05
Q1
VIN
VIN
10k
PWMIN1-3
VIN
D1-D3: ON SEMICONDUCTOR MBRB2515L
L1-L3: WÜRTH 7443310470
LED1-LED3: LUMINUS PT39
LPWR: COILTRONICS SD25-470
Efficiency vs VIN
M1, M3, M5: VISHAY SILICONIX Si4174DY
M2, M4, M6: INFINEON BSC130P03LS
Q1: DIODES INC. FMMTL718
RT1: MURATA NCP15XH103J03RC
1000:1 PWM Dimming at 120Hz
PWMIN1-3 = 5V
PWM
5V/DIV
95
EFFICIENCY (%)
GATE3 SENSEP3 SENSEN3
ISN1-3
200k
18.7k
D3
0.01Ω
0.1µF
0.1µF
GATE1 SENSEP1 SENSEN1
VIN
VIN
100
ISN3
COUT3
22µF
×2
M6
TG3
L2
4.7µH
0.1µF
CIN4
10µF
0.05Ω
TG2
L1
4.7µH
M1
5A
20k
ISP3
0.05Ω
COUT1
M2 22µF ×2
75k
FBH3
20k
0.05Ω
TG1
LED3
BLUE
FBH2
20k
5A
LED2
GREEN
75k
90
IL1
5A/DIV
85
ILED
5A/DIV
80
75
70
–15
VIN = –12V
–14
–13
–12
–11
VIN (V)
–10
5µs/DIV
3797 TA05c
–9
3797 TA05b
3797f
32
For more information www.linear.com/LT3797
LT3797
TYPICAL APPLICATIONS
Wide Input Range, Triple Buck-Boost LED Driver
VIN
2.5V TO 40V
(60V TRANSIENT,
41V INTERNAL
OVLO PROTECTION)
4.7µF
×3
L1
22µH
1A
L2
24V 22µH
1A
L3
24V 22µH
562k
562k
4.7µF M2
FBH1
4.7µF M4
1µF
ISP3
1µF
1µF
M3
M5
0.015Ω
0.1µF
SENSEN1 SENSEP1 GATE1
0.015Ω
0.1µF
SENSEN2 SENSEP2 GATE2
0.1µF
SENSEN3 SENSEP3 GATE3 TG1-3
VIN
1µF
0.25Ω
D3
ISP2
ISP1
M1
ISN1-3
47.5k
LT3797
ISP1-3
EN/UVLO
49.9k
TG3
ISN3
24.9k
0.25Ω
D2
4.7µF M6
FBH3
TG2
ISN2
24.9k
0.25Ω
D1
562k
FBH2
TG1
ISN1
24.9k
0.015Ω
1A
24V
FBH1-3
VREF CTRL1-3 PWM1-3
OVLO
VIN
357k
RT
SYNC SS1-3
RT
35.7k
400kHz
75k
FLT1-3 SW1 SW2
LPWR
47µH
0.1µF
BOOST
0.1µF
INTVCC
GND
VC1-3
2k
10µF
10nF
3797 TA06
D1-D3: DIODES INC. PDS5100
L1-L3: COILTRONICS HC9-220R
LPWR: COILTRONICS SD25-470
M1, M3, M5: INFINEON BSC160N10NS3G
M2, M4, M6: VISHAY SILICONIX Si4401BDY
Efficiency and Output Current vs VIN
100
2.2
PWM1-3 = 2V
EFFICIENCY
1.4
80
OUTPUT CURRENT
70
1.0
60
0.6
0
5
10
15
20
25
PWM
5V/DIV
1.8
30
35
40
OUTPUT CURRENT (A)
EFFICIENCY (%)
90
50
1000:1 PWM Dimming at 120Hz
IL1
1A/DIV
ILED
1A/DIV
VIN = 24V
2µs/DIV
3797 TA06c
0.2
VIN (V)
3797 TA06b
3797f
For more information www.linear.com/LT3797
33
LT3797
TYPICAL APPLICATIONS
Triple SEPIC LED Driver
L1A
CDC1
4.7µF
50V
•
COUT1
4.7µF L2A
•
×2
ISP1
D1
CDC2
4.7µF
50V
0.25Ω
L1B
COUT2
4.7µF
×2
ISP2
D2
•
CIN1-3
4.7µF
×3
COUT3
4.7µF
×2
ISP3
D3
0.25Ω
L3B
ISN3
M6
M4
•
M1
M5
M3
1A
24V
0.015Ω
SENSEN1 SENSEP1 GATE1 TG1
1A
24V
0.1µF
0.015Ω
SENSEN2 SENSEP2 GATE2 TG2
1A
24V
0.1µF
SENSEN1 SENSEP1 GATE1 TG1 ISN1-3
VIN
CIN4
1µF
•
ISN2
M2
0.1µF
CDC3
4.7µF
50V
0.25Ω
L2B
ISN1
0.015Ω
L3A
•
VIN
2.5V TO 40V
(60V TRANSIENT,
41V INTERNAL
OVLO PROTECTION)
ISP1-3
47.5k
LT3797
24.9k
EN/UVLO
49.9k
FBH1-3
OVLO VREF CTRL1-3 PWM1-3
VIN
RT
SYNC SS1-3
RT
35.7k
400kHz
487k
FLT1-3 SW1 SW2
0.1µF
BOOST
LPWR
47µH
0.1µF
INTVCC
GND
VC1-3
2k
10µF
562k
10nF
75k
3797 TA07
D1-D3: ON SEMICONDUCTOR MBRS3100
L1-L3: WÜRTH ELEKTRONIK 748870220
LPWR: COILTRONICS SD25-470
M1, M3, M5: INFINEON BSC160N10NS3G
M2, M4, M6: VISHAY SILICONIX Si4401BDY
Efficiency and Output Current vs VIN
100
2.00
PWM1-3 = 2V
95
1.75
EFFICIENCY
1.50
85
1.25
OUTPUT CURRENT
80
1.00
75
0.75
70
0.50
65
0.25
60
0
5
10
15
20
25
30
35
40
PWM
5V/DIV
OUTPUT CURRENT (A)
90
EFFICIENCY (%)
1000:1 PWM Dimming at 120Hz
IL1A+IL1B
2A/DIV
ILED
1A/DIV
VIN = 24V
2µs/DIV
3797 TA07c
0
VIN (V)
3797 TA07b
3797f
34
For more information www.linear.com/LT3797
LT3797
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
UKG Package
Variation UKG52(47)
52-Lead Plastic QFN (7mm × 8mm)
(Reference LTC DWG # 05-08-1874 Rev Ø)
7.50 ±0.05
6.10 ±0.05
5.50 REF
52
41
0.70 ±0.05
40
1
5.70 ±0.05
6.50 REF
7.10 ±0.05 8.50 ±0.05
4.70 ±0.05
27
14
PACKAGE OUTLINE
15
26
0.25 ±0.05
0.50 BSC
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
TOP VIEW
52 51 50
49
7.00 ± 0.10
48 47 46 45 44
43
42 41
0.75 ± 0.05
0.00 – 0.05
PIN 1 TOP MARK
(SEE NOTE 6)
1
41
R = 0.115
TYP
5.50 REF
52
0.40 ± 0.10
40
40
1
2
8.00 ± 0.10
3
38
4
37
5
36
6
35
7
8
33
9
32
10
31
11
30
PIN 1 NOTCH
R = 0.30 TYP OR
0.35 × 45°C
CHAMFER
5.70 ±0.10
6.50 REF
12
4.70 ±0.10
28
27
14
27
R = 0.10
TYP
0.200 REF
0.00 – 0.05
0.75 ± 0.05
15
16
17
19
20
21 22
23 24
25
14
26
15
0.25 ± 0.05
0.50 BSC
BOTTOM VIEW—EXPOSED PAD
(UKG52(47)) QFN REV Ø 0410
26
SIDE VIEW
NOTE:
1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE, IF PRESENT
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE
3797f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection
of its circuits
as described
herein will not infringe on existing patent rights.
For more
information
www.linear.com/LT3797
35
LT3797
TYPICAL APPLICATION
Dual Buck Mode LED Driver with a Boost Pre-Regulator
L1
10µH
VIN
2.5V TO 40V
(60V TRANSIENT,
41V INTERNAL
OVLO PROTECTION)
VOUT1
REGULATED AT 20V WHEN VIN < 20V
FOLLOWS VIN WHEN VIN > 20V
D1
CIN
4.7µF
50V
×2
COUT1
4.7µF
50V
×4
ISP2
ISP3
0.25Ω
0.25Ω
ISN2
TG2
M3
TG3
LED1+
COUT2
4.7µF 1A
50V 16V
×2
ISN3
M5
Efficiency and Output
Current vs VIN
COUT3
1A 4.7µF
16V 50V
×2
100
274k
M1
M2
20k
274k
L3
22µH
20k
FBH3
D3
M4
FBH1
301k
0.068Ω
VREF
0.012Ω
0.1µF
0.1µF
0.068Ω
0.1µF
90
1.4
OUTPUT CURRENT
80
1.0
70
0.6
OUTPUT CURRENT (A)
L2
22µH
FBH2
D2
EFFICIENCY
EFFICIENCY (%)
20k
1.8
PWM1-3 = 2V
N/C
CIN4
1µF
GATE1 SENSEP1 SENSEN1
VIN
TG1 ISP1 ISN1 PWM1 CTRL1 SENSEN2 SENSEP2 GATE2
GATE3 SENSEP3 SENSEN3
49.9k
LT3797
OVLO
60
0
5
10
ISP2-3
EN/UVLO
49.9k
TG1-3
ISN2-3
15
20 25
VIN (V)
VIN
CTRL2-3
PWM2-3
RT
357k
SYNC SS1-3
35.7k
400kHz
75k
FLT1-3 SW1 SW2
0.1µF
BOOST
L4
47µH
INTVCC
0.1µF
GND
10µF
35
40
0.2
3797 TA08b
FBH2-3
VREF
30
VC1-3
4.7k
10nF
3797 TA08
D1: VISHAY SILICONIX 12CWQ06FN
D2, D3: VISHAY SILICONIX 30BQ060
L1: COILTRONICS HC9-100
L2, L3: WURTH 74437349220
L4: COILTRONICS SD25-470
M1: INFINEON BSC110N06NS3-G
M2, M4: VISHAY SILICONIX Si4850EY
M3, M5: VISHAY SILICONIX Si7415DN
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT3476
Quad Output 1.5A, 2MHz High Current LED Driver with
1000:1 Dimming
VIN: 2.8V to 16V, VOUT(MAX) = 36V, PWM Dimming =1000:1,
ISD < 10μA, 5mm × 7mm QFN-10 Package
LT3492
60V, 2.1MHz 3-Channel (ILED = 600mA) Full-Featured
LED Driver
VIN: 3V to 30V (40VMAX), VOUT(MAX) = 45V, PWM Dimming = 3000:1,
ISD < 1μA, 4mm × 5mm QFN-28 and TSSOP Package
LT3496
45V, 2.1MHz 3-Channel (ILED = 750mA) Full-Featured
LED Driver
VIN: 3V to 30V (40VMAX), VOUT(MAX) = 45V, PWM Dimming = 3000:1,
ISD < 1μA, 4mm × 5mm QFN-28 and TSSOP Package
LT3795
110V LED Controller with Spread Spectrum Frequency
Modulation
VIN: 4.5V to 110V, VOUT(MAX) = 110V, ISD < 10μA, TSSOP-28E Package
LT3595
45V, 2MHz 16-Channel Full-Featured LED Driver
VIN: 4.5V to 55V, VOUT(MAX) = 45V PWM Dimming = 5000:1, ISD < 1μA,
5mm × 9mm QFN-56 Package
LT3596
60V Step-Down LED Driver
VIN: 6V to 60V, PWM Dimming = 10000:1, ISD < 1μA,
5mm × 8mm QFN-52 Package
LT3598
44V, 1.5A, 2.5MHz Boost 6-Channel LED Driver
VIN: 3V to 30V (40VMAX), VOUT(MAX) = 44V, PWM Dimming = 1000:1,
ISD < 1μA, 4mm × 4mm QFN-24 Package
LT3599
2A Boost Converter with Internal 4-String 150mA LED
Ballaster
VIN: 3V to 30V, VOUT(MAX) = 44V, PWM Dimming = 1000:1, ISD < 1μA,
5mm × 5mm QFN-32 and TSSOP-28 Packages
LT3754
16-Channel × 50mA LED Driver with 60V Boost
Controller and PWM Dimming
VIN: 6V to 40V, VOUT(MAX) = 45V, PWM Dimming = 3000:1, ISD < 1μA,
5mm × 5mm QFN-32 Package
3797f
36
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
For more information www.linear.com/LT3797
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com/LT3797
LT 1113 • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 2013