Minilogic ML1565 Step-up-down dc-dc convert Datasheet

ML1565
ML1565
Step-Up-Down DC-DC Converts
Digital Still camera Power Supply
High-Efficiency, 5-channel
General Description
Features
The ML1565 provides a complete power-supply solution
for digital still and video cameras through the integration
of ultra-high-efficiency step-up/step-down DC-to-DC converters along with three auxiliary step-up controllers. The
ML1565 is targeted for applications that use either 2 or 3
alkaline or NiMH batteries as well as those using a single
lithium-ion (Li+) battery.
The step-up DC-to-DC converter accepts inputs from 0.7V
to 5.5V and regulates a resistor-adjustable output from
2.7V to 5.5V. It uses internal MOSFETs to achieve
95% efficiency. Adjustable operating frequency facilitates
design for optimum size, cost, and efficiency.
The step-down DC-to-DC converter can produce output
voltages as low as 1.25V and also utilizes internal
MOSFETs to achieve 95% efficiency. An internal softstart
ramp minimizes surge current form the battery. The
converter can operate from the step-up output providing
buck-boost capability with up to 90% compound efficiency,
or it can run directly from the battery if buck-boost
operation is not needed. The ML1565 features auxiliary
step-up controllers that power CCD, LCD, motor actuator,
and backlight circuits. The ML1565 is available in a
space-saving QFN-32 thin package.
Ordering Information
Item
Package Mark
Shipping
ML1565QFNG
QFN-32
160pcs/ Tray
ML1565
Applications
■ Pin Configuration
Step-up DC-to-DC Converter:
95% Efficient
3.3V (Fixed) or 2.7V to 5.5V (Adjustable) Output
Voltage
Step-Down DC-to-DC Converter:
Operate form Battery for 95% Efficient Buck
Combine with Step-Up for 90% Efficient Buck-Boost
Adjustable Output Down to 1.25V
Three Auxiliary PWM Controllers
Up to 1MHz Operating Frequency
1µA Shutdown Mode
Internal Soft-Start control
Overload Protection
Compact 32-Pin, 5mmx5mm Thin QFN Package
Digital Still Cameras
Digital video Cameras
PDAs
■ Typical operating Circuit
ML1565
ML1565
--
P1/21
Rev. C, Sep 2005
ML1565
■ Function Diagram
Absolute Maximum Ratings (VOUTSU=3.3V, TA=0℃
to + 85℃unless otherwise noted.)
Parameter
Ratings
OUTSU_, INSD, SDOK, ON_, FB_, FBSEL_ to GND
PGND to GND
DL_ to PGND
LXSU Current(Note 1)
LXSU Current(Note 1)
REF, OSC, COMP_ to GND
Continuous Power Dissipation(TA = +70℃)
32-Pin Thin QFN (derate 22mW/℃above +70℃)
-0.3V to +6V
-0.3V to +0.3V
-0.3V to OUTSU + 0.3V
3.6A
2.25A
-0.3V to OUTSU + 0.3V
Operating Temperature Range
Junction Temperature
Storage Temperature Range
Lead Temperature (soldering, 10s)
-40℃to + 85℃
+150℃
-65℃ to + 150℃
+300℃
1700mW
Unit
V
V
V
A
A
V
mW
℃
℃
℃
℃
Note 1: LXSU has internal clamp diodes to OUTSU and PGND, and LXSD has internal clamp diodes to
INSD and PGND.
Applications that forward bias these diodes should take care not to exceed the devices power
dissipation limits.
P2/21
Rev. C, Sep 2005
ML1565
Electrical Characteristics (VOUTSU = 3.3V, TA = 0℃
to +85℃,unless otherwise noted.)
General
Parameter
Condition
Min
Input Voltage Range
(Note 2)
ILOAD<1mA, TA = +25℃ startup voltage tempco is
-2300ppm/℃ (typ) (Note 3)
0.7
Minimum Startup Voltage
Typ.
0.9
Overload Protection Fault Interval
100k
Thermal Shutdown
Thermal Shutdown Hysteresis
160
20
Max
Unit
5.5
V
1.1
V
OSC
cycles
℃
℃
Shutdown Supply Current into
OUTSU
ONSU=ONSD=ON1=ON2=ON3=0; OUTSU=3.6V
0.1
5
µA
Step-Up DC-to-DC Supply
Current into OUTSU
ONSU=3.35V, FBSU=1.5V
(does not include switching losses)
290
400
µA
Step-Up Plus 1 AUX Supply
Current into OUTSU
ONSU=ON_=3.35V, FBSU=1.5V, FB_=1.5V
(does not include switching losses)
420
600
µA
Step-Up Plus Step-Down Supply
Current into OUTSU
Reference Output Voltage
Reference Load Regulation
Reference Line Regulation
OSC Discharge Trip Level
OSC Discharge Resistance
OSC Discharge Pulse Width
OSC Frequency
ONSU=ONSD=3.35V, FBSU=1.5V, FBSD=1.5V
(does not include switching losses)
IREF = 20µA
10µA<1REF<200µA
2.7<OUTSU<5.5V
Rising edge
OSC = 1.5V, IOSC = 3mA
470
650
µA
1.225
1.25
4.5
1.3
1.25
52
300
400
1.27
10
5
1.275
80
V
mV
mV
V
Ω
ns
kHz
1.23
ROSC = 40kΩ, COSC = 100pF
Step-Down DC-DC Converter
Parameter
FBSD Regulation Voltage
OUTSD Regulation Voltage
FBSD to COMPSD
Transconductance
FBSD Input Leakage current
Burst Mode Trip Level
Current-Sense Amplifier
Transresistance
LXSD Leakage Current
Switch On-Resistance
Condition
Min
Typ.
Max
Unit
FBSELSD = GND
1.231
1.48
1.25
1.5
1.269
1.52
V
V
FBSD = COMPSD
80
135
185
µS
-100
110
+1
160
+100
190
nA
mA
FBSD = 1.25V
(Note 6)
0.60
XLXSD = 5.5V, OUTSU = 5.5V
VLXSD = 0V, OUTSU = 5.5V
N-channel
P-channel
P-channel Current Limit
N-channel Turn-off Current
0.7
Soft-Start Interval
SDOK Output Low Voltage
SDOK Operating Voltage Range
0.01
0.01
95
150
0.79
20
V/A
20
20
150
250
1.0
4096
FBSD = 0.4V; 0.1mA into SDOK pin
0.002
1.0
P3/21
0.1
5.5
µA
mΩ
A
mA
OSC
cycles
V
V
Rev. C, Sep 2005
ML1565
Step-Up DC-DC Converter
Parameter
Condition
Min
Typ.
Max
Unit
Step-Up Startup-to-Normal
Operating Threshold
Rising or falling edge (Note 4)
2.30
2.5
2.60
V
Step-Up Startup-to-Normal
Operating Threshold Hysteresis
Step-Up Voltage Adjust Range
FBSU Regulation Voltage
OUTSU Regulation Voltage
FBSU to COMPSU Tranconductance
FBSU Input Leakage Current
Burst Mode Trip Level
Current-Sense Amplifier
Transresistance
Step-Up Maximum Duty Cycle
OUTSU Leakage Current
LXSU Leakage Current
Switch On-Resistance
N-Channel Current limit
P-Channel Turn-Off Current
Startup Current Limit
Startup tOFF
Startup Frequency
80
FBSELSU = GND
FBSU = COMPSU
FBSU = 1.25V
(Note 6)
2.7
1.231
3.296
80
-100
150
1.25
3.35
135
+1
200
mV
5.5
1.269
3.404
185
+100
265
0.3
FBSU = 1V
VLXSD = 0V, OUTSU = 5.5V
VLXSU = VOUT = 5.5V
N-channel
P-channel
80
1.6
OUTSU = 1.8V (Note 5)
OUTSU = 1.8V
OUTSU = 1.8V
85
0.01
0.01
95
150
2
20
800
700
200
V
V
V
µS
nA
mA
V/A
90
20
20
150
250
2.4
%
µA
µA
mΩ
A
mA
mA
ns
kHz
Auxiliary DC-DC Controllers (Aux 1, 2, and 3)
Parameter
Condition
Min
Typ.
Max
Unit
Maximum Duty Cycle
FB_ Regulation Voltage
FB_ to COMP_
Transconductance
FB_ = 1V
FB_ = COMP_
FB_ = COMP_
80
1.231
85
1.25
90
1.269
%
V
80
135
185
µS
FB_ Input Leakage Current
AUX1 Output Regulation Voltage
DL_ Driver Resistance
FB_ = 1.25V
FBSEL1 = GND. FB1 connected to AUX1 output
Output high
Output low
Sourcing or sinking
-100
4.93
+1
5
3
2
0.5
4096
+100
5.07
10
5
nA
V
Parameter
Condition
Min
Typ.
Max
Unit
Input Low Level
1.1V < OUTSU < 1.8V (ONSU only)
1.8V < OUTSU < 5.5V
0.2
0.4
V
DL_ Drive Current
Soft-Start Interval
Ω
A
OSC cycle
Logic Inputs (ON_, FBSEL_)
Input High Level
FBSEL_ Input Leakage Current
ON_ Impedance to GND
1.1 < OUTSU < 1.8V (ONSU only)
1.8 < OUTSU < 5.5V
FBSEL = 3.6V, OUTSU = 3.6V
FBSEL = GND, OUTSU = 3.6V
ON_ = 3.35V
P4/21
VOUTSU
-0.2
1.6
-100
-100
V
0
0
330
+100
+100
nA
kΩ
Rev. C, Sep 2005
ML1565
Electrical Characteristics (VOUTSU = 3.3V, TA = -40℃, to + 85℃ unless otherwise specified)
General
Parameter
Condition
Min
Input Voltage Range
(Note 2)
ILOAD < ImA, TA = +25℃ startup voltage tempco is
-2300ppm/℃ (typ) (Note 3)
0.7
Minimum Startup Voltage
Typ.
Max
Unit
5.5
V
1.1
V
5
µA
Shutdown Supply current into
OUTSU
ONSU=ONSD=ON1=ON2=ON3=0; OUTSU=3.6V
Step-Up DC-to-DC Supply
Current into OUTSU
ONSU=3.35V, FBSU=1.5V
(does not include switching losses)
400
µA
Step-Up Plus 1AUX Supply
Current into OUTSU
ONSU=ON_=3.35V, FBSU=1.5V, FB_=1.5V
(does not include switching losses)
600
µA
Step-Up Plus Step-Down Supply
Current into OUTSU
Reference Output Voltage
Reference Load Regulation
Reference Line Regulation
OSC Discharge Trip Level
OSC Discharge Resistance
ONSU=ONSD=3.35V, FBSU=1.5V, FBSD=1.5V
(does not include switching losses)
IREF = 20µA
10µA<IREF<200µA
2.7V<OUTSU<5.5V
Rising edge
OSC = 1.5V, IOSC = 3mA
650
µA
1.27
10
5
1.275
80
V
mV
mV
V
Ω
1.23
1.225
Step-Down DC-DC Converter
Parameter
FBSD Regulation Voltage
OUTSD Regulation Voltage
FBSD to COMPSD
Transconductance
FBSD Input Leakage Current
Burst Mode Trip Level
LXSD Leakage Current
Switch On-Resistance
P-Channel Current Limit
SDOK Output Low Voltage
SDOK Operating Voltage Range
Condition
Max
Unit
FBSELSD = GND
1.225
1.47
Min
1.275
1.53
V
V
FBSD = COMPSD
80
185
µS
-100
110
+100
195
20
20
150
250
1.0
0.1
5.5
nA
mA
µA
µA
mΩ
mΩ
A
V
V
FBSD = 1.25V
(Note 6)
VLXSD = 5.5V, OUTSU = 5.5V
VLXSD = 0V, OUTSU = 5.5V
N-channel
P-channel
0.7
FBSD = 0.4V; 0.1mA into SDOK pin
1.0
P5/21
Typ.
Rev. C, Sep 2005
ML1565
Step-Up DC-DC Converter
Parameter
Condition
Min
Step-Up Startup-to-Normal
Operating Threshold
Rising or falling edge (Note 4)
Step-Up Voltage Adjust Range
FBSU Regulation Voltage
OUTSU Regulation Voltage
Max
Unit
2.30
2.60
V
FBSELSU = GND
2.7
1.225
3.283
5.5
1.275
3.417
V
V
V
FBSU to COMPSU Transconductance
FBSU = COMPSU
80
185
µS
FBSU Input Leakage Current
Burst Mode Trip Level
Step-Up Maximum Duty Cycle
OUTSU Leakage Current
LXSU Leakage Current
FBSU = 1.25V
(Note 6)
FBSU = 1V
VLX = 0V, OUTSU = 5.5V
VLXSU = VOUT = 5.5V
N-channel
P-channel
-100
150
80
+100
275
90
20
20
150
250
2.4
nA
mA
%
µA
µA
Switch On-Resistance
N-Channel Current limit
Typ.
1.6
mΩ
A
Auxiliary DC-DC Controllers (Aux 1, 2, and 3)
Parameter
Condition
Max
Unit
Maximum Duty Cycle
FB_ Regulation Voltage
FB_ to COMP_
Transconductance
FB_ Input Leakage Current
AUX1 Output Regulation Voltage
FB_ = 1V
FB_ = COMP_
80
1.225
90
1.275
%
V
FB_ = COMP_
80
185
µS
-100
4.90
+100
5.10
10
5
nA
V
Max
Unit
0.2
0.4
V
DL_ Driver Resistance
Min
FB_ = 1.25V
FBSEL1 = GND. FB1 connected directly to AUX1 output
Typ.
Output high
Output low
Ω
Logic Inputs (ON_, FBSEL)
Parameter
Condition
Input Low Level
1.1V < OUTSU < 1.8V (ONSU only)
1.8V < OUTSU < 5.5V
Input High Level
Min
1.1V < OUTSU < 1.8V (ONSU only)
Typ.
VOUTSU
-0.2
1.6
-100
-100
V
1.8V < OUTSU < 5.5V
FBSEL = 3.6V, OUTSU = 3.6V
+100
FBSEL_ Input Leakage Current
nA
FBSEL = GND, OUTSU = 3.6V
+100
Note 2: The IC is powered from the OUTSU output.
Note 3: Since the part is powered from OUTSU, a Schottky rectifier, connected from the input battery to OUTSU, is required
for low-voltage startup.
Note 4: The step-up regulator operates in startup mode until this voltage is reached. Do not apply full load current during
startup.
Note 5: The step-up current limit in startup refers to the LXSU switch current limit, not an output current limit.
Note 6: The burst mode current threshold is the transition point between fixed-frequency PWM operation and burst mode
operation (where switching rate varies with load). The spec is given in terms of inductor current. In terms of output
current, the burst mode transition varies with input/output voltage ratio and inductor value. For step-up, the transition
output current is approximately 1/3 the inductor current when stepping from 2V to 3.3V. For step-down, the transition
current in terms of output current is approximately 3/4 the inductor current when stepping down from 3.3V to 1.8V.
P6/21
Rev. C, Sep 2005
ML1565
Pin Description
Pin1
COMP1
Pin2
FB1
Pin3
PGNDA
Pin4
LXSD
Pin5
INSD
Pin6
ONSD
Pin7
COMPSD
Pin8
FBSD
Pin9
ON1
Pin10
ON2
Pin11
ON3
Pin12
ONSU
Pin13
REF
Pin14 FBSU
Pin15
COMPSU
Auxiliary Controller 1 Compensation Node. Connect a series RC from COMP1 To GND to compensate
the control loop. COMP1 is actively driven to GND in shutdown and thermal limit.
Auxiliary Controller 1 Feedback Input. For 5V output, short FBSEL1 to GND and connect FB1 to the
output voltage. For other output voltages, connect FBSEL1 to OUTSU and connect a resistive voltage
divider from the step-up converter output to FB1 to GND. The FB1 feedback threshold is then 1.25V.
This pin is high impedance in shutdown.
Power Ground. Connect PGNDA and PGNDB together and to GND with short trace as close to the IC
as possible.
Step-Down Converter Power-Switching Node. Connect LXSD to the step-down converter inductor.
LXSD is the drain of the P-channel switch and N-channel synchronous rectifier. LXSD is high
impedance in shutdown.
Step-Down Converter Input. INSD can connect to OUTSU, effectively making OUTSD a buck-boost
output from the battery. Bypass to GND with a 1µF ceramic capacitor if connected to OUTSU. INSD
may also be connected to the battery, but should not exceed OUTSU by more than a Schottky diode
forward voltage. Bypass INSD with a 10µF ceramic capacitor when connecting to the battery input. A
10kΩ internal resistance connects OUTSU and INSD.
Step-Down Converter On/Off Control Input. Drive ONSD high to turn on the step-down converter. This
pin has an internal 330kΩ pulldown resistor. ONSD does not start until OUTSU is in regulation.
Step-Down Converter Compensation Node. Connect a series RC from COMPSD to GND to
compensate the control loop. COMPSD is pulled to GND in normal shutdown and during thermal
shutdown (see the Step-Down Compensation section).
Step-Down Converter Feedback Input. For a 1.5V output, short FBSELSD to GND and connect FBSD
to OUTSD. For other voltages, short FBSELSD to OUTSU and connect a resistive voltage-divider
from OUTSD to FBSD to GND. The FBSD feedback threshold is 1.25V. This pin is high impedance in
shutdown.
Auxiliary Controller 1 On/Off control Input Drive ON1 high to turn on. This pin has an internal 330kΩ
pulldown resistor. ON1 cannot start until OUTSU is in regulation.
Auxiliary Controller 2 On/Off control Input Drive ON2 high to turn on. This pin has an internal 330kΩ
pulldown resistor. ON2 cannot start until OUTSU is in regulation.
Auxiliary Controller 3 On/Off control Input Drive ON3 high to turn on. This pin has an internal 330kΩ
pulldown resistor. ON3 cannot start until OUTSU is in regulation.
Step-up Converter On/Off Control. Drive ONSU high to turn on the step-up converter. All other control
pins are locked out until 2ms after the step-up output has reached its final value. This pin has an
internal 330kΩ resistance to GND.
Reference Output. Bypass REF to GND with a 0.1µF or greater capacitor. The maximum allowed load
on REF is 200µA. REF is actively pulled to GND when all converters are shut down.
Step-Up Converter Feedback Input. To regulate OUTSU to 3.35V, connect FBSELSU to GND. FBSU
may be connected to OUTSU or GND. For other output voltages, connect FBSELSU to OUTSU and
connect a resistive voltage-divider from OUTSU to FBSU to GND. The FBSU feedback threshold is
1.25V. This pin is high impedance in shutdown.
Step-Up Converter Compensation Node. Connect a series RC from COMPSU to GND to compensate
the control loop. COMPSD is pulled to GND in normal shutdown and during thermal shutdown (see
the Step-Down Compensation section).
P7/21
Rev. C, Sep 2005
ML1565
Pin Description
Pin16 FBSELSU
Pin17 FBSELSD
Pin18 FBSEL1
Pin19
OSC
Pin20 PGNDB
Pin21 LXSU
Pin22
OUTSUA
Pin23 SDOK
Pin24
COMP3
Pin25 FB3
Pin26
OUTSUB
Pin27
DL3
Pin28
DL2
Pin29
DL1
Pin30
Pin31
GND
COMP2
Pin32 FB2
Exposed EP
Pad
Step-Up Feedback Select Pin. With FBSELSD = GND, OUTSU regulates to 3.35V. With FBSELSD =
OUTSU, FBSU regulates to a 1.25V threshold for use with external feedback resistors. This pin is high
impedance in shutdown.
Step-Down Feedback Select Pin. With FBSELSD = GND, FBSD regulates to 1.5V. With FBSELSD =
OUTSU, FBSD regulates to a 1.25V for use with external feedback resistors. This pin is high
impedance in shutdown.
Auxiliary Controller 1 Feedback Select Pin. With FBSEL1 = GND and FB1 regulates to 5V. With
FBSEL1=OUTSU, FB1 regulates to 1.25V for use with external feedback resistors. This pin is high
impedance in shutdown.
Oscillator Control. Connect a timing capacitor from OSC to GND and a timing resistor from OSC to
OUTSU to set the oscillator frequency between 100kHz and 1MHz. This pin is high impedance in
shutdown.
Power Ground. Connect PGNDA and PGNDB together and to GND with short trace as close to the IC
as possible.
Step-Up Converter Power-Switching Node. Connect LXSU to the step-up converter inductor. LXSU is
high impedance in shutdown.
Step-Up Converter Output. OUTSUA is the power output of the step-up converter. Connect OUTSUA
to OUTSUB at the IC.
This open-drain output goes high impedance when the step-down has successfully completed
soft-start.
Auxiliary Controller 3 Compensation Node. Connect a series resistor-capacitor form COMP3 to GND
to compensate the control loop. COMP3 is actively driven to GND in shutdown and thermal limit.
Auxiliary Controller 3 Feedback Input. Connect a resistive voltage-divider from the output voltage to
FB3 to GND. The FB3 feedback threshold is 1.25V. This pin is high impedance in shutdown.
Step-Up Converter Output. OUTSUB powers the ML1565 and is the sense input when FBSELSU is
GND and the output is 3.3V. Connect OUTSUA to OUTSUB.
Auxiliary Controller 3 Gate-Drive Output. Connect the gate of an N-channel MOSFET to DL3. DL3
swings from GND to OUTSU and supplies up to 500mA. DL3 is driven to GND in shutdown and
thermal limit.
Auxiliary Controller 2 Gate-Drive Output. Connect the gate of an N-channel MOSFET to DL2. DL2
swings from GND to OUTSU and supplies up to 500mA. DL2 is driven to GND in shutdown and
thermal limit.
Auxiliary Controller 1 Gate-Drive Output. Connect the gate of an N-channel MOSFET to DL1. DL1
swings from GND to OUTSU and supplies up to 500mA. DL1 is driven to GND in shutdown and
thermal limit.
Quiet Ground. Connect GND to PGND as close to the IC as possible.
Auxiliary Controller 2 compensation Node. Connect a series resistor-capacitor from COMP2 to GND to
compensate the control loop. COMP2 is actively driven to GND in shutdown and thermal limit.
Auxiliary Controller 2 Feedback Input. Connect a resistive voltage-divider form the output voltage to
FB2 to GND to set the output voltage. The FB2 feedback threshold is 1.25V. This pin is high
impedance in shutdown.
Exposed Underside Metal Pad. This pad must be soldered to the PC board to achieve package
thermal and mechanical ratings. The exposed pad is electrically connected to GND.
P8/21
Rev. C, Sep 2005
ML1565
■ Typical Application Circuit
ML1565
P9/21
Rev. C, Sep 2005
ML1565
Detailed Description (1)
The ML1565 is a complete digital still camera power conversion IC. It can accept input from a variety of sources
including single-cell Li+ batteries, 2-cell alkaline or NiMH
batteries, as well as systems designed to accept both
battery types. The ML1565 includes five DC-to-DC converter channels to generate all required voltages.
1) Synchronous rectified step-up DC-to-DC converter
with on-chip MOSFETs--This typically supplies 3.3V
for main system power.
2) Synchronous rectified step-down DC-to-DC converter
with on-chip MOSFETs---Powering the stepdown
from the step-up output provides efficient (up to 90%)
buck-boost functionality that supplies a regulated
output when the battery voltage is above or below the
output voltage. The step-down can also be powered
from the battery.
3) Auxiliary DC-to-DC Controller 1-- Typically used for
5V output for motor, strobe, or other functions as
required.
4) Auxiliary DC-to-DC Controller 2--Typically supplies
LCD bias voltages with either a multi-output flyback
transformer, or boost converter with charge pump
inverter. Alternately may power white LEDs for LCD
backlighting.
5) Auxiliary DC-to-DC Controller 3 Typically supplies
CCD bias voltages with either a multi-output flyback
transformer, or boost converter with charge pump
inverter.
All ML1565 DC-to-DC converter channels employ fixed
frequency PWM operation. In addition to multiple
DC-to-DC channels, the ML1565 also includes overload
protection, soft-start circuitry, adjustable PWM operating
frequency, and a power-OK(POK) output to signal when
the step-down converter output voltage (for CPU core) is
in regulation.
Step-up DC-to-DC Converter
The step-up DC-to-DC converter channel generates a
2.7V to 5.5V output voltage range from a 0.9V to 5.5V
battery input voltage. An internal switch and synchronous
rectifier allow conversion efficiencies as high as 95% while
reducing both circuit size and the number of external
components. Under moderate to heavy loading,
P10/21
The converter operates in a low-noise PWM mode with
constant frequency. Switching harmonics generated by
fixed-frequency operation are consistent and easily filtered.
The step-up is a current-mode PWM. An error signal (at
COMPSU) represents the difference between the feedback
voltage and the reference. The error signal programs the
inductor current to regulate the output voltage. At light
loads (under 75mA when boosting from 2V to 3.3V),
efficiency is enhanced by burst mode in which switching
occurs only as needed to service the load. In this mode, the
inductor current peak is limited to typically 200mA for each
pulse.
Step-Down DC-to-DC Converter
The step-down DC-to-DC converter channel is optimized
for generating output voltages down to 1.25V. Lower output
voltages can be set by adding an additional resistor (see
the Applications Information section). An internal switch
and synchronous rectifier allow conversion efficiencies as
high as 95% while reducing both circuit size and the
number of external components. Under moderate to heavy
loading, the converter operates in a low-noise PWM mode
with constant frequency. Switching harmonics generated
by fixed-frequency, operation are consistent and easily
filtered. The step-down is a current-mode PWM. An error
signal (at COMPSD) represents the difference between the
feedback voltage and the reference. The error signal
programs the inductor current to regulate the output
voltage. At light loads (under 120 mA), efficiency is
enhanced by burst mode in which switching occurs only as
needed to service the load. In this mode, the inductor
current peak is limited to 150mA (typ) for each pulse.
The step-down remains inactive until the step-up DC-to-DC
is in regulation. This means that the step-down DC-to-DC
on/off pin (ONSD) is overridden by ONSU. The soft-start
sequence for the step-down begins 1024 OSC cycles after
the step-up output is in regulation. If the step-up,
step-down, or any of the auxiliary controllers remains
faulted for 200ms, all channels turn off. The step-down also
features and open-drain SDOK output that goes low when
the output is in regulation.
Rev. C, Sep 2005
ML1565
Detailed Description (2)
● Buck-Boost Operation
The step-down input can be powered from the output of
the step-up. By cascading these two channels, the
step-down output can maintain regulation even as the
battery voltage falls below the step-down output voltage.
This is especially useful when trying to generate 3.3V
from 1-cell Li+ inputs, or 2.5V from 2-cell alkaline or NiMH
inputs, or when designing a power supply that must
operate from both Li+ and alkaline/NiMH inputs.
Compound efficiencies of up to 90% can be achieved
when the step-up and step-down are operated in series.
Note that the step-up output supplies both the step-up
load and the step-down input current when the step-down
is powered from the step-up. The step-down input current
reduces the available step up output current for other
loads.
Direct Battery Step-down Operation
The step-down converter can also be operated directly
from the battery as long as the voltage at INSD does not
exceed OUTSU by more than a Schottky diode forward
voltage. When using this connection, connect a Schottky
diode from the battery input to OUTSU to INSD, which
adds a small additional current drain (of approximately
(VOUTSU – VINSD) / 10kΩ from OUTSU when INSD is not
connected directly to OUTSU.
Step-down direct battery operation improves efficiency for
the step-down output (up to 95%), but limits the upper limit
of the output voltage to 200mV less than the minimum
battery voltage. In 1-cell Li+ designs (with a 2.7V min), the
output can be set up to 2.5V. In 2-cell alkaline or
P11/21
NiMH designs, the output may be limited to 1.5V or 1.8V,
depending on the minimum allowed cell voltage.
The step-down can only be briefly operated in dropout since
the ML1565 fault protection detects the out-of-regulation
condition and activates after 100,000 OSC cycles, or 200ms
at 500kHz. At that point, all ML1565 channels shut down.
Auxiliary DC-to-DC Controllers
The three auxiliary controllers operate as fixed-frequency
voltage-mode PWM controllers. They do not have internal
MOSFETs, so output power is determined by external
components. The controllers regulate output voltage by
modulating the pulse width of the DL_ drive signal to an
external N-channel MOSFET switch.
Figure 3 shows a functional diagram of an AUX controller
channel. A sawtooth oscillator signal at OSC governs timing.
At the start of each cycle, DL_ goes high, turning on the
external N-FET switch. The switch then turns off when the
internally level-shifted sawtooth rises above COMP_ or when
the maximum duty cycle is exceeded. The switch remains off
until the start of the next cycle. A transconductance error
amplifier forms an integrator at COMP_ so that DC high-loop
gain and accuracy can be maintained.
The auxiliary controllers do not start until the step-up
DC-to-DC output is in regulation. If the step-up, step-down or
any of the auxiliary controllers remains faulted for 100,000
OSC cycles, the all ML1565 channels latch off.
Rev. C, Sep 2005
ML1565
Detailed Description (3)
Maximum Duty Cycle
Reference
The ML1565 auxiliary PWM controllers have a guaranteed
maximum duty cycle of 80%. That is to say that all
controllers can achieve at least 80% and typically reach
85%. In boost designs that employ continuous current, the
maximum duty cycle limits the boost ratio such that:
1-VIN / VOUT ≤ 80%
With discontinuous inductor current, no such limit exists
for the input/output ratio since the inductor has time to fully
discharge before the next cycle begins.
The ML1565 has internal 1.250V reference. Connect a
0.1µF ceramic bypass capacitor from REF to GND within
0.2in (5mm) of the REF pin. REF can source up to 200µA
and is enabled whenever ONSD is high and OUTSD is
above 2.5V. If the application requires that REF be loaded
beyond 200µA, it may be buffered with a unity-gain
amplifier or op amp.
Fault Protection
All ML1565 DC-to-DC converter channels employ fixed
frequency PWM operation. The operating frequency is set
by an RC network at the OSC pin, The range of usable
setting is 100kHz to 1MHz.
The ML1565 has robust fault and overload protection.
After power-up, the device is set to detect an out-of
regulation state that could be caused by an overload or
short, If any DC-to-DC converter channel (step-up.
Step-down, or any of the auxiliary controllers) remains
faulted for 100,000 clock cycles, then ALL outputs latch off
until the step-up DC-to-DC converter is reinitialized by the
ONSU pin, or by cycling of input power. The fault-detection
circuitry for any channel is disabled during its initial turn-on
soft-start sequence.
Note that output of the step-up, or that of any auxiliary
channel set up in boost configuration, does not fall to 0V
during shutdown or fault. This is due to the current path
from the battery to the output that remains even when the
channel is off. This path exists through the boost inductor
and the synchronous rectifier body diode. An auxiliary
boost channel falls to the input voltage minus the rectifier
drop during fault and shutdown. OUTSU falls to the input
voltage minus the synchronous rectifier body diode drop
during shutdown, and also during fault if the input voltage
exceeds 2.5V. If the input voltage is less than 2.5V,
OUTSU remains at 2.5V due to operation of the startup
oscillator, but can source only limited current.
Oscillator
The oscillator uses a comparator, a 300ns one-shot, and
an internal N-FET switch in conjunction with an external
timing resistor and capacitor (Figure 4). When the switch is
open, the capacitor voltage exponentially approaches the
step-up output voltage from zero with a time constant given
by the RoscCosc product. The comparator output switches
high when the capacitor voltage reaches VREF(1.25V). In
turn, the one-shot activates the internal MOSFET switch to
discharge the capacitor within a 300ns interval, and the
cycle repeats. Note that the oscillation frequency changes
as the main output voltage ramps upward following startup.
The oscillation frequency is constant once the main output
is in regulation.
ML1565
Figure 4 Master Oscillator
P12/21
Rev. C, Sep 2005
ML1565
Detailed Description (4)
Design Procedure (1)
Low-Voltage Startup Oscillator
Setting the Switching Frequency
The ML1565 internal control and reference-voltage
circuitry receive power from OUTSU and do not function
when OUTSU is less than 2.5V. To ensure low-voltage
startup, the step-up employs a low-voltage startup
oscillator that activates at 0.9V. The startup oscillator
drives the internal N-channel MOSTFET at LXSU until
OUTSU reaches 2.5V, at which point voltage control is
passed to the current-mode PWM circuitry.
Once in regulation, the ML1565 operates with inputs as
low as 0.7V since internal power for the IC is supplied by
OUTSU. At low input voltages, the ML1565 can have
difficulty starting into heavy loads.
Choose a switching frequency to optimize external
component size or circuit efficiency for any particular
ML1565 application. Typically, switching frequencies
between 300kHz and 600kHz offer a good balance
between component size and circuit efficiency. Higher
frequencies generally allow smaller components and lower
frequencies give better conversion efficiency. The switching
frequency is set with an external timing resistor (Rosc) and
capacitor (Cosc). At the beginning of a cycle, the timing
capacitor charges thought the resistor until it reaches VREF.
The charge time, t1, is:
t1 = -RoscCosc In [ 1 - 1.25 / VOUTSU ]
Soft-Start
The ML1565 step-down and AUX_ channels feature a
soft-start function that limits inrush current and prevents
excessive battery loading at startup by ramping the output
voltage to the regulation voltage. This is achieved by
increasing the internal reference inputs to the controller
transconductance amplifiers from 0V to the 1.25V
reference voltage over 4096 oscillator cycles (8ms at
500kHz) when initial power is applied or when a channel
is enabled. Soft-start is not included in the step-up
converter in order to avoid limiting startup capability with
loading.
Shutdown
Table 1. Voltage Setting Summary
Channel
FB THRESHOLD
FB THRESHOLD
FB_
(FBSEL_LOW)
(FBSEL_HIGH)
FBSU
3.35V
1.25V
FBSD
1.5V
FB1
5V
Always 1.25V
FB2
(FBSEL is not provided for these
FB2
channels)
The capacitor voltage is then given time (t2 = 300ns) to
discharge. The oscillator frequency is
fOSC = 1 / (t1 + t2)
The step-up converter is activated with a high input at
ONSU. The step-down and auxiliary DC-to-DC
converters1, 2, and 3 activate with a high input at ONSD,
ON1, ON2, and ON3, respectively. The auxiliary
controllers and step-down cannot be activated until
OUTSU is in regulation. For automatic startup, connect
ON_ to OUTSU or a logic level greater than 1.6V.
fOSC can operate from 100kHz to 1MHz. Choose Cosc
between 47pF and 470pF. Determine Rosc from the
equation:
P13/21
Rev. C, Sep 2005
Rosc = (300ns – 1 / fOSC) / ( Cosc In [ 1-1.25/VOUTSU] )
See the Typical operating Characteristics for fOSC versus
ROSC using different values of COSC.
ML1565
Design Procedure (2)
Setting Output voltages
Step-Up Component Selection
The ML1565 step-up / step-down converters and the
AUX1 controllers have both factory-set and adjustable
output voltages. These are selected by FBSEL_ for the
appropriate channel. When FBSEL_ is low, the channel
output regulates at its preset voltage. When FBSEL_ is
high, the channel regulates FB_ at 1.25V for use with
external feedback resistors.
When setting the voltage for auxiliary channels 2 and 3, or
when using external feedback at FBSU, FBSD, or FB1,
connect a resistive voltage-divider from the output voltage
to the corresponding FB_ input. The FB_ input bias
current is less than 100nA, so choose the low-side
(FB_-to-GND) resistor (RL), to be 100kΩ or less. Then
calculate the high-side (output-to-FB_) resistor (RH) using:
The external components required for the step-up are
an inductor, input and output filter capacitor, and
compensation RC. Typically, the inductor is selected to
operate with continuous current for best efficiency. An
exception might be if the step-up ratio, (VOUT / VIN ), is
greater than 1/(1 – DMAX), where DMAX is the maximum
PWM duty factor of 80%.
When using the step-up channel to boost from a low input
voltage, loaded startup is aided by connecting a Schottky
diode from the battery to OUTSU. See the Minimum
Startup Voltage vs. Load Current graph in the Typical
Operating characteristics.
RH = RL [ (VOUT / 1.25) -1 ]
General Filter Capacitor Selection
The input capacitor in DC-to-DC converter reduces current
peaks drawn from the battery, or other input power source,
and reduces switching noise in the controller. The
impedance of the input capacitor at the switching
frequency should be less than that of the input source so
that high-frequency switching currents do not pass through
the input source.
The output capacitor keeps output ripple small and ensure
control-loop stability. The output capacitor must also have
low impedance at the switching frequency. Ceramic,
polymer, and tantalum capacitors are suitable, with
ceramic exhibiting the lowest ESR and high-frequency
impedance.
Output ripple with a ceramic output capacitor is
approximately:
Step-Up Inductor
In most step-up designs, a reasonable inductor value
(LIDEAL) can be derived from the following equation. Which
sets continuous peak-to-peak inductor current at one-half
the DC inductor current:
LIDEAL = [ 2 VIN(MAX) D (1-D) ] / (IOUT fOSC)
where D is the duty factor given by:
D = 1 – ( VIN / VOUT )
Given LIDEAL , the consistent peak-to peak inductor
current is 0.5IOUT / (1-D).The peak inductor current,
IIND(PK)=1.25IOUT / (1-D). Inductance values smaller than
LIDEAL can be used to reduce inductor size. However, if
much smaller values are used, the inductor current rises
and a larger output capacitance may be required to
suppress output ripple.
VRIPPLE = IL( PEAK ) [ 1 / (2πfOSCCOUT) ]
If the capacitor has significant ESR, the output ripple
component due to capacitor ESR is:
VRIPPLE = IL( PEAK ) ESR
Output capacitor specifics are also discussed in the
Step-Up Compensation section and the Step-Down
Compensation section.
P14/21
Rev. C, Sep 2005
ML1565
Design Procedure (3)
If we select L = 3.3µH then:
fRHPZ = 3.35(2/3.35)2 / (2π x 4.7 x 10-6 x 0.5) = 115kHZ
Step-Up Component Selection
Step-Up compensation
The inductor and output capacitor are usually chosen first
in consideration of performance, size, and cost. The
compensation resistor and capacitor are then chosen to
optimize control-loop stability. In some cases it may help to
readjust the inductor or output capacitor value to get
optimum results. For typical designs, the component
values in the circuit of Figure 1 yield good results.
The step-up converter employs current-mode control,
thereby simplifying the control-loop compensation. When
the converter operates with continuous inductor current
(typically the case), a right-half-plane zero (RHPZ)
appears in the loop-gain frequency response. To ensure
stability, the control-loop gain should crossover (drop
below unity gain) at a frequency (fC) much less than that of
the right-half-plane zero.
The relevant characteristics for step-up channel
compensation are:
1) Transconductance (from FBSU to COMPSU), gmEA
(135µS)
2) Current-sense amplifier transresistance, RCS, (0.3V/A)
3) Feedback regulation voltage, VFB(1.25V)
4) Step-up output voltage, VSUOUT, in V
5) Output load equivalent resistance, RLOAD,
in Ω = VSUOUT / ILOAD
The key steps for step-up compensation are:
1) Place fC sufficiently below the RHPZ and calculate Cc.
2) Select Rc based on the allowed load-step transient. Rc
sets a voltage delta on the COMP pin that corresponds
to load current step.
3) Calculate the output filter capacitor (COUT) required to
allow the Rc and Cc selected.
4) Determine if Cp is required (if calculated to be>10pF).
For continuous conduction, the right-plane zero
frequency (fRHPZ) is given by:
fRHPZ
2
= VOUTSU (1-D) / (2π L ILOAD)
where D = the duty cycle = 1 – (VIN / VOUT), L is the inductor
value, and ILOAD is the maximum output current. Typically
target crossover (fc) for 1/6 the RHPZ. For example, if we
assume VIN = 2V, VOUT = 3.35V, and IOUT = 0.5A, then
RLOAD = 6.7Ω.
Small, High-Efficiency, Five-channel
Digital Still Camera Power Supply
Choose fc = 200kHz. Calculate Cc:
Cc = (VFB/ VOUT)(RLOAD/ RCS)(gm/ 2π fc)(1-D)
=(1.25/3.35)(6.7/0.3) x (135µS/6.28 x 20kHz)
(2/3.35) = 5.35nF
Choose 6.8nF. Now select Rc such that transient droop
requirements are met. For example, if 4% transient droop
is allowed, the input to the error amplifier moves 0.04 x
1.25V, or 50mV. The error amp output drives 50mV x
135µS, or 6.75µA, across Rc to provide transient gain.
Since the current-sense transresistance is 0.3 V/A, the
value of Rc that allows the required load step swing:
Rc = 0.3 IIND(PK) / 6.75µA
In a step-up DC-to-DC converter, if LIDEAL is used, output
current relates to inductor current by:
IIND(PK) = 1.25 IOUT/ (1-D) = 1.25 IOUT VOUT / VIN
Thus, for a 400mA output load step with VIN = 2V and VOUT
= 3.35V:
Rc = [1.25(0.3 x 0.4 x 3.35) / 2]] / 6.75µA = 37kΩ
Note that the inductor does not limit the response in this
case since it can ramp at 2V/3.3µH, or 606mA/µs. The
output filter capacitor is then chosen so that the COUT RLOAD
pole cancels the Rc Cc Zero.
COUT RLOAD = Rc Cc
For example:
COUT = 37KΩ x 6.8nF / 6.7 = 37.5µF
Since a reasonable value for COUT is 47µF rather than 37.5,
choose 47µF and rescale Rc:
Rc = 47µF x 6.7 / 6.8nF = 46.3kΩ
which provides a slightly higher transient gain and
consequently less transient droop than previously selected.
If the output filter capacitor has significant ESR, a zero
occurs at:
ZESR = 1 / (2π COUT RESR)
If ZESR > fc, it can be ignored, as is typically the case with
ceramic output capacitors. If ZESR is less than fc, it should
be cancelled with a pole set by capacitor Cp connected
from COMPSU to GND:
Cp = COUT RESR / RC
If Cp is calculated to be < 10pF, it can be omitted.
P15/21
Rev. C, Sep 2005
ML1565
Design Procedure (4)
Step-Down Component Selection
Step-Down Inductor
The external components required for the step-down are
an inductor, input and output filter capacitors, and
compensation RC network. The ML1565 step-down
converter provides best efficiency with continuous inductor
current. A reasonable inductor value (LIDEAL) can be
derived from:
LIDEAL = 2(VIN) D (1-D) / (IOUT fOSC)
which sets the peak-to-peak inductor current at 1/2 the DC
inductor current. D is the duty cycle:
D = VOUT / VIN
Given LIDEAL, the peak-to-peak inductor current variation is
0.5 IOUT. The absolute peak inductor current is 1.25 IOUT.
Inductance values smaller than LIDEAL can be used to
reduce inductor size. However, if much smaller values are
used, inductor current rises and a large output
capacitance may be required to suppress output ripple.
Larger values than LIEDAL can be used to obtain higher
output current, but with typical larger inductor size.
Step-Down Compensation
The relevant characteristics for step-down compensation
are:
1) Transconductance (from FBSD to COMPSD), gmEA
(135µS)
2) Step-down slope compensation pole,
PSLOPE = VIN / (π L)
3) Current-sense amplifier transresistance,Rcs, (0.6V/A)
4) Feedback regulation voltage, VFB(1.25V)
5) Step-down output voltage, VSD, in V
6) Output load equivalent resistance,
RLOAD in Ω = VOUTSD / ILOAD
The key steps for step-down compensation are:
1) Set the compensation RC zero to cancel the RLOAD
COUT pole.
2) Set the loop crossover below the lower of 1/5 the
slope compensation pole, or 1/5 the switching
frequency.
If we assume VIN = 3.35V, VOUT = 1.5V, and
IOUT = 350mA, then RLOAD = 4.3Ω
If we select L = 4.7µH and fOSC = 440kHz,
Choose 3.3nF. Now select Rc such that transient droop
requirements are met. For example, if 4% transient droop
is allowed, the input to the error amplifier moves 0.04 x
1.25V, or 50mV. The error amp output drives 50mV x
135µS, or 6.75µA across Rc to provide transient gain.
Since the current-sense transresistance is 0.6V/A, the
value of Rc that allows the required load step swing:
RC=0.6 IIND(PK) / 6.75µA
In a step-down DC-to-DC converter, if LIDEAL is used, output
current relates to inductor current by:
IIND(OK) = 1.25 IOUT
Thus, for a 250mA output load step with VIN = 3.35V and
VOUT = 1.5V:
Rc = (1.25 x 0.6 x 0.25) / 6.75µA = 27.8kΩ
Choose 27kΩ. Note that the inductor does not limit the
response in this case since it can ramp at
(VIN-VOUT)/4.7µH, or (3.35 – 1.5)/4.7µH = 394mA/µs
The output filter capacitor is then chosen so that the
COUTRLOAD pole cancels the RcCc zero:
COUTRLOAD = RcCc
For example: COUT = 27kΩ x 3.3nF / 4.3 = 20.7µF
Choose 22µF. If the output filter capacitor has significant
ESR, a zero occurs at:
ZESR = 1 / (2πCOUTRESR)
If ZESR > fc, it can be ignored, as is typically the case with
ceramic output capacitors. If ZESR is less than fc, it should
be cancelled with a pole set by capacitor Cp connected
from COMPAD to GND.
Cp = COUTRESR / Rc
If Cp is calculated to be < 10pF, it can be omitted.
Auxiliary Controller Component Selection
Diode
For most auxiliary applications, a Schottky diode rectifies
the output voltage. The Schottky diode’s low forward
voltage and fast recovery time provide the best
performance in most applications. Silicon signal diodes
(such as 1N4148) are sometimes adequate in low-current
(<10mA) high-voltage (>10V) output circuits where the
output voltage is large compared to the diode forward
voltage.
PSLOPE = VIN / (πL) = 214kHz, so choose fc = 40kHz and
calculate Cc:
Cc = (VFB / VOUT)(RLOAD / RCS) (gm / 2π fc)
=(1.25/1.5)(4.3/0.6) x (135µS/(6.28 x 40kHz))
= 3.2nF
P16/21
Rev. C, Sep 2005
ML1565
Design Procedure (5)
A discontinuous current boost has a single pole at:
fP = (2VOUT – VIN) / (2π RLOAD COUT VOUT)
Auxiliary controller Component Selection
External MOSFET
All ML1565 auxiliary controllers drive external logic-level
N-channel MOSFETs. Significant MOSFET selection
parameters are:
1) On-resistance (RDS(ON))
2) Maximum drain-to-source voltage (VDS(MAX))
3) Total gate charge (QG)
4) Reverse transfer capacitance (CRSS)
DL_ swings between OUTSU and GND. Use a MOSFET
with on-resistance specified at or below the main output
voltage. The gate charge, QG, includes all capacitance
associated with charging the gate and helps to predict
MOSFET transition time between on and off states.
MOSFET power dissipation is a combination of
on-resistance and transition losses. The on-resistance
loss is:
PRDSON = D IL2RDS(ON)
where D is the duty cycle, IL is the average inductor
current, and RDS(ON) is MOSFET on-resistance. The
transition loss is approximately:
PTRANS = (VOUT IL fOSC tT) / 3
were VOUT is the output voltage, IL is the average inductor
current, fOSC is the switching frequency, and tT is the
transition time. The transition time is approximately QG / IG,
where QG is the total gate charge, and IG is the gate drive
current (typically 0.5A). the total power dissipation in the
MOSFET is:
PMOSFET = PRDSON + PTRANS
Auxiliary Compensation
The auxiliary controllers employ voltage-mode control to
regulate their output voltage. Optimum compensation
somewhat depends on whether the design uses
continuous or discontinuous inductor current.
Choose the integrator capacitor such that the unity-gain
Crossover (fC) occurs at fOSC / 10 or lower. Note that for
many auxiliary circuits, such as those powering motors,
LEDs, or other loads that do not require fast transient
response, it is often acceptable to over compensate by
setting fC at fOSC / 20 or lower. Cc is then determined by:
Cc = [2VOUTVIN / (2VOUT – VIN) VRAMP]]
[VOUT / (K(VOUT – VIN))1/2 [VFB / VOUT]
(gM / 2π fC))]
where K = 2 L fOSC / RLOAD, and VRAMP is the internal slope
compensation voltage ramp of 1.25V. The CcRc zero is
then used to cancel the fP pole, so:
Rc = RLOAD COUT VOUT / [ (2VOUT – VIN) Cc]
Continuous Inductor Current
Continuous inductor current can sometimes improve boost
efficiency by lowering the ratio between peak inductor
current and output current. It does this at the expense of a
larger inductance value that requires larger size for a given
current rating. With continuous inductor current boost
operation, there is a right-plane zero at:
fRHPZ = (1-D)2 RLOAD /(2πL)
where (1-D) = VIN / VOUT (in a boost converter). A complex
pole pair is located at:
f0 = VOUT /[2π VIN (L COUT)1/2]
If the zero due to the output capacitor capacitance and
ESR is less than 1/10 the righe-plane zero:
ZCOUT = 1 / (2πCOUT RESR) < fRHPZ / 10
Choose Cc such that the crossover frequency fC occurs at
ZCOUT. The ESR zero provides a phase boost at crossover.
Cc = (VIN / VRAMP)(VFB / VOUT)(gM / (2πZCOUT))
Choose Rc to place the integrator zero, 1/(2πRcCc), at f0 to
cancel one of the pole pairs:
Rc = VIN (L COUT)1/2 / (VOUT Cc)
Discontinuous Inductor Current
When the inductor current falls to zero on each switching
cycle, it is described as discontinuous. The inductor is not
utilized as efficiently as with continuous current. This often
has little negative impact in light-load applications since
the coil losses may already be low compared to other
losses. A benefit of discontinuous inductor current is more
flexible loop compensation and no maximum duty-cycle
restriction on boost ratio.
To ensure discontinuous operation, the inductor must have
a sufficiently low inductance to fully discharge on each
cycle. The occurs when:
L < [ VIN2 (VOUT – VIN) / VOUT3 ] [ RLOAD / (2fOSC) ]
P17/21
If ZCOUT is not less than fRHPZ / 10 (as is typical with ceramic
output capacitors) and continuous conduction is required,
then cross the loop over before fRHPZ and f0:
fC < f0/10, and fC < fRHPZ / 10
In that case:
Cc = (VIN/ VRAMP) (VFB/ VOUT) (gM / 2π fC)
Place 1/ (2πRcCc) = 1 / (2π RLOADCOUT), so that Rc =
RLOADCOUT / Cc or reduce the inductor value for
discontinuous operation.
Rev. C, Sep 2005
ML1565
Applications Information (1)
LED, LCD, and Other Boost Applications
SEPIC Buch-Boost
Any auxiliary channel can be used for a wide variety of
step-up applications. These include generation 5V or
some other voltage for motor or actuator drive, generating
15V or similar voltage for LCD bias, or generating a
step-up current source to efficiently drive a series array of
white LEDs for display backlighting. Figures 5 and 6 show
examples of these applications.
The ML1565 ’s internal switch step-up and step-down can
be cascaded to make a high-efficiency buck-boost
converter, but it may sometimes be desirable to build a
second buck-boost converter with an AUX_ controller. One
type of step-up/step-down converter is the SEPIC (Figure
7). Inductors L1 and L2 can be separate inductors or
wound on a single core and coupled like a transformer.
Typically, a coupled inductor improves efficiency since
some power is transferred through the coupling, causing
less power to pass through the coupling capacitor (C2).
Likewise, C2 should have low ESR to improve efficiency.
The ripple current rating must be greater than the larger of
the input and output currents. The MOSFET (Q1)
drain-to-source voltage rating, and the rectifier (D1)
reverse-voltage rating must exceed the sum of the input
and output voltages. Other types of step-up/step-down
circuits are a flyback converter and a step-up converter
followed by a linear regulator.
Figure 5. Using an AUX_ Controller Channel to
Generate LCD Bias
PART OF
ML 1565
Figure 7. Auxiliary SEPIC Configuration
ML1565
Figure 6. AUX_ Channel Powering a White LED
Step-Up Current Source
P18/21
Rev. C, Sep 2005
ML1565
Applications Information (2)
Multiple Output Flyback Circuits
Some applications require multiple voltages from a single
converter channel. This is often the case when generating
voltages for CCD bias or LCD power. Figure 8 shows a
two-output flyback configuration with AUX_ controller. The
controller drives an external MOSFET that switches the
transformer primary. Two transformer secondaries
generate the output voltages. Only one positive output
voltage can be feedback, so the other voltages are set by
the turn ratio of the transformer secondaries. The load
stability of the other secondary voltages depends on
transformer leakage inductance and winding resistance.
Voltage regulation is best when the load on the secondary
with feedback is small when compared to the load on the
one that is. Regulation also improves if the load current
range is limited. Consult the transformer manufacturer for
the proper design for a given application.
ML1565
Figure 9. ±15V Output Using a Boost with
Charge-Pump Inversion
When the MOSFET turns on, C1 discharges through D3,
thereby charging C3 to VOUT- minus the drop across D3 to
create roughly the same voltage as VOUT+ at VOUT - but with
inverted polarity. If different magnitudes are required for the
positive and negative voltages, a linear regulator can be
used at one of the outputs to achieve the desired voltages.
ML1565
Figure 8. +15V and -7.5V CCD Bias with Transformer
Boost with Charge Pump for Positive and Negative
Outputs
Negative output voltages can be produced without a
transformer, using a charge-pump circuit with an auxiliary
controller as shown in Figure 9. When MOSFET Q1 turns
off, the voltage at its drain rises to supply current to VOUT+.
At the same time, C1 charges to the voltage VOUT+
through D1.
P19/21
Rev. C, Sep 2005
ML1565
Applications Information (3)
Using SDOK for Power Sequencing
SDOK goes low when the step-down reaches regulation.
Some microcontrollers with low-voltage cores require that
the high-voltage (3.3V) I/O rail not be powered up until the
core has a valid supply. The circuit in Figure 11
accomplishes this by driving the gate of a PFET
connected between the 3.3V output and the
microcontroller I/O supply. Alternately, power sequencing
may be implemented by connecting RC networks to the
appropriate converter ON_ inputs.
where VSD is the output voltage, VFBSD is 1.25V, and VSU is
the step-up output voltage. Note that any available voltage
that is higher than 1.25V can be used as the connection
point for R3 on Figure 12 and for the VSD term in the
equation. Since there are multiple solutions for R1, R2, and
R3, the above equation cannot be written in terms of one
resistor. The best method for determining resistor values is
to enter the above equation into a spreadsheet and test
estimated resistors’ values. A good starting point is with
100kΩ at R2 and R3.
ML1565
ML1565
Figure 12. Setting OUTSD for Outputs Below 1.25V
Figure 11. Using SDOK to Gate 3.3V Power to CPU
Designing a PC Board
After the Core Voltage is OK
Setting OUTSD Below 1.25V
The step-down feedback voltage is 1.25V when FBSELSD
is high. With a standard two-resistor feedback network,
the output voltage may be set to values between 1.25V
and the input voltage. If a step-down output voltage less
than 1.25V is desired, it can be set by adding a third
feedback resistor from FB to a voltage higher than 1.25V
(the step-up output is a convenient voltage for this) as
shown in Figure 12. The equation governing output
voltage shown in Figure 12 is:
0= [(VSD – VFBSD) / R1] +[(0 – VFBSD) / R2]
+ [(VSU – VFBSD) / R3]
P20/21
Good PC board layout is important to achieve optimal
performance from the ML1565. Poor design can cause
excessive conducted and/or radiated noise.
Conductors carrying discontinuous currents, and any
high-current path should be made as short and wide as
possible. A separate low-noise ground plane containing the
reference and signal grounds should connect to the
power-ground plane at only one point to minimize the effect
of power-ground currents. Typically, the ground planes are
best joined right at the IC.
Keep the voltage feedback network very close to the IC,
preferably within 0.2in (5mm) of the FB_ pin. Nodes with
high dV/dt (switching nodes) should be kept as small as
possible and should be routed away from high-impedance
nodes such as FB_.
Rev. C, Sep 2005
ML1565
Package Information
PKG.
QFN-32L 5mm x 5mm
Symbol
Min.
Typ.
Max.
M1
4.90
5.00
5.10
M2
4.90
5.00
5.10
B1
0.50 BSC.
B2
(ND-1)xB1
B3
(NE-1)xB1
B4
0.30
0.40
0.50
B5
0.25
-
-
B6
3.00
3.10
3.20
B7
3.00
3.10
3.20
B8
0.20
0.25
0.30
H1
0.70
0.75
0.80
H2
0
0.02
0.05
H3
0.20 REF
JEDEC
The information presented in this document does not form part of any quotation or contract, is believed to be accurate and
reliable and may be changed without notice. No liability will be accepted by the publisher for any consequence of its use.
P21/21
Rev. C, Sep 2005