NOT RECOMMENDED FOR NEW DESIGNS NO RECOMMENDED REPLACEMENT contact our Technical Support Center at 1-888-INTERSIL or www.intersil.com/tsc HIP6311A DATASHEET FN9035 Rev 1.00 July 2004 Microprocessor CORE Voltage Regulator Multi-Phase Buck PWM Controller The HIP6311A Multi-Phase Buck PWM control IC together with HIP6601A, HIP6602A or HIP6603A companion gate drivers form a precision voltage regulation system for advanced microprocessors. The HIP6311A controls microprocessor core voltage regulation by driving 2 to 4 synchronous-rectified buck channels in parallel. The multiphase buck topology takes advantage of interleaving phases to increase ripple frequency and reduce input and output ripple currents. Resulting in fewer components, reduced component ratings, lower power dissipation, and smaller implementation area. The HIP6311A control IC features a 5 bit digital-to-analog converter (DAC) that adjusts the core output voltage from 1.100V to 1.850V with an unsurpassed system accuracy of ±0.5% over temperature. The HIP6311A uses a lossless current sensing approach in which the voltage developed across the on-resistance of the lower MOSFETs during conduction is sampled. Current sensing provides the required signals for precision droop, channel-current balancing, load sharing, and over-current protection. Another feature of this control IC is the PGOOD monitor which is held low until the core voltage increases to within 8% of the programmed voltage. An over-voltage condition is detected when the output voltage exceeds 115% of the programmed VID. This results in the converter shutting down and PGOOD being pulled low. During an under-voltage condition (output voltage 10% below the programmed VID), PGOOD transitions low, but the converter continues to operate. TEMP. (oC) PACKAGE • Microprocessor Voltage Identification Input - 5-Bit VID Input - 1.100V to 1.850V in 25mV Steps - Programmable Droop Voltage • RDS(on) Current Sensing - Accurate Channel Current Balancing - Loss less Current Sampling - Low-Cost Implementation • Fast Transient Response • Digital Soft Start • Over Current Protection • Selection of 2, 3, or 4 Phase Operation • 50kHz to 1.5MHz Switching Frequency • Pb-free available Applications • Desktop Motherboards • Voltage Regulator Modules • Servers and Workstations Pinout PKG. NO. HIP6311ACB 0 to 70 20 Ld SOIC M20.3 HIP6311ACBZ (See Note) 0 to 70 20 Ld SOIC (Pb-free) M20.3 HIP6311ACBZA (See Note) 0 to 70 20 Ld SOIC (Pb-free) M20.3 *Add “-T” suffix to part number for tape and reel packaging. NOTE: Intersil Pb-free products employ special Pb-free material sets; molding compounds/die attach materials and 100% matte tin plate termination finish, which is compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J Std-020B. FN9035 Rev 1.00 July 2004 • Precision CORE Voltage Regulation - ±0.5% System Accuracy Over Temperature HIP6311A (SOIC) TOP VIEW Ordering Information PART NUMBER Features VID4 1 20 VCC VID3 2 19 PGOOD VID2 3 18 PWM4 VID1 4 17 ISEN4 VID0 5 16 ISEN1 COMP 6 15 PWM1 FB 7 14 PWM2 FS/DIS 8 13 ISEN2 GND 9 12 ISEN3 VSEN 10 11 PWM3 Page 1 of 16 HIP6311A Block Diagram VCC PGOOD POWER-ON RESET (POR) + VSEN THREE STATE UV X 0.9 - OV LATCH CLOCK AND SAWTOOTH GENERATOR S + OVP X1.15 + - + + PWM1 PWM - SOFTSTART AND FAULT LOGIC FS/EN - + PWM2 PWM - - COMP + + PWM - VID0 PWM3 - VID1 VID2 + D/A VID3 + VID4 - - E/A CURRENT FB CORRECTION + PWM4 PWM - PHASE NUMBER CHANNEL DETECTOR ISEN1 I_TOT + + OC I_TRIP + ISEN2 + + ISEN3 ISEN4 GND FN9035 Rev 1.00 July 2004 Page 2 of 16 HIP6311A Simplified Power System Diagram SYNCHRONOUS RECTIFIED BUCK CHANNEL VSEN PWM 1 SYNCHRONOUS RECTIFIED BUCK CHANNEL PWM 2 MICROPROCESSOR HIP6311A SYNCHRONOUS RECTIFIED BUCK CHANNEL PWM 3 PWM 4 VID SYNCHRONOUS RECTIFIED BUCK CHANNEL Functional Pin Description converter. Pulling this pin to ground disables the converter and three states the PWM outputs. See Figure 10. VID4 1 20 VCC VID3 2 19 PGOOD VID2 3 18 PWM4 VID1 4 17 ISEN4 VID0 5 16 ISEN1 VSEN (Pin 10) COMP 6 15 PWM1 FB 7 14 PWM2 Power good monitor input. Connect to the microprocessorCORE voltage. FS/DIS 8 13 ISEN2 GND 9 12 ISEN3 VSEN 10 11 PWM3 VID4 (Pin 1), VID3(Pin 2), VID2 (Pin 3), VID1(Pin 4) and VID0 (Pin 5) Voltage Identification inputs from microprocessor. These pins respond to TTL and 3.3V logic signals. The HIP6311A decodes VID bits to establish the output voltage. See Table 1. COMP (Pin 6) Output of the internal error amplifier. Connect this pin to the external feedback and compensation network. FB (Pin 7) Inverting input of the internal error amplifier. FS/DIS (Pin 8) Channel frequency, FSW, select and disable. A resistor from this pin to ground sets the switching frequency of the FN9035 Rev 1.00 July 2004 GND (Pin 9) Bias and reference ground. All signals are referenced to this pin. PWM1 (Pin 15), PWM2 (Pin 14), PWM3 (Pin 11) and PWM4 (Pin 18) PWM outputs for each driven channel in use. Connect these pins to the PWM input of a HIP6601/2/3 driver. For systems which use 3 channels, connect PWM4 high. Two channel systems connect PWM3 and PWM4 high. ISEN1 (Pin 16), ISEN2 (Pin 13), ISEN3 (Pin 12) and ISEN4 (Pin 17) Current sense inputs from the individual converter channel’s phase nodes. Unused sense lines MUST be left open. PGOOD (Pin 19) Power good. This pin provides a logic-high signal when the microprocessor CORE voltage (VSEN pin) is within specified limits and Soft-Start has timed out. VCC (Pin 20) Bias supply. Connect this pin to a 5V supply. Page 3 of 16 HIP6311A Typical Application - 2 Phase Converter Using HIP6601 Gate Drivers +12V BOOT VIN = +5V PVCC UGATE +5V VCC PWM PHASE DRIVER HIP6601 COMP FB LGATE GND VCC VSEN +VCORE PWM4 PGOOD PWM3 VID4 PWM2 VID3 PWM1 VID2 VID1 +12V BOOT VIN = +5V PVCC UGATE MAIN CONTROL HIP6311A PHASE VCC VID0 FS/DIS ISEN4 NC ISEN3 NC PWM DRIVER HIP6601 LGATE GND ISEN2 GND FN9035 Rev 1.00 July 2004 ISEN1 Page 4 of 16 HIP6311A Typical Application - 4 Phase Converter Using HIP6602 Gate Drivers BOOT1 +12V VIN = +12V UGATE1 L01 VCC PHASE1 LGATE1 +5V DUAL DRIVER HIP6602 FB PVCC BOOT2 COMP +5V VIN +12V VCC VSEN UGATE2 L02 ISEN1 PGOOD PWM1 VID4 PWM2 VID3 ISEN2 VID2 VID1 PHASE2 PWM1 PWM2 LGATE2 GND MAIN CONTROL HIP6311A +VCORE VID0 ISEN3 FS/DIS PWM3 PWM4 GND +12V BOOT3 VIN+12V ISEN4 UGATE3 L03 VCC PHASE3 LGATE3 DUAL DRIVER HIP6602 PVCC BOOT4 UGATE4 PWM3 +5V VIN +12V L04 PHASE4 PWM4 LGATE4 GND FN9035 Rev 1.00 July 2004 Page 5 of 16 HIP6311A Absolute Maximum Ratings Thermal Information Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+7V Input, Output, or I/O Voltage . . . . . . . . . . GND -0.3V to VCC + 0.3V ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.5KV Thermal Resistance (Typical, Note 1) JA (oC/W) SOIC Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 87 Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . .150oC Maximum Storage Temperature Range . . . . . . . . . -65oC to 150oC Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . .300oC (SOIC - Lead Tips Only) Recommended Operating Conditions Supply Voltage. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V 5% Ambient Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . 0oC to 70oC CAUTION: Stress above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. NOTE: 1. JA is measured with the component mounted on a low effective thermal conductivity test board in free air. See Tech Brief 379 for details. Electrical Specifications Operating Conditions: VCC = 5V, TA = 0oC to 70oC, Unless Otherwise Specified PARAMETER TEST CONDITIONS MIN TYP MAX UNITS - 10 15 mA INPUT SUPPLY POWER Input Supply Current RT = 100k, Active and Disabled Maximum Limit POR (Power-On Reset) Threshold VCC Rising 4.25 4.38 4.5 V VCC Falling 3.75 3.88 4.00 V System Accuracy Percent system deviation from programmed VID Codes -0.5 - 0.5 % DAC (VID0 - VID4) Input Low Voltage DAC Programming Input Low Threshold Voltage - - 0.8 V DAC (VID0 - VID4) Input High Voltage DAC Programming Input High Threshold Voltage 2.0 - - V VID Pull-Up VIDx = 0V or VIDx = 3V 10 20 40 A Frequency, FSW RT = 100k, 1% 245 275 305 kHz Adjustment Range See Figure 10 0.05 - 1.5 MHz Disable Voltage Maximum voltage at FS/DIS to disable controller. IFS/DIS = 1mA. - - 1.0 V DC Gain RL = 10K to ground - 72 - dB Gain-Bandwidth Product CL = 100pF, RL = 10K to ground - 18 - MHz Slew Rate CL = 100pF, Load = 400A - 5.3 - V/s Maximum Output Voltage Load = 400A 3.6 4.1 - V Minimum Output Voltage Load = -400A - 0.16 0.5 V Full Scale Input Current - 50 - A Over-Current Trip Level - 82.5 - A REFERENCE AND DAC CHANNEL GENERATOR ERROR AMPLIFIER ISEN POWER GOOD MONITOR Under-Voltage Threshold VSEN Rising - 0.92 - VDAC Under-Voltage Threshold VSEN Falling - 0.90 - VDAC PGOOD Low Output Voltage IPGOOD = 4mA - 0.18 0.4 V 1.12 1.15 1.2 VDAC - 2 - % PROTECTION Over-Voltage Threshold VSEN Rising Percent Over-Voltage Hysteresis VSEN Falling after Over-Voltage FN9035 Rev 1.00 July 2004 Page 6 of 16 HIP6311A RIN FB VIN HIP6311A ERROR AMPLIFIER + COMPARATOR CORRECTION + - Q1 PWM CIRCUIT + L01 PWM1 HIP6601 IL1 - Q2 PHASE PROGRAMMABLE REFERENCE DAC + CURRENT RISEN1 ISEN1 SENSING I AVERAGE CURRENT AVERAGING VCORE + + SENSING CORRECTION RLOAD VIN PHASE - COUT RISEN2 ISEN2 CURRENT COMPARATOR + - Q3 PWM CIRCUIT L02 PWM2 HIP6601 IL2 Q4 FIGURE 1. SIMPLIFIED BLOCK DIAGRAM OF THE HIP6311A VOLTAGE AND CURRENT CONTROL LOOPS FOR A TWO POWER CHANNEL REGULATOR Operation Figure 1 shows a simplified diagram of the voltage regulation and current control loops. Both voltage and current feedback are used to precisely regulate voltage and tightly control output currents, IL1 and IL2, of the two power channels. The voltage loop comprises the Error Amplifier, Comparators, gate drivers and output MOSFETS. The Error Amplifier is essentially connected as a voltage follower that has as an input, the Programmable Reference DAC and an output that is the CORE voltage. Voltage Loop Feedback from the CORE voltage is applied via resistor RIN to the inverting input of the Error Amplifier. This signal can drive the Error Amplifier output either high or low, depending upon the CORE voltage. Low CORE voltage makes the amplifier output move towards a higher output voltage level. Amplifier output voltage is applied to the positive inputs of the Comparators via the Correction summing networks. Out-ofphase sawtooth signals are applied to the two Comparators inverting inputs. Increasing Error Amplifier voltage results in FN9035 Rev 1.00 July 2004 increased Comparator output duty cycle. This increased duty cycle signal is passed through the PWM CIRCUIT with no phase reversal and on to the HIP6601, again with no phase reversal for gate drive to the upper MOSFETs, Q1 and Q3. Increased duty cycle or ON time for the MOSFET transistors results in increased output voltage to compensate for the low output voltage sensed. Current Loop The current control loop works in a similar fashion to the voltage control loop, but with current control information applied individually to each channel’s Comparator. The information used for this control is the voltage that is developed across rDS(ON) of each lower MOSFET, Q2 and Q4, when they are conducting. A single resistor converts and scales the voltage across the MOSFETs to a current that is applied to the Current Sensing circuit within the HIP6311A. Output from these sensing circuits is applied to the current averaging circuit. Each PWM channel receives the difference current signal from the summing circuit that compares the average sensed current to the individual channel current. When a power channel’s current is greater than the average current, Page 7 of 16 HIP6311A the signal applied via the summing Correction circuit to the Comparator, reduces the output pulse width of the Comparator to compensate for the detected “above average” current in that channel. Droop Compensation In addition to control of each power channel’s output current, the average channel current is also used to provide CORE voltage “droop” compensation. Average full channel current is defined as 50A. By selecting an input resistor, RIN, the amount of voltage droop required at full load current can be programmed. The average current driven into the FB pin results in a voltage increase across resistor RIN that is in the direction to make the Error Amplifier “see” a higher voltage at the inverting input, resulting in the Error Amplifier adjusting the output voltage lower. The voltage developed across RIN is equal to the “droop” voltage. See the “Current Sensing and Balancing” section for more details. Applications and Convertor Start-Up Each PWM power channel’s current is regulated. This enables the PWM channels to accurately share the load current for enhanced reliability. The HIP6601, HIP6602 or HIP6603 MOSFET driver interfaces with the HIP6311A. For more information, see the HIP6601, HIP6602 or HIP6603 data sheets. The HIP6311A is capable of controlling up to 4 PWM power channels. Connecting unused PWM outputs to VCC automatically sets the number of channels. The phase relationship between the channels is 360o/number of active PWM channels. For example, for three channel operation, the PWM outputs are separated by 120o . Figure 2 shows the PWM output signals for a four channel system. PWM 1 PWM 2 PWM 3 PWM 4 FIGURE 2. FOUR PHASE PWM OUTPUT AT 500kHz Power supply ripple frequency is determined by the channel frequency, FSW, multiplied by the number of active channels. FN9035 Rev 1.00 July 2004 For example, if the channel frequency is set to 250kHz and there are three phases, the ripple frequency is 750kHz. The IC monitors and precisely regulates the CORE voltage of a microprocessor. After initial start-up, the controller also provides protection for the load and the power supply. The following section discusses these features. Initialization The HIP6311A usually operates from an ATX power supply. Many functions are initiated by the rising supply voltage to the VCC pin of the HIP6311A. Oscillator, Sawtooth Generator, SoftStart and other functions are initialized during this interval. These circuits are controlled by POR, Power-On Reset. During this interval, the PWM outputs are driven to a three state condition that makes these outputs essentially open. This state results in no gate drive to the output MOSFETs. Once the VCC voltage reaches 4.375V (+125mV), a voltage level to insure proper internal function, the PWM outputs are enabled and the Soft-Start sequence is initiated. If for any reason, the VCC voltage drops below 3.875V (+125mV). the POR circuit shuts the converter down and again three states the PWM outputs. Soft-Start After the POR function is completed with VCC reaching 4.375V, the Soft-Start sequence is initiated. Soft-Start, by its slow rise in CORE voltage from zero, avoids an over-current condition by slowly charging the discharged output capacitors. This voltage rise is initiated by an internal DAC that slowly raises the reference voltage to the error amplifier input. The voltage rise is controlled by the oscillator frequency and the DAC within the HIP6311A, therefore, the output voltage is effectively regulated as it rises to the final programmed CORE voltage value. For the first 32 PWM switching cycles, the DAC output remains inhibited and the PWM outputs remain three stated. From the 33rd cycle and for another, approximately 150 cycles the PWM output remains low, clamping the lower output MOSFETs to ground, see Figure 3. The time variability is due to the Error Amplifier, Sawtooth Generator and Comparators moving into their active regions. After this short interval, the PWM outputs are enabled and increment the PWM pulse width from zero duty cycle to operational pulse width, thus allowing the output voltage to slowly reach the CORE voltage. The CORE voltage will reach its programmed value before the 2048 cycles, but the PGOOD output will not be initiated until the 2048th PWM switching cycle. The Soft-Start time or delay time, DT = 2048/FSW. For an oscillator frequency, FSW, of 200kHz, the first 32 cycles or 160s, the PWM outputs are held in a three state level as explained above. After this period and a short interval described above, the PWM outputs are initiated and the voltage rises in 10.08ms, for a total delay time DT of 10.24ms. Figure 3 shows the start-up sequence as initiated by a fast rising 5V supply, VCC, applied to the HIP6311A. Note the short Page 8 of 16 HIP6311A rise to the three state level in PWM 1 output during first 32 PWM cycles. 12V ATX SUPPLY DELAY TIME PWM 1 OUTPUT PGOOD PGOOD VCORE VCORE 5 V ATX SUPPLY 5V VCC VIN = 5V, CORE LOAD CURRENT = 31A FREQUENCY 200kHz ATX SUPPLY ACTIVATED BY ATX “PS-ON PIN” FIGURE 5. SUPPLY POWERED BY ATX SUPPLY VIN = 12V FIGURE 3. START-UP OF 4 PHASE SYSTEM OPERATING AT 500kHz Figure 4 shows the waveforms when the regulator is operating at 200kHz. Note that the Soft-Start duration is a function of the Channel Frequency as explained previously. Also note the pulses on the COMP terminal. These pulses are the current correction signal feeding into the comparator input (see the Block Diagram on page 2.) V COMP Note that Figure 5 shows the 12V gate driver voltage available before the 5V supply to the HIP6311A has reached its threshold level. If conditions were reversed and the 5V supply was to rise first, the start-up sequence would be different. In this case the HIP6311A will sense an over-current condition due to charging the output capacitors. The supply will then restart and go through the normal Soft-Start cycle. Fault Protection The HIP6311A protects the microprocessor and the entire power system from damaging stress levels. Within the HIP6311A both Over-Voltage and Over-Current circuits are incorporated to protect the load and regulator. Over-Voltage DELAY TIME PGOOD VCORE 5V VCC VIN = 12V FIGURE 4. START-UP OF 4 PHASE SYSTEM OPERATING AT 200kHz Figure 5 shows the regulator operating from an ATX supply. In this figure, note the slight rise in PGOOD as the 5V supply rises.The PGOOD output stage is made up of NMOS and PMOS transistors. On the rising VCC, the PMOS device becomes active slightly before the NMOS transistor pulls “down”, generating the slight rise in the PGOOD voltage. FN9035 Rev 1.00 July 2004 The VSEN pin is connected to the microprocessor CORE voltage. A CORE over-voltage condition is detected when the VSEN pin goes more than 15% above the programmed VID level. The over-voltage condition is latched, disabling normal PWM operation, and causing PGOOD to go low. The latch can only be reset by lowering and returning VCC high to initiate a POR and Soft-Start sequence. During a latched over-voltage, the PWM outputs will be driven either low or three state, depending upon the VSEN input. PWM outputs are driven low when the VSEN pin detects that the CORE voltage is 15% above the programmed VID level. This condition drives the PWM outputs low, resulting in the lower or synchronous rectifier MOSFETS to conduct and shunt the CORE voltage to ground to protect the load. If after this event, the CORE voltage falls below the overvoltage limit (plus some hysteresis), the PWM outputs will three state. The HIP6601 family drivers pass the three state information along, and shuts off both upper and lower MOSFETs. This prevents “dumping” of the output capacitors back through the lower MOSFETs, avoiding a possibly Page 9 of 16 HIP6311A destructive ringing of the capacitors and output inductors. If the conditions that caused the over-voltage still persist, the PWM outputs will be cycled between three state and VCORE clamped to ground, as a hysteretic shunt regulator. Under-Voltage The VSEN pin also detects when the CORE voltage falls more than 10% below the VID programmed level. This causes PGOOD to go low, but has no other effect on operation and is not latched. There is also hysteresis in this detection point. CORE Voltage Programming The voltage identification pins (VID0, VID1, VID3, and VID4) set the CORE output voltage. Each VID pin is pulled to VCC by an internal 20A current source and accepts opencollector/open-drain/open-switch-to-ground or standard lowvoltage TTL or CMOS signals. Table 1 shows the nominal DAC voltage as a function of the VID codes. The power supply system is 0.5% accurate over the operating temperature and voltage range. Over-Current TABLE 1. VOLTAGE IDENTIFICATION CODES In the event of an over-current condition, the over-current protection circuit reduces the average current delivered to less than 25% of the current limit. When an over-current condition is detected, the controller forces all PWM outputs into a three state mode. This condition results in the gate driver removing drive to the output stages.The HIP6311A goes into a wait delay timing cycle that is equal to the Soft-Start ramp time. PGOOD also goes “low” during this time due to VSEN going below its threshold voltage.To lower the average output dissipation, the Soft-Start initial wait time is increased from 32 to 2048 cycles, then the Soft-Start ramp is initiated. At a PWM frequency of 200kHz, for instance, an over-current detection would cause a dead time of 10.24ms, then a ramp of 10.08ms. At the end of the delay, PWM outputs are restarted and the Soft-Start ramp is initiated. If a short is present at that time, the cycle is repeated. This is the hiccup mode. Figure 6 shows the supply shorted under operation and the hiccup operating mode described above. Note that due to the high short circuit current, over-current is detected before completion of the start-up sequence so the delay is not quite as long as the normal Soft-Start cycle. SHORT APPLIED HERE PGOOD SHORT CURRENT 50A/Div VID4 VID3 VID2 VID1 VID0 VDAC 1 1 1 1 1 Off 1 1 1 1 0 1.100 1 1 1 0 1 1.125 1 1 1 0 0 1.150 1 1 0 1 1 1.175 1 1 0 1 0 1.200 1 1 0 0 1 1.225 1 1 0 0 0 1.250 1 0 1 1 1 1.275 1 0 1 1 0 1.300 1 0 1 0 1 1.325 1 0 1 0 0 1.350 1 0 0 1 1 1.375 1 0 0 1 0 1.400 1 0 0 0 1 1.425 1 0 0 0 0 1.450 0 1 1 1 1 1.475 0 1 1 1 0 1.500 0 1 1 0 1 1.525 0 1 1 0 0 1.550 0 1 0 1 1 1.575 0 1 0 1 0 1.600 HICCUP MODE. SUPPLY POWERED BY ATX SUPPLY CORE LOAD CURRENT = 31A, 5V LOAD = 5A SUPPLY FREQUENCY = 200kHz, V IN = 12V 0 1 0 0 1 1.625 0 1 0 0 0 1.650 ATX SUPPLY ACTIVATED BY ATX “PS-ON PIN” 0 0 1 1 1 1.675 FIGURE 6. SHORT APPLIED TO SUPPLY AFTER POWER-UP 0 0 1 1 0 1.700 0 0 1 0 1 1.725 0 0 1 0 0 1.750 0 0 0 1 1 1.775 0 0 0 1 0 1.800 0 0 0 0 1 1.825 0 0 0 0 0 1.850 FN9035 Rev 1.00 July 2004 Page 10 of 16 HIP6311A RIN RFB Cc COMP FB VIN HIP6311A - CORRECTION + - PWM CIRCUIT + L01 Q1 PWM VCORE HIP6601 IL Q2 + - PHASE DIFFERENCE + REFERENCE DAC RLOAD COMPARATOR GENERATOR COUT SAWTOOTH ERROR AMPLIFIER CURRENT ISEN RISEN SENSING CURRENT SENSING FROM OTHER CHANNELS TO OTHER CHANNELS AVERAGING TO OVER CURRENT TRIP + ONLY ONE OUTPUT STAGE SHOWN INDUCTOR CURRENT(S) FROM OTHER CHANNELS COMPARATOR REFERENCE FIGURE 7. SIMPLIFIED FUNCTIONAL BLOCK DIAGRAM SHOWING CURRENT AND VOLTAGE SAMPLING Current Sensing and Balancing Overview The HIP6311A samples the on-state voltage drop across each synchronous rectifier FET, Q2, as an indication of the inductor current in that phase, see Figure 7. Neglecting AC effects (to be discussed later), the voltage drop across Q2 is simply rDS(ON)(Q2) x inductor current (IL). Note that IL, the inductor current, is either 1/2, 1/3, or 1/4 of the total current (ILT), depending on how many phases are in use. The voltage at Q2’s drain, the PHASE node, is applied to the RISEN resistor to develop the IISEN current to the HIP6311A ISEN pin. This pin is held at virtual ground, so the current through RISEN is IL x rDS(ON)(Q2) / RISEN. The IISEN current provides information to perform the following functions: 1. Detection of an over-current condition 2. Reduce the regulator output voltage with increasing load current (droop) 3. Balance the IL currents in multiple channels Over-Current, Selecting RISEN The current detected through the RISEN resistor is averaged with the current(s) detected in the other 1, 2, or 3 channels. The aver- FN9035 Rev 1.00 July 2004 aged current is compared with a trimmed, internally generated current, and used to detect an over-current condition. The nominal current through the RISEN resistor should be 50A at full output load current, and the nominal trip point for over-current detection is 165% of that value, or 82.5A. Therefore, RISEN = IL x rDS(ON) (Q2) / 50A. For a full load of 25A per phase, and an rDS(ON) (Q2) of 4m, RISEN = 2k. The over-current trip point would be 165% of 25A, or ~ 41A per phase. The RISEN value can be adjusted to change the overcurrent trip point, but it is suggested to stay within 25%of nominal. Droop, Selection of RIN The average of the currents detected through the RISEN resistors is also steered to the FB pin. There is no DC return path connected to the FB pin except for RIN, so the average current creates a voltage drop across RIN. This drop increases the apparent VCORE voltage with increasing load current, causing the system to decrease VCORE to maintain balance at the FB pin. This is the desired “droop” voltage used to maintain VCORE within limits under transient conditions. With a high dv/dt load transient, typical of high performance microprocessors, the largest deviations in output voltage occur Page 11 of 16 HIP6311A RIN should be selected to give the desired “droop” voltage at the normal full load current 50A applied through the RISEN resistor (or at a different full load current if adjusted as under “Over-Current, Selecting RISEN” above). 25 20 AMPERES at the leading and trailing edges of the load transient. In order to fully utilize the output-voltage tolerance range, the output voltage is positioned in the upper half of the range when the output is unloaded and in the lower half of the range when the controller is under full load. This droop compensation allows larger transient voltage deviations and thus reduces the size and cost of the output filter components. 15 10 5 0 RIN = Vdroop/50A For a Vdroop of 80mV, RIN = 1.6k The AC feedback components, RFB and Cc, are scaled in relation to RIN. FIGURE 8. TWO CHANNEL MULTIPHASE SYSTEM WITH CURRENT BALANCING DISABLED Current Balancing The detected currents are also used to balance the phase currents. The balancing circuit can not make up for a difference in rDS(ON) between synchronous rectifiers. If a FET has a higher rDS(ON), the current through that phase will be reduced. 25 20 AMPERES Each phase’s current is compared to the average of all phase currents, and the difference is used to create an offset in that phase’s PWM comparator. The offset is in a direction to reduce the imbalance. 15 10 5 0 Figures 8 and 9 show the inductor current of a two phase system without and with current balancing. Inductor Current The inductor current in each phase of a multi-phase Buck converter has two components. There is a current equal to the load current divided by the number of phases (ILT / n), and a sawtooth current, (iPK-PK) due to switching. The sawtooth component is dependent on the size of the inductors, the switching frequency of each phase, and the values of the input and output voltage. Ignoring secondary effects, such as series resistance, the peak to peak value of the sawtooth current can be described by: iPK-PK = (VIN x VCORE - VCORE2) / (L x FSW x VIN) Where: VCORE VIN L FSW = DC value of the output or VID voltage = DC value of the input or supply voltage = value of the inductor = switching frequency Example: For VCORE VIN L FSW Then iPK-PK = 4.3A FN9035 Rev 1.00 July 2004 = 1.6V, = 12V, = 1.3H, = 250kHz, FIGURE 9. TWO CHANNEL MULTIPHASE SYSTEM WITH CURRENT BALANCING ENABLED The inductor, or load current, flows alternately from VIN through Q1 and from ground through Q2. The HIP6311A samples the on-state voltage drop across each Q2 transistor to indicate the inductor current in that phase. The voltage drop is sampled 1/3 of a switching period, 1/FSW, after Q1 is turned OFF and Q2 is turned on. Because of the sawtooth current component, the sampled current is different from the average current per phase. Neglecting secondary effects, the sampled current (ISAMPLE) can be related to the load current (ILT) by: ISAMPLE = ILT / n + (VINVCORE - 3VCORE2) / (6L x FSW x VIN) Where: ILT = total load current n = the number of channels Example: Using the previously given conditions, and For ILT = 100A, n =4 Then ISAMPLE = 25.49A Page 12 of 16 HIP6311A As discussed previously, the voltage drop across each Q2 transistor at the point in time when current is sampled is rDSON (Q2) x ISAMPLE. The voltage at Q2’s drain, the PHASE node, is applied through the RISEN resistor to the HIP6311A ISEN pin. This pin is held at virtual ground, so the current into ISEN is: ISENSE = ISAMPLE x rDS(ON) (Q2) / RISEN. RIsen = ISAMPLE x rDS(ON) (Q2) / 50A Example: From the previous conditions, where ILT = 100A, ISAMPLE = 25.49A, rDS(ON) (Q2) = 4m Then: RISEN = 2.04K and ICURRENT TRIP = 165% Short circuit ILT = 165A. Channel Frequency Oscillator The channel oscillator frequency is set by placing a resistor, RT, to ground from the FS/DIS pin. Figure 10 is a curve showing the relationship between frequency, FSW, and resistor RT. To avoid pickup by the FS/DIS pin, it is important to place this resistor next to the pin. If this pin is also used to disable the converter, it is also important to locate the pull-down device next to this pin. There are two sets of critical components in a DC-DC converter using a HIP6311A controller and a HIP6601 gate driver. The power components are the most critical because they switch large amounts of energy. Next are small signal components that connect to sensitive nodes or supply critical bypassing current and signal coupling. The power components should be placed first. Locate the input capacitors close to the power switches. Minimize the length of the connections between the input capacitors, CIN, and the power switches. Locate the output inductors and output capacitors between the MOSFETs and the load. Locate the gate driver close to the MOSFETs. The critical small components include the bypass capacitors for VCC and PVCC on the gate driver ICs. Locate the bypass capacitor, CBP , for the HIP6311A controller close to the device. It is especially important to locate the resistors associated with the input to the amplifiers close to their respective pins, since they represent the input to feedback amplifiers. Resistor RT, that sets the oscillator frequency should also be located next to the associated pin. It is especially important to place the RSEN resistor(s) at the respective terminals of the HIP6311A. 1,000 500 200 100 50 RT (k) causes voltage spikes across the interconnecting impedances and parasitic circuit elements. These voltage spikes can degrade efficiency, radiate noise into the circuit and lead to device over-voltage stress. Careful component layout and printed circuit design minimizes the voltage spikes in the converter. Consider, as an example, the turnoff transition of the upper PWM MOSFET. Prior to turnoff, the upper MOSFET was carrying channel current. During the turnoff, current stops flowing in the upper MOSFET and is picked up by the lower MOSFET. Any inductance in the switched current path generates a large voltage spike during the switching interval. Careful component selection, tight layout of the critical components, and short, wide circuit traces minimize the magnitude of voltage spikes. Contact Intersil for evaluation board drawings of the component placement and printed circuit board. 20 10 5 2 1 10 20 50 100 200 500 1,000 2,000 5,000 10,000 CHANNEL OSCILLATOR FREQUENCY, FSW (kHz) FIGURE 10. RESISTANCE RT vs FREQUENCY Layout Considerations MOSFETs switch very fast and efficiently. The speed with which the current transitions from one device to another FN9035 Rev 1.00 July 2004 A multi-layer printed circuit board is recommended. Figure 11 shows the connections of the critical components for one output channel of the converter. Note that capacitors CIN and COUT could each represent numerous physical capacitors. Dedicate one solid layer, usually the middle layer of the PC board, for a ground plane and make all critical component ground connections with vias to this layer. Dedicate another solid layer as a power plane and break this plane into smaller islands of common voltage levels. Keep the metal runs from the PHASE terminal to inductor LO1 short. The power plane should support the input power and output power nodes. Use copper filled polygons on the top and bottom circuit layers for the phase nodes. Use the remaining printed circuit layers for small signal wiring. The wiring traces from the driver IC to the MOSFET gate and source should be sized to carry at least one ampere of current. Page 13 of 16 HIP6311A +5VIN USE INDIVIDUAL METAL RUNS FOR EACH CHANNEL TO HELP ISOLATE OUTPUT STAGES +12V CBP LOCATE NEXT TO IC PIN(S) CBP CT VCC CBOOT CIN LOCATE NEAR TRANSISTOR LO1 HIP6601 VCORE PHASE COMP FS/DIS COUT RT HIP6311A RFB LOCATE NEXT TO FB PIN PWM VCC PVCC FB LOCATE NEXT TO IC PIN RSEN RIN VSEN ISEN KEY ISLAND ON POWER PLANE LAYER ISLAND ON CIRCUIT PLANE LAYER VIA CONNECTION TO GROUND PLANE FIGURE 11. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS Component Selection Guidelines Output Capacitor Selection The output capacitor is selected to meet both the dynamic load requirements and the voltage ripple requirements. The load transient for the microprocessor CORE is characterized by high slew rate (di/dt) current demands. In general, multiple high quality capacitors of different size and dielectric are paralleled to meet the design constraints. Modern microprocessors produce severe transient load rates. High frequency capacitors supply the initially transient current and slow the load rate-of-change seen by the bulk capacitors. The bulk filter capacitor values are generally determined by the ESR (effective series resistance) and voltage rating requirements rather than actual capacitance requirements. High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. Use only specialized low-ESR capacitors intended for switching-regulator applications for the bulk capacitors. The bulk capacitor’s ESR determines the output ripple voltage and the initial voltage drop following a high slew-rate transient’s edge. In most cases, multiple capacitors of small case size perform better than a single large case capacitor. Bulk capacitor choices include aluminum electrolytic, OSCon, Tantalum and even ceramic dielectrics. An aluminum electrolytic capacitor’s ESR value is related to the case size FN9035 Rev 1.00 July 2004 with lower ESR available in larger case sizes. However, the equivalent series inductance (ESL) of these capacitors increases with case size and can reduce the usefulness of the capacitor to high slew-rate transient loading. Unfortunately, ESL is not a specified parameter. Consult the capacitor manufacturer and measure the capacitor’s impedance with frequency to select a suitable component. Output Inductor Selection One of the parameters limiting the converter’s response to a load transient is the time required to change the inductor current. Small inductors in a multi-phase converter reduces the response time without significant increases in total ripple current. The output inductor of each power channel controls the ripple current. The control IC is stable for channel ripple current (peak-to-peak) up to twice the average current. A single channel’s ripple current is approximately: V IN – V OUT V OUT I = -------------------------------- ---------------V IN F SW L The current from multiple channels tend to cancel each other and reduce the total ripple current. Figure 12 gives the total ripple current as a function of duty cycle, normalized to the parameter Vo LxF SW at zero duty cycle. To determine the total ripple current from the number of channels and the duty cycle, multiply the y-axis value by Vo LxF SW . Small values of output inductance can cause excessive power dissipation. The HIP6311A is designed for stable operation for ripple currents up to twice the load current. However, for this condition, the RMS current is 115% above the value shown in the following MOSFET Selection and Page 14 of 16 HIP6311A the high frequency decoupling and bulk capacitors to supply the RMS current. Small ceramic capacitors should be placed very close to the drain of the upper MOSFET to suppress the voltage induced in the parasitic circuit impedances. Considerations section. With all else fixed, decreasing the inductance could increase the power dissipated in the MOSFETs by 30%. For bulk capacitance, several electrolytic capacitors (Panasonic HFQ series or Nichicon PL series or Sanyo MV-GX or equivalent) may be needed. For surface mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to the capacitor surge current rating. These capacitors must be capable of handling the surgecurrent at power-up. The TPS series available from AVX, and the 593D series from Sprague are both surge current tested. SINGLE CHANNEL 0.8 VO / (LX FSW) RIPPLE CURRENT (APEAK-PEAK) 1.0 0.6 2 CHANNEL 0.4 3 CHANNEL 0.2 MOSFET Selection and Considerations 4 CHANNEL 0 0 0.1 0.2 0.3 0.4 0.5 DUTY CYCLE (VO/VIN) FIGURE 12. RIPPLE CURRENT vs DUTY CYCLE Input Capacitor Selection The important parameters for the bulk input capacitors are the voltage rating and the RMS current rating. For reliable operation, select bulk input capacitors with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. The capacitor voltage rating should be at least 1.25 times greater than the maximum input voltage and a voltage rating of 1.5 times is a conservative guideline. The RMS current required for a multi-phase converter can be approximated with the aid of Figure 13. CURRENT MULTIPLIER 0.5 SINGLE CHANNEL 0.4 0.3 2 CHANNEL The equations assume linear voltage-current transitions and do not model power loss due to the reverse-recovery of the lower MOSFETs body diode. The gate-charge losses are dissipated by the Driver IC and don't heat the MOSFETs. However, large gate-charge increases the switching time, tSW which increases the upper MOSFET switching losses. Ensure that both MOSFETs are within their maximum junction temperature at high ambient temperature by calculating the temperature rise according to package thermal-resistance specifications. A separate heatsink may be necessary depending upon MOSFET power, package type, ambient temperature and air flow. 2 0.2 I O r DS ON V OUT I O V IN t SW F SW P UPPER = ------------------------------------------------------------ + ---------------------------------------------------------V IN 2 3 CHANNEL 0.1 0 In high-current PWM applications, the MOSFET power dissipation, package selection and heatsink are the dominant design factors. The power dissipation includes two loss components; conduction loss and switching loss. These losses are distributed between the upper and lower MOSFETs according to duty factor (see the following equations). The conduction losses are the main component of power dissipation for the lower MOSFETs, Q2 and Q4 of Figure 1. Only the upper MOSFETs, Q1 and Q3 have significant switching losses, since the lower device turns on and off into near zero voltage. 4 CHANNEL 0 0.1 2 I O r DS ON V IN – V OUT P LOWER = --------------------------------------------------------------------------------V IN 0.2 0.3 0.4 0.5 DUTY CYCLE (VO/VIN) FIGURE 13. CURRENT MULTIPLIER vs DUTY CYCLE First determine the operating duty ratio as the ratio of the output voltage divided by the input voltage. Find the Current Multiplier from the curve with the appropriate power channels. Multiply the current multiplier by the full load output current. The resulting value is the RMS current rating required by the input capacitor. Use a mix of input bypass capacitors to control the voltage overshoot across the MOSFETs. Use ceramic capacitance for FN9035 Rev 1.00 July 2004 A diode, anode to ground, may be placed across Q2 and Q4. These diodes function as a clamp that catches the negative inductor swing during the dead time between the turn off of the lower MOSFETs and the turn on of the upper MOSFETs. The diodes must be a Schottky type to prevent the lossy parasitic MOSFET body diode from conducting. It is usually acceptable to omit the diodes and let the body diodes of the lower MOSFETs clamp the negative inductor swing, but efficiency could drop one or two percent as a result. The diode's rated reverse breakdown voltage must be greater than the maximum input voltage. Page 15 of 16 HIP6311A Small Outline Plastic Packages (SOIC) N INDEX AREA 0.25(0.010) M H M20.3 (JEDEC MS-013-AC ISSUE C) B M 20 LEAD WIDE BODY SMALL OUTLINE PLASTIC PACKAGE E INCHES -B- 1 2 SYMBOL 3 L SEATING PLANE -A- h x 45o A D -C- e C 0.10(0.004) C A M B S 1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of Publication Number 95. MILLIMETERS MIN MAX NOTES A 0.0926 0.1043 2.35 2.65 - 0.0040 0.0118 0.10 0.30 - B 0.013 0.0200 0.33 0.51 9 C 0.0091 0.0125 0.23 0.32 - D 0.4961 0.5118 12.60 13.00 3 E 0.2914 0.2992 7.40 7.60 4 0.050 BSC 1.27 BSC - H 0.394 0.419 10.00 10.65 - h 0.010 0.029 0.25 0.75 5 L 0.016 0.050 0.40 1.27 6 N NOTES: MAX A1 e A1 B 0.25(0.010) M µ MIN 20 0o 20 8o 0o 7 8o Rev. 0 12/93 2. Dimensioning and tolerancing per ANSI Y14.5M-1982. 3. Dimension “D” does not include mold flash, protrusions or gate burrs. Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side. 4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per side. 5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area. 6. “L” is the length of terminal for soldering to a substrate. 7. “N” is the number of terminal positions. 8. Terminal numbers are shown for reference only. 9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater above the seating plane, shall not exceed a maximum value of 0.61mm (0.024 inch) 10. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact. © Copyright Intersil Americas LLC 2001-2004. All Rights Reserved. All trademarks and registered trademarks are the property of their respective owners. For additional products, see www.intersil.com/en/products.html Intersil products are manufactured, assembled and tested utilizing ISO9001 quality systems as noted in the quality certifications found at www.intersil.com/en/support/qualandreliability.html Intersil products are sold by description only. Intersil may modify the circuit design and/or specifications of products at any time without notice, provided that such modification does not, in Intersil's sole judgment, affect the form, fit or function of the product. Accordingly, the reader is cautioned to verify that datasheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com FN9035 Rev 1.00 July 2004 Page 16 of 16