AD AD844JR-16-REEL7 60 mhz, 2000 v/us monolithic op amp Datasheet

a
FEATURES
Wide Bandwidth: 60 MHz at Gain of –1
Wide Bandwidth: 33 MHz at Gain of –10
Very High Output Slew Rate: Up to 2000 V/s
20 MHz Full Power Bandwidth, 20 V p-p, RL = 500 Fast Settling: 100 ns to 0.1% (10 V Step)
Differential Gain Error: 0.03% at 4.4 MHz
Differential Phase Error: 0.158 at 4.4 MHz
High Output Drive: 650 mA into 50 Load
Low Offset Voltage: 150 mV Max (B Grade)
Low Quiescent Current: 6.5 mA
Available in Tape and Reel in Accordance with
EIA-481A Standard
60 MHz, 2000 V/s
Monolithic Op Amp
AD844
CONNECTION DIAGRAMS
8-Lead Plastic (N),
and Cerdip (Q) Packages
NULL 1
–IN 2
+IN 3
–VS 4
AD844
8
7
6
16-Lead SOIC
(R) Package
NC 1
NULL
+VS
OFFSETNULL 2
OUTPUT
TOP VIEW
5 TZ
(Not to Scale)
PRODUCT DESCRIPTION
The AD844 is a high-speed monolithic operational amplifier
fabricated using Analog Devices’ junction isolated complementary bipolar (CB) process. It combines high bandwidth and very
fast large signal response with excellent dc performance. Although
optimized for use in current to voltage applications and as an
inverting mode amplifier, it is also suitable for use in many
noninverting applications.
The AD844 can be used in place of traditional op amps, but its
current feedback architecture results in much better ac performance, high linearity and an exceptionally clean pulse response.
This type of op amp provides a closed-loop bandwidth which is
determined primarily by the feedback resistor and is almost independent of the closed-loop gain. The AD844 is free from the slew
rate limitations inherent in traditional op amps and other
current-feedback op amps. Peak output rate of change can be over
2000 V/µs for a full 20 V output step. Settling time is typically
100 ns to 0.1%, and essentially independent of gain. The AD844
can drive 50 Ω loads to ± 2.5 V with low distortion and is short
circuit protected to 80 mA.
The AD844 is available in four performance grades and three
package options. In the 16-lead SOIC (R) package, the AD844J is
specified for the commercial temperature range of 0°C to 70°C.
The AD844A and AD844B are specified for the industrial temperature range of –40°C to +85°C and are available in the cerdip (Q)
16 NC
15 OFFSETNULL
–IN 3
14 V+
NC 4
13 NC
+IN 5
12 OUTPUT
NC 6
11 TZ
V– 7
10 NC
TOP VIEW
9 NC
(Not to Scale)
NC 8
APPLICATIONS
Flash ADC Input Amplifiers
High-Speed Current DAC Interfaces
Video Buffers and Cable Drivers
Pulse Amplifiers
AD844
NC = NO CONNECT
package. The AD844A is also available in an 8-lead plastic
mini-DIP (N). The AD844S is specified over the military temperature range of –55°C to +125°C. It is available in the 8-lead
cerdip (Q) package. “A” and “S” grade chips and devices processed
to MIL-STD-883B, REV. C are also available.
PRODUCT HIGHLIGHTS
1. The AD844 is a versatile, low cost component providing an
excellent combination of ac and dc performance.
2. It is essentially free from slew rate limitations. Rise and fall
times are essentially independent of output level.
3. The AD844 can be operated from ± 4.5 V to ± 18 V power
supplies and is capable of driving loads down to 50 Ω, as well
as driving very large capacitive loads using an external network.
4. The offset voltage and input bias currents of the AD844 are
laser trimmed to minimize dc errors; VOS drift is typically
1 µV/°C and bias current drift is typically 9 nA/°C.
5. The AD844 exhibits excellent differential gain and differential phase characteristics, making it suitable for a variety of
video applications with bandwidths up to 60 MHz.
6. The AD844 combines low distortion, low noise and low drift
with wide bandwidth, making it outstanding as an input
amplifier for flash A/D converters.
REV. D
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 2001
AD844–SPECIFICATIONS (@ T = 25C and V = 15 V dc, unless otherwise noted)
A
Model
INPUT OFFSET VOLTAGE1
TMIN–TMAX
vs. Temperature
vs. Supply
Initial
TMIN–TMAX
vs. Common Mode
Initial
TMIN–TMAX
INPUT BIAS CURRENT
–Input Bias Current1
TMIN–TMAX
vs. Temperature
vs. Supply
Initial
TMIN–TMAX
vs. Common Mode
Initial
TMIN–TMAX
+Input Bias Current1
TMIN–TMAX
vs. Temperature
vs. Supply
Initial
TMIN–TMAX
vs. Common Mode
Initial
TMIN–TMAX
Conditions
S
Min
AD844J/A
Typ Max
Min
AD844B
Typ
Max
AD844S
Min Typ Max
Unit
50
75
1
300
500
50
75
1
150
200
5
50
125
1
300
500
5
µV
µV
µV/°C
4
4
20
4
4
10
10
4
4
20
20
µV/V
µV/V
10
10
35
10
10
20
20
10
10
35
35
µV/V
µV/V
200
800
9
450
1500
150
750
9
250
1100
15
200
1900
20
450
2500
30
nA
nA
nA/°C
175
220
250
175
220
200
240
175
220
250
300
nA/V
nA/V
90
110
150
350
3
160
90
110
100
300
3
110
150
200
500
7
90
120
100
800
7
160
200
400
1300
15
nA/V
nA/V
nA
nA
nA/°C
5 V–18 V
VCM = ± 10 V
5 V–18 V
VCM = ± 10 V
400
700
5 V–18 V
80
100
150
80
100
100
120
80
120
150
200
nA/V
nA/V
90
130
150
90
130
120
190
90
140
150
200
nA/V
nA/V
50
10
65
50
10
65
50
10
65
Ω
MΩ
VCM = ± 10 V
INPUT CHARACTERISTICS
Input Resistance
–Input
+Input
Input Capacitance
–Input
+Input
Input Voltage Range
Common Mode
7
7
2
2
± 10
7
2
2
± 10
2
2
± 10
pF
pF
V
INPUT VOLTAGE NOISE
f ≥ 1 kHz
2
2
2
nV/√Hz
INPUT CURRENT NOISE
–Input
+Input
f ≥ 1 kHz
f ≥ 1 kHz
10
12
10
12
10
12
pA/√Hz
pA/√Hz
3.0
1.6
4.5
MΩ
MΩ
pF
OPEN LOOP TRANSRESISTANCE
VOUT = ± 10 V
RLOAD = 500 Ω
TMIN–TMAX
Transcapacitance
2.2
1.3
3.0
2.0
4.5
2.8
1.6
3.0
2.0
4.5
2.2
1.3
DIFFERENTIAL GAIN ERROR2
f = 4.4 MHz
0.03
0.03
0.03
%
DIFFERENTIAL PHASE ERROR2
f = 4.4 MHz
0.15
0.15
0.15
Degree
60
33
60
33
60
33
MHz
MHz
0.005
0.005
0.005
%
100
100
100
100
100
100
ns
ns
110
100
110
100
110
100
ns
ns
FREQUENCY RESPONSE
Small Signal Bandwidth
Gain = –13
Gain = –104
TOTAL HARMOMIC DISTORTION
SETTLING TIME
10 V Output Step
Gain = –1, to 0.1%5
Gain = –10, to 0.1%6
2 V Output Step
Gain = –1, to 0.1%5
Gain = –10, to 0.1%6
f = 100 kHz,
2 V rms5
± 15 V Supplies
± 5 V Supplies
–2–
REV. D
AD844
AD844J/A
Typ Max
Model
Conditions
Min
OUTPUT SLEW RATE
Overdriven
Input
1200
FULL POWER BANDWIDTH
VOUT = 20 V p-p5
VOUT = 2 V p-p5
OUTPUT CHARACTERISTICS
Voltage
Short Circuit Current
TMIN–TMAX
Output Resistance
VS = ± 15 V
VS = ± 5 V
THD = 3%
RLOAD = 500 Ω
AD844B
Typ
1200
2000
20
20
10
Open Loop
POWER SUPPLY
Operating Range
Quiescent Current
TMIN–TMAX
2000
Min
Max
AD844S
Min Typ Max
Unit
1200 2000
V/µs
20
20
11
80
60
15
10
± 4.5
± 18
7.5
8.5
6.5
7.5
11
80
60
15
± 4.5
6.5
7.5
10
± 18
7.5
8.5
20
20
MHz
MHz
11
80
60
15
±V
mA
mA
Ω
+4.5
6.5
8.5
± 18
7.5
9.5
V
mA
mA
NOTES
1
Rated performance after a 5 minute warmup at T A = 25°C.
2
Input signal 285 mV p-p carrier (40 IRE) riding on 0 mV to 642 mV (90 IRE) ramp. R L= 100 Ω; R1, R2 = 300 Ω.
3
Input signal 0 dBm, C L = 10 pF, RL = 500 Ω, R1 = 500 Ω, R2 = 500 Ω in Figure 2.
4
Input signal 0 dBm, C L =10 pF, R L = 500 Ω, R1 = 500 Ω, R2 = 50 Ω in Figure 2.
5
CL = 10 pF, RL = 500 Ω, R1 = 1 kΩ, R2 = 1 kΩ in Figure 2.
6
CL = 10 pF, RL = 500 Ω, R1 = 500 Ω, R2 = 50 Ω in Figure 2.
Specifications subject to change without notice. All min and max specifications are guaranteed.
ABSOLUTE MAXIMUM RATINGS 1
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V
Power Dissipation2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.1 W
Output Short Circuit Duration . . . . . . . . . . . . . . . . . Indefinite
Common-Mode Input Voltage . . . . . . . . . . . . . . . . . . . . . ± VS
Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . 6 V
Inverting Input Current
Continuous . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 mA
Transient . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 mA
Storage Temperature Range (Q) . . . . . . . . . –65°C to +150°C
Storage Temperature Range (N, R) . . . . . . . –65°C to +125°C
Lead Temperature Range (Soldering 60 sec) . . . . . . . . . 300°C
ESD Rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1000 V
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device at these or any other conditions above those
indicated in the operational sections of this specification is not implied. Exposure
to absolute maximum rating conditions for extended periods may affect device
reliability.
2
8-Lead Plastic Package:
θJA = 90°C/W
8-Lead Cerdip Package:
θJA = 110°C/W
16-Lead SOIC Package:
θJA = 100°C/W
METALIZATION PHOTOGRAPH
Contact factory for latest dimensions.
Dimension shown in inches and (mm).
ORDERING GUIDE
Model
Temperature
Range
Package
Option*
AD844AN
AD844ACHIPS
AD844AQ
AD844BQ
AD844JR-16
AD844JR-16-REEL
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
0°C to 70°C
0°C to 70°C
AD844JR-16-REEL7
0°C to 70°C
AD844SCHIPS
AD844SQ
AD844SQ/883B
5962-8964401PA
–55°C to +125°C
–55°C to +125°C
–55°C to +125°C
–55°C to +125°C
N-8
Die
Q-8
Q-8
R-16
13" Tape
and Reel
7" Tape
and Reel
Die
Q-8
Q-8
Q-8
*N
= Plastic DIP; Q = Cerdip; R = Small Outline IC (SOIC).
REV. D
–3–
AD844–Typical Characteristics (T = 25C and V = 15 V, unless otherwise noted)
A
–60
60
50
40
5
RL =
–70
TRANSRESISTANCE – M
HARMONIC DISTORTION – dB
70
–3dB BANDWIDTH – MHz
S
–80
1V rms
–90
–100
–110
2ND HARMONIC
4
RL = 500
3
2
RL = 50
1
–120
30
10
15
5
SUPPLY VOLTAGE – V
0
20
–130
100
3RD HARMONIC
10k
1k
INPUT FREQUENCY – Hz
20
20
MAGNITUDE OF THE OUTPUT VOLTAGE – V
TPC 2. Harmonic Distortion vs.
Frequency, R1 = R2 = 1 kΩ
10
5
0
0
10
15
5
SUPPLY VOLTAGE – V
10
5
0
0
10
15
5
SUPPLY VOLTAGE – V
7
VS = 15V
6
VS = 5V
4
–60 –40 –20
20
0 20 40 60 80 100 120 140
TEMPERATURE – C
TPC 6. Quiescent Supply
Current vs. Temperature and
Supply Voltage
TPC 5. Output Voltage Swing
vs. Supply Voltage
40
100
VS = 15V
35
OUTPUT IMPEDANCE – INPUT BIAS CURRENT – A
8
5
TPC 4. Noninverting Input Voltage
Swing vs. Supply Voltage
1
IBP
0
–1
–2
–50
IBN
0
50
100
TEMPERATURE – C
150
9
15
20
2
50
100
TEMPERATURE – C
10
RL = 500
TA = 25C
SUPPLY CURRENT – mA
15
0
TPC 3. Transresistance vs.
Temperature
–3dB BANDWIDTH – MHz
MAGNITUDE OF THE OUTPUT VOLTAGE – V
TPC 1. –3 dB Bandwidth vs.
Supply Voltage R1 = R2 = 500 Ω
TA = 25C
0
–50
100k
10
5V SUPPLIES
1
0.1
30
VS = 5V
25
20
15
150
TPC 7. Inverting Input Bias Current (IBN) and Noninverting Input
Bias Current (IBP) vs. Temperature
0.01
10k
100k
1M
10M
100M
FREQUENCY – Hz
TPC 8. Output Impedance vs.
Frequency, Gain = –1, R1 = R2 = 1 kΩ
–4–
10
–60 –40 –20
0
20 40
60 80 100 120 140
TEMPERATURE – C
TPC 9. –3 dB Bandwidth vs.
Temperature, Gain = –1,
R1 = R2 = 1 kΩ
REV. 0
AD844
Inverting Gain-of-1 AC Characteristics
+VS
R1 = R2 = 500
–210
0
R1
PHASE – Degrees
0.22F
–180
6
4.7
VIN
–
AD844
VOUT
+
RL
CL
GAIN – dB
R1 = R2 = 1k
R2
–6
–12
R1 = R2 = 500
–240
–270
R1 = R2 = 1k
–300
–18
0.22F
4.7
–24
100k
–VS
TPC 10. Inverting Amplifier,
Gain of –1 (R1 = R2)
–330
1M
10M
FREQUENCY – Hz
100M
0
TPC 11. Gain vs. Frequency for
Gain = –1, RL = 500 Ω, CL = 0 pF
25
FREQUENCY – MHz
50
TPC 12. Phase vs. Frequency
Gain = –1, RL = 500 Ω, CL = 0 pF
TPC 14. Small Signal Pulse
Response, Gain = –1, R1 = R2 = 1 kΩ
TPC 13. Large Signal Pulse
Response, Gain = –1, R1 = R2 = 1 kΩ
Inverting Gain-of-10 AC Characteristics
26
–180
+VS
RL = 500
4.7
20
–210
RL = 50
VIN
GAIN – dB
500
50
–
AD844
PHASE – Degrees
0.22F
14
8
VOUT
–240
RL = 500
–270
RL = 50
+
2
RL
–300
CL
–4
100k
0.22F
4.7
1M
10M
FREQUENCY – Hz
100M
–330
0
25
FREQUENCY – MHz
–VS
TPC 15. Gain of –10 Amplifier
REV. D
TPC 16. Gain vs. Frequency,
Gain = –10
–5–
TPC 17. Phase vs. Frequency,
Gain = –10
50
AD844
Inverting Gain-of-10 Pulse Response
TPC 19. Small Signal Pulse
Response, Gain = –10, RL = 500 Ω
TPC 18. Large Signal Pulse
Response, Gain = –10, RL = 500 Ω
Noninverting Gain-of-10 AC Characteristics
26
–180
RL = 500
20
0.22F
450
–
50
AD844
VIN
+
0.22F
VOUT
RL
CL
–210
14
PHASE – Degrees
4.7
GAIN – dB
+VS
RL = 50
8
2
–240
RL = 500
RL = 50k
–270
–300
4.7
–4
100k
–VS
TPC 20. Noninverting Gain of
+10 Amplifier
1M
10M
FREQUENCY – Hz
100M
TPC 21. Gain vs. Frequency,
Gain = +10
TPC 23. Noninverting Amplifier Large
Signal Pulse Response, Gain = +10,
RL = 500 Ω
–330
0
25
FREQUENCY – MHz
50
TPC 22. Phase vs. Frequency,
Gain = +10
TPC 24. Small Signal Pulse
Response, Gain = +10, RL = 500 Ω
–6–
REV. D
AD844
UNDERSTANDING THE AD844
Response as an Inverting Amplifier
The AD844 can be used in ways similar to a conventional op
amp while providing performance advantages in wideband
applications. However, there are important differences in the
internal structure which need to be understood in order to
optimize the performance of the AD844 op amp.
Figure 2 shows the connections for an inverting amplifier.
Unlike a conventional amplifier the transient response and the
small signal bandwidth are determined primarily by the value of
the external feedback resistor, R1, rather than by the ratio of
R1/R2 as is customarily the case in an op amp application. This
is a direct result of the low impedance at the inverting input. As
with conventional op amps, the closed loop gain is –R1/R2.
Open Loop Behavior
Figure 1 shows a current feedback amplifier reduced to essentials. Sources of fixed dc errors such as the inverting node bias
current and the offset voltage are excluded from this model and
are discussed later. The most important parameter limiting the
dc gain is the transresistance, Rt, which is ideally infinite. A finite
value of Rt is analogous to the finite open loop voltage gain in a
conventional op amp.
The current applied to the inverting input node is replicated by
the current conveyor so as to flow in resistor Rt. The voltage
developed across Rt is buffered by the unity gain voltage follower.
Voltage gain is the ratio Rt / RIN. With typical values of Rt = 3 MΩ
and RIN = 50 Ω, the voltage gain is about 60,000. The open
loop current gain is another measure of gain and is determined
by the beta product of the transistors in the voltage follower
stage (see Figure 4); it is typically 40,000.
+1
IIN
RIN
Rt
Ct
+1
IIN
Figure 1. Equivalent Schematic
The important parameters defining ac behavior are the transcapacitance, Ct, and the external feedback resistor (not shown).
The time constant formed by these components is analogous to
the dominant pole of a conventional op amp, and thus cannot
be reduced below a critical value if the closed loop system is to
be stable. In practice, Ct is held to as low a value as possible
(typically 4.5 pF) so that the feedback resistor can be maximized
while maintaining a fast response. The finite RIN also affects the
closed loop response in some applications as will be shown.
The closed loop transresistance is simply the parallel sum of R1
and Rt. Since R1 will generally be in the range 500 Ω to 2 kΩ
and Rt is about 3 MΩ the closed loop transresistance will be
only 0.02% to 0.07% lower than R1. This small error will often
be less than the resistor tolerance.
When R1 is fairly large (above 5 kΩ) but still much less than Rt,
the closed loop HF response is dominated by the time constant
R1Ct. Under such conditions the AD844 is over-damped and
will provide only a fraction of its bandwidth potential. Because
of the absence of slew rate limitations under these conditions,
the circuit will exhibit a simple single pole response even under
large signal conditions.
In Figure 2, R3 is used to properly terminate the input if desired.
R3 in parallel with R2 gives the terminated resistance. As R1 is
lowered, the signal bandwidth increases, but the time constant
R1Ct becomes comparable to higher order poles in the closed
loop response. Therefore, the closed loop response becomes
complex, and the pulse response shows overshoot. When R2 is
much larger than the input resistance, RIN, at Pin 2, most of the
feedback current in R1 is delivered to this input; but as R2
becomes comparable to RIN, less of the feedback is absorbed at
Pin 2, resulting in a more heavily damped response. Consequently, for low values of R2 it is possible to lower R1 without
causing instability in the closed loop response. Table I lists
combinations of R1 and R2 and the resulting frequency response
for the circuit of Figure 2. TPC 13 shows the very clean and fast
± 10 V pulse response of the AD844.
R1
VIN
R2
R3
OPTIONAL
VOUT
RL
The open loop ac gain is also best understood in terms of the
transimpedance rather than as an open loop voltage gain. The
open loop pole is formed by Rt in parallel with Ct. Since Ct is
typically 4.5 pF, the open loop corner frequency occurs at
about 12 kHz. However, this parameter is of little value in
determining the closed loop response.
REV. D
AD844
Figure 2. Inverting Amplifier
–7–
CL
AD844
Table I.
R1
ISIG
Gain
R1
R2
BW (MHz)
GBW (MHz)
–1
–1
–2
–2
–5
–5
–10
–10
–20
–100
+100
1 kΩ
500 Ω
2 kΩ
1 kΩ
5 kΩ
500 Ω
1 kΩ
500 Ω
1 kΩ
5 kΩ
5 kΩ
1 kΩ
500 Ω
1 kΩ
500 Ω
1 kΩ
100 Ω
100 Ω
50 Ω
50 Ω
50 Ω
50 Ω
35
60
15
30
5.2
49
23
33
21
3.2
9
35
60
30
60
26
245
230
330
420
320
900
CL
Figure 3. Current-to-Voltage Converter
Circuit Description of the AD844
A simplified schematic is shown in Figure 4. The AD844 differs
from a conventional op amp in that the signal inputs have
radically different impedance. The noninverting input (Pin 3)
presents the usual high impedance. The voltage on this input is
transferred to the inverting input (Pin 2) with a low offset
voltage, ensured by the close matching of like polarity transistors operating under essentially identical bias conditions. Laser
trimming nulls the residual offset voltage, down to a few
tens of microvolts. The inverting input is the common emitter
node of a complementary pair of grounded base stages and
behaves as a current summing node. In an ideal current feedback op amp the input resistance would be zero. In the AD844
it is about 50 Ω.
The AD844 works well as the active element in an operational
current to voltage converter, used in conjunction with an external scaling resistor, R1, in Figure 3. This analysis includes the
stray capacitance, CS, of the current source, which might be a
high speed DAC. Using a conventional op amp, this capacitance
forms a “nuisance pole” with R1 which destabilizes the closed
loop response of the system. Most op amps are internally compensated for the fastest response at unity gain, so the pole due
to R1 and CS reduces the already narrow phase margin of the
system. For example, if R1 were 2.5 kΩ a CS of 15 pF would
place this pole at a frequency of about 4 MHz, well within the
response range of even a medium speed operational amplifier.
In a current feedback amp this nuisance pole is no longer determined by R1 but by the input resistance, RIN. Since this is about
50 Ω for the AD844, the same 15 pF forms a pole 212 MHz
and causes little trouble. It can be shown that theresponse of
this system is:
A current applied to the inverting input is transferred to a
complementary pair of unity-gain current mirrors which deliver
the same current to an internal node (Pin 5) at which the full
output voltage is generated. The unity-gain complementary
voltage follower then buffers this voltage and provides the load
driving power. This buffer is designed to drive low impedance
loads such as terminated cables, and can deliver ± 50 mA into a
50 Ω load while maintaining low distortion, even when operating at supply voltages of only ± 6 V. Current limiting (not
shown) ensures safe operation under short circuited conditions.
K R1
(1 + sTd )(1 + sTn )
7 +VS
where K is a factor very close to unity and represents the finite
dc gain of the amplifier, Td is the dominant pole and Tn is the
nuisance pole:
K =
VOUT
RL
Response as an I-V Converter
VOUT = – Isig
AD844
CS
IB
Rt
Rt + R1
+IN 3
2 –IN
TZ
5
6 OUT
Td = KR1Ct
Tn = RINCS (assuming RIN << R1)
IB
Using typical values of R1 = 1 kΩ and Rt = 3 MΩ, K is 0.9997;
in other words, the “gain error” is only 0.03%. This is much less
than the scaling error of virtually all DACs and can be absorbed,
if necessary, by the trim needed in a precise system.
4 –VS
Figure 4. Simplified Schematic
In the AD844, Rt is fairly stable with temperature and supply
voltages, and consequently the effect of finite “gain” is negligible unless high value feedback resistors are used. Since that
would result in slower response times than are possible, the
relatively low value of Rt in the AD844 will rarely be a significant source of error.
–8–
REV. D
AD844
It is important to understand that the low input impedance at
the inverting input is locally generated, and does not depend on
feedback. This is very different from the “virtual ground” of a
conventional operational amplifier used in the current summing
mode which is essentially an open circuit until the loop settles.
In the AD844, transient current at the input does not cause
voltage spikes at the summing node while the amplifier is settling.
Furthermore, all of the transient current is delivered to the
slewing (TZ) node (Pin 5) via a short signal path (the grounded
base stages and the wideband current mirrors).
+VS
4.7
OFFSET
TRIM
CPK 3nF
VIN
The offset voltage of the AD844 is laser trimmed to the 50 µV
level and exhibits very low drift. In practice, there is an additional offset term due to the bias current at the inverting input
(IBN) which flows in the feedback resistor (R1). This can optionally be nulled by the trimming potentiometer shown in Figure 5.
REV. D
RL
–VS
Figure 5. Noninverting Amplifier Gain = 100, Optional
Offset Trim Is Shown
46
VS = 15V
40
GAIN – dB
Since current feedback amplifiers are asymmetrical with regard
to their two inputs, performance will differ markedly in noninverting and inverting modes. In noninverting modes, the large
signal high speed behavior of the AD844 deteriorates at low
gains because the biasing circuitry for the input system (not
shown in Figure 4) is not designed to provide high input voltage
slew rates.
The AD844 provides very clean pulse response at high noninverting gains. Figure 5 shows a typical configuration providing a
gain of 100 with high input resistance. The feedback resistor is
kept as low as practicable to maximize bandwidth, and a peaking capacitor (CPK) can optionally be added to further extend
the bandwidth. Figure 6 shows the small signal response with
CPK = 3 nF, RL = 500 Ω, and supply voltages of either ± 5 V or
± 15 V. Gain bandwidth products of up to 900 MHz can be achieved
in this way.
+
4.7
Response as a Noninverting Amplifier
Noninverting Gain of 100
AD844
0.22F
This inherent advantage would be lost if the voltage follower
used to buffer the output were to have slew rate limitations. The
AD844 has been designed to avoid this problem, and as a result
the output buffer exhibits a clean large signal transient response,
free from anomalous effects arising from internal saturation.
However, good results can be obtained with some care. The
noninverting input will not tolerate a large transient input; it
must be kept below ±1 V for best results. Consequently this mode
is better suited to high gain applications (greater than ×10).
TPC 20 shows a noninverting amplifier with a gain of 10 and a
bandwidth of 30 MHz. The transient response is shown in
TPCs 23 and 24. To increase the bandwidth at higher gains, a
capacitor can be added across R2 whose value is approximately
the ratio of R1 and R2 times Ct.
0.22F
8
–
R2
4.99
The current available to charge the capacitance (about 4.5 pF)
at TZ node, is always proportional to the input error current, and
the slew rate limitations associated with the large signal response
of op amps do not occur. For this reason, the rise and fall times
are almost independent of signal level. In practice, the input
current will eventually cause the mirrors to saturate. When using
± 15 V supplies, this occurs at about 10 mA (or ± 2200 V/µs).
Since signal currents are rarely this large, classical “slew rate”
limitations are absent.
R1
499
20
34
VS = 5V
28
22
16
100k
1M
10M
20M
FREQUENCY – Hz
Figure 6. AC Response for Gain = 100, Configuration
Shown in Figure 5
USING THE AD844
Board Layout
As with all high frequency circuits considerable care must be
used in the layout of the components surrounding the AD844.
A ground plane, to which the power supply decoupling capacitors are connected by the shortest possible leads, is essential
to achieving clean pulse response. Even a continuous ground
plane will exhibit finite voltage drops between points on the
plane, and this must be kept in mind in selecting the grounding
points. Generally speaking, decoupling capacitors should be
taken to a point close to the load (or output connector) since
the load currents flow in these capacitors at high frequencies.
The +IN and –IN circuits (for example, a termination resistor
and Pin 3) must be taken to a common point on the ground
plane close to the amplifier package.
Use low impedance capacitors (AVX SR305C224KAA or
equivalent) of 0.22 µF wherever ac coupling is required. Include
either ferrite beads and/or a small series resistance (approximately 4.7 Ω) in each supply line.
–9–
AD844
Input Impedance
At low frequencies, negative feedback keeps the resistance at the
inverting input close to zero. As the frequency increases, the
impedance looking into this input will increase from near zero to
the open loop input resistance, due to bandwidth limitations,
making the input seem inductive. If it is desired to keep the
input impedance flatter, a series RC network can be inserted
across the input. The resistor is chosen so that the parallel sum
of it and R2 equals the desired termination resistance. The
capacitance is set so that the pole determined by this RC network
is about half the bandwidth of the op amp. This network is not
important if the input resistor is much larger than the termination used, or if frequencies are relatively low. In some cases, the
small peaking that occurs without the network can be of use in
extending the –3 dB bandwidth.
Schottky diodes, to create the error signal and limit the input
signal to the oscilloscope. For measuring settling time, the ratio
of R6/R5 is equal to R1/R2. For unity gain, R6 = R5 = 1 kΩ,
and RL = 500 Ω. For the gain of –10, R5 = 50 Ω, R6 = 500 Ω
and R L was not used since the summing network loads the
output with approximately 275 Ω. Using this network in a
unity-gain configuration, settling time is 100 ns to 0.1% for a
–5 V to +5 V step with CL = 10 pF.
TO SCOPE
(TEK 7A11 FET PROBE)
R5
R6
D1
D2
R1
Driving Large Capacitive Loads
Capacitive drive capability is 100 pF without an external network. With the addition of the network shown in Figure 7, the
capacitive drive can be extended to over 10,000 pF, limited by
internal power dissipation. With capacitive loads, the output
speed becomes a function of the overdriven output current
limit. Since this is roughly ± 100 mA, under these conditions,
the maximum slew rate into a 1000 pF load is ± 100 V/µs.
Figure 8 shows the transient response of an inverting amplifier
(R1 = R2 = 1 kΩ) using the feed forward network shown in
Figure 7, driving a load of 1000 pF.
AD844
VOUT
CL
750
VIN
R2
VOUT
RL
CL
D1, D2 IN6263 OR EQUIV. SCHOTTKY DIODE
Figure 9. Settling Time Test Fixture
DC Error Calculation
Figure 10 shows a model of the dc error and noise sources for
the AD844. The inverting input bias current, IBN, flows in the
feedback resistor. IBP, the noninverting input bias current, flows
in the resistance at Pin 3 (RP), and the resulting voltage (plus
any offset voltage) will appear at the inverting input. The total
error, VO, at the output is:

R1 
VO = (IBP RP + VOS + IBN RIN )1 +
 + I BN R1

R2 
22pF
Figure 7. Feed Forward Network for Large Capacitive
Loads
AD844
R3
Since IBN and IBP are unrelated both in sign and magnitude,
inserting a resistor in series with the noninverting input will not
necessarily reduce dc error and may actually increase it.
R1
R2
+
VN
RIN
~
INN
IBN
INP
IBP
RP
VOS
AD844
Figure 8. Driving 1000 pF CL with Feed Forward Network
of Figure 7
Figure 10. Offset Voltage and Noise Model for the AD844
Settling Time
Settling time is measured with the circuit of Figure 9. This
circuit employs a false summing node, clamped by the two
–10–
REV. D
Applications–AD844
Noise
0.3
Noise sources can be modeled in a manner similar to the dc bias
currents, but the noise sources are INN, INP, VN, and the amplifier
induced noise at the output, VON, is:
DIFFERENTIAL PHASE – Degree
IRE = 7.14mV
2
R1 
2
2 
2
VON = ((Inp RP ) + Vn )1 +
 + (Inn R1)
R2 

Overall noise can be reduced by keeping all resistor values to a
minimum. With typical numbers, R1 = R2 = 1 kΩ, R P = 0,
Vn = 2 nV/√Hz, Inp = 10 pA/√Hz, Inn = 12 pA/√Hz, V ON
calculates to 12 nV/√Hz. The current noise is dominant in
this case, as it will be in most low gain applications.
Video Cable Driver Using 5 Volt Supplies
2
50
6
–0.2
54
72
90
IRE = 7.14mV
0.04
VOUT
RL
50
2.2F
36
0.06
ZO = 50
4
18
Figure 11c. Differential Phase for the Circuit of Figure 11a
DIFFERENTIAL PHASE – Degree
7
–0.1
VOUT – IRE
2.2F
50
0
0
+5V
VIN
0.1
–0.3
The AD844 can be used to drive low impedance cables. Using
± 5 V supplies, a 100 Ω load can be driven to ± 2.5 V with low
distortion. Figure 11a shows an illustrative application which
provides a noninverting gain of 2, allowing the cable to be
reverse-terminated while delivering an overall gain of +1 to the
load. The –3 dB bandwidth of this circuit is typically 30 MHz.
Figure 11b shows a differential gain and phase test setup. In
video applications, differential-phase and differential-gain
characteristics are often important. Figure 11c shows the variation in phase as the load voltage varies. Figure 11d shows the
gain variation.
3
0.2
300
0.02
0
–0.02
–0.04
–0.06
0
–5V
18
36
54
72
90
VOUT – IRE
300
Figure 11d. Differential Gain for the Circuit of Figure 11a
Figure 11a. The AD844 as a Cable Driver
HP8753A
NETWORK
ANALYZER
RF OUT
R
EXT
TRIG
SYNC OUT
HP3314A
STAIRCASE
GENERATOR
IN
OUT
OUT
HP11850C
SPLITTER
50
(TERMINATOR)
VIN
VIN
CIRCUIT
UNDER
TEST
High Speed DAC Buffer
VOUT
OUT
470
OUT
Figure 11b. Differential Gain/Phase Test Setup Figure
REV. D
The AD844 performs very well in applications requiring
current-to-voltage conversion. Figure 12 shows connections for
use with the AD568 current output DAC. In this application the
bipolar offset is used so that the full-scale current is ± 5.12 mA,
which generates an output of ± 5.12 V usingdecoupling and
grounding techniques to achieve the full 12-bit accuracy and
realize the fast settling capabilities of the system. The unmarked
capacitors in this figure are 0.1 µF ceramic (for the 1 kΩ application resistor on the AD568. Figure 13 shows the full-scale
transient response. Care is needed in power supply example,
AVX Type SR305C104KAA), and the ferrite inductors should
be about 2.5 µH (for example, Fair-Rite Type 2743002122).
The AD568 data sheet should be consulted for more complete
details about its use.
–11–
AD844
1
MSB
REFCOM 23
3
–15V 22
4
IBPD 21
6
AD568 RL
*
*
*
ACOM 18
8
LCOM 17
9
SPAN 16
10
SPAN 15
11
THCOM 14
LSB
–15V
VOUT
AD844
19
7
12
*
IOUT 20
5
DIGITAL
INPUTS
+15V
+15V 24
2
VTH 13
RI
ANALOG
SUPPLY
GROUND
GROUND
100pF
–5V
*0.22F
POWER SUPPLY
BYPASS CAPACITORS
TOP VIEW
(Not to Scale)
DIGITAL
SUPPLY
Figure 12. High Speed DAC Amplifier
10
10
0.22F
INPUTS
VX
0 TO 3V
VY
2V FS
3nF
+VS
TYP+6V
@15A
0.22F
1
16
TOP VIEW
(Not to Scale)
AD844
OUTPUT
VW
AD539
I/P
GND
Figure 13. DAC Amplifier Full-Scale Transient Response
0.22F
8
9
0.22F
10
20 MHz Variable Gain Amplifier
The AD844 is an excellent choice as an output amplifier for the
AD539 multiplier, in all of its connection modes. (See AD539
data sheet for full details.) Figure 14 shows a simple multiplier
providing the output:
V V
VW = – X Y
2V
where VX is the “gain control” input, a positive voltage of from
0 V to 3.2 V (max) and VY is the “signal voltage,” nominally
± 2 V FS but capable of operation up to ± 4.2 V. The peak output in this configuration is thus ± 6.7 V. Using all four of the
internal application resistors provided on the AD539 in parallel
results in a feedback resistance of 1.5 kΩ, at which value the
bandwidth of the AD844 is about 22 MHz, and is essentially
independent of VX. The gain at VX = 3.16 V is +4 dB.
*VX AND VY INPUTS MAY OPTIONALLY
BE TERMINATED – TYPICALLY BY USING
A 50 OR 75 RESISTOR TO GROUND.
VW =
–VXVY
2V
10
–VS
TYP+6V
@15A
Figure 14. 20 MHz VGA Using the AD539
Figure 15 shows the small signal response for a 50 dB gain
control range (VX = 10 mV to 3.16 V). At small values of VX,
capacitive feedthrough on the PC board becomes troublesome,
and very careful layout techniques are needed to minimize this
problem. A ground strip between the pins of the AD539 will be
helpful in this regard. Figure 16 shows the response to a 2 V
pulse on VY for VX = 1 V, 2 V, and 3 V. For these results, a load
resistor of 500 Ω was used and the supplies were ± 9 V. The
multiplier will operate from supplies between ±4.5 V and ±16.5 V.
Disconnecting Pins 9 and 16 on the AD539 alters the denominator in the above expression to 1 V, and the bandwidth will be
approximately 10 MHz, with a maximum gain of 10 dB. Using
only Pin 9 or Pin 16 results in a denominator of 0.5 V, a bandwidth of 5 MHz and a maximum gain of 16 dB.
–12–
REV. D
AD844
+4
VX = 3.15V
–6
VX = 1.0V
GAIN – dB
–16
VX = 0.316V
–26
VX = 0.10V
–36
VX = 0.032V
–46
–56
100k
1M
10M
60M
FREQUENCY – Hz
Figure 15. VGA AC Response
REV. D
Figure 16. VGA Transient Response with VX = 1 V, 2 V, and 3 V
–13–
AD844
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
Mini-DIP (N) Package
(N-8)
0.430 (10.92)
0.348 (8.84)
8
5
PIN 1
0.280 (7.11)
0.240 (6.10)
4
1
0.325 (8.25)
0.300 (7.62)
0.100 (2.54)
BSC
0.060 (1.52)
0.015 (0.38)
0.210
(5.33)
MAX
0.195 (4.95)
0.115 (2.93)
0.130
(3.30)
MIN
0.160 (4.06)
0.115 (2.93)
0.015 (0.381)
0.008 (0.204)
0.022 (0.558) 0.070 (1.77) SEATING
0.014 (0.356) 0.045 (1.15) PLANE
Cerdip (Q) Package
(Q-8)
0.055 (1.4)
MAX
0.005 (0.13)
MIN
8
5
0.310 (7.87)
0.220 (5.59)
PIN 1
1
4
0.100 (2.54) BSC
0.320 (8.13)
0.290 (7.37)
0.405 (10.29) MAX
0.060 (1.52)
0.015 (0.38)
0.200 (5.08)
MAX
0.150
(3.81)
MIN
0.200 (5.08)
0.125 (3.18)
SEATING
0.023 (0.58) 0.070 (1.78) PLANE
0.014 (0.36) 0.030 (0.76)
0.015 (0.38)
0.008 (0.20)
15
0
16-Lead SOIC (R) Package
(R-16)
0.4133 (10.50)
0.3977 (10.00)
9
16
0.2992 (7.60)
0.2914 (7.40)
PIN 1
0.4193 (10.65)
0.3937 (10.00)
8
1
0.050 (1.27)
BSC
0.0118 (0.30)
0.0040 (0.10)
0.1043 (2.65)
0.0926 (2.35)
8
0.0192 (0.49) SEATING
0
0.0125 (0.32)
PLANE
0.0138 (0.35)
0.0091 (0.23)
–14–
0.0291 (0.74)
45
0.0098 (0.25)
0.0500 (1.27)
0.0157 (0.40)
REV. D
AD844
Revision History
Location
Page
Data Sheet changed from REV. B to REV. C.
Edits to SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2
Edits to ABSOLUTE MAXIMUM RATINGS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Edits to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
REV. D
–15–
–16–
PRINTED IN U.S.A.
C00897–0–11/01(D)